JP7063004B2 - Self-diagnosis device - Google Patents

Description

The present invention relates to a self-diagnosis device that self-diagnoses a signal processing circuit.

In recent years, many technologies such as collision prevention and automatic driving have been proposed, and a technology for measuring the distance from an own device to a target using radar technology is drawing attention. For example, the applicant has proposed a millimeter-wave band radar device for a mobile body as a device for measuring the distance from the own device to a target.

This device generates, for example, a signal in the high frequency region of the millimeter wave band and sends and receives it to and from the outside. When testing this type of device, it is conceivable to input an external signal source in the millimeter wave band to test the receiver, but it is desired to realize a self-diagnosis function that enables mass production testing at a lower cost. There is. The semiconductor integrated circuit is provided with a BIST (Built-In-Self Test) function for performing an internal test in order to reduce the cost required for the test at the time of shipment. This BIST function is disclosed in, for example, Patent Document 1.

According to the technique described in Patent Document 1, as a method of measuring a gain without using an external measuring instrument such as a spectrum analyzer, a received signal is received, mixed with a local signal from an oscillator, and an intermediate frequency signal is output. It is equipped with a frequency mixer and a phase control unit that extracts a part of the local signal, sets different phases, and outputs it as a received signal, and is a frequency mixer for different phases set by the phase control unit. The DC voltage in the intermediate frequency signal output from is detected, and the amount of change in the output DC offset of the frequency mixer is obtained to calculate the gain of the receiver.

Japanese Unexamined Patent Publication No. 2013-98693

In the technique described in Patent Document 1, since the frequencies of the high frequency signal input to the frequency mixer and the frequency of the local signal are set to be equal, only the gain measurement process using the DC offset by self-mixing can be supported. Therefore, it is not possible to realize a gain inspection other than the DC offset with high accuracy.

An object of the present disclosure is to provide a self-diagnosis apparatus capable of performing self-diagnosis with high accuracy even at frequencies other than DC.

According to the invention according to claim 1, a high frequency amplifier that amplifies a high frequency signal input to a high frequency input through an antenna, a PLL that generates a first local signal, and a high frequency signal are down-converted by the first local signal. It comprises a first frequency mixer and an intermediate frequency amplifier that amplifies an intermediate frequency signal output by the first frequency mixer, from a high frequency input to a high frequency amplifier, a first frequency mixer, an intermediate frequency. It is intended for self-diagnosis devices that perform self-diagnosis using signals obtained through an amplifier.

At this time, the same PLL generates a first local signal and also generates a second local signal for self-diagnosis and all self-diagnosis frequency signals that satisfy a frequency condition lower than the second local signal and exceeding DC. Therefore, these first and second local signals and self-diagnosis frequency signals can be generated with high frequency accuracy. The second frequency mixer up-converts the self-diagnosis frequency signal to the second local signal generated by the PLL to generate two tones and input them to the high frequency input.

The high frequency amplifier then amplifies the two tone output of the second frequency mixer, and the first frequency mixer downconverts the two tone output input to the high frequency input by the first local signal. Therefore, theoretically, the first frequency mixer outputs the frequency component of the self-diagnosis frequency signal as a mixed signal. At this time, since the same PLL generates all the first and second local signals and the self-diagnosis frequency signal, the first frequency mixer can be down-converted to one tone with high frequency accuracy.
The intermediate frequency amplifier amplifies the intermediate frequency signal output by the first frequency mixer. Therefore, the self-diagnosis device can perform self-diagnosis with high accuracy at a frequency exceeding DC by performing self-diagnosis using the signal obtained through this intermediate frequency amplifier.

Electrical configuration diagram illustrating the first embodiment Explanatory diagram of frequencies before and after down conversion (2 tones) Part 1 of the self-diagnosis processing flow chart Part 2 of the self-diagnosis processing flow chart Part 3 of the self-diagnosis processing flow chart Part 4 of the self-diagnosis processing flow chart Explanatory diagram of the phase relationship between two tones when the phase changes Output theory result after synthesizing the desired wave and image wave after down conversion when the phase is changed by the phase shifter Output evaluation result Evaluation result of evaluating the gain at the intermediate frequency Electrical configuration diagram of the second frequency mixer in the second embodiment (two-tone generation) Evaluation result of two-tone mixed signal Electrical configuration diagram for explaining the third embodiment Explanatory diagram of frequencies before and after down conversion (1 tone) Electrical configuration diagram of the second frequency mixer in the fourth embodiment (1 tone generation) Evaluation result of 1 tone signal Change in output DC offset voltage of first frequency mixer due to phase change An electrical configuration diagram illustrating a fifth embodiment An electrical configuration diagram illustrating a sixth embodiment An electrical configuration diagram illustrating a seventh embodiment

Hereinafter, some embodiments of the self-diagnosis device will be described with reference to the drawings. In each of the embodiments described below, configurations that perform the same or similar operations are designated by the same or similar reference numerals, and the description thereof will be omitted as necessary. In the following embodiments, the same or similar configurations are described by adding the same reference numerals to the tens place and the ones place of the reference numerals.

(First Embodiment)
1 to 7 show explanatory views of the first embodiment. As shown in FIG. 1, the receiver 1 includes a coupler 2, a high frequency amplifier 3, first and second frequency mixers 4 and 5, PLL 6, an intermediate frequency amplifier 7, and a phase shifter 8. The receiving antenna 9 and the controller 10 are provided in the chip of the semiconductor integrated circuit. When the receiver 1 has such a configuration, it has a function as a self-diagnosis device.

The controller 10 is composed of a control entity such as a logic circuit or a microcomputer, has various control functions for variable control of the gain of the intermediate frequency amplifier 7 and control of the output frequency of the PLL 6, and the phase shifter 8 transfers a signal. It has a control function as a phase shift adjustment unit that controls the phase φ.

Normally, the PLL 6 inputs a reference clock CLK generated by a reference oscillation circuit (not shown), and uses this reference clock CLK as a first local signal having a frequency of fLO (for example, several tens of GHz, 40 GHz, 80 GHz). Generate LO1. The high frequency amplifier 3 is composed of, for example, a low noise amplifier, and amplifies a high frequency signal in the millimeter wave band input to the high frequency input RxIN through the receiving antenna 9. The gain of the high frequency amplifier 3 is set to, for example, about several dB.
The first frequency mixer 4 downconverts the high frequency signal amplified by the high frequency amplifier 3 by the first local signal LO1. The intermediate frequency amplifier 7 is configured so that the amplification degree can be changed by the controller 10, and the signal down-converted by the first frequency mixer 4 is amplified in the range of, for example, several dB to several tens dB, and the intermediate frequency is used. Output as signal IFOUT.

On the other hand, the receiver 1 has a self-diagnosis function. In order to realize this self-diagnosis function, the PLL 6 has a function of generating a second local signal LO2 of the frequency fLO2 together with the first local signal LO1 of the frequency fLO, and further, the second local signal LO2. It has a function to generate a self-diagnosis clock signal (corresponding to a self-diagnosis frequency signal) BIST_CLK having a frequency fBIST_CLK (for example, 20 MHz) that satisfies a frequency condition lower than the frequency fLO2 and exceeding DC. This PLL 6 uses the reference clock CLK of the reference oscillation circuit (not shown) to adjust parameters such as the multiplication factor of the reference clock CLK, thereby adjusting these first and second local signals LO1 and LO2, as well as self. Generates all diagnostic clock signals BIST_CLK.

Since the same PLL6 generates all of these signals, these first and second local signals LO1 and LO2 and the self-diagnosis clock signal BIST_CLK have frequency characteristics against frequency fluctuations and external environment fluctuations of the reference clock CLK. Highly correlated with change.

When the receiver 1 self-diagnoses this receiving system circuit, the PLL 6 outputs the second local signal LO2 of the frequency fLO2 (= fLO) to the second frequency mixer 5, and the second local signal. The self-diagnosis clock signal BIST_CLK of the frequency fBIST_CLK that satisfies the frequency condition lower than LO2 is output. Then, the second frequency mixer 5 up-converts the self-diagnosis clock signal BIST_CLK to the second local signal LO2 generated by the PLL 6, so that the two tones of the frequency fLO2 ± fBIST_CLK are within the predetermined frequency bandwidth. Can be generated.

FIG. 1 illustrates the main frequency components output to the main part when the second frequency mixer 5 generates and outputs a two-tone signal. Further, on the left side of FIG. 2, the output PBIST of the mixed signal of the two-tone frequency fLO2 ± fBIST_CLK by the second frequency mixer 5 is shown.

The phase shifter 8 is provided, for example, on the path from the second frequency mixer 5 to the high frequency input RxIN, and when a two-tone signal is input from the second frequency mixer 5, the reference phase (for example, 0 °) is obtained. Two tones are phase-shifted to each of the two tones, and this signal is output to the high-frequency input RxIN.

The coupler 2 is configured as a high frequency input RxIN, and a two-tone signal having a frequency of fLO ± fBIST_CLK output from the phase shifter 8 is coupled by the high frequency input RxIN and input to the high frequency amplifier 3. The high frequency amplifier 3 amplifies the two-tone signal and outputs it to the first frequency mixer 4.

The first frequency mixer 4 mixes this amplified two-tone signal with the first local signal LO1 of frequency fLO. Then, the intermediate frequency amplifier 7 amplifies the mixed signal mixed by the first frequency mixer 4 and outputs it as an intermediate frequency signal IFOUT. In this case, the first frequency mixer 4 mixes the signal of frequency fLO2 ± fBIST_CLK with the first local signal LO1 of frequency fLO and outputs it as an intermediate frequency signal IFOUT.

In principle, the first frequency mixer 4 downconverts the output PBIST (RF +) on the high frequency side and the output PBIST (RF−) on the low frequency side to the same frequency, and is therefore shown on the right side of FIG. As described above, the frequency of the mixed signal is output as fBIST_CLK.

3A to 3D show a self-diagnosis processing flow chart. At the time of self-diagnosis, as shown in FIG. 3A, for example, the controller 10 sets the frequency fLO2 of the second local signal LO2 to, for example, 80 GHz, and sets the frequency fBIST_CLK of the self-diagnosis clock signal BIST_CLK to, for example, 20 MHz. Then, the second frequency mixer 5 couples the signal of the frequency fLO2 ± fBIST_CLK (80 GHz ± 20 MHz) to the high frequency input RxIN by using the coupler 2. After that, as shown in FIG. 3B, the controller 10 adjusts the phase Δφ by the phase shifter 8 to perform variable gain control of one tone output from the first frequency mixer 4 as an intermediate frequency signal IFOUT. ..

At this time, the controller 10 utilizes the image interference between the two tones of the frequency fLO2 ± fBIST_CLK and adjusts the phase Δφ of the phase shifter 8 so as to minimize the signal level of the output of the first frequency mixer 4. It is desirable to control. As shown in FIG. 3C, the output of the intermediate frequency band by the first frequency mixer 4 is shown. By adjusting and controlling the phase Δφ of the phase shifter 8, the controller 10 adjusts the level from, for example, -30 dBm to a minimum of -40 dBm. Change. Then, as shown in FIG. 3D, at the time of self-diagnosis, the gain variable range of the intermediate frequency amplifier 7 can be widened as much as possible, for example, from a minimum of 0 dB to a maximum of 30 dB. As a result, it is possible to realize a gain variable range that is difficult to realize only by the gain (for example, several dB) by the high frequency amplifier 3 that amplifies the millimeter wave band.

FIG. 4 shows the phase relationship between the two tones when the phase Δφ changes. FIG. 4 shows the two tones generated by the second frequency mixer 5 as a desired wave (Desired tone) and an image wave (Image tone), respectively. The phase shifter 8 can change the phase Δφ from 0 ° to 360 °.

FIG. 5 shows the output theoretical result after the first frequency mixer 4 synthesizes these two tones. As shown in FIG. 5, the output can be maximized when the phase Δφ of the phase shifter 8 is 90 ° and 270 °.

Theoretically, when the first frequency mixer 4 down-converts a signal having two tones of the same phase, for example, the frequencies of the desired wave and the image wave after the down-conversion are the same frequency and are out of phase with each other, that is, Notationally ± fBIST_CLK.

On the other hand, when the controller 10 changes the phase Δφ by the phase shifter 8, the desired wave and the image wave simultaneously change their phases in opposite directions, so that the phase change between the two tones is doubled. .. Therefore, in theory, by setting the phase Δφ of the phase shifter 8 to 90 ° or 270 °, the phases at the frequency fBIST_CLK after down-converting the desired wave and the image wave can be in phase with each other. This allows these two tones to strengthen each other to maximize the output.
On the contrary, the controller 10 sets the phase Δφ of the phase shifter 8 to 0 ° and 180 ° so that the phases at the frequency fBIST_CLK after down-converting the desired wave and the image wave are in an opposite phase relationship with each other. Can be done. Thereby, these desired waves and image waves weaken each other, so that the output can be minimized.

Further, FIG. 6 shows the evaluation results based on these theories. In FIG. 6, the horizontal axis is the phase Δφ of the phase shifter 8, and the vertical axis is the output power (dBm) of the mixed signal of the first frequency mixer 4. The phase at which the output power peaks depends on the initial phase value of the first local signal LO1 supplied by the first frequency mixer 4 at the time of down conversion. In the notation condition of FIG. 6, unlike the notation condition of FIG. 5, it was confirmed that the output power peaked when the phase Δφ was 0 °.

Considering the evaluation result of FIG. 6 relative to the above-mentioned theoretical value, the maximum level is reached when the phase Δφ = 0 ° and 180 ° of the phase shifter 8, and when the phase Δφ = 90 ° and 270 °. It becomes the minimum level. As shown in FIG. 6, when the phase shifter 8 changes the phase Δφ and compares the maximum level at Δφ = 0 ° with the minimum level at Δφ = 90 °, an output change exceeding 20 dB can be confirmed. .. By using this output change at the time of self-diagnosis, self-diagnosis can be performed while increasing the inspectable range.

FIG. 7 shows the evaluation result of evaluating the gain of the intermediate frequency amplifier 7 in the frequency domain. In FIG. 7, the frequency fBIST_CLK of the self-diagnosis clock signal BIST_CLK is set to 22 MHz, and the gain of the intermediate frequency amplifier 7 is observed using a spectrum analyzer when the gain is changed to low gain IFlo, medium gain IFmid, and high gain IFhi. The example is shown.

As is clear from the measurement example of FIG. 7, since the same PLL6 generates all the first and second local signals LO1 and LO2 and the self-diagnosis clock signal BIST_CLK, the first frequency mixer It was confirmed that the frequency of one tone after the down conversion by 4 could be matched with fBIST_CLK within an allowable range, and the frequency accuracy could be extremely high.

That is, after the second frequency mixer 5 generates two tones on the high frequency side, the first frequency mixer 4 down-converts the signal after the phase shift by the phase shifter 8 to tone the frequency fBIST_CLK. This tone can be made into one tone even though it produces. This is because the outputs of the first and second local signals LO1 and LO2 can be synchronized by the same PLL6. That is, it can be guaranteed that the offset frequency fBIST_CLK used at the time of up-conversion always becomes the same offset frequency fBIST_CLK even when down-converted to the intermediate frequency band.

<Summary and effect of this embodiment>
As described above, according to the present embodiment, at the time of self-diagnosis, the same PLL6 generates all the self-diagnosis clock signals BIST_CLK together with the first and second local signals LO1 and LO2, and the second frequency mixing. The instrument 5 up-converts the self-diagnosis clock signal BIST_CLK to the second local signal LO2 generated by the PLL 6 to generate two tones, and causes the coupler 2 to input the high frequency input RxIN.
Therefore, at the time of self-diagnosis, the first frequency mixer 4 down-converts the two tones input to the high frequency input RxIN by the first local signal LO1, so that one tone of the frequency fBIST_CLK becomes high frequency accuracy. Can be output. As a result, when the receiver 1 self-diagnoses using the signal obtained from the high frequency input RxIN through the high frequency amplifier 3, the first frequency mixer 4, and the intermediate frequency amplifier 7, it is possible to perform self-diagnosis at a frequency other than DC with high accuracy. You can self-diagnose. In addition, self-diagnosis can be performed without complicating the scale of the chip of the semiconductor integrated circuit.

In particular, the intermediate frequency amplifier 7 is configured to variably amplify the intermediate frequency signal after down-conversion by the first frequency mixer 4, from the high frequency input RxIN to the high frequency amplifier 3 and the first frequency mixer 4. , And when self-diagnosis is performed using the signal obtained through the intermediate frequency amplifier 7, the amount of change in AC gain of the intermediate frequency amplifier 7 can be self-diagnosed with high accuracy.

Further, the phase shifter 8 is configured in the path from the second frequency mixer 5 to the high frequency input RxIN, and changes the phase Δφ between the two tones output by the second frequency mixer 5. It is configured as follows. Therefore, the gain of the output of the first frequency mixer 4 can be adjusted.

Further, by adjusting the phase Δφ by the phase shifter 8, the controller 10 can also perform variable gain control of one tone output from the first frequency mixer 4. Therefore, the inspectable range of the reception gain of the receiver 1 and the gain of the intermediate frequency amplifier 7 can be expanded while being configured as compact as possible. At this time, it is desirable that the controller 10 adjusts the phase Δφ of the phase shifter 8 so as to make the signal level of the output of the first frequency mixer 4 the lowest by utilizing the image interference. Then, the variable gain range of the intermediate frequency amplifier 7 can be widened as much as possible.

(Modification 1 of the first embodiment)
The phase shifter 8 has a function as an amplitude adjuster that attenuates the amplitude of the two-tone signal output by the second frequency mixer 5 with an attenuation degree α and amplifies the two-tone signal. May be. Further, an amplitude adjuster for adjusting the amplitude of the two-tone signal may be provided in the front stage or the rear stage of the phase shifter 8. In this case, the controller 10 may have a function as an amplitude adjusting unit for adjusting and controlling the attenuation degree α or the amplification degree of the amplitude of the two-tone signal. Then, the controller 10 can control the output of the first frequency mixer 4 with variable gain.

(Modification 2 of the first embodiment)
Although the description is omitted in the first embodiment described above, when a low-pass filter (not shown) is connected to the subsequent stage of the intermediate frequency amplifier 7, the controller 10 sets the frequency fBIST_CLK of the self-diagnosis clock signal BIST_CLK. By changing it, it is possible to check the frequency characteristics such as the cutoff frequency of the low-pass filter and its gain.

(Second Embodiment)
8 and 9 show additional explanatory views of the second embodiment. In the first embodiment, the second frequency mixer 5 is shown by a block diagram and described, but in the present embodiment, a desirable electrical configuration example of the second frequency mixer 5 will be described.

FIG. 8 shows an electrical configuration diagram of a second frequency mixer 105 that replaces the second frequency mixer 5. As shown in FIG. 8, the second frequency mixer 105 inputs the amplification stage 11 that amplifies the second local signal LO2 with a predetermined amplification degree and the self-diagnosis clock signal BIST_CLK to the gate as an input terminal. It is composed of a switch stage 12 including first to fourth transistors T1 to T4, and a Gilbert mixer which outputs a high frequency to a load 13. The first to fourth transistors T1 to T4 are each composed of, for example, a MOS transistor.

The gates of the first and fourth transistors T1 and T4 are commonly connected, and the gates of the second and third transistors T2 and T3 are commonly connected. Further, the source as the energizing terminal on the amplification stage 11 side of the first and third transistors T1 and T3 and the source as the energizing terminal on the amplification stage 11 side of the second and fourth transistors T2 and T4 are , Each is connected in common. Further, the drain as the energizing terminal on the load 13 side of the first and second transistors T1 and T2 and the drain as the energizing terminal on the load 13 side of the third and fourth transistors T3 and T4, respectively. It is a common connection.

Further, in the present embodiment, DC bias Vb is input to the gates of the first and fourth transistors T1 and T4 and the gates of the second and third transistors T2 and T3, respectively, through resistors 14 and 15, respectively. .. The switches SW1 and SW2 are connected to the supply points of the DC bias Vb for the gates of the second and third transistors T2 and T3, and the controller 10 controls the switches SW1 and SW2 on and off. The DC bias Vb can be switched and input. In the present embodiment, the controller 10 turns off the switch SW1 and turns on the switch SW2 to input the DC bias Vb to the gates of the transistors T1 and T3.

A self-diagnosis clock signal BIST_CLK is input to the gates of the second and third transistors T2 and T3 through the DC cutoff capacitor 16. Further, a self-diagnosis clock signal BIST_CLK is input to the gates of the first and fourth transistors T1 and T4 through the DC cutoff capacitor 17. Further, a second local signal LO2 is input to the sources of the first and second transistors T1 and T2 via the amplification stage 11. A second local signal LO2 is input to the sources of the third and fourth transistors T3 and T4 via the amplification stage 11. Therefore, the first to fourth transistors T1 to T4 can output a signal obtained by mixing the second local signal LO2 and the self-diagnosis clock signal BIST_CLK to the high frequency input RxIN through the load 13.

FIG. 9 shows the evaluation result of the two-tone mixed signal by the second frequency mixer 105. As shown in FIG. 9, the second frequency mixer 105 outputs the leakage signal of the second local signal LO2 and other harmonic signals in addition to the two tones which are the fundamental waves of the mixed signal. Since the two-tone signal of the fundamental wave has a larger output than other signals, it has been confirmed that the two-tone mixed signal can be made practically suitable.
As described above, according to the present embodiment, since the configuration of the Gilbert mixer is used as the second frequency mixer 105, it can be made suitable for practical use.

Note that the switches SW1 and SW2 shown in FIG. 8 have a configuration corresponding to a switching unit for generating one tone, as shown in the fourth embodiment described later, and in the present embodiment, these switches SW1 and SW2 are configured. It is not necessary to provide SW2, and a circuit configuration may be used in which the portion provided with the switch SW1 is opened and the portion provided with the switch SW2 is conductive.

(Third Embodiment)
10 and 11 show additional explanatory views of the third embodiment. In the present embodiment, the second frequency mixer 5 has a function of generating two tones, and the second frequency mixer 5 has a function of generating one tone by using the second local signal LO2.

The receiver 201 shown in FIG. 10 includes a second frequency mixer 205 instead of the second frequency mixer 5. Similar to the second frequency mixer 5 described in the above-described embodiment, the second frequency mixer 205 has a self-diagnosis clock signal BIST_CLK of frequency fBIST_CLK on the second local signal LO2 of frequency fLO2 generated by PLL6. Has a function to generate two tones by up-converting. Since this description is the same as that of the above-described embodiment, the description thereof will be omitted.

Further, the second frequency mixer 205 is configured to generate a one-tone signal of the frequency fLO2 by passing through the second local signal LO2 of the frequency fLO2 (= fLO) generated by the PLL6. There is. On the left side of FIG. 11, the output PBIST of the frequency fLO2 of this one tone is shown.

When a one-tone signal is input from the second frequency mixer 205, the phase shifter 8 shown in FIG. 10 shifts the one-tone signal by the phase φ and outputs this signal to the high-frequency input RxIN. The coupler 2 is configured as a high frequency input RxIN, and couples a one-tone signal of frequency fLO2 output from the phase shifter 8 to be input to the high frequency amplifier 3.

The high frequency amplifier 3 amplifies this one-tone signal and outputs it to the first frequency mixer 4. The first frequency mixer 4 mixes the first local signal LO1 of the frequency fLO output by the PLL 6 with the one-tone signal of the frequency fLO2 amplified by the high frequency amplifier 3. In this case, since the first frequency mixer 4 mixes the signal of frequency fLO2 with the signal of frequency fLO and outputs it as an intermediate frequency signal IFOUT, as shown on the right side of FIG. 11, DC (DC) is used as a mixed signal. Output the component.

At this time, the controller 10 can adjust the amplitude of the output PBIST by adjusting the output of the PLL 6, for example. Further, as shown in the first modification of the first embodiment, when the phase shifter 8 has an amplitude adjusting function for adjusting the amplitude attenuation α and the amplification degree, or when the amplitude adjuster is separately provided. Can control the output of the first frequency mixer 4 in a variable gain by adjusting and controlling the attenuation α or the amplification degree of the amplitude of the signal of one tone by the controller 10. This allows the DC offset self-diagnosis test to be performed over a wide gain range.

As described above, according to the present embodiment, the second frequency mixer 205 is provided with a function of generating one tone. As a result, in addition to the effects shown in the above-described embodiment, the DC offset self-diagnosis test can be performed, and the degree of freedom in selecting the inspection frequency can be improved. That is, AC gain inspection can be performed and DC offset inspection can be performed.

(Fourth Embodiment)
12 to 14 show additional explanatory views of the fourth embodiment. In the third embodiment, the second frequency mixer 205 has been described using the block configuration, but in the present embodiment, a practical example having the configuration shown in FIG. 8 is shown. FIG. 12 shows the same configuration as the configuration of the second frequency mixer 105 shown in FIG. 8, and also serves as an operation explanatory diagram when generating one tone.

When the controller 10 generates one tone by the second frequency mixer 5, the switch SW1 is turned on and the switch SW2 is turned off. As a result, the gates of the transistors T2 and T3 can be brought to the ground potential through the resistor 15 while applying the DC bias Vb to the gates of the transistors T1 and T4. At this time, since the transistors T2 and T3 are always turned off, the functions of the transistors T2 and T3 can be completely disabled. This makes it possible to invalidate the input of the self-diagnosis clock signal BIST_CLK. The controller 10 may stop and control the output of the self-diagnosis clock signal BIST_CLK from the PLL 6.

Since the DC bias Vb is applied to the gates of the transistors T1 and T4, the transistors T1 and T4 can be effectively functioned. Since the second frequency mixer 105 at this time has the same circuit as the cascode connection amplifier, the second local signal LO2 input to the sources of the transistors T1 and T4 can be amplified and output to the load 13. FIG. 13 shows the evaluation result of the one-tone signal. As shown in FIG. 13, it was confirmed that a one-tone signal can be generated to such an extent that harmonics can be sufficiently ignored.

FIG. 14 shows a change in the output DC offset voltage (pp) of the first frequency mixer 4 corresponding to the change in the phase φ of the phase shifter 8 when the output of this one tone is applied. FIG. 14 shows the result of self-diagnosis by connecting an A / D converter (not shown) to the output of the first frequency mixer 4 and reading the DC offset voltage (pp).
The horizontal axis represents the phase shift code corresponding to the amount of the phase φ of the phase shifter 8, and the vertical axis represents the DC offset voltage. Further, Gmax is a DC offset voltage value when the configuration of the present embodiment is applied and the gain of the high frequency amplifier 3 is the maximum gain, and Gmin is a DC offset voltage value when the gain of the high frequency amplifier 3 is the minimum gain. Is shown. As shown in FIG. 14, the DC offset voltage can be self-diagnosed.

When the controller 10 changes the phase φ of the phase shifter 8, the first frequency mixer 4 mixes the frequency fLO2 (= fLO) signal from the high frequency amplifier 3 and the frequency fLO signal by PLL6. do. Therefore, the first frequency mixer 4 changes the amplitude of the DC offset output by strengthening or weakening these input signals. At this time, the controller 10 can self-diagnose the DC offset voltage by reading the DC output of the first frequency mixer 4.

The technique described in Patent Document 1 is configured to detect the DC offset voltage, but it has been confirmed that the maximum response range when the gain changes is also small and the inspectable range is narrow. There is. In the present embodiment, the verification result of the gain of the DC offset can be confirmed as in the prior art. That is, according to the configuration of the present embodiment, it becomes possible to flexibly deal with the self-diagnosis process of the DC offset.

As described above, according to the present embodiment, the controller 10 can generate one tone by turning on the switch SW1 of the second frequency mixer 105 and turning off the switch SW2. The controller 10 can supply / stop the DC bias Vd for, for example, the second and third transistors T2 and T3 by switching the switches SW1 and SW2 on / off, for example, and the second frequency mixer 105. It is possible to switch between the two-tone generation function and the one-tone generation function. Therefore, it is possible to flexibly support the self-diagnosis process of DC offset.

(Fifth Embodiment)
FIG. 15 shows an additional explanatory diagram of the fifth embodiment. In this embodiment, an example when applied to the millimeter-wave radar system S for a mobile body will be described, but the parts having the same or similar functions as those in the third or fourth embodiment are described with the same reference numerals. Will be omitted to explain the different parts.
The millimeter wave radar system S for a moving body is mounted so as to be able to transmit radar waves in front of the moving body, for example, and the transmitter 21 outputs the radar waves in the millimeter wave (for example, 80 GHz band: 76 GHz) band toward the target T. Then, the reflected wave WR1 is received by using the receiver 301. The target T is, for example, another moving body such as a preceding vehicle, a roadside object on the road, or the like.

The millimeter-wave radar system S includes a multiplier 25, a phase shifter 26, an amplifier 27, a coupler 28, a detector 29 for detecting a transmission signal from the coupler 28, and a transmission antenna 30 as a transmitter 21. This millimeter wave radar system S includes a receiving antenna 9, a coupler 2, a high frequency amplifier 3, a first and second frequency mixer 4,205, an intermediate frequency amplifier 7, a phase shifter 8, a multiplier 22, 23, and a variable amplifier. 24, PLL 6 is provided as a receiver 301, and a controller 10 is further provided. The receiver 301 includes a double multiplier 22 between the PLL 6 and the first frequency mixer 4, and a double multiplier 23 between the PLL 6 and the second frequency mixer 205.
The PLL 6 includes a VCO 6a, and in response to a frequency command commanded by the controller 10, for example, a modulated signal that gradually increases / decreases the frequency in time is generated using the reference clock CLK, and this modulated signal is output to the multiplier 25. do.

The multiplier 25 multiplies the output of the PLL 6 by, for example, two, the phase shifter 26 shifts this signal, and the amplifier 27 amplifies it. The transmitter 21 transmits and outputs the signal amplified by the variable amplifier 24 through the transmission antenna 30. As shown in FIG. 15, the radar transmission wave WT1 is reflected by the target T to generate a reflected wave WR1. This reflected wave WR1 is input to the receiving antenna 9 as a received wave.

The receiver 301 is configured to include the multipliers 22 and 23 and the variable amplifier 24 in the configuration shown in FIG. 1 of the above-described embodiment. The gain of the variable amplifier 24 can be controlled by the controller 10. Therefore, by setting the signal generation frequency of the PLL 6 to a frequency of about 1/2 (for example, 38 GHz) as compared with the above-described embodiment, the same effect as that of the receiver 1 of the above-mentioned embodiment can be obtained.

Information about the target T (for example, own machine and target T) by executing signal processing based on the intermediate frequency signal IFOUT acquired by the receiver 301 or the processing signal of the subsequent stage by an external microcomputer (not shown). The distance R between and, the relative velocity, the direction) is acquired.

In this millimeter wave radar system S, the radar transmission wave WT1 is reflected by the target T located at the distance R to generate the reflected wave WR1. Further, if the component Tb mounted on a moving body such as a bumper or a windshield is installed at a distance Rb closer than this distance R, a reflected wave WR2 reflected by this component Tb is also generated. Therefore, when the reflected wave WR2 is input to the receiving antenna 9, it becomes noise.

In order to detect and cancel this noise level, the controller 10 switches the output of the second frequency mixer 205 to a mode for outputting one tone of frequency fLO2 (= fLO) and outputs the noise level. Good to detect. Then, when the transmission frequency of the radar transmission wave WT1 is the frequency fLO, the reception frequencies of the reflected waves WR1 and WR2 also become the frequency fLO, and the controller 10 shifts the phase according to the intensity and phase of the reflected wave WR2 of this frequency fLO. By controlling the phase Δφ of the device 8 and the gain of the variable amplifier 24, the noise caused by the reflected wave WR2 can be reduced, whereby the noise can be detected and the noise can be canceled.

According to the present embodiment, since the second frequency mixer 205 has a function of outputting one tone of frequency fLO, it can be practically used even when applied to the millimeter wave radar system S. Moreover, it can have a function as a noise canceller.

(Sixth Embodiment)
FIG. 16 shows an additional explanatory diagram of the sixth embodiment. FIG. 16 is an electrical configuration diagram of the receiver 401 shown in place of the receiver 1 of FIG. As shown in the receiver 401 of FIG. 16, the phase shifter 408 may be provided in the output path of the second local signal LO2 from the PLL 6 to the second frequency mixer 5. The phase shifter 408 shifts the phase Δφ of the second local signal LO2 having the frequency fLO2 under the control of the controller 10.
Theoretically, the second frequency mixer 5 can generate and output a two-tone signal having a frequency of fLO2 ± fBIST_CLK, as in the configuration of FIG. Other configurations are the same as those in the first embodiment described above, and have the same effects.

(7th Embodiment)
FIG. 17 shows an additional explanatory diagram of the seventh embodiment. FIG. 17 is an electrical configuration diagram of the receiver 501 shown in place of FIG. As shown in the receiver 501 of FIG. 17, the phase shifter 508 may be provided in the output path of the self-diagnosis clock signal BIST_CLK from the PLL 6 to the second frequency mixer 5. Theoretically, the second frequency mixer 5 can generate and output a two-tone signal having a frequency of fLO2 ± fBIST_CLK, as in the configuration of FIGS. 1 and 16. Other configurations are the same as those of the above-described embodiment, and have the same effects as those of the above-mentioned embodiment.

(Other embodiments)
The present disclosure is not limited to the above-described embodiment, but can be variously modified and implemented, and can be applied to various embodiments without departing from the gist thereof. For example, the following modifications or extensions are possible.

The phase shifters 8, 408 and 508 may be provided as needed. In the first embodiment, the same PLL 6 produces and outputs the frequency fLO of the first local signal LO1 generated and output to the first frequency mixer 4, and the second local signal LO2 generated and output to the second frequency mixer 5. Although the frequency fLO2 of is the same, it may be set so as to be different from each other. For example, when the frequencies fLO and fLO2 of the first and second local signals LO1 and LO2 output by the same PLL6 are output at different frequencies, either or both of them are multiplied by a multiplier (not shown). Alternatively, it may be configured to divide the frequency by a frequency divider (not shown) and output. In such a case, it is sufficient that the frequency of the mixed signal to be mixed and output by the first frequency mixer 4 can be set to the frequency fBIST_CLK at the time of self-diagnosis.

The configurations and functions of the plurality of embodiments described above may be combined. An embodiment in which a part of the above-described embodiment is omitted as long as the problem can be solved can also be regarded as an embodiment. In addition, any conceivable embodiment can be regarded as an embodiment as long as it does not deviate from the essence of the invention specified by the wording described in the claims.

Although the present disclosure has been described in accordance with the embodiments described above, it is understood that the present disclosure is not limited to such embodiments or structures. The present disclosure also includes various variations and variations within a uniform range. In addition, various combinations and forms, as well as other combinations and forms, including one element, more, or less, are within the scope and scope of the present disclosure.

In the drawings, 1,201,301,401,501 are receivers (self-diagnosis devices), 3 is a high frequency amplifier, 4 is a first frequency mixer, 5,105,205 are second frequency mixers, and 6 is. PLL, 7 is an intermediate frequency amplifier, 8,408,508 is a phase shifter, 10 is a controller (phase shift adjustment unit, amplitude adjustment unit, switching unit), 10 is an amplification stage, 12 is a switch stage, and 13 is a load. T1 to T4 are first to fourth transistors, and S is a mobile millimeter-wave radar system.

Claims (9)

A high frequency amplifier (3) that amplifies a high frequency signal input to a high frequency input through an antenna, a PLL (6) that generates a first local signal, and a first that downconverts the high frequency signal by the first local signal. The frequency mixer (4) and the intermediate frequency amplifier (7) for amplifying the intermediate frequency signal output by the first frequency mixer are provided, and the high frequency input is provided with the high frequency amplifier and the first frequency amplifier. A self-diagnosis device (1,201, 301, 401, 501) that self-diagnoses using a signal obtained through a frequency mixer and the intermediate frequency amplifier.
With the PLL
With a second frequency mixer (5,105,205),
The PLL generates the first local signal and also generates a second local signal for self-diagnosis and all self-diagnosis frequency signals that satisfy a frequency condition lower than the second local signal and exceeding DC. Configured,
The second frequency mixer is a self-diagnosis device that up-converts the self-diagnosis frequency signal to the second local signal generated by the PLL to generate two tones and input them to the high-frequency input. The self-diagnosis apparatus according to claim 1, further comprising a phase shifter (8) that is located in a path from the second frequency mixer to the high frequency input and changes the phase of the two tones. The self-diagnosis apparatus according to claim 1 or 2, wherein the second frequency mixer has a function of generating one tone by using the second local signal as well as a function of generating the two tones. A phase shifter (8) for changing the phase of the two tones is provided.
The self-diagnosis according to any one of claims 1 to 3, further comprising a phase shift adjusting unit (10) for variable gain control of the output of the first frequency mixer by adjusting the phase with the phase shifter. Device. The self-diagnosis apparatus according to claim 4, wherein the phase shift adjusting unit uses image interference to adjust the phase of the phase shift mixer so as to minimize the signal level of the output of the first frequency mixer. An amplitude adjuster (8) for adjusting the amplitude of the two tones is provided.
The invention according to any one of claims 1 to 5, further comprising an amplitude adjusting unit (10) for variable gain control of the output of the first frequency mixer by adjusting the amplitude of the two tones with the amplitude adjuster. Self-diagnosis device. The second frequency mixer (105) is
An amplification stage (11) for amplifying the second local signal and a switch stage (12) including first to fourth transistors (T1 to T4) for inputting the self-diagnosis frequency signal to the input terminal are provided. The Gilbert mixer provided generates the two tones in the load (13) and inputs them to the high frequency input.
The input terminals of the first and fourth transistors are commonly connected, the input terminals of the second and third transistors are commonly connected, and a DC bias is applied to the input terminals of the second and third transistors. Switching input is possible,
The energizing terminals on the amplification stage side of the first and third transistors are commonly connected, and the energizing terminals on the amplification stage side of the second and fourth transistors are commonly connected.
The energization terminals on the load side of the first and second transistors are commonly connected, and the energization terminals on the load side of the third and fourth transistors are commonly connected. Ori,
The self-diagnosis apparatus according to any one of claims 1 to 6, further comprising a switching unit (10) for switching between the two-tone generation function and the one-tone generation function by supplying / stopping the DC bias. The self-diagnosis apparatus according to claim 1, wherein the phase shifter (408) is provided in the output path of the second local signal from the PLL to the second frequency mixer. The self-diagnosis apparatus according to claim 1, further comprising a phase shifter (508) in the output path of the self-diagnosis frequency signal from the PLL to the second frequency mixer.

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