JP6815637B2 - Electron spin resonance device - Google Patents

Description

The present invention relates to an electron spin resonance apparatus capable of imaging free radicals in a living body using electron spin resonance (EPR: Electron Paramagnetic Resonance).

In general, the EPR measurement method is a method of measuring an atom or molecule having an unpaired electron such as a free radical by utilizing the motion of the magnetic moment of the electron. By EPR measurement, for example, free radicals in small animals can be imaged.

The continuous wave EPR (CW-EPR) method is mainly adopted in the electron spin resonance apparatus. The CW-EPR method is a method of measuring an EPR signal by making the frequency of a high frequency (microwave) supplied to a resonator constant and performing magnetic field sweeping. In the EPR measurement, when EPR absorption occurs, the matching of the resonator is deviated and a reflected wave is generated, and the EPR signal can be acquired by detecting the intensity and phase of the reflected wave. At this time, in order to increase the sensitivity, the measurement is further applied with magnetic field modulation.

However, since the reflected wave is generated by the mismatch of the resonator, it is also generated by the mismatch of the resonator due to the movement of a small animal or the like. That is, in order to measure the reflected wave due to EPR absorption, it is necessary to suppress the reflected wave due to the movement of a small animal or the like.

Therefore, an electron spin resonance device incorporating an automatic tuning circuit that tunes the resonance frequency of the resonator to a high frequency frequency supplied to the resonator and an automatic matching circuit that matches the impedance of the resonator has been proposed ( For example, see Patent Document 1). According to the technique of Patent Document 1, EPR measurement can be stably performed.

Japanese Unexamined Patent Publication No. 2007-10331

In the technique of Patent Document 1, the deviation from the resonance frequency of the resonator is detected by FM-modulating the high frequency, and the resonance frequency of the resonator is stabilized by negative feedback. Further, the matching adjustment signal is AM-modulated and the phase detection is performed to detect the degree of matching of the resonator, and negative feedback is performed to stabilize the matching of the resonator.

However, in order to suppress the reflected wave by using these modulation methods, an analog control circuit including a modulation frequency generation circuit, a phase detection circuit, a feedback signal generation circuit, etc. is used, and the configuration of the electron spin resonance device is complicated. It has become. As the configuration of the electron spin resonance apparatus becomes complicated, there are problems such as deterioration of the SN ratio, which is the ratio of the signal (Signal) and noise (Noise) of the EPR signal, and the cost increase of the electron spin resonance apparatus.

Therefore, the present invention has been made in view of the above problems, and an object of the present invention is to provide an electron spin resonance apparatus having a simplified configuration.

In order to achieve the above object, the electron spin resonance device according to one embodiment of the present invention has a resonance portion having a matching tuning circuit and a resonator, and an oscillator, and resonates a high frequency based on the oscillation frequency of the oscillator. The high-frequency output unit that outputs to the oscillator and the I (In-phase) signal and Q (Quadrature-phase) signal that have the intensity information and phase information of the high-frequency reflected wave reflected by the oscillator are the oscillators, respectively. The matching tuning circuit includes a control unit that negatively feeds back to the I signal and the Q signal, and adjusts the impedance of the resonance unit based on one of the I signal and the Q signal, and based on the other of the I signal and the Q signal. The resonance frequency of the oscillator is tuned to the high frequency.

According to this, the I baseband component (called an I signal) and the Q baseband component (called a Q signal) having the intensity information and the phase information of the reflected wave are directly matched and tuned to the resonator (resonant part), respectively. It is negatively fed back to the matching tuning circuit that performs the matching, and the matching and tuning of the resonator (resonant portion) can be performed at the same time. That is, since the conventionally used AM modulation and FM modulation are not used, the analog control circuit becomes unnecessary, and the configuration of the electron spin resonance device can be simplified. Therefore, the SN ratio can be improved, free radicals can be imaged with high accuracy, and the cost of the electron spin resonance apparatus can be reduced.

Further, the electron spin resonance device may further include a phase device that adjusts the phase of the reflected wave to approximately 45 degrees or approximately 225 degrees with respect to the phase of the high frequency incident on the resonator.

According to this, the intensities of the negatively fed I signal and the Q signal can be made substantially the same.

Further, the electron spin resonance device further includes a band path filter for filtering the reflected wave and an AD converter for digitally converting the filtered reflected wave, and the high frequency output unit further oscillates the oscillator. It has a first divider that divides the frequency by m (m is an odd number) × n (n is an integer), and a second divider that divides the oscillation frequency of the oscillator by 4 × n. As a high frequency based on the oscillation frequency of the oscillator, a high frequency of the frequency divided by the second divider is output to the resonator, and the AD converter outputs the frequency divided by the first divider. As the sampling frequency, the filtered reflected wave may be digitally converted.

According to this, bandpass subsampling can be used in which the signal is limited to a desired band by a bandpass filter and information of a frequency higher than the sampling frequency can be extracted, and DC offset and the like can be removed. Further, by using the first frequency divider and the second frequency divider instead of the general-purpose frequency generator, it is possible to generate a high frequency with low phase noise and the clock signal for the AD converter, which are synchronized with each other. it can.

Further, the control unit may be configured by an FPGA (Field Programmable Gate Array).

According to this, for example, a DSP (Digital Signal Processor) such as a DDC (Digital Down Controller), a phaser, and a digital filter used in an electron spin resonance apparatus can be realized by an FPGA. Further, in recent years, since FPGA has been realized at low cost, the cost of the electron spin resonance apparatus can be further reduced.

According to the present invention, an electron spin resonance apparatus having a simplified configuration can be realized.

Therefore, it is expected that the electron spin resonance apparatus according to the present invention will contribute to the spread of EPR technology and will contribute to redox research in the fields of life science and medicine by being used by many life science researchers.

It is a circuit block diagram which shows an example of the electron spin resonance apparatus which concerns on embodiment. It is a block diagram which shows an example of FPGA which concerns on embodiment. It is a circuit block diagram which shows an example of the matching tuning circuit which concerns on embodiment. It is a figure which shows the relationship between the phase of a reflected wave, and the coupling state of a resonator. It is a figure which shows the intensity of the reflected wave at the time of matching and tuning by the matching tuning circuit which concerns on embodiment. It is a figure which shows the EPR spectrum acquired by the electron spin resonance apparatus and the conventional electron spin resonance apparatus which concerns on embodiment.

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings. It should be noted that all of the embodiments described below show a specific example of the present invention. Numerical values, shapes, materials, components, arrangement positions of components, connection forms, etc. shown in the following embodiments are examples, and are not intended to limit the present invention. Further, among the components in the following embodiments, the components not described in the independent claims indicating the highest level concept of the present invention will be described as arbitrary components.

FIG. 1 is a circuit configuration diagram showing an example of an electron spin resonance apparatus 1 according to an embodiment.

The electron spin resonance device 1 is a device for performing EPR measurement by the CW-EPR method, and is a control unit 10, a high frequency output unit 20, a high frequency acquisition unit 30, a resonance unit 40, a magnetic circuit unit 60, a circulator 70, and an LNA ( It is equipped with a Low Noise Amplifier) 80 and a PC (Personal Computer) 90.

The control unit 10 includes an FPGA 100, DA converters (DACs) 11a to 11h, and a USB-IO port 12.

The FPGA 100 is a programmable device that forms an arbitrary circuit configuration by a written program. For example, a DSP such as a DDC, a phaser, or a digital filter used in a general electron spin resonance apparatus is realized by the FPGA 100. The operation of the FPGA 100 will be described in detail with reference to FIG. 2, which will be described later.

DACs 11a to 11h are electronic circuits that convert digital data acquired from the FPGA 100 into analog data. The DAC 11a outputs a magnetic field modulation signal to the amplifier 61a described later. The DAC 11b outputs a signal for performing a magnetic field sweep to the sweep unit 63a described later. The DAC 11c outputs a matching control (MC: Matching Control) signal to the resonator 41 (matching tuning circuit 50) described later. The DAC 11d outputs a tuning control (TC: Tuning Control) signal to the resonator 41 (matching tuning circuit 50). The DACs 11e to 11g output a signal for generating a gradient magnetic field to the gradient magnetic field control unit 62a described later. The DAC 11h supplies a voltage corresponding to the oscillation frequency to the oscillator 21 described later.

The USB-IO port 12 is a connection port into which a USB cable or the like is inserted, and connects the PC 90 or the like to the control unit 10.

The PC 90 is a computer connected to the control unit 10 and transfers data with the control unit 10 (FPGA100). The PC 90 generates digital data for sweeping the magnetic field and digital data for generating the gradient magnetic field, and these data are stored in the RAM (Random Access Memory) of the FPGA 100. Further, the PC 90 performs free radical imaging by processing the EPR data received from the control unit 10 (FPGA100).

The high frequency output unit 20 has an oscillator 21, a first divider 22, a second divider 23, an attenuator 24, and a phaser 25, and resonates a high frequency based on the oscillation frequency of the oscillator 21 via the circulator 70. Output to the device 41. Further, the high frequency output unit 20 outputs a clock signal based on the oscillation frequency of the oscillator 21 to an AD converter (ADC) 33 described later.

The oscillator 21 is, for example, a VCO (Voltage Controlled Oscillator) that controls the oscillation frequency according to the supplied voltage, and the digital data generated by the FPGA 100 and corresponding to the target oscillation frequency is converted into an analog voltage by the DAC 11h. It is output to the oscillation frequency control port of the oscillator 21.

The first frequency divider 22 is a frequency divider that divides the oscillation frequency of the oscillator 21 by m (m is an odd number) × n (n is an integer), and outputs a clock signal to the ADC 33 via the phase controller 25. To do.

The second divider 23 is a divider that divides the oscillation frequency of the oscillator 21 by 4 × n, and outputs a high frequency to the circulator 70 via the attenuator 24.

The resonance frequency of the EPR depends on the external magnetic field. In this embodiment, since the external magnetic field is about 27 mT as described later, about 750 MHz is used as the high frequency. For example, when n = 1 (the division ratio of the second frequency divider 23 is 1/4), 3 GHz is used as the oscillation frequency of the oscillator 21 in order for the high frequency to be 750 MHz. Similarly, when n = 2 (the division ratio of the second divider 23 is 1/8), 6 GHz is used as the oscillation frequency of the oscillator 21 in order for the high frequency to be 750 MHz. In the present embodiment, as the second frequency divider 23, a frequency divider that divides the oscillation frequency of the oscillator 21 by, for example, 4 (n = 1) is used, and the oscillation frequency of the oscillator 21 is 3 GHz. ..

Further, since the ADC 33 is an AD converter of 500 Mbps as described later, the frequency of the clock signal needs to be 500 MHz or less in order to drive the ADC 33. In the present embodiment, since 3 GHz is used as the oscillation frequency of the oscillator 21, the minimum m × n of the first frequency divider 22 is 7 (the frequency of the clock signal is 428.6 MHz). For example, when m × n = 5, the frequency of the clock signal is 600 MHz, and the ADC 33 cannot be driven. When the oscillation frequency of the oscillator 21 is 6 GHz (n = 2), the minimum m × n of the first frequency divider 22 is 14.

For example, when an oscillator with an oscillation frequency of 750 MHz is used, in order to set the frequency of the clock signal to 428.6 MHz, 3 GHz is created using a 4x multiplier, and then the 1st divider 22 is used. It is necessary to divide by 7. In this case, since the multiplier has a large conversion loss, additional phase noise, and the like, the performance of the electron spin resonance apparatus may deteriorate. On the other hand, the electron spin resonance device 1 does not use a multiplier, has an oscillator 21 having an oscillation frequency of 3 GHz, a second frequency divider 23 having a frequency division ratio of 1/4, and a frequency division ratio of 1/4. Since the 1/7 first frequency divider 22 is used, it is possible to prevent the performance of the electron spin resonance apparatus 1 from deteriorating. Further, by using the second frequency divider 23, it is possible to reduce the high frequency phase noise generated by dividing the frequency from a high frequency. For example, the phase noise is reduced by 6 dB by changing the frequency twice. In the present embodiment, since the division ratio of the second frequency divider 23 is 1/4, the phase noise can be reduced by 12 dB.

The attenuator 24 is, for example, a digital step attenuator, which attenuates the high frequency input from the second divider 23 to an intensity corresponding to the control data supplied from the FPGA 100, and outputs the attenuator to the circulator 70.

The phase device 25 adjusts the phase of the clock signal output to the ADC 33. The phase of the clock signal can be changed, for example, by 360 degrees or more. Although the details will be described later, the phase device 25 adjusts the phase of the detected reflected wave to approximately 45 degrees or approximately 225 degrees with respect to the phase of the high frequency incident on the resonator 41 by adjusting the phase of the clock signal. To do.

The circulator 70 supplies a high frequency from the high frequency output unit 20 to the resonator 41, and transmits the reflected wave reflected by the resonator 41 to the high frequency acquisition unit 30 via the LNA 80. The circulator 70 has an isolation characteristic of at least 30 dB or more in the 750 MHz band as an isolation characteristic between the input port connected to the high frequency output unit 20 and the output port connected to the LNA 80.

The resonance unit 40 has a resonator 41 and a matching tuning circuit 50, as shown in FIG. 3 described later. In FIG. 1, the resonator 41 and the matching tuning circuit 50 are not shown. The resonator 41 is, for example, a loop gap resonator having a one-turn coil, and a sample such as a small animal is placed inside. A high frequency is supplied to the resonator 41 from the circulator 70 via the matching tuning circuit 50 and the coaxial cable. The reflected wave reflected from the resonator 41 in which the sample is placed is returned to the circulator 70 via the matching tuning circuit 50 and the coaxial cable.

The magnetic circuit unit 60 includes a modulation coil 61, an amplifier 61a for amplifying power supplied to the modulation coil 61, a gradient coil 62, a gradient magnetic field control unit 62a, a sweep coil 63, and a magnetic field sweep control unit 63a. The magnetic circuit unit 60 has, for example, a 27 mT permanent magnet (not shown).

The modulation coil 61 is a coil that applies a modulation magnetic field to the sample in order to realize highly sensitive EPR measurement, and the magnetic field modulation signal from the DAC 11a is amplified and supplied to the amplifier 61a. The modulation coil 61 is, for example, a saddle coil and is arranged so as to surround the resonator 41.

The gradient coil 62 is a coil for generating a gradient magnetic field. The gradient magnetic field control unit 62a controls the current flowing through the gradient coil 62 in response to a signal from the DACs 11e to 11g for generating a spatially linear magnetic field in the x, y, and z directions.

The sweep coil 63 is a coil for performing magnetic field sweep. The magnetic field sweep control unit 63a controls the current flowing through the sweep coil 63 in response to a signal for performing magnetic field sweep from the DAC 11b. The sweep coil 63 is, for example, a Helmholtz coil.

The high frequency acquisition unit 30 acquires the reflected wave output from the circulator 70 and amplified by the LNA 80. The high frequency acquisition unit 30 has an LNA 31, a bandpass filter (BPF) 32, and an ADC 33.

The LNA 31 re-amplifies the reflected wave amplified by the LNA 80.

The BPF 32 is a filter that filters the reflected wave amplified by the LNA 31, and is, for example, a SAW (Surface Acoustic Wave) filter. The BPF 32 limits the frequency of the reflected wave to, for example, 740 MHz to 760 MHz in order to perform bandpass subsampling.

The ADC 33 digitally converts the reflected wave filtered by the BPF 32 and outputs the digital data to the FPGA 100. The ADC 33 is, for example, a 500 Mbps, 12-bit high-speed AD converter. The ADC 33 samples the reflected wave at a sampling frequency corresponding to the clock signal from the first frequency divider 22. In the present embodiment, the sampling frequency is 4/7 times the high frequency. As the sampling frequency of the ADC 33, a frequency corresponding to digital down-conversion using bandpass subsampling is selected.

Next, the details of the operation of the FPGA 100 will be described with reference to FIG.

FIG. 2 is a block diagram showing an example of the FPGA 100 according to the embodiment.

The FPGA 100 has DDC unit 101, downsampling unit 102, attenuator 103, FIR (Finite Impulse Response) low-pass filter (LPF) 104, addition / subtractor 105, downsampling unit 106, FIRBPF 107, DDC unit 108, as functional components. It has an integrator 110, an attenuator 111, an NCO (Numerical Controlled Oscillator) 112, and an attenuator 113. In the present embodiment, the FPGA 100 is provided in the control unit 10, but these functional components included in the FPGA 100 may be distributed and arranged in, for example, the control unit 10 and the high frequency acquisition unit 30. Specifically, the DDC unit 101 and the downsampling unit 102 may be realized by the FPGA provided in the high frequency acquisition unit 30, and other functional components may be realized by the FPGA 100.

The DDC unit 101 obtains an I baseband component (called an I signal) and a Q baseband component (called a Q signal) having amplitude information and phase information of the reflected wave (digital data) output from the ADC 33. , The digital down-conversion process, which is the process of frequency-converting the digital data to around 0 Hz, is performed. The DDC unit 101 divides the digital data into two data by performing the first thinning process on the digital data. The DDC unit 101 further divides the one data into two data by performing the second thinning process on one of the two data obtained by the first thinning process. Then, the DDC unit 101 acquires, for example, an I signal by subtracting each of the two data obtained by the second thinning process. The DDC unit 101 acquires, for example, a Q signal by performing the same processing on the other data of the two data obtained by the first thinning process. By each thinning process, the sampling frequency of the data becomes 1/4 of the sampling frequency of the ADC 33, and the DC offset caused by the ADC 33 is canceled by the subtraction.

The phase of the reflected wave is adjusted to approximately 45 degrees or approximately 225 degrees with respect to the phase of the high frequency incident on the resonator 41 by, for example, adjusting the phase of the clock signal of the ADC 33 using the phase device 25. As will be described later, the phase of the reflected wave may be similarly adjusted by signal processing by the phase device 103 mounted on the FPGA 100. In this case, the high frequency output unit 20 does not have to have the phase device 25. By these methods, the intensities of the I signal and the Q signal are adjusted to be substantially the same.

The downsampling unit 102 lowers the sampling frequency by integrating and thinning out the sampling frequencies of the I signal and the Q signal down-converted to around 0 Hz. For example, the downsampling unit 102 outputs an I signal and a Q signal whose frequency is 1/64 of the sampling frequency of the ADC 33.

The phase device 103 is a complex number divider that performs processing represented by (I + jQ) × (cosθ + jsinθ). However, j is an imaginary unit, and θ is a phase quantity to be shifted. The phase device 103 adjusts the phase of the reflected wave to approximately 45 degrees or approximately 225 degrees with respect to the phase of the high frequency incident on the resonator 41 in order to make the signal intensities of the I signal and the Q signal substantially the same. The phase shift amount can be controlled by transferring a control signal from the PC 90 to the FPGA 100. Further, by integrating the difference between the absolute values of the I signal and the Q signal and negatively feeding back to the phase device 103, the phase of the reflected wave is approximately 45 degrees or approximately 225 degrees with respect to the phase of the high frequency incident on the resonator 41. Can be automatically controlled.

The FIRLPF 104 is a low frequency pass type digital filter, and is processed by the downsampling unit 106 by removing noise of a high frequency component that is more than twice the frequency of magnetic field modulation by the modulation coil 61 before the downsampling 106. It reduces the folding component of high frequency noise caused by.

The I signal and Q signal output from the FIRLPF 104 are negatively fed back to the resonator 40 in order to suppress the reflected wave due to the mismatch caused by the impedance change of the resonator 41 generated by the movement of the sample (small animal or the like). However, the I signal and Q signal output from FIRLPF104 are also used for generating the EPR signal. First, the process for generating the EPR signal will be described.

The adder / subtractor 105 performs subtraction and addition of the I signal and the Q signal. The reflected wave generated by EPR absorption is 90 degrees out of phase with the reflected wave generated by EPR dispersion. Further, the reflected wave due to EPR dispersion has the same phase as the reflected wave due to inconsistency due to fluctuations in the characteristics of the resonator 41 due to temperature, movement of animals, and the like. Since the phase is adjusted by using, for example, a phase device 103 so that the intensities of the I signal and the Q signal are almost the same, the reflected wave generated by EPR absorption remains due to the subtraction of the I signal and the Q signal. , The reflected wave generated by EPR dispersion disappears. On the other hand, by adding the I signal and the Q signal, the reflected wave generated by the EPR dispersion remains, and the reflected wave generated by the EPR absorption disappears. However, by adjusting the phase while checking the measured EPR spectrum, it is possible to reliably extract only the EPR absorption in the end.

Although the reflected wave from the resonator 41 is suppressed by the matching tuning circuit 50 described later, since the modulated magnetic field is generated by the modulation coil 61, the EPR signal (reflected wave generated by EPR absorption and reflected wave generated by EPR dispersion). ) Is a high frequency signal. Therefore, when the bandwidth of the signal negatively fed back to the resonance portion 40 is smaller than the frequency of the modulated magnetic field, the EPR signal at the frequency of the modulated magnetic field remains unsuppressed. For example, since the frequency of the modulated magnetic field is 105 kHz, which is higher than the frequency of movement of small animals and the like, the reflected wave due to the movement of small animals and the like disappears, and the high frequency EPR signal remains. Then, the EPR signal is digitally bandpass subsampled and demodulated by the downsampling unit 106, FIRBPF107, and DDC unit 108.

The downsampling unit 106 performs thinning processing so that the modulated EPR signal output from the adder / subtractor 105 becomes an EPR signal sampled at a frequency four times the modulation frequency. For example, the downsampling unit 106 performs thinning processing so that the modulated EPR signal becomes an EPR signal sampled at a frequency of 1/1024 of the sampling frequency in the ADC 33.

The FIRBPF 107 is a band-passing type digital filter applied to the modulated EPR signal output from the downsampling unit 106 and passing a signal near the modulation frequency. The FIRBPF 107 is, for example, a digital filter having 301 taps, and limits the bandwidth to about 5 kHz in order to perform digital bandpass subsampling.

The DDC unit 108 demodulates the modulated EPR signal and performs a digital down-conversion process for obtaining the I signal and the Q signal. Further, the DDC unit 108 divides the EPR signal into two data by performing the first thinning process on the EPR signal. The DDC unit 108 further divides the one data into two data by performing the second thinning process on one of the two data obtained by the first thinning process. Then, the DDC unit 108 acquires, for example, an I signal by subtracting each of the two data obtained by the second thinning process. The DDC unit 108 acquires, for example, a Q signal by performing the same processing on the other data of the two data obtained by the first thinning process. For example, the DDC unit 108 outputs an EPR signal down-converted to around 0 Hz at a sampling frequency of 1/4096 of the sampling frequency of the ADC 33.

In order to synchronize the timing of the digital bandpass subsampling and the magnetic field modulation, the NCO 112 generates a magnetic field modulation signal using the clock signal of the ADC 33, and the attenuation unit 113 adjusts the strength of the magnetic field modulation signal. Thereby, the phase and intensity of the modulated magnetic field generated from the modulation coil 61 can be adjusted.

In this way, the EPR signal is demodulated and the EPR data is transferred to the PC90.

Next, a process for suppressing the reflected wave due to the mismatch caused by the impedance change of the resonator 41 generated by the movement of the sample (small animal or the like) will be described.

The integrator 110 is required to match the reflected wave with the control target of zero reflected wave, and integrates the I signal and the Q signal output from FIRLPF104.

The attenuation unit 111 adjusts the intensities of the integrated I signal and Q signal.

Then, the I signal is analog-converted by the DAC 11c. The analog-converted I signal is used as an MC signal for matching control of the resonator 41. Further, the Q signal is analog-converted by the DAC11d. The analog-converted Q signal is used as a TC signal for tuning control of the resonator 41. The voltage range of the MC signal and the TC signal is, for example, 0V to 19V. The MC signal and the TC signal are input to the matching tuning circuit 50.

FIG. 3 is a circuit configuration diagram showing an example of the matching tuning circuit 50 according to the embodiment. As shown in FIG. 3, the MC signal induces a reverse voltage in the varicap diodes 51 and 52. That is, the capacitance values of the varicap diodes 51 and 52 connected in parallel to the resonator 41 change according to the voltage value of the MC signal, and the impedance of the resonator 40 changes. Specifically, impedance matching with a component connected to the resonance portion 40, a coaxial cable, or the like is performed according to the voltage value of the MC signal. Further, the TC signal induces a reverse voltage in the varicap diode 53. That is, the capacitance value of the varicap diode 53 changes according to the voltage value of the TC signal, and the resonance frequency of the resonator 41 changes. Specifically, the resonance frequency of the resonator 41 and the high frequency are tuned according to the voltage value of the TC signal.

A CR filter is formed by the resistor 54a and the capacitor 55a connected to the port to which the MC signal is input, and the resistor 54b and the capacitor 55b connected to the port to which the TC signal is input. Each CR filter has a cutoff frequency of, for example, 1.6 kHz, smoothes the MC signal and the TC signal, and narrows the control band. Further, each CR filter has a role of filtering the voltage noise so that the voltage noise is not applied to each varicap diode when the voltage noise is induced in the wiring by the magnetic field modulation. Further, the inductors 56a to 56c block high frequencies and conduct low frequencies including direct current. That is, the inductors 56a to 56c serve as a low-pass filter.

In this way, the matching tuning circuit 50 adjusts the impedance of the resonance portion 40 (for example, 50Ω) based on one of the I signal and the Q signal (I signal in the present embodiment), and the other of the I signal and the Q signal. The resonance frequency of the resonator 41 is tuned to a high frequency based on (Q signal in this embodiment). As a result, the resonator 41 is in a critical coupling state, and the reflected wave from the resonator 41 is suppressed.

Since the MC signal and the TC signal are treated independently, it is preferable that they have the same intensity. This is because when the strength of one of the MC signal and the TC signal is high, the strength of the other is low. This will be described with reference to FIG.

FIG. 4 is a diagram showing the relationship between the phase of the reflected wave and the coupling state of the resonator 41.

As shown in FIGS. 4A and 4C, when the resonator 41 is in the over-coupling state or the under-coupling state, the reflected wave from the resonator 41 is generated. The phase of the reflected wave is 0 degrees or 180 degrees depending on the coupling state, based on the phase of the incident wave. However, in general, the phase of the reflected wave detected by the ADC 33 changes depending on the length of the cable connected to the resonance portion 40 or the component connected to the resonance portion 40, so that the phase device 25 or the phase device is used. By using 103, the phase of the reflected wave can be set to a desired phase θ. That is, the phase can be compensated by the phase device 25 or the phase device 103. The intensities of the I signal and the Q signal change according to the phase θ of the reflected wave, and the phase θ is approximately 45 degrees as shown in FIG. 4 (a) or as shown in FIG. 4 (c). When the phase θ is about 225 degrees, the intensities of the I signal and the Q signal are about the same. Then, when the I signal and the Q signal are negatively fed back to the matching tuning circuit 50, the resonator 41 is in a critical coupling state as shown in FIG. 4B, and the reflected wave from the resonator 41 is released. For example, it becomes zero. Here, the reflected wave when matching and tuning are performed by the matching tuning circuit 50 will be described with reference to FIG.

FIG. 5 is a diagram showing the intensity of the reflected wave when matching and tuning are performed by the matching tuning circuit 50 according to the embodiment. The horizontal axis represents the high frequency frequency incident on the resonator 41, and the vertical axis represents the intensity of the high frequency reflected wave reflected by the resonator 41. When the voltage values of the MC signal and the TC signal were the first condition, the result shown by the solid line in FIG. 5 was obtained, and when the voltage values were the second condition, the result shown by the broken line in FIG. 5 was obtained. The first condition and the second condition are different conditions, but are conditions when the resonator 41 is in the critical coupling state. As shown in FIG. 5, in both the first condition and the second condition, the intensity of the reflected wave is reduced at a specific frequency, and the matching and tuning of the matching tuning circuit 50 based on the I signal and the Q signal is correct. You can see that it is being done.

As described above, the electron spin resonator 1 obtains I signals and Q signals for matching and tuning that control the resonator 41 from the phase and intensity information included in the high-frequency reflected wave reflected by the resonator 41. By generating and negatively feeding those signals to the resonance unit 40 (matching tuning circuit 50) separately, the resonator 41 is automatically matched and automatically tuned without using a modulation method such as FM modulation or AM modulation. Can be done. Therefore, an analog control circuit including a modulation frequency generation circuit, a phase detection circuit, a feedback signal generation circuit, and the like becomes unnecessary, and the configuration of the electron spin resonance device 1 can be simplified.

The reflected wave in the low frequency band (for example, the frequency band based on the movement of a small animal or the like) is suppressed by the matching and tuning of the matching tuning circuit 50 based on the I signal and the Q signal, but the frequency of the modulated magnetic field (for example, 105 kHz or the like). ) The reflected wave remains unsuppressed. Further, since the high frequency, clock signal and modulated magnetic field signal are generated from the oscillator 21, all of these signals are synchronized, and the demodulation of the EPR signal becomes easy.

Further, the high frequency output unit 20 has an oscillator 21 having an oscillation frequency of 4 × n (n is an integer) times a high frequency (for example, 750 MHz) (for example, 3, 6, 12 GHz), and has a second frequency division. By dividing the oscillation frequency by the device 23, a high frequency with low phase noise can be generated. Further, the high frequency output unit 20 easily generates a clock signal for the ADC 33 in which the oscillation frequency is divided by the first frequency divider 22 by m (m is an odd number) × n (for example, 7, 14, 28, etc.). it can.

Further, the analog control circuit becomes unnecessary, and the configuration of the electron spin resonance device 1 is simplified, so that the SN ratio of the EPR signal can be improved.

FIG. 6 is a diagram showing an EPR spectrum acquired by the electron spin resonance apparatus 1 and the conventional electron spin resonance apparatus according to the embodiment. The solid line shown in FIG. 6 shows the EPR spectrum acquired by the electron spin resonance apparatus 1, and the broken line shown in FIG. 6 shows the EPR spectrum acquired by the conventional electron spin resonance apparatus. These EPR spectra are first-order differential spectra measured based on the reflected waves due to EPR absorption. As shown in FIG. 6, the electron spin resonance apparatus 1 has an SN ratio improved by about 1.8 times as compared with the conventional electron spin resonance apparatus. By simplifying the configuration of the electron spin resonance device 1 in this way, the SN ratio of the EPR signal can be improved.

Although the electron spin resonance apparatus of the present invention has been described above based on the embodiment, the present invention is not limited to this embodiment. As long as the gist of the present invention is not deviated, various modifications that can be conceived by those skilled in the art are applied to the present embodiment, and other embodiments constructed by combining some components in the embodiment are also within the scope of the present invention. Included in.

For example, in the above embodiment, the reflected wave is converted into digital data and processed, but analog data may be processed as it is.

Further, for example, in the above embodiment, the matching tuning circuit 50 matches the impedance of the resonator 41 based on the I signal, and tunes the resonance frequency of the resonator 41 to a high frequency frequency based on the Q signal. Not limited to this. For example, the matching tuning circuit 50 may match the impedance of the resonator 41 based on the Q signal and tune the resonance frequency of the resonator 41 to a high frequency based on the I signal.

Further, for example, in the above embodiment, the matching tuning circuit 50 has varicap diodes 51 to 53 whose capacitance values change according to the MC signal and the TC signal, but the present invention is not limited to this. For example, the matching tuning circuit 50 may have other variable capacitors whose capacitance values change according to the MC signal and the TC signal.

The electron spin resonance apparatus according to the present invention includes imaging of free radicals in the body of a small animal or the like, food inspection (generation of radicals due to redox), evaluation of exposure radiation dose (generation of unpaired electrons in a substance by ionizing radiation). It can be used for industrial or medical purposes such as.

1 Electron spin resonance device 10 Control unit 11a to 11h DAC
12 USB-IO port 20 High frequency output unit 21 Oscillator 22 1st frequency divider 23 2nd frequency divider 24 Attenuator 25 Phaser 30 High frequency acquisition unit 31, 80 LNA
32 BPF
33 ADC
40 Resonant 41 Resonator 50 Matching tuning circuit 51-53 Barractor diode 54a, 54b Resistance 55a, 55b Capacitor 56a-56c Inductor 60 Magnetic circuit part 61 Modulation coil 61a Amplifier 62 Gradient coil 62a Gradient coil control part 63 Sweep coil 63a Control unit 70 Circulator 90 PC
100 FPGA
101, 108 DDC section 102, 106 Downsampling section 103 Phaser 104 FIRLPF
105 Adder / Subtractor 107 FIRBPF
110 Integrator 111, 113 Attenuator 112 NCO

Claims (4)

A resonator having a matching tuning circuit and a resonator,
A high-frequency output unit that has an oscillator and outputs a high frequency based on the oscillation frequency of the oscillator to the resonator.
A control unit that negatively feeds back an I (In-phase) signal and a Q (Quadrature-phase) signal having intensity information and phase information of the high-frequency reflected wave reflected by the resonator to the resonator. Prepare,
The matching tuning circuit adjusts the impedance of the resonance portion based on one of the I signal and the Q signal, and sets the resonance frequency of the resonator based on the other of the I signal and the Q signal to the high frequency. Synchronize with,
Electron spin resonance device. The electron spin resonator further comprises a phaser that adjusts the phase of the reflected wave to approximately 45 degrees or approximately 225 degrees with respect to the phase of the high frequency incident on the resonator.
The electron spin resonance apparatus according to claim 1. The electron spin resonance device further
A bandpass filter that filters the reflected wave and
Equipped with an AD converter that digitally converts the filtered reflected wave,
The high frequency output unit further
A first frequency divider that divides the oscillation frequency of the oscillator by m (m is an odd number) x n (n is an integer), and
It has a second frequency divider that divides the oscillation frequency of the oscillator by 4 × n.
As a high frequency based on the oscillation frequency of the oscillator, a high frequency signal having a frequency divided by the second frequency divider is output to the resonator.
The AD converter digitally converts the filtered reflected wave using the frequency divided by the first frequency divider as the sampling frequency.
The electron spin resonance apparatus according to claim 1 or 2. The control unit is composed of an FPGA (Field Programmable Gate Array).
The electron spin resonance apparatus according to any one of claims 1 to 3.

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