JP6569583B2 - AC motor control device - Google Patents

Description

  The present invention relates to an AC motor control device that controls energization of an AC motor.

2. Description of the Related Art Conventionally, in an AC motor control device that performs feedback control based on a phase current detection value detected by a current sensor, a technique for reducing the influence of higher-order components superimposed on the primary component of phase current is known.
For example, the control device for an AC motor disclosed in Patent Literature 1 extracts a primary component by expanding a phase current detection value by Fourier series, and performs feedback control based on the extracted primary current. At this time, for example, the Fourier coefficient is calculated by integrating the calculated values based on the values sampled at the sampling interval obtained by dividing one electrical cycle by the division number N.

Japanese Patent No. 5741966

  Patent Document 1 describes that the division number N may be changed depending on the rotation speed or the electric frequency of the AC motor in order to ensure the detection accuracy of the primary current. According to the knowledge of this prior art, by setting the division number N to the maximum within the range that can be processed according to the rotation speed, the Fourier coefficient is calculated using more sampling values, and the detection accuracy of the primary current is improved. Seems to be able to improve.

However, depending on the relationship between the degree and the division number N of the high-order components, when sampling the phase current in the division number N, if it becomes impossible to high-order components of a particular order to distinguish the first-order component of the phase current There is. For example, the primary component and the 11th and 13th components cannot be distinguished from 12 sampling values in one electrical cycle.

  Then, the high-order component that should be removed by the filter is erroneously recognized as the primary component. As a result, the detected value of the primary current calculated by Fourier series expansion becomes larger than the true value. Therefore, if the division number N is determined based on only the rotation speed as in the conventional technique, an error occurs in the amplitude of the primary current, and the detection accuracy may be reduced.

  The present invention was created in view of the above points, and an object thereof is to provide a control device for an AC motor that detects a primary current with high accuracy in sampling by a filter.

The control apparatus for an AC motor of the present invention includes an inverter (20) that converts DC power into AC power by operation of a plurality of switching elements (21-26) and supplies the AC power to the AC motor (80), and a fed back phase current. And an inverter control unit (30) for controlling the energization of the AC motor.
In the present specification, the “AC motor” includes an AC drive motor, a generator, and a motor generator. For example, the AC motor is used as a main machine of a hybrid vehicle or an electric vehicle, and generates torque for driving drive wheels. Applicable to motor generators. Further, for example, a control device that controls energization of the motor generator corresponds to the “control device for an AC motor”.

The inverter control unit includes a filter (51), a spectrum calculation unit (401, 403, 404), and a division number setting unit (53, 54).
The spectrum calculation unit detects or estimates the spectrum of the phase current flowing through the AC motor.
The filter samples the phase current at a sampling interval obtained by dividing one electrical cycle by “the number of divisions N (N is an integer of 2 or more)”, and extracts a primary component of the phase current based on the sampled phase current value. .
If the division number setting section, obtained by sampling the phase current in the division number N, the higher order components of the specific orders spectrum calculating unit is determined based on the spectrum of the phase current detected or estimated, first-order component of the phase current The division number N is set so as to avoid being indistinguishable .

Preferably, the filter of the present invention is a primary current calculation unit (51) that extracts a primary component obtained by Fourier-phase expansion of a phase current detection value as a function of an electrical angle and calculates a primary current of the phase.
The primary current calculation unit calculates a Fourier coefficient by integrating a calculated value based on a phase current detection value sampled at a sampling interval over one electrical cycle, and calculates a primary current based on the Fourier coefficient.

Thus, in the present invention, in the sampling by the filter, the division number N is set based on the spectrum of the phase current. At this time, the division number N is set so that, when the phase current is sampled by the division number N, a high-order component of a specific order cannot be distinguished from the primary component of the phase current . Thereby, it is possible to prevent a high-order component of a specific order from being erroneously recognized as a primary component, and to improve the detection accuracy of the primary current.

The schematic block diagram of the MG drive system to which the control apparatus of the alternating current motor of each embodiment is applied. The control block diagram of the inverter control part by 1st, 2nd embodiment. FIG. 3 is a control block diagram of a modulator according to the first and second embodiments. The figure explaining the division | segmentation number in a Fourier coefficient calculation process. The figure which shows a detected value when the (a) 11th-order component of a phase current and the (b) 13th-order component are sampled by the division | segmentation number 12 (sampling interval 30deg). The figure which shows the relationship between the filter characteristic of division number 12, and a phase current spectrum. The figure which shows a detected value when (a) 11th-order component of a phase current and (b) 13th-order component are sampled by the division | segmentation number 24 (sampling interval 15deg). The figure which shows the relationship between the filter characteristic of division number 24, and a phase current spectrum. The control block diagram of the electric current process part by 1st Embodiment. The figure explaining the division number setting example by 1st Embodiment. The flowchart of the division number setting process by 1st Embodiment. The control block diagram of the electric current processing part by 2nd Embodiment. The figure explaining the division number setting example by 2nd Embodiment. The flowchart of the division number setting process by 2nd Embodiment. The control block diagram of the inverter control part by 3rd Embodiment. The control block diagram of the modulator by 3rd Embodiment. The control block diagram of the electric current process part by 3rd Embodiment. The figure which shows the example of the division | segmentation number map which a pulse pattern memory | storage part memorize | stores. The control block diagram of the inverter control part by 4th Embodiment. The control block diagram of the modulator by a 4th embodiment. The control block diagram of the inverter control part by 5th Embodiment.

Hereinafter, a plurality of embodiments of an AC motor control device will be described with reference to the drawings. In the plurality of embodiments, substantially the same configuration is denoted by the same reference numeral, and description thereof is omitted. The following first to fifth embodiments are collectively referred to as “this embodiment”.
The control apparatus for an AC motor according to this embodiment is an apparatus that controls energization of an MG that is a three-phase AC motor in a system that drives a motor generator (hereinafter referred to as “MG”) that is a power source of a hybrid vehicle or an electric vehicle. . “MG” and “MG control device” in each embodiment correspond to “AC motor” and “AC motor control device” recited in the claims.

[System configuration]
First, the overall configuration of an MG drive system to which the MG control device of each embodiment is applied will be described with reference to FIG. FIG. 1 illustrates a system including one MG.
MG drive system 90 is a rechargeable secondary battery der Luba Tteri 11 system supplies the MG80 DC power into a three-phase AC power by an inverter 20. In the MG drive system 90, the MG control device 10 mainly includes an inverter 20 and an inverter control unit 30 .
The MG control device 10 may be applied to an MG drive system including a converter that boosts the voltage of the battery 11 and outputs the boosted voltage to the inverter 20. Further, the MG control apparatus 10 can be similarly applied to an MG drive system including two or more MGs.

  The MG 80 is, for example, a permanent magnet type synchronous three-phase AC motor. In the present embodiment, the MG 80 has a function as an electric motor that generates torque for driving driving wheels of a hybrid vehicle and a function as a generator that recovers energy by generating electric power transmitted from the engine and driving wheels.

A current sensor for detecting a phase current is provided in a current path connected to the two-phase winding among the three-phase windings 81, 82, and 83 of the MG 80. In the example of FIG. 1, current sensors 87 and 88 for detecting phase currents Iv and Iw are provided in current paths connected to the V-phase winding 82 and the W-phase winding 83, respectively, and the remaining U-phase current Iu is estimated based on Kirchhoff's law. In other embodiments, any two-phase current may be detected, and a three-phase current may be detected. Or you may employ | adopt the technique which estimates the other two-phase electric current based on the electric current detection value of one phase.
The electrical angle θe of the MG 80 is detected by a rotation angle sensor 85 such as a resolver.

  In the inverter 20, six switching elements 21-26 of upper and lower arms are bridge-connected. Specifically, the switching elements 21, 22, and 23 are upper-arm switching elements of the U phase, the V phase, and the W phase, respectively, and the switching elements 24, 25, and 26 are below the U phase, the V phase, and the W phase, respectively. This is an arm switching element. The switching elements 21-26 are made of, for example, IGBTs, and are connected in parallel with reflux diodes that allow a current from the low potential side to the high potential side.

Inverter 20 converts DC power into three-phase AC power by switching elements 21-26 operating in accordance with gate signals UU, UL, VU, VL, WU, WL from inverter control unit 30. Then, phase voltages Vu, Vv, and Vw corresponding to the voltage command calculated by the inverter control unit 30 are applied to the phase windings 81, 82, and 83 of the MG 80. The smoothing capacitor 15 smoothes the system voltage Vsys input to the inverter 20. The system voltage Vsys corresponds to an “inverter voltage” recited in the claims.
The voltage sensor 27 detects the system voltage Vsys.

  The inverter control unit 30 is configured by a microcomputer or the like, and includes a CPU, a ROM, an I / O (not shown), a bus line that connects these configurations, and the like. The microcomputer executes control by software processing by executing a program stored in advance by the CPU or hardware processing by a dedicated electronic circuit.

The inverter control unit 30 acquires the system voltage Vsys, the two-phase currents Iv and Iw, and the electrical angle θe detected by each sensor. Further, the inverter control unit 30 acquires the electrical angular velocity ω [deg / s] obtained by time-differentiating the electrical angle θe by the differentiator 86. Note that a differentiator 86 may be provided inside the inverter control unit 30.
The electrical angular velocity ω is converted to a rotational speed Nr [rpm] by multiplying by a proportionality constant. In this specification, “the rotational speed converted from the electrical angular velocity ω” is omitted, and is appropriately referred to as “the rotational speed ω”, and “the rotational speed ω” and “the rotational speed Nr” are used together. In particular, the “rotation speed Nr” is used in a scene where the division number upper limit value Nlim is calculated by a current processing unit described later. In other motor control descriptions, “rotational speed ω” is mainly used.

Further, the inverter control unit 30 receives the torque command Trq ^* from the host control circuit.
Based on such information, the inverter control unit 30 calculates gate signals UU, UL, VU, VL, WU, and WL for operating the inverter 20. The inverter 20 operates the switching elements 21-26 in accordance with the gate signals UU, UL, VU, VL, WU, WL, thereby converting DC power input from the battery 11 into AC power and supplying the AC power to the MG 80.

[Configuration of inverter control unit]
Hereinafter, the configuration of the inverter control unit 30 will be described for each embodiment. As a reference numeral of the inverter control unit of each embodiment, the number of the embodiment is added to the third digit following “30” for distinction. In addition, as the codes of the modulator, the spectrum calculation unit, and the current processing unit included in the inverter control unit of each embodiment, the number of the embodiment is assigned to the third digit following “60”, “40”, and “50”, respectively. To distinguish. However, when those configurations are substantially the same as the configuration of the above-described embodiment, the reference numerals of the above-described configuration are used.

(First embodiment)
The configuration of the inverter control unit and the modulator common to the first and second embodiments will be described with reference to FIGS.
As shown in FIG. 2, the inverter control unit 301 includes a dq conversion unit 31, a torque estimation unit 32, a torque feedback control unit 340, a current command calculation as a general feedback control (“FB control” in the figure). 35, a current feedback control unit 380, a modulator 601, and a gate signal generation unit 79. Moreover, the inverter control part 301 contains the spectrum calculating part 401 and the current process part 501 of 1st Embodiment, or the current process part 502 of 2nd Embodiment as a characteristic structure. Hereinafter, matters common to the current processing units 501 and 502 of the first and second embodiments will be described as “current processing unit 501”.

  First, a general feedback control configuration will be described. This embodiment is established in a configuration in which at least one phase of the current flowing through the three-phase windings 81, 82, 83 of the MG 80 is fed back. Therefore, the inverter control unit 301 includes only the current feedback control unit 380 and does not need to include the torque feedback control unit 340. Or as shown in FIG. 2, you may provide only the torque feedback control part 340 of the structure which calculates the torque estimated value Trq_est based on a feedback current.

However, since a configuration including both the torque feedback control unit 340 and the current feedback control unit 380 is generally used as a configuration that is actually applied to the hybrid vehicle MG drive system 90, the configuration is representative here. Will be described as a preferred embodiment.
In this configuration, the feedback method for calculating the voltage vector is switched according to the modulation factor calculated from the calculated amplitude Vr of the voltage vector and the system voltage Vsys. That is, there are a case where the torque feedback control unit 340 and the current feedback control unit 380 cooperate to calculate a voltage vector, and a case where the current feedback control unit 380 calculates a voltage vector independently.

The torque feedback control unit 340 includes a torque subtracter 33 and a controller 34.
The current feedback control unit 380 includes a current subtracter 36, a controller 37, a controller 38, and a voltage amplitude / phase calculation unit 39. Among these, the controller 37, the controller 38, and the voltage amplitude / phase calculating unit 39 are selectively provided according to the two feedback methods described above.

First, a configuration common to both feedback systems will be described.
The dq conversion unit 31 converts the primary currents Iv1s and Iw1s output from the current processing unit 501 into dq-axis currents Id and Iq based on the electrical angle θe and feeds them back to the current subtractor 36.
Based on the torque command Trq ^* , the current command calculation unit 35 calculates dq-axis current commands Id ^* and Iq ^* using a map or a mathematical formula so that, for example, the maximum torque per current can be obtained.
The current subtractor 36 of the current feedback control unit 380 calculates current deviations ΔId and ΔIq between the dq axis current commands Id ^* and Iq ^* and the dq axis currents Id and Iq fed back from the dq conversion unit 31.

Next, a configuration when the torque feedback control unit 340 and the current feedback control unit 380 cooperate to calculate a voltage vector will be described.
Torque estimation unit 32 calculates torque estimated value Trq_est using equation (1) based on dq-axis currents Id and Iq and the motor constant of MG80.
Trq_est = p × {Iq × ψ + (Ld−Lq) × Id × Iq} (1)
However,
p: number of pole pairs of MG ψ: counter electromotive voltage constant Ld, Lq: d-axis inductance, q-axis inductance

The torque subtractor 33 of the torque feedback control unit 340 calculates a torque deviation ΔTrq between the torque command Trq ^* and the estimated torque value Trq_est. The controller 34 calculates the voltage phase φ by PI calculation so that the torque deviation ΔTrq converges to 0, and outputs it to the modulator 601.
The controller 37 of the current feedback control unit 380 calculates the voltage amplitude Vr by PI calculation so that the current deviations ΔId and ΔIq converge to 0, and outputs the voltage amplitude Vr to the modulator 601.

Next, a configuration when the current feedback control unit 380 calculates a voltage vector independently will be described.
The controller 38 of the current feedback control unit 380 calculates the dq axis voltage commands Vd ^* and Vq ^* by PI calculation so that the current deviations ΔId and ΔIq converge to 0. The voltage amplitude / phase calculation unit 39 converts the dq axis voltage commands Vd ^* and Vq ^* into a voltage amplitude Vr and a voltage phase φ, and outputs them to the modulator 601. FIG. 2 shows an example in which the voltage phase φ is defined with reference to the d axis, but the voltage phase may be defined with reference to the q axis.

Thus, the modulator 601 receives the voltage amplitude Vr and the voltage phase φ calculated by any feedback method. Further, the modulator 60 1, system voltage Vsys, the electrical angle .theta.e, information such as rotational speed ω is input.
Based on this information, the modulator 601 outputs a pulse pattern or a PWM signal to the gate signal generation unit 79 as an output waveform of a pulse voltage for operating the inverter 20.
The gate signal generation unit 79 generates gate signals UU, UL, VU, VL, WU, WL based on the pulse pattern or PWM signal output from the modulator 601 and outputs the generated signals to the switching elements 21-26 of the inverter 20. .

As shown in FIG. 3, the modulator 601 includes a modulation rate calculation unit 61, a method switching unit 62, a synchronization number setting unit 63, a pulse pattern selection unit 64, a pulse pattern storage unit 65, a PWM signal generation unit 66, and a carrier frequency setting. Part 67.
Among them, the synchronization number setting unit 63, the pulse pattern selection unit 64, and the pulse pattern storage unit 65 are blocks that generate pulse patterns, and the PWM signal generation unit 66 and the carrier frequency setting unit 67 are blocks that generate PWM signals. It is. In the first embodiment, the modulator 601 may include at least one block. In this specification, it is assumed that the pulse pattern includes a pattern that outputs a rectangular wave of one pulse in one electrical cycle.

The modulation factor calculation unit 61 calculates the modulation factor m from the ratio between the voltage amplitude Vr output from the current feedback control unit 380 and the system voltage Vsys by Expression (2).
m = 2√ (2/3) × (Vr / Vsys) (2)
The method switching unit 62 switches the voltage waveform specifying method based on the modulation factor m and the like.
The synchronization number setting unit 63, the pulse pattern selection unit 64, and the pulse pattern storage unit 65 will be described in a third embodiment to be described later.

The PWM signal generation unit 66 compares the phase voltage calculated based on the output of the current feedback control unit 380 with a carrier wave as a method for specifying the voltage waveform output from the inverter 20 and generates a PWM signal. The voltage waveform output from the inverter 20 is specified.
Specifically, the PWM signal is generated by comparing the duty in which the phase voltage is converted with a carrier wave such as a triangular wave.
The carrier frequency setting unit 67 sets a carrier frequency (hereinafter referred to as “carrier frequency”) Fc used by the PWM signal generation unit 66.

Returning to FIG. 2, the spectrum calculation unit 401 and the current processing unit 501 which are characteristic configurations of the present embodiment will be described next.
The spectrum calculation unit of each embodiment detects or estimates a spectrum of one or more phase currents flowing through the MG 80. In particular, the spectrum calculation unit 401 of the first embodiment acquires the phase current detection values Iv and Iw detected by the current sensors 87 and 88, and detects the spectrum of the phase current.

In addition to the phase current detection values Iv and Iw detected by the current sensors 87 and 88, the electrical angle θe and the rotation speed Nr are input to the current processing unit 501.
By the way, as described in Patent Document 1 (Japanese Patent No. 5741966) which is a prior art, in the MG control device, a high-order component is superimposed on a phase current to be fed back or the phase current is offset. There is. Therefore, feedback control is performed using the primary current calculation value extracted by Fourier series expansion of the phase current detection value, thereby reducing noise due to high-order components and suppressing torque fluctuation and power fluctuation due to phase current offset. It becomes.

  The current processing unit 501 of the present embodiment is obtained by adding other functional units based on the basic configuration of the “primary current calculation unit” of the prior art. The current processing unit 501 acquires the information on the phase current spectrum detected by the spectrum calculation unit 401 as a feature that the conventional technology does not have, and based on the information, the “division number N (N is 2 or more) Integer) ”. Then, the current processing unit 501 outputs the primary currents Iv1s and Iw1s calculated based on the acquired information to the dq conversion unit 31.

The overall configuration of the current processing unit 501 will be described later with reference to FIG. Before that, the “primary current calculation unit 51” of the current processing unit 501 will be described here.
The primary current calculation unit 51 extracts a primary component obtained by expanding the Fourier series of the phase current detection value as a function of the electrical angle, and calculates the primary current of the phase, as in the prior art. It functions as a “filter” described in the range. The primary current calculation unit 51 calculates a Fourier coefficient by accumulating a calculated value based on a phase current detection value sampled at a sampling interval obtained by dividing one electrical period by a division number N over one electrical period, and the Fourier coefficient Based on this, the primary current is calculated.

  In Patent Document 1, “sampling interval obtained by dividing one period of electricity by the number of divisions N” is expressed as “interval of integrated angles”. The concept of the integrated angle, the calculation formula of the Fourier coefficient, the calculation formula of the primary current using the Fourier coefficient, and the like are based on the disclosure of Patent Document 1, and detailed description thereof is omitted in this specification. Further, the “electrical angle k cycle” described in Patent Document 1 is considered as “k = 1” in the present specification.

  In the second embodiment of the prior art, the interval between the integrated angles is set to a constant interval obtained by equally dividing one electrical cycle by the division number N. For example, as shown in FIG. 4, when the number of divisions N is 12, the interval between the integrated angles is 30 degrees. In this case, the Fourier coefficient is calculated based on the data of 12 electrical angles θe and V-phase current Iv per electrical cycle.

Further, paragraph [0084] of Patent Document 1 describes as follows.
“When the number of divisions N is fixed, the period of integration timing varies according to the number of rotations Nr. Therefore, the number of divisions N depends on the number of rotations Nr of the AC motor or the electrical frequency so that the accuracy of integration can be appropriately secured. Specifically, the division number N may be decreased as the rotational speed Nr or the electrical frequency is increased, and the division number N may be increased as the rotational speed Nr or the electrical frequency is decreased.
According to the knowledge of this prior art, by setting the division number N to the maximum within the range that can be processed according to the rotation speed, the Fourier coefficient is calculated using more sampling values, and the detection accuracy of the primary current is improved. Seems to be able to improve.

However, depending on the relationship between the degree and the division number N of the high-order component, it may not be possible to calculate the primary current correctly. This will be described with reference to FIGS.

FIGS. 5A and 5B show detection values when the 11th-order component and the 13th-order component of the phase current are sampled at the division number 12 (that is, the sampling interval 30 deg). The locus of the 12 points sampled draws a first order sine wave . For this reason, from 12 sampled values into electric one cycle, the first-order component and the 11th, it is impossible to distinguish between 13-order component.

FIG. 6 is a diagram showing this phenomenon by the relationship between the phase current spectrum and the filter characteristics.
The filter characteristic indicates the “attenuation rate of the amplitude after the filter processing relative to the amplitude before the filter processing” of each order component in the case of the division number 12. For example, the attenuation rate of −20 [dB] means that the amplitude before the filter is attenuated to 1/10 after the filter. An attenuation factor of 0 [dB] means that the amplitude before the filter is not attenuated after the filter.
FIG. 6 shows characteristic lines connecting the attenuation rate points calculated every 0.25th order from the 0th order to the 30th order. Excluding the first order, the attenuation rate is 0 [dB] in the 11th, 13th, 23rd, and 25th orders. On the other hand, the absolute value of the attenuation factor is large in the section between the first order and the eleventh order, between the thirteenth order and the twenty-third order, and from the 25th order to the 30th order.

The phase current spectrum is a virtual spectrum shown for explanation, and actually corresponds to a frequency spectrum detected by means such as fast Fourier transform (FFT).
In FIG. 6, it is assumed that the spectrum amplitudes of the 11th and 13th components included in the phase current detection value are relatively large. The order components having relatively small amplitudes other than the first, eleventh, and thirteenth orders merely indicate that “other order components” exist, and the order values themselves have no meaning. .

As described above, when the attenuation factor of the filter characteristic is 0 [dB] for the 11th and 13th components having a relatively large spectrum amplitude, the 11th and 13th components are erroneously recognized as the primary components. The As a result, the amplitude of the erroneously recognized high-order component is added to the amplitude of the primary current. Therefore, the amplitude A _{sns of the} primary current detected by the filter has an error on the plus side with _respect to the true amplitude A _real, so that the detection accuracy of the primary current is lowered.

Next, FIGS. 7A and 7B show current values when the 11th-order component and the 13th-order component are sampled at the division number 24 (that is, sampling interval 15 deg), respectively. In this case, the locus of the 24 sampled points follows the current waveform before the filter to some extent and can be distinguished from the current waveform of the primary component.
As shown in FIG. 8, according to the filter characteristics in the case of the division number of 24, the attenuation factors of the 11th and 13th components are on the negative side from −30 [dB], and the absolute value exceeds 30. This means that the 11th order component and the 13th order component are appropriately removed when the phase current detection value is sampled by the division number 24. As a result, the amplitude A _{sns of the} primary current detected by the filter matches the true amplitude A _real with high accuracy.

In short, when setting the division number N of the Fourier series expansion, it is not only necessary to make a determination based on the rotational speed Nr as in the prior art, but the relationship between the amplitude of the phase current spectrum and the attenuation factor of the order component of interest. It is important to consider.
Therefore, this embodiment is characterized by the configuration and operation of the current processing unit for appropriately setting the division number N used by the primary current calculation unit 51 for Fourier series expansion.

A configuration of the current processing unit 501 of the first embodiment and an example of setting the number of divisions by the current processing unit 501 will be described with reference to FIGS. 9, 10, and 11.
In the setting example shown in FIG. 10, the division number “N = 24” set for the first time is inappropriate, so the division number is changed to “N = 18” for the second time. Thus, in the first embodiment, the division number N is set based on the idea of trial and error that “set and change if inappropriate”.

The flowchart of FIG. 11 shows a division number setting process routine according to the first embodiment. In the description of the flowchart below, the symbol “S” means a step.
“S2-1”, “S7-2”, etc. in FIG. 10 correspond to the steps with the same numbers in FIG. For example, “S2-1” means “S2” of the first loop, and “S7-2” means “S7” of the second loop.

As shown in FIG. 9, the current processing unit 501 includes a primary current calculation unit 51, a spectrum comparison unit 52, a division number setting unit 53, an attenuation rate acquisition unit 55, a non-attenuation component specifying unit 56, and a division number limiting unit 57. Have.
The general description of the primary current calculation unit 51 is as described above. As shown in FIG. 9, the primary current calculation unit 51 receives, for example, information on the detected values Iv and Iw of the V-phase and W-phase currents and the electrical angle θe, and develops a Fourier series based on these information 1 Extract the next component. In this example, V-phase and W-phase primary currents Iv1s and Iw1s are calculated and output to the dq converter 31 as feedback currents.

The spectrum comparison unit 52 compares the amplitude of the phase current spectrum detected by the spectrum calculation unit 401 with the determination threshold Ath. In the first embodiment, the spectrum amplitude for the non-attenuating component specified by the non-attenuating component specifying unit 56 is compared with the determination threshold Ath.
The division number setting unit 53 first sets a certain division number N. For example, the division number N may be calculated from only the rotation speed Nr as in the prior art, or the value used in the previous control may be used. From the viewpoint of improving the detection accuracy, in principle, the division number N is preferably set as large as possible within a processable range according to the rotation speed Nr.
Thereafter, the division number setting unit 53 changes the division number N when it is determined that the set division number N is inappropriate based on the comparison result of the spectrum amplitude by the spectrum comparison unit 52.

The division number limiting unit 57 calculates the upper limit value Nlim of the division number based on the processing time Tc of the current processing unit 501 and the rotation speed Nr of the MG 80, and sets the division number N that can be set by the division number setting unit 53 as the upper limit value Nlim. Restrict to: Here, symbols are defined as follows.
Tc: Processing time [s]
T: Time of one electrical cycle [s]
p: Number of pole pairs of MG [−]
Nr: number of revolutions [rpm]
θs: Sampling interval [deg] (= 360 / N)
θslim: Lower limit of sampling interval [deg]

Depending on the rotation speed Nr of the MG 80, the time T of one electrical cycle is calculated by the equation (3).
T = 60 / (p × Nr) (3)
Since the division number N needs to be equal to or less than the number of times that the processing of the processing time Tc can be repeated within the time T of one electrical cycle, Equation (4) is obtained.
N ≦ T / Tc = 60 / (p × Nr × Tc) = Nlim (4)
Equation (4) can be rewritten as Equation (5) for the lower limit value θslim of the sampling interval.
θs ≧ (360/60) × p × Nr × Tc = 6 (p × Nr × Tc) = θslim
... (5)

The attenuation rate acquisition unit 55 acquires the attenuation rate of the amplitude after the filter processing with respect to the amplitude before the filter processing, for one or more high-order components sampled at the division number N set by the division number setting unit 53. .
As illustrated in FIGS. 5 to 8, the attenuation rate of each order component is theoretically determined by the relationship between the order of the higher order component and the division number N. Therefore, the attenuation rate acquisition unit 55 may store, for example, a map that defines the relationship between the division number N and the attenuation rate for each order of higher-order components.

  The non-attenuating component specifying unit 56 specifies a high-order component of the order whose absolute value of the attenuation rate acquired by the attenuation rate acquiring unit is less than the threshold value Rth as the “non-attenuating component”. Hereinafter, the magnitude comparison with the threshold value Rth is expressed by the absolute value of the attenuation rate, and the description of “absolute value” is omitted as appropriate. Therefore, “the attenuation factor is less than the threshold value Rth” means that the attenuation factor is closer to 0 than the negative threshold value Rth, that is, hardly attenuates.

Next, a procedure for setting the division number N will be described along the flowchart of FIG. 11 with reference to FIGS.
First, the overall flow of the routine will be described, and then the specific example shown in FIG. 10 will be compared.
In S1, the division number limiting unit 57 acquires the processing time Tc and the rotation number Nr, and calculates the division number upper limit value Nlim based on the information. It is assumed that the processing time Tc and the rotational speed Nr do not change during one routine.

In S2, the division number setting unit 53 sets the division number N. Note that the notes in the broken line frame will be described later.
In S3, it is determined whether the set division number N is equal to or less than the upper limit value Nlim. If NO in S3, the process returns to S2, and the division number setting unit 53 resets the division number N. If YES in S3, the process proceeds to S4.

In S4 and S5, the spectrum calculation unit 401 acquires a current of one phase or more and detects a spectrum.
In S6, the attenuation rate acquisition unit 55 and the non-attenuation component specifying unit 56 acquire the attenuation rate according to the division number N, and specify the “non-attenuation component” whose attenuation rate is less than the threshold value Rth.

In S7, the spectrum comparison unit 52 determines whether the spectrum amplitude of the non-attenuating component is less than the determination threshold Ath. If NO in S7, the process returns to S2, and the division number setting unit 53 resets the division number N. If YES in S7, the process proceeds to S10.
In S10, the primary current calculation unit 51 calculates a Fourier coefficient using the determined division number N, and calculates a primary current by Fourier series expansion.

Next, a specific example will be described with reference to FIG. In S3 of this example, YES is determined for both the first time and the second time. In S7, it is determined NO in the first S7-1, and then YES is determined in the second S7-2.
In the first S2-1, the division number setting unit 53 sets the division number N to 24 (that is, the sampling interval 15 deg). In S6-1, the 23rd-order component and the 25th-order component are specified as non-attenuating components by the filter characteristic of “N = 24”.

In S7-1, the spectrum comparison unit 52 compares the spectrum amplitude with the determination threshold Ath for the 23rd-order component and the 25th-order component. Then, since the spectrum amplitudes of the 23rd-order component and the 25th-order component are equal to or greater than the determination threshold Ath, “NO” is determined in S7-1. Therefore, the non-adoption of the division number 24 is determined.
Next, in the second S2-2, the division number setting unit 53 changes the division number N from 24 to 18 (that is, the sampling interval 20 deg) and resets it. In S6-2, the 17th-order component and the 19th-order component are specified as non-attenuating components by the filter characteristic of “N = 18”.

In S7-2, the spectrum comparison unit 52 compares the spectrum amplitude with the determination threshold value Ath for the 17th-order component and the 19th-order component. Then, since the spectrum amplitudes of the 17th order component and the 19th order component are less than the determination threshold Ath, “YES” is determined in S7-2. Thus, the division number 18 is adopted.
As shown in this example, the larger the division number N, the better. For example, the detection accuracy may be improved by reducing the division number N from 24 to 18.

Referring to FIG. 11 again, the notes described in the dashed frame in S2 will be described. This note defines the reset condition for the division number N when NO is determined in S7 and the process returns to S2.
When the division number setting unit 53 changes the division number N from a relatively large value to a small value, the division number N (x) after the change is changed to “other than a divisor, the division number N (x−1) before the change. Value.

For example, as shown in the example of FIG. 10, when the division number N is set to 24 in the first loop, the spectrum amplitudes of the 23rd and 25th components that are non-attenuating components are equal to or greater than the determination threshold Ath. Suppose the case of NO is determined.
In this case, as shown in FIG. 6, in the filter characteristics of the division number 12 which is a divisor of 24, the 23rd order component and the 25th order component are also non-attenuating components and are not removed. Therefore, even if the division number N is set to 12 in the second loop, it does not meet the positive condition of S7. This is the same when the other 24 divisors are changed to 8, 6, 4, 3, 2.
Therefore, when the division number N (x) after the change is set, the division number N (x) after the change is efficiently removed by excluding the divisors of the division number N (x−1) before the change in advance. Can be reset.

(effect)
The effect of the first embodiment will be described.
(1) First Embodiment In the following embodiments, in the phase current sampling by the primary current calculation unit 51, the division number N is set based on the spectrum of the phase current. At this time, the division number N is set so that, when the phase current is sampled by the division number N, a high-order component of a specific order cannot be distinguished from the primary component of the phase current . Thereby, it is possible to prevent a high-order component of a specific order from being erroneously recognized as a primary component, and to improve the detection accuracy of the primary current.

(2) The non-attenuating component specifying unit 56 specifies the higher-order component of the order whose absolute value of the attenuation rate acquired by the attenuation rate acquiring unit 55 is less than the threshold value Rth as the “non-attenuating component”.
In addition, the division number setting unit 53 of the first embodiment changes the division number N when the spectrum amplitude of the phase current is equal to or greater than the determination threshold Ath for the non-attenuating component specified according to the set division number N. Reset it.
In this way, the spectral amplitude in the entire frequency range is not evaluated loosely, but by evaluating the amplitude intensively with respect to the non-attenuating component having a large influence as a filter characteristic, the division number N is appropriately set, and the primary current Detection accuracy can be improved.

(3) When the division number setting unit 53 changes the division number N from a relatively large value to a small value, the division number N (x) after the change is approximately equal to the division number N (x−1) before the change. Set to a value other than a number. Thereby, the changed division number N (x) can be efficiently reset.
(4) The division number limiting unit 57 calculates the upper limit value Nlim of the division number based on the processing time Tc of the current processing unit 501 and the rotation number Nr of the MG 80, and sets the division number N that can be set by the division number setting unit 53. It is limited to the upper limit value Nlim or less. Thereby, the failure of control can be prevented.

(Second Embodiment)
A second embodiment will be described with reference to FIGS.
In the second embodiment, the configurations of the inverter control unit 301 and the modulator 601 are the same as the configurations of the first embodiment shown in FIGS. 2 and 3. Further, FIGS. 12, 13, and 14 of the second embodiment correspond to FIGS. 9, 10, and 11 of the first embodiment, respectively.

The phase current spectrum and filter characteristics shown in FIG. 13 are the same as those shown in FIG. In FIG. 14, the same step numbers are assigned to substantially the same steps as in FIG. 11, and a part of the description is omitted. “S8” and “S9” in FIG. 13 correspond to steps with the same numbers in FIG.
In the second embodiment, the division number setting unit 54 selects the number of divisions according to the order of higher-order components in which the phase current spectrum amplitude is equal to or greater than the determination threshold from among a plurality of division number N candidates stored in advance. “Select” N.

As illustrated in FIG. 12, the current processing unit 502 according to the second embodiment does not include the non-attenuating component specifying unit 56 with respect to the current processing unit 501 according to the first embodiment. In addition, the attenuation rate acquisition unit 55 notifies the division number setting unit 54 of the acquired attenuation rate.
Hereinafter, FIGS. 12 to 14 will be referred to together.
In S <b> 1 of FIG. 14, the division number limiting unit 57 calculates the division number upper limit value Nlim. In S4 and S5, the spectrum calculation unit 401 acquires a current of one phase or more and detects a spectrum.

  In S8, the spectrum comparison unit 52 compares the spectrum amplitude of each order component with the determination threshold Ath over the entire frequency range, and identifies a “high amplitude component” for which the spectrum amplitude is greater than or equal to the determination threshold Ath. In the example shown in FIG. 13, the 23rd-order component and the 25th-order component are specified as high-amplitude components. The spectrum comparison unit 52 notifies this identification result to the division number setting unit 54.

The division number setting unit 54 stores a plurality of candidates for the division number N in advance. Among them, the attenuation factor of the high amplitude component is read from the attenuation factor acquisition unit 55 with respect to the filter characteristics of the division number N that is equal to or less than the upper limit value Nlim. In the example of FIG. 13, the attenuation rates of the 23rd order component and the 25th order component are evaluated.
Then, when the division number N is 24, the attenuation rate of the 23rd-order component and the 25th-order component is less than the threshold value Rth and is not removed by the filter. On the other hand, when the division number N is 18, the attenuation factors of the 23rd-order component and the 25th-order component are equal to or greater than the threshold value Rth and are removed by a filter. For the other filter characteristics of the division number N, the attenuation factors of the 23rd-order component and the 25th-order component are similarly evaluated.

As a result, in S9, the division number setting unit 54 selects “N = 18” as the division number N that is “the upper limit value Nlim or less and the attenuation factor of the high amplitude component is the threshold value Rth or more”.
The primary current calculation unit 51 uses the selected number of divisions 18 to expand the phase currents Iv and Iw by Fourier series and extracts the primary current.
The second embodiment has the effects (1) and (4) of the first embodiment in common. In the second embodiment, since the division number N is selected in a single sequence without repeating a plurality of processing loops, variations in processing time can be reduced.

(Third embodiment)
A third embodiment will be described with reference to FIGS.
As shown in FIG. 15, in the inverter control unit 303 of the third embodiment, the spectrum calculation unit 403 estimates the phase current spectrum based on the voltage waveform of the pulse pattern acquired from the modulator 603. That is, the third embodiment is different from the first embodiment in the spectrum calculation configuration by the spectrum calculation unit 403. Note that the configuration of the current processing unit of the third embodiment is common to the current processing units 501 and 502 of the first or second embodiment.

  Since the modulator 603 of the third embodiment is characterized by the action of a block that generates a pulse pattern, the letter “pulse pattern” is circled in FIG. In FIG. 16, a block for generating a PWM signal is indicated by a two-dot chain line. As shown in FIG. 16, the voltage waveform selected by the pulse pattern selection unit 64 of the modulator 603 is output to the spectrum calculation unit 403.

Here, the synchronization number setting unit 63, the pulse pattern selection unit 64, and the pulse pattern storage unit 65 that are omitted in the description of the first embodiment will be described.
The synchronization number setting unit 63 sets the synchronization number s, that is, the maximum number of pulses in one electrical cycle, based on the rotation number ω.
The pulse pattern selection unit 64 uses a plurality of pulse patterns stored in the pulse pattern storage unit 65 based on the modulation rate m calculated by the modulation rate calculation unit 61 and the synchronization number s set by the synchronization number setting unit 63. Select one of the pulse patterns.

The pulse pattern storage unit 65 stores a plurality of pulse patterns in advance as voltage waveforms that can be output by the inverter 20. Note that the pulse pattern storage unit 65 may store a voltage waveform generated by carrier wave comparison by PWM control as a pulse pattern.
In FIG. 16, the input / output relationship between the pulse pattern selection unit 64 and the pulse pattern storage unit 65 is represented by a bidirectional arrow.

Each pulse pattern defines the number of synchronizations in one electrical cycle, the position and width of each pulse, and the amplitude of each order higher order component included in the phase current is determined according to these values. Therefore, the spectrum calculation unit 403 can estimate the spectrum of the phase current based on the voltage waveform of the pulse pattern selected by the pulse pattern selection unit 64.
As shown in FIG. 17, even if the division number setting unit 53 of the current processing unit 501 similar to that of the first embodiment sets the division number N while changing it based on the phase current spectrum estimated by the spectrum calculation unit 403. Good. Alternatively, the division number setting unit 54 of the current processing unit 502 similar to the second embodiment may select the division number N instead.

Furthermore, a modification of the third embodiment will be described.
In this modification, the pulse pattern storage unit 65 stores the division number N used by the primary current calculation unit 51 for sampling the phase current for each pulse pattern.
For example, as illustrated in FIG. 18, the pulse pattern storage unit 65 stores a map in which the synchronization number s and the modulation rate m are associated with the voltage waveform of the pulse pattern and the division number N. The division number “N _ * #” in FIG. 18 indicates the division number N corresponding to the pulse pattern selected when the modulation factor m is the value in the * th row and the synchronization number s is the value in the #th column. means.

In this map, the relationship between the pulse pattern and the division number N may be stored as a default at the time of manufacture. Alternatively, the pulse pattern actually used and the division number N set at that time may be associated and learned each time.
The correspondence relationship between the synchronization number s, the modulation rate m, and the division number N is not limited to one-to-one. For example, the first division number N1 may correspond to the modulation rate m in a certain range, and the second division number N2 may correspond to the modulation rate m in other ranges.

Further, as shown by broken lines in FIGS. 15, 16, and 17, in this modification, when the pulse pattern selection unit 64 selects a pulse pattern, the division number N stored in the pulse pattern storage unit 65 is the current. This is notified to the division number setting unit 53 of the processing unit 501. Therefore, it is considered that the spectrum calculation unit 403 does not always need to estimate the phase current spectrum.
However, at least in the stage of storing the relationship between the pulse pattern and the division number N, the division number N is set based on the technical idea that “the amplitude of the phase current spectrum of the non-attenuating component is less than the determination threshold”. Therefore, the configuration of this modified example also corresponds to the “configuration including a spectrum calculation unit” recited in the claims.

(Fourth embodiment)
A fourth embodiment will be described with reference to FIGS. 19 and 20.
As shown in FIG. 19, in the inverter control unit 304 of the fourth embodiment, the spectrum calculation unit 404 estimates the phase current spectrum based on the carrier frequency Fc and the modulation factor m acquired from the modulator 604. That is, the fourth embodiment differs from the first and third embodiments in the spectrum calculation configuration by the spectrum calculation unit 404. Note that the configuration of the current processing unit of the fourth embodiment is the same as that of the current processing units 501 and 502 of the first or second embodiment.

  Since the modulator 604 of the fourth embodiment is characterized by the action of a block that generates a PWM signal, the letter “PWM signal” is circled in FIG. 19. In FIG. 20, a block for generating a pulse pattern is indicated by a two-dot chain line. As shown in FIG. 20, the carrier frequency Fc set by the carrier frequency setting unit 67 of the modulator 604 and the modulation rate m calculated by the modulation rate calculation unit 61 are output to the spectrum calculation unit 404.

  In PWM control, the number of synchronizations of the voltage waveform and the position and width of each pulse are defined according to the carrier frequency Fc and the modulation factor m, so that the amplitude of the higher-order component of each order included in the phase current is determined. Therefore, the spectrum calculation unit 404 can estimate the spectrum of the phase current based on the carrier frequency Fc and the modulation factor m.

(Fifth embodiment)
A fifth embodiment will be described with reference to FIG. The inverter control unit 305 of the fifth embodiment includes a current processing unit 59 in which a current processing unit 501 and a dq conversion unit 31 are integrated with respect to the inverter control unit 301 of the first embodiment shown in FIG. The current processing unit 59 extracts the primary component from the phase current detection values Iv and Iw, and executes the calculation up to the dq conversion as a series of processes. The operational effects of the fifth embodiment are the same as those of the first embodiment.
Similarly, in other embodiments, the current processing unit and the dq conversion unit may be integrated.

(Other embodiments)
(A) In the above embodiment, a “primary current calculation unit” that extracts a primary component by Fourier series expansion is used as the “filter” described in the claims. However, the present invention is not limited to this, and may be applied to filters other than the Fourier series expansion.
(B) The spectrum calculation method by the spectrum calculation unit is not limited to the method described in the above embodiment, and the phase current spectrum may be detected or estimated by any method.

(C) In the above embodiment, the spectrum calculation unit 401 and the like are included in the inverter control unit 30 and the like. However, it only means that there is a functional relevance and is not limited to a form physically provided on the same substrate, for example.
(D) The number of phases of the AC motor driven in the system to which the present invention is applied is not limited to three phases, and may be any number of phases. Further, the AC motor is not limited to a permanent magnet type synchronous motor, and may be an induction motor or other synchronous motor.

(E) The control apparatus for an AC motor according to the present invention is not limited to an MG drive system for a hybrid vehicle or an electric vehicle, and may be applied to an AC motor drive system for any application, such as for a general machine.
As mentioned above, this invention is not limited to the said embodiment at all, In the range which does not deviate from the meaning of invention, it can implement with a various form.

10 ... MG control device (control device for AC motor),
20 ... Inverter, 21-26 ... Switching element,
301, 303, 304, 305... Inverter control unit,
401, 403, 404 ... spectrum calculation unit,
51... Primary current calculation unit (filter),
53, 54... Division number setting unit,
70: Spectrum calculation unit,
80: MG (AC motor).

Claims (12)

An inverter (20) that converts DC power into AC power by operation of the plurality of switching elements (21-26) and supplies the AC power to the AC motor (80);
An inverter control unit (30) for operating the inverter based on the fed back phase current and controlling the energization of the AC motor;
With
The inverter control unit
A spectrum calculator (401, 403, 404) for detecting or estimating a spectrum of a phase current flowing in the AC motor;
A filter (51) for sampling a phase current at a sampling interval obtained by dividing one electrical cycle by a division number N (N is an integer of 2 or more), and extracting a primary component of the phase current based on the sampled phase current value; ,
When sampling the phase currents in the division number, high order components of a particular order that the spectrum calculating unit is determined based on the spectrum of the phase current detected or estimated, distinguishing the primary component of the phase current A division number setting unit (53, 54) for setting the number of divisions , so as to avoid being unable to
A control apparatus for an AC electric motor. The filter is a primary current calculation unit (51) that extracts a primary component obtained by expanding a Fourier series of a phase current detection value as a function of an electrical angle and calculates a primary current of the phase.
The primary current calculator is
2. The AC motor according to claim 1, wherein a Fourier coefficient is calculated by integrating calculated values based on phase current detection values sampled at the sampling interval over one electrical cycle, and a primary current is calculated based on the Fourier coefficient. Control device. The inverter control unit
An attenuation rate acquisition unit (55) for acquiring an attenuation rate of the amplitude after the filter processing with respect to the amplitude before the filter processing for one or more high-order components sampled at the division number set by the division number setting unit The control device for an AC motor according to claim 1, further comprising: The inverter control unit
A non-attenuating component specifying unit (56) for specifying, as a non-attenuating component, a higher-order component of the order whose absolute value of the attenuation rate acquired by the attenuation rate acquiring unit is less than a threshold (Rth);
The division number setting unit (53)
4. The AC motor according to claim 3, wherein for the non-attenuating component specified according to the set number of divisions, the division number is changed and reset when a spectrum amplitude of a phase current is equal to or greater than a determination threshold (Ath). 5. Control device. When the division number setting unit changes the division number from a relatively large value to a small value,
The control apparatus for an AC motor according to claim 4, wherein the division number (N (x)) after the change is set to a value other than a divisor of the division number (N (x-1)) before the change. The division number setting unit (54)
The division number is selected from among a plurality of division number candidates stored in advance according to the order of the high-amplitude component, which is a high-order component whose phase current spectral amplitude is equal to or greater than the determination threshold (Ath). The control apparatus of the alternating current motor as described in any one of Claims 1-3. The inverter control unit
A pulse pattern storage unit (65) that stores a plurality of pulse patterns in advance as voltage waveforms that can be output by the inverter;
Based on the number of rotations of the AC motor and the modulation factor calculated from the ratio between the voltage amplitude (Vr) and the inverter voltage, any one of the pulse patterns stored in the pulse pattern storage unit is selected. A pulse pattern selection unit (64) to select;
The control device for an AC motor according to any one of claims 1 to 6, further comprising: The pulse pattern storage unit
The control apparatus for an AC motor according to claim 7, wherein the division number used by the filter for sampling of phase current is stored for each pulse pattern. The spectrum calculation unit (401)
The AC motor according to any one of claims 1 to 8, wherein one or more phase current detection values detected by one or more current sensors (87, 88) are acquired and a spectrum of the phase current is detected. Control device. The spectrum calculation unit (403)
The control apparatus for an AC motor according to claim 7 or 8, wherein a spectrum of a phase current is estimated based on a voltage waveform of a pulse pattern selected by the pulse pattern selection unit. The inverter control unit
A PWM signal generator (66) for comparing the phase voltage calculated by the current feedback control and the carrier wave to generate a PWM signal and specifying the voltage waveform output by the inverter;
The spectrum calculation unit (404)
7. The phase current spectrum is estimated based on a carrier frequency used for generating the PWM signal and a modulation rate calculated from a ratio between a voltage amplitude (Vr) and an inverter voltage. The control apparatus for an AC motor described in the paragraph. The inverter control unit
The upper limit value (Nlim) of the division number is calculated based on the processing time (Tc) of the control device and the rotation speed (Nr) of the AC motor, and the division number that can be set by the division number setting unit is set to the upper limit. The control device for an AC electric motor according to any one of claims 1 to 11, further comprising a division number limiting unit (57) for limiting to a value less than or equal to the value.

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