JP6395672B2 - Radar equipment - Google Patents

Description

  This invention reproduces a radio wave image with high resolution for a target by aperture synthesis of a radar reception signal that repeatedly transmits and receives radio waves while changing the relative positional relationship between the target to be observed and the radar. The present invention relates to a radar device.

Image radars such as SAR (Synthetic Aperture Radar), which is a synthetic aperture radar, and ISAR (Inverse SAR), which is a reverse synthetic aperture radar, observe targets at different times while changing the relative positions of the target and the radar. The received signal obtained by observation is synthesized (aperture synthesis) in consideration of the radar direction (expected angle) in the coordinate system fixed to the target, thereby achieving high resolution of the image.
In the SAR, observation of a stationary target (for example, a ground surface, a ground structure, a stationary vehicle, etc.) is mainly assumed, and the change of the prospective angle is obtained by changing the position of the radar.
On the other hand, ISAR mainly assumes observation of a moving target (for example, an aircraft, a vehicle, a ship, etc.), and the radar itself is not necessarily used by utilizing the movement of the target (for example, movement of position, rotation, shaking, etc.). Even if does not change the position, a change in the angle of view is obtained.

  In each observation regarding a certain reflection point on the target, the received signal at the relative time relative to the transmission time of the radio wave is reflected from the propagation path determined by the relative positional relationship between the radar and the target at the time of the observation, that is, from the transmission antenna. A time delay corresponding to the time required for the radio wave to travel through the propagation path to the receiving antenna via the point is generated, and the transmission signal is given as an amplitude-multiplied transmission signal according to the shape and material of the reflection point. If there are multiple reflection points on the target, they are given as a superposition of them. Hereinafter, the elapsed time from the radio wave transmission time is referred to as fast time. In addition, a time delay caused by a radio wave moving along the propagation path is referred to as a propagation delay time. The distribution (profile) of the received signal (or a signal obtained by increasing the resolution of the received signal by pulse compression or the like as necessary) with respect to the fast time reflects the influence of the propagation delay time of each reflection point. In view of this, it will be referred to as a delay profile hereinafter. A profile obtained by multiplying the fast time by the speed of light can be regarded as a profile for the propagation path length. Furthermore, in the case of monostatic observation in which the positions of the transmitting antenna and the receiving antenna match, 1/2 of the propagation path length of each reflection point becomes the distance to each reflection point, so the delay profile A profile obtained by multiplying the fast time by (light speed / 2) can be regarded as a profile with respect to a distance (range) from the radar. This profile is well known as a range profile.

The history (history) for each observation of each delay profile obtained in each observation is a two-dimensional distribution with the fast time and the time of each observation as two axes. The transmission time of the radio wave in each observation is referred to as slow time. Of these two types of time giving the delay history, the frequency obtained by Fourier transform (FT) of the fast time is referred to as a fast frequency, and the frequency obtained by Fourier transforming the slow time is referred to as a Doppler frequency.
Hereinafter, the fast time axis and the fast frequency axis may be collectively referred to as a fast axis, and the slow time axis and the Doppler frequency axis may be collectively referred to as a slow axis.
The profile for the fast frequency obtained by FT of the delay profile is called a delay spectrum. A history of delay spectrum obtained by FT delay history in the fast frequency direction is referred to as delay spectrum history. Further, a distribution obtained by FT delay history in the slow time direction is referred to as a delayed Doppler distribution.

  As one method for performing aperture synthesis, the delay history is FT in the slow time direction, so that “the received signal relating to each reflection point is set to one fast time on the delay Doppler distribution with the fast time and Doppler frequency as axes. A method of forming an image at a point of Doppler frequency. However, in order to form an image by such a method, it is necessary that at least the change in the propagation delay of each reflection point during observation is equal to or less than the size of the resolution cell in the fast time axis direction. If this is exceeded, the target signal will exist across multiple resolution cells, which contributes to image blurring. However, depending on the relative motion of the target and the radar, the size of the resolution cell, the length of the observation time, etc., movement beyond the resolution cell may occur. Therefore, it is necessary to compensate for the movement of the resolution cell, in other words, the change of the propagation delay time with respect to the slow time by some method, and to cancel the change.

In addition to the image radar as described above, each delay profile obtained by a plurality of observations is accumulated in a Doppler cell corresponding to the target speed by FT in the slow time direction to improve the signal-to-noise ratio. As a result, processing for improving the target detection performance may be performed.
The purpose of this processing is not to perform aperture synthesis, but is processing equivalent to the above aperture synthesis in terms of processing. This process is sometimes called coherent integration, and the above aperture synthesis can be regarded as a kind of coherent integration.
Even when this processing is performed, when blur occurs due to the movement of the target in the fast time direction, that is, when a signal stacking loss occurs, the signal pair is compared with the case where the signal stacks correctly. Since the noise ratio is degraded, the target detection performance is degraded. In order to avoid this problem, it is necessary to estimate the change of the propagation delay time with respect to the slow time and compensate so as to cancel this change, as in the case of the image radar.

  Non-Patent Document 1 below describes a radio wave propagation delay with respect to a slow time in the delay history by resampling a signal on the delay spectrum history in the slow time direction at a sampling interval corresponding to each fast frequency of the delayed spectrum history. A method for compensating for a first-order change in time has been proposed. This method is called KT (Keystone Transform).

Perry, R.P .; DiPietro, R.C .; Fante, R., “SAR imaging of moving targets,” Aerospace and Electronic Systems, IEEE Transactions on, vol.35, no.1, pp.188,200, Jan 1999

  Since the conventional radar apparatus is configured as described above, it can compensate for the primary change in the propagation delay time of the radio wave with respect to the slow time in the delay history. In such a case, there is a problem that the target detection performance may be deteriorated due to the influence of the second or higher order change.

  The present invention has been made to solve the above-described problems, and provides a radar apparatus that can prevent deterioration in target detection performance even when a propagation delay time has a second or higher order change. With the goal.

  A radar apparatus according to the present invention provides a fast signal that is an elapsed time from a transmission time of a radio wave as a reception signal of the radio wave from a radar that repeatedly transmits and receives the radio wave while changing a relative positional relationship with a target to be observed. A signal acquisition circuit that acquires a delay history that is a two-dimensional distribution of time and a slow time that is a transmission time of radio waves, and a delay spectrum that is obtained by Fourier-transforming the delay history acquired by the signal acquisition circuit in the fast time direction Time change compensation that compensates for changes in radio wave propagation delay time with respect to slow time in the delay history by resampling the signal on the delay spectrum history in the slow time direction at sampling intervals corresponding to each fast frequency in the history Circuit, and an unnecessary signal reduction circuit is a time variation compensation circuit. The delay history after compensation is divided into two at the fast frequency, and each of the divided delay histories is Fourier-transformed in the slow time direction to generate a delayed Doppler distribution, respectively, and the complex conjugate of one delayed Doppler distribution and the other The delay Doppler distribution is multiplied.

  According to the present invention, the unnecessary signal reduction circuit divides the delay history after compensation by the time change compensation circuit into two at the fast frequency, and performs Fourier transform on each of the divided delay histories in the slow time direction, thereby delay Doppler distribution. Are generated and multiplied by the complex conjugate of one delayed Doppler distribution and the other delayed Doppler distribution, so that unnecessary signals other than the target signal can be reduced. Therefore, even when a change of the second or higher order of the propagation delay time occurs, there is an effect that it is possible to prevent the target detection performance from being deteriorated.

It is a block diagram which shows the radar apparatus by Embodiment 1 of this invention. It is a hardware block diagram in case a radar apparatus is comprised with a computer. It is a block diagram which shows the unnecessary signal reduction circuit 7 of the radar apparatus by Embodiment 1 of this invention. It is explanatory drawing which shows an example of a delay Doppler image. It is explanatory drawing which shows an example of the delay Doppler image by which the Doppler band was expanded by upsampling. It is explanatory drawing which shows an example of the effect by multiplication of a weight function. It is a block diagram which shows the KT compensation circuit 6 of the image radar apparatus by Embodiment 2 of this invention. 4 is a configuration diagram showing a received signal scaling conversion circuit 45 of the KT compensation circuit 6. FIG. It is explanatory drawing which shows the merge addition process by the KT compensation circuit. It is a block diagram which shows the radar apparatus by Embodiment 3 of this invention.

  Hereinafter, in order to describe the present invention in more detail, modes for carrying out the present invention will be described with reference to the accompanying drawings.

Embodiment 1 FIG.
1 is a block diagram showing a radar apparatus according to Embodiment 1 of the present invention.
In FIG. 1, a signal acquisition circuit 1 is a fast time that is an elapsed time from a radio wave transmission time as a radio wave reception signal from a radar that repeatedly transmits and receives radio waves while changing a relative positional relationship with a target to be observed. And a delay history that is a two-dimensional distribution with the slow time that is the transmission time of the radio wave in each observation.
Specifically, after radiating a high-frequency signal generated by the transmitter from the transmitting antenna in the radar as a radio wave toward the target, the reflected wave of the radio wave reflected and returned by the target is incident on the receiving antenna. Then, the receiver obtains a delay history that is a two-dimensional distribution of the fast time and the slow time of the radio wave by repeatedly performing the process of obtaining the received signal by detecting and demodulating the reflected wave of the radio wave. In addition, processing to increase the resolution of the fast time axis (or the range axis obtained by multiplying the fast time axis by (light speed / 2)) by pulse-compressing the received signal using radio wave transmission signals as necessary. Also implement.

However, the radar does not have to be mounted separately on the transmission antenna and the reception antenna, but on a transmission / reception antenna that performs transmission and reception in a time division manner, and a transmission / reception switch that switches between the transmission signal and the reception signal. May be. Also, the radar may be a bistatic configuration in which the transmission system and the reception system are arranged at different positions, or a radar that uses existing radio waves flying in space like a broadcast wave as a transmission wave. Also good.
When using an existing radio wave as a transmission wave, a first reception system that receives a target scattered wave and a second reception system that receives a direct wave from a transmission station are prepared. The high resolution of the fast time axis is realized by cross-correlation. In this case, the propagation delay time is a value based on the propagation delay time of the direct wave path. Even if the existing radio wave used is a continuous wave instead of the pulse waveform used in general image radar, a different slow time can be obtained by cutting the reflected wave or direct wave at an appropriate time width or time interval. The received signal at can be obtained.

The relative motion specifying circuit 2 collects the motion specifications of the target from the external device, specifies the relative motion with the target from the motion specifications, and estimates the change in the propagation delay time of the radio wave with respect to the slow time from the relative motion. In addition, as an estimation error of the change in the propagation delay time, the predicted value s _{e1 of the} primary change in the propagation delay time of the radio wave with respect to the slow time, and the predicted value s _{e2 of the} secondary change in the propagation delay time of the radio wave with respect to the slow time Is a circuit for specifying
As an external device, for example, a target tracking device which is a radar for tracking a target, a database storing target motion parameters, and the like can be considered, and the motion parameters such as target position, speed and acceleration can be obtained from the external device. collect. Further, when the own radar apparatus is not stationary and is mounted on the radar platform, the motion specifications of the own radar apparatus are collected from the motion sensor and position sensor mounted on the radar platform.
By obtaining the difference between the motion specification of the target and the motion specification of the own radar apparatus, the relative motion with respect to the target can be specified.

The preprocessing propagation delay compensation amount estimation circuit 3 is a circuit that estimates a propagation delay compensation amount that cancels a change in the propagation delay time estimated by the relative motion identification circuit 2.
The preprocessing propagation delay compensation circuit 4 is a circuit that compensates the delay history acquired by the signal acquisition circuit 1 using the propagation delay compensation amount estimated by the preprocessing propagation delay compensation amount estimation circuit 3.

The up-sampling circuit 5 is a circuit that up-samples the delay spectrum history obtained by Fourier transforming the delay history compensated by the preprocessing propagation delay compensation circuit 4 in the fast time direction in the slow time direction.
That is, the upsampling circuit 5 sets N as the number of upsampling according to the predicted value _{se1 of the} primary change of the propagation delay time of the radio wave with respect to the slow time specified by the relative motion specifying circuit 2, and the delay spectrum history This is a circuit for multiplying the sampling frequency of the delay spectrum history by N times by inserting (N−1) 0 cells between cells in the slow time direction in FIG.

The KT compensation circuit 6 is improved in the slow time direction by the upsampling circuit 5 by the improvement technique of the KT disclosed in the above Non-Patent Document 1 or the general KT disclosed in the following Non-Patent Document 2. By resampling the signal on the delay spectrum history in the slow time direction at the sampling interval corresponding to each fast frequency of the sampled delay spectrum history, the change of the propagation delay time of the radio wave with respect to the slow time in the delay history can be changed. It is a time change compensation circuit to compensate.
As a result, the change in the primary propagation delay time is eliminated for the true image that is the target image, but the change in the primary propagation delay time is not eliminated for the false image other than the true image.
[Non-Patent Document 2]
Daiyin Zhu; Li, Yong; Zhaoda Zhu, “A Keystone Transform Without Interpolation for SAR Ground Moving-Target Imaging,” Geoscience and Remote Sensing Letters, IEEE, vol.4, no.1, pp.18, 22, Jan. 2007

The unnecessary signal reduction circuit 7 divides the delay history in which the change of the propagation delay time of the radio wave with respect to the slow time is compensated by the KT compensation circuit 6 into two at the fast frequency, and Fourier-transforms each of the divided delay histories in the slow time direction. Thus, the delay Doppler distribution is generated, and the complex conjugate of one delay Doppler distribution and the other delay Doppler distribution are multiplied to reduce unnecessary signals other than the target signal.
The propagation delay Doppler specifying circuit 8 detects the target image from the delay Doppler distribution in which the unnecessary signal is reduced by the unnecessary signal reduction circuit 7, and the position on the fast time where the target image exists and the target Doppler frequency. Is a circuit for specifying

The primary change compensation amount calculation circuit 9 compensates the primary change compensation amount for compensating the primary change of the propagation delay time of the radio wave with respect to the slow time in the delay history from the target Doppler frequency specified by the propagation delay Doppler specifying circuit 8. Is a circuit for calculating.
The primary change compensation circuit 10 uses the primary change compensation amount calculated by the primary change compensation amount calculation circuit 9 to use the radio wave for the slow time remaining in the delay history compensated by the preprocessing propagation delay compensation circuit 4. This circuit compensates for a primary change in the propagation delay time.

The reference signal extraction circuit 11 is based on the fast time where the target image specified by the propagation delay Doppler specifying circuit 8 exists from the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit 10. This circuit extracts all slow time signals at positions as reference signals.
The propagation delay compensation amount precise estimation circuit 12 has a propagation delay time remaining in the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit 10 from the reference signal extracted by the reference signal extraction circuit 11. Propagation delay compensation amount that estimates the phase change that occurs due to the effect of the change, eliminates the folding of the phase change every 2π, converts the phase change after the elimination of the folding into a propagation delay change, and cancels the propagation delay change Is a circuit for estimating

The propagation delay change compensation circuit 13 uses the propagation delay compensation amount estimated by the propagation delay compensation amount precise estimation circuit 12 and remains in the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit 10. This circuit compensates for a change in propagation delay time.
The imaging circuit 14 is a circuit that performs imaging by Fourier-transforming the delay history after compensation by the propagation delay change compensation circuit 13 in the slow time direction.

In the example of FIG. 1, the signal acquisition circuit 1, the relative motion identification circuit 2, the preprocessing propagation delay compensation amount estimation circuit 3, the preprocessing propagation delay compensation circuit 4, the upsampling circuit 5, and the KT compensation circuit, which are constituent elements of the radar apparatus. 6, unnecessary signal reduction circuit 7, propagation delay Doppler identification circuit 8, primary change compensation amount calculation circuit 9, primary change compensation circuit 10, reference signal extraction circuit 11, propagation delay compensation amount precise estimation circuit 12, propagation delay change compensation It is assumed that each of the circuit 13 and the imaging circuit 14 is configured by dedicated hardware. As the dedicated hardware, for example, a semiconductor integrated circuit in which a CPU (Central Processing Unit) is mounted, a one-chip microcomputer, a semiconductor integrated circuit in which a GPU (Graphics Processing Unit) is mounted can be considered.
However, the radar apparatus of the first embodiment may be configured with a computer.
FIG. 2 is a hardware configuration diagram when the radar apparatus is configured by a computer.
When the radar apparatus is configured by a computer, the signal acquisition circuit 1, the relative motion specifying circuit 2, the preprocessing propagation delay compensation amount estimation circuit 3, the preprocessing propagation delay compensation circuit 4, the upsampling circuit 5, the KT compensation circuit 6, and unnecessary. A signal reduction circuit 7, a propagation delay Doppler identification circuit 8, a primary change compensation amount calculation circuit 9, a primary change compensation circuit 10, a reference signal extraction circuit 11, a propagation delay compensation amount precise estimation circuit 12, a propagation delay change compensation circuit 13, and A program describing the processing contents of the imaging circuit 14 may be stored in the memory 21 of the computer, and the processor 22 of the computer may execute the program stored in the memory 21.

FIG. 3 is a block diagram showing the unnecessary signal reduction circuit 7 of the radar apparatus according to Embodiment 1 of the present invention.
In FIG. 3, the delay Doppler distribution generation circuit 31 converts the delay history compensated for the change in the propagation delay time of the radio wave with respect to the slow time by the KT compensation circuit 6 into a low-frequency delay history and a high-frequency delay history at a fast frequency. By dividing into two and Fourier transforming the low-frequency delay history in the slow time direction, a low-frequency delay Doppler distribution is generated at the fast frequency, and the high-frequency delay history is Fourier-transformed in the slow time direction. This is a circuit that generates a high-frequency delayed Doppler distribution at a fast frequency.

The delay Doppler distribution conjugate multiplication circuit 32 multiplies the complex conjugate of one delay Doppler distribution and the other delay Doppler distribution among the low-frequency and high-frequency delay Doppler distributions generated by the delay Doppler distribution generation circuit 31. Circuit.
The Doppler axis inverse Fourier transform circuit 33 is a circuit that obtains a two-dimensional distribution of fast time and slow time by performing inverse Fourier transform on the delayed Doppler distribution after conjugate multiplication by the delayed Doppler distribution conjugate multiplier circuit 32 in the Doppler axis direction.
The Doppler width calculation circuit 34 calculates the second-order change in radio wave propagation delay time with respect to the slow time specified by the relative motion specifying circuit 2 from the predicted value s _e2 on the delay Doppler distribution after conjugate multiplication by the delay Doppler distribution conjugate multiplier circuit 32. This is a circuit for calculating the Doppler width of the target image.

The weight function multiplication circuit 35 sets a slow time width that is inversely proportional to the Doppler width calculated by the Doppler width calculation circuit 34, and the target signal is present in the two-dimensional distribution obtained by the Doppler axis inverse Fourier transform circuit 33. This is a circuit that sets a weighting function that passes a signal of a slow time in the slow time width and prevents a signal of a slow time other than the slow time in which the target signal is present.
The weight function multiplication circuit 35 is a circuit that multiplies the set weight function by the two-dimensional distribution obtained by the Doppler axis inverse Fourier transform circuit 33.
The slow time axis Fourier transform circuit 36 Fourier-transforms the two-dimensional distribution multiplied by the weight function by the weight function multiplication circuit 35 in the slow time direction, thereby obtaining a delayed Doppler distribution in which unnecessary signals other than the target signal are suppressed. It is a circuit to obtain.

Next, the operation will be described.
A radar (not shown) repeatedly transmits radio waves toward a target to be observed.
That is, the radar radiates a radio wave that is a high-frequency signal generated by the transmitter from the transmission antenna toward the target.
In addition, when the reflected wave of the radio wave reflected and returned from the target is incident on the receiving antenna, the radar detects a received wave of the radio wave and demodulates it to obtain a received signal.
The signal acquisition circuit 1 receives, as a radio wave reception signal from the radar, a delay history that is a two-dimensional distribution of a fast time τ that is an elapsed time from a radio wave transmission time and a slow time η that is a radio wave transmission time in each observation. To get.

Hereinafter, the delay history acquired by the signal acquisition circuit 1 is represented by G _tt (τ, η), the fast time τ is Fourier transformed (FT: Fourier Transform), the fast frequency in the baseband is ρ, and the slow time η is FT. Let the Doppler frequency be denoted by γ.
Further, the delay spectrum history obtained by performing the FT of the delay history G _tt (τ, η) in the fast time τ direction is expressed as G _ft (ρ, η), and the delay history G _tt (τ, η) is expressed as the slow time η. The delayed Doppler distribution obtained by performing FT in the direction is represented by G _tf (τ, γ).
Of the two subscripts ( _tt , _ft , _tf ) in _Gtt (τ, η), _Gft (ρ, η) and _Gtf (τ, γ), the left subscript indicates the fast axis. If the fast axis is the time domain, it is “t”, and if it is the frequency domain, it is “f”. The subscript on the right side indicates the slow axis, and is “t” if the slow axis is the time domain and “f” if the slow axis is the frequency domain.

式（１）において、Ｃは光速である。 The relative motion specifying circuit 2 collects the target motion parameters from the external device, specifies the relative motion with the target from the motion parameters, and changes the propagation delay time s of the radio wave from the relative motion to the slow time η. _Pre (η) is estimated.
For example, in a coordinate system common to the radar and the target, when the time during which the radar is observing the target is η, the radar position vector is r _rdr (η) and the target position vector is r _tgt (η). The propagation delay time s (η) can be calculated by the following equation (1). In the text of the specification, the vector “r” is described in thin letters, but in the expression (1), “r” is described in bold letters.

In formula (1), C is the speed of light.

上記のようにして、各スロータイムηでの伝搬遅延時間ｓ（η）が分かれば、伝搬遅延時間の変化ｓ_ｐｒｅ（η）が求まる。
ここでは、相対運動特定回路２が、伝搬遅延時間の変化ｓ_ｐｒｅ（η）を推定するようにしているが、目標の運動諸元を格納しているデータベースなどから、伝搬遅延時間ｓ（η）の概算値を取得して、伝搬遅延時間の変化ｓ_ｐｒｅ（η）を求めるようにしてもよい。 If the position of the radar transmitter and the receiver are different, _assuming that the transmitter position vector is r _tra (η) and the receiver position vector is r _rec (η), the propagation delay time s (η) is It can be calculated by the following equation (2).

As described above, if the propagation delay time s (η) at each slow time η is known, the change s _pre (η) of the propagation delay time can be obtained.
Here, although the relative motion specifying circuit 2 estimates the change s _pre (η) of the propagation delay time, the propagation delay time s (η) is obtained from a database storing the target motion specifications. May be obtained to obtain the change s _pre (η) of the propagation delay time.

ここでは、残存する伝搬遅延時間の変化ｓ_ｐｒｅ（η）によるスロータイムηの経過に伴う反射点のファストタイム方向の分解能セルを超えた移動であるマイグレーションは、スロータイムηの１次成分のみで与えられるものとする。 When the relative motion specifying circuit 2 estimates the change s _pre (η) of the propagation delay time of the radio wave with respect to the slow time η, the accuracy of the target tracking device and database which are external devices, the motion sensor and position mounted on the radar platform Information indicating the accuracy of the sensor is acquired, and the estimation error s _e (η) of the change s _pre (η) of the propagation delay time is specified based on the accuracy. Needless to say, if their accuracy is poor, the estimation error s _e of the propagation delay time change s _{pre (η) (η)} is increased, if you prefer those accuracy, the propagation delay time change s _{pre (η)} The estimation error s _e (η) becomes smaller.
In the first embodiment, the relative motion specifying circuit 2 uses the estimated error s _e (η) of the change s _pre (η) of the propagation delay time as the estimated value of the primary change in the propagation delay time of the radio wave with respect to the slow time η. s _e1 and the predicted value s _{e2 of the} secondary change of the propagation delay time of the radio wave with respect to the slow time η are specified.
The estimation error s _e (η) of the propagation delay time change s _pre (η) is represented by the sum of the first and second order terms relating to the slow time η, as shown in the following equation (3).

Here, the migration, which is the movement of the reflection point beyond the resolution cell in the fast time direction with the passage of the slow time η due to the change s _pre (η) of the remaining propagation delay time, is only the primary component of the slow time η. Shall be given.

The preprocessing propagation delay compensation amount estimation circuit 3 cancels the propagation delay time change s _pre (η) when the relative motion identification circuit 2 estimates the change s _pre (η) of the radio wave propagation delay time with respect to the slow time η. The propagation delay compensation amount q _pre (η) is estimated.
That is, the preprocessing propagation delay compensation amount estimation circuit 3 uses the propagation delay compensation amount q _pre (η as a result of inverting the sign of the change s _pre (η) of the propagation delay time as shown in the following equation (4). ).
q _pre (η) = − s _pre (η) (4)

そして、前処理伝搬遅延補償回路４は、補償後の遅延スペクトルヒストリＧ_ｆｔ（ρ，η）をファスト周波数方向に逆フーリエ変換（ＩＦＴ：Ｉｎｖｅｒｓｅ ＦＴ）することで補償後の遅延ヒストリＵ_ｔｔ（τ，η）を求める。 Preprocessing propagation delay compensation circuit 4, when the pretreatment propagation delay compensation amount estimating circuit 3 estimates a propagation delay compensation amount q _{pre (eta),} using the propagation delay compensation amount q _{pre (eta),} signal acquisition circuit 1 The delay history G _tt (τ, η) acquired by the above is compensated, and the compensated delay history U _tt (τ, η) is output.
That is, the preprocessing propagation delay compensation circuit 4 obtains the delay spectrum history G _ft (ρ, η) by performing an FT on the delay history G _tt (τ, η) acquired by the signal acquisition circuit 1 in the slow time direction, As shown in the following equation (5), the delay spectrum history G _ft (ρ, η) is compensated using the propagation delay compensation amount q _pre (η) estimated by the preprocessing propagation delay compensation amount estimation circuit 3. .
In equation (5), the delay spectrum history G _ft (ρ, η) before compensation is represented by G _{in, ft} (ρ, η), and the delayed spectrum history G _ft (ρ, η) after compensation is represented by G _{out, ft.} (Ρ, η).

Then, the preprocessing propagation delay compensation circuit 4 performs an inverse Fourier transform (IFT: Inverse FT) on the compensated delay spectrum history G _ft (ρ, η) in the fast frequency direction, so that the compensated delay history U _tt (τ , Η).

In the delay history U _tt (τ, η), when the accuracy of the preprocessing compensation by the preprocessing propagation delay compensation circuit 4 is low, the reflection point moves beyond the resolution cell in the fast time direction as the slow time η elapses. There may be a migration that is.
In the KT compensation circuit 6 to be described later, the compensation for eliminating the migration occurring in the delay history U _tt (τ, η) is estimated by estimating the compensation amount in the slow time η direction according to the fast frequency ρ of the delay spectrum history. There is an advantage that can be realized without requiring.
However, in order for a general KT process to operate correctly, it is necessary that at least the target Doppler frequency γ does not return due to the pulse repetition frequency. Therefore, for a target whose propagation delay time is large and the Doppler frequency γ is turned back, only the primary change in the apparent Doppler frequency γ is compensated, and the primary change in the target true Doppler frequency γ is compensated. Not.

Therefore, the upsampling circuit 5 FTs the delay history U _tt (τ, η) after compensation by the preprocessing propagation delay compensation circuit 4 in the fast time τ direction in order to widen the application limit related to the Doppler frequency γ in the KT process. The delay spectrum history U _ft (ρ, η) obtained in this way is up-sampled in the slow time η direction.
Specifically, the delay spectrum history U _ft (ρ, η) is up-sampled as follows.

The fast (fast time τ, fast frequency ρ) axis and slow axis (slow time η, Doppler frequency γ) of the delay history U _tt (τ, η) after compensation by the preprocessing propagation delay compensation circuit 4 are discretized. It shall be.
When performing upsampling of the delayed spectrum history U _ft (ρ, η) by N (N is an integer of 1 or more) times in the slow time η direction, the original delayed spectrum history U _ft (ρ, η) (N−1) zero cells are inserted between each cell of slow time η.
As a result, the apparent sampling frequency is increased, and the bandwidth of the Doppler frequency γ is expanded N times even in the delay spectrum Doppler distribution obtained by FT delay spectrum history after cell insertion in the slow time η direction. Is done.

Here, FIG. 4 is an explanatory diagram showing an example of a delayed Doppler image, where (a) is a delayed Doppler image that is blurred due to the influence of relative motion between the radar and the target, and (b) is an ideal image that is not blurred. Shows a delayed Doppler image.
FIG. 4 shows an example in which images of two targets (target A and target B) exist on the same image.
FIG. 5 is an explanatory diagram showing an example of a delayed Doppler image in which the Doppler band is expanded by upsampling. (A) shows a delayed Doppler image immediately after upsampling, and (b) shows a delayed Doppler image after KT processing. ing.

For example, when the original delayed Doppler distribution is an image as shown in FIG. 4A, if the delayed spectrum history U _ft (ρ, η) is up-sampled N times in the slow time η direction, FIG. As shown in a), for each target, an image that is repeated with a period 1 / F _p [s] of the pulse repetition frequency F _p [Hz] in the Doppler frequency γ direction, that is, a replica of the same shape is N ( In the example of FIG. 5, N = 5) occurs.
At this point, the images of the two targets (target A and target B) are still blurred, but one of these replicas is present at the position of the target true Doppler frequency γ.
Therefore, when KT processing described later is applied to an image that does not exist at the position of the target true Doppler frequency γ (hereinafter referred to as “false image”), as shown in FIG. The image existing at the position of the Doppler frequency γ (hereinafter referred to as “true image”) forms an image and becomes so blurred that the distance from the position of the true Doppler frequency γ increases.
In the example of FIG. 5, the third true image from the left is formed for the target A, and the fourth true image from the left is formed for the target B.
Therefore, the generation of the target image in which the primary component is correctly compensated can be substituted by the cut out of the formed true image.

In this example, the delay spectrum history U _ft (ρ, η) is up-sampled N times in the slow time η direction. However, as a method of determining N, the true Doppler frequency γ is surely included. A method of setting as large a value as possible in the system can be considered.
This method has an advantage that the KT process can be operated correctly when a sufficiently large N that can cover the target Doppler frequency γ of any relative motion that is expected to be observed can be secured. However, increasing N may cause a problem in terms of processing load because the data handled increases. Therefore, when it is possible to predict to the extent that the magnitude of the primary change in the propagation delay time is based on prior information, it is useful in terms of reducing the processing load to set N to a level that just includes the Doppler frequency γ.
Therefore, the upsampling circuit 5 sets N as the number of upsampling according to the predicted value s _{e1 of the} primary change of the propagation delay time s (η) of the radio wave with respect to the slow time η specified by the relative motion specifying circuit 2. To do.

式（６）において、Ｆ_ｃはレーダから放射される電波である送信信号の中心周波数である。
これにより、真像に関しては、ＫＴ補償回路６が正しく動作して、マイグレーションが補償される。 Since the magnitude of the Doppler frequency γ corresponding to the predicted value s _e1 of the primary change is | F _c s _e1 | = F _c | s _e1 | [Hz], considering the sign of the Doppler frequency γ, By setting N so as to satisfy the following expression (6), the target true image can be contained within the expanded Doppler width.

In Equation (6), F _c is the center frequency of the transmission signal that is a radio wave radiated from the radar.
As a result, for the true image, the KT compensation circuit 6 operates correctly and the migration is compensated.

When the up-sampling circuit 5 up-samples the delay spectrum history U _ft (ρ, η) in the slow time η direction, the KT compensation circuit 6 receives each fast frequency ρ of the delayed spectrum history U _ft (ρ, η) after the up-sampling. By resampling the signal on the delay spectrum history U _ft (ρ, η) in the slow time η direction at a sampling interval corresponding to, the propagation of radio waves with respect to the slow time η in the delay history U _tt (τ, η) Compensates for changes in delay time.
Here, in the KT process, when the delay spectrum history U _ft (ρ, η) is expressed in the form of the following expression (7), for example, the fast frequency ρ is expressed by the following expression (8). This is a technique for obtaining the compensated delay spectrum history U _ft (ρ, F _c η / (F _c + ρ)) by changing the scale according to the resampling and resampling. In the equation (8), the change component with respect to the slow time η does not depend on the fast frequency ρ. Therefore, on the delay history corresponding to the compensated delay spectrum history U _ft (ρ, F _c η / (F _c + ρ)). The migration at the same time is also eliminated.

式（７）（８）において、ｓ_０は目標が存在しているファストタイム上の位置である０次の伝搬遅延時間である。
以上により、予測値ｓ_ｅ１が示す伝搬遅延時間の１次変化とドップラー周波数γが整合する真像はマイグレーションが正しく補償される。しかし、偽像では、伝搬遅延時間の１次変化とドップラー周波数γが整合しないので、マイグレーションが正しく補償されない。

In equations (7) and (8), s ₀ is a zero-order propagation delay time that is a position on the fast time where the target exists.
As described above, the _{true image in} which the primary change in the propagation delay time indicated by the predicted value s _e1 matches the Doppler frequency γ is correctly compensated for migration. However, in the false image, the primary change in the propagation delay time and the Doppler frequency γ do not match, so that the migration is not compensated correctly.

As a result, the true image on the delayed Doppler distribution U _tf (τ, γ) correctly accumulates signals, while the false image does not accumulate as much as the true image. As a result, the detection performance of the target signal, which is a true image in a noisy environment, is improved, and an effect of enabling discrimination of the true image from among the true image and the false image is expected. Therefore, it is also useful to provide a path that goes from the KT compensation circuit 6 to the propagation delay Doppler specifying circuit 8 without passing through the unnecessary signal reduction circuit 7.
However, if the propagation delay time s (η) includes a second-order component, a change in the Doppler frequency γ occurs due to a second-order phase change in the received signal. A situation that does not rise (situation that blurs in the direction of the Doppler axis) occurs. As a result, there is a concern that the detection performance of a true image in a noisy environment may deteriorate, or the performance of discriminating a true image from among the inclusion of a true image and a false image.

The unnecessary signal reduction circuit 7 is provided to prevent degradation of the true image detection performance and discrimination performance, and divides the delay history after KT compensation by the KT compensation circuit 6 into two by the fast frequency ρ. Each delayed history is subjected to FT in the slow time η direction to generate a delayed Doppler distribution, and by multiplying the complex conjugate of one delayed Doppler distribution and the other delayed Doppler distribution, Reduce unnecessary signals.
Hereinafter, the unnecessary signal reduction processing by the unnecessary signal reduction circuit 7 will be described in detail.

The delay Doppler distribution generation circuit 31 of the unnecessary signal reduction circuit 7 divides the delay history after the KT compensation by the KT compensation circuit 6 into a low-frequency delay history and a high-frequency delay history at the fast frequency ρ.
Next, the delay Doppler distribution generation circuit 31 generates a low-frequency-side delay Doppler distribution at the fast frequency ρ by performing FT on the low-frequency-side delay history in the slow time η direction. Is delayed in the slow time η direction to generate a high-frequency delayed Doppler distribution at the fast frequency ρ.

式（９）（１０）において、Ｂは遅延スペクトルヒストリＰ_ｆｔ（ρ，η）の帯域幅である。 Here, when the delay spectrum history obtained by performing the FT compensation delay history in the fast time τ direction is represented by P _ft (ρ, η), the low-frequency side delay spectrum history L _ft (ρ, η) is expressed by the following equation: The delay spectrum history H _ft (ρ, η) on the high frequency side is expressed as in the following equation (10).

In equations (9) and (10), B is the bandwidth of the delayed spectrum history P _ft (ρ, η).

Next, the delay Doppler distribution generation circuit 31 performs an IFT on the low-frequency side delay spectrum history L _ft (ρ, η) in the fast frequency ρ direction, and then performs an FT in the slow time η direction, thereby reducing the low-frequency side delay spectrum history L _ft (ρ, η). A delayed Doppler distribution L _tf (τ, γ) is generated.
Further, the high frequency side delay spectrum history H _ft (ρ, η) is subjected to IFT in the fast frequency ρ direction, and then subjected to FT in the slow time η direction, whereby the high frequency side delay Doppler distribution H _tf (τ, γ ) Is generated.

Here, the position of the true image on the low-frequency side delayed Doppler distribution L _tf (τ, γ) and the high-frequency side delayed Doppler distribution H _tf (τ, γ) will be described.
As described above, the migration has already been eliminated by the effect of the KT process. Therefore, these delayed Doppler distributions L _tf (τ, γ) and H _tf (τ, γ) do not spread in the fast time τ direction. However, due to the influence of the acceleration component of the relative motion (secondary component of the propagation delay time), the Doppler frequency γ direction depends on the acceleration and has a width that is substantially equal on the low frequency side and the high frequency side. . Strictly speaking, as will be described later, a slightly different spread is caused by the difference between the center frequency on the low frequency side and the center frequency on the high frequency side. For a simple target with one reflection point, the spread is represented by a generally rectangular shape.

Originally, the Doppler frequency γ at each fast frequency ρ that changes depending on the primary change of the propagation delay time and each fast frequency ρ is eliminated by the effect of compensation by KT processing, and the dependence on each fast frequency ρ is eliminated. Te, because determined by the center frequency F _c of the delay spectrum history, the position of the upper position and the Doppler frequency γ on fast time τ a true image is present, consistent with the low frequency side and high frequency side. In addition, the change of the phase on the true image with respect to the Doppler frequency γ substantially coincides between the low frequency side and the high frequency side.
Accordingly, the true Doppler frequency of the phase on the true image is obtained by multiplying one of the true image on the low-frequency side and the true image on the high-frequency side by the complex conjugate image of the other true image. Changes to γ can be canceled.

これにより、共役乗算後の遅延ドップラー分布Ｍ_ｔｆ（τ，γ）をドップラー軸方向にＩＦＴすることで得られる遅延ヒストリ（以下、「共役乗算後の遅延ヒストリ」と称する）上の真像の分布が、元のドップラー位置によらず、必ず０スロータイム付近に局在化する。
ここでは、低域側の遅延ドップラー分布Ｌ_ｔｆ（τ，γ）の複素共役Ｌ_ｔｆ（τ，γ）^＊を高域側の遅延ドップラー分布Ｈ_ｔｆ（τ，γ）に乗算している例を示しているが、高域側の遅延ドップラー分布Ｈ_ｔｆ（τ，γ）の複素共役Ｈ_ｔｆ（τ，γ）^＊を低域側の遅延ドップラー分布Ｌ_ｔｆ（τ，γ）に乗算しても、同様に位相をキャンセルさせることができる。 When the delayed Doppler distribution generation circuit 31 generates the low-frequency delayed Doppler distribution L _tf (τ, γ) and the high-frequency delayed Doppler distribution H _tf (τ, γ), As shown in Equation (11), the complex conjugate L _tf (τ, γ) ^* of the low-frequency side delayed Doppler distribution L _tf (τ, γ) is represented by the high-frequency side delayed Doppler distribution H _tf (τ, γ). And a delayed Doppler distribution M _tf (τ, γ) after conjugate multiplication in which the change of the phase on the true image with respect to the Doppler frequency γ is canceled is output.

Thus, the true image distribution on the delay history (hereinafter referred to as “delay history after conjugate multiplication”) obtained by performing IFT on the delayed Doppler distribution M _tf (τ, γ) after conjugate multiplication in the Doppler axis direction. However, it is localized near the zero slow time regardless of the original Doppler position.
Here, an example is multiplied to the low frequency side of the delay Doppler distribution L _{tf (τ,} γ) the complex conjugate of L _{tf (τ,} γ) ^* the high band side delay Doppler distribution H _{tf (τ,} γ) shows, but the high frequency side delay Doppler distribution H _{tf (τ,} γ) complex conjugate H _{tf (τ,} γ) of be multiplied by the ^* in the low-frequency side delay Doppler distribution L _{tf (τ,} γ) Similarly, the phase can be canceled.

When the Doppler axis inverse Fourier transform circuit 33 receives the delayed Doppler distribution M _tf (τ, γ) after conjugate multiplication from the delayed Doppler distribution conjugate multiplier circuit 32, the Doppler axis inverse Fourier transform circuit 33 converts the delayed Doppler distribution M _tf (τ, γ) into the Doppler axis direction. To obtain a two-dimensional distribution of fast time τ and slow time η.
As a result, for example, when the true image on the original delayed Doppler distribution is given by a rectangular distribution rect ((γ−γ _c ) / W _M ) whose width in the Doppler axis direction is W _M [Hz], after conjugate multiplication The distribution on the delay history can be expressed in the form of sinc (W _M η) when a constant multiplication and a phase irrelevant to the discussion are omitted. γ _c is the central Doppler.
However, sinc (x) = sinπx / (πx), that is, a station with a center of 0 slow time and a width of approximately 1 / W _M [s] (Null-to-Null in 2 / W _M [s]). It becomes natural.
On the other hand, the noise spreads over all slow time regions, and the false image is affected by the difference in phase change with respect to the Doppler frequency and the Doppler frequency shift between the high frequency image and the low frequency image. It does not stay only in the vicinity, but diffuses to a slow time region away from zero. Therefore, unnecessary signals other than the true image can be suppressed by utilizing the difference in these characteristics.

このとき、Ｆ_Ｌ＜Ｆ_Ｈであることを踏まえると、共役乗算後の遅延ドップラー分布Ｍ_ｔｆ（τ，γ）上の真像のドップラー幅Ｗ_Ｍは、下記の式（１３）で与えられる。

The Doppler width calculation circuit 34 performs the conjugate multiplication output from the delay Doppler distribution conjugate multiplication circuit 32 from the predicted value s _{e2 of the} secondary change of the propagation delay time of the radio wave with respect to the slow time η specified by the relative motion specifying circuit 2. The Doppler width W _M of the true image on the delayed Doppler distribution M _tf (τ, γ) is calculated.
For example, the center frequency F _L of the delay history on the low frequency side is F _L = F _c − (B / 4), the center frequency F _H of the delay history on the high frequency side is F _H = F _c + (B / 4), Assuming that the observation slow time width is T, the change of the true image Doppler frequency γ in each delay history, that is, each Doppler width W _i (i = L, H) on the delay Doppler distribution is expressed by the following formula ( 12).

At this time, considering that F _L <F _H , the true image Doppler width W _M on the delayed Doppler distribution M _tf (τ, γ) after conjugate multiplication is given by the following equation (13).

Weighting function multiplying circuit 35, the Doppler width calculating circuit 34 calculates the Doppler width W _M of the true image, delay Doppler distribution after conjugate multiplication M _{tf (τ,} γ) the signal of the true image on is localized slow time width is inversely proportional to the Doppler width W _M, to set a slow time width α / W _M that is inversely proportional to the Doppler width W _M. α is an arbitrary constant.
Weighting function multiplying circuit 35, setting the slow time width alpha / W _M, of the two-dimensional distribution obtained by the Doppler Jikugyaku Fourier transform circuit 33, near 0 slow-time signal true image is localized , I.e., a slow function having a slow time width α / W _M , while passing a signal other than the vicinity of the zero slow time where the true image signal is not localized. Set.
The shape of the weight function may be a rectangular distribution, or may be a general one including other distributions such as Hanning, Taylor, Chebyshev, and Gauss.
Here, although the set of slow time width alpha / W _M is inversely proportional to the Doppler width W _M, it may be fixed for bandpass around 0 slow time in the weight function. Further, it may be changed for each target category (for example, ship, aircraft, automobile, etc.) to be observed.
When the weight function multiplying circuit 35 sets a weight function that passes a signal near 0 slow time and blocks the passage of signals other than near 0 slow time, the weight function is obtained by the Doppler axis inverse Fourier transform circuit 33. Multiply the two-dimensional distribution.
Thereby, in the two-dimensional distribution multiplied by the weighting function, noise and false image signals are suppressed.

The slow time axis Fourier transform circuit 36 delays the Doppler in which unnecessary signals other than the true image signal are suppressed by performing the two-dimensional distribution multiplied by the weight function by the weight function multiplication circuit 35 in the slow time η direction. Get the distribution.
Since unnecessary signals are significantly suppressed by multiplication of the weight function on the slow time axis, the visibility of the true image, that is, the detection performance of the true image is improved compared to before multiplication of the weight function.

FIG. 6 is an explanatory diagram showing an example of the effect of multiplication of the weight function.
The upper part of FIG. 6 shows the Doppler distribution at the fast time τ where the true image before multiplication by the weight function exists, and the lower part of FIG. 6 shows the fast time τ where the true image after multiplication by the weight function exists. Shows the Doppler distribution at.
In the example of FIG. 6, the secondary change in the propagation delay time assumed at the time of signal generation increases from the left to the right. The width of the weighting coefficient is a constant value determined from the width corresponding to the second largest secondary change from the right.

Thus, it is difficult to detect a true image signal when the secondary change of the propagation delay time is large before the multiplication of the weight function, but the secondary change of the propagation delay time is large after the multiplication of the weight function. Even in this case, the effect of detecting a true image signal is confirmed.
In addition, despite the aim of the second relatively large secondary change from the right, other secondary changes can be detected well. In particular, when the secondary change in the propagation delay time is small, the width of the true image signal increases on the slow time axis, and thus the weight when assuming a secondary change larger than this, that is, the slow time width is narrow. The weights also block some of the true image signals.
However, when the secondary change of the propagation delay time in which such a situation occurs is small, as shown in the upper diagram, the accumulation in the original Doppler distribution has already been large, so that the detection performance is maintained. .
Based on this, it is useful to configure the weight using the maximum assumed secondary change, and this is adopted as one of the criteria for setting the passing slow time width in the present invention. In addition, instead of using the maximum value, there is also a method of determining a secondary change that maintains the detection performance as much as possible within the range of the assumed secondary change of the target by a prior simulation with the parameters shown here. This is useful, and is also adopted in the present invention as one of the methods for setting the passage slow time width.

When the propagation delay Doppler specifying circuit 8 receives the delay Doppler distribution in which the unnecessary signal is reduced from the unnecessary signal reduction circuit 7, it detects a true image from the delay Doppler distribution.
When detecting the true image from the delay Doppler distribution, the propagation delay Doppler specifying circuit 8 specifies the 0th-order propagation delay time _sest0 that is the position on the fast time τ where the true image exists, and also detects the true image Doppler. _{Specify the} frequency _Fdop .

The primary variation compensation amount calculation circuit 9, the propagation delay Doppler particular circuit 8 identifies the Doppler frequency F _dop true image, the primary change s _est1 propagation delay time corresponding to the Doppler frequency F _dop true image, true Since the image Doppler frequency _Fdop is associated with the following equation (14), the first-order variation compensation for canceling the first-order variation _sest1 of the propagation delay time as shown in the following equation (15): The quantity q _est (η) is calculated.

When the primary change compensation amount calculation circuit 9 calculates the primary change compensation amount q _est (η), the primary change compensation circuit 10 uses the primary change compensation amount q _est (η) to perform a preprocessing propagation delay. The first-order change s _est1 of the propagation delay time remaining in the delay history U _tt (τ, η) compensated by the compensation circuit 4 is compensated, and the delay history X after the first-order change compensation (after the first-order coarse compensation) _tt (τ, η) is output.
The primary change compensation amount q _est (η) used by the primary change compensation circuit 10 for the compensation process of the delay history U _tt (τ, η) is the same as the pre-processing propagation delay compensation circuit 4 and the delay history G _tt (τ, η) Is different from the compensation amount q _pre (η) used in the compensation process of FIG. 1, but the compensation process itself is the same as that of the pre-process propagation delay compensation circuit 4, and therefore the delay history U _tt (τ, A detailed description of the compensation process of η) will be omitted.

When the reference signal extraction circuit 11 receives the delay history X _tt (τ, η) after the primary change compensation (after the primary coarse compensation) from the primary change compensation circuit 10, the delay history X _tt (τ, η) From all the signals of the slow time η in the 0th-order propagation delay time _sest0 that is the position on the fast time τ where the true image specified by the propagation delay Doppler specifying circuit 8 exists is referred to as the reference signal Y (η). Extract as
That is, the reference signal extraction circuit 11 extracts a data string arranged in the slow time η direction of the fast time cell where the true image exists as the reference signal Y (η) as shown in the following equation (16).

When the reference signal extraction circuit 11 extracts the reference signal Y (η), the propagation delay compensation amount estimation circuit 12 extracts the reference signal Y (η) from the reference signal Y (η) after primary change compensation by the primary change compensation circuit 10 (primary order). A phase change caused by a change in propagation delay time remaining in the delay history X _tt (τ, η) after the coarse compensation is estimated.
There are various methods for estimating the phase change. For example, after extracting the phase change of the reference signal Y (η), there is a method of connecting the phase change by unwrapping the return of the phase every 2π. Conceivable.
Further, a method of smoothing by applying a least square fitting to the unwrapped phases can be considered.
In addition, the reference signal Y (η) is subjected to a short-time Fourier transform with an appropriate slow time width to obtain a change in the Doppler frequency distribution with respect to the slow time η, and the change is well-known instantaneous Doppler frequency change and phase A method of converting to a phase based on the relationship of change is conceivable.
In addition, “various methods proposed for estimating phase change” used in SAR / ISAR phase compensation, fluctuation compensation autofocus, and the like may be used.

When the phase change estimation circuit 12 estimates the phase change from the reference signal Y (η), the propagation delay compensation amount precise estimation circuit 12 eliminates the return of the phase change every 2π and cancels the phase change after the return elimination φ _high ( η) is estimated.
As a method of estimating the phase compensation amount φ _high (η) that cancels the phase change after the aliasing is eliminated, for example, various kinds of second-order or higher-order phase compensation amounts are assumed, and each phase compensation amount is used as a reference signal Y Compensation of (η) is performed, and the Doppler distribution is generated by Fourier transforming each compensation result. There is a method of adopting the phase compensation amount related to the Doppler distribution having the maximum peak among the Doppler distributions as the estimation result of the second or higher order phase compensation amount.
This estimation method is particularly useful when the noise or clutter level is high and it is difficult to directly obtain the trace of the phase change from the reference signal Y (η).

Propagation delay compensation amount fine estimation circuit 12, the estimated amount of phase compensation phi _{high (eta),} as shown in the following equation (17), 2 of the propagation delay time corresponding to the phase compensation amount phi _{high (eta)} A compensation amount q _high (η) for canceling the change after the next is calculated.

Further, the propagation delay compensation amount precise estimation circuit 12 calculates the primary change of the propagation delay time from the peak position F _peak [Hz] of the Doppler frequency in the Doppler distribution where the peak is maximum, as shown in the following equation (18). A compensation amount q _1in (η) for compensating for is calculated.

The propagation delay compensation amount precise estimation circuit 12 includes a compensation amount q _high (η) for canceling the second and higher order changes in the propagation delay time and a compensation amount q _1in (η for compensating the first order change in the propagation delay time. ) Is calculated, the sum of these compensation amounts is calculated as a propagation delay compensation amount q _{high + 1in} (η) that cancels the propagation delay change, as shown in the following equation (19).
q _{high + 1in} (η) = q _high (η) + q _1in (η) (19)

When the propagation delay compensation amount precise estimation circuit 12 estimates the propagation delay compensation amount q _{high + 1in} (η), the propagation delay change compensation circuit 13 uses the propagation delay compensation amount q _{high + 1in} (η) to use the primary change compensation circuit 10. The change in the propagation delay time remaining in the delay history X _tt (τ, η) after the first-order change compensation (after the first-order coarse compensation) is compensated. That is, the second or higher-order change in the propagation delay time and the first-order change in the propagation delay time that cannot be compensated for by the primary change compensation circuit 10 are compensated.
The propagation delay compensation amount q _{high + 1in} (η) used by the propagation delay change compensation circuit 13 for the compensation process of the delay history X _tt (τ, η) is the same as the preprocessing propagation delay compensation circuit 4 of the delay history G _tt (τ, η). Although it differs from the compensation amount q _pre (η) used for the compensation process, the compensation process itself is the same as that of the pre-process propagation delay compensation circuit 4, and therefore the delay history X _tt (τ, η) by the propagation delay change compensation circuit 13. The detailed description of the compensation processing in () is omitted.
The delay history Z _tt (τ, η) after propagation delay compensation (after fine compensation) by the propagation delay change compensation circuit 13 is output to the imaging circuit 14.

Upon receiving the delay history Z _tt (τ, η) after propagation delay compensation (after fine compensation) by the propagation delay change compensation circuit 13, the imaging circuit 14 uses the delay history Z _tt (τ, η) as a slow time η. Radar image is obtained by FT in the direction.
In the processing up to this point, the migration on the fast time τ under observation and the change in the Doppler frequency γ have already been eliminated, so that the signals accumulate coherently, and as a result, the target detection performance is improved. This is equivalent to obtaining an image formed by eliminating the blurring factor from the viewpoint of the radar image.

  As apparent from the above, according to the first embodiment, the unnecessary signal reduction circuit 7 divides the delay history after KT compensation by the KT compensation circuit 6 into two by the fast frequency ρ, and each of the divided delay histories is divided. The delay Doppler distribution is generated by performing FT in the slow time η direction, and the complex conjugate of one delay Doppler distribution is multiplied by the other delay Doppler distribution. Can be reduced. For this reason, even when a second-order or higher-order change in the propagation delay time occurs, there is an effect that it is possible to prevent the target detection performance from being deteriorated.

  Also, according to the first embodiment, the upsampling circuit 5 converts the delay history compensated by the preprocessing propagation delay compensation circuit 4 in the fast time τ direction to the delay spectrum history obtained in the slow time η direction. Therefore, even with respect to the target where the Doppler frequency γ is turned back, the primary change in the propagation delay time of the radio wave with respect to the slow time η can be correctly compensated.

In the first embodiment, a radar apparatus that obtains a radar image by FT delay history Z _tt (τ, η) after propagation delay compensation (after fine compensation) by propagation delay change compensation circuit 13 in the slow time η direction. As described above, depending on the purpose of the radar apparatus, it is not necessary to obtain a radar image. For example, it may be sufficient to obtain the position on the fast time τ where the target exists and the target Doppler frequency γ.
In such a case, the primary change compensation amount calculation circuit 9, the primary change compensation circuit 10, the reference signal extraction circuit 11, the propagation delay compensation amount precise estimation circuit 12, the propagation delay change compensation circuit 13 and the imaging circuit 14 are provided. It may be omitted.
If it can be expected that the Doppler frequency γ will not be folded, the upsampling circuit 5 can be omitted. When preprocessing compensation is not required, the preprocessing propagation delay compensation amount estimation circuit 3 and the preprocessing propagation delay compensation circuit 4 can be omitted.

ここで得られるのは、前処理補償後に残存するラジアル速度であり、真のラジアル速度については、前処理補償時に補償した伝搬遅延時間のうち、伝搬遅延時間の１次変化に相当するラジアル速度を加える必要がある。即ち、前処理伝搬遅延補償量推定回路３により推定された伝搬遅延補償量ｑ_ｐｒｅ（η）から換算されたラジアル速度を加える必要がある。目標のラジアル速度は目標の追尾等にも有用な情報となる。 In the first embodiment, the propagation delay Doppler specifying circuit 8 shows a circuit that specifies the true image Doppler frequency _Fdop . However, by effectively utilizing the property that the aliasing of the Doppler frequency γ is removed. The Doppler frequency _Fdop may be converted into the radial velocity v _rad [m / s] by the following equation (20).

What is obtained here is the radial speed remaining after the pre-processing compensation. For the true radial speed, the radial speed corresponding to the primary change of the propagation delay time out of the propagation delay time compensated during the pre-processing compensation is obtained. Need to add. That is, it is necessary to add a radial speed converted from the propagation delay compensation amount q _pre (η) estimated by the preprocessing propagation delay compensation amount estimation circuit 3. The target radial speed is useful information for tracking the target.

In the first embodiment, the propagation delay compensation amount precise estimation circuit 12 calculates the compensation amount q _high (η) for canceling the second-order or more change in the propagation delay time. The sign of the quantity q _high (η) may be inverted to change the propagation delay time to a higher order change, and the higher order change may be converted into acceleration. That is, a secondary component is extracted from a high-order change in the propagation delay time, and the acceleration is obtained by multiplying the secondary component by c / 2. However, if acceleration is also compensated at the time of preprocessing compensation in the same way as radial velocity, it is necessary to add acceleration corresponding to a secondary change in propagation delay time. That is, it is necessary to add an acceleration converted from the propagation delay compensation amount q _pre (η) estimated by the preprocessing propagation delay compensation amount estimation circuit 3.
In an application that continuously detects and images a target while shifting the observation time little by little, a configuration in which the target radial velocity and acceleration are fed back to the relative motion specifying circuit 2 is useful.

In such a configuration, the relative motion specifying circuit 2 estimates the radial speed converted from the Doppler frequency _Fdop specified by the propagation delay Doppler specifying circuit 8 as the target true radial speed and the preprocessing propagation delay compensation amount estimation. The sum of the radial velocity converted from the propagation delay compensation amount q _pre (η) estimated by the circuit 3 is calculated, and the compensation amount estimated by the propagation delay compensation amount precise estimation circuit 12 as the target true acceleration. The sum of the acceleration converted from q _high (η) and the acceleration converted from the propagation delay compensation amount q _pre (η) estimated by the preprocessing propagation delay compensation amount estimation circuit 3 is calculated, and the target true Using the radial velocity and acceleration, the relative motion with the target is specified, and the change in the propagation delay time of the radio wave with respect to the slow time η is estimated from the relative motion. It may be.

  By adopting the above configuration, it is not necessary to estimate the amount of compensation, and thus compensation for migration on the fast time axis of a received signal in a situation where it is difficult to estimate the amount of compensation, for example, a situation where there is a lot of noise or clutter. The KT processing useful for the above is applied to the target where the Doppler frequency γ is turned up by upsampling, and further, the Doppler frequency γ of the target signal is affected by the higher-order change in the propagation delay time such as acceleration of relative motion. As a result, on the delay Doppler distribution after the migration compensation by the above KT processing, the detection performance deterioration that occurs when the target image at the true Doppler position before the aliasing removal blurs in the Doppler axis direction, and thereafter Unnecessary signal reduction circuit eliminates more precise estimation of compensation amount due to failure to extract reference data where target signal exists in stage The effect of 7 can be avoided or reduced.

Further, the unnecessary signal reduction circuit 7 is omitted, and the target propagation delay time and Doppler frequency are specified directly by the processing of the propagation delay Doppler specifying circuit 8 for the delay Doppler distribution corresponding to the output of the KT compensation circuit 6. Assuming a process with a simple configuration, by adding an unnecessary signal reduction circuit 7 to this process, an effect of reducing false images and other unnecessary signals (noise, clutter) in addition to the true image can be newly obtained. As a result, it is expected that the detection performance of the true image is improved and the discrimination performance of the true image from the true image and the false image is improved.
In addition, when it is expected that the Doppler frequency γ is not aliased, the target image on the delayed Doppler distribution can be obtained after the KT processing without the upsampling circuit 5. Since it becomes an image, the unnecessary signal suppression effect by the unnecessary signal reduction circuit 7 can be obtained similarly.

  Further, in a situation where it is difficult to detect a signal even after migration compensation due to the influence of noise and clutter, the position on the fast time where the target signal exists and the position on the Doppler frequency can be specified. It is possible to identify the fast time including the target signal after compensation and the extraction of the reference data string including the target signal necessary for estimating the precise compensation amount of the propagation delay time. There is an advantage that can be made possible. As a result, it is possible to estimate the amount of compensation based on the magnitude of the peak of the Doppler distribution obtained by applying various assumed compensation amounts to the reference data sequence. There is also an advantage that the accuracy of compensation amount estimation is improved.

Embodiment 2. FIG.
In the first embodiment, the KT compensation circuit 6 has a sampling interval corresponding to each fast frequency ρ of the delay spectrum history U _ft (ρ, η) up-sampled in the slow time η direction by the up-sampling circuit 5. By re-sampling the signal on the delay spectrum history U _ft (ρ, η) in the direction of the slow time η, the change in the propagation delay time of the radio wave with respect to the slow time η in the delay history U _tt (τ, η) is compensated Is explained.

In the second embodiment, the KT compensation circuit 6 is used to expand signals in the slow time η direction and the Doppler frequency γ direction that are generated during the process of resampling the signal on the delay spectrum history U _ft (ρ, η). to deal with both ends of the delay spectrum history U _{ft ([rho,} eta) size of the margin which corresponds to the expansion width of the signal on the delay (area value 0) spectrum history U _{ft ([rho,} eta) on the signal Then, resampling of the signal is performed.
Note that the signal width of the signal on the delay spectrum history U _ft (ρ, η) is enlarged during the resampling process because the scaling theory (SP: Scaling Principle) disclosed in Non-Patent Document 2 above. This is a peculiar phenomenon when resampling using. However, even when resampling using SP, if a correct margin is set, the signal on the delayed spectrum history U _ft (ρ, η) returns to the original signal width after the resampling process.

FIG. 7 is a block diagram showing a KT compensation circuit 6 of an image radar apparatus according to Embodiment 2 of the present invention.
In FIG. 7, the margin-minimized secondary phase coefficient determination circuit 41 has a limit slow time width (a range in which no folding signal is generated) even when the received signal scaling conversion circuit 45 performs delay history compensation processing. The maximum slow time width) and the Doppler frequency width (the maximum Doppler frequency width in the range where no aliasing signal is generated) and the ratio band that is the ratio of the transmission bandwidth to the center frequency of the radio wave. This is a circuit for determining a secondary phase coefficient b to be determined.
The secondary phase signal parameter setting circuit 42 uses the secondary phase coefficient b determined by the margin-minimized secondary phase coefficient determination circuit 41 to quadratic the change in the slow time η direction and the Doppler frequency γ direction of the radio wave. This is a circuit for setting the slow time widths w ₁ , w ₂ , w ₃ , w ₄ of the secondary phase signal having a phase change of.

The minimum margin adding circuit 43 sets a margin size according to the secondary phase coefficient b determined by the margin minimizing secondary phase coefficient determining circuit 41, and adds the margin having the size to both ends of the signal on the delay spectrum history. Circuit.
That is, the minimum margin adding circuit 43 calculates the Doppler frequency width W ₃ ^(max) of the secondary phase signal using the secondary phase coefficient b determined by the margin minimizing secondary phase coefficient determining circuit 41, and the delay spectrum. The margin of size (w ₂ −d _s ) is set on the delay spectrum history so that the slow time width of the signal on the history matches the slow time width w ₂ set by the secondary phase signal parameter setting circuit 42. The signal is added to both ends of the slow time of the signal, and then the signal after the margin is added is FT in the slow time direction, and the Doppler frequency width of the signal after FT coincides with the Doppler frequency width W ₃ ^{(max) of the} secondary phase signal. As described above, a process of adding a margin of size (W ₃ ^(max) −D _s ) to both ends of the Doppler frequency γ of the signal after FT is performed.

The secondary phase signal generation circuit 44 is based on the slow time widths w ₁ , w ₂ , w ₃ , w ₄ of the secondary phase signal set by the secondary phase signal parameter setting circuit 42, and the slow time η and the fast frequency ρ. Is a circuit that generates four types of secondary phase signals Q ₁ , Q ₂ , Q ₃ , and Q ₄ whose sampling points are the same as the signals on the delay spectrum history to which the margin is added by the minimum margin adding circuit 43. is there. The four types of secondary phase signals Q ₁ , Q ₂ , Q ₃ , and Q ₄ are related to each other in phase change.
The received signal scaling conversion circuit 45 includes signals on the delay spectrum history to which a margin is added by the minimum margin adding circuit 43 and four types of secondary phase signals Q ₁ , Q ₂ , Q generated by the secondary phase signal generating circuit 44. ₃ , Q ₄ is a circuit that compensates for a change in the propagation delay time of the radio wave with respect to the slow time in the delay history by performing KT processing based on SP.

The margin removal circuit 46 is a circuit that removes the margin added to the signal in which the change of the propagation delay time of the radio wave with respect to the slow time η is compensated by the reception signal scaling conversion circuit 45.
In other words, the margin removal circuit 46 is the reverse operation of the minimum margin addition circuit 43, and the received signal scaling conversion circuit 45 performs FT in the direction of the slow time η in the direction in which the change in the propagation delay time of the radio wave with respect to the slow time η is compensated. From the delay Doppler distribution obtained in this way, a signal having a Doppler frequency width D _s centered on the 0 Doppler frequency is extracted, and a signal having the Doppler frequency width D _s is subjected to IFT in the Doppler frequency γ direction, This is a circuit that obtains a delay history in which the margin of both the slow time η and the Doppler frequency γ is removed by extracting a signal having a slow time width d _s centered on 0 slow time.

The received signal shaping circuit 47 is a circuit that corrects the difference in the slow time width for each fast frequency ρ caused by the delay history compensation and outputs the delay history after the slow time width correction to the unnecessary signal reduction circuit 7.
That is, the reception signal shaping circuit 47 cuts out a signal having a slow time width T used for imaging in advance from the delay history from which the margin is removed by the margin removal circuit 46, centering on the zero slow time. A signal having a slow time width T is output to the unnecessary signal reduction circuit 7.

FIG. 8 is a block diagram showing the received signal scaling conversion circuit 45 of the KT compensation circuit 6.
In FIG. 8, the slow time FT unit 51 FTs the signal G _ft (ρ, η) on the delay spectrum history to which the margin is added by the minimum margin adding circuit 43 in the slow time η direction, and the signal G _ff (after FT) The process of outputting ρ, γ) is performed.
Slow time FT unit 52 FT secondary phase signal is the secondary phase signal _{Q 1} generated by the generating circuit _{44 Q} 1ft _(ρ, η) slow time eta direction, the signal after FT _{Q 1ff} ([rho, The process of outputting γ) is performed.
The multiplication circuit 53 multiplies the signal G _ff (ρ, γ) output from the slow time FT unit 51 by the secondary phase signal Q _1ff (ρ, γ) output from the slow time FT unit 52. carry out.

The slow time IFT unit 54 performs an IFT on the multiplication result G _ff (ρ, γ) × Q _1ff (ρ, γ) of the multiplication circuit 53 in the Doppler frequency γ direction, and a signal X _1ft (ρ, η) after the IFT (delay spectrum) A result of convolution operation G _ft (ρ, η) * Q _{1 ft} (ρ, η) of the _historical signal G _ft (ρ, η) and the secondary phase signal Q _1ft (ρ, η) is output. Perform the process.
Multiplier circuit 55 is a slow time IFT unit for the signal _{X 1 ft} after IFT output from 54 (ρ, η), 2-order phase signal generated by the generation circuit 44 secondary phase signal _{Q 2 Q 2 ft} ( A process of multiplying (ρ, η) is performed.

The slow time FT unit 56 FTs the multiplication result X _2ft (ρ, η) = X _1ft (ρ, η) × Q _2ft (ρ, η) of the multiplication circuit 55 in the slow time η direction, and a signal X _2ff after the FT A process of outputting (ρ, γ) is performed.
Slow time FT section 57 is FT which is a secondary phase signal _{Q 3} generated by the secondary phase signal generating circuit 44 _{Q 3ft (ρ,} η) slow time eta direction, the signal after FT _{Q 3ff} ([rho, The process of outputting γ) is performed.
The multiplication circuit 58 multiplies the post-FT signal X _2ff (ρ, γ) output from the slow time FT unit 56 by the secondary phase signal Q _3ff (ρ, γ) output from the slow time FT unit 57. Perform the process.

The slow time IFT unit 59 performs an IFT on the multiplication result X _2ff (ρ, γ) × Q _3ff (ρ, γ) of the multiplication circuit 58 in the Doppler frequency γ direction, and a signal X _3ft (ρ, η) after the IFT (multiplication circuit) 55 corresponding to a convolution operation result X _2ft (ρ, η) * Q _3ft (ρ, η) of the multiplication result X _2ft (ρ, η) of 55 and the secondary phase signal Q _3ft (ρ, η). Perform the process.
The multiplication circuit 60 outputs the post-IFT signal X _3ft (ρ, η) output from the slow time IFT unit 59 to the secondary phase signal Q ₄ generated by the secondary phase signal generation circuit 44, Q _4ft ( ρ, η) is multiplied, and the multiplication result X _3ft (ρ, η) × Q _4ft (ρ, η) is output as U _ft ^(margin) (ρ, η).

Next, the operation will be described.
Since the processing contents other than the KT compensation circuit 6 are the same as those in the first embodiment, only the processing contents of the KT compensation circuit 6 will be described here.

FIG. 9 is an explanatory diagram showing merge addition processing by the KT compensation circuit 6.
The horizontal axis in the upper left original data is the slow time axis or Doppler frequency axis of the input signal.
Since the aliasing problem occurs in both the slow time axis and the Doppler frequency axis, here, these are collectively referred to as the slow axis and will be described in a unified manner.
First, consider the case where the SP process is simply applied to the KT process.
At a certain stage of the processing process, the slow axis widens, so that a signal exceeding the original width (slow time width d _s or Doppler frequency width D _s ) is generated, and the signal exceeding the original width is The folded signal is superimposed as shown on the left side of FIG.

On the other hand, as shown on the right side of FIG. 9, if a margin of a sufficient slow axis width (a margin of a size corresponding to an enlarged width exceeding the original width) is added, a certain stage of the processing process Thus, even if the slow axis is widened, the folding signal is not generated, so that the folding of the folding signal can be avoided.
However, as the margin width is increased, the processing load increases as the data capacity increases.
Therefore, from the viewpoint of reducing the processing load, it is desirable to set the margin size to the minimum necessary size.

For this reason, the margin-minimized secondary phase coefficient determination circuit 41 sets the secondary phase coefficient b so that the margin size becomes the minimum necessary size.
That is, since the spread width of the signal varies depending on the secondary phase coefficient b and the necessary margin also varies depending on the secondary phase coefficient b, the margin-minimized secondary phase coefficient determination circuit 41 performs the signal spread in each processing stage. After estimating the characteristics of the width and the required margin, the secondary phase coefficient b is set so that the margin size becomes the minimum necessary size.

That is, the margin-minimized secondary phase coefficient determination circuit 41 does not generate a return signal even when the received signal scaling conversion circuit 45 performs the resampling process on the signal on the delay spectrum history G _ft (ρ, η). In order to obtain a margin size necessary and sufficient to achieve the limit, the limit slow time width d _s , the limit Doppler frequency width D _s, and the ratio band ξ (= B / F _c ) are expressed by the following equations ( 21), the optimum design value b ^(opt) of the secondary phase coefficient b is calculated.

式（２２）〜（２５）において、β_１，β_２，β_３は１と等しいか、１より僅かに大きな定数である。 When the margin-minimized secondary phase coefficient determination circuit 41 calculates the optimal design value b ^(opt) of the secondary phase coefficient b, the secondary phase signal parameter setting circuit 42 determines the optimal design value b of the secondary phase coefficient b. ^By substituting ^(opt) into the following equations (22) to (25), the optimum value w _k ^(opt) of the slow time width of the secondary phase signal Q _kft (ρ, η ⁾ is calculated. k = 1, 2, 3, 4.

In the expressions (22) to (25), β ₁ , β ₂ , and β ₃ are constants that are equal to or slightly larger than 1.

最小マージン付加回路４３は、ドップラー周波数幅の最適値Ｗ_３ ^{（ｍａｘ，ｏｐｔ）}を算出すると、アップサンプリング回路５によりアップサンプリングされた遅延スペクトルヒストリＧ_ｆｔ（ρ，η）に対して、スロータイム幅が最適値ｗ_２ ^{（ｏｐｔ）}になり、ドップラー周波数幅が最適値Ｗ_３ ^{（ｍａｘ，ｏｐｔ）}になるように、値が０の領域であるマージンを各軸上で加える処理を行う。
具体的には、以下のようにマージンを付加する。 Minimum margin adding circuit 43, when the margin minimize secondary phase coefficient determining circuit 41 calculates the optimum design value b ^(opt) of the secondary phase coefficient b, the optimum design value b of the secondary phase coefficient b ^(opt) Is substituted into the following equation (26 ⁾ to calculate the optimum value W ₃ ^{(max, opt)} of the Doppler frequency width.

When the minimum margin adding circuit 43 calculates the optimum value W ₃ ^{(max, opt)} of the Doppler frequency width, the slow margin width 43 with respect to the delay spectrum history G _ft (ρ, η) up-sampled by the up-sampling circuit 5 Is set to the optimum value w ₂ ^(opt) , and a margin of a value 0 region is added on each axis so that the Doppler frequency width becomes the optimum value W ₃ ^{(max, opt)} .
Specifically, a margin is added as follows.

First, the minimum margin adding circuit 43 has an insufficient slow time width (w at both ends of the delay spectrum history G _ft (ρ, η) so that the slow time width matches the optimum value w ₂ ^(opt). ₂ ^(opt) −d _s ) is added.
Next, the minimum margin addition circuit 43 FTs the delay spectrum history G _ft (ρ, η) after the margin addition in the slow time η direction, and the delay spectrum Doppler distribution G _ff (ρ, ρ) which is the delay spectrum history after FT. The missing Doppler frequency width at both ends of the delayed spectral Doppler distribution G _ff (ρ, γ) so that the Doppler frequency width of γ) matches the Doppler frequency width W ₃ ^{(max, opt) of the} secondary phase signal. A margin having the same size as (W ₃ ^{(max, opt)} −D _s ) is added.

Here, a margin of the same size as the insufficient slow time width (w ₂ ^(opt) −d _s ) is added to both ends of the delay spectrum history G _ft (ρ, η), and then the delay spectrum Doppler distribution G _Although an example is shown in which margins having the same size as the missing Doppler frequency width (W ₃ ^{(max, opt)} −D _s ) are added to both ends of _ff (ρ, γ), the slow time width is w _2. while ensuring an empty data area ^(opt), the center delay spectrum history G _{ft (ρ,} η) of the data area after placing, the delay spectrum history G _{ft (ρ,} η) and slow time FT in the η direction, and delay spectrum Doppler G _ff (ρ, γ) is arranged at the center of an empty data area having a separately secured Doppler frequency width of W ₃ ^{(max, opt)} . You may make it do.

The secondary phase signal generation circuit 44 determines the slow time η and the fast frequency ρ based on the optimum values w ₁ ^{(opt) to} w ₄ ^(opt) of the slow time width calculated by the secondary phase signal parameter setting circuit 42. Are the same as the delay spectrum history G _ft (ρ, η) to which the margin is added by the minimum margin adding circuit 43, and the four kinds of secondary phase signals Q _1ft (ρ, η), Q _2ft (ρ, η), Q _3ft (ρ, η), Q _4ft (ρ, η) are generated.
That is, the secondary phase signal generation circuit 44 sets the optimum values w ₁ ^{(opt) to} w ₄ ^(opt) of the slow time width calculated by the secondary phase signal parameter setting circuit 42 to the following equations (27) to (30 ) To generate secondary phase signals Q _1ft (ρ, η), Q _2ft (ρ, η), Q _3ft (ρ, η), and Q _4ft (ρ, η). At this time, the slow time width and the step width are determined by the minimum margin adding circuit 43.
In equation (27) - _(30), simplified to are denoted as ^w 1 ^{_(opt) to w 4} a ^(opt) _w 1 ~w _4. Further, the optimum design value b ^(opt) of the secondary phase coefficient b is ^simply expressed as b.

The received signal scaling conversion circuit 45 includes a delay spectrum history G _ft (ρ, η) to which a margin is added by the minimum margin addition circuit 43 and four types of secondary phase signals Q _1ft generated by the secondary phase signal generation circuit 44. By performing KT processing based on SP using (ρ, η), Q _2ft (ρ, η), Q _3ft (ρ, η), and Q _4ft (ρ, η), the distance between the target and the radar The primary change of the propagation delay time of the radio wave with respect to the slow time η generated due to the relative motion of is compensated.
Hereinafter, the processing content of the received signal scaling conversion circuit 45 will be specifically described.

When the slow time FT unit 51 of the received signal scaling conversion circuit 45 receives the delay spectrum history G _ft (ρ, η) with a margin added from the minimum margin addition circuit 43, the delay spectrum history G _ft (ρ, η) Is delayed in the slow time η direction, and the delayed spectrum Doppler distribution G _ff (ρ, γ), which is the delayed spectrum history after FT, is output to the multiplier circuit 53.
When receiving the secondary phase signal Q _1ft (ρ, η) from the secondary phase signal generation circuit 44, the slow time FT unit 52 performs an FT on the secondary phase signal Q _1ft (ρ, η) in the slow time η direction. , The post-FT secondary phase signal Q _1ff (ρ, γ) is output to the multiplier circuit 53.

The multiplication circuit 53 uses the delay spectrum Doppler distribution G _ff (ρ, γ) output from the slow time FT unit 51 and the secondary phase signal Q _1ff (ρ, γ) output from the slow time FT unit 52 as frequencies. Multiplication is performed on the axis, and the multiplication result G _ff (ρ, γ) × Q _1ff (ρ, γ) is output to the slow time IFT unit 54.
Slow time IFT unit 54, the multiplication result of the multiplier circuit _{53 G ff (ρ, γ)} × Q 1ff (ρ, γ) and IFT of the Doppler frequency gamma direction, the IFT results at which _{X 1ft (ρ, η)} The result is output to the multiplication circuit 55. This X _1ft (ρ, η) is a result of convolution operation G _ft (ρ, η) * Q _1ft (ρ, η) of the delayed spectrum history G _ft (ρ, η) and the secondary phase signal Q _1ft (ρ, η). ).

Multiplier circuit 55, _{X 1ft (ρ, η)} is IFT results from the slow time IFT section 54 receives the, the IFT is the result _{X 1ft (ρ, η)} and output from the secondary phase signal generation circuit 44 The secondary phase signal Q _2ft (ρ, η) is multiplied on the time axis, and the multiplication result X _2ft (ρ, η) = X _1ft (ρ, η) × Q _2ft (ρ, η) is the slow time FT. To the unit 56.
The slow time FT unit 56 performs the FT on the multiplication result X _2ft (ρ, η) of the multiplication circuit 55 in the slow time η direction, and outputs the X _2ff (ρ, γ) as the FT result to the multiplication circuit 58.

When the slow time FT unit 57 receives the secondary phase signal Q _3ft (ρ, η) from the secondary phase signal generation circuit 44, the slow time FT unit FT performs the FT on the secondary phase signal Q _3ft (ρ, η) in the slow time η direction. , The post-FT secondary phase signal Q _3ff (ρ, γ) is output to the multiplier circuit 58.
The multiplication circuit 58 uses the FT result X _2ff (ρ, γ) output from the slow time FT unit 56 and the secondary phase signal Q _3ff (ρ, γ) output from the slow time FT unit 57 on the frequency axis. The multiplication result X _2ff (ρ, γ) × Q _3ff (ρ, γ) is output to the slow time IFT unit 59.

The slow time IFT unit 59 performs an IFT on the multiplication result X _2ff (ρ, γ) × Q _3ff (ρ, γ) of the multiplication circuit 58 in the Doppler frequency γ direction, and X _3ft (ρ, η) as the IFT result is obtained. The result is output to the multiplication circuit 60. This X _3ft (ρ, η) is the result of convolution operation X _2ft (ρ, η) * Q _3ft () of the multiplication result X _2ft (ρ, η) of the multiplication circuit 55 and the secondary phase signal Q _3ft (ρ, η). corresponds to ρ, η).
Multiplier circuit 60, _{X 3ft (ρ, η)} is IFT results from the slow time IFT section 59 receives the, its _{X 3ft (ρ, η)} and the secondary phase signal output from the secondary phase signal generation circuit 44 Q _4ft (ρ, η) is multiplied on the time axis, and the multiplication result X _3ft (ρ, η) × Q _4ft (ρ, η) is used as a delay spectrum history U _ft ^(margin) (ρ, η) and margin Output to the removal circuit 46.
Although the signal width is increased by the processing of the multiplication circuits 53 and 55, the correct margin is set, so that the signal width is restored to the original width by the processing of the multiplication circuits 58 and 60.

When the margin removal circuit 46 receives the delay spectrum history U _ft ^(margin) (ρ, η) in which the primary change of the propagation delay time of the radio wave with respect to the slow time η is compensated from the reception signal scaling conversion circuit 45, the delay The ^margin added to the spectrum history U _ft ^(margin) (ρ, η) is removed.
That is, the margin removal circuit 46 is the reverse operation of the minimum margin addition circuit 43, and first, a delayed Doppler distribution obtained by FT delay spectrum history U _ft ^(margin) (ρ, η) in the slow time η direction. Then, a signal having a Doppler frequency width D _s centered on the 0 Doppler frequency is extracted.
Next, the margin removal circuit 46 extracts a signal having a slow time width d _s centered on 0 slow time from a delay spectrum history obtained by performing an IFT on a signal having a Doppler frequency width D _s , so that the slow time η And a delay spectrum history from which the margin of both the Doppler frequency γ is removed.

The received signal shaping circuit 47 corrects the difference in the slow time width for each fast frequency ρ, which is caused by the KT processing based on the SP in the KT compensation circuit 6 (compensation for the change in the propagation delay time of the radio wave with respect to the slow time η). The delay history corrected for the difference in the slow time width is output to the unnecessary signal reduction circuit 7.
That is, the reception signal shaping circuit 47 extracts a signal having a slow time width T [s] used for imaging in advance from the delay history from which the margin has been removed by the margin removal circuit 46, centering on the 0 slow time. Then, the extracted signal of the slow time width T is output to the unnecessary signal reduction circuit 7. Alternatively, in the delay history from which the margin is removed by the margin removal circuit 46, weighting is performed so that the value outside the range of the slow time width T is set to zero with the zero slow time as the center.
In the expression in which the slow time is discretized, data corresponding to the number of samples (number of pulses) H used for imaging is finally cut out, and the delay spectrum history U _ft after shaping like the following equation (31) is obtained. ^(Last) (ρ, h) is obtained.

In the delay spectrum history U _ft ^(last) (ρ, h) obtained through the processing of each block described above, the primary change in the propagation delay time of the radio wave with respect to the slow time η is eliminated by the effect of the KT processing. . Further, the difference in the slow time width for each fast frequency ρ caused by the adverse effect of applying SP is also eliminated.

Embodiment 3 FIG.
In the first and second embodiments, the propagation delay compensation amount precise estimation circuit 12 estimates the propagation delay compensation amount q _{high + 1in} (η) that cancels the propagation delay change, and the propagation delay change compensation circuit 13 performs the propagation delay compensation amount precise estimation circuit. 12, using the propagation delay compensation amount q _{high + 1 in} (η) estimated by 12, it remains in the delay history X _tt (τ, η) after the primary change compensation by the primary change compensation circuit 10 (after the primary coarse compensation). Although the compensation for the change in the propagation delay time is shown, the influence of the change in the propagation delay time remaining in the delay history X _tt (τ, η) after the primary change compensation (after the primary coarse compensation) is shown. Is estimated, and a phase compensation amount for canceling the phase change is estimated. Using the phase compensation amount, a delay history X _tt (τ, τ, τ) after primary change compensation (after primary coarse compensation) is estimated. η) to compensate for the phase It may be.

FIG. 10 is a block diagram showing a radar apparatus according to Embodiment 3 of the present invention. In FIG. 10, the same reference numerals as those in FIG.
The phase compensation amount estimation circuit 15 uses the delay history X _tt (τ) after the primary change compensation (after the primary coarse compensation) by the primary change compensation circuit 10 from the reference signal Y (η) extracted by the reference signal extraction circuit 11. , Η) is a circuit that estimates a phase change that occurs due to the effect of a change in propagation delay time remaining, and estimates a phase compensation amount that cancels the phase change.
The phase compensation circuit 16 uses the phase compensation amount estimated by the phase compensation amount estimation circuit 15 to use the delay history X _tt (τ, η) after the primary change compensation (after the primary coarse compensation) by the primary change compensation circuit 10. ).

In the example of FIG. 10, the signal acquisition circuit 1, the relative motion identification circuit 2, the preprocessing propagation delay compensation amount estimation circuit 3, the preprocessing propagation delay compensation circuit 4, the upsampling circuit 5, and the KT compensation circuit that are components of the radar apparatus. 6, unnecessary signal reduction circuit 7, propagation delay Doppler identification circuit 8, primary change compensation amount calculation circuit 9, primary change compensation circuit 10, reference signal extraction circuit 11, phase compensation amount estimation circuit 15, phase compensation circuit 16, and image It is assumed that each of the digitizing circuits 14 is configured by dedicated hardware. As dedicated hardware, for example, a semiconductor integrated circuit on which a CPU is mounted, a one-chip microcomputer, a semiconductor integrated circuit on which a GPU is mounted, and the like can be considered.
However, the radar apparatus of the third embodiment may be configured with a computer.
When the radar apparatus is configured by a computer, the signal acquisition circuit 1, the relative motion specifying circuit 2, the preprocessing propagation delay compensation amount estimation circuit 3, the preprocessing propagation delay compensation circuit 4, the upsampling circuit 5, the KT compensation circuit 6, and unnecessary. Signal reduction circuit 7, propagation delay Doppler identification circuit 8, primary change compensation amount calculation circuit 9, primary change compensation circuit 10, reference signal extraction circuit 11, phase compensation amount estimation circuit 15, phase compensation circuit 16, and imaging circuit 14 2 may be stored in the memory 21 of the computer shown in FIG. 2, and the processor 22 of the computer may execute the program stored in the memory 21.

Next, the operation will be described.
Since this embodiment is the same as Embodiments 1 and 2 except that a phase compensation amount estimation circuit 15 and a phase compensation circuit 16 are mounted instead of the propagation delay compensation amount precise estimation circuit 12 and the propagation delay change compensation circuit 13. Only the processing contents of the phase compensation amount estimation circuit 15 and the phase compensation circuit 16 will be described.

When the reference signal extraction circuit 11 extracts the reference signal Y (η), the phase compensation amount estimation circuit 15 performs primary change compensation (primary coarse compensation) from the reference signal Y (η) by the primary change compensation circuit 10. The phase change θ _high (η) generated by the influence of the change in the propagation delay time remaining in the later) delay history X _tt (τ, η) is estimated.
The method for estimating the phase change θ _high (η) is the same as the method for estimating the phase change by the propagation delay compensation amount precise estimation circuit 12 of FIG. 1, but the phase compensation amount estimation circuit 15 uses the propagation delay compensation amount precise estimation. Unlike the circuit 12, the phase change θ _high (η) is allowed to be turned back every 2π.
When estimating the phase change θ _high (η) from the reference signal Y (η), the phase compensation amount estimating circuit 15 inverts the sign of the phase change θ _high (η) as shown in the following equation (32). _Is calculated as a phase compensation amount φ _highcmp (η).

Phase compensating circuit 16, the phase compensation quantity estimator circuit 15 calculates a phase compensation amount _φ highcmp (η), as shown in the following equation (33), using the phase compensation amount _φ highcmp (η), 1 The phase of the delay history X _tt (τ, η) after the primary change compensation (after the primary coarse compensation) by the secondary change compensation circuit 10 is compensated, and the delay history Z _tt (τ, η) after phase compensation is imaged Output to the circuit 14.

According to the third embodiment, since the phase compensation amount estimation circuit 15 and the phase compensation circuit 16 are mounted instead of the propagation delay compensation amount precise estimation circuit 12 and the propagation delay change compensation circuit 13, the above embodiment is described. In addition to the same effects as 1 and 2, the following unique effects are also obtained.
In the third embodiment, since the compensation of the propagation delay time is changed to mere phase compensation, an effect of simplifying the processing can be obtained. In addition, since the phase change θ _high (η) is allowed to be turned back every 2π, a general phase estimation method that estimates the phase in a form that allows the turn back every 2π can be applied. An effect is obtained.

  In the present invention, within the scope of the invention, any combination of the embodiments, or any modification of any component in each embodiment, or omission of any component in each embodiment is possible. .

  1 signal acquisition circuit, 2 relative motion identification circuit, 3 preprocessing propagation delay compensation amount estimation circuit, 4 preprocessing propagation delay compensation circuit, 5 upsampling circuit, 6 KT compensation circuit (time variation compensation circuit), 7 unnecessary signal reduction circuit , 8 Propagation delay Doppler identification circuit, 9 Primary change compensation amount calculation circuit, 10 Primary change compensation circuit, 11 Reference signal extraction circuit, 12 Propagation delay compensation amount precise estimation circuit, 13 Propagation delay change compensation circuit, 14 Imaging circuit , 15 phase compensation amount estimation circuit, 16 phase compensation circuit, 21 memory, 22 processor, 31 delay Doppler distribution generation circuit, 32 delay Doppler distribution conjugate multiplication circuit, 33 Doppler axis inverse Fourier transform circuit, 34 Doppler width calculation circuit, 35 weight Function multiplication circuit, 36 Slow time axis Fourier transform circuit, 41 Margin minimized secondary phase Number determination circuit, 42 secondary phase signal parameter setting circuit, 43 minimum margin addition circuit, 44 secondary phase signal generation circuit, 45 reception signal scaling conversion circuit, 46 margin removal circuit, 47 reception signal shaping circuit, 51 slow time FT section , 52 slow time FT section, 53 multiplication circuit, 54 slow time IFT section, 55 multiplication circuit, 56 slow time FT section, 57 slow time FT section, 58 multiplication circuit, 59 slow time IFT section, 60 multiplication circuit.

Claims (14)

From a radar that repeatedly transmits and receives radio waves while changing the relative positional relationship with the target to be observed, the received signal of the radio waves is a fast time that is an elapsed time from the transmission time of the radio waves and a transmission time of the radio waves A signal acquisition circuit for acquiring a delay history that is a two-dimensional distribution with a certain slow time;
Resampling the signal on the delay spectrum history in the slow time direction at the sampling interval corresponding to each fast frequency of the delay spectrum history obtained by Fourier transforming the delay history acquired by the signal acquisition circuit in the fast time direction A time change compensation circuit that compensates for a change in radio wave propagation delay time with respect to the slow time in the delay history;
The delay history after compensation by the time variation compensation circuit is divided into two at a fast frequency, and each of the divided delay histories is Fourier transformed in the slow time direction to generate a delay Doppler distribution, A radar apparatus comprising: an unnecessary signal reduction circuit that multiplies the complex conjugate and the other delayed Doppler distribution.   The time variation compensation circuit sets a margin of a size corresponding to the expansion width of the signal on the delay spectrum history that expands in the slow time direction and the Doppler frequency direction during the resampling process on the delay spectrum history. 2. The radar apparatus according to claim 1, wherein the signal is resampled after being added to both ends of the signal. An upsampling circuit that upsamples a delay spectrum history obtained by Fourier transforming the delay history acquired by the signal acquisition circuit in a slow time direction;
The time variation compensation circuit resamples a signal on the delay spectrum history in a slow time direction at a sampling interval corresponding to each fast frequency of the delay spectrum history upsampled by the upsampling circuit. The radar apparatus according to claim 1 or 2.   The upsampling circuit sets N as the number of upsampling according to the change in the propagation delay time of the radio wave with respect to the slow time, and (N−1) cells between the cells in the slow time direction in the delay spectrum history. 4. The radar apparatus according to claim 3, wherein the sampling frequency of the delay spectrum history is increased N times by inserting zero cells.   The upsampling circuit obtains a predicted value of a primary change in propagation delay time of the radio wave with respect to the slow time, and a product of twice the predicted value of the primary change and the center frequency of the radio wave transmitted from the radar. 5. The radar apparatus according to claim 4, wherein a value larger than a value obtained by dividing the frequency by the frequency at which the radio wave transmission is repeated is set as the upsampling score.   A propagation delay Doppler specifying circuit for detecting the target from the delayed Doppler distribution after multiplication by the unnecessary signal reduction circuit and specifying the position on the fast time where the target exists and the target Doppler frequency; The radar device according to any one of claims 1 to 5, wherein: First-order change compensation amount calculation for calculating a first-order change compensation amount for compensating for a first-order change in radio wave propagation delay time with respect to a slow time in a delay history from a target Doppler frequency specified by the propagation delay Doppler specifying circuit. Circuit,
Using the primary change compensation amount calculated by the primary change compensation amount calculation circuit, a primary change that compensates for the primary change of the propagation delay time of the radio wave with respect to the slow time in the delay history acquired by the signal acquisition circuit. A compensation circuit;
Signals of all slow times at positions on the fast time where the target specified by the propagation delay Doppler specifying circuit exists from the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit. A reference signal extraction circuit for extracting a signal as a reference signal;
Phase change caused by the influence of the change in the propagation delay time remaining in the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit from the reference signal extracted by the reference signal extraction circuit And estimating the propagation delay compensation amount that cancels out the propagation delay change and estimates the propagation delay compensation amount that cancels the propagation delay change. Circuit,
Using the propagation delay compensation amount estimated by the propagation delay compensation amount precise estimation circuit, a change in the propagation delay time remaining in the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit. A propagation delay change compensation circuit for compensating
The radar apparatus according to claim 6, further comprising: an imaging circuit configured to perform imaging by performing Fourier transform on a delay history after compensation by the propagation delay change compensation circuit in a slow time direction. First-order change compensation amount calculation for calculating a first-order change compensation amount for compensating for a first-order change in radio wave propagation delay time with respect to a slow time in a delay history from a target Doppler frequency specified by the propagation delay Doppler specifying circuit. Circuit,
Using the primary change compensation amount calculated by the primary change compensation amount calculation circuit, a primary change that compensates for the primary change of the propagation delay time of the radio wave with respect to the slow time in the delay history acquired by the signal acquisition circuit. A compensation circuit;
Signals of all slow times at positions on the fast time where the target specified by the propagation delay Doppler specifying circuit exists from the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit. A reference signal extraction circuit for extracting a signal as a reference signal;
Phase change caused by the influence of the change in the propagation delay time remaining in the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit from the reference signal extracted by the reference signal extraction circuit A phase compensation amount estimating circuit for estimating a phase compensation amount for canceling the phase change, and
A phase compensation circuit that compensates the phase of the delay history in which the primary change of the propagation delay time is compensated by the primary change compensation circuit, using the phase compensation amount estimated by the phase compensation amount estimation circuit;
The radar apparatus according to claim 6, further comprising: an imaging circuit configured to image the delay history after compensation by the phase compensation circuit by performing a Fourier transform in a slow time direction. The unnecessary signal reduction circuit includes:
By dividing the delay history after compensation by the time change compensation circuit into a low frequency side delay history and a high frequency side delay history at a fast frequency, and Fourier transforming the low frequency side delay history in the slow time direction. A delay that generates a low-frequency side delay Doppler distribution at the fast frequency and generates a high-frequency side delay Doppler distribution at the fast frequency by Fourier transforming the high-frequency side delay history in a slow time direction. A Doppler distribution generation circuit;
A delay Doppler distribution conjugate multiplication circuit for multiplying the complex conjugate of one delay Doppler distribution and the other delay Doppler distribution among the low-frequency side and high-frequency side delay Doppler distributions generated by the delay Doppler distribution generation circuit;
A Doppler axis inverse Fourier transform circuit that obtains a two-dimensional distribution of fast time and slow time by inverse Fourier transforming the delayed Doppler distribution after conjugate multiplication by the delayed Doppler distribution conjugate multiplier circuit in the Doppler axis direction;
Of the two-dimensional distribution obtained by the Doppler axis inverse Fourier transform circuit, a slow time other than the slow time in which the target signal exists by passing the signal in the slow time in which the target signal exists is passed. A weighting function multiplication circuit that sets a weighting function that prevents the signal from passing through and multiplies the two-dimensional distribution by the weighting function;
A slow time axis Fourier transform circuit that obtains a delayed Doppler distribution in which unnecessary signals other than the target signal are suppressed by Fourier transforming the two-dimensional distribution multiplied by the weight function by the weight function multiplying circuit in the slow time direction. The radar apparatus according to any one of claims 1 to 8, wherein the radar apparatus comprises: The unnecessary signal reduction circuit acquires a predicted value of a secondary change in the propagation delay time of the radio wave with respect to the slow time, and delay Doppler distribution after conjugate multiplication by the delayed Doppler distribution conjugate multiplier circuit from the predicted value of the secondary change A Doppler width calculation circuit for calculating a Doppler width of the target image on
The weighting function multiplication circuit sets a slow time width inversely proportional to the Doppler width calculated by the Doppler width calculation circuit, and the target signal of the two-dimensional distribution obtained by the Doppler axis inverse Fourier transform circuit is A weight function is set to pass a signal of a slow time in the slow time width that exists and to block a signal of a slow time other than the slow time in which the target signal exists. The radar device according to claim 9. The Doppler width calculating circuit center frequency F _c of the delay spectrum history obtained by Fourier transforming the delay history after compensation by the time variation compensating circuit in the fast time _direction, the bandwidth of the delay spectrum history is B, _Assuming that the predicted value of the secondary change is s _e2 , and the delay time width of the delay history after compensation by the time change compensation circuit is T, on the delay Doppler distribution after conjugate multiplication by the delay Doppler distribution conjugate multiplier circuit The radar apparatus according to claim 10, wherein a Doppler width of the target image is calculated by F _c − (B / 4) · T · se ₂ . In the delay Doppler distribution generation circuit, the center frequency of the delay spectrum history obtained by Fourier-transforming the delay history after compensation by the time change compensation circuit in the fast time direction is F _c , and the bandwidth of the delay spectrum history is B As a result, a signal having a center frequency of F _c − (B / 4) and a bandwidth of B / 2 is extracted from the delay spectrum history as a low frequency side delay spectrum history, and the center frequency is F _c +. In (B / 4), a signal having a bandwidth of B / 2 is extracted as a high-frequency side delay spectrum history, and the low-frequency side and high-frequency side delay spectrum history is subjected to inverse Fourier transform in the fast frequency direction. By performing Fourier transform in the slow time direction, low-frequency and high-frequency delayed Doppler distributions are generated at the fast frequency. The radar apparatus according to claim 9, wherein. Relative motion identification that collects motion specifications of the target from an external device, identifies relative motion with the target from the motion specifications, and estimates changes in radio wave propagation delay time relative to the slow time from the relative motion Circuit,
A pre-processing propagation delay compensation amount estimation circuit for estimating a propagation delay compensation amount that cancels a change in propagation delay time estimated by the relative motion identification circuit;
And a preprocessing propagation delay compensation circuit that compensates the delay history acquired by the signal acquisition circuit using the propagation delay compensation amount estimated by the preprocessing propagation delay compensation amount estimation circuit. The radar apparatus according to any one of claims 1 to 12. Relative motion identification that collects motion specifications of the target from an external device, identifies relative motion with the target from the motion specifications, and estimates changes in radio wave propagation delay time relative to the slow time from the relative motion Circuit,
A pre-processing propagation delay compensation amount estimation circuit for estimating a propagation delay compensation amount that cancels a change in propagation delay time estimated by the relative motion identification circuit;
Using a propagation delay compensation amount estimated by the preprocessing propagation delay compensation amount estimation circuit, and a preprocessing propagation delay compensation circuit that compensates for a delay history obtained by the signal acquisition circuit;
In the relative motion specifying circuit, the Doppler frequency is already specified by the propagation delay Doppler specifying circuit, and the propagation delay compensation amount is already estimated by the preprocessing propagation delay compensation amount estimation circuit, the Doppler frequency and the The radar apparatus according to claim 6, wherein a relative motion with respect to the target is specified using a propagation delay compensation amount, and a change in a propagation delay time of the radio wave with respect to the slow time is estimated from the relative motion.

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