JP6344421B2 - Bandpass type ΔΣ modulator, apparatus and method - Google Patents

Description

  The present invention relates to a transmitter, a receiver, a communication system, and a radio base station apparatus.

The present invention relates to a bandpass ΔΣ modulator, an apparatus, and a method .

  For example, as shown in FIG. 20, in the optical transmitter 101 of the optical fiber communication system 100, a digital electric signal to be transmitted is converted into an optical signal by an electric-optical converter (E / O; optical link) 101a. , To the optical transmission line (optical fiber) 102. The optical receiver 103 obtains a digital electric signal by converting an optical signal transmitted from the optical transmission path 102 into an electric signal by an optical-electric converter (O / E; optical link) 103a.

  In general, a laser diode is used for the electrical-optical converter 101a, and a photodiode is used for the optical-electrical converter 103a. In many cases, the input / output characteristics of these elements 101a and 103a are not sufficiently linear. Therefore, an optical fiber communication system can be used to transmit a value quantized with a stepwise value such as a binarized value, but to transmit a continuous value such as an analog signal. Not very suitable.

  In an optical fiber communication system, when a continuous value such as an analog signal is directly subjected to optical intensity modulation by a laser diode 101a and optical transmission is attempted, a special electro-optical converter 101a and optical- It is necessary to use the electrical converter 103a or correct the signal, resulting in an expensive system.

Therefore, in many cases, even when an analog signal is transmitted through an optical transmission line (optical fiber), a digital signal pulse train is transmitted through the optical transmission line (optical fiber) 102 as shown in FIG. For this purpose, in the optical transmitter 101, an analog signal to be transmitted is first converted into a digital signal by an AD converter (ADC) 101b.
The parallel digital signal output from the AD converter 101b is converted into a serial bit string, and a pulse train of an optical signal corresponding to the serial bit string is generated by the electro-optical converter 101a for optical transmission. Send to path 102.
The optical receiver 103 acquires the analog signal by converting the digital signal indicated by the received pulse train into an analog signal by the DA converter (DAC) 103b.

On the other hand, as a technique for transmitting an analog signal in an optical fiber communication system, as described above, in addition to the method of transmitting an analog signal by modulating the optical intensity, the pulse width of transmitting the analog signal by modulating the pulse width. There is a modulation method (PWM; Plus Width Modulation).
The pulse width modulation method represents the waveform of an analog signal by the pulse width. Since a pulse train is transmitted through the optical transmission path, the pulse width modulation method is less susceptible to non-linearity in the transmission system.

Takao Wabo and Akira Yasuda (original author Richard Schreier, Gabor C. Temes) Introduction to ΔΣ analog / digital converters (Understanding Delta-Sigma Data Converters), Maruzen Co., Ltd., 2007

  When an analog signal is PWM-modulated, the amplitude of the analog signal is expressed by the pulse width. Therefore, if an attempt is made to modulate with an appropriate accuracy, the sampling rate is much higher than the frequency of the analog signal (for example, 128 times the frequency of the analog signal). ) Is required.

This drawback is less of a problem when the analog signal is a signal with a relatively low frequency (for example, an audio signal).
However, when the analog signal is a modulated wave obtained by modulating a carrier wave (unmodulated wave) with a transmission signal such as a radio frequency (RF) signal, the frequency of the carrier wave itself is very high, which is a big problem. .

  For example, in the field of mobile communication, a radio frequency (carrier frequency) of about 1 GHz may be used. If pulse width modulation is performed on a 1 GHz radio frequency signal (RF signal), a very high sampling rate of 128 GS / s, which is 128 times the radio frequency (1 GHz), is required. Sampling and transmitting at such a high speed is unrealistic from the viewpoint of practical use.

Accordingly, an object of the present invention and Turkey suppressing sampling rate.

(1) The present invention is a band-pass ΔΣ modulator that performs band-pass ΔΣ modulation by digital signal processing on a digital modulation wave generated by digital modulation, and the sampling frequency of the band-bus ΔΣ modulator Is a band-pass ΔΣ modulator having a frequency determined by the bandwidth of the digital modulation wave. Another aspect of the present invention is a method for performing bandpass ΔΣ modulation by digital signal processing on a digital modulation wave generated by digital modulation, wherein the sampling frequency of the bandbus ΔΣ modulation is the digital modulation It is a method that is a frequency determined by the bandwidth of the wave. From another point of view, the present invention is a transmitter that transmits a modulated wave, in which a transmission signal is added to an unmodulated wave, to a signal transmission path, and performs bandpass ΔΣ modulation on the modulated wave A transmitter comprising: a band-pass ΔΣ modulator; and an output unit that outputs a quantized signal output from the band-pass ΔΣ modulator to the signal transmission path as the modulated wave. is there. When performing bandpass ΔΣ modulation on a modulated wave, it is sufficient if the sampling frequency is sufficiently large with respect to the signal band of the modulated wave, so even if the frequency of the modulated wave (frequency of the unmodulated wave) is large, There is no need to increase the sampling rate.

(2) The frequency of the unmodulated wave is preferably set to satisfy the following expression.
f ₀ ′ = f ₀ −n × fs
However,
f ₀ : reception frequency on the receiver side fs: sampling frequency of the band-pass ΔΣ modulator f ₀ ′: frequency of the unmodulated wave n: integer

(3) It is preferable that n is an integer having an absolute value of 1 or more. In this case, the reception frequency f ₀ of the receiver can be larger than the sampling frequency fs.

(4) It is preferable to further include a band extending unit that extends a signal band of the modulated wave given to the bandpass ΔΣ modulator. In this case, the sampling frequency can be set high by extending the band.

(5) Preferably, the band extending unit extends the band of the modulated wave by inserting a zero signal outside the signal band of the modulated wave. In this case, the bandwidth can be easily expanded.

(6) It is preferable that a digital modulation unit that generates the modulated wave by digital signal processing is further provided, and the digital modulated wave generated by the digital modulation unit is given to the bandpass ΔΣ modulator. In this case, both the generation of the modulated wave and the generation of the quantized signal can be performed by digital signal processing.

(7) It is preferable that the digital modulation unit is a digital quadrature modulation unit.

(8) The modulated wave is preferably a radio frequency modulated wave.

(9) From another viewpoint, the present invention is a receiver for receiving a signal transmitted from a signal transmission path, and is a bandpass type ΔΣ modulation for a modulated wave in which a transmission signal is added to an unmodulated wave. An input unit that receives the quantized signal generated by performing from the signal transmission path, and an analog bandpass filter that receives the quantized signal received by the input unit as an input. It is a featured receiver. An analog modulated wave can be obtained by passing a quantized signal generated by performing bandpass ΔΣ modulation on the modulated wave through an analog bandpass filter.

(10) The analog bandpass filter is preferably set so as to satisfy the following expression.
= Fc = f ₀ '+ n × fs
However,
fc: center frequency of the passband of the analog bandpass filter fs: sampling frequency of the ΔΣ modulator f ₀ ′: frequency of the unmodulated wave n: integer

(11) The n is preferably an integer having an absolute value of 1 or more.

(12) It is preferable that an analog circuit for processing the modulated wave is further provided, and an output of the analog bandpass filter is given to the analog circuit.

(13) It is preferable that an antenna for outputting radio waves is provided, and the antenna also functions as the analog bandpass filter. In this case, the analog bandpass filter can be omitted.

(14) From another viewpoint, the present invention provides the transmitter according to any one of (1) to (8) and the receiver according to any one of (9) to (13). And a communication system.

(15) The present invention viewed from another viewpoint is a radio base station apparatus comprising a base station main body and a remote radio head connected to the base station main body via a signal transmission path, the base station main body Includes the transmitter according to any one of (1) to (8), and the remote radio head includes the receiver according to any one of (9) to (13). Wireless base station apparatus.

1 is a configuration diagram of a communication system according to an embodiment. It is a block diagram of a ΔΣ modulator. This is a primary low-pass type ΔΣ modulator. (A) is an output spectrum of the low pass type ΔΣ modulator, and (b) is an output spectrum of the band pass type ΔΣ modulator. (A) is an output spectrum of the low pass type ΔΣ modulator, and (b) is an output spectrum of the band pass type ΔΣ modulator. (A) is an output spectrum of the low pass type ΔΣ modulator, and (b) is an output spectrum of the band pass type ΔΣ modulator. (A) is a polar coordinate indicating the operation of the low-pass ΔΣ modulator, and (b) is a polar coordinate indicating the operation of the bandpass ΔΣ modulator. This is a secondary low-pass ΔΣ modulator obtained by conversion from a primary low-pass ΔΣ modulator. This is a low-pass ΔΣ modulator having a CRFB structure. This is a bandpass type ΔΣ modulator obtained by conversion from a low pass type ΔΣ modulator having a CRFB structure. FIG. 6 is an output spectrum waveform diagram of a bandpass ΔΣ modulator for θ ₀ = π / 4. It is an output spectrum waveform diagram of a band pass type ΔΣ modulator for θ ₀ = 3π / 4. It is an output spectrum waveform diagram of a band pass type ΔΣ modulator for θ ₀ = 5π / 4. It is an output spectrum waveform diagram of a band pass type ΔΣ modulator for θ ₀ = 7π / 4. It is a block diagram of the communication system which has a zone | band extension part in a transmitter. It is explanatory drawing of the extended signal band. It is a figure which shows the pass characteristic of the extended signal band and a band pass filter. It is a figure which shows a main signal component and a harmonic component. It is a figure which shows a radio base station apparatus. It is a block diagram of the conventional optical fiber communication system. It is a block diagram of the conventional optical fiber communication system.

Hereinafter, preferred embodiments of the present invention will be described with reference to the drawings.
[1. Communications system]
FIG. 1 shows a communication system 1 according to the embodiment. The communication system 1 includes a transmitter 2 and a receiver 3. The communication system 1 is configured as an optical fiber communication system in which the transmission line 4 is an optical fiber (optical transmission line).
The transmitter 2 can be used as, for example, an OLT (station side device) in a PON (Passive Optical Network). The receiver 3 can be used as, for example, an ONU (home-side device) in the PON.
Note that the communication system is not limited to the optical fiber communication system. For example, the transmission path 4 may be an electric cable (electric transmission path) instead of an optical fiber (optical transmission path).

  The transmitter 2 includes a digital signal processing unit 21. The digital signal processing unit 21 outputs a quantized signal (here, 1-bit quantized signal; pulse signal) that is a digital signal. The digital signal processing unit 21 is output to the optical transmission line 4 by an electro-optical converter (optical link) 22 that is an output unit of a quantized signal. When the transmission line 4 is an electric transmission line, the output unit 22 may be a converter that converts the voltage of the quantized signal. Further, when conversion of voltage or the like is not necessary, the connection terminal to the transmission line 4 is regarded as an output unit.

  The digital signal processing unit 21 includes a baseband unit 23 that outputs a baseband signal (IQ signal) that is a transmission signal, an orthogonal modulator 24 that performs orthogonal modulation on the baseband signal, and a bandpass ΔΣ modulator 25. I have.

The baseband unit 23 outputs IQ baseband signals (I signal and Q signal) as digital data.
The quadrature modulator 24 modulates the carrier wave (non-modulated wave) according to the change of the IQ baseband signal, and outputs a modulated wave (orthogonal modulated wave) in which the IQ baseband signal is added to the carrier wave. The quadrature modulator 24 is configured as a digital quadrature modulator that performs quadrature modulation by digital signal processing. Accordingly, the quadrature modulator 24 outputs a modulated wave (digital modulated wave) in a digital signal format expressed by multi-bit digital data (discrete values). The output modulated wave is given to the bandpass type ΔΣ modulator 25.

A normal radio frequency can be adopted as the frequency of the carrier wave. The radio frequency is preferably 30 MHz or more, more preferably 300 MHz or more, and further preferably 1 GHz or more.
The signal bandwidth of the modulated wave is not particularly limited, but is preferably a narrow band sufficiently small with respect to the carrier frequency. The signal bandwidth is preferably in the range of 5 MHz to 20 MHz, for example.

The modulator 24 that generates the modulated wave is not limited to the quadrature modulator, and may be a modulator of another method for generating the modulated wave.
Further, since the transmitter 2 according to the embodiment includes the quadrature modulator 24, the transmitter 2 itself has a function of generating a carrier wave, but does not have a function of generating a carrier wave. May be. For example, the transmitter 2 may accept a carrier wave generated by an external device of the transmitter 2 as an input, and give the carrier wave to the bandpass ΔΣ modulator 25.

The bandpass type ΔΣ modulator 25 performs bandpass type ΔΣ modulation on the modulation signal output from the quadrature modulator 24 and outputs a 1-bit quantized signal (pulse signal). The bandpass type ΔΣ modulator 25 is set so that its center frequency matches the frequency of the carrier wave.
Note that the quantized signal output from the bandpass ΔΣ modulator 25 does not have to be 1 bit. The quantized signal output from the ΔΣ modulator 25 may be smaller than the number of bits of digital data input to the ΔΣ modulator 25.

The quantized signal (ΔΣ modulation signal) output from the bandpass type ΔΣ modulator 25 (digital signal processing unit 21) is converted into an optical pulse signal by the electro-optical converter 22. The quantized signal is output to the transmission line 4 as a modulated wave.
On the receiver 3 side, a modulated wave of an analog signal can be acquired from this quantized signal (pulse signal). The ΔΣ modulation will be described later.

  In the transmitter 2 of the present embodiment, both the quadrature modulator 24 and the bandpass ΔΣ modulator 25 are configured as digital circuits that perform modulation by digital signal processing. Therefore, it is advantageous that an analog circuit need not be used before the electro-optical converter 22 while handling a modulated wave having a high frequency. Note that the band-pass ΔΣ modulator 25 can output a pulse signal in the same manner even when an analog signal is input instead of a digital signal, so that the quadrature modulator 24 can also be configured with an analog circuit.

The receiver 3 includes an opto-electric converter 31, an analog bandpass filter 32, and an analog circuit 33.
The receiver 3 receives an optical pulse signal (quantized signal) transmitted from the transmission line 4 by an optical-electrical converter 31 as an input unit. The photoelectric converter 31 converts the received optical pulse signal into an electrical pulse signal and outputs it.

When the transmission line 4 is an electric transmission line, the input unit 31 may be a converter that converts the voltage of the quantized signal. When no conversion of voltage or the like is necessary, the connection terminal to the transmission line 4 is regarded as an input unit.
Further, the input unit 31 may obtain a signal obtained by shaping the signal in the transmission path 4 by comparing the signal received from the transmission path 4 with a reference value.

The analog band pass filter 32 is set with a pass band centered on the frequency of the carrier wave, and passes a band near the frequency of the carrier wave (a band slightly wider than the band of the transmission signal).
For example, if the carrier frequency used for the quadrature modulator 24 of the transmitter 2 is 1 GHz, the bandpass filter 32 is set with 1 GHz as the center frequency of the passband.
The analog band filter 32 removes quantization noise that has been noise-shaped (described later) by ΔΣ modulation.

  When a quantized signal (ΔΣ modulation signal) obtained by performing ΔΣ modulation on the modulated wave is applied to the input of the bandpass filter 32, the modulated wave is converted into an analog signal (continuous wave) from the output of the bandpass filter 32. ) Is output. That is, the analog bandpass filter 32 generates and outputs an analog RF signal corresponding to the digital RF signal input to the bandpass ΔΣ modulator 25.

  The output of the band pass filter 32 is given to the analog circuit 33. The analog circuit 33 is not particularly limited as long as it is a circuit that processes an analog signal. For example, the analog circuit 33 can be a circuit of an RF unit in a wireless receiver.

The communication system of this embodiment generally transmits a modulated wave from the transmitter 2 to the receiver 3, but a pulse signal is transmitted in the transmission path 4 between the transmitter 2 and the receiver 3. Since it flows, there is almost no signal degradation due to the influence of the transmission path 4. Therefore, the modulated wave can be transmitted with high quality.
In addition, since a pulse signal (digital signal) flows in the transmission path 4, even if the signal is deteriorated in the transmission path 4, the signal can be corrected on the receiving side using a digital signal processing technique. Therefore, also from this point, an analog signal (modulated wave) can be transmitted with high quality.

[2. Bandpass ΔΣ modulation]
[2.1 Basic configuration of ΔΣ modulator]
As shown in FIG. 2, the ΔΣ modulator 25 includes a loop filter 27 and a quantizer 28 (see Non-Patent Document 1).
In the ΔΣ modulator 25 shown in FIG. 2, an input (modulated wave in this embodiment) U is given to the loop filter 27. The output Y of the loop filter 27 is supplied to a quantizer (for example, 1-bit quantizer) 28. The output (quantized signal) V of the quantizer 28 is given as another input to the loop filter 27.

The characteristic of the delta-sigma modulator 25 can be represented by a signal transfer function (STF) and a noise transfer function (NTF; Noise Transfer Function).
That is, when the input of the ΔΣ modulator 25 is U, the output of the ΔΣ modulator 25 is V, and the quantization noise is E, the characteristics of the ΔΣ modulator 25 are expressed in the z region as follows. is there.

  Therefore, given the desired NTF and STF, the transfer function of the loop filter 27 can be obtained.

Such ΔΣ modulation is a kind of oversampling modulation, and is generally a technique used for AD conversion or DA conversion.
In ΔΣ modulation, noise shaping (Noise Shaping) is performed in which the quantization noise in the signal band is moved outside the signal band to greatly reduce the quantization noise in the signal band.

FIG. 3 shows a block diagram of the linear z-domain model of the first-order low-pass ΔΣ modulator 125. Reference numeral 127 represents a loop filter portion, and reference numeral 128 represents a quantizer. When the input to the ΔΣ modulator 125 is U (z), the output is V (z), and the quantization noise is E (z), the characteristics of the ΔΣ modulator 125 are expressed in the z region. It is as follows.
V (z) = U (z) + (1-z ⁻¹ ) E (z)

That is, in the first-order low-pass ΔΣ modulator 125 shown in FIG. 3, the signal transfer function STF (z) = 1 and the noise transfer function NTF (z) = 1−z ⁻¹ .

[2.2 Low-pass ΔΣ modulation and bandpass ΔΣ modulation]
In general, the term “ΔΣ modulation” refers to low-pass ΔΣ modulation.
In the low-pass ΔΣ modulation, as shown in FIG. 4A, noise shaping is performed so that the low-frequency quantization noise moves to a higher frequency side and the low-frequency quantization noise is attenuated. That is, in the low-pass type Δ modulation, the noise transfer function (NTF) has a characteristic of blocking passing noise at a low frequency (near 0 Hz).

  In order to perform oversampling, the frequency of the signal subjected to ΔΣ modulation needs to be sufficiently smaller than the sampling frequency fs of the low-pass ΔΣ modulator. In other words, a sufficiently high sampling frequency fs is required for the signal frequency. For example, a sampling frequency fs of about 128 times the signal frequency is required.

  On the other hand, in the bandpass type ΔΣ modulation, as shown in FIG. 4B, the noise transfer function (NTF) blocks the passing noise at a frequency higher than 0 Hz.

In the band-pass ΔΣ modulation, not the signal frequency f ₀ but the signal bandwidth f _B should be sufficiently smaller than the sampling frequency fs.
Therefore, in the bandpass type ΔΣ modulation, the frequency (center frequency) f ₀ of the signal subjected to ΔΣ modulation may be equal to or lower than the sampling frequency fs.
In other words, in the band-pass ΔΣ modulation, the sampling frequency fs may be sufficiently larger than the signal bandwidth f _B. For example, a sufficiently large sampling frequency fs of about 64 times the signal bandwidth f _B is sufficient.

Here, for example, it is assumed that the carrier frequency f _{0 of} radio communication is 1 GHz and the signal band f _B is 20 MHz. When low-pass ΔΣ modulation is performed on a modulation wave (RF signal) of such a radio frequency, the maximum frequency of the modulation wave is about 1 GHz, so that the maximum frequency of the modulation wave is 1 GHz. A sufficiently large (about 128 times) sampling frequency fs (= about 128 GHz) is required. As described above, the low-pass ΔΣ modulation is not practical because the sampling frequency (sampling speed) becomes too high, as in the PWM modulation.

On the other hand, in the band-pass ΔΣ modulation, the sampling frequency fs may be sufficiently larger than the signal bandwidth f _B , so that if the signal band is an RF signal of 20 MHz, 20 MHz × 64 = 1. A sampling frequency fs of about 28 GHz (sampling rate = 1.28 GS / S) may be used. If the signal band is 5 MHz, a sampling frequency of 320 MHz (sampling speed = 320 MS / s) may be used.
Thus, the band-pass type ΔΣ modulation is advantageous because the sampling frequency (sampling speed) can be reduced.

[2.3 Contrast with DA conversion using ΔΣ modulation]
In DA conversion using ΔΣ modulation, a digital signal that is input to the DA converter is supplied to the ΔΣ modulator, and oversampling and noise shaping are performed. The low-bit quantized signal output from the ΔΣ modulator is converted into an analog signal by passing through an analog filter that cuts components outside the signal band. This analog signal becomes the output of the DA converter.

The ΔΣ modulator 25 provided in the transmitter 2 of the present embodiment performs oversampling and noise shaping on the input digital signal (modulated wave), similarly to the case where it is used in the DA converter.
However, the low-bit quantized signal output from the delta-sigma modulator 25 of the present embodiment is supplied to the electro-optical converter 22 that is an output unit without passing through the analog filter as it is. It becomes an optical pulse signal.

  The optical pulse signal is received by an optical-electrical converter 31 that is an input unit of the receiver 3, becomes an electric pulse signal, and is given to an analog filter (analog bandpass filter) 32. The analog bandpass filter 32 of the receiver 3 outputs a modulated wave that is an analog signal by cutting a component outside the signal band of the modulated wave.

[2.4 Design of bandpass ΔΣ modulator]
[2.4.1 Conversion formula]
According to Non-Patent Document 1, a low pass type ΔΣ modulator can be converted into a band pass type ΔΣ modulator by performing the following conversion on the low pass type ΔΣ modulator.

By replacing z in the z region model of the low-pass ΔΣ modulator 125 with z ′ = − z ² in accordance with the above conversion formula, a band-pass ΔΣ modulator can be obtained.

Using the above conversion equation, an n-order low-pass ΔΣ modulator (n is an integer of 1 or more) can be converted to a 2n-order band-pass Σ modulator.
For example, the frequency characteristic of the first-order low-pass ΔΣ modulator 125 is as shown in FIG. The frequency characteristic of the secondary bandpass ΔΣ modulator obtained by converting the primary lowpass ΔΣ modulator 125 with the above conversion formula is as shown in FIG. In FIG. 5, the horizontal axis θ is the normalized frequency.

  Although the signal transfer function and noise transfer function of the bandpass ΔΣ modulator obtained by the above conversion equation have the same gain as the low-pass ΔΣ modulator 125 before conversion, the frequency characteristics shown in FIG. The frequency characteristic shown in FIG. 5A is compressed by half and folded.

  The bandpass ΔΣ modulator obtained by the above conversion formula has the same stability characteristics and SNR characteristics as the low-pass ΔΣ modulator 125 before conversion operating at the same oversampling ratio.

However, in the above conversion formula, as shown in FIG. 5B, only a band-pass type ΔΣ modulator for a frequency of 1/4 of the sampling frequency fs (normalized frequency θ = ± π / 2) can be obtained. That is, in the above conversion formula, only a bandpass ΔΣ modulator having a quarter frequency (normalized frequency θ = ± π / 2) of the sampling frequency fs and the center frequency f ₀ of the quantization noise stop band can be obtained.

The inventor has found a conversion formula for obtaining a bandpass type ΔΣ modulator having a desired frequency f ₀ (θ = θ ₀ ) as a center frequency f ₀ from a low pass type ΔΣ modulator. The conversion formula is as shown in the following formula (3), for example.
here,
θ ₀ = 2π × (f ₀ / fs)

The conversion formula of Formula (2) relates to a specific frequency θ ₀ = π / 2, but the conversion formula of Formula (3) is generalized to an arbitrary frequency (θ ₀ ).

[2.4.2 Concept of conversion formula]
In the low-pass type ΔΣ modulator, assuming that z = e ^jωT = 1, the absolute value of z ′ for conversion to the band-pass type ΔΣ modulator should be 1 while maintaining the characteristics of the low-pass type modulator. It is.
If | z ′ | = 1, the magnitude (amplitude) of the signal that has passed through the element z changes, and the characteristics are deteriorated compared to the low-pass ΔΣ modulator before conversion.
Note that z ′ may be 1 or −1. This is because z ′ = 1 and z ′ = − 1 are merely a relationship in which the phases are reversed, and the signal magnitude is not changed.

Therefore, z ′ for obtaining a bandpass ΔΣ modulator while maintaining the characteristics of the lowpass ΔΣ modulator without deterioration is a function f _cnv (z, θ ₀ ) including z and θ _0. For any z, θ ₀ , a function f _cnv (z, θ ₀ ) in which the absolute value of f _cnv (z, θ ₀ ) is always 1 may be used.

If such a function f _cnv (z, θ ₀ ) is found, a low-pass ΔΣ modulator and a band-pass ΔΣ modulator for a desired frequency f ₀ (θ ₀ ) can be obtained.

The present inventor _finds such a function z ′ = f _cnv (z, θ ₀ ) as follows and generalizes a conversion formula z → z ′ (formula (3)) obtained by generalizing formula (2). Obtained.

First, conversion from a low pass type ΔΣ modulator to a band pass type ΔΣ modulator having a desired frequency f ₀ (θ = θ ₀ ) as a center frequency f ₀ is considered as shown in FIG. Become. FIG. 6 is a generalization of FIG. 5 at an arbitrary frequency f ₀ (θ = θ ₀ ).
As shown in FIG. 6B, the center frequency of the noise stop band of the band-pass ΔΣ modulator is f ₀ (θ ₀ = 2π × (f ₀ / fs)).

here,
Therefore, we consider in the frequency domain. T is a sampling period.

In addition, ωT in Equation (4) is
It is.

As shown in FIG. 6A, the low-pass ΔΣ modulator operates at f ₀ = 0 (θ = 0). Therefore, the present inventor considered that the following equation (6) holds in the low-pass type ΔΣ modulator with respect to the equation (4).
That is, it can be considered that the low-pass type ΔΣ modulator operates at e ^{j0 as} shown in FIG.

From the equation (6), the following equation (7) is obtained.

On the other hand, the bandpass ΔΣ modulator operates as a complex conjugate pair at θ ₀ and −θ ₀ as shown in FIGS. 6B and 7B.
Therefore, considering the fact that the bandpass type ΔΣ modulator has a complex conjugate pair based on the formula (7) in the low-pass type Δ modulator, the following formula (8) is obtained.

The present inventor obtained z ′ = f _cnv (z, θ ₀ ) using the formula (8).
That is, first, the above equation (8) is modified as follows to obtain equation (10) in which the value of the right side (one side) is 1.

Equation (10), the value of the expression of the left-hand (the other side) is, any z, the theta _0, it is clear that it is always identity as the left-hand side value = 1.
Therefore, the left side of the equation (10) is a function f _cnv (z, θ ₀ ) _whose value is always 1 for any z and θ ₀ .

From Expression (10), z ′ in the conversion expression z → z ′ for converting from the low pass type to the band pass type is as follows.
From the above equation (11), the conversion equation of equation (3) is obtained.
In the above equation (3), when θ ₀ = π / 2 (when f ₀ = fs / 4) is set, it can be seen that it is equivalent to the conversion equation of equation (2).
Further, the low-pass ΔΣ converter has θ ₀ = 0. When θ ₀ = 0, the conversion equation of Equation (3) becomes z → z, and it can be seen that Equation (3) does not deform the low-pass ΔΣ converter.

Since z ′ = f _cnv (z, θ ₀ ) may be −1 (because the absolute value is 1), z ′ may be in the following format.

Moreover, even if the denominator and the numerator of z ′ = f _cnv (z, θ ₀ ) are replaced, it becomes 1 or −1, and therefore z ′ may be in the following format.

Of course, the expression format of the expression z ′ = f _cnv (z, θ ₀ ) whose absolute value is always 1 for any z, θ ₀ is not limited to that illustrated. Regarding f _cnv (z, θ ₀ ), there are various expression formats. This means that the equation can be transformed from the equation (8) in order to obtain an identity with a value of one side of 1 or −1. It is clear that this is not the case.

[Example of 2.5 bandpass ΔΣ modulator]
[2.5.1 First Example]
FIG. 8 shows a second-order bandpass ΔΣ modulator 25 obtained by converting the first-order lowpass ΔΣ modulator 125 shown in FIG. 3 using the conversion equation (3).
In the conversion from FIG. 3 to FIG. 8, for the convenience of notation, the following conversion equation with a = cos θ ₀ in Equation (3) was used.

[2.5.2 Second Example]
FIG. 9 shows a low-pass ΔΣ modulator 125 having a CRFB structure loop filter 127 described in Non-Patent Document 1. In FIG. 9, reference numeral 128 denotes a quantizer.

When the low-pass type ΔΣ modulator 125 shown in FIG. 9 is converted by the conversion formula (3), a bandpass type ΔΣ modulator 25 shown in FIG. 10 is obtained. Also here, for convenience of description, in equation (3), a = cos θ ₀ is set.

Z in (1 / (z-1)) and (z / (z-1)) in FIG. 9 is converted by a conversion formula. Expressions after conversion of (1 / (z-1)) and (z / (z-1)) are as follows, respectively.

[2.5.3 Others]
The conversion to the band-pass type ΔΣ modulator can be applied to other high-order low-pass type ΔΣ modulators (for example, the CIFB structure, the CRFF structure, the CIFF structure, etc. described in Non-Patent Document 1).

[2.6 Output results]
FIGS. 11 to 14 show a case where θ ₀ = π / 4 (FIG. 11) and θ ₀ = 3π / 4 in the bandpass ΔΣ modulator of the second example (FIG. 10) (FIG. 12). , Θ ₀ = 5π / 4 (FIG. 13), and θ ₀ = 7π / 4 (FIG. 14).

As shown in FIGS. 11 to 14, at each frequency of θ ₀ = π / 4, 3π / 4, 5π / 4, and 7π / 4, a signal appears at a desired θ ₀ , and θ ₀ = ± π It can be seen that bandpass ΔΣ modulators for frequencies other than / 2 are obtained.

Conventionally, a design method for a bandpass ΔΣ modulator that performs bandpass ΔΣ modulation on an arbitrary frequency f ₀ has not been established. However, by using a conversion equation such as Equation (3), the desired carrier frequency f ₀ can be set as the noise rejection band of the noise transfer function (NTF), and the bandpass ΔΣ with respect to the desired carrier frequency f ₀ . A bandpass ΔΣ modulator that performs modulation can be designed.

[3. Bandwidth expansion]
FIG. 15 shows a configuration in which a band extension unit 29 is added to the receiver 3 of the communication system 1 of FIG. In the communication system 1 of FIG. 15, the points that are not described are the same as those of FIG.

As described above, in the band-pass ΔΣ modulation, the signal band f _B only needs to be sufficiently small with respect to the sampling frequency fs. For example, if the signal band is an RF signal having a frequency of 20 MHz, the sampling frequency fs (sampling speed = 1.28 GS / S) may be 20 MHz × 64 = 1.28 GHz.

Here, when the above-described sampling frequency fs (1.28 GHz) can be further increased, the signal band f _B can be afforded. Therefore, as shown in FIG. 15, a band extending unit 29 is provided to reduce the signal band f _B. It is preferable to expand.
As shown in FIG. 16, the band extending unit 29 inserts zero signals on both sides of the original signal band f _B (= 20 MHz) by upsampling, and doubles the signal band (f _B ′ = 40 MHz). As the signal band f _B ′ is expanded, the sampling frequency fs of the bandpass ΣΔ modulator 25 is also doubled to 2.56 GHz.

Thus, when the band f _B is expanded, the sampling frequency fs increases. Here, the carrier frequency f _0, it is possible to select the following values sampling frequency fs, the sampling frequency fs increases, also increases the range of selection of the carrier frequency f _0.

Here, in LTE (Long Term Evolution), which is a wireless communication standard, for example, the signal band f _B = 20 MHz and the carrier frequency f ₀ is 2 GHz. In this case, if the sampling frequency fs is determined based on only the signal band f _B = 20 MHz for the band-pass ΔΣ modulation, the sampling frequency fs = 1.28 GHz = 20 MHz × 64. However, the sampling frequency fs = 1.28 GHz is inappropriate because it is smaller than the carrier frequency f ₀ = 2 GHz.

However, when the sampling frequency fs is determined based on the expanded signal band f _B ′ = 40 MHz, the sampling frequency fs = 2.56 GHz = 40 MHz × 64. In this case, the sampling frequency fs = 2.56 MHz is appropriate because it is larger than the carrier frequency f ₀ = 2 GHz.

Further, the signal band portions f _B1 and f _B2 expanded by the band extending unit 29 are substantially portions where no signal exists. Therefore, as shown in FIG. 17, the band pass filter 32 of the receiver 3 may use the signal band f _B before expansion as the pass band, and the entire extended signal band f _B ′ is the pass band. It does not have to be.
Moreover, since the roll-off of the bandpass filter 32 can be widened using the extended signal band portions f _B1 and f _B2 , the design of the bandpass filter 32 is facilitated.

[4. Use of harmonics]
Since the output of the ΔΣ modulator 25 is a quantized signal (pulse signal), a harmonic component due to aliasing exists in addition to the main signal component.
By utilizing this harmonic component, the frequency of the modulated wave received on the receiver 3 side can be increased while the carrier frequency f ₀ ′ and the sampling frequency fs are kept low on the transmitter 2 side.

For example, it is assumed that the frequency (reception frequency) f ₀ desired to be received by the receiver 3 is 2 GHz. When harmonics are not used as in the communication system 1 described so far, the transmitter 2 side sets the carrier frequency (frequency of unmodulated wave) f ₀ to 2 GHz and the sampling frequency fs to a value larger than 2 GHz. It is necessary to.

However, as shown in FIG. 18, when a modulated wave having a carrier frequency f ₀ ′ is input to the bandpass ΔΣ modulator 25, the output (quantized signal) of the bandpass ΔΣ modulator 25 is the carrier frequency. In addition to having a main signal component centered on f ₀ ′, it also has a harmonic component of f ₀ = n × fs + f ₀ ′ (n is an integer having an absolute value of 1 or more) by folding.
By actively receiving this harmonic component on the receiver 3 side, while processing is performed on the relatively low frequency f ₀ ′ on the transmitter 2 side, the carrier frequency f is on the receiver 3 side. It becomes possible to receive a modulated wave having a high frequency of ₀ = n × fs + f ₀ ′.

Specifically, when the reception frequency f ₀ of the receiver 3 side and 2 GHz, the center frequency fc of the passband of the analog bandpass filter in the receiver 3 is also set to 2 GHz. That is, the receiver 3 receives a modulated wave having a center frequency f ₀ of 2 GHz.
In this case, assuming that the sampling rate fs of the bandpass type Δ modulator 25 of the transmitter 2 is 1.5 GHz (<f ₀ ), the frequency f ₀ ′ of the carrier wave (unmodulated wave) in the quadrature modulator 24 is
f ₀ ′ = f ₀ −fs = 2 GHz−1.5 GHz = 500 MHz
It's okay.

Therefore, the transmitter 2 actually handles a modulated wave having a center frequency (carrier frequency) f ₀ ′ of 500 MHz, but when viewed from the receiver 3 side, the transmitter 2 is as if the center frequency (carrier wave) It can be considered that the frequency) f ₀ is transmitting a modulated wave of 2 GHz. As a result, a modulated wave having a frequency higher than the sampling frequency in the transmitter 2 can be transmitted.

When the main signal component shown in FIG. 18 is transmitted as a signal having a frequency (reception frequency) desired by the receiver 3, and the harmonic component shown in FIG. 18 is transmitted as a signal having a frequency (reception frequency) desired by the receiver 3. In summary, the frequency f ₀ ′ of the carrier wave (unmodulated wave) used in the quadrature modulator 24 of the transmitter 2 satisfies the following expression.
f ₀ ′ = f ₀ −n × fs
However,
f ₀ : reception frequency on the receiver 3 side fs: sampling frequency of the bandpass type ΔΣ modulator 25 f ₀ ′: frequency of carrier wave (unmodulated wave) of the quadrature modulator 24 n: integer

In the above equation, when n = 0, the main signal component shown in FIG. 18 is transmitted as a signal of the frequency (reception frequency) desired by the receiver 3, and the other cases are the harmonics shown in FIG. This is a case where the component is transmitted as a signal of a frequency (reception frequency) desired by the receiver 3.
As n increases, the harmonic component gradually decreases, so n = ± 1 (particularly n = 1) is preferable.

Further, the center frequency fc of the pass band of the analog bandpass filter 32 of the receiver 3 satisfies the following expression.
fc = f ₀ '+ n × fs
However,
fc: center frequency of pass band of analog bandpass filter 32 fs: sampling frequency of bandpass ΔΣ modulator 25 f ₀ ′: frequency of carrier wave (unmodulated wave) of quadrature modulator 24 n: integer

Also in the above formula, when n = 0, the main signal component shown in FIG. 18 is transmitted as a signal of the frequency (reception frequency) desired by the receiver 3, and the other cases are the harmonics shown in FIG. In this case, the wave component is transmitted as a signal having a frequency (reception frequency) desired by the receiver 3.
As n increases, the harmonic component gradually decreases, so n = ± 1 (particularly n = 1) is preferable.

[5. Application to radio base station equipment]
FIG. 19 shows a variation of the embodiment of the radio base station apparatus 41 using the communication system 1 described above.

A radio base station apparatus 41 shown in FIG. 19 includes a base station main body 42 and a remote radio head 43 that is connected to the base station main body 42 via a signal transmission path (optical transmission path or electrical transmission path) 44. And.
In the radio base station apparatus 41 having a remote radio head, the remote radio head 43 having the antenna 35 can be installed on the building roof while the base station main body 42 is installed inside the building, and the degree of freedom of installation is high.

  Here, in the conventional radio base station apparatus, the base station apparatus main body is configured as a radio apparatus control unit (REC; Radio Equipment Control) that performs baseband signal processing, control and management in the digital domain, and the remote radio head is It is configured as a radio apparatus (Radio Equipment) that performs radio signal processing (modulation, amplification, etc.) in the analog domain.

  In the conventional radio base station apparatus, the digital baseband signal is transmitted from the base station body (REC) to the remote radio head (RE) via the transmission path. Therefore, the remote radio head (RE) needs a circuit for modulating (orthogonal modulation) the digital baseband signal transmitted from the base station body (REC). Moreover, in the radio base station apparatus, a plurality of remote radio heads may be connected in parallel or in series to one base station body (REC). In this case, an orthogonal modulation circuit is provided in each remote radio head. Necessary.

  On the other hand, in the radio base station apparatus 41 of the present embodiment, as shown in FIG. 19, the base station main body 42 includes the transmitter 2 in the communication system 1 of the present embodiment, and the remote radio head includes the present embodiment. The receiver 3 in the communication system 1 is provided.

  In the conventional base station body, the signal output from the baseband unit 23 in the transmitter 2 of the present embodiment is transmitted as an optical pulse signal. In the base station body 42 of the present embodiment, digital signal processing is performed. By including the quadrature modulator 24 and the band-pass ΔΣ modulator 25 as the unit 21, the quadrature-modulated modulated wave (RF signal) is transmitted as a quantized signal (optical pulse) signal.

Therefore, the remote radio head 43 including the receiver 3 does not need to modulate (orthogonal modulation) the signal received from the base station main body 42, and the circuit scale of the remote radio head can be reduced. This is particularly advantageous when a plurality of remote radio heads 43 are connected to one base station body 42 (in parallel or in series).
Note that the modulated wave (RF signal) received by the antenna 35 is subjected to ΔΣ modulation (bandpass ΔΣ modulation) by the remote radio head 43 and transmitted to the base station main body 42.

  19A to 19D, the configuration of the base station main body 42 is common. The base station main body 42 has other functions necessary for the radio base station apparatus 41 in addition to the function as the transmitter 2 of the present embodiment.

  The remote radio head 43 in FIG. 19A is configured to output an analog modulated wave (RF signal) output from the analog bandpass filter 32 from the antenna 35 without passing through an amplifier. Such a connection method is also possible when a high wireless output is not required.

  The remote radio head 43 in FIG. 19B amplifies the analog modulated wave (RF signal) output from the analog bandpass filter 32 by the amplifier (analog amplifier) 36 and outputs it from the antenna 35. In this case, since the analog modulated wave is amplified by the amplifier 36, a high wireless output can be obtained.

  The remote radio head 43 in FIG. 19C amplifies the quantized signal (1 bit pulse signal) before passing through the analog bandpass filter 32 with the digital amplifier 37 (class S amplifier), and then the analog band. An analog modulated wave (RF signal) amplified by passing through the pass filter 32 is obtained. The digital amplifier 37 amplifies the quantized signal (1-bit pulse signal) as it is. The digital amplifier 37 is highly efficient because it operates in a saturated state.

  The remote radio head 43 in FIG. 19D corresponds to the remote radio head 43 in FIG. 19C in which the analog bandpass filter 32 is omitted. The antenna 35 of the remote radio head shown in FIG. 19D has a characteristic of blocking the passage of signals in a band other than the vicinity of the center frequency (carrier frequency) of the RF signal (modulated wave). That is, the antenna 35 has the same function as the analog bandpass filter 32 and also serves as the analog bandpass filter 32.

In the remote radio head 43 of FIG. 19D, the quantized signal (1 bit pulse signal) amplified by the digital amplifier 37 is band-limited by the band-pass filter function of the antenna 35 to become an analog modulated wave. And radiated as a radio wave from the antenna 35.
When a high output is not required, the amplifier 37 may be omitted in FIG. 19D as in FIG. 19A. In this case, both the amplifier 37 and the bandpass filter 32 are omitted, which is advantageous.

Here, in this embodiment and the conventional radio base station apparatus, the transmission speed of the signal transmitted from the base station body to the remote radio head will be considered.
In the case of LTE, for example, the signal bandwidth f _{B of} the IQ baseband signal is 5 MHz, the sampling rate of the IQ baseband signal is 7.68 MS / s, the I signal is 20 bits, and the Q signal is 20 bits. Therefore, the sampling rate for transmitting the digital IQ baseband signal serially from the base station body to the remote radio head is (20 bits + 20 bits) × 7.68 MS / s = 307.2 MS / s, and this sampling rate is the transmission rate. It becomes.

On the other hand, in the radio base station apparatus 41 according to the present embodiment, the sampling rate may be about 64 times the signal bandwidth f _B = 5 MHz of the IQ baseband signal, and is 320 MS / s, and this sampling rate becomes the transmission rate. . In this case, the carrier frequency can be freely selected as long as it is 320 MHz or less. As shown in FIG. 18, when using a harmonic, a carrier frequency of 320 MHz or more can be selected.

  As described above, the conventional radio base station apparatus transmits an IQ baseband signal at a transmission rate of 307.2 Mb / s, whereas the radio base station apparatus 41 of the present embodiment is almost the same as the conventional one. Since modulation wave transmission (RF signal transmission) can be performed at a transmission speed of about 320 Mb / s, it is advantageous.

[6. Addendum]
The embodiment disclosed this time should be considered as illustrative in all points and not restrictive. The scope of the present invention is defined by the terms of the claims, rather than the meanings described above, and is intended to include any modifications within the scope and meaning equivalent to the terms of the claims.

DESCRIPTION OF SYMBOLS 1 Communication system 2 Transmitter 3 Receiver 4 Transmission path 21 Digital signal processing part 22 Output part 23 Baseband part 24 Orthogonal modulator 25 Band pass delta-sigma modulator 27 Loop filter 28 Quantizer 29 Band extension part 31 Input part 32 Analog Bandpass filter 33 Analog circuit 35 Antenna 36 Analog amplifier 37 Digital amplifier 41 Radio base station device 42 Base station body 43 Remote radio head 44 Transmission path 125 Low-pass type ΔΣ modulator 127 Loop filter 128 Quantizer

Claims (7)

Against digital modulated digital RF signals, a band-pass by the digital signal processing in the sampling frequency higher than the frequency of the carrier wave transmitted by radio to a frequency determined by the bandwidth of the RF signal A bandpass ΔΣ modulator that performs ΔΣ modulation and outputs a quantized signal including the component of the RF signal, wherein the frequency of the RF signal is the frequency of the carrier wave. Modulator. A device for digital signal processing,
An apparatus comprising the bandpass ΔΣ modulator according to claim 1. The apparatus according to claim 2, wherein the apparatus includes a digital modulation unit that performs the digital modulation. A bandpass ΔΣ modulator that performs bandpass ΔΣ modulation by digital signal processing on a digital modulated wave generated by digital modulation, wherein the sampling frequency of the bandpass ΔΣ modulator is the frequency of the digital modulated wave A bandpass ΔΣ modulator that is a frequency determined by the bandwidth;
A band extending unit that extends a signal band of the digital modulation wave supplied to the bandpass ΔΣ modulator. The apparatus according to claim 4, wherein the band extending unit extends the band of the digital modulated wave by inserting a zero signal outside the signal band of the modulated wave. The apparatus according to claim 3 , wherein the digital modulation unit is a digital quadrature modulation unit. Against digital modulated digital RF signals, a band-pass by the digital signal processing in the sampling frequency higher than the frequency of the carrier wave transmitted by radio to a frequency determined by the bandwidth of the RF signal A method of performing ΔΣ modulation and outputting a quantized signal including a component of the RF signal, wherein a frequency of the RF signal is a frequency of the carrier wave .