JP6277087B2 - Power converter - Google Patents

Description

  Embodiments described herein relate generally to a power conversion apparatus.

  In recent years, devices using DC power, such as digital home appliances and OA (office automation) devices, have become widespread. Generally, these devices are equipped with a power converter for obtaining DC power from a commercial AC power supply. While the demand for power saving of electronic devices is increasing, power converters mounted on each device are indispensable for reducing loss.

  As a circuit to be applied to a power converter that obtains direct current from alternating current, a system composed of three circuit elements such as a diode rectifier circuit, a power factor correction circuit (PFC), and an insulated DC / DC converter is generally used. Are known.

  In this method, the diode rectifier circuit rectifies an alternating voltage and converts it into a direct voltage. Since the amplitude of the voltage rectified by the diode rectifier circuit varies greatly as in the case of the AC voltage, it is necessary to connect a smoothing capacitor to smooth the voltage.

  However, when a smoothing capacitor is connected, the diode rectifier circuit operates such that the diode conducts only when the AC voltage is higher than that of the smoothing capacitor. For this reason, the current flowing into the diode rectifier circuit from the AC power supply has a waveform with a poor power factor having an amplitude only near the peak of the AC voltage. Therefore, a PFC circuit is connected between the diode rectifier circuit and the smoothing capacitor to improve the power factor.

  A DC power supply can be obtained from an AC power supply using only a diode rectifier circuit and a PFC circuit, but the voltage required at the load is often lower than the AC power supply voltage, such as 5 V or 12 V, particularly in the case of electronic equipment. For this reason, a step-down DC / DC converter is connected to step down to a desired voltage.

  As described above, by using the diode rectifier circuit, the PFC circuit, and the DC / DC converter, it is possible to configure a power converter that has a good input power factor and that can obtain a stable output voltage.

Jun-Ho Kim et al, “Analysis and Design of Boost-LLC Converter for High Power Density AC-DC Adapter”, ECCE Asia 2013, pp. 6-11

  However, since the conventional power conversion device has a multistage configuration using three circuits, the loss of the entire device is the sum of the losses generated in each circuit. For this reason, in the conventional power converter, even if the conversion efficiency of each circuit is good, the conversion efficiency of the whole apparatus will fall.

  The problem to be solved by the present invention is to provide a power conversion device that can improve the conversion efficiency of the entire device while having the functions of a rectifier circuit, a PFC circuit, and a DC / DC conversion circuit.

The power converter of the embodiment includes a low-pass filter, a first inductor, a transformer, an H bridge, a first rectifier, a power supply voltage detector, a first current detector, a capacitor voltage detector, an output voltage detector, and a controller. To do.

  The low-pass filter includes a filter inductor and a filter capacitor that are connected in series to an AC power source to form a closed loop.

  One end of the first inductor is connected to a connection point between the filter inductor and the filter capacitor.

  The transformer includes a transformer primary winding having one end connected to a connection point between the AC power supply and the filter capacitor, and a transformer secondary winding electromagnetically coupled to the transformer primary winding.

  The H bridge connects a first switch and a second switch in series via the other end of the first inductor, and a third switch and a fourth switch via the other end of the transformer primary winding. Are connected in series, and the first switch, the third switch, the second switch, and the fourth switch are connected to form a closed loop, and smoothing capacitors are connected to both ends of the third switch and the fourth switch. Connected.

  The first rectifier is connected to the transformer secondary winding.

The first current detection means detects a first inductor current flowing through the first inductor .

  The power source current detecting means detects an AC power source current that flows to the AC power source.

  The capacitor voltage detection means detects a capacitor voltage of the smoothing capacitor.

  The output voltage detection means detects an output voltage output from the first rectification means to a load.

It said control means, a detected value of the AC power supply voltage, the detection value of the first inductor current, and on the basis of the detected value of the capacitor voltage, to flow a first inductor current synchronized with the AC power supply voltage, and Based on the detected value of the output voltage, a set of the first switch and the fourth switch, a set of the second switch and the third switch so as to eliminate a deviation from the output voltage command value with respect to the output voltage. A switching signal for alternately opening and closing is supplied to the set of the first switch and the fourth switch and the set of the second switch and the third switch .

Wherein the control means comprises an amplitude command value calculating means, and sine wave generating means, and the current command value calculating means, the width sending means, the command value generation means, and determination means, and a signal generating means.

The amplitude command value calculation means calculates an amplitude command value of an AC power supply current to be supplied to the AC power supply based on the detected value of the capacitor voltage. The sine wave generating means generates a sine wave having the same phase as the AC power supply voltage based on the detected value of the AC power supply voltage. The current command value calculation means calculates a current command value of the AC power supply current based on the amplitude command value and the sine wave. The width sending means detects a deviation between the detected value of the output voltage and the output voltage command value, and sends a value corresponding to the deviation as a width. The command value generating means generates an arc extinguishing command value and an ignition command value by giving the current command value the width. The determination means determines whether or not the detected value of the first inductor current falls within a range between the extinction command value and the ignition command value. The signal generation means generates the switching signal so as to switch the opening and closing at a timing when the detected value of the first inductor current is out of the range according to the determination result .

It is a schematic diagram which shows the power converter device which concerns on 1st Embodiment, and its periphery structure. It is a schematic diagram which shows the structure of the control part in the embodiment. It is a schematic diagram which shows the structure of the alternating current command value calculation part in the embodiment. It is a schematic diagram which shows the modification of the alternating current command value calculation part in the embodiment. It is a schematic diagram for demonstrating the operation | movement in the embodiment. It is a schematic diagram for demonstrating the operation | movement in the embodiment. It is a schematic diagram which shows the structure of the gate signal generation part in 3rd Embodiment. It is a schematic diagram which shows the structure of the gate signal generation part in 3rd Embodiment. It is a schematic diagram which shows the power converter device which concerns on 4th Embodiment, and its periphery structure. It is a schematic diagram which shows the structure of the control part in the embodiment. It is a schematic diagram for demonstrating the operation | movement in the embodiment. It is a schematic diagram which shows the structure of the power converter device relevant to each embodiment. It is a wave form diagram showing the voltage and current waveform in the device. It is a schematic diagram which shows the equivalent circuit in each frequency of the same apparatus. It is a wave form diagram which shows the voltage and electric current waveform in the zero crossing point of the alternating current power supply voltage of the apparatus. It is a wave form diagram which shows the voltage and electric current waveform in the bottom point of the alternating current power supply voltage of the apparatus. It is a schematic diagram which shows the control structure of the apparatus. It is a figure which shows the frequency-gain characteristic of the apparatus. It is a figure which shows the experimental conditions of the same apparatus. It is a figure which shows the constant used for each experiment example of the same apparatus. It is a wave form diagram which shows the voltage and electric current waveform at the time of steady state in the 1st experiment example of the same apparatus. It is a wave form diagram which shows the voltage and electric current waveform in the zero crossing point of the alternating current power supply voltage of the said 1st experiment example. It is a wave form diagram which shows the voltage and current waveform in the bottom point of the alternating current power supply voltage of the said 1st experiment example. It is a figure which shows the characteristic of the efficiency with respect to the output voltage of each experimental example, and a frequency. It is a figure which shows the breakdown of the loss at the time of the output of 100 W of each experiment example.

  Hereinafter, each embodiment will be described with reference to the drawings, but before that, an outline of the configuration and operation related to each embodiment will be described.

FIG. 12 is a schematic diagram showing a configuration of a power conversion device related to each embodiment. As shown, an H-bridge inverter, transformer, inductor, and LC harmonic filter are connected in series to a single-phase AC power source. The secondary side of the transformer has two windings, and is connected to a smoothing capacitor _Cout and a load via a rectifier circuit. The smoothing capacitor C _DC of the H-bridge is charged with a voltage obtained by boosting the AC power supply voltage V _AC. The H bridge outputs an AC output voltage V _ab by switching. The LC harmonic filter (L _f , C _f ) is a filter circuit installed so that the high frequency first inductor current I _L1 generated in the H bridge does not flow to the AC power supply. The first inductor L ₁ is provided for soft switching of the H bridge by resonance with the capacitor C _f of the LC harmonic filter.

Next, the principle of AC-DC conversion of such a power converter is described. The power converter shown in FIG. 12 realizes the PFC operation for controlling the AC power supply current with one H-bridge inverter and the control of the power transmission to the secondary side of the transformer. FIGS. 13A to 13F show AC power supply voltage, AC power supply current, and voltage / current waveforms output from the inverter. The inverter outputs an output voltage V _ab synchronized with the AC power source, and a transformer primary-side current I _L1 flows through the primary-side circuit. In the current _IL1 , the carrier frequency component of the inverter is superimposed on the _AC power supply current IAC. The secondary current I _T2 can be expressed by the equations (1) to (3) using the exciting current _Im and the number of turns n ₂ and n ₁ . _Two transformer secondary windings having a winding number n ₂ are connected in series and the current I _T2 is rectified, so that I _T2 always flows only on the positive side. The current I _{T2 including} the ripple of the carrier frequency component is smoothed by the capacitor _Cout . The capacitor _Cout outputs a constant output voltage _Vout .

  FIG. 14A shows an equivalent circuit of the power conversion device at an AC power supply frequency (50 Hz), and FIG. 14B shows an equivalent circuit of the power conversion device at a carrier frequency (several tens of kHz to several hundreds of kHz).

In FIG. 14A, the load on the secondary side of the transformer is replaced with a variable impedance Z. The H-bridge inverter is modulated and controlled in a sine wave shape of 50 Hz, and an AC power supply current I _AC flows depending on the differential voltage between the inverter output voltage V _ab and the AC power supply voltage V _AC and the load impedance Z. The PFC operation is realized by controlling the _AC power supply current I _AC to be in phase with the _AC power supply voltage V _AC . PFC, i.e., the charge is also realized from the AC power source voltage V _AC by controlling to flow only active current to H-bridge capacitor C _DC.

On the other hand, FIG. 14B is an equivalent circuit of the carrier frequency component, and the high frequency first inductor current I _L1 passes through the capacitor C _f of the harmonic filter and is transmitted to the secondary side by the transformer. Since the voltage V _ab output from the inverter is sinusoidally modulated unlike the conventional LLC converter, the ON ratio differs between the vicinity of the zero cross and the bottom (peak) of the AC power supply voltage. Near the zero cross, the pulse is 50%, and the bottom (peak) is thick (thin). Since the power transmitted by the transformer is considered to be the effective value of the harmonics of the applied voltage, when the same carrier frequency and the same DC voltage _VDC are used, the harmonics are maximum near the zero cross (maximum power transmission), and the harmonics are at the bottom (peak). Minimum (minimum power transmission).

Next, the zero volt switching (ZVS) operation principle of the power converter will be described. FIG. 15 and FIG. 16 show voltage / current waveforms at various parts in the carrier cycle of the inverter. 15A to 15F show waveforms at the zero cross point of the _AC power supply voltage V _AC , and FIGS. 16A to 16F show waveforms at the bottom point of the _AC power supply voltage V _AC . The first inductor current I _L1 at an arbitrary time t in the figure is expressed by equation (4).

V _Lf is expressed by equation (5). When the cutoff frequency of L _f and C _f is designed so that the LC harmonic filter can sufficiently prevent the outflow of harmonics to the AC power source, dI _AC is much smaller than dI _L1 , so that V _{Lf ≈0} Can be simplified. V _L1 shown in FIGS. 15E and 16E is simplified.

Here, the ZVS operation of the H-bridge inverter will be described. Now, in a situation where V _ab > 0, that is, the b-phase high-side switch S ₃ is on, and when the first inductor current I _L1 <0, which is the H-bridge output current, when S ₃ is off, I _L1 is through the parasitic diode of S ₃ during the dead time period. Thereafter, when the low-side switch S ₄ is turned on, ZVS cannot be achieved, and a large recovery loss occurs during the reverse recovery time of the S ₃ parasitic diode. In contrast, V _ab> 0 for I _L1 during the application flows S ₄ during the dead time period after S ₃ off if increased to positive, ZVS can be achieved during subsequent S ₄ on. In view of FIGS. 15 and 16, V _ab > 0 is from time t ₀ to t ₂ , and it is _sufficient that I _L1 does not become negative during this period. That is, in the second stage of the equation (4), the equation (6) in which I _L1 (t ₂ ) at t = t ₂ becomes positive is one of the conditions. Similarly, considering that V _ab <0 is being applied, equation (7) is a condition, and when equations (6) and (7) are established simultaneously, ZVS is always established.

Compared with the conventional LLC converter, the expressions (6) and (7) are not always 0 (= 50% pulse width) in the modulation rate of the H-bridge inverter that defines the _AC power supply voltage V _AC and the times t _{0 to} t _4. The point is different.

  Next, the control configuration of the power converter will be described with reference to FIG. Specifically, an AC power supply current control unit and a part that controls a DC output voltage will be described.

In FIG. 17, a portion surrounded by a broken line Control for PFC is a control unit that realizes a PFC operation for matching the phase of the AC power supply current I _AC with the AC power supply voltage V _AC . V _{DC_ref} is an H bridge capacitor voltage command value and is compared with the detected capacitor voltage V _DC to generate an AC power source current amplitude command value I _{AC_amp_ref} by PI calculation. Further, an AC power supply voltage phase ωt is detected from the AC power supply voltage V _AC by a PLL (phase-locked loop), and I _{AC_amp_ref} and sin ωt are multiplied to calculate an AC power supply current command value I _{AC_ref} . I _{AC_ref} is compared with the detected AC power supply current I _AC to generate an H-bridge output voltage command value V _{ab_ref} by PI calculation. The H bridge output voltage command value V _{ab_ref} is _divided by the H bridge capacitor voltage V _DC to become a modulation command value D. The modulation command value D is compared with a triangular wave carrier to generate a PWM pulse. The A-phase and B-phase pulses are the same signal with high and low reversed.

The effective current is controlled by generating the AC power supply current command value I _{AC_ref} in the same phase as the AC power supply voltage V _AC so that the capacitor voltage V _DC becomes a constant command value, and the PFC operation is realized.

On the other hand, in FIG. 17, a portion surrounded by a broken line Control for DC-DC is a control unit that controls the DC output voltage _Vout to a constant command value. The output voltage _Vout on the transformer secondary side is detected and compared with the output voltage command value _{Vout_ref} to perform PI calculation. Result of the PI calculation is the carrier frequency f _c.

Expression (8) is a transfer function G (s) of V _out with respect to the inverter output voltage V _ab in the high-frequency component equivalent circuit shown in FIG.

The resistance of the output load is multiplied by the number of turns n ₁ and n ₂ to obtain a converted value on the primary side.

FIG. 18 shows the frequency-gain characteristic of the equation (8). The constants are L _f = 1.5 mH, C _f = 0.45 mF, L ₁ = 200 uH, L _m = 200 uH, n ₁ / n ₂ = 5.5, and R = 5.76 W. In the equation (8) and FIG. 18, since the fluctuation of the AC power supply voltage V _AC of 50 Hz is not taken into consideration, attention should be paid to the fact that the _AC power supply voltage V _AC has a characteristic at zero crossing. In the LC resonance characteristics shown in FIG. 18, the power converter uses only the right side (high frequency side) from the resonance point. For this reason, it is necessary to select the constant of each part so that it may become the minimum frequency _{fc_min} with the assumed maximum output _Pmax . Then, to adjust the carrier frequency f _c by the magnitude of the load current, keep the output voltage V _out constant. Incidentally, the amount of power appearing to transformer secondary, the size V _DC of the harmonic output of the H-bridge inverter, the gain characteristics of the based on the carrier frequency f _c (8) formula, and determined by the harmonic effective ratio alpha. The harmonic effective ratio α is related to the modulation rate of the inverter, becomes maximum at a modulation rate of 0 where the pulse width is 50%, and decreases as the modulation rate approaches 1 or −1. Since the modulation factor fluctuates at ω = 50 Hz by the control of the _AC power supply current I _AC , the above-described harmonic effective ratio α fluctuates at the double frequency 2ω = 100 Hz. Further, the capacitor voltage V _DC has a 2ω ripple centered on the average value V _{DC —} avg when a sine wave effective current having an effective value I _ACrms is passed to the AC power supply current.

That is, when the carrier frequency f _c is constant, the output voltage V _out is subjected to two influences of the capacitor voltage V _DC and harmonic effective ratio alpha, ripples 2ω are generated. The output voltage controller outputs a constant output voltage _Vout by frequency modulation so as to suppress these 2ω ripples.

  Next, the configuration of an experimental example of the power conversion device will be described.

FIG. 19 shows experimental conditions, and FIG. 20 shows constants of capacitors, inductors, and transformers. The constants are shown for two types, the first experimental example and the second experimental example. The input was a single phase 100V _rms and the output was a maximum of 100W. The operating switching frequency is a minimum of 25 kHz and a maximum of 100 kHz. Each switch of the H bridge uses a SiC-MOSFET with R _ON = 50 mW, and the secondary rectifier element uses an SBD (Schottky Barrier Diode) diode NTSJ30U100CTG with V _f = 0.4V. Since the experiment example was before establishing the ZVS condition, SiC was used. However, in reality, it is possible to use a Super Junction-MOSFET having a large Q _rr and a withstand voltage of about 600V. Similarly, a large core of 50 mm is used for the inductor and transformer.

Next, characteristics of PFC operation and output voltage control will be described. FIG. 21 shows voltage / current waveforms of each part of the power conversion device in a steady state using the constants of the first experimental example. It can be seen that the AC power supply current I _AC is in phase with the _AC power supply voltage V _AC , and the distortion is small. The output voltage V _out is being controlled to the command value 24V, can also be confirmed that the remaining ripple 1V _pp about 2 [omega. This is presumably because the gain of the carrier frequency control of the output voltage is low.

Next, the ZVS characteristic will be described. 22 and 23 show waveforms of the H-bridge output voltage V _ab , the transformer excitation voltage V _m , the AC power supply current V _AC , and the first inductor current I _{L1 in} the carrier cycle using the constants of the first experimental example.

FIG. 22 shows the zero crossing of the AC power supply voltage / current described in FIG. 15, and FIG. 23 shows the vicinity of the bottom described in FIG. At the time of zero crossing, the first inductor current I _L1 flows evenly and positively, and it can be seen that the ZVS operation is the same as that of a general LLC converter.

On the other hand, near the bottom of the AC power supply voltage, the current ripple toward the negative side increases and the positive side decreases. However, while the H bridge output voltage V _ab outputs a positive voltage, I _L1 > 0, which satisfies the ZVS condition described in the equation (6). That is, I _L1 changes from negative to positive during application of the positive voltage of the H-bridge output voltage V _ab , and can change from positive to negative during application of the negative voltage. In any switching, the four elements constituting the bridge are ZVS. It is working.

  Next, efficiency and loss analysis will be described.

  FIG. 24 shows the efficiency and frequency characteristics with respect to the output power of the power converter at the experimental parameters shown in FIGS. When comparing the results of the first and second experimental examples, the efficiency of the second experimental example is as high as 92% at the maximum.

  The operating carrier frequency has also increased. Since the power device has a ZVS operation, it is presumed that the difference in carrier frequency mainly affects the loss of the magnetic component.

FIG. 25 shows a breakdown of the loss of the power converter when the output is 100 W in the experimental example. Each loss is measured from the voltage and current of each part, and calculated from the on-resistance specification value of the power device, the iron loss specification with respect to the magnetic flux density of the core, and the measurement value of the high-frequency resistance including skin and proximity effects. Since the conversion is performed in one stage, the loss ratio of the primary side H-bridge (power device) is very small. Inductive losses differ greatly for magnetic components. The main factor is the difference in iron loss. Although the magnetic flux density of the inductor is expressed by Equation (10), the inductance L ₁ is higher in the _first experimental example than in the second experimental example. Further, the effective value of the first inductor current I _L1 is determined by the output P _out , the output voltage V _out , and the transformer turns ratio n ₁ / n ₂ when the exciting current _Im is the same as shown in the equation (11). The same level. Further, in the first and second experimental examples, the core cross-sectional areas S are substantially equal, but the number of turns n _L1 of L ₁ is larger in the core of the first experimental example.

  As a result, the magnetic flux density B of the first experimental example is higher than the magnetic flux density B of the second experimental example, and the core volume of both experimental examples is approximately the same, so the iron loss of the first experimental example is considered to have increased. It is done.

  On the other hand, the loss of SBD, which is the rectifier on the secondary side, accounts for a large proportion of over 4%. By replacing the rectifier with a low-voltage MOSFET, synchronous rectification, and optimal design of magnetic components, the efficiency can be increased to about 96% It is thought that it is raised.

  In summary, according to the above power converter, AC-DC conversion, which has been conventionally converted in a multi-stage configuration, can be performed in one stage, and all switches can be operated in ZVS (Zero Volt Switching). Here, although the said power converter device performs PFC control and DC-DC conversion by 1 step | paragraph, the isolation | separation system of each control is as having mentioned above. And the ZVS operation | movement in the AC-DC conversion operation | movement by the power converter device provided with PFC in one step was demonstrated by the experiment example. It was confirmed that a maximum conversion efficiency of 92% was obtained, and as a result of loss analysis, it was found that the loss of the secondary side rectifier element accounted for the majority.

  Regarding this, in addition to synchronous rectification on the secondary side, optimization of magnetic components and operating carrier frequency can be expected to improve conversion efficiency.

  The above is the outline of the configuration and operation of the power conversion device related to each embodiment. Then, the power converter device which concerns on each embodiment is demonstrated concretely.

<First Embodiment>
FIG. 1 is a schematic diagram showing a power conversion device according to the first embodiment and its peripheral configuration. The power conversion apparatus 100 includes a main circuit unit that converts an _AC power supply voltage VAC obtained from the AC power supply V1 into a DC voltage _Vout and supplies power to the load 5, and a control unit 200 connected to the main circuit unit. I have.

The main circuit section includes a low-pass filter (L _f , C _f ), a first inductor L ₁ , an H bridge (S _{1 to} S ₄ , D _{1 to} D ₄ , C _DC ), a transformer (Tn ₁ , Tn ₂ ), and a diode. D ₅ , D ₆ , capacitor C _out , input voltage detector 1, power supply current detector 2, capacitor voltage detector 3, and output voltage detector 4.

The low-pass filter includes an inductor L _f (for filter) connected in series to the AC power source V1, and a capacitor C _f (for filter) connected in series between the inductor L _f and the AC power source V1. Yes. That is, the inductor L _f and the capacitor C _f are connected in series to the AC power supply V1 to form a closed loop. Incidentally, the term "filter" is a convenient modifier for distinguishing a capacitor C _DC of the first inductor L ₁ and H-bridge of the primary side of the transformer in the claims, omitted or other modifications It may be changed to a word (eg, “0th” or “first”).

The first inductor L ₁ has one end connected to a connection point between the inductor L _f and the capacitor C _f , and the other end connected to a connection point M ₁ of the H bridge.

The transformer includes a transformer primary winding T _n1 having one end connected to a connection point between the AC power supply V1 and the capacitor C _f, and a transformer secondary winding T _n2 electromagnetically coupled to the transformer primary winding T _n1. Have

The H bridge connects the first switch S ₁ and the second switch S ₂ in series via the other end of the first inductor L ₁ , and the third switch via the other end of the transformer primary winding T _n1. S ₃ and the fourth switch S ₄ are connected in series, and the first switch S ₁ and the third switch S ₃ and the second switch S ₂ and the fourth switch S ₄ are connected to form a closed loop, and the third switch a switch S ₃ to both ends of the fourth switch S ₄ formed by connecting the (smoothing) capacitor C _DC. The term “smoothing” is a convenient modifier for distinguishing from the capacitor C _f of the low-pass filter in the claims, and is omitted or changed to another modifier (eg, “second”, etc.). May be.

Specifically, the H bridge includes first and second switches S ₁ and S ₂ connected in series with each other, a capacitor _CDC, and third and fourth switches S ₃ and S ₄ directly connected to each other. They are connected to each other in parallel.

Each of the switches S _{1 to} S ₄ is a self-extinguishing type switching element, and here, an N-type field effect transistor (MOSFET) is used. The diodes D _{1 to} D ₄ connected in antiparallel with the switches S _{1 to} S ₄ may be replaced with body diodes such as MOSFETs. The gate terminals of the switches S _{1 to} S ₄ are connected to the control unit 200.

The source terminal of the first switch S ₁ is connected to the drain terminal of the second switch S ₂ via the connection point M ₁ .

The source terminal of the third switch S ₃ is connected to the drain terminal of the fourth switch S ₄ via the connection point M ₂ .

The first drain terminal of the switch S ₁ has one end of the capacitor C _DC, is connected to the drain terminal of the third switch S _3.

The source terminal of the second switch S ₂ has the other end of the capacitor C _DC, is connected to the source terminal of the fourth switch S _4.

Such an H-bridge switches the switches S _{1 to} S ₄ based on the voltage V _DC charged in the capacitor C _DC , thereby generating an arbitrary voltage (V _ab ) with the capacitor voltage V _DC as the upper limit. Can output. As a method for outputting a voltage by switching, a known PWM (Pulse Width Modulation), PDM (Pulse Density Modulation), or the like can be used as appropriate. Various known methods such as carrier comparison and hysteresis comparator can be applied to the PWM and PDM generation methods. Although will be described later charging method and the voltage control method thereof to the capacitor C _DC of the H-bridge, it is possible to charge and control of the capacitor C _DC using an AC power source voltage V _AC.

The output of such an H-bridge has a connection point M ₁ between the first and second switches S ₁ and S ₂ connected in series, and a connection point M between the third and fourth switches S ₃ and S ₄ connected in series. Obtained as a voltage V _ab between ₂ . Hereinafter also referred to this voltage V _ab and H-bridge output voltage V _ab.

The primary winding T _{n1 of the} transformer is connected in series between the connection point M ₂ of the H bridge and the connection point of the capacitor C _f and the AC power supply V 1.

The anode terminals of the fifth diode D ₅ and the sixth diode D ₆ are connected to both ends of the secondary winding T _n2 of the transformer, respectively. The cathode terminals of the fifth diode D ₅ and the sixth diode D ₆ are connected to each other. A capacitor _Cout is connected between the connection point of each cathode terminal and the midpoint of the secondary winding _Tn2 of the transformer. These diodes D ₅ and D ₆ and capacitor C _out constitute a rectifying / smoothing circuit (first rectifying means). A load 5 is connected in parallel across the capacitor _Cout .

Input voltage detection unit 1 is connected in parallel to both ends of the AC power supply V1, and detects the AC power supply voltage V _AC input from the AC power supply V1 to the main circuit, control V _AC detection value indicating the instantaneous value of V _AC Output to the unit 200. The input voltage detection unit 1 constitutes a power supply voltage detection unit that detects the _AC power supply voltage V _AC of the AC power supply V1.

Power supply current detection unit 2 is connected in series between the AC power supply V1 and the low-pass filter inductor L _f, the AC power supply to detect the AC power source current I _AC flowing through the V1, I _AC detection value indicating the instantaneous value of I _AC Is output to the control unit 200. The power supply current detection unit 2 constitutes a power supply current detection unit that detects an _AC power supply current I _AC that flows to the AC power supply V1.

Capacitor voltage detecting unit 3 is connected in parallel across the capacitor C _DC of the H-bridge, it detects the voltage V _DC of the capacitor C _DC, and outputs a V _DC detection value indicating the instantaneous value of V _DC to the control unit 200. Capacitor voltage detecting unit 3 constitute a capacitor voltage detecting means for detecting the capacitor voltage V _DC (for smoothing) capacitor C _DC.

Output voltage detection unit 4 is connected in parallel across the load 5 side of the capacitor C _out, detects the output voltage V _out of the capacitor C _out, output V _out detection value indicating the instantaneous value of V _out to the controller 200 To do. Output voltage detection unit 4 constitute an output voltage detecting means for detecting a first rectifying means _{(D 5, D 6, C} out consists rectifying smoothing circuit) the output voltage V _out from) is output to the load 5 .

Based on the detected value of AC power supply voltage V _AC , the detected value of AC power supply current I _AC , and the detected value of capacitor voltage V _DC , control unit 200 causes _AC power supply current I _AC synchronized with _AC power supply voltage V _AC to flow. as such, and based on the detected value of the output voltage V _out, so as to eliminate the deviation V _{Out_dif} between the output voltage command value V _{Out_ref} for the output voltage V _out, the first switch S ₁ and the fourth switch S ₄ pairs And control means for supplying switching signals S _{1_PWM to} S _{4_PWM} for alternately opening and closing the pair of the second switch S ₂ and the third switch S ₃ to the first switch S _{1 to} the fourth switch S ₄ doing.

Specifically, the control unit 200 is based on the detection values received from the detection units 1 to 4, for example, functions shown in a portion surrounded by broken lines (Control for PFC, Control for DC-DC) in FIG. 17. Thus, the switching signals S _{1_PWM to} S _{4_PWM} for controlling the main circuit section are generated. The switching signals S _{1_PWM to} S _{4_PWM} are individually output from the control unit 200 to the gate terminals of the switches S _{1 to} S ₄ . The first and fourth switches S ₁ and S ₄ are conductive while the switching signals S _{1_PWM} and S _{4_PWM} are supplied to the gate terminals. The second and third switches S ₂ and S ₃ are conductive while the switching signals S _{2_PWM} and S _{3_PWM} are supplied to the gate terminals.

Here, as shown in FIG. 2, the control unit 200 includes a V _{DC_ref} setting unit 201, a subtraction unit 202, a voltage controller (AVR) 203, an AC command value calculation unit 204, a subtraction unit 205, and a current controller (ACR). 206, a division unit 207, and a switching signal generation unit 208.

V _{dc_ref} setting unit 201 outputs the voltage command value V _{dc_ref} preset capacitor C _DC to the subtraction unit 202.

The subtraction unit 202 subtracts the voltage command value V _{DC_ref} received from the V _{DC_ref} setting unit 201 from the V _DC detection value received from the capacitor voltage detection unit 3, and the obtained deviation V _{DC_dif} (= V _DC -V _{DC_ref} ) Is output to the voltage controller (AVR) 203.

The voltage controller (AVR) 203 generates an amplitude command value I _{AC_amp_ref} of the AC power supply current I _AC based on the deviation V _{DC_dif} received from the subtracting unit 202 by PI calculation as shown in FIG. _{AC_amp_ref} is output to the AC command value calculation unit 204.

The AC command value calculation unit 204 is based on the amplitude command value I _{AC_amp_ref} received from the voltage controller (AVR) 203 and the V _AC detection value received from the input voltage detection unit 1, and the current command value I related to the inductor L _{f. Generate AC_ref} .

For example, as shown in FIG. 3, the AC command value calculating section 204, the PLL211 from V _AC detection value, to detect the V _AC power supply voltage phase .omega.t, the sine wave generator 212, the power supply voltage phase .omega.t the same phase A sine wave sin ωt is generated. Thereafter, the AC command value calculation unit 204 multiplies the amplitude command value I _{AC_amp_ref} and the sine wave sin ωt by the multiplication unit 213 to calculate a current command value I _{AC_ref} having the same phase as V _AC . Here, for example, as shown in FIG. 4, the constant multiplication unit 210 multiplies the V _AC detection value (eg, amplitude 200 V) by a constant K (eg, 1/200) to obtain the obtained K · _VAC detection value. (Eg, amplitude 1V) may be sent to the PLL 211.

Anyway, the AC command value calculating section 204, as shown in FIG. 5, the amplitude command value I _{AC_amp_ref,} by multiplying a sine wave sin .omega.t of V _AC and the same phase, current command V _AC in phase Calculate the value I _{AC_ref} . Thereafter, the AC command value calculation unit 204 outputs the current command value I _{AC_ref} to the subtraction unit 205.

The subtraction unit 205 subtracts the I _AC detection value received from the power supply current detection unit 2 from the current command value I _{AC_ref} received from the AC command value calculation unit 204, and obtains the obtained deviation I _{AC_dif} (= I _{AC_ref} −I _AC ) To the current controller (ACR) 206.

The current controller (ACR) 206 generates an H bridge output voltage command value V _{ab_ref} by PI calculation as shown in FIG. 17 based on the deviation I _{AC_dif} received from the subtraction unit 205, and generates an H bridge output voltage command value V _{ab_ref} is output to the division unit 207.

The division unit 207 divides the H-bridge output voltage command value V _{ab_ref} received from the current controller (ACR) 206 by the _VDC detection value received from the capacitor voltage detection unit 3, and obtains a modulation command value D (= V _{ab_ref} / V _DC ) is output to the switching signal generator 208. The “modulation command value D” may be called “voltage ratio D”. The “divider” may be referred to as “ratio calculator”.

Switching signal generation unit 208 controls switching signals S _{1_PWM to} S for controlling the main circuit unit based on modulation command value D received from division unit 207 and V _out detection value received from output voltage detection unit 4. _{4_PWM} is generated.

Here, the switching signal generation unit 208 determines the switching frequency based on the magnitude of the deviation V _{out_dif from} the output voltage command value V _{out_ref} based on the detected value of the output voltage V _out , and _performs switching according to the switching frequency. constitute a signal generating means for generating a signal S _{1_PWM} ~S _{4_PWM.} The switching frequency is determined so as to eliminate the deviation _{Vout_dif} .

The signal generating means, based on the detected value of the output voltage V _out, and the frequency calculating means for calculating a carrier frequency f _c on the basis of the magnitude of the deviation V _{Out_dif} between the output voltage command value V _{Out_ref,} carrier frequency f _c Carrier generating means for generating the carrier signal S _car having the first and second generating means for generating the switching signals S _{1_PWM to} S _{4_PWM} based on the carrier signal S _car .

Specifically, the switching signal generation unit 208 generates the switching signals S _{1_PWM to} S _{4_PWM} by the function shown in the part surrounded by the broken line (Control for DC-DC) in FIG. 17, for example. For example, the switching signal generation unit 208 subtracts a preset output voltage command value V _{out_ref} from the V _out detection value, and based on the obtained deviation V _{out_dif} (= V _out −V _{out_ref} ), as shown in FIG. by such PI calculation to obtain the carrier frequency f _c. The carrier frequency f _c is determined so as to eliminate the deviation _V out_dif. The value of the carrier frequency f _c corresponds to the value of the switching frequency. Therefore, determining the carrier frequency f _c is equivalent to determining the switching frequency.

Further, the switching signal generating unit 208 generates a carrier signal S _car of the triangular wave carrier having a carrier frequency f _c obtained are compared with the carrier signal S _car and modulation instruction value D.

When the comparison result is S _car <D, the switching signal generation unit 208 includes “1” switching signals S _{1_PWM} and S _{4_PWM} for firing and “0” switching signals S _{2_PWM} and S _{3_PWM for firing} . Is generated. The switching signals S _{2_PWM} and S _{3_PWM} are inverted signals of the switching signal S _{1_PWM} or S _{4_PWM} .

When the comparison result is S _car > D, the switching signal generation unit 208 includes “0” switching signals S _{1_PWM} and S _{4_PWM} for _extinction and “1” switching signals S _{2_PWM} and S _{3_PWM for firing} . Is generated.

The switching signals S _{1_PWM to} S _{4_PWM} are individually output from the switching signal generator 208 to the gate terminals of the switches S _{1 to} S ₄ .

  Next, the operation of the power conversion device configured as described above will be described.

As shown in FIG. 1, the circuit configuration is a configuration in which an AC power source V1, an inductor L _f , a first inductor L ₁ , H bridge connection points M ₁ and M ₂ , and a transformer primary winding T _n1 are connected in series. . A capacitor C _f is connected in parallel to the AC power source V1 and the inductor L _f . The inductor L _f and the capacitor C _f constitute a low pass filter.

Therefore, as shown in FIG. 6, the H bridge output voltage V _ab and the AC power supply are connected to the inductor L _f , the first inductor L ₁ and the transformer primary winding T _n1 connected in series with the H bridge and the AC power supply V1. the difference between the voltage V _AC is applied.

At this time, in a direction to cancel the AC power source voltage V _AC from the H-bridge will be described when outputting the AC power source voltage V output voltage V _ab of _AC equivalent exchange.

The H bridge output voltage V _ab includes a harmonic component due to switching and a low frequency component of 50 Hz or 60 Hz of the output voltage. The low frequency component of the H-bridge output voltage V _ab has the same frequency of 50 Hz and 60 Hz as the AC power supply V1, but the switching frequency of each of the switches S _{1 to} S ₄ is several tens to several hundreds of kilohertz which is sufficiently higher than that. Is used.

A difference between the _AC power supply voltage V _AC and the H-bridge output voltage V _ab is applied to the inductor L _f , the first inductor L _1, and the transformer primary winding T _n1 that are connected in series. However, during the H-bridge and an inductor L _f capacitor C _f it is connected.

For inductor L _f, the high frequency component of the H-bridge output voltage V _ab is for attenuating by a capacitor C _f, H-bridge output voltage of 50Hz or 60Hz contained in V _ab between the low frequency component and an AC power source voltage V _AC The difference is applied. Since the inductor L _f is an inductance having a value sufficiently larger than the exciting inductance appearing in the first inductor L ₁ and the transformer primary winding T _n1 , most of the low frequency component is in the inductor L _f . Applied.

The magnitude of the difference between the low-frequency component of the H-bridge output voltage V _ab applied to the inductor L _f and the AC power supply voltage V _AC can be arbitrarily controlled by adjusting the H-bridge output voltage V _ab . Therefore, the AC power supply current I _AC flowing through the inductor L _f , that is, the magnitude of the current flowing from the AC power supply V1 into the power converter 100 can be controlled by manipulating the low frequency component of the H bridge output voltage V _Ab .

The control unit 200 uses each of the switching signals S _{1_PWM to} S _{4_PWM} so as to flow an AC power supply current I _AC synchronized with the AC power supply voltage V _AC in order to cause the main circuit unit to perform a rectifier and PFC operation using this characteristic. Supply to switches S _{1 to} S ₄ .

On the other hand, the difference between the AC power supply voltage V _AC and the H-bridge output voltage V _ab is similarly applied to the first inductor L ₁ and the transformer primary winding T _n1 . However, the H bridge outputs an AC output voltage V _ab having a magnitude that cancels the _AC power supply voltage V _AC .

Therefore, a high frequency component of the H bridge output voltage V _ab is applied to the first inductor L ₁ and the transformer primary winding T _n1 . The high frequency component applied to the transformer primary winding T _n1 generates an excitation voltage in the transformer secondary winding T _n2 . This excitation voltage is rectified and smoothed by the diodes D ₅ and D ₆ and the capacitor C _out and supplied to the load 5 as a DC output voltage V _out . In this way, the power conversion device 100 operates as a power supply device for AC / DC conversion. The “power conversion device” may be called a “power supply device”.

Here, the excitation voltage generated in the transformer secondary winding T _n2 is determined by the magnitude of the high frequency component applied to the transformer primary winding T _n1 . The high frequency component of the H-bridge output voltage V _ab is applied to the transformer primary winding T _n1 and the first inductor L ₁ . For this reason, the harmonic component applied to the transformer primary winding T _n1 is a voltage divided by the first inductor L ₁ .

Since the high frequency component applied to the first inductor L ₁ and the transformer primary winding T _n1 is a high frequency voltage generated by switching of the H bridge, the frequency of this high frequency voltage is the switching frequency of each of the switches S _{1 to} S ₄ . It depends on.

Since the first inductor L ₁ has a constant inductance, the first inductor L ₁ has a characteristic that the impedance changes depending on the frequency of the applied voltage.

Using this characteristic, the control unit 200 adjusts the voltage applied to the transformer primary winding T _n1 by adjusting the voltage V _L1 applied to the first inductor L ₁ by controlling the switching frequency, and outputs it. The output voltage _Vout is controlled so as to eliminate the deviation from the voltage command value _{Vout_ref} .

In summary, the control unit 200 determines the AC power supply current I synchronized with the AC power supply voltage V _AC based on the detected value of the AC power supply voltage V _AC , the detected value of the AC power supply current I _AC , and the detected value of the capacitor voltage V _DC. as flow _AC, and based on the detected value of the output voltage V _out, so as to eliminate the deviation V _{Out_dif} between the output voltage command value V _{Out_ref} for the output voltage V _out, the first switch S ₁ and the fourth switch S Switching signals S _{1_PWM to} S _{4_PWM} for alternately opening and closing the set of _{4 and} the set of the second switch S ₂ and the third switch S ₃ are _supplied to the first switch S _{1 to} the fourth switch S ₄ .

Thereby, the control unit 200 performs the rectifier and the PFC operation so as to flow the _AC power supply current I _AC synchronized with the AC power supply voltage V _AC and eliminates the deviation V _{out_dif} from the output voltage command value V _{out_ref.} The output voltage _Vout is controlled.

Here, the charging and voltage control method thereof to the capacitor C _DC of the H-bridge supplementary explained.

At start-up, the capacitor _CDC is not charged at all.

When the AC power supply V1 is connected to the power conversion apparatus 100 in this state, the AC power supply voltage V _AC is applied to the H bridge output via the inductors L _f and L ₁ . Since state capacitor voltage V _DC is no, the diode D ₁ to D ₄ connected in parallel to each switch S ₁ to S ₄ by the AC power source voltage V _AC to be applied is fired, the relative capacitor C _DC Diodes D _{1 to} D ₄ operate as a full-wave rectifier circuit.

Thereby, the capacitor _CDC is charged to about the peak value of the _AC power supply voltage V _AC . When the capacitor _CD is charged to the _AC power supply voltage V _AC , the H bridge can output the output voltage V _ab corresponding to the _AC power supply voltage V _AC . As described above, H for operating the bridge output voltage V _ab is controllable AC power source current I _AC flowing through the inductor L _f, the AC power source voltage V _AC and controlling the AC power source current I _AC so that the same phase By doing so, the active power can be supplied to the power conversion apparatus 100.

Here, the inductor L _f has an impedance of only several percent with respect to the rated capacity of the power conversion device 100. For this reason, the voltage applied to the inductor L _f for controlling the _AC power supply current I _AC is about several percent of the _AC power supply voltage V _AC .

H-bridge output voltage V _ab includes a component that cancels _AC power supply voltage V _AC and a voltage that is applied to inductor L _f . Sine for the voltage applied to the inductor L _f is about several% of the AC supply voltage V _AC, the majority of the H-bridge output voltage V _ab is the serve as component for canceling an AC supply voltage V _AC, approximating to the AC power source voltage V _AC Wave voltage.

Therefore, most of the active power obtained by controlling the AC power supply current I _AC having the same phase as the AC power supply voltage V _AC to flow into the power conversion device 100 is supplied to the H bridge. Therefore, when the AC power supply current I _AC is supplied in the direction in which power is supplied from the AC power supply V1 to the H bridge, the capacitor voltage V _DC of the H bridge increases.

On the other hand, when the AC power supply current I _AC is supplied in a direction in which power is supplied from the H bridge to the AC power supply V1, the capacitor voltage V _DC of the H bridge decreases.

In this way, the charging of the H bridge capacitor C _DC and the capacitor voltage V _DC are controlled by controlling the _AC power source current I _AC in phase with the AC power source voltage V _AC .

As described above, according to the present embodiment, the _AC synchronized with the _AC power supply voltage V _AC based on the detected value of the AC power supply voltage V _AC , the detected value of the AC power supply current I _AC , and the detected value of the capacitor voltage V _DC. The switching signals S _{1_PWM to} S _{4_PWM} are switched to the switches S ₁ to S so that the power supply current I _AC flows and the deviation from the output voltage command value V _{out_ref} is eliminated based on the detected value of the output voltage V _out. supplied to the S _4. In this case, to determine the switching frequency based on the basis of the detected value of the output voltage V _out to the size of the deviation between the output voltage command value V _{Out_ref,} generates switching signal S _{1_PWM} ~S _{4_PWM} in accordance with the switching frequency .

  Thereby, the conversion efficiency of the entire apparatus can be improved while having the functions of the rectifier circuit, the PFC circuit, and the DC / DC conversion circuit.

Supplementally, the AC power supply current I _AC is controlled by manipulating the low frequency component of the H bridge output voltage V _ab by connecting the AC power supply V1 and the inductors L _f , L ₁ , H bridge, and transformer in series. Thus, the function of the PFC circuit can be realized, and the capacitor voltage V _DC of the H bridge can be controlled using the _AC power supply current I _AC , thereby realizing the function of the rectifier circuit.

Further, by changing the switching frequency of the H bridge, the output voltage V _out supplied to the load 5 can be controlled, and the function of the DC / DC conversion circuit can be realized.

  As described above, this embodiment can realize three circuit functions such as a conventional rectifier circuit, a PFC circuit, and a DC / DC converter circuit with one circuit stage number (H bridge), and achieves low loss, improved conversion efficiency, and reduced component count. Reduction can be realized.

<Second Embodiment>
Next, a power converter according to a second embodiment will be described with reference to FIGS. 1 and 6.

The second embodiment describes in detail the portion of the first embodiment that controls the output voltage _Vout supplied to the load 5 by controlling the switching frequency. In the following, parts different from the first embodiment will be mainly described.

As described above, the voltage applied to the first inductor L ₁ and the transformer primary winding T _n1 is a high-frequency voltage generated by the switching of the H bridge, and the frequency of this high-frequency voltage is determined by each of the switches S _{1 to} S ₄ . Varies with switching frequency.

Since the first inductor L ₁ has a constant inductance, the impedance changes depending on the frequency of the applied voltage. Therefore, when the switching frequency of the H bridge changes, the impedance of the first inductor L ₁ also changes.

On the other hand, since the exciting inductance L _m appears in the transformer primary winding T _n1, it has frequency characteristics like the first inductor L ₁ . However, since a resistive load is connected to the transformer secondary winding T _n2 , the impedance of the transformer when viewed from the transformer primary winding T _n1 depends on the frequency of the voltage applied to the transformer primary winding T _n1 . It does n’t change that much. This is compared to the impedance of the excitation inductance L _m at the switching frequency, the load resistance is because sufficiently small.

Therefore, increasing the switching frequency, a first impedance of the inductor L ₁ becomes relatively larger than the transformer of the impedance, the high-frequency voltage applied to the transformer primary winding T _n1 decreases.

Conversely, lowering the switching frequency, a first impedance of the inductor L ₁ becomes relatively small, high frequency voltage applied to the transformer primary winding T _n1 rises.

Therefore, the output voltage _Vout supplied to the load 5 can be controlled by changing the switching frequency of the H bridge and adjusting the high frequency voltage applied to the transformer primary winding T _n1 .

  As described above, according to the present embodiment, it is possible to obtain the same functions and effects as those of the first embodiment.

<Third Embodiment>
FIG. 7 is a schematic diagram illustrating a configuration of a switching signal generation unit applied to the power conversion device according to the third embodiment.

  The third embodiment is a specific example or a modification of the first or second embodiment, and describes a part related to the switching signal generation unit 208 in detail. Hereinafter, parts different from the first or second embodiment will be mainly described.

Here, the switching signal generation unit 208 includes a V _{out_ref} setting unit 221, a subtraction unit 222, a voltage controller (AVR) 223, a carrier generation unit 224, a comparator 225, a NOT circuit 226, a constant multiplication unit 227, a comparator 228, and a NOT circuit. 229.

Here, the V _{out_ref} setting unit 221 _outputs a preset voltage command value V _{out_ref} of the capacitor C _out to the subtraction unit 222.

The subtraction unit 222 subtracts the voltage command value V _{out_ref} received from the V _{out_ref} setting unit 221 from the V _out detection value received from the output voltage detection unit 4 and obtains the obtained deviation V _{out_dif} (= V _out −V _{out_ref} ). Is output to the voltage controller (AVR) 223.

Voltage controller (AVR) 223, based on the deviation V _{Out_dif} received from the subtraction unit 222, a PI calculation as shown in FIG. 17, to generate a carrier frequency f _c, the output carrier frequency f _c to the carrier generator 224 To do.

Carrier generator 224 generates a carrier signal S _car of the triangular wave carrier having a carrier frequency f _c received from voltage controller (AVR) 223, and outputs the carrier signal S _car to the inverting input terminal of the comparators 225,228 To do.

The comparator 225 compares the modulation command value D received at the non-inverting input terminal from the division unit 207 with the carrier signal S _car received at the inverting input terminal from the carrier generation unit, and “1” or “ The switching signal S _{1_PWM} having a value of “0” is output to the first switch S ₁ and the NOT circuit 226.

The NOT circuit 226 inverts the switching signal S _{1_PWM} received from the comparator 225 to generate the switching signal S _2, and outputs the switching signal S _{2_PWM} to the second switch S ₂ .

  The constant multiplication unit 227 multiplies the modulation command value D received from the division unit 207 by a constant “−1”, and outputs the obtained modulation command value “−D” to the non-inverting input terminal of the comparator 228.

The comparator 228 compares the modulation command value “−D” received at the non-inverting input terminal from the constant multiplication unit 227 and the carrier signal S _car received from the carrier generation unit at the inverting input terminal. The switching signal S _{3_PWM} having a value of “0” or “1” is output to the third switch S ₃ and the NOT circuit 229.

The NOT circuit 229 inverts the switching signal S _{3_PWM} received from the comparator 228 to generate the switching signal S _4, and outputs this switching signal S _{4_PWM} to the fourth switch S ₄ .

  Next, a method for obtaining a desired voltage at the output of the H-bridge by the switching signal generator 208 configured as described above will be described.

As described above, the H-bridge output voltage V _ab is obtained using the capacitor voltage V _DC as a power source. Therefore, in the control unit 200, the H bridge output voltage command value V _{ab_ref} is divided by the V _DC detection value received from the capacitor voltage detection unit 3, and the obtained modulation command value D (= V _{ab_ref} / V _DC ) is obtained. This is output to the switching signal generator 208.

The switching signal generation unit 208 uses a carrier comparison method as a method of obtaining a pulse wave switching signal from the modulation command value D. For example, with the configuration shown in FIG. 7, a carrier signal S _car composed of a triangular wave carrier that periodically changes in the range of −1 to +1 is generated, and the carrier signal S _car and the modulation command value D are compared by the comparators 225 and 228. To do.

The larger the modulation command value D than the carrier signal S _car, the switching signal S _{1_PWM} having a value of "1", outputs an S _{4_PWM,} the switching signal S _{2_PWM} having a value of "0", and outputs the S _{3_PWM} .

The smaller the modulation command value D than the carrier signal S _car, the switching signal S _{1_PWM} having a value of "0", and outputs the S _{4_PWM,} the switching signal S _{2_PWM} having a value of "1", and outputs the S _{3_PWM} .

In the power conversion apparatus 100, in response to the switching signal S _{1_PWM} ~S _{4_PWM,} by igniting the switches S ₁ to S ₄ of the H-bridge at "1", extinguished by "0", the desired H The bridge output voltage V _ab is obtained.

Further, by changing the period of the carrier signal _{S car (= 1 / f c} ), it is possible to vary the switching frequency of the switches S ₁ to S _4.

Case of only changing the carrier frequency f _c, has no effect on the H-bridge output voltage V _ab. Therefore, by changing the carrier frequency f _c , the switching frequency can be changed without affecting the low frequency component of the H-bridge output voltage V _ab (that is, the component corresponding to the _AC power supply current I _AC ). it is possible to control the output voltage V _out.

  As described above, according to the present embodiment, it is possible to obtain the same functions and effects as those of the first embodiment.

<Fourth Embodiment>
FIG. 8 is a schematic diagram illustrating a power conversion apparatus according to the fourth embodiment and its peripheral configuration.

The fourth embodiment is a modification of the first to third embodiments, and switches each of the switches S _{1 to} S ₄ based on a first inductor current I _L1 flowing through the first inductor L _1. It is.

Here, in the power converter 200a, in the configuration shown in FIG. 1, instead of the power source current detecting unit 2 (power source current detecting means) that detects the _AC power source current I _AC , the first current flowing through the first inductor L ₁ is used. A circuit current detection unit 2a (first current detection means) for detecting the inductor current I _L1 is provided. Incidentally, the circuit current detector 2a, is connected in series between the first inductor L ₁ and the connection point M _1, detects the first inductor current I _L1 flowing in the first inductor L _1, the I _L1 detection value It outputs to the control part 200a.

Accordingly, the control unit 200a performs control based on the I _L1 detection value instead of the I _AC detection value.

For example, as shown in FIG. 9, such a control unit 200a includes a V _{DC_ref} setting unit 201, a subtraction unit 202, a voltage controller (AVR) 203, an AC command value calculation unit 204, a V _{DC_ref} setting unit 231, a differential amplification. Part 232, width adjusting part 233, comparators 234 and 235, SR-F / F 236, and NOT circuit 237.

Here, the V _{DC_ref} setting unit 201, the subtraction unit 202, the voltage controller (AVR) 203, and the AC command value calculation unit 204 have the same functions as described above. However, the output destination of the current command value I _{AC_ref} obtained by the AC command value calculation unit 204 is the width adjustment unit 233. The V _{DC_ref} setting unit 201, the subtracting unit 202, and the voltage controller (AVR) 203 calculate an amplitude command value for calculating the amplitude command value I _{AC_amp_ref} of the _AC power supply current I _AC based on the detected value of the capacitor voltage V _DC. Means. Of the AC command value calculating section 204, for example PLL211 and sine wave generating unit 212 shown in FIG. 3, based on the detected value of the AC power supply voltage V _AC, a sine wave sin .omega.t of the AC power source voltage V _AC in phase The sine wave generation means to generate is comprised. Further, for example, the multiplication unit 213 illustrated in FIG. 3 in the AC command value calculation unit 204 calculates a current command value I _{AC_ref} of the AC power supply current based on the amplitude command value I _{AC_amp_ref} and the sine wave sin ωt. It constitutes a value calculation means.

The V _{out_ref} setting unit 231 _outputs a preset voltage command value V _{out_ref} of the capacitor C _out to the differential amplification unit 232.

The differential amplifying unit (width sending means) 232 detects a deviation between the voltage command value V _{out_ref} received from the V _{out_ref} setting unit 231 and the V _out detection value received from the output voltage detecting unit 4, and responds to the deviation. The value is output to the width adjustment unit 233 as the width dif1.

The width adjusting unit 233 constitutes command value generating means for generating the arc extinguishing command value envUP and the ignition command value envDN by giving the current command value I _{AC_ref} a width dif1.

Specifically, as shown in FIG. 10, the width adjusting unit 233 adds and subtracts the width dif1 received from the differential amplifier 232 to the current command value I _{AC_ref} received from the AC command value calculating unit 204. The arc extinguishing command value and the firing command value are generated. The extinguishing command value is a value envUP obtained by adding the width dif1 to the current command value I _{AC_ref} , and the firing command value is a value envDN obtained by subtracting the width dif1 from the current command value I _{AC_ref} .

envUP = I _{AC_ref} + dif1
envDN = I _{AC_ref} −dif1
The width adjusting unit 233 outputs the firing command value envDN to the non-inverting input terminal of the comparator 234 and outputs the extinguishing command value envUP to the inverting input terminal of the comparator 235.

Comparator 234 and 235, the AC power source current using the detection value of the first inductor current I _L1 instead of the detection value of the I _AC, the detection value extinguishing command value envUP and firing command value of the first inductor current I _L1 EnvDN The determination means which determines whether it is settled in the range is established.

Specifically, the comparator 234 compares the ignition command value envDN received at the non-inverting input terminal from the width adjustment unit 233 with the I _L1 detection value received at the inverting input terminal from the circuit current detection unit 2a. _{When the L1} detection value is lower than the firing command value envDN, a set signal _{S_set} having a value of “1” is output to the SR-F / F 236. Specifically, I _L1 <When EnvDN, the set signal S _{_set} of "1" is output, I _L1> when EnvDN, the set signal S _{_set} "0" is output.

The comparator 235 compares the arc extinguishing command value envUP received at the inverting input terminal from the width adjustment unit 233 with the I _L1 detection value received at the non-inverting input terminal from the circuit current detection unit 2a, and the I _L1 detection value is erased. When it is higher than the arc command value envUP, a reset signal _{R_reset} having a value of “1” is output to the SR-F / F 236. Specifically, when I _L1 > envUP, a reset signal _{R_reset} of “1” is output, and when I _L1 <envUP, a reset signal _{R_reset} of “0” is output.

The SR-F / F 236 and the NOT circuit 237 have a timing at which the detected value of the first inductor current I _L1 is out of the range (envUP ≦ I _L1 ≦ envDN) according to the determination result by the determination means (234, 235). The second generation means for generating the switching signals S _{1_PWM to} S _{4_PWM} so as to switch between opening and closing is configured.

Specifically, the SR-F / F 236 sends the switching signals S _{2_PWM} , S _{3_PWM} to the NOT circuit 237 and the second and third switches S ₂ , S ₂ , _responsive to the signals _{S_set} , _{R_reset} received from the comparators 234, 235. and outputs it to the S _3.

Specifically, when _{S _set = 0, R _reset =} 0, do not alter the switching signal S _{2_PWM,} the status of the S _{3_PWM.}

When _{S_set} = 1 and _{R_reset} = 0, “1” switching signals S _{2_PWM} and S _{3_PWM} are output.

When _{S_set} = 0 and _{R_reset} = 1, switching signals _{S2_PWM} and _{S3_PWM} of “0” are output.

The NOT circuit 237 generates the switching signals S _{1_PWM} and S _{4_PWM} by inverting the switching signals S _{2_PWM} and S _{3_PWM} received from the SR-F / F ₂₃₆ , and _{uses the} switching signals S _{1_PWM} and S _{4_PWM} as the first and fourth switches. Output to S ₁ and S ₄ .

  Next, the operation of the power conversion device configured as described above will be described.

As described above, the difference between the AC power supply voltage V _AC and the H-bridge output voltage V _ab is applied to the first inductor L ₁ and the transformer primary winding T _n1 . That is, the voltage V _L1 applied to the first inductor L ₁ changes according to the H-bridge output voltage V _ab, and as a result, the first inductor current I _L1 flowing through the first inductor L ₁ changes. Here, a method of switching each of the switches S _{1 to} S ₄ of the H bridge according to the change of the first inductor current I _L1 will be described.

In order to generate the switching signals S 1 — _{PWM to} S 4 — _PWM , an I _L1 detection value indicating an instantaneous value of the first inductor current I _L1 flowing in the first inductor L ₁ connected in series to the H bridge is used. The detected value of the first inductor current I _L1 is positive in the direction flowing from the AC power supply V1 toward the H bridge. The first inductor current detection value is compared with the current desired to flow through the first inductor L ₁ , that is, the current command value I _{AC_ref of the AC} power supply current I _AC flowing into the power converter 100. If the first inductor current I _L1 of the first inductor L ₁ is smaller _than the current command value I _{AC_ref} , the H bridge is switched in the direction in which the first inductor current I _L1 increases.

Taking FIG. 8 as an example, the second and third switches S ₂ and S ₃ are fired, and the first and fourth switches S ₁ and S ₄ are extinguished. When the first inductor current value I _L1 reaches the current command value I _{AC_ref} , the second and third switches S ₂ and S ₃ are extinguished, the first and fourth switches S ₁ and S ₄ are fired, Switching is performed in a direction that reduces the first inductor current I _L1 .

However, if the switching operation is determined by comparing one current command value I _{AC_ref} with the detected value I _{L1 of} the first inductor current I _L1 , a continuous switching operation occurs and oscillation occurs.

Therefore, the control unit 200a prepares an ignition command value envDN for igniting the second and third switches S ₂ and S ₃ and an arc extinguishing command value envUP for extinguishing the first inductor current. By comparing with the detected value of I _L1 , the timing for starting and extinguishing the switch is widened.

Here, the firing command value envDN and the extinguishing command value envUP are values _having a deviation dif1 with respect to the input current command value I _{AC_ref} . This is a so-called hysteresis comparator configuration.

By using the configuration of the hysteresis comparator, the first inductor current I _L1 shown in FIG. 10 is switching-controlled so as to return between the two command values envUP and envDN while following the current command value I _{AC_ref} . Such a first inductor current I _L1 includes a low frequency component of the current command value I _{AC_ref and} a high frequency component due to aliasing.

The control unit 200a constantly compares the command values envUP, envDN and the I _L1 detection value. For example, if the control unit 200a detects that the I _L1 detection value is outside the extinguishing command value envUP, the four switches S _{1 to} S The state of ₄ is inverted, and the slope of the first inductor current I _L1 is inverted.

In addition, when the control unit 200a detects that the detected value of I _L1 is out of the ignition command value envDN, the control unit 200a reverses the states of the _four switches S _{1 to} S ₄ again, and sets the slope of the first inductor current I _L1 . Invert again.

Accordingly, the control unit 200a can control the H-bridge output voltage V _ab by generating the H-bridge switching signals S 1 — _{PWM to} S 4 — _PWM so as to follow the current command value I _{AC — ref} . At this time, the switching frequency can be adjusted by adjusting the deviations dif1 between the ignition command value envDN and the extinguishing command value envUP, and the output voltage _Vout can be controlled.

As described above, according to the present embodiment, the configuration in which the switches S _{1 to} S ₄ are switched based on the first inductor current I _L1 flowing through the first inductor L ₁ is the same as that of the first embodiment. An effect can be obtained.

<Fifth Embodiment>
FIG. 11 is a schematic diagram showing a power conversion device according to the fifth embodiment and its peripheral configuration.

The fifth embodiment is a modification of the first to third embodiments, instead of the output voltage detecting unit 4 for detecting the output voltage V _DC of the capacitor C _DC, a voltage detection provided in trans A circuit (second voltage detection means) is provided. The fifth embodiment may be a modification of the fourth embodiment, but here, a case of a modification of the first to third embodiments will be described as an example.

Here, the voltage detection circuit includes a transformer tertiary winding T _n3 electromagnetically coupled to the transformer primary winding T _n1 and a second rectifier (D ₇₎ connected to the transformer tertiary winding T _n3. , D _8, C _det) and ,, second output voltage detection unit 4a that detects an output voltage V _out of _the rectifying means of the output voltage (V _out ') the first rectifying means _{(D 5, D 6, C} out) (Second voltage detector).

Specifically, the anode terminals of the seventh diode D ₇ and the eighth diode D ₈ are connected to both ends of the transformer tertiary winding T _n3 , respectively. The cathode terminals of the seventh diode D ₇ and the eighth diode D ₈ are connected to each other. A connecting point of each cathode terminal, between the anode terminal of the eighth diode D _8, capacitor C _out is connected. At both ends of the capacitor C _det, the output voltage detection unit 4a that detects a voltage V _out 'of the capacitor C _det are connected in parallel.

The output voltage detector 4 a detects the voltage V _out ′ of the capacitor C _det and outputs the detected value V _out ′ to the controller 200. Control unit 200, converted to V _out detection value due constant multiplier section V _out 'detection value (not shown), based on the V _out detection values obtained, operates as before. Depending on the design value of the voltage detection circuit provided in the transformer, the control unit 200 inputs the V _out ′ detection value as the V _out detection value, and operates in the same manner as described above based on the V _out detection value. It is also possible to do.

According to the above configuration, the voltage V _out ′ obtained by rectifying the voltage of the transformer tertiary winding T _n3 is fed back to the control unit 200 to change the switching frequency, and the output voltage supplied to the load 5 V _out is controlled. The voltage appearing on the secondary and tertiary sides of the transformer is determined by the voltage applied to the transformer primary side. Therefore, each voltage appearing on the secondary side and the tertiary side of the transformer exhibits the same behavior.

Therefore, the output voltage V _out supplied to the secondary load 5 is indirectly detected by detecting the voltage V _out ′ obtained by rectifying and smoothing the voltage of the transformer tertiary winding T _n3. Is possible.

According to the present embodiment as described above, instead of the output voltage detecting unit 4 for detecting the output voltage V _DC of the capacitor C _DC, even with a circuit for voltage detection is provided in the transformer, the first to The same effects as those of the fourth embodiment can be obtained.

In addition to this, in general, since the control unit is often configured with the primary side potential as a reference, components such as an insulation amplifier are required to detect the secondary side voltage _Vout . In contrast, in the present embodiment, the transformer tertiary winding T _n3 is added to detect the voltage V _out ′, and the detected value V _out ′ is input to the control unit 200.

  Therefore, according to the present embodiment, the same control as in the first to third embodiments can be realized without additional components such as an insulation amplifier.

  The present embodiment is not limited to the first to third embodiments, and can be similarly applied when applied to the fourth embodiment. In this case, the same function and effect can be obtained.

<Other embodiments>
In each embodiment, the single-phase AC power supply V1 is used, but the AC power supply V1 is not limited to a single phase. It is also possible to use a three-phase or higher-phase AC power source.

In each embodiment, FETs (field effect transistors) S1 to S4 are used as switches. However, the present invention is not limited to this. For example, a semiconductor element such as a bipolar transistor, IGBT, GaN, or SiC may be used. Alternatively, the H-bridge switches S _{1 to} S ₄ may be configured by a combination of a mechanical switch such as a relay and a diode.

In addition, although some embodiment of this invention was described, these embodiment is shown as an example and is not intending limiting the range of invention. These novel embodiments can be implemented in various other forms, and various omissions, replacements, and changes can be made without departing from the scope of the invention. These embodiments and modifications thereof are included in the scope and gist of the invention, and are included in the invention described in the claims and the equivalents thereof.
Hereinafter, the invention described in the scope of claims of the present application will be appended.
[1] A low-pass filter having a filter inductor and a filter capacitor connected in series to an AC power source to form a closed loop, and a first inductor having one end connected to a connection point between the filter inductor and the filter capacitor A transformer having a transformer primary winding having one end connected to a connection point between the AC power supply and the filter capacitor; and a transformer secondary winding electromagnetically coupled to the transformer primary winding; A first switch and a second switch are connected in series via the other end of the first inductor, and a third switch and a fourth switch are connected in series via the other end of the transformer primary winding. The first switch, the third switch, the second switch, and the fourth switch are connected to form a closed loop, and the third switch And an H bridge formed by connecting a smoothing capacitor to both ends of the fourth switch, a first rectifying means connected to the transformer secondary winding, and a power supply voltage detecting means for detecting an AC power supply voltage of the AC power supply. A power supply current detecting means for detecting an AC power supply current flowing to the AC power supply, a capacitor voltage detecting means for detecting a capacitor voltage of the smoothing capacitor, and an output voltage output from the first rectifying means to the load. Based on an output voltage detection means, a detection value of the AC power supply voltage, a detection value of the AC power supply current, and a detection value of the capacitor voltage, an AC power supply current synchronized with the AC power supply voltage is caused to flow, and Based on the detected value of the output voltage, the set of the first switch and the fourth switch and the second switch so as to eliminate the deviation from the output voltage command value with respect to the output voltage. A control means for supplying a switching signal for alternately opening and closing the switch and the third switch set to the first switch and the fourth switch set and the second switch and the third switch set; And the control means determines a switching frequency based on a deviation from the output voltage command value based on the detected value of the output voltage, and generates a switching signal corresponding to the switching frequency. A power conversion device comprising means.
[2] In the power conversion device according to attachment 1, the signal generation unit calculates a carrier frequency based on a magnitude of deviation from the output voltage command value based on the detected value of the output voltage. A power conversion device comprising: means; carrier generation means for generating a carrier signal having the carrier frequency; and first generation means for generating the switching signal based on the carrier signal.
[3] In the power conversion device according to attachment 1, in place of the power supply current detection unit, the power conversion device includes first current detection unit that detects a first inductor current flowing in the first inductor, and the control unit includes the capacitor Amplitude command value calculation means for calculating an amplitude command value of the AC power supply current based on the detected voltage value, and a sine wave in phase with the AC power supply voltage based on the detected value of the AC power supply voltage A sine wave generating means; a current command value calculating means for calculating a current command value of the AC power supply current based on the amplitude command value and the sine wave; and a detection value of the output voltage and the output voltage command value. Width sending means for detecting a deviation and sending a value corresponding to the deviation as a width; command value generating means for generating the extinguishing command value and the firing command value by giving the current command value the width; AC power supply The determination value for determining whether or not the detected value of the first inductor current falls within the range between the extinction command value and the ignition command value is used instead of the detected value of the first inductor current. And second generation means for generating the switching signal so as to switch the opening and closing at a timing when the detected value of the first inductor current deviates from the range according to the result of the determination. A power conversion device.
[4] In the power conversion device according to any one of appendix 1 to appendix 3, a second voltage detection unit is provided instead of the output voltage detection unit, and the second voltage detection unit includes the transformer primary. A transformer tertiary winding electromagnetically coupled to the winding, a second rectifier connected to the transformer tertiary winding, and an output voltage of the second rectifier detected as an output voltage of the first rectifier And a second voltage detector.

DESCRIPTION OF SYMBOLS 1 ... Input voltage detection part, 2 ... Power supply current detection part, 2a ... Circuit current detection part, 3 ... Capacitor voltage detection part, 4, 4a ... Output voltage detection part, 5 ... Load, 100, 100a ... Power converter, 200 , 200a ... control unit, 201, 231 ... V _{DC_ref} setting unit, 202, 205, 222 ... subtraction unit, 203, 223 ... voltage controller (AVR), 204 ... AC command value calculation unit, 206 ... current controller (ACR) 207, a division unit, 208, a switching signal generation unit, 210, 227, a constant multiplication unit, 211, a PLL, 212, a sine wave generation unit, 213, a multiplication unit, 221, a V _{out_ref} setting unit, 224, a carrier generation unit. 225, 228, 234, 235, comparators, 226, 229, 237, NOT circuits, 232, differential amplifiers, 233, width adjusting units, 236, SR-F / F.

Claims (3)

A low-pass filter having a filter inductor and a filter capacitor connected in series to an AC power source to form a closed loop;
A first inductor having one end connected to a connection point between the filter inductor and the filter capacitor;
A transformer having a transformer primary winding having one end connected to a connection point between the AC power supply and the filter capacitor; and a transformer secondary winding electromagnetically coupled to the transformer primary winding;
A first switch and a second switch are connected in series via the other end of the first inductor, and a third switch and a fourth switch are connected in series via the other end of the transformer primary winding. The first switch, the third switch, the second switch, and the fourth switch are connected to form a closed loop, and a smoothing capacitor is connected to both ends of the third switch and the fourth switch. The bridge,
First rectifying means connected to the transformer secondary winding;
Power supply voltage detecting means for detecting an AC power supply voltage of the AC power supply;
First current detecting means for detecting a first inductor current flowing in the first inductor;
Capacitor voltage detecting means for detecting a capacitor voltage of the smoothing capacitor;
Output voltage detection means for detecting an output voltage output from the first rectification means to the load;
Based on the detected value of the AC power supply voltage, the detected value of the first inductor current , and the detected value of the capacitor voltage, the first inductor current synchronized with the AC power supply voltage is allowed to flow and the output voltage is detected. Based on the value, the set of the first switch and the fourth switch and the set of the second switch and the third switch are alternately opened and closed so as to eliminate the deviation from the output voltage command value with respect to the output voltage. Control means for supplying a switching signal for the first switch and the fourth switch, and the second switch and the third switch;
Comprising
The control means includes
Based on the detected value of the capacitor voltage, an amplitude command value calculating means for calculating an amplitude command value of the AC power supply current flowing to the AC power supply,
A sine wave generating means for generating a sine wave in phase with the AC power supply voltage based on the detected value of the AC power supply voltage;
Based on the amplitude command value and the sine wave, current command value calculation means for calculating a current command value of the AC power supply current;
A width sending means for detecting a deviation between the detected value of the output voltage and the output voltage command value, and sending a value corresponding to the deviation as a width;
Command value generating means for generating an arc extinguishing command value and an ignition command value by giving the current command value the width;
Determining means for determining whether or not the detected value of the first inductor current falls within a range between the extinction command value and the ignition command value;
Signal generating means for generating the switching signal so as to switch the opening and closing at a timing when the detection value of the first inductor current deviates from the range according to a result of the determination ;
A power conversion device comprising: A low-pass filter having a filter inductor and a filter capacitor connected in series to an AC power source to form a closed loop;
A first inductor having one end connected to a connection point between the filter inductor and the filter capacitor;
A transformer having a transformer primary winding having one end connected to a connection point between the AC power supply and the filter capacitor; and a transformer secondary winding electromagnetically coupled to the transformer primary winding;
A first switch and a second switch are connected in series via the other end of the first inductor, and a third switch and a fourth switch are connected in series via the other end of the transformer primary winding. The first switch, the third switch, the second switch, and the fourth switch are connected to form a closed loop, and a smoothing capacitor is connected to both ends of the third switch and the fourth switch. The bridge,
First rectifying means connected to the transformer secondary winding;
Power supply voltage detecting means for detecting an AC power supply voltage of the AC power supply;
A power source current detecting means for detecting an AC power source current flowing to the AC power source;
Capacitor voltage detecting means for detecting a capacitor voltage of the smoothing capacitor;
A transformer tertiary winding electromagnetically coupled to the transformer primary winding;
A second rectifying means connected to the transformer tertiary winding;
Second voltage detecting means for detecting an output voltage of the second rectifying means ;
Based on the detected value of the AC power supply voltage, the detected value of the AC power supply current, and the detected value of the capacitor voltage, the AC power supply current synchronized with the AC power supply voltage is caused to flow, and the detected value of the output voltage is set. Based on the above, the set of the first switch and the fourth switch and the set of the second switch and the third switch are alternately opened and closed so as to eliminate the deviation from the output voltage command value with respect to the output voltage. Control means for supplying a switching signal to the set of the first switch and the fourth switch and the set of the second switch and the third switch;
Comprising
The control unit includes a signal generation unit that determines a switching frequency based on a magnitude of deviation between the detected value of the output voltage and the output voltage command value and generates a switching signal according to the switching frequency. The power converter characterized by this. A low-pass filter having a filter inductor and a filter capacitor connected in series to an AC power source to form a closed loop;
A first inductor having one end connected to a connection point between the filter inductor and the filter capacitor;
A transformer having a transformer primary winding having one end connected to a connection point between the AC power supply and the filter capacitor; and a transformer secondary winding electromagnetically coupled to the transformer primary winding;
A first switch and a second switch are connected in series via the other end of the first inductor, and a third switch and a fourth switch are connected in series via the other end of the transformer primary winding. The first switch, the third switch, the second switch, and the fourth switch are connected to form a closed loop, and a smoothing capacitor is connected to both ends of the third switch and the fourth switch. The bridge,
First rectifying means connected to the transformer secondary winding;
Power supply voltage detecting means for detecting an AC power supply voltage of the AC power supply;
First current detecting means for detecting a first inductor current flowing in the first inductor;
Capacitor voltage detecting means for detecting a capacitor voltage of the smoothing capacitor;
A transformer tertiary winding electromagnetically coupled to the transformer primary winding;
A second rectifying means connected to the transformer tertiary winding;
Second voltage detecting means for detecting an output voltage of the second rectifying means ;
Based on the detected value of the AC power supply voltage, the detected value of the first inductor current , and the detected value of the capacitor voltage, the first inductor current synchronized with the AC power supply voltage is allowed to flow and the output voltage is detected. Based on the value, the set of the first switch and the fourth switch and the set of the second switch and the third switch are alternately opened and closed so as to eliminate the deviation from the output voltage command value with respect to the output voltage. Control means for supplying a switching signal for the first switch and the fourth switch, and the second switch and the third switch;
Comprising
The control means includes
Based on the detected value of the capacitor voltage, an amplitude command value calculating means for calculating an amplitude command value of the AC power supply current flowing to the AC power supply,
A sine wave generating means for generating a sine wave in phase with the AC power supply voltage based on the detected value of the AC power supply voltage;
Based on the amplitude command value and the sine wave, current command value calculation means for calculating a current command value of the AC power supply current;
A width sending means for detecting a deviation between the detected value of the output voltage and the output voltage command value, and sending a value corresponding to the deviation as a width;
Command value generating means for generating an arc extinguishing command value and an ignition command value by giving the current command value the width;
Determining means for determining whether or not the detected value of the first inductor current falls within a range between the extinction command value and the ignition command value;
Signal generating means for generating the switching signal so as to switch the opening and closing at a timing when the detection value of the first inductor current deviates from the range according to a result of the determination ;
A power conversion device comprising: