JP6023301B2 - Electronic devices, capacitance sensors and touch panels - Google Patents

Description

The present invention relates to an electronic device, a capacitance sensor and a touch panel, and more particularly to an electronic device using an AM modulation / demodulation system, a capacitance sensor and a touch panel.

A touch panel is a device that detects the position coordinates pointed to by an indicator such as a finger or a pen or the presence or absence of a pointing action. Currently, a liquid crystal display device (Liquid Crystal Display, LCD) or a plasma display device is usually used. (Plasma Display Panel, PDP), used in combination with display devices such as organic EL displays.

A user-friendly human interface is realized by inputting the output of the touch panel to a computer and controlling the display contents of the display device or the device by the computer. Currently, touch panels are widely used in daily life such as game machines, personal digital assistants, ticket vending machines, automated teller machines (ATMs), and car navigation systems. In addition, with the increasing performance of computers and the spread of network connection environments, the services provided by electronic devices have diversified, and the need for display devices equipped with touch panels continues to grow.

As one type of touch panel, there is a surface capacitance type touch panel. The surface capacitance type touch panel is connected to (a) a surface resistor and (b) the surface resistor, and an AC voltage (sinusoidal voltage) is applied to the surface resistor as excitation to flow through the surface resistor. It consists of a drive / detection circuit that measures and outputs current.

Specifically, the surface capacitance type touch panel is composed of a transparent substrate, a transparent surface resistor formed on the surface thereof, and a thin insulating film formed on the upper surface thereof. This surface resistor is called a position detection conductive film. When driving this type of touch panel, AC voltage is applied to the four corners of the position detection conductive film. When the touch panel is touched with a finger or a pointer (hereinafter referred to as a finger or the like), a capacitor is formed by the capacitance coupling between the position detection conductive film and the finger or the like. A minute current flows through this capacitor to a finger or the like. This current flows from each corner of the position detection conductive film to a point touched by a finger or the like. Based on the current detected by the drive / detection circuit, the signal processing circuit calculates the presence / absence of touch of the finger or the like and the coordinates of the touch position of the finger or the like. Specifically, the signal processing circuit detects the presence or absence of touch based on the sum of the currents at the four corners of the position detection conductive film. Further, the coordinates of the touch position are calculated based on the ratio of the currents at the four corners of the position detection conductive film.

Patent Documents 1 to 5 disclose a touch panel based on such a surface capacitance type operating principle.

According to Patent Document 1, when the display panel and the touch panel are operated in combination, an AC voltage is applied to the touch panel during the non-display period of the display panel in order to prevent a decrease in position detection accuracy due to a drive signal of the display panel. At the same time, it is provided with a counter electrode driving means for applying the same AC voltage to the counter electrode of the display panel.

Further, Patent Document 2 states, "Safety is achieved by increasing the AC voltage vibration level when the noise is large, decreasing the AC voltage vibration level when the noise is small, and switching to another voltage vibration frequency in the case of specific frequency noise. The S / N ratio is improved, the noise resistance is excellent, and the electrically safe touch panel device is shown.

In addition, in Patent Document 3, "the phase and AC voltage when a finger touches the panel is used as a contact vector signal, and the scalar amount calculated by using the cosine theorem from the phase difference and amplitude of both signals is the original finger. As the AC signal touched by, in detecting the touch position, the AC voltage due to the parasitic signal when the finger is not near the surface resistor, or the signal from the finger of the capacitive grounded human body or the resistant grounded human body. Eliminate the phase difference ".

Subsequently, Patent Document 4 states, "In the arithmetic circuit, the output of the long sensor line LSLi and the output of the short sensor line SSLi are input, and the signal component S is calculated by using the difference (Delta) and the wiring capacitance ratio Kc. "Ask" is disclosed.

On the other hand, Patent Document 5 states that "these four nodes are designated by the symbols Na, Nb, Nc, and Nd, respectively. Each terminal of the current detection circuit described later is connected to these nodes. Also disclosed, "The single-pole double-throw switches 21a to 21d are connected to the nodes Na to Nd via the current detection circuits 13a to 13d. Two contacts of the single-pole double-throw switches 21a to 21d. An AC voltage source 22 is connected to one, and a storage capacitance line drive circuit is connected to the other (that is, the node described as COM in FIG. 4). As an example, a sinusoidal wave is used as the waveform of the AC voltage. Can be done. "

JP-A-2007-334606 Japanese Unexamined Patent Publication No. 2006-106853 Japanese Unexamined Patent Publication No. 2010-86285 Japanese Unexamined Patent Publication No. 2011-13757 Japanese Unexamined Patent Publication No. 2011-14109

The following analysis was made by the inventor of the present invention. The touch panel described in Patent Document 1 has the following five problems.

The first problem is that it is vulnerable to external noise (change in electric field, noise with capacitance coupling). Although Patent Document 1 states that the position detection accuracy is prevented from being lowered by the drive signal of the display panel, external noise derived from other than the drive signal of the display panel, for example, is arranged above the touch surface of the touch panel. It is susceptible to noise emitted from fluorescent lamps that include an inverter circuit.

One of the reasons is based on the operating principle of the touch panel. That is, since the surface capacitance type touch panel detects the capacitance of the capacitor formed between the position detection conductive film and the finger, the electromagnetic field is shielded between the position detection conductive film and the finger. It is not possible to form a shield electrode for this purpose. Therefore, the touched surface of the position detection conductive film must have a structure that is vulnerable to external noise. Then, as the size of the touch panel increases, it becomes more susceptible to external noise.

Another reason is that the number of noise sources is increasing. For example, inverter-type fluorescent lamps developed to reduce flicker have been accepted by the market and their number is increasing. Alternatively, in chargers and AC adapters for mobile devices, switching power supplies developed for improving the conversion efficiency of power supply voltage have come to be widely adopted. The noise generated by these devices interferes with the normal operation of the devices that detect capacitance.

The second problem is that the noise cannot be removed by the bandpass filter when the excitation frequency of the touch panel and the frequency of the noise match or are close to each other.

The fundamental frequency of the noise or the frequency of its harmonics exemplified above is equal to or near the excitation frequency of the touch panel. On the other hand, the synchronous detection circuit described in Patent Document 1 is said to perform filtering in order to remove noise having a frequency different from the excitation frequency. Therefore, in this method of decomposing and selecting the observation signal by frequency, noise cannot be removed when the excitation frequency and the noise frequency match.

Further, when the noise frequency is near the excitation frequency, the noise is mixed through the attenuation region (or transition region) existing between the pass region and the blocking region of the bandpass filter. That is, a feasible filter has a certain limit on its frequency resolution and therefore cannot remove noise at frequencies near the excitation frequency.

The third problem is that when the touch detection operation period is limited to the non-display period (non-address period) or the like, the frequency component is reduced and noise near the frequency of the true signal cannot be removed. For example, in the case of the periodogram spectrum estimation method, when the target signal consists of two sinusoidal signals with the same amplitude,

Ｔ：信号取得期間
となるスペクトルピークを分離できるとされる。

T: It is said that the spectral peaks during the signal acquisition period can be separated.

In this case, when the signal acquisition period T is 500 microseconds, Δf is 2 kHz, and when the true signal is 100 kHz and the noise is 99 kHz, it is considered that both cannot be decomposed by frequency.

The fourth problem is that the noise removal effect by averaging is reduced, and the S / N is reduced. For example, when an observation signal with Poisson distribution noise superimposed is acquired many times and the noise is offset by averaging to reduce the noise, the amount of noise reduction is said to be proportional to the square root of the number of acquisitions. .. That is, when the signal acquisition period is limited to a short time such as a non-display period (non-address period), the noise removal effect by averaging decreases, and the S / N decreases.

The fifth problem is that when the applicant of the present application applies a structure in which a polarizing plate is present between the position detection conductive film and the finger, as shown in Japanese Patent Application No. 2009-163401, the position detection conductive film and the position detection conductive film The capacitance formed between the finger and the finger becomes smaller, and the S / N decreases. Further, when a protective glass or the like is inserted between the position detection conductive film and the finger, the S / N is similarly lowered.

Therefore, electronic devices and capacitance sensors that have the same signal frequency and noise frequency, or can remove noise in the vicinity that cannot be decomposed by conventional frequency resolution, and can accurately detect the presence or absence of touch and the touch position. And providing a touch panel is an issue.

In order to solve the above problems, the electronic device 120 according to the present invention has an excitation generating unit 102 that generates an intermittent sine wave signal with the sensor system 101 and gives it to the sensor system, and an amplitude that is an output of the sensor system. The demodulator includes a demodulator 105 that demolishes the modulated signal, and the demodulator is _one of the response x 1 (t) of the sensor system during the period when the excitation generating unit outputs a sine wave, and at least immediately before and after the response x 1 (t). , The demodulated signal D (t) is generated by using both _{the response z 1} (t) of the sensor system during the period when the excitation generating unit is not outputting the sine wave.

Further, in the electronic device of the present invention, X is a vector obtained from the amplitude and phase of the frequency component of the sine wave calculated from the response of the sensor system during the period when the excitation generator outputs the sine wave, and the excitation is performed. When the vector obtained from the amplitude and phase of the frequency component of the sine wave calculated from the response of the sensor system during the period when the generator is not outputting the sine wave is N, the demodulated signal is | X-N. It also has a demodulator that is a constant multiple of |.

Further, in the electronic device of the present invention, X is a vector obtained from the amplitude and phase of the frequency component of the sine wave calculated from the response of the sensor system during the period when the excitation generator outputs the sine wave, and immediately before that. And when the vectors obtained from the amplitude and phase of the frequency component of the sine wave calculated from the response of the sensor system during the period when the excitation generator immediately after that does not output the sine wave are Y and Z, respectively, The demodulated signal is a constant multiple of | X-M |, where M is an average vector of Y and Z.

On the other hand, in order to solve the above-mentioned problems, the capacitance sensor according to the present invention is configured to include the electronic device, and applies a voltage to the surface resistor and the surface resistor connected to the surface resistor. It has a sensor system consisting of a drive / detection circuit that applies and measures and outputs the current flowing through the surface resistor, and detects the capacitance of the capacitor formed by the surface resistor and the indicator. By doing so, the touch state or coordinates of the indicator are detected.

Further, the capacitance sensor of the present invention is configured to include the electronic device, and further includes a display device, and during the non-address period of the display device, the excitation generating unit outputs a sine wave. The demodulated signal is generated by using both the response of the sensor system during the period when the sine wave is output and the response of the sensor system during the period when the sine wave is not output. Generate.

On the other hand, in order to solve the above-mentioned problems, the touch panel according to the present invention is configured to include the electronic device, and a voltage is applied to the surface resistor and the surface resistor connected to the surface resistor. It has a sensor system consisting of a drive / detection circuit that measures and outputs the current flowing through the surface resistor, and detects the capacitance of the capacitor formed by the surface resistor and the indicator. , Detects the touch state or coordinates of the indicator.

Further, the touch panel of the present invention is configured to include the electronic device, and further includes a display device, and during the non-address period of the display device, a period during which the excitation generator outputs a sine wave and a sine wave. A demodulated signal is generated by using both the response of the sensor system during the period during which the sine wave is output and the response of the sensor system during the period during which the sine wave is not output.

Further, in order to solve the above problems, the electronic device according to the present invention includes an arithmetic amplifier, a resistor connected between the output terminal and the inverting input terminal of the arithmetic amplifier, and an inverting input terminal of the arithmetic amplifier. An electronic device that detects the capacitance of a conductor having a conductor connected to the conductor and an excitation generator that generates an intermittent sinusoidal signal and gives it to the non-inverting input terminal of the arithmetic amplifier. The demodulator includes a demodulator that demolishes the amplitude-modulated signal that is the output of the arithmetic amplifier, and the demodulator is either immediately before or after the response of the electronic device during the period when the excitation generating unit outputs a sine wave. A demodulated signal is generated by using both the response of the electronic device during the period when the excitation generating unit does not output a sine wave.

In this specification, the capacitance sensor includes a touch sensor.

In addition, although it is stated in the present specification and claims that the excitation generator outputs a sine wave, it should be noted that the output in this case is not limited to a single frequency sine wave. Keep it. All signals can be represented as sinusoidal series of different frequencies (Fourier series expansion). That is, when the excitation generator outputs, for example, a square wave, the square wave is a series of sine waves of different frequencies. In this case, the signal may be processed to obtain a demodulated signal by paying attention to the sine wave having the fundamental frequency of the square wave. As described above, even when the excitation generating unit outputs a rectangular wave, it is included in the present invention. For the same reason, the excitation generator is included in the present invention regardless of the output of any alternating current.

By implementing the electronic device, capacitance sensor, touch sensor and touch panel according to the present invention, the following five effects can be obtained.

The first effect is that the sine wave is stopped to acquire noise, so that noise can be acquired accurately regardless of the presence or absence of a finger (presence or absence of touch).

The second effect is accurate because the "noise" signal processing path acquired by stopping the sine wave and the "true signal + noise" signal processing path acquired by applying the sine wave are the same. It is possible to acquire noise.

The third effect is to subtract the vectors of "true signal + noise" and "noise", so even if the true signal and noise have the same frequency, the true signal can be obtained accurately. Is what you can do.

The fourth effect is the front noise (noise acquired when the excitation generator stops before outputting the sine wave) and the rear noise (noise acquired when the excitation generator stops after outputting the sine wave). By using the average vector, it is possible to remove noise in the vicinity of the true signal beyond the frequency resolution.

The fifth effect is that by using the average vector of the front noise and the rear noise, the noise can be removed accurately even if the noise amplitude fluctuates.

The present invention makes it possible to provide a touch panel and an electronic device that are resistant to external noise and have a high S / N due to the effects of the above five points.

It is a block diagram of the electronic device of this invention. It is a block diagram of the capacitance sensor of this invention. It is a timing chart of the capacitance sensor of this invention. It is a graph of the sensor system output voltage for demonstrating the calculation of this invention. It is a vector figure which shows the operation of the demodulation part of this invention. It is a demodulation part input voltage for demonstrating the operation of this invention. It is a vector figure for demonstrating the demodulation part of this invention. It is a vector figure for demonstrating the demodulation part of this invention. It is a vector figure for demonstrating the demodulation part of this invention. It is a block diagram of the capacitive touch panel of this invention. It is a timing chart of the capacitive touch panel of this invention. It is a signal waveform of the capacitive touch panel of this invention. It is a block diagram explaining the signal processing of the capacitive touch panel of this invention. It is a signal waveform of the capacitive touch panel of this invention.

(Embodiment 1)
The capacitance sensor of the present invention will be described. A general capacitance sensor realizes its function by omitting the position detection function from the touch panel function shown in the background technology. Since the position detection function is omitted, it is possible to use a surface conductor or simply a conductor instead of the surface resistor.

(Constitution)
FIG. 2 shows a block diagram of the capacitance sensor 100 of the present invention, and FIG. 1 shows a block diagram of the electronic device 120 of the present invention, which is an abstraction of the capacitance sensor of the present invention. Capacitance sensor 100 shown in FIG. 2 is configured to detect the capacitance of the capacitor C _in described in FIG. The capacitance sensor as input capacitance and the excitation of the capacitor C _in, a sensor system 101 for outputting a signal corresponding to the capacitance of the capacitor C _in, the excitation generating unit 102 for generating a vibration該励, It has a sinusoidal wave generation unit 103 connected to an excitation generation unit and a DC generation unit 104. The output of the sensor system is input to the demodulation unit 105, and the demodulation unit generates a demodulation signal.

The excitation generator generates an intermittent sinusoidal signal. As illustrated in FIG. 2, the means for generating an intermittent sine wave signal includes a sine wave generation unit 103 and a DC generation unit 104, and there is a means for switching between them. However, it is not limited to this means. As another means, for example, a DA converter may be used, and the digital signal given to the DA converter may be a signal obtained by discretizing an intermittent sine wave.

The sensor system is composed of an operational amplifier 110, a resistor R _f inserted in the feedback path thereof, and a capacitor C _f, and further includes an adder 111 that subtracts the output voltage and the excitation voltage of the operational amplifier 110.

Assuming that this operational amplifier 110 is an ideal operational amplifier, the excitation voltage input to the sensor system 101 is V ₁ , and the output voltage of the sensor system is V ₂ , the frequency response H (jω) of this sensor system can be obtained from the figure. By solving the circuit equation, the following equation is obtained.

ここで、ωは励振の角周波数、ｊは虚数単位をあらわす。
上式より、このセンサシステムの振幅応答｜Ｈ（ｊω）｜は、

Here, ω represents the angular frequency of excitation, and j represents the imaginary unit.
From the above equation, the amplitude response | H (jω) | of this sensor system is

となる。

Will be.

As shown in Equation 3, the amplitude of the output of the sensor system 101 is proportional to the capacitance of the capacitor C _in.

The output of the sensor system is consistent with the frequency of its frequency excitation, the amplitude varies in accordance with the capacitance of the capacitor C _in, the sensor system can be said that the amplitude modulation system.

When FIG. 2 is abstracted, it is represented as shown in FIG. Here, the input S (t) of the sensor system can be not only an electric signal such as a voltage or a current, but also a capacitance as shown in this embodiment.

(motion)
The operation of the capacitance sensor of the present invention will be described with reference to FIG.

The excitation generation unit 102 generates an intermittent sinusoidal voltage as shown in the waveform at the top of FIG. 3, that is, the output voltage of the excitation generation unit. This is supplied to the sensor system 101 as excitation. In this example, the frequency of the sine wave is 100 kHz. The sensor system in response to the electrostatic capacitance of the excitation and capacitor C _in, 2-th waveform of FIG. 3, that is, as shown in the sensor system output voltage, and outputs a voltage f (t). As shown in the figure, the response of the sensor system during the period when the excitation generator 102 outputs a sine wave is x ₁ (t) and x ₂ (t), and the excitation generator 102 is the sensor system during the period when the wave is stopped. Let the output voltages be z ₁ (t) and z ₂ (t).

According to Equation 3, the amplitude of the output voltage of the sensor system during the period when the excitation generator is stopped is zero. However, in reality, noise is mixed in and it does not become zero. Often, such as a touch sensor or a touch panel, the capacitance of the capacitor C _in shown in FIG. 2 is the capacitance of the capacitor formed by the indicator body (finger) surface resistor and, a capacitor C _in External noise (change in electric field, noise with capacitance coupling) is easily mixed in the surface resistors that form a part. _{The reason why z 1} (t) and z ₂ (t) are not zero in FIG. 3 is that they show the influence of this noise. When the external noise is stationary, the external noise is mixed regardless of whether the excitation is a sine wave or a stopped wave (DC), so noise is also generated at x ₁ (t) and x ₂ (t). Is mixed. That is, x ₁ (t) and x ₂ (t) contain a signal in which noise is added to the true signal (true signal + noise), and z ₁ (t) and z ₂ (t) contain only noise. It is appearing.

Importantly the inventors have found _{is, z 1 (t), z} 2 (t) does not depend on the capacitance of the capacitor C _in, it is to represent external noise. That is, in the case of a touch sensor or a touch panel, only noise appears regardless of the presence or absence of a finger as an indicator. This is because the impedance of the capacitor C _in which is formed between the finger and the position detecting conductive film, sufficiently high relative to the impedance of the sensor system, noise mixed in the position detecting conductive film, with or without a finger, This is because it flows into the sensor system as an electric current.

Then, the noise mixed in the sensor system output voltage during the period when the excitation generator is outputting a sine wave and the noise mixed in the sensor system output voltage during the period when the excitation generator before and after that is stopped are mixed. A correlation was found with noise.

The demodulation unit 105 receives the output signal of the sensor system 101 and takes advantage of the above characteristics to remove noise. And the observed signal _x 1 (t) containing the true signal + noise, the observed signal contains only noise _z 1 _{(t), x} true signal 1 (t), where the true signal _x 1 (t) is An example of finding the amplitude of

The demodulation unit 105 periodically reads a signal value from the sensor system output voltage f (t) at each time interval Δt and converts it into discrete-time signals f (iΔt) and i ∈ Z (Z: set of integers). x ₁ (t) is sampled in this way to x ₁ (iΔt), i = 0, 1, 2, ... N-1, and z ₁ (t) is sampled to z ₁ (iΔt), i. = 0, 1, 2, ... Q-1 is obtained.

Of the discrete Fourier transform Dk of x ₁ (iΔt), if the Dk corresponding to 100 kHz, which is the frequency of the excitation sine wave, is X _1,

ｊは虚数単位、Ｎはサンプル数、と、複素数Ｘ_１を求めることができる。複素数Ｘ_１は、ベクトルＸ_１≡（Ｒｅ｛Ｘ_１｝，Ｉｍ｛Ｘ_１｝）、Ｒｅ｛Ｘ_１｝は複素数Ｘ_１の実部，Ｉｍ｛Ｘ_１｝は複素数Ｘ_１の虚部と記述でき、２次元ベクトルＸ_１で表現することもできる。そしてこれらは同値である。

It is possible to obtain _{the complex number X 1 such} that j is an imaginary unit and N is the number of samples. The complex number X ₁ is described as the vector X ₁ ≡ (Re {X ₁ }, Im {X ₁ }), Re {X ₁ } is described as the real part of the _{complex number X 1 , and Im {X 1} } is described as the imaginary part of the complex number X _1. It can, can be expressed by 2-dimensional vectors X _1. And these are equivalent.

Similarly, _{of the discrete Fourier transform Dk of z 1} (iΔt), if the Dk corresponding to 100 kHz, which is the frequency of the sine wave, is Z _1.

ｊは虚数単位、Ｑはサンプル数、と、複素数Ｚ_１を求めることができる。複素数Ｚ_１は、ベクトルＺ_１≡（Ｒｅ｛Ｚ_１｝，Ｉｍ｛Ｚ_１｝）と、２次元ベクトルＺ_１で表現することもできる。そしてこれらは同値である。

It is possible to obtain _{the complex number Z 1 such} that j is an imaginary unit and Q is the number of samples. The complex number Z ₁ can also be expressed by the vector Z ₁ ≡ (Re {Z ₁ }, Im {Z ₁ }) and the two-dimensional vector Z _1. And these are equivalent.

_{Next, the vector X 1} − vector Z ₁ is calculated on the assumption that the 100 kHz component of the noise contained in the observation signal x ₁ (t) is the same as the 100 kHz component of the observation signal z ₁ (t). .. Then, the magnitude of | X ₁ −Z ₁ | is set _{as the amplitude of the true signal of x 1} (t), and the demodulated signal D (t) is set as the output of the demodulation unit.

Using the observation signal model, the operation of the demodulation unit described above will be described by applying specific numerical values.

The model of the observation signal is shown in FIG. When the model of the observed signal is expressed as f (t), f (t) is the next signal obtained by adding _{a true signal (V sig} ) having a 2V amplitude and a noise (V _{noise) having a 1V amplitude.}

Δｔ＝０．１マイクロ秒としてサンプリングし、ｆ（ｔ）をｆ（ａΔｔ）、ａ＝０，１，２，・・・４９９９と離散化した。

Sampling was performed with Δt = 0.1 microsecond, and f (t) was discretized as f (aΔt), a = 0,1,2, ... 4999.

x ₁ (iΔt) and z ₁ (iΔt) are the signals shown in FIG. _The length of _x 1 _(iΔt) (Time), i.e. t _{1 '-t 1,} considering that the extract components of 100kHz after an integer multiple of the period of 100kHz, i.e. n x 10 microseconds, n Is preferably a positive integer.

_{Specifically, x 1 (iΔt), i} = 0~1999 the f (aΔt), a = 1000~2999 , and then, and the t _{1 '-t 1} and 200 microseconds (n = 20).

Further, it is desirable that the start time t2 of z _{1 (iΔt) is t 1} + m x 10 μsec, and m is a positive integer, and further, _{the length (time) of z 1} (t), that is, t _{2 ′} −. It is desirable that t ₂ is an integral multiple of the period of 100 kHz, that is, w x 10 microseconds, and w is a positive integer.

_{Specifically, z 1 (iΔt), i} = 0~1999 the f (aΔt), and _{a = 3000~4999, t 2 = t} 1 +200 microsecond (m = 20), the t _{2 '-t 2} It was set to 200 microseconds (w = 20).

When X ₁ and Z ₁ were calculated, the following results were obtained.

上の複素数をベクトルと捉え、ベクトルＸ_１、ベクトルＺ_１、および、ベクトルＸ_１−ベクトルＺ_１を複素平面上にプロットすると、図５のようになる。

Taking the above complex number as a vector _{, and plotting the vector X 1} , the vector Z ₁ , and the vector X ₁ − vector Z ₁ on the complex plane, the result is as shown in FIG.

Note that the magnitude of the vector X ₁ -vector Z ₁ is 1.0 as shown, and here, the magnitude of each vector in FIG. 5 is 1/2 of the amplitude of the signal at 100 kHz. From the X ₁ -vector Z ₁ , it was calculated that the true signal amplitude was 2V. On the other hand, the amplitude of the calculated _{x 1 (iΔt) 2x | X} 1 | Z 1 | | (1.0V) such as the amplitude of _2x the amplitude of (1.5V) and the calculated _z 1 _(iΔt) It is difficult to derive the true signal amplitude (2V) based solely on information.

The amplitude of the amplitude of the _x 1 _(iΔt) (1.5V) and _z 1 _(iΔt) (1.0V) _was determined amplitude of 100kHz components _{x 1 (iΔt), z 1} (iΔt) each signal Is equivalent to that. In other words, it is not possible to obtain the true signal amplitude only by removing noise using conventional frequency separation.

In the above, an example in which one value of the demodulated signal D (t) is obtained _{by calculating X 1} and Z _{1 from x 1} (iΔt) and z ₁ (iΔt) and calculating | X ₁ −Z _{1 |.} showed that. For the next value of D (t), as shown in FIG. 3, _{X 2} and Z _{2 are calculated from x 2} (t) and z ₂ (t), and | X _2- Z ₂ | is calculated. The demodulated signal D (t) is obtained by calculating the subsequent values of D (t) in the same manner.

There are two actions and effects, and the first effect is to stop the sine wave and acquire noise, so regardless of the presence or absence of the finger, or when the presence or absence of the finger changes, or when the pressing pressure of the finger changes. even if the capacitance of the capacitor C _in is changed by changing, is the ability to obtain accurate noise.

The second effect is to subtract the vectors of "true signal + noise" and "noise", so even if the true signal and noise have the same frequency, the true signal can be obtained accurately. Is what you can do.

(Embodiment 2)
In the first embodiment, the observation _{signal z 1} (iΔt) is used to obtain the true signal amplitude of _{the observation signal x 1 (iΔt).} That is, the noise was removed by using _{the noise z 1} (iΔt) observed after the observation signal x ₁ (iΔt) in time. In Embodiment 2, Embodiment obtaining the amplitude of the true signal by using the front and rear of the noise observed signals x _{1 (iΔt)} observed signal x _{1 (iΔt),} will be mainly described operation of the demodulation unit.

FIG. 6 shows a model f (aΔt) of an observation signal in which the input signal of the demodulation unit 105 is discretized, a = 0, 1, 2, ..., Δt = 0.4 microseconds.

f (aΔt) is the sum of a true signal (V _{sig ) having an amplitude of 1 V and a noise (V noise} ) of 99 kHz whose amplitude changes in proportion to the passage of time. This is expressed by a mathematical formula as follows.

ｙ（ｉΔｔ）、ｘ（ｉΔｔ）、ｚ（ｉΔｔ）はそれぞれ、ｆ（ａΔｔ）から次のように切り出した信号とした。

y (iΔt), x (iΔt), and z (iΔt) are signals cut out from f (aΔt) as follows.

y (iΔt), i = 0 to 399 are f (aΔt), a = 3800 to 4199, and x (iΔt), i = 0 to 1624 are f (aΔt), a = 4250 to 5874, and z (iΔt). , I = 0 to 299 were set to f (aΔt), and a = 6000 to 6299.

Here, for convenience, y (iΔt) is named forward noise, and z (iΔt) is named backward noise.

In the demodulation unit, using the same method as in the first embodiment, the complex numbers Y _m and Z _m are obtained from the observed signals y (iΔt) and z (iΔt) by the following equations.

Δｔはサンプリング周期，ｊは虚数単位。

Δt is the sampling period and j is the imaginary unit.

The vectors Y _m and Z _m obtained here are schematically shown in FIG.

Then the vector _{Y m,} from the vector _{Z m,} the noise vector Y at time _{t 1} and time t _{1 ',} guess Z. The guessing method is as follows. _{Let Y m be} the noise vector when the time is (t ₀ + t _0' ) / 2, and _{Z m be} the noise vector when the time is (t ₂ + t _{2') / 2.}

From Y _m to Z _m, approximate amplitude and phase changes of the vector in proportion to time, obtain a noise vector Y, Z at time t ₁ and time t _{1 '.} FIG. 7 schematically shows the relationship between _{Y m,} Z _{m and Y, Z.}

Next, these average vectors M are calculated from the vectors Y and Z. The calculation of the average vector M will be described with reference to FIG.

The vector representation and the complex number representation have the same value as described above, and the calculation formula of M in the complex number is as follows.

ここで、Ｔは図６のｔ_１’−ｔ_１、Ａ_Ｓ，θ_ＳはベクトルＹの振幅と位相、Ａ_Ｅ，θ_ＥはベクトルＺの振幅と位相をあらわす。図９（ａ）に、図６のモデル信号から、上記にしたがって求めたＹ、Ｚ及びＭを示す。

Here, T is t _{1 '-t 1} of FIG. 6, _{A S,} θ _S is the vector Y of the amplitude and phase, _{A E,} θ _E represents the amplitude and phase of the vector Z. FIG. 9A shows Y, Z, and M obtained from the model signal of FIG. 6 according to the above.

Next, X is obtained and X-M is calculated in the same manner as in the first embodiment. X is expressed by the following equation.

ここで、Δｔはサンプリング周期，ｊは虚数単位で計算される。

Here, Δt is calculated in the sampling period, and j is calculated in the imaginary unit.

X obtained from x (iΔt) in FIG. 6, M obtained earlier, and XM are shown in FIG. 9 (b).

From FIG. 9B, | X-M | is 0.5, and it should be noted that this value represents 1/2 of the amplitude of the true signal, and the amplitude of the true signal of 1.0 V is correctly obtained. It was confirmed that That is, it was shown that the noise is accurately removed even when the noise of 99 kHz, which is extremely close to the excitation frequency of 100 kHz, is mixed.

Further, in general, when the signal acquisition period is limited as in x (iΔt) this time, the frequency resolution is lowered and noise near the frequency of the true signal cannot be removed. On the other hand, as shown in the present embodiment, by using the noise y (iΔt) in front of x (iΔt) and the noise z (iΔt) in the rear, the noise in the vicinity frequency is removed beyond the frequency resolution. I was able to.

Further, as shown in the present embodiment, even when the amplitude of the noise depends on the time, the noise can be removed with high accuracy by using the average vector M.

(Action / effect)
The following two can be mentioned as the action and effect.

First, by using the average vector calculated from the front noise and the rear noise, it is possible to remove noise in the vicinity frequency beyond the frequency resolution.

Secondly, by using the average vector calculated from the front noise and the rear noise, it is possible to accurately remove the noise even if the amplitude of the noise fluctuates.

The capacitive touch panel of the present invention will be described.

(Constitution)
FIG. 10 shows the configuration of the capacitive touch panel 130 of the present invention. The touch panel shown in FIG. 10, by using the electrostatic capacitance of the capacitor C _in which is formed between the finger and the surface resistor 131, detects the presence or absence of a touch and a touch position.

For the surface resistor 131, an ITO (Indium-tin-oxide) film was used. The ITO film is a solid film having a uniform sheet resistance value, here 800 ohms, on a glass substrate (not shown). An insulator, here, a polarizing plate 132 used for forming a liquid crystal display device, was attached onto the ITO film using an acid-free glue.

Wiring is connected to the four corners of the ITO film 131. Each wire is connected to four sensor systems 101 as shown in FIG. The configuration of the sensor system is the same as that of the first embodiment. The output voltage of the excitation generation unit 102 is input to the four sensor systems, and the output of each sensor system is given to the demodulation unit 105 (demodulation unit 0 to demodulation unit 3).

The output of the demodulation unit is transmitted to a block (not shown) including a signal processing circuit, and the block including the signal processing circuit calculates the presence / absence of touch and the touch position based on the output value of the demodulation unit.

(motion)
The operation of the capacitive touch panel of the present invention will be described with reference to FIG.

The capacitive touch panel of the present invention is arranged on the display surface of a liquid crystal display (LCD) and is driven so as to avoid driving noise of the LCD.

The non-address explicit signal of FIG. 11 is a signal that clearly indicates the non-address period of the LCD, and is a signal that is set to a high level during the non-address period. Here, the non-address period refers to a period during which the scanning lines of the LCD are not scanned, and refers to a period after the selection of the last scanning line is completed and before the first scanning line is selected.

One of the driving features of the present invention is to have a period (t _{1 to t 1'} ) in which a sine wave is applied to the excitation to detect a touch during the non-address period, and the sine wave is stopped to make noise. it is to have a period to obtain the _{(t 0} ~t 0 _{'and t 2} ~t _2').

By acquiring noise during the non-address period, this noise includes external noise, but does not include LCD drive noise. As a result, it is possible to accurately estimate and remove the noise mixed in during the touch detection period (t _{1 to t 1').}

The excitation generation unit 102 generates an intermittent sinusoidal voltage as shown in the second waveform from the top of FIG. This is the excitation of the sensor system. In order to obtain the output voltage of the excitation generator of FIG. 11, _{a sine wave having a frequency of 100 kHz and an amplitude of 1.5 V pp} (1.5 volt peak to peak) is given to the excitation generator by the sine wave generator 103. And a DC voltage of DC = 1.2V is given by the DC generator 104. Then, the excitation generator outputs an intermittent sinusoidal voltage having an offset of 1.2 V, a frequency of 100 kHz, and an amplitude of 1.5 V _pp. During the period when the sine wave is stopped, a voltage of DC = 1.2V is output.

The voltage generated by the excitation generator is applied to four sensor systems 101-here, for convenience, the sensor system of ch0, the sensor system of ch1, the sensor system of ch2, and the sensor system of ch3 are distinguished. The voltage generated by the excitation generation unit 102 is applied to the non-inverting input terminal of the operational amplifier 110 in the sensor system, and this voltage appears at the inverting input terminal by the imaginary short operation of the operational amplifier. That is, when the excitation generation unit 102 outputs a voltage having a frequency of 100 kHz and an amplitude of 1.5 V _pp, a voltage having a frequency of 100 kHz and an amplitude of 1.5 V _pp is applied to the ITO 131.

When the capacitance of the capacitor C _in is formed, an alternating current flows from each sensor system to the human body via the _{conductances G0 to G3 and the capacitor C in,} which are determined according to the position of the finger.

The output of each sensor system is a superposition of noise on an intermittent sinusoidal voltage whose amplitude is determined by the magnitude of this alternating current. The sensors system to select a representative sensor systems ch1, as shown in FIG. 11 the output voltage as f _{1 (t).}

The operation of the demodulation unit 105 will be described using ch1 as an example.

_{In the output voltage f 1} (t) of the sensor system of ch 1, the demodulation unit 105b of ch _{1 has y n} (t), x _n (t), z _n (t), n as integers as shown in FIG. The _{amplitude D 1} (t) of the true signal of _{x n} (t) is output by using the signal of.

The demodulation unit 105b _{samples the output voltage f 1} (t) of the sensor system at a sampling interval Δt = 0.4 microseconds _{, and obtains f 1} (aΔt), where a is an integer with a sample number.

x ₁ (iΔt), y ₁ (iΔt), and z ₁ (iΔt) are _{signals cut out from f 1} (aΔt) as follows. y ₁ (iΔt), i = 0 to 399 are f (aΔt), a = 3801 to 4200, and x ₁ (iΔt), i = 0 to 1624 are f (aΔt), a = 4251 to 5875, and z ₁ (IΔt) and i = 0 to 399 were set to f (aΔt) and a = 6001 to 6400.

In this embodiment, in order to accurately estimate the rotation of the noise phase, _{the period corresponding to y 1} (t) and z ₁ (t) is divided into four segments, and the vector of the 100 kHz component is calculated for each segment. ..

Specifically, it is shown in the following formulas 18 to 25.

つぎに、前方ノイズ、後方ノイズの振幅と位相を求める。

Next, the amplitude and phase of the front noise and the rear noise are obtained.

For the amplitude, first calculate the average value of the segments as follows. The forward noise amplitude | Y _m | and the rear noise amplitude | Z _m | are, respectively.

位相は、まず、数式１８から数式２５で得られた結果から、次の通り、各セグメントの位相を計算する。

For the phase, first, the phase of each segment is calculated from the results obtained by the formulas 18 to 25 as follows.

angle [Y _1,1 ], angle [Y _{1, 2,} ], angle [Y ₁ , 3], angle [Y ₁ , 4], and angle [Z ₁ , 1], angle [Z _{1, 2,} ], angle [Z _1,3 ], angle [Z _1,4 ], where angle [Y _1,1 ] indicates the phase of _{Y 1,1.}

The phase calculated above is limited to the range ± π. If this is left as it is, it is inconvenient to estimate the phase, so 2nπ and n are integers as appropriate, and the phases are smoothly connected.

This operation is easy to understand by actually observing the phase transition of the 100 kHz component of the sensor system output mixed with the external noise of the inverter circuit of the fluorescent lamp.

FIG. 12 shows a waveform when the capacitive touch panel of the present invention is driven near an inverter circuit of a fluorescent lamp. The top is the ITO voltage, the second is the waveform obtained by sampling the sensor system output of ch1, and the third is the 100 kHz amplitude calculated from each segment when 100 samples are taken as one segment, and the bottom graph. Indicates the phase of 100 kHz calculated from each segment when 100 samples are taken as one segment. The graph at the bottom is the result of adding 2nπ and n to an integer to the phase limited to the range of ± π, and connecting the phases smoothly.

From this result, it can be seen that the phase change is smooth, and it is possible to smoothly connect the phases by adding 2nπ and n as an integer as appropriate.

Further, it was obtained from _{the inclinations of the four phases angle [Y 1} , 1], angle [Y _{1, 2,} ], angle [Y ₁ , 3], and angle [Y _{1, 4} ] obtained from the front noise, and the rear noise. By using angle [Z ₁ , 1], angle [Z _{1, 2,} ], angle [Z ₁ , 3], and angle [Z _{1, 4} ], which phase is in the process from front noise to rear noise. Guess how much it has rotated in the direction.

Regarding the phase, the angle after performing the above two processes, that is, the process of removing the limitation in the range of ± π and the process of estimating the rotation direction and the amount from the inclination of the phases of the front noise and the rear noise, is angled. [Y _1,1 ]', angle [Y _{1, 2} ]', angle [Y _{1, 3} ]', angle [Y _{1, 4} ]' and angle [Z _{1, 1} ]', angle [Z _{1, 2,} ]', Angle [Z _1,3 ]', angle [Z _1,4 ]', and the front noise phase angle [Y _m ] and the rear noise phase angle [Z _m ] are calculated as follows. ..

なお、図１２の上から３番目のグラフにより、前方ノイズと後方ノイズとを直線でつなぐ近似で、ｘ（ｔ）の期間に混入しているノイズの振幅も推測可能であることも分かる。

From the third graph from the top of FIG. 12, it can be seen that the amplitude of the noise mixed in the period of x (t) can be estimated by the approximation of connecting the front noise and the rear noise with a straight line.

The vector Y _m is determined by | Y _m | and the angle [Y _m ] obtained above, and the vector Z _m is determined by | Z _m | and the angle [Z _m].

Then, according to the procedure described in embodiment _2, Y m, from _{Z m,} the noise vector Y at time _{t 1} and time t _{1 ',} guess Z.

Subsequently, according to the procedure described in embodiment 2, to calculate an average vector M ₁ of the vectors Y, Z.

Further, the vector X ₁ is obtained, and X ₁ −M ₁ is calculated. X is represented by the following equation.

ここで、Δｔはサンプリング周期，ｊは虚数単位で計算される。
ベクトルＸ_１−Ｍ_１の大きさ｜Ｘ_１−Ｍ_１｜は復調部１０５ｂの出力Ｄ_１（ｔ）として、図１１に示すように出力される。

Here, Δt is calculated in the sampling period, and j is calculated in the imaginary unit.
The magnitude | X _1- M ₁ | of the vector X ₁ −M ₁ is output as the output D ₁ (t) of the demodulation unit 105b as shown in FIG.

Even in the next non-address period, as shown in FIG. 11, _{| X 2-} M _{2 | is calculated from y 2} (t), x ₂ (t), and z ₂ (t) and used as the output of the demodulation unit. ..

Hereinafter, _{| X n −} M _{n | is calculated from y n} (t), x _n (t), and z _n (t) in the same manner, and is used as the output of the demodulation unit.

Next, from the output voltage f ₁ (t) of _{the sensor system described above, Y 1} , ₁ , Y ₁ , 2, Y 1, 3, ..., X ₁ , ..., Z ₁ , _3, A block diagram of the signal processing unit for obtaining, Z _{1, 4 will be described with reference to FIG.}

The output f (t) of the sensor system 101 of FIG. 13 _{corresponds to the output voltage f 1} (t) of the sensor system described with reference to FIG. f (t) is supplied to the sampler 140 and is converted into discrete-time signals f (aΔt), a = 0, 1, 2, ..., Every time interval Δt = 0.4 microseconds. f (aΔt) is input to two multipliers (multiplier I 141a and multiplier Q 141b). The multiplier I 141a sequentially multiplies f (aΔt) with cos (ωaΔt), a = 0,1,2,3 ..., Ω = 2π100 kHz, and sequentially outputs the result at each time interval Δt. .. Similarly, the multiplier Q 141b sequentially multiplies f (aΔt) by sin (ωaΔt), a = 0,1,2,3 ..., Ω = 2π100 kHz, and sequentially multiplies the result at each time interval Δt. Output.

The cos (ωaΔt) of the multiplier I uses the output of the sine wave generator 103, and the sin (ωaΔt) of the multiplier Q passes the output of the sine wave generator through the phase shifter 145 of -90 degrees. Use the converted signal.

The outputs of the multiplier I 141a and the multiplier Q 141b are input to the multiplier I 142a and the multiplier Q 142b, respectively, and the multiplier adds the signals input during the active period of the control signal given by the controller 146. To do.

For example, in order to obtain Y1, 1, the controller gives an active signal to the integrator during the period when the value of a of f (aΔt) is 3801 to 3900. As a result, the multiplier I 142a becomes

を計算する。つまり、数式１７で示したＹ_１，１の実部の１００倍の値が計算される。

To calculate. That is, a value 100 times the real part of _{Y 1} , 1 shown in Equation 17 is calculated.

The signals integrated for a predetermined period are taken into the register I 143a and the register Q 143b, respectively, and are multiplied by 1 / N (N is the integrated number of samples) by the multiplier 144 connected to the register.

Through this process, the multiplier I 144a becomes the fruit of _{Y 1} , ₁ , Y ₁ , 2, Y 1, 3, ..., X ₁ , ..., Z ₁ , 3, Z _{1, 4.} Part, that is, Re {Y ₁ , 1}, Re {Y _{1, 2,} }, Re {Y ₁ , 3}, ..., Re {X ₁ }, ..., Re {Z ₁ , 3}, The values of Re {Z _1,4 } are output in sequence, and the multiplier Q 144b is Y ₁ , ₁ , Y ₁ , 2, Y 1, 3, ..., X ₁ , ..., Z ₁ , 3, -1 times the imaginary part of Z ₁ , 4, that is, -Im {Y ₁ , 1}, _{-Im {Y 1, 2,} }, _{-Im {Y 1, 3} }, ..., -Im {X _The values of 1}, ..., -Im {Z _1,3 }, _{and -Im {Z 1,4} } are output in sequence.

These values are sequentially input to a computer (not shown), and the magnitude and phase are calculated.

Next, the experimental results of the case where the present invention is used and the conventional case, that is, noise removal using only frequency separation will be described.

As for the configuration of the experiment, the touch panel of FIG. 10 was prepared, and the inverter circuit of the inverter type fluorescent lamp was arranged 30 cm above the touch panel. Observing the output of the sensor system, it is clear that noise from the inverter circuit is mixed in.

The measurement was performed for about 10 seconds, and about 5 seconds after the start of the measurement, the center of the touch panel was touched with a finger. The experimental results are shown in FIG.

FIG. 14B is an experimental result when the present invention is used. One of | Xn-Mn |, which is the output of D1 (t), is plotted as one point, and the plots of 653 points are connected by a straight line. It is.

On the other hand, FIG. 14A shows the experimental result of noise removal using only frequency separation. Specifically, the amplitude of the 100 kHz component of the output signal of the sensor system during the period of the sine wave with the excitation of 100 kHz is | Xn. It was obtained by −0 |.

By implementing the present invention, if the magnitude of the signal difference with and without touch is the signal S and the standard deviation without touch is the noise N, the conventional S / N = 1.36, whereas in the present invention, 3. An improvement in S / N of 87 and 9 dB was confirmed.

It can be applied to capacitance sensors, touch panels, touch sensors, and other electronic devices that use AM modulation / demodulation systems.

100: Capacitance sensor 101: Sensor system 102: Excitation generation unit 103: Sine wave generation unit 104: DC generation unit 105, 105a, 105b, 105c, 105d: Demodulation unit 110: Operational amplifier 111: Adder 120: Electronic device 130: Capacitive touch panel 131: Surface resistor (ITO)
132: Polarizing plate 140: Sampler 141: Multiplier, 141a: Multiplier I, 141b Multiplier Q
142: Integrator, 142a: Integrator I, 142b: Integrator Q
143: Register, 143a: Register I, 143b: Register Q
144: Multiplier, 144a: Multiplier I, 144b: Multiplier Q
145: Phase shifter 146: Controller

Claims (9)

With the sensor system
An excitation generator that generates an intermittent AC signal and gives it to the sensor system,
A demodulator that demodulates the amplitude-modulated signal that is the output of the sensor system,
Including
The demodulator
The response of the sensor system during the period when the excitation generator outputs alternating current,
An electronic device characterized in that a demodulated signal is generated by using at least one of the responses immediately before and after the period and the response of the sensor system during the period when the excitation generating unit does not output alternating current. .. Calculated from the response of the sensor system during the period when the excitation generator outputs alternating current.
Let X be a vector obtained from the amplitude and phase of the frequency component of the alternating current.
When N is a vector obtained from the amplitude and phase of the frequency component of the AC, which is calculated from the response of the sensor system during the period when the excitation generator is not outputting AC.
The electronic device according to claim 1, wherein the demodulated signal is a constant multiple of | XN |. Calculated from the response of the sensor system during the period when the excitation generator outputs alternating current.
Let X be a vector obtained from the amplitude and phase of the frequency component of the alternating current.
When the vectors obtained from the amplitude and phase of the frequency component of the AC calculated from the response of the sensor system during the period immediately before and immediately after the excitation generator is not outputting the AC are Y and Z, respectively.
The electronic device according to claim 1, wherein the demodulated signal is a constant multiple of | X-M |, where M is an average vector of Y and Z. Has a surface resistor and
The sensor system is composed of a drive / detection circuit connected to the surface resistor, which applies a voltage to the surface resistor and measures and outputs a current flowing through the surface resistor.
The electronic device according to any one of claims 1 to 3, which detects the touch state or coordinates of the indicator by detecting the capacitance of the capacitor formed by the surface resistor and the indicator. Capacitance sensor. It is configured to include a display device
The non-address period of the display device has a period in which the excitation generator outputs alternating current and a period in which the alternating current is not output.
The static according to claim 4, wherein a demodulated signal is generated by using both the response of the sensor system during the period when the alternating current is output and the response of the sensor system during the period when the alternating current is not output. Capacitive sensor. Has a surface resistor and
The sensor system is composed of a drive / detection circuit connected to the surface resistor, which applies a voltage to the surface resistor and measures and outputs a current flowing through the surface resistor.
The touch panel including the electronic device according to claims 1 to 3, which detects the touch state or the coordinates of the indicator by detecting the capacitance of the capacitor formed by the surface resistor and the indicator. It is configured to include a display device
The non-address period of the display device has a period in which the excitation generating unit outputs an alternating current and a period in which the alternating current is not output.
The first to third claims, wherein a demodulated signal is generated by using both the response of the sensor system during the period when the alternating current is output and the response of the sensor system during the period when the alternating current is not output. Electronics. It is configured to include a display device
The non-address period of the display device has a period in which the excitation generator outputs alternating current and a period in which the alternating current is not output.
The touch panel according to claim 6, wherein a demodulated signal is generated by using both the response of the sensor system during the period when the alternating current is output and the response of the sensor system during the period when the alternating current is not output. An operational amplifier, a resistor connected between the output terminal and the inverting input terminal of the operational amplifier, and
A conductor connected to the inverting input terminal of the operational amplifier and
An electronic device that detects the capacitance of a conductor, including an excitation generator that generates an intermittent AC signal and supplies it to the non-inverting input terminal of the operational amplifier.
A demodulator that demodulates the amplitude modulated signal that is the output of the operational amplifier.
Including
The demodulator
The response of the operational amplifier during the period when the excitation generator outputs alternating current,
An electronic device characterized in that a demodulated signal is generated by using both the response of the operational amplifier during the period when the excitation generating unit does not output alternating current, at least immediately before and immediately after that.

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