JP5962593B2 - Current detector - Google Patents

Description

  The present invention relates to a current detection device that outputs a signal value corresponding to a change in line current.

  In recent years, in the power line communication field such as in-vehicle, communication means has been adopted in which a signal current that changes according to the signal value is sent to the power line that supplies power to the load, and the signal value is regenerated by capturing the current change of the power line on the receiving side. Has been. For example, a configuration is conceivable in which power is supplied from an electronic control device to a sensor through a power supply line, and an A / D conversion value of a sensor signal is transmitted from the sensor to the electronic control device through the same power supply line. In addition to the signal current, a load current always flows through the power line. For this reason, a circuit for detecting a change in the signal current flowing through the power line and converting it into digital data is required.

  The current detection device described in Patent Document 1 includes a detection resistor provided on a line through which a load current flows, a current-voltage conversion circuit that converts the current flowing through the detection resistor into a voltage, a conversion voltage, and a reference voltage. And a voltage comparison circuit (A / D converter) for comparing and converting into digital data.

US Pat. No. 8,237,449B2

  In the above configuration, since a loss occurs when a load current flows through the detection resistor, it is necessary to set the resistance value to a very small value. As a result, there is a concern that a minute error existing in each circuit greatly affects the detection accuracy of the signal value and causes a communication error. In general, in a semiconductor circuit, resistance and transistor pair characteristics vary during manufacturing. Therefore, an offset voltage is generated in the amplifier constituting the current-voltage conversion circuit described above, and an error occurs in the conversion voltage. In addition, since an offset voltage exists in the voltage comparison circuit, there is a possibility that the detection error further increases.

  The present invention has been made in view of the above circumstances, and an object of the present invention is to provide current detection capable of detecting a signal value with high accuracy by eliminating the influence of errors generated in a current-voltage conversion circuit and errors generated in a voltage comparison circuit. To provide an apparatus.

  The current detection device according to claim 1 outputs a signal value corresponding to a change in the current flowing through the line, based on a magnitude relationship between the amount of change in the current flowing through the line and the current threshold value. The current detection device includes a current-voltage conversion circuit, a threshold addition circuit, a voltage comparison circuit, and a control circuit. The current-voltage conversion circuit includes a detection resistor, a reference resistor, and an amplifier circuit provided on the line. The amplifier circuit causes a current corresponding to the line current to flow through the reference resistor by negative feedback amplification using the voltage across the detection resistor as an input.

  The threshold value adding circuit includes a constant current circuit that outputs a threshold setting current corresponding to the current threshold value, and an addition switch provided in a path from the constant current circuit to the reference resistor. The voltage comparison circuit includes a capacitor, a comparator that compares the voltage of the reference resistor with the voltage of the capacitor, and a feedback switch that forms a negative feedback path from the output terminal of the comparator to the input terminal to which the capacitor is connected. .

  The control circuit includes a threshold setting period for setting a threshold voltage on the capacitor, a signal value detection period for generating a signal value by comparing the threshold voltage with a voltage generated at the reference resistor according to the line current Execute.

  The control circuit turns on the addition switch and the feedback switch during the threshold setting period. At this time, a current corresponding to the line current, a threshold setting current corresponding to the current threshold, and an error current generated in the current-voltage conversion circuit flow through the reference resistor. The capacitor is set with a threshold voltage corresponding to the current obtained by combining the line current and the threshold setting current. This threshold voltage includes an error (for example, a voltage corresponding to an offset current) generated in the current-voltage conversion circuit and an error (for example, an offset voltage) generated in the voltage comparison circuit.

  The control circuit turns off the addition switch and the feedback switch during the signal value detection period. At this time, the voltage of the reference resistor is compared with the threshold voltage, and a signal value is generated based on whether or not the line current has changed by more than the current threshold. Since the above-mentioned error is included in the threshold voltage, the error generated in the current-voltage conversion circuit and the error generated in the voltage comparison circuit are canceled in the comparison operation. Therefore, the signal value can be detected with high accuracy without being affected by these errors.

  According to a second aspect of the present invention, a threshold value adding circuit and a voltage comparing circuit are provided for each of a plurality of current threshold values having different sizes. The control circuit sequentially selects one current threshold value from the plurality of current threshold values during the threshold setting period, and adds an addition switch of the threshold value addition circuit corresponding to the selected current threshold value. Turn on the feedback switch of the voltage comparison circuit. As a result, the threshold voltage corresponding to the selected current threshold is set in the capacitor of the voltage comparison circuit corresponding to the selected current threshold.

  The control circuit turns off all the addition switches and feedback switches during the signal value detection period. As a result, the voltage generated in the reference resistor in accordance with the line current and the threshold voltages having different magnitudes are compared in parallel to generate a signal value. Since a plurality of current threshold values are used, a signal value having a multi-value level such as ternary, quaternary,.

  According to the third aspect of the present invention, one end of the reference resistor is set to the ground potential. The amplifier circuit includes an amplifier, a first resistor connected between the high potential side terminal of the detection resistor and the non-inverting input terminal of the amplifier, and between the low potential side terminal of the detection resistor and the inverting input terminal of the amplifier. A second resistor is connected, and a transistor is connected between the non-inverting input terminal of the amplifier and the other end of the reference resistor, and uses the output voltage of the amplifier as a control voltage. According to this configuration, a current obtained by multiplying the line current by (resistance value of the detection resistor) / (resistance value of the first resistor) flows through the reference resistor by negative feedback amplification.

  According to the fourth aspect of the present invention, one end of the reference resistor is set to the ground potential. The amplifier circuit includes an amplifier, first and second transistors having a common control terminal, a bias circuit that applies a bias voltage so that a bias current smaller than the line current flows through the first and second transistors, and a detection circuit Between the first and second resistors provided between the high potential side terminal of the resistor and the first transistor and between the low potential side terminal of the detection resistor and the second transistor, respectively, and between the first transistor and the ground potential The third and fourth resistors provided between the second transistor and the ground potential are connected between the non-line side terminal of the first resistor and the other end of the reference resistor to control the output voltage of the amplifier. A third transistor serving as a voltage. The resistance values of the first and second resistors are equal to each other, and the resistance values of the third and fourth resistors are also equal to each other. The non-ground side terminals of the third and fourth resistors are connected to the non-inverting input terminal and the inverting input terminal of the amplifier, respectively.

  According to this configuration, the voltages at the current-carrying terminals (for example, sources) on the first and second resistance sides of the first and second transistors are equalized by negative feedback amplification. Therefore, a current obtained by multiplying the line current by (resistance value of the detection resistor) / (resistance value of the first resistor) flows through the reference resistor.

  According to the means of claim 5, the first, second and third transistors have a withstand voltage corresponding to the voltage of the line, and the amplifier has a withstand voltage according to the voltages of the third and fourth resistors. . Thereby, the withstand voltage of the amplifier can be made lower than the voltage of the line. By reducing the withstand voltage of the amplifier, the characteristic variation of the elements constituting the amplifier is reduced, so that errors generated in the current-voltage conversion circuit can be reduced.

  According to a sixth aspect of the present invention, the first and second transistors are in cascode connection. In this case, for example, the breakdown voltage of the transistors arranged on the third and fourth resistance sides may be set high, and the breakdown voltage of the transistors arranged on the first and second resistance sides may be set low. As a result, variations in the characteristics of the first and second transistors are reduced, and errors generated in the current-voltage conversion circuit can be reduced.

Configuration diagram of a current detection unit showing the first embodiment Block diagram showing a schematic electrical configuration of the airbag device Current waveform diagram when signal current flows through power line Waveform diagram when receiving signal values FIG. 1 equivalent diagram showing the second embodiment 4 equivalent diagram FIG. 1 equivalent view showing the third embodiment FIG. 1 equivalent view showing the fourth embodiment

In each embodiment, substantially the same parts are denoted by the same reference numerals and description thereof is omitted.
(First embodiment)
A first embodiment will be described with reference to FIGS. As shown in FIG. 2, the vehicle airbag device 11 includes an airbag ECU 12 (hereinafter referred to as ECU 12), a sensor module 13, and an airbag 14. The airbag 14 attached to the steering and instrument panel is deployed when the vehicle collides front to protect the occupant, and the airbag 14 attached to the driver side door or the passenger side door Deploy to protect passengers in the event of a collision.

  The ECU 12 that controls the deployment of the airbag supplies power to the sensor module 13 through power lines 15 and 16 (corresponding to tracks) and communicates with the sensor module 13 through the power lines 15 and 16. That is, the sensor module 13 changes the current that flows through the power supply lines 15 and 16 according to the detected digital data of the sensor signal. The ECU 12 has a binary (0, 1) value corresponding to the change in the current flowing in the power supply lines 15 and 16 based on the magnitude relationship between the amount of change in the current flowing in the power supply lines 15 and 16 and a current threshold value described later. The signal value is sequentially received, and the digital data of the sensor signal is constructed.

  In the ECU 12, a DC power supply 17 having a voltage value of 20 V, for example, is connected between the power supply lines 15 and 16. A detection resistor Rs is provided on the power supply line 15 on the high potential side. The higher the resistance value of the detection resistor Rs, the larger the voltage amplitude of the detection resistor Rs. However, since the load current supplied to the sensor module 13 flows through the detection resistor Rs, the loss also increases. Therefore, the detection resistance Rs is set to a resistance value as small as possible while ensuring a margin for a communication error. The current detection unit 19 including the reception unit 18 and the detection resistor Rs will be described later. The ECU 12 includes a microcomputer, an ignition device, and the like (not shown), and performs collision detection and airbag 14 deployment control based on the received digital data of the sensor signal.

  In the sensor module 13, the power supply circuit 20 reduces the voltage between the power supply lines 15 and 16 to generate a control voltage. The sensor unit 21 includes a capacitive sensor element whose capacitance value changes according to acceleration, for example, and samples and holds a sensor signal obtained by CV conversion and outputs the sensor signal. The A / D converter 22 converts this sensor signal into digital data composed of a bit string of a predetermined length.

  A constant current circuit 24 with a control terminal is connected between the power supply lines 15 and 16. The current conversion unit 23 configured by a gate array or the like controls on / off of the output of the constant current circuit 24 corresponding to each bit value of the digital data. The constant current circuit 24 stops the current output when the bit value is 0, for example, and outputs a signal current having a constant value when the bit value is 1.

  The waveform of the line current Iin flowing through the power supply line 15 is, for example, as shown in FIG. A constant load current (for example, 8 mA) consumed by the sensor module 13 continues to flow through the power line 15, and a signal current (for example, 24 mA) corresponding to the bit value (signal value) of the digital data is superimposed on the power line 15. Flows. In the waveform example shown in FIG. 3, the communication frame is composed of 10 bits, and a start bit is added before the communication frame. When energization (power supply) from the ECU 12 to the sensor module 13 is stopped, the current Iin flowing through the power supply line 15 becomes zero.

  As shown in FIG. 1, the current detection unit 19 of the ECU 12 includes a current-voltage conversion circuit 25, a threshold addition circuit 26, a voltage comparison circuit 27, and a control circuit 28. The current-voltage conversion circuit 25 includes a detection resistor Rs, a reference resistor Ra, and an amplifier circuit 29 provided on the power supply line 15. The amplifying circuit 29 inputs a voltage across the detection resistor Rs and amplifies it by negative feedback, thereby causing a current Ia corresponding to the current Iin to flow through the reference resistor Ra. One end of the reference resistor Ra is connected to the power supply line 16 and has a ground potential.

  The amplifier circuit 29 includes an operational amplifier 30, an N-channel MOS transistor Ma, a first resistor R1, and a second resistor R2. The resistor R1 is connected between the high potential side terminal (DC power supply 17 side) of the detection resistor Rs and the non-inverting input terminal of the operational amplifier 30. The resistor R2 is connected between the low potential side terminal (sensor module 13 side) of the detection resistor Rs and the inverting input terminal of the operational amplifier 30. The drain and source of the transistor Ma are connected between the non-inverting input terminal of the operational amplifier 30 and the reference resistor Ra, and the transistor Ma operates using the output voltage of the operational amplifier 30 as a gate voltage (control voltage).

  The threshold value adding circuit 26 includes a constant current circuit 32 and an addition switch 33 connected in series between a control power supply line 31 for supplying a control voltage VDD and an output node Na of the current / voltage conversion circuit 25. Yes. The constant current circuit 32 outputs a threshold setting current Idetect / n corresponding to the current threshold Idetect as will be described later. Addition switch 33 is turned on when signal φ1 is at the H level.

  The voltage comparison circuit 27 includes a capacitor C1, a comparator 34, a feedback switch 35, and an output switch 36. One end of the capacitor C1 is connected to the power supply line 16 and has a ground potential. The comparator 34 compares the voltage Va of the reference resistor Ra and a voltage Vc (a threshold voltage Vth described later) of the capacitor C1. The feedback switch 35 is connected between the output terminal and the inverting input terminal of the comparator 34, and is turned on when the signal φ1 is at the H level to form a negative feedback path. At this time, the voltage comparison circuit 27 operates as a sample hold circuit.

  The output switch 36 is provided to prevent the output terminal 37 from outputting a voltage having an intermediate level other than the H / L binary level. The output switch 36 is turned on when the signal φ2 is at the H level. The control circuit 28 generates and outputs the signals φ1 and φ2 described above. The control circuit 28 may be constituted by a microcomputer mounted on the ECU 12.

Next, the operation of this embodiment will be described with reference to FIG. The current-voltage conversion circuit 25 of the ECU 12 causes the current Ia expressed by the expression (1) corresponding to the current Iin flowing through the detection resistor Rs to flow through the reference resistor Ra due to the negative feedback amplification action of the operational amplifier 30. The offset current Ioffset is an error current component generated by the offset voltage of the operational amplifier 30.
Ia = Iin × (Rs / R1) + Ioffset (1)

When defined as in the equation (2), the voltage Va of the reference resistor Ra is as in the equation (3).
Rs / R1 = 1 / n (2)
Va = Ra × (Iin / n) + Ra × Ioffset (3)

  The sensor module 13 receives a load current from the ECU 12 through the power supply lines 15 and 16, and transmits a signal value by changing a current flowing through the power supply lines 15 and 16 according to each bit value of the digital data of the sensor signal. . In order to receive this signal value, the control circuit 28 of the ECU 12 sets the threshold voltage Vth in the capacitor C1 during the threshold setting period in which the signal value is not transmitted from the sensor module 13, and the signal in which the signal value is transmitted. A change in the current Iin flowing through the detection resistor Rs is detected during the value detection period.

  The control circuit 28 sets the signal φ1 to the H level and the signal φ2 to the L level during the threshold setting period, turns on the addition switch 33 and the feedback switch 35, and turns off the output switch 36. Here, the amount of change in the current Iin when it is determined that the level (H / L) of the signal value has changed is defined as a current threshold Idetect. In order to maximize the margin for the noise current flowing in the power supply line 15, the current threshold Idetect is preferably set to ½ of the signal current output from the constant current circuit 24.

The constant current circuit 32 outputs a threshold setting current Idetect / n obtained by multiplying the current threshold Idetect by 1 / n. At this time, the voltage Va of the reference resistor Ra is as shown in Equation (4).
Va = Ra × (Iin + Idetect) / n + Ra × Ioffset (4)

In the threshold setting period, the voltage comparison circuit 27 operates as a sample hold circuit. The comparator 34 also has an offset voltage Voffset. At this time, the threshold voltage Vth charged in the capacitor C1 is expressed by the equation (5). The first term of the equation (5) is a voltage corresponding to the current obtained by adding the current threshold Idetect to the current Iin flowing through the power supply line 15. The second term and the third term are voltages due to an offset error that inevitably occurs in the current-voltage conversion circuit 25 and the voltage comparison circuit 27.
Vth = Ra × (Iin + Idetect) / n + Ra × Ioffset + Voffset (5)

  In the signal value detection period, the control circuit 28 sets the signal φ1 to the L level and the signal φ2 to the H level, turns off the addition switch 33 and the feedback switch 35, and turns on the output switch 36. The voltage comparison circuit 27 compares the voltage Va of the reference resistor Ra shown by the equation (3) with the threshold voltage Vth shown by the equation (5) and outputs a signal value.

  When the current Iin flowing through the detection resistor Rs is equal to the current value (current value corresponding to the signal value 0) flowing during the threshold setting period, the voltage Va is expressed by the following equation (3). Since the voltage Va is lower than the threshold voltage Vth shown by the equation (5), the signal value output from the comparator 34 is L level.

  When the signal value transitions from 0 to 1, the current Iin flowing through the detection resistor Rs increases by the signal current from the current value flowing during the threshold setting period. In the middle of this, when the current Iin is increased by the current threshold Idetect from the current value flowing during the threshold setting period, the voltage Va becomes a value represented by the equation (4). At this time, since both the voltage Va and the threshold voltage Vth have the term Ra × Ioffset, the error due to the offset current Ioffset generated in the current-voltage conversion circuit 25 is cancelled. Further, since the threshold voltage Vth has a term of Voffset, an error due to the offset voltage Voffset of the comparator 34 is also cancelled. As a result, the signal value output from the comparator 34 changes from the L level to the H level. Thereafter, the same applies when the current Iin decreases.

  As described above, the current detection unit 19 of the present embodiment has the threshold setting current Idetect corresponding to the current threshold Idetect to the reference resistor Ra through which the current corresponding to the current Iin flows during the threshold setting period. / N flow. The voltage Va of the reference resistor Ra at this time includes an offset current Ioffset that always flows through the reference resistor Ra. Further, the current detection unit 19 receives the voltage Va and operates the voltage comparison circuit 27 as a sample hold circuit. As a result, the threshold voltage Vth including the offset current Ioffset of the current-voltage conversion circuit 25 and the offset voltage Voffset of the voltage comparison circuit 27 is set in the capacitor C1.

  By setting the threshold voltage Vth in this way, it is possible to cancel the offset error generated in the current-voltage conversion circuit 25 and the voltage comparison circuit 27 in the comparison operation during the signal value detection period. Therefore, the current detector 19 can detect with high accuracy that the signal value has changed when the current Iin actually changes by the current threshold Idetect. Thereby, it is possible to prevent the occurrence of a communication error due to the shift of the current threshold value due to the influence of the offset error described above.

  Since the current threshold value Idetect can be accurately set to ½ of the signal current, the margin for noise current can be maximized. As a result, the resistance value of the detection resistor Rs can be lowered and the loss generated in the detection resistor Rs can be reduced as compared with the conventional configuration that is affected by the error described above. When there is a situation in which the noise current flows in a biased manner with respect to the signal value (H / L), the current threshold Idetect is set to be shifted from 1/2 of the signal current in order to prevent the margin from decreasing. May be.

(Second Embodiment)
A second embodiment will be described with reference to FIGS. 5 and 6. The current detection unit 41 receives a ternary (0, 1, 2) signal value corresponding to a change in the current Iin flowing through the power supply line 15, and outputs the signal value as a combination of the bit values B0 and B1. That is, the signal value 0 is (B1, B0) = (0, 0), the signal value 1 is (B1, B0) = (0, 1), and the signal value 2 is (B1, B0) = (1, 1). is there.

  In order to detect a ternary signal value from the current Iin, the current detection unit 41 includes two sets of threshold addition circuits 26A and 26B and two sets of voltage comparison circuits in addition to the current-voltage conversion circuit 25 and the control circuit 28. 27A and 27B are provided. The configurations of the threshold addition circuits 26A and 26B and the voltage comparison circuits 27A and 27B are the same as those of the threshold addition circuit 26 and the voltage comparison circuit 27 described above, respectively. In addition, in order to distinguish the components in these two sets of circuits, suffixes a and b are added to the reference numerals.

  The addition switch 33a and the feedback switch 35a are turned on when the signal φ3 is at the H level. Addition switch 33b is turned on when signal φ4 is at the H level. The feedback switch 35b is turned on when the signal φ1 is at the H level. The output switches 36a and 36b are turned on when the signal φ2 is at the H level.

  When the signal value is 0, only the load current flows through the power supply line 15. When the signal value is 1 or 2, the signal current is superimposed on the load current through the power line 15. The signal current when the signal value is 2 is larger than the signal current when the signal value is 1. Also in this embodiment, in order to maximize the margin for the noise current, 1/2 of the signal current when the signal value is 1 is set as the current threshold Idetect1, and the signal current and the signal value when the signal value is 1 are The median of the signal current at 2 is the current threshold Idetect2. The constant current circuits 32a and 32b output threshold setting currents Idetect1 / n and Idetect2 / n, respectively.

  The control circuit 28 sets the signal φ2 to the L level during the threshold setting period and turns off the output switches 36a and 36b. As described above, this is to prevent a voltage having an intermediate level other than the binary level of H / L from being output to the output terminals 37a and 37b. In the first half of the threshold setting period, the control circuit 28 sets the signals φ1 and φ3 to the H level and the signal φ4 to the L level, and turns on only the addition switch 33a and the feedback switches 35a and 35b. As a result, the threshold voltage Vth1 obtained by replacing Idetect in the equation (5) with Idetect1 is set in the capacitors C1a and C1b.

  In the latter half of the threshold setting period, the control circuit 28 sets the signals φ1 and φ4 to the H level and the signal φ3 to the L level, and turns on only the addition switch 33b and the feedback switch 35b. As a result, the threshold voltage Vth2 is set in the capacitor C1b by replacing Idetect in the equation (5) with Idetect2. Capacitor C1a holds threshold voltage Vth1. The feedback switch 35b may be changed to turn on when the signal φ4 is at the H level.

  Subsequently, in the signal value detection period, the control circuit 28 sets the signals φ1, φ3, and φ4 to the L level and the signal φ2 to the H level, turns off the addition switches 33a and 33b and the feedback switches 35a and 35b, and sets the output switches 36a and 36b. Turn on. As a result, the voltage comparison circuit 27A compares the voltage Va of the reference resistor Ra shown by the equation (3) with the threshold voltage Vth1, and outputs a bit value B0. The voltage comparison circuit 27B compares the voltage Va of the reference resistor Ra shown by the equation (3) with the threshold voltage Vth2 and outputs a bit value B1. According to the present embodiment, it is possible to receive signal values having three levels while maintaining the same effect as in the first embodiment.

(Third embodiment)
A third embodiment will be described with reference to FIG. The current detection unit 51 is obtained by replacing the current-voltage conversion circuit 25 with a current-voltage conversion circuit 52 in the current detection unit 19 shown in FIG. The current-voltage conversion circuit 25 and the current-voltage conversion circuit 52 are different in the configuration of the amplifier circuit.

  The amplifying circuit 53 of the current-voltage conversion circuit 52 includes an operational amplifier 30, a bias circuit 54, P-channel MOS transistors M1 and M2, gates (control terminals) connected in common, an N-channel MOS transistor Ma, and a first The resistor R1 to the fourth resistor R4 are configured. Transistors M1 to M3 correspond to first to third transistors, respectively.

  The resistor R1 is connected between the high potential side terminal (DC power supply 17 side) of the detection resistor Rs and the source of the transistor M1. The resistor R2 is connected between the low potential side terminal (sensor module 13 side) of the detection resistor Rs and the source of the transistor M2. The resistors R3 and R4 are connected between the drains of the transistors M1 and M2 and the power supply line 16, respectively. The resistance values of the resistors R1 and R2 are equal to each other, and the resistance values of the resistors R3 and R4 are also equal to each other.

  The non-ground side terminals (the drains of the transistors M1 and M2) of the resistors R3 and R4 are connected to the non-inverting input terminal and the inverting input terminal of the operational amplifier 30, respectively. The drain and source of the transistor Ma are connected between the source of the transistor M1 and the reference resistor Ra, and the transistor Ma operates using the output voltage of the operational amplifier 30 as the gate voltage (control voltage).

  The bias circuit 54 applies a gate voltage to the transistors M1 and M2 so that a bias current sufficiently smaller than the current Iin flows through the transistors M1 and M2. The gate of the P-channel MOS transistor Mb is connected in common with the drain and the gates of the transistors M1 and M2. A resistor Rb is connected between the power supply line 15 and the source of the transistor Mb, and a constant current circuit 55 is connected between the drain of the transistor Mb and the power supply line 16. The resistance value of the resistor Rb is equal to the resistance values of the resistors R1 and R2, but may be different.

  Next, the operation and effect of the current-voltage conversion circuit 52 will be described. Due to the negative feedback amplification by the operational amplifier 30, the voltage values of the resistors R3 and R4 are controlled to be equal except for the offset voltage of the operational amplifier 30, and the currents flowing through the transistors M1 and M2 are also equal. Since the transistors M1 and M2 are formed on the semiconductor substrate so as to have the same characteristics, the source potentials of the transistors M1 and M2 also coincide with each other under the condition that the gate potential is equal to the drain potential and the drain current is also equal.

  In this case, the current-voltage conversion circuit 52 is designed so that most of the voltage between the power supply lines 15 and 16 is borne by the transistors M1, M2, Ma, and Mb. That is, the transistors M1, M2, Ma, and Mb are configured by elements having a withstand voltage corresponding to the power supply voltage, and the operational amplifier 30 is configured by a circuit having a withstand voltage lower than the power supply voltage. Thereby, the offset voltage of the operational amplifier 30 becomes smaller than that in the first embodiment. As a result, the offset current Ioffset when the current Ia shown by the equation (1) flows through the reference resistor Ra becomes very small.

  The operations of the threshold addition circuit 26 and the voltage comparison circuit 27 in the threshold setting period and the signal value detection period are the same as those in the first embodiment described with reference to FIG. According to the present embodiment, the same effect as in the first embodiment can be obtained, and the offset error itself of the current-voltage conversion circuit 52 can be reduced, so that a signal value can be obtained with higher accuracy based on the change in the current Iin. be able to.

(Fourth embodiment)
The current detector 61 shown in FIG. 8 is obtained by changing the transistors M1 and M2 of the current detector 51 shown in FIG. The amplifier circuit 63 of the current-voltage conversion circuit 62 includes transistors M11 and M12 connected in cascode instead of the transistor M1, and includes transistors M21 and M22 connected in cascode instead of the transistor M2.

  The bias circuit 64 applies a bias voltage Vb1 to the gates of the transistors M11 and M21, and applies a bias voltage Vb2 to the gates of the transistors M12 and M22. Here, the bias voltages Vb1 and Vb2 are set so that the drain-source voltages of the transistors M12 and M22 are higher than the drain-source voltages of the transistors M11 and M21.

  Due to the same action as in the third embodiment, the source potentials of the transistors M11 and M21 become equal. In the present embodiment, the transistors M11 and M21 can be configured with lower breakdown voltage elements by setting the drain-source voltage as described above. As a result, the characteristic difference between the transistors M11 and M21 is further reduced, and the source potentials of the transistors M11 and M21 are more accurately matched. Therefore, the offset current Ioffset flowing through the current-voltage conversion circuit 62 is further reduced, and a signal value can be obtained with higher accuracy based on the change in the current Iin.

(Other embodiments)
As mentioned above, although preferred embodiment of this invention was described, this invention is not limited to embodiment mentioned above, A various deformation | transformation and expansion | extension can be performed within the range which does not deviate from the summary of invention.

  In the second embodiment, the configuration of receiving ternary signal values has been described. However, by increasing the number of threshold addition circuits 26 and voltage comparison circuits 27, signal values having multilevel levels of four or more levels can be obtained. Can receive. The second embodiment and the modification with four or more values can be applied to the third and fourth embodiments.

The current detection units 19, 41, 51, 61 described above are not limited to being applied to the airbag device 11 in which the ECU 12 and the sensor module 13 are combined. The present invention can be applied to any system in which the change in the load current flowing through the power supply lines 15 and 16 in each cycle including the threshold setting period and the signal value detection period is smaller than the change in the signal current.
The threshold addition circuit includes a configuration for adding a negative current, that is, a configuration for subtracting a positive current. This configuration can be applied to the case where the constant current circuit 24 outputs a negative current.

  In the drawings, 15 and 16 are power lines (lines), 19, 41, 51 and 61 are current detection units (current detection devices), 25, 52 and 62 are current-voltage conversion circuits, and 26, 26A and 26B are threshold values. 27, 27A and 27B are voltage comparison circuits, 28 is a control circuit, 29, 53 and 63 are amplifier circuits, 30 is an operational amplifier (amplifier), 32, 32a and 32b are constant current circuits, 33, 33a and 33b are Addition switch, 34, 34a and 34b are comparators, 35, 35a and 35b are feedback switches, 54 and 64 are bias circuits, Rs is a detection resistor, Ra is a reference resistor, R1, R2, R3 and R4 are first switches Reference numerals 1, 2, 3, 4, C 1 are capacitors, M 1, M 11, M 12 are first transistors, M 2, M 21, M 22 are second transistors, and Ma is a third transistor.

Claims (6)

Based on the magnitude relationship between the amount of change in the current flowing through the line (15, 16) and the current threshold value, the current detection device (19, 41, 51, 61)
An amplifying circuit (29) for causing a current corresponding to the line current to flow through the reference resistor by negative feedback amplification using a detection resistor (Rs), a reference resistor (Ra), and a voltage across the detection resistor provided in the line as inputs. , 53, 63), a current-voltage conversion circuit (25, 52, 62),
A constant current circuit (32, 32a, 32b) for outputting a threshold setting current corresponding to the current threshold, and an addition switch (33, 33a, 33b) provided in a path from the constant current circuit to the reference resistance ) Including a threshold value adding circuit (26, 26A, 26B),
A capacitor (C1), a comparator (34, 34a, 34b) for comparing the voltage of the reference resistor and the voltage of the capacitor, and a negative feedback path from the output terminal of the comparator to the input terminal to which the capacitor is connected A voltage comparison circuit (27, 27A, 27B) comprising feedback switches (35, 35a, 35b) forming
In the threshold setting period, by turning on the addition switch and the feedback switch, a threshold voltage corresponding to a current obtained by adding the current threshold to the line current is set in the capacitor, and a signal value In the detection period, by turning off the addition switch and the feedback switch, the control circuit generates the signal value by comparing the threshold voltage with the voltage generated in the reference resistor according to the line current (28). A current detection device comprising: The threshold value adding circuit (26A, 26B) and the voltage comparing circuit (27A, 27B) are provided corresponding to a plurality of current threshold values having different sizes, respectively.
The control circuit sequentially selects one current threshold value from the plurality of current threshold values during the threshold setting period, and the threshold value adding circuit corresponding to the selected current threshold value By turning on the addition switch and the feedback switch of the voltage comparison circuit, the threshold voltage corresponding to the selected current threshold is set in the capacitor of the voltage comparison circuit corresponding to the selected current threshold In the signal value detection period, by turning off all of the addition switch and the feedback switch, the threshold voltage having a magnitude different from the voltage generated in the reference resistor according to the line current The current detection device according to claim 1, wherein the signal value is generated by comparing the signal values. One end of the reference resistor is a ground potential,
The amplifier circuit (29)
An amplifier (30);
A first resistor (R1) connected between a high potential side terminal of the detection resistor and a non-inverting input terminal of the amplifier;
A second resistor (R2) connected between a low potential side terminal of the detection resistor and an inverting input terminal of the amplifier;
3. A transistor (Ma) connected between a non-inverting input terminal of the amplifier and the other end of the reference resistor and having an output voltage of the amplifier as a control voltage. The current detection device described. One end of the reference resistor is a ground potential,
The amplifier circuit (53, 63)
An amplifier (30);
First and second transistors (M1, M2) having control terminals connected in common;
A bias circuit (54, 64) for applying a bias voltage so that a bias current smaller than the line current flows to the first and second transistors;
A first resistor (R1) provided between the high potential side terminal of the detection resistor and the first transistor and between the low potential side terminal of the detection resistor and the second transistor, and having the same resistance value. And a second resistor (R2),
A third resistor (R3) and a fourth resistor (R4) provided between the first transistor and the ground potential and between the second transistor and the ground potential, respectively, and having the same resistance value;
A third transistor (Ma) connected between the non-line side terminal of the first resistor and the other end of the reference resistor and having an output voltage of the amplifier as a control voltage;
3. The current detection device according to claim 1, wherein the non-ground side terminals of the third and fourth resistors are connected to a non-inverting input terminal and an inverting input terminal of the amplifier, respectively.   The first, second, and third transistors have a withstand voltage according to the voltage of the line, and the amplifier has a withstand voltage according to the voltages of the third and fourth resistors. Item 5. The current detection device according to Item 4.   6. The current detection device according to claim 4, wherein the first and second transistors are cascode-connected.

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