JP5729833B2 - Base station apparatus, radio communication method, and radio communication system - Google Patents

Description

  The present invention uses the limited frequency resources and reduces the total transmission power on the transmission side while reducing the total transmission power on the transmission side in an environment in which strict restrictions on the circuit design are imposed by increasing the transmission distance in wireless transmission. In addition, the present invention relates to a base station apparatus, a radio communication method, and a radio communication system that improve signal-to-noise power ratio (SNR) and enable efficient transmission with high power consumption and large capacity.

  With the spread of the Internet in recent years, lines using optical fibers are already available in 90% of all households. In this way, the trend toward broadband has certainly progressed, but in reality there are areas where profitability is not expected due to the laying of optical lines, so how to eliminate the broadband zero area? There is currently no way to find a way to solve the problem. Such an area where profitability is not expected by laying an optical line is called an unprofitable area (conditionally disadvantaged area).

  As a countermeasure in such unprofitable areas, it is advantageous to use a wireless line. For example, a service using a wireless standard called WiMAX (Worldwide interoperability for microwave access) (registered trademark, the same shall apply hereinafter) is used. For this reason, a measure called “Region WiMAX” has been implemented, in which 10 [MHz] is reserved for the frequency channel for this purpose, and the WiMAX service using this frequency channel is applied mainly to conditionally disadvantaged areas. In WiMAX used for this measure, for example, a base station performs signal transmission with a large transmission power of about 10 [W], and as a result, an area with a radius of about 3 km can be covered by one station. .

In general, in an environment where the line of sight is clear, the received signal strength is inversely proportional to the square of the distance due to propagation between the transmitting station and the receiving station. In the case of out of line of sight, the received signal strength is inversely proportional to the 3rd to 4th power of the distance, which places more severe restrictions on the circuit design. Even if a line of sight is assumed, in order to extend the transmission distance by a factor of 2, it is necessary to increase the transmission power to 2 ² = 4 times, and a transmission amplifier with higher linearity is required. However, such a transmission amplifier is expensive, and when such a transmission amplifier is used, the power efficiency is remarkably lowered, so that the power consumption increases rapidly.

  In recent years, environmental problems have attracted particular attention, and low power consumption of infrastructure including radio is required, and inefficient communication using a high-output transmission amplifier is not preferable. As a method for solving such a problem, for example, as described in Non-Patent Document 1, coherent transmission with a plurality of relay stations interposed is effective. In Non-Patent Document 1, non-regenerative relay is assumed for relay, but the point of this coherent transmission depends on whether the form of relay is “non-regenerative relay” or “regenerative relay”. However, the transmission is performed so that the signals are combined in the same phase on the receiving side. As another form when performing such coherent transmission, there is a distributed antenna system as described in Non-Patent Document 2, for example.

A distributed antenna system is a radio module or remote base station in which a plurality of antennas installed in a distributed manner in one control station (strictly speaking, an antenna is combined with a device that performs optical / electrical conversion, signal amplification, etc.) ), And the control station and each antenna are connected by an optical fiber or the like.
As another form, it is possible to adopt a configuration (wireless relay system) in which a plurality of relay stations are wirelessly connected to one base station. In this case, the base station serves as a control station, and the relay station serves as an antenna or a radio module, which constitutes a distributed antenna system as a whole, but differs in that the base station and the relay station are connected by radio. is there.
In any case, coherent transmission is performed in which a plurality of antennas (relay stations) transmit on the receiving terminal side so that the signals are combined in the same phase. The detailed description will be given below.

[Outline of coherent transmission system in the prior art]
(Wireless relay system)
FIG. 2 is a diagram showing an outline of a wireless relay system in the prior art.
As shown in the figure, the wireless relay system includes a transmission station 901, N ₁ relay stations 902-1 to 902 -N _1, and a reception station 903. The transmitting station 901, once transmitted to the relay station _{902-1~902-N 1} radio packets destined receiving station 903. The relay stations 902-1 to 902-N ₁ perform various reception signal processing on the signal received from the transmission station 901, and reproduce (restore) the wireless packet transmitted by the transmission station 901. Then, each relay station _{902-1~902-N 1} transmits to the receiving station 903 the same radio packet reproduced at the same time. In this case, each relay station _{902-1~902-N 1,} as the signal, each transmitted are received by the same phase at the receiving station 903 adjusts the phase of the transmission signal. In the receiving station 903, all signals transmitted from the relay station _{902-1~902-N 1} is received is synthesized on the transmission path. At this time, the signal transmitted from the relay station 902-1~902-N ₁ is, if the received at comparable received power at the receiving station 903, the signal after synthesis, is synthesized the N ₁ times the amplitude with respect to the previous signal. Further, the received power is proportional to the square of the amplitude _(N ¹⁾ is ^doubled.

Here, a comparison is made between the case where the number of relay stations 902 in the wireless relay system is one and the case of N ₁ stations. In order to make the evaluation condition fair, a single relay station 902 transmits the transmission power as P when relaying by _one station, and relay stations 902-1 to 902-N when relaying by N1 station. ₁ is compared with the transmission power of P / N ₁ (the condition that the total transmission power is constant). When transmitted from the relay station _{902-1~902-N 1} of N ₁ station, signals transmitted from the relay station _{902-1~902-N 1} is synthesized in the transmission path, the relay station 902-1～902- Compared with the received signal from any _one of N ₁ stations, the amplitude of the received signal at the receiving station 903 is N ₁ times, and as a result, the total received power is (N ₁ ) ² times. However, when transmission is performed at N ₁ station, the transmission power per relay station 902 is 1 / N ₁ when transmission is performed at a single relay station 902. Therefore, the received power is (1 / N ₁ ) × (N ₁ ) ² = N ₁ times.
That is, although the total transmission power of the relay stations 902-1 to 902-N ₁ is constant, the reception power at the reception station 903 is N ₁ times as compared with the case of relaying at one station, and the line gain is It is possible to earn 10 × Log ₁₀ N ₁ [dB].

(Distributed antenna system)
FIG. 3 is a diagram showing an overview of a distributed antenna system in the prior art.
As shown in the figure, the distributed antenna system includes remote base stations 912-1 to 912-3 that form three cells 911-1 to 911-3 that perform cooperative communication, and a plurality of terminal devices 913-1. 913-6 and a control station 914 connected to each of the remote base stations 912-1 to 912-3 via an optical fiber 915. The optical fiber 915 connecting each remote base station 912-1 to 912-3 and the control station 914 may be a coaxial cable or the like. In addition, here, the description will be made assuming three cells 911-1 to 911-3 and three remote base stations 911-1 to 912-3, but in general, the number may be other than three.

Each remote base station 912-1 to 912-3 communicates with each terminal device 913-1 to 913-6 located in the cell formed by each using the same frequency channel. The control station 914 controls the remote base stations 912-1 to 912-3 via the optical fiber 915. In order to perform communication using the same frequency channel, each terminal device 913-1 to 913-6 can simultaneously receive signals transmitted from a plurality of remote base stations 912-1 to 912-3. For example, the terminal device 913-4 can receive signals from all the remote base stations 912-1 to 912-3.
Here, if the channel information between each of the remote base stations 912-1 to 912-3 and the terminal device 913-4 is known, each of the remote base stations 912-1 to 912-3 has the terminal device 913-3. 4, transmission weight multiplication can be performed so that signals transmitted from the remote base stations 912-1 to 912-3 have the same phase in the terminal device 913-4. In this case, since the signal received by the terminal device 913-4 is synthesized in phase, the received power increases. As a result, the communication characteristics in the terminal device 913-4 are improved. Such control of signal processing for performing in-phase synthesis is all performed by the control station 914, and the remote base stations 912-1 to 912-3 operate according to instructions from the control station 914.

  In the distributed antenna system, the control station 914 and each remote base station 912-1 to 912-3 are connected by an optical fiber 915, and a signal transferred on the optical fiber 915 is transmitted to each remote base station 912-2. In 1 to 912-3, an electrical signal to be transmitted on the wireless line is generated by performing optical / electrical conversion, and this is transmitted from the antenna after processing such as signal amplification. By utilizing such control, the signal processing functions for in-phase synthesis on the receiving side are aggregated in the control station 914 that has grasped all channel information, and as a result, each remote base station 912-1- It is possible to improve the communication quality while avoiding the uncertainty of the phase control in 912-3.

  In the strict sense of the distributed antenna system, each of the remote base stations 912-1 to 912-3 simultaneously performs multi-user MIMO (Multiple MIMO) that performs spatial multiplexing on the same frequency as a plurality of terminal devices 913-1 to 913-6. It is possible to further improve the characteristics by using (Input Multiple Output) technology. The control when using the multi-user MIMO technology is performed by using multiple transmission antennas, so that in-phase synthesis of desired signals in the terminal device 913 and null control for removing interference signals between different terminal devices 913, Is basically control based on coherent transmission.

[Overview of channel feedback in coherent transmission]
In order to perform coherent transmission, it is necessary to grasp the channel state between the transmitting and receiving stations. In order to ensure that signals transmitted from a plurality of transmitting stations or relay stations reach the receiving station in the same phase, the transmitting station and the relay station grasp the state of the channel between the receiving station and the channel. This is because a signal is transmitted using a transmission weight according to the state.

  FIG. 4 is a flowchart showing channel feedback processing in the prior art. There are roughly two types of channel feedback methods in the prior art. Here, “(A) a direct method” for directly acquiring the channel estimation result of the forward link and “(B) an indirect method” for performing conversion estimation using information on the backward link will be described.

In general, the channel information of the forward link and the backward link in the reverse direction do not match. This is because the combination of the high-power amplifier on the transmission side and the low-noise amplifier on the reception side used in the forward link is different from the combination of the high-power amplifier on the transmission side and the low-noise amplifier on the reception side used in the backward link. This is because the complex phase and amplitude differ between the information and the channel information of the backward link.
However, by performing a conversion process (calibration process), which will be described later, it is possible to estimate conversion of the forward link information from the backward link channel information. In the following description, when the “remote base station” and the “relay station” in the above description are not distinguished, they are referred to as “wireless modules”.

FIG. 4A is a flowchart showing the processing of the direct method. As shown in the figure, in the direct method, when channel information estimation is started (step S901), a radio packet including a channel estimation preamble signal and the like is transmitted from each radio module to the terminal device (step S902). ).
The terminal apparatus receives the wireless packet transmitted from each wireless module, and performs channel estimation using a preamble signal or the like included in the received wireless packet (step S903). In the terminal device, this channel estimation result is accommodated in a “radio packet for accommodating control information” and transmitted to the radio module (step S904).
The wireless module receives the “radio packet for accommodating control information” transmitted from the terminal device, and acquires channel information (step S905). Further, the wireless module stores the received channel information in a memory, constructs a database related to the channel information (step S906), and ends the process (step S907).

FIG. 4B is a flowchart showing processing of an indirect method. As shown in the figure, in the indirect method, when channel information estimation is started (step S908), a radio packet including a preamble signal for channel estimation is transmitted from the terminal device to the radio module (step S909). .
The wireless module receives a wireless packet transmitted from the terminal device, and performs channel estimation using a preamble signal included in the wireless packet (step S910). The wireless module performs conversion processing on the estimation result of the channel information in the backward link, and acquires the channel information on the forward link side (step S911).

The conversion process for calculating the channel information in the forward link from the channel information in the backward link can be performed by using a coefficient that corrects the difference between the high power amplifier in the forward link and the low noise amplifier in the backward link. is there. Specifically, the conversion process in step S911 can be performed by multiplying the channel information in the backward link by a coefficient for correcting the difference between the high power amplifier and the low noise amplifier.
Further, the radio module stores the channel information in the backward link received from the terminal device and the channel information in the forward link obtained by the conversion process in a memory, and constructs a database for storing the channel information (step S912). The process is terminated (step S913).

  In this way, the channel information is acquired in advance, and generally the transmission weight is calculated based on this channel information when actually communicating. In addition, since channel information changes with time, it is common to update periodically according to a situation, for example. Further, in the above, the channel information may be stored as a database in a control station other than the wireless module.

  Taking a distributed antenna system as an example, this transmission weight calculation processing is generally not performed individually by each wireless module, but is collectively performed by a control station in a centralized manner. In particular, when communication is performed on the same frequency channel simultaneously with a plurality of terminal devices by multiuser MIMO, the transmission weight cannot be calculated unless all channel information is used. However, if the transmission is limited to one-to-one communication with one terminal device instead of multi-user MIMO, a transmission weight that cancels the rotation of the complex phase on the transmission path obtained from the channel information (that is, Since the complex phase becomes a constant when the channel information and the transmission weight are multiplied in all the wireless modules, it is possible to perform processing individually in the wireless modules.

  Here, a description will be given taking as an example the case of using an OFDM (Orthogonal Frequency Division Multiplexing) modulation scheme and an SC-FDE (Single Carrier Frequency Domain Equalization) scheme. In the OFDMA (Orthogonal Frequency Division Multiple Access) method, processing in the physical layer basically uses the OFDM modulation method, and therefore the following description treats OFDM and OFDMA as equivalent methods. I will decide.

[Phased array antenna technology]
As a technique similar to coherent transmission, there is a phased array antenna technique using many antenna elements (for example, Non-Patent Document 3).
FIG. 5 is a diagram showing the principle of the phased array antenna. The figure shows a phased array antenna in which five antenna elements 961-1 to 961-5 are linearly arranged with a distance d therebetween. In the phased array antenna, when the directivity in the angle θ direction is formed with respect to the arrangement direction of the antenna elements 961-1 to 961-5, there is a difference in path length for each antenna element 961-1 to 961-5 with respect to the direction. In consideration of dCosθ, adjustment may be made to each of the signals transmitted and received using each of the antenna elements 961-1 to 961-5 so as to perform in-phase synthesis.

Here, when the wavelength of a signal to be transmitted / received is λ, a signal whose phase is shifted by ((2πdCosθ) / λ) between adjacent antenna elements 961-1 to 961-5 is output in the direction of the angle θ. A directivity can be formed. This phase difference ((2πdCosθ) / λ) may be given to a signal to be transmitted / received in an analog manner using a phase shifter or may be given in digital signal processing.
In the phased array antenna, the antenna gain with respect to a predetermined angle direction can be obtained in this way. In general, since a sublobe showing a fine wave of undulation is generated around the main lobe direction in which the directivity gain is maximized, in order to reduce the influence and stably operate the main lobe, the antenna element 961- The interval d between 1 and 961-5 is set to λ / 2 or less.

  However, as the distance between the antenna elements 961-1 to 961-5 is shortened with respect to the wavelength λ, the antenna elements 961-1 to 961-5 are coupled to each other and various factors, compared with the case of simple in-phase synthesis. The gain is greatly reduced. In this case, the signals transmitted and received from the individual antenna elements 961-1 to 961-5 are different from the waves that can simply add the amplitudes as independent waves at the transmission and reception points, as if many antenna elements 961-1 to 961-5. As a whole, one virtual antenna element is configured, and one signal (wave) is transmitted from the virtual antenna element. In this respect, it can be regarded as a phenomenon different from coherent transmission in which simple in-phase synthesis is realized.

[About multi-user MIMO technology]
(Outline of multi-user MIMO)
Coherent transmission and phased array antenna technology are basically technologies that improve channel gain, and in order to increase the channel capacity when covering a wide service area with one base station device, another wireless communication Technology is required. On the other hand, since frequency resources are limited, here, for example, a multi-user MIMO technique studied in Non-Patent Document 4 will be described as a technique for using limited resources with high frequency utilization efficiency.

  FIG. 6 is a schematic diagram illustrating a configuration example of a multi-user MIMO system. As shown in the figure, the multi-user MIMO system includes a base station device 801 and terminal devices 802-1, 802-2, and 802-3 (terminal devices # 1 to # 3). There are actually a large number of terminal devices 802 accommodated in one base station device 801, but several of these are selected (terminal devices 802-1 to 802-3 in the figure) to perform communication. Each terminal device 802 generally has a smaller number of transmission / reception antennas than the base station device 801. For example, a case where communication (downlink) from the base station apparatus 801 to the terminal apparatus 802 will be described.

  Base station apparatus 801 forms a plurality of directional beams using a large number of antenna elements. For example, consider a case where three MIMO channels are allocated to each of the terminal devices 802-1 to 803 and nine signal sequences are transmitted as a whole. At that time, the signal transmitted to the terminal device 802-1 is adjusted so that the directivity gain becomes extremely low in the direction of the terminal device 802-2 and the terminal device 802-3. As a result, the terminal device 802 2 and the terminal device 802-3. Similarly, the signal transmitted to the terminal device 802-2 is adjusted so that the directivity gain becomes extremely low in the direction of the terminal device 802-1 and the terminal device 802-3. The same processing is performed on the terminal device 802-3. The reason for performing the directivity control in this way is that, for example, in the terminal device 802-1, there is no way of knowing information of signals received by the terminal device 802-2 and the terminal device 802-3. A cooperative reception process is not possible. That is, in the reception process of only the terminal device 802-1 having only three, it is very strict to separate all nine signal sequences. Therefore, interference separation is performed in advance on the transmission side so that each terminal device 802-1 to 802-3 does not receive signals from other terminal devices 802.

The above is the outline of the existing multi-user MIMO system. Next, a method for forming a directional beam will be described below. Here, a case will be described in which base station apparatus 801 includes nine antenna elements, and each terminal apparatus 802-1 to 802-3 includes three antenna elements. For example, in FIG. 6, channel information between the j-th (j = 1,..., 9) antenna element of the base station apparatus 801 and the first antenna element of the terminal apparatus 802-1 is denoted as h _1j . . Each antenna element of the base station apparatus 801 (j = 1, ..., 9) and, the first row vectors _{h 1} using the channel information of the antenna element _{(h 11} of the terminal apparatus _{802-1, h} 12, h ₁₃ ,..., H ₁₈ , h ₁₉ ). Similarly, channel information between the j-th antenna element of the base station apparatus 801 and the second antenna element and the third antenna element of the terminal apparatus 802-1 is denoted as h _2j and h _3j and corresponds. The row vectors h ₂ and h ₃ are denoted as (h ₂₁ , h ₂₂ , h ₂₃ ,..., H ₂₈ , h ₂₉ ) and (h ₃₁ , h ₃₂ , h ₃₃ ,..., H ₃₈ , h ₃₉ ). Pretend similar serial number to the antenna elements of the terminal apparatus 802-2 and the terminal device 802-3, the row vector _{h 4 ~h 9 (h 41,} h 42, h 43, ..., h 48, h 49) To (h ₉₁ , h ₉₂ , h ₉₃ ,..., H ₉₈ , h ₉₉ ).

In addition, nine systems of signals transmitted by the base station apparatus 801 are expressed as t ₁ to t ₉ , and a column vector having these components as T x ^[all] = (t ₁ , t ₂ , t ₃ ,..., T ₈ , t ₉ ) ^T Here, the letter T on the right shoulder indicates transposition of a vector or a matrix. Similarly, received signals at the nine antenna elements of the terminal devices 802-1 to 80-3 are denoted as r ₁ to r ₉ , and a column vector having these components as components Rx ^[all] = (r ₁ , r ₂ , r ₃ ,..., r ₈ , r ₉ ) ^T Finally, a matrix having the row vectors h _{1 to} h ₉ as the first to ninth row components is denoted as an overall channel information matrix H ^[all] .
In this case, the relationship of the following formula (1) is established for the entire multiuser MIMO system.

  On the other hand, in order to perform transmission directivity control, a transmission weight matrix W of 9 rows and 9 columns is introduced, and the equation (1) is rewritten as the following equation (2).

Furthermore, the transmission weight matrix W is decomposed into column vectors _{w 1 ~w 9, W = (} w 1, w 2, w 3, ..., w 8, w 9) If the denoted ^"H in the formula (2) ^{[ all]} · W ”can be expressed as the following equation (3).

Here, for example, the multiplication of the six row vectors h _{4 to} h ₉ and the three column vectors w _{1 to} w ₃ (the sum of the multiplication of each component, which differs from the inner product in the case of a complex vector) is all zero Consider that the values of w ₁ to w ₃ are selected. At the same time, the multiplication of the row vector _h 1 to h ₃ and _h 7 to h ₉ column vector _w 4 to w _6, multiplication becomes all zero row vector _h 1 to h ₆ column vector _w 7 to w ₉ Thus, the values of w ₄ to w ₉ are selected.
Then, the 9 × 9 matrix H ^[all] · W shown in Equation (3) can be expressed as the following Equation (4) using a 3 × 3 partial matrix.

In Equation (4), H ^[1] , H ^[2] , and H ^[3] are 3-by-3 matrices, and “0” is a 3-by-3 matrix with all components zero. By selecting a transformation matrix that satisfies such conditions as the transmission weight matrix W, the equation (4) can be decomposed into three relational expressions represented by the following equations (5-1) to (5-3).

^{_{Here, Tx [1] = (t}} 1, t 2, t 3) T, Tx [2] = (t 4, t 5, t 6) T, Tx [3] = (t 7, t 8, t ^{_{9) T, Rx [1]}} = (r 1, r 2, r 3) T, Rx [2] = (r 4, r 5, r 6) T, Rx [3] = (r 7, r 8, r ₉ ) ^T In this way, it can be considered that one base station apparatus performs MIMO communication on a one-to-one basis, that is, so-called single-user MIMO communication is communicating in three systems simultaneously in parallel.

Next, an example of a method for determining the transmission weight vectors w _{1 to} w ₉ will be described below. As a procedure, transmission weight vectors w _{1 to} w ₃ for the terminal device 802-1 are determined, and transmission weight vectors w _{4 to} w _{6 for} the terminal device 802-2 and transmission weight vectors w ₇ for the terminal device 802-3 are sequentially set. to determine the ~w _9.
First, as a first step, six basis vectors e _{4 to} e ₉ in a six-dimensional subspace spanned by six row vectors h _{4 to} h ₉ for the terminal devices 802-2 and 802-3 are obtained. There are various methods other than the Gram Schmidt orthogonalization method. The Gram Schmidt orthogonalization method will be described as an example here.
First, paying attention to one row vector h ₄ , a vector having an absolute value of 1 in this direction is set as a base vector e ₄ . The basis vector e ₄ is expressed as the following equation (6).

( ^H ₄ h ₄ ^H ) in Equation (6) is a scalar quantity that means the square of the absolute value of the same vector, and division by the square root of this value means normalization of the row vector h ₄ . “H ₄ ^H ” is a Hermitian conjugate vector for the row vector h ₄ , and is a vector obtained by transposing the row and column and taking the complex conjugate of each component.
Next, focusing on the row vector h _5, after obtaining the row vector h _{5 'canceling} the basis vectors e ₄ direction component from among the row vectors further normalized. The row vector h ₅ ′ and the basis vector e ₅ are expressed by the following expressions (7-1) and (7-2).

( ^H ₅ e ₄ ^H ) in Equation (7-1) means the projection of the row vector h _{5 in} the direction of the base vector e ₄ . The same processing is performed as in the following equation (8-1) and the following equation (8-2).

Here, the range of the summation of Σ in the equation (8-1) is the summation for the integer i of 4 ≦ i ≦ (j−1) (j is an integer of 5 to 9). That is, it means that the component in the defined vector direction that has already been determined is canceled. In this way, six basis vectors e _{4 to} e ₉ can be obtained.
Next, as a second step, transmission weight vectors w _{1 to} w ₃ for the terminal device 802-1 are obtained. First, the components of the 6-dimensional subspace spanned by the base vectors e _{4 to} e ₉ are canceled from the row vectors h _{1 to} h ₃ . Specifically, it is represented by the following formula (9).

Here, j in the formula (9) is an integer of 1 to 3, and the range of the sum of Σ is the sum for the integer i of 4 ≦ i ≦ 9. The three-dimensional space spanned by the three vectors of the row vectors h ₁ ′ to h ₃ ′ thus obtained is orthogonal to any of the above-described row vectors h _{4 to} h ₉ . If three vectors in this three-dimensional space (not necessarily an orthogonal vector) are selected and the complex conjugate vector of the vector is set as transmission weight vectors w _{1 to} w ₃ , another terminal device 802-2, Interference with 802-3 can be suppressed.
Note that any method may be used for selecting the three vectors. For example, if three orthogonal vectors that form a unitary matrix obtained by performing singular value decomposition are used, the sub-spaces that do not interfere with other terminal devices 802 are used. The eigenmode transmission limited to 1 is possible, and efficient transmission becomes possible.

Finally, as a third step, if the same processing is performed for the terminal device 802-2 and the terminal device 802-3, finally the entire transmission weight vectors w _{1 to} w ₉ can be obtained. .
The above is how to obtain the transmission weight matrix W.

FIG. 7 is a flowchart showing a procedure for calculating the transmission weight matrix W in the multiuser MIMO system. First, in calculating the transmission weight matrix W, channel information matrices H for all the terminal devices 802 are acquired (step S801). When a serial number is assigned to the destination terminal device 802 and the variable indicating the serial number is k, k is first initialized (step S802). Further, k is counted up (step S803), and partial channel information (herein expressed as H ^main for convenience) corresponding to the terminal device 802 (# 1) corresponding to the current value indicated by k is extracted (step S804). ), And a partial channel information matrix (indicated here as H ^sub for convenience) for the other terminal device 802 is extracted (step S805).

Further, an orthogonal basis vector of a subspace spanned by each row vector of the partial channel matrix H ^sub is calculated, and this is set as a basis vector {e _j } (step S806). Next, as processing corresponding to Equation (9), the component related to the basis vector {e _j } obtained in step S806 from the partial channel information matrix H ^main for the terminal device 802 (# 1) of interest is canceled, Is a matrix to H ^main (step S807). Here, in step S807, H with “˜ (tilde)” added thereto is denoted as “˜H”. Hereinafter, similarly, in a mathematical expression or the like, when a character such as “^ (hat)” is written on the character, the symbol is written before the character.

Further, an arbitrary orthogonal basis vector of the subspace spanned by the row vectors of the matrix to H ^main is calculated, and this is set as the basis vector {e _i } (step S808). Here, as the arbitrary base vector, for example, a vector constituting the right singular matrix when the matrix ~ H ^main is subjected to singular value decomposition may be selected. Thereafter, a transmission weight vector {w _j } relating to the signal of the terminal device 802 (# 1) is determined as a Hermitian conjugate vector (a column vector obtained by transposing the complex conjugate vector) of each vector of the basis vector {e _i } (step S809). ).

Here, it is determined whether or not the transmission weight vectors of all destination terminal devices 802 have been determined (step S810), and if there are remaining terminal devices 802, the processing from step S803 to step S809 is repeated. If the transmission weight vectors of all the terminal devices 802 have been determined, the transmission weight matrix W is determined as a matrix having the transmission weight vector {w _j } as each column vector (step S811), and the process ends.
Since channel information generally differs for each frequency component, if a wideband signal, for example, a signal using the OFDM modulation method, a similar transmission weight is calculated for each frequency component, that is, for each subcarrier. Become. Here also, since the terminal device 802-1～802-3 has taken explain if provided by three elements of the antenna each example, the orthogonal basis vectors of the subspace spanned by each row vector of the ^{to H main} step S808 In the case where the terminal device includes only one antenna, the step S808 simply corresponds to normalizing a row vector corresponding to ~ H ^main .

(Multi-user MIMO device configuration example)
FIG. 8 is a schematic block diagram showing an example of the configuration of the base station apparatus 80 in the multiuser MIMO system. As shown in the figure, the base station apparatus 80 includes a transmission unit 81, a reception unit 85, an interface circuit 87, a MAC layer processing circuit 88, and a communication control circuit 820. The MAC layer processing circuit 88 has a scheduling processing circuit 881.
The base station apparatus 80 inputs / outputs data to / from an external device or network via the interface circuit 87. The interface circuit 87 detects data to be transferred on the wireless line among the input data, and outputs the detected data to the MAC layer processing circuit 88. The MAC layer processing circuit 88 performs processing related to the MAC layer in accordance with an instruction from the communication control circuit 820 that performs management control of the operation of the entire base station apparatus 80. Here, the processing related to the MAC layer includes conversion of data input / output by the interface circuit 87 and data transmitted / received on the wireless line, addition of header information of the MAC layer, and the like. In this process, the scheduling processing circuit 881 performs various scheduling processes including a combination of terminal devices that simultaneously perform spatial multiplexing in multiuser MIMO transmission. The scheduling processing circuit 881 outputs the scheduling result to the communication control circuit 820.
In multi-user MIMO, signals are transmitted to a plurality of terminal devices at a time, so that a plurality of signal sequences are output from the MAC layer processing circuit 88 to the transmission unit 81.

FIG. 9 is a schematic block diagram illustrating an example of the configuration of the transmission unit 81 in the base station apparatus 80 in the multiuser MIMO system. As shown in the figure, the transmission unit 81 includes transmission signal processing circuits 811-1 to 811-L (L is an integer of 2 or more) and addition synthesis circuits 812-1 to 812-K (K is an integer of 2 or more). ), IFFT (Inverse Fast Fourier Transform) & GI (Guard Interval) assigning circuits 813-1 to 813 -K, and a D / A (Digigal / Analogue) converter 814- 1-814-K, local oscillator 815, mixers 816-1 to 816-K, filters 817-1 to 817-K, high power amplifiers (HPA) 818-1 to 818-K, and antenna element 819 -1 to 819-K and a transmission weight processing unit 830. The transmission signal processing circuits 811-1 to 811-L and the transmission weight processing unit 830 are connected to the communication control circuit 820 shown in FIG.
The transmission weight processing unit 830 includes a channel information acquisition circuit 831, a channel information storage circuit 832, and a multiuser MIMO (MU-MIMO) transmission weight calculation circuit 833.
Here, the subscript L of the transmission signal processing circuits 811-1 to 811-L in FIG. The subscript K of the circuits from the adder / synthesizer circuit 812-1 to 812-K to the antenna elements 819-1 to 819-K represents the number of antenna systems provided in the base station apparatus 80.

  In multi-user MIMO, in order to transmit signals to a plurality of terminal devices at a time, a plurality of signal sequences are input from the MAC layer processing circuit 88 to the transmission unit 81, and the input plurality of signal sequences are transmitted signal processing. Input to the circuits 811-1 to 811-L. The transmission signal processing circuits 811-1 to 811-L, when data to be transmitted to each destination terminal device (data input # 1 to #L) are input from the MAC layer processing circuit 88, wireless transmission is performed via a wireless line. A packet is generated and modulated. Here, for example, if the OFDM modulation method is used, the signal of each signal series is subjected to modulation processing for each frequency component. Further, the baseband signal subjected to modulation processing is multiplied by a transmission weight for each frequency component. The signal multiplied by the transmission weight corresponding to each of the antenna elements 819-1 to 819 -K is subjected to the remaining signal processing as necessary, and is added and synthesized as a sampling signal of the transmission signal in the baseband 812-1. 812-K.

  The signals input to the adder / synthesizers 812-1 to 812-K are synthesized for each frequency component. The synthesized signal is converted from a signal on the frequency axis into a signal on the time axis by IFFT & GI adding circuits 83-1 to 813-K, and further, a guard interval is inserted or between OFDM symbols (block for SC-FDE). (Between transmission blocks) is processed, and for each antenna element 819-1 to 819-K, a D / A converter 814-1 to 814-K converts a digital sampling data into a baseband analog signal. Is converted to Further, each analog signal is multiplied by the local oscillation signal input from the local oscillator 815 by the mixers 816-1 to 816-K and up-converted to a radio frequency signal. Here, since the signal is included in the frequency component outside the band of the channel to be transmitted in the up-converted signal, the frequency component outside the band is removed by the filters 817-1 to 817-K, and the electrical signal to be transmitted is transmitted. A typical signal. The generated signals are amplified by the high power amplifiers 818-1 to 818 -K and transmitted from the antenna elements 819-1 to 819 -K.

In FIG. 9, after adding and synthesizing the signals of the respective frequency components by the adding and synthesizing circuits 812-1 to 812-K, processing such as IFFT processing, insertion of guard intervals, waveform shaping, and the like is performed. The signal processing circuits 811-1 to 811-L may perform these processes, and the IFFT & GI giving circuits 83-1 to 813-K may be omitted. In this case, the remaining signal processing as necessary after transmission weight multiplication in the transmission signal processing circuits 811-1 to 811-L refers to IFFT processing, insertion of guard intervals, waveform shaping, and the like.
Note that the transmission weights multiplied by the transmission signal processing circuits 811-1 to 811 -L are acquired from the multiuser MIMO transmission weight calculation circuit 833 provided in the transmission weight processing unit 830 during signal transmission processing. In the transmission weight processing unit 830, channel information is separately acquired by the channel information acquisition circuit 831 and stored in the channel information storage circuit 832 while being updated sequentially. At the time of signal transmission, the multiuser MIMO transmission weight calculation circuit 833 reads channel information corresponding to the destination station from the channel information storage circuit 832 and calculates a transmission weight based on the read channel information. Multi-user MIMO transmission weight calculation circuit 833 outputs the calculated transmission weight to transmission signal processing circuits 811-1 to 811-L.
Further, the communication control circuit 820 manages control related to the entire communication such as management of the destination station and overall timing control. The communication control circuit 820 outputs information indicating the destination station and the like to the transmission weight processing unit 830 that performs signal processing related to the calculation of the transmission weight described above.

FIG. 10 is a schematic block diagram illustrating an example of the configuration of the reception unit 85 in the base station apparatus 80 in the multiuser MIMO system. As shown in the figure, the base station apparatus 80 includes antenna elements 851-1 to 851-K, low noise amplifiers (LNA) 852-1 to 852-K, a local oscillator 853, and mixers 854-1 to 854-. K, filters 855-1 to 855-K, A / D (analogue / digital) analog converters 856-1 to 856-K, and FFT (Fast Fourier Transform) circuit 857-1 ˜857-K, reception signal processing circuits 858-1 to 858 -L, and a reception weight processing unit 860. Reception signal processing circuits 858-1 to 858 -L and reception weight processing unit 860 are connected to communication control circuit 820 shown in FIG. 8.
Reception weight processing section 860 includes channel information estimation circuit 861 and multiuser MIMO (MU-MIMO) reception weight calculation circuit 862.

  Signals received by the antenna elements 851-1 to 851-K are amplified by the low noise amplifiers 852-1 to 852-K. The amplified signal and the local oscillation signal output from the local oscillator 853 are multiplied by mixers 854-1 to 854-K, and the amplified signal is down-converted from a radio frequency signal to a baseband signal. Since the down-converted signal includes frequency components outside the frequency band to be received, the out-of-band components are removed by the filters 855-1 to 855-K. The signal from which the out-of-band component is removed is converted into a digital baseband signal by the A / D converters 856-1 to 856-K. All digital baseband signals are input to FFT circuits 857-1 to 857 -K, and signals on the time axis are converted into signals on the frequency axis (separated into signals of each frequency component) at a predetermined symbol timing. The signals separated into the frequency components are input to the reception signal processing circuits 858-1 to 858 -L and also input to the channel information estimation circuit 861.

  In the channel information estimation circuit 861, the antenna element of each terminal device and the base station device 80 based on a known signal for channel estimation separated into each frequency component (such as a preamble signal added to the head of the radio packet). Channel information between each of the antenna elements 851-1 to 851-K is estimated for each frequency component, and the estimation result is output to the multiuser MIMO reception weight calculation circuit 862. Multi-user MIMO reception weight calculation circuit 862 calculates reception weights to be multiplied for each frequency component based on the input channel information. At this time, reception weights for synthesizing signals received by the antenna elements 851-1 to 851-K are different for each signal series, and reception signal processing circuits 858-1 to 858-L corresponding to the signal series to be extracted. Input to each.

In the reception signal processing circuits 858-1 to 858 -L, the signal for each frequency component input from the FFT circuits 857-1 to 847 -K is multiplied by the reception weight input from the multiuser MIMO reception weight calculation circuit 862. Then, the signals received by the antenna elements 851-1 to 851-K are added and synthesized for each frequency component. The reception signal processing circuits 858-1 to 858 -L perform demodulation processing on the added and combined signals and output the reproduced data to the MAC layer processing circuit 88.
Here, different received signal processing circuits 858-1 to 858 -L perform signal processing of different signal sequences. The MAC layer processing circuit 88 performs processing related to the MAC layer (for example, conversion between data input / output to / from the interface circuit 87 and data transmitted / received on the wireless line, termination of header information of the MAC layer, etc.). Do. In this process, the scheduling processing circuit 881 performs various scheduling processes including a combination of terminal apparatuses that simultaneously perform spatial multiplexing in multiuser MIMO transmission, and outputs a scheduling result to the communication control circuit 820. The received data processed by the MAC layer processing circuit 88 is output to an external device or network via the interface circuit 87.
In addition, the communication control circuit 820 manages control related to overall communication such as management of a transmission source terminal device and overall timing control. In addition, information indicating a transmission source terminal device or the like is input from the communication control circuit 820 to the reception weight processing unit 860 that performs signal processing related to the calculation of the reception weight.

  As for signal reception, as in the case of transmission, in the wideband system using the OFDM modulation scheme or SC-FDE scheme, the above-described reception weight multiplication is performed for each frequency component. That is, the signals output from the A / D converters 856-1 to 856-K are subjected to FFT in the FFT circuits 857-1 to 857-K and separated into frequency components, and channel information is obtained for each separated frequency component. The signal processing in the estimation circuit 861 and the reception signal processing in the reception signal processing circuits 858-1 to 858 -L are performed.

(Multi-user MIMO transmission processing)
FIG. 11 is a flowchart showing a transmission process of the base station apparatus 80 in multiuser MIMO. In multi-user MIMO, feedback of downlink channel information, which is performed separately from data transmission, is periodically performed. When the channel information acquisition circuit 831 acquires channel information in the downlink (step S831), the channel information storage circuit 832 stores the channel information of each frequency component for each terminal device (step S832). The processes in step S831 and step S832 are performed sequentially.

When signal transmission processing from the base station apparatus 80 is started (step S821), the multiuser MIMO transmission weight calculation circuit 833 obtains channel information of each frequency component corresponding to the terminal apparatus that is the destination from the channel information storage circuit 832. Read (step S822).
Based on the read channel information, the multiuser MIMO transmission weight calculation circuit 833 calculates a transmission weight for multiuser MIMO for each frequency component by the processing described above (step S823). Separately from the processing of step S822 and step S823, the transmission signal processing circuits 811-1 to 811-L perform each frequency for each destination station by performing transmission signal processing such as various modulation processing on the data to be transmitted for each destination. A component transmission signal is generated (step S824).

  The transmission signal processing circuits 811-1 to 811-L multiply the generated transmission signal by the transmission weight calculated by the multiuser MIMO transmission weight calculation circuit 833 in step S823 (step S825). The transmission signal processing circuits 811-1 to 811-L perform a series of signal processing, and the adder / synthesizing circuits 812-1 to 812-L each terminal device of each frequency component for each of the antenna elements 819-1 to 819-L. Addition synthesis is performed on the transmission signal addressed to the signal, and the IFFT & GI adding circuits 83-1 to 813-K convert the signal on the frequency axis to the signal on the time axis, and further insert a guard interval or between OFDM symbols (SC- If it is FDE, it performs processing such as waveform shaping between the blocks in the block transmission) and outputs it to the D / A converters 814-1 to 814-K (steps S826-1 to S826-K). The signals output from the IFFT & GI adding circuits 813-1 to 813 -K are subjected to signal processing in the high power amplifiers 818-1 to 818 -K from the D / A converters 814-1 to 814 -K, and the antenna element 819. -1 to 819-K (steps S827-1 to S827-K), and the process ends (steps S828-1 to S828-K).

  Note that the processing in steps S827-1 to S827-K includes up-conversion processing from a baseband signal to a radio frequency, removal of frequency components of a band by a filter, signal amplification by a high power amplifier, and the like.

(Multi-user MIMO reception processing)
FIG. 12 is a flowchart showing reception processing of the base station apparatus 80 in multiuser MIMO. First, when reception processing is started (step S840), signals are received by the first to Kth antenna elements 851-1 to 851-K (steps S841-1 to S841-K). Here, reception includes processing for performing analog / digital conversion on a received signal or a signal obtained by down-converting the received signal. Subsequent signal processing means processing on a digitized received signal.
Subsequently, the received signals corresponding to the antenna elements 851-1 to 851-K are subjected to signal processing such as separation into frequency components by the FFT circuits 857-1 to 857-K (steps S841-1 to S842). -K). Furthermore, the channel information estimation circuit 861 performs channel estimation of each frequency component based on the reception state of the preamble signal having a known pattern attached to the wireless packet (steps S843-1 to S843-K). Here, the attenuation of the signal on the propagation path and the rotation state of the complex phase are grasped. In the channel estimation performed in steps S843-1 to S843-K, channel estimation is performed individually for each spatially multiplexed signal sequence, as shown in steps S843-1, S843-2,. Need to do. This individual channel estimation needs to be performed in a state in which the signals transmitted from the respective transmission source terminal devices can be separated. Taking the OFDM modulation method as an example, generally, a preamble signal for channel estimation having the same number of symbols as the number of spatial multiplexing is required. Each terminal apparatus performs signal transmission with the same number of symbols as the number of spatial multiplexing (or more) and assigns a different pattern of preamble signals, and the base station apparatus 80 uses the difference in pattern to perform step transmission. Individual channel estimation is performed in S843-1 to S843-K.

The multiuser MIMO reception weight calculation circuit 862 uses the channel information estimated by the channel information estimation circuit 861 to calculate individual appropriate reception weights for each spatially multiplexed signal sequence and each frequency component (step S844). Further, the reception signal processing circuits 858-1 to 858 -L multiply the reception signal calculated for each signal series and each frequency component by the reception signal of each antenna element separated for each frequency component (step S <b> 845-). 1-S845-K).
Here, since reception weights are prepared for each spatially multiplexed signal sequence, the multiplication results in steps S845-1 to 845-L are different results for each spatially multiplexed signal sequence. The signals of the respective signal sequences are obtained by adding and synthesizing the signals of the antenna elements 851-1 to 851-K for each frequency component (steps S846-1 to S846-L). Processing from signal sequence signal processing (step S847-1) to signal processing of the Lth signal sequence (step S847-L) is performed, and the processing ends (steps S848-1 to S848-L).

  Here, for the sake of simplicity, an example in which a linear reception weight is used is shown, but in general, nonlinear signal processing such as MLD (Maximum Likelihood Detection) may be performed for MIMO. In this case, the processes in steps S845-1 to S845-L, steps S846-1 to S846-L, and steps S847-1 to S847-L are integrally performed with nonlinear signal detection processing. Further, the linear reception weight can be calculated by a method similar to the transmission weight calculation process shown in FIG. In addition, it is also possible to use a reception weight using a pseudo inverse matrix or an MMSE weight. Here, the number of spatially multiplexed signal sequences is described as L for the number K of antenna elements 851-1 to 851-K used for reception, but in general, K and L need to match. If the value of L is equal to or less than the value of K, a number of signal series signals can be spatially multiplexed.

  As described above, a typical feature of multi-user MIMO is that, in the reception processing in the base station apparatus 80 in the uplink, the reception weight is set for each reception based on the channel information between the transmission side and the reception side. And the latest channel information in the downlink transmission process is read (step S822), and the transmission weight is calculated based on the read channel information (step S823). In other words, the transmission weight and the reception weight are calculated every time transmission or reception is performed. This is due to channel time variation, and in order to obtain good channel estimation accuracy, it is necessary to periodically perform feedback processing of channel information. As the channel feedback period is set shorter, transmission / reception of control information for channel feedback becomes necessary and the overhead increases. Furthermore, when the base station apparatus 80 receives a spatially multiplexed signal, it is necessary to individually perform channel estimation for a plurality of terminal apparatuses, and thus a desired number of orthogonal preambles are required. In general, it is preferable that the preamble signal patterns themselves are orthogonal, but if such a pattern cannot be set, overhead of the same number of symbols as the number of spatial multiplexing is required, and the number of spatial multiplexing increases. Accordingly, the overhead also increases.

[Requirements for actual system]
In the coherent transmission and distributed antenna system described above, the channel information needs to be known on the transmission side. Therefore, in an actual system, it is necessary to clear the following requirements.

(Requirement 1)
For example, consider a case where a line gain of 20 [dB] is gained using a radio module of 100 stations. In communication, since a circuit such as a wireless communication device is designed on the assumption that the line gain is improved by 20 [dB], when performing channel estimation between one wireless module and a terminal device, 20 [ dB] Channel estimation must be performed in a degraded environment. For example, if the required SNR in actual communication is 10 [dB], channel estimation must be performed in a noise-dominated environment with an SNR of −10 [dB]. However, in such an environment where noise is dominant, even if the transmission weight is obtained from the estimated extremely uncertain channel information, in-phase synthesis cannot be realized.

  Note that the distributed antenna system is assumed to be a terminal device that exists in a region where a plurality of cells overlap as shown in FIG. That is, only a few remote base stations that are involved in transmission / reception in a distributed antenna system are geographically relatively close to the terminal device. As a result, the SNR does not become low, and the above-mentioned channel estimation accuracy problem occurs in the first place. It wasn't. Further, in Non-Patent Document 1 in which coherent transmission using a plurality of relay stations is described, as described in the chapter “Summary”, various control achievement methods including channel information estimation methods are described. In this document, it is clearly stated that “don't mention it”. In other words, the author recognizes that it is difficult to realize coherent transmission at present, and Non-Patent Document 1 makes a claim that there is a possibility that a beneficial effect can be obtained if these various problems can be solved. It is inferred that Thus, in the prior art, a method for performing feedback of channel information in an ultra-low SNR region necessary for coherent transmission has not been established. Therefore, it is required that these technologies be established in an actual system.

(Requirement 2)
Assuming an environment where there is no constant traffic such as in urban areas, the channel conditions change over time. Even if the channel estimation accuracy is at a desired level and channel feedback is possible, the channel feedback period should be set relatively long, considering the decrease in transmission efficiency due to the overhead required for channel feedback. As a result, a transmission / reception weight based on channel information past the actual transmission / reception time is used. However, since the optimum transmission / reception weight changes due to channel fluctuations, the expected line gain may not be obtained, and communication may become unstable. Therefore, in an actual system, establishment of a countermeasure technique for this channel time variation is required.

  As described above, in order to perform coherent transmission via a plurality of wireless modules or a plurality of antenna elements, the above-mentioned “measures for reducing accuracy of channel information in an environment where reception power is low” (requirement condition 1 ), “Technique for countermeasures against communication destabilization due to channel fluctuation” (Requirement 2) so that signals are synthesized in the same phase in the terminal device as the receiving side Therefore, a new technique for adjusting a signal transmitted from each wireless module or each antenna element is required. Similar to the transmission side, there are exactly the same requirements in the received signal processing for signals received by each wireless module or each antenna element.

(Requirement 3)
Even if the above requirements can be cleared, if a high line gain such as 20 [dB] can be obtained, a very wide area can be collectively set as a service area. Therefore, frequency resources must be shared by many terminal devices located in a wide area. As the area becomes wider and the number of terminal devices sharing frequency resources increases, the throughput per terminal device is consequently reduced. In order to increase the throughput per terminal device to a predetermined value or more, it is necessary to increase the throughput of the entire system. However, since frequency resources are limited, the frequency band used for communication cannot be expanded. In other words, it is necessary to improve the throughput per terminal device by increasing the frequency utilization efficiency. In other words, it is necessary to establish a technology for increasing the capacity of the system in such an environment.

  Further, when a very large number of signals are spatially multiplexed by multi-user MIMO transmission, it is necessary to separate the spatially multiplexed signals at least in the uplink and then perform channel estimation of individual paths on the receiving side. In order to perform such channel estimation, at least spatially multiplexed number of orthogonal preamble signals are required. In general, it is preferable that the preamble signal patterns themselves are orthogonal, but if such a pattern cannot be set, an overhead of the same number of symbols as the number of spatial multiplexing is required. This lowers the efficiency of the MAC layer and lowers the frequency utilization efficiency. That is, the conventional countermeasure technique for (Requirement 3) creates a new problem. Therefore, in an actual system, it is possible to suppress a decrease in the efficiency of the MAC layer due to a realistic calculation load and overhead including channel feedback, a preamble for channel estimation, etc. There is a need to effectively implement spatial multiplexing.

Yusuke Hara et al., "Reduction of power consumption by coherent transmission", IEICE Society Conference BS-3-1, September 2009 Daiki Matsuda et al., “A Study on Channel Capacity of Distributed Antenna System Using Maximum Ratio Transmission”, IEICE RCS2007-196, pp. 61-66, February 2008 Takehiko Tsukiji, “Introduction to Radio and Antenna Engineering”, General Electronic Publishing Company, pp. 166-168, March 2002 Taiji Takatori et al., "Application of downlink multi-user MIMO technology to next-generation high-speed wireless access systems" IEICE Transactions B, Communication J93-B (9), pp 1127-1139, September 2010

As described above, the multi-user MIMO technique is effective for (Requirement 3). However, in order to significantly increase the throughput, it is necessary to increase the number of spatial multiplexing. A new condition arises.
For example, in multi-user MIMO transmission using an extremely large number (for example, 100) of antenna elements, a matrix of “total number of transmission antenna elements” × “total number of reception antenna elements” is used in calculation of transmission weights and reception weights. In other words, the influence on the calculation of the transmission weight and the reception weight required for each data transmission / reception increases as the matrix size increases. In general, the amount of calculation processing such as inverse matrix calculation and singular value decomposition (specifically, because the circuit scale of a multiplication circuit is larger than that of an addition circuit when it is configured as a circuit, evaluation is based on the number of multiplications or divisions. Is said to increase in proportion to the cube of the matrix size. Except for desk studies and some experimental studies, the current technology level is considered to have an upper limit of about 4 multiplexing for multiuser MIMO, and the number of antenna elements in that case is also 4-6. It is about a book. When the number of antenna elements that is one digit or more larger than the number of antenna elements used in this generally assumed multi-user MIMO is used, the required amount of calculation becomes 1000 times or more. Further, in a general mobile environment, channel time fluctuation cannot be ignored, and it is necessary to calculate a transmission weight or a reception weight every time data is transmitted / received. That is, there is a problem in that it takes a long time to calculate the transmission weight and the reception weight, and it is difficult to efficiently perform spatial multiplexing.

  The present invention has been made in view of such a situation, and improves frequency use efficiency while performing stable communication when performing high-order spatial multiplexing using a large number of antenna elements in a base station apparatus. The present invention provides a base station apparatus, a wireless communication method, and a wireless communication system.

  In order to solve the above problem, the present invention comprises a base station apparatus having a plurality of antenna elements and a plurality of terminal apparatuses that perform radio communication with the base station apparatus, and the base station apparatus and at least two of the above-mentioned base station apparatuses A base station apparatus in a radio communication system capable of performing spatial multiplexing transmission at the same time on the same frequency component with a terminal apparatus, wherein the terminal apparatus performs spatial multiplexing transmission at the same time on the same frequency component Channel information of each frequency component in the uplink between the terminal device and each of the plurality of antenna elements based on a training signal received from the terminal device for each of the terminal devices An uplink channel information acquisition unit to acquire, and each frequency component for each terminal device acquired by the uplink channel information acquisition unit. A reception weight calculation unit that calculates reception weights for spatial multiplexing transmission at each frequency component for each of the plurality of antenna elements based on channel information in the uplink or a physical quantity calculated from the channel information, and the reception Reception signal correction based on the reception weight storage unit that stores the reception weight calculated by the weight calculation unit, the reception weight calculated by the reception weight calculation unit, and the channel information acquired by the uplink channel information acquisition unit A reception signal correction information calculation unit that calculates information; a reception signal correction information storage unit that stores reception signal correction information calculated by the reception signal correction information calculation unit; and for each antenna element, the antenna element The received signal received from the terminal device is separated for each frequency component, and the separated frequency component Is multiplied by the reception weight of each frequency component corresponding to the terminal device read from the reception weight storage unit, and the multiplied signals are added and synthesized across all or some of the antenna elements. A received signal processing unit that acquires the obtained summed composite signal for each frequency component, and an added composite signal acquired by the received signal processing unit corresponding to the terminal device that performs spatial multiplexing transmission on the same frequency component at the same time A reception signal correction unit that performs correction based on the reception signal correction information stored in the reception signal correction information storage unit, and the reception signal processing unit uses the signal corrected by the reception signal correction unit. A base station apparatus that detects a transmitted signal.

第ｋ次（０≦ｋ，ｋは整数）と第（ｋ＋１）次のベクトルの間の漸化式は次式（Ｂ）で与えられ、

式（Ｂ）の漸化式により目標とする次数α次（１≦α、αは整数）までのベクトルを算出した後、ベクトルδＲｘ^（0）からベクトルδＲｘ^（α）までのベクトル和を算出し、算出されたベクトルを前記補正した周波数成分ごとの信号として前記受信信号処理部に出力することを特徴とする。 In the invention described above, the i-th terminal device (1 ≦ i ≦ N, i) is connected to the terminal device of N (N is a positive integer) station that performs spatial multiplexing simultaneously at a predetermined frequency component at a certain time. Is a channel information vector in the uplink configured by channel information in the predetermined frequency component between the antenna element and the plurality of antenna elements included in the device, and h _Ui is a reception weight vector corresponding to the channel information vector W _Ui in the received signal correction unit, the received signal correction information is a matrix W _Ud whose component is the reciprocal of a product w _i h _i of two vectors corresponding to the i th terminal device. ⁻¹ , j (1 ≦ j ≦ N, i ≠ j, j is an integer) and the channel information vector and the reception weight vector corresponding to the i-th terminal device. The value given by the calculation -w _Ui h _Uj / w _Ui h _Ui is the (i, j) component and the diagonal component (i = j) is a matrix ΔW _U given by all zeros. Assuming that a signal vector whose component is an additive composite signal acquired by the received signal processing unit corresponding to the terminal device that simultaneously performs spatial multiplexing in frequency components is Rx, the 0th-order initial vector δRx ⁽⁰⁾ is Given in (A),

The recurrence formula between the kth order (0 ≦ k, k is an integer) and the (k + 1) th order vector is given by the following formula (B):

After calculating the vector up to the target order α-order (1 ≦ α, α is an integer) by the recurrence formula of Formula (B), the vector sum from vector δRx ^{( 0)} to vector δRx ^(α) is calculated. The calculated vector is output to the received signal processing unit as a signal for each corrected frequency component.

第ｋ次（０≦ｋ，ｋは整数）と第（ｋ＋１）次のベクトルの間の漸化式は次式（Ｄ）で与えられ、

式（Ｄ）の漸化式により目標とする次数α次（１≦α、αは整数）までのベクトルを算出した後、ベクトルδＲｘ^（0）からベクトルδＲｘ^（α）までのベクトル和を算出し、算出されたベクトルを前記補正した周波数成分ごとの信号として前記受信信号処理部に出力し、前記第ｉ端末装置に関して前記受信ウエイト算出部にて算出されるある周波数成分の受信ウエイトｗ_Ｕｉは次式（Ｅ）で与えられる

ことを特徴とする。 In the invention described above, the i-th terminal device (1 ≦ i ≦ N, i) is connected to the terminal device of N (N is a positive integer) station that performs spatial multiplexing simultaneously at a predetermined frequency component at a certain time. Is a channel information vector in the uplink configured by channel information in the predetermined frequency component between the antenna element and the plurality of antenna elements included in the device, and h _Ui is a reception weight vector corresponding to the channel information vector the upon a w _Ui, in the received signal correction unit, and the reception signal correction information, the j (1 ≦ j ≦ N, i ≠ j, j is an integer) corresponding to the terminal device and the i in all the channel information vector and the value given by the vector operation of receiving weight vector -w _Ui h _Uj a second (i, j) component and to and diagonal components (i = j) is zero A Erareru matrix [Delta] W _U, placing the signal vector to the additive synthesis signals the reception signal processing unit corresponding to the terminal device that performs spatial multiplexing simultaneously at a predetermined frequency component at a certain time to get the ingredients and Rx, The zeroth-order initial vector δRx ⁽⁰⁾ is given by the following equation (C):

The recurrence formula between the k th order (0 ≦ k, k is an integer) and the (k + 1) th order vector is given by the following formula (D):

After calculating the vector up to the target order α-order (1 ≦ α, α is an integer) by the recurrence formula of Formula (D), the vector sum from vector δRx ^{( 0)} to vector δRx ^(α) is calculated. The calculated vector is output to the reception signal processing unit as a signal for each corrected frequency component, and the reception weight w _Ui of a certain frequency component calculated by the reception weight calculation unit with respect to the i-th terminal device is Given by equation (E)

It is characterized by that.

  In the above-described invention, for each antenna element, a calibration coefficient used for calculating downlink channel information from uplink channel information between the antenna element and the terminal device is calculated for each frequency component. And each of the channel information in the uplink of each frequency component for each of the antenna elements calculated by the uplink channel information acquisition unit by the uplink channel information acquisition unit. A downlink channel information calculation unit that calculates the channel information in the downlink by multiplying the calibration coefficient corresponding to the combination of, and each of the antenna elements in the downlink calculated by the downlink channel information calculation unit. A transmission weight calculation unit for calculating a transmission weight for spatial multiplexing transmission of each frequency component for each of the antenna elements, and a transmission weight for storing the reception weight calculated by the transmission weight calculation unit A transmission signal correction information calculation unit that calculates transmission signal correction information based on the storage unit, the transmission weight calculated by the transmission weight calculation unit, and the channel information calculated by the downlink channel information calculation unit; A transmission signal correction information storage unit for storing transmission signal correction information calculated by the transmission signal correction information calculation unit, and a transmission signal to be transmitted to the terminal device for each of the terminal devices selected as data transmission destinations. Performs spatial multiplexing transmission at the same time on the same frequency component with the transmission signal processing unit that separates the signal for each frequency component A transmission signal correction unit that corrects a signal for each frequency component separated by the transmission signal processing unit corresponding to the terminal device based on transmission signal correction information stored in the transmission signal correction information storage unit, The transmission signal processing unit reads the transmission weight of each frequency component corresponding to the terminal device that is a data transmission destination from the transmission weight storage unit, and the transmission signal correction unit corrects the transmission weight of the frequency component to be read. The signal for each frequency component is multiplied and transmitted from the plurality of antenna elements.

第ｋ次（０≦ｋ，ｋは整数）と第（ｋ＋１）次のベクトルの間の漸化式は次式（Ｇ）で与えられ、

式（Ｇ）の漸化式により目標とする次数α次（１≦α、αは整数）までのベクトルを算出した後、ベクトルδＴｘ^（0）からベクトルδＴｘ^（α）までのベクトル和を算出し、算出されたベクトルを前記補正した周波数成分ごとの信号として前記送信信号処理部に出力することを特徴とする。 In the invention described above, the i-th terminal device (1 ≦ i ≦ N, i) is connected to the terminal device of N (N is a positive integer) station that performs spatial multiplexing simultaneously at a predetermined frequency component at a certain time. Is a channel information vector in the downlink composed of channel information in the predetermined frequency component between the antenna element and the plurality of antenna elements included in the base station apparatus, h _Di , and a transmission weight corresponding to the channel information vector When the vector is _wDi , in the transmission signal correction unit, the transmission signal correction information is the jth corresponding to the jth terminal device (1 ≦ j ≦ N, i ≠ j, j is an integer). And a value −h _Di w _Dj / h _Di w _Di given by the vector operation of the channel information vector and the transmission weight vector corresponding to the i th terminal device ( i, j) component and to and diagonal components (i = j) is a matrix [Delta] W _D all given by zero, the transmission signal processing corresponding to the terminal device that performs spatial multiplexing simultaneously at a predetermined frequency component at a certain time If a signal vector whose component is a signal for each frequency component separated by the part is denoted by Tx, the 0th-order initial vector δTx ⁽⁰⁾ is given by the following equation (F):

The recurrence formula between the k th order (0 ≦ k, k is an integer) and the (k + 1) th order vector is given by the following formula (G):

After calculating the vector up to the target order α order (1 ≦ α, α is an integer) by the recurrence formula of Expression (G), the vector sum from the vector δTx ^{( 0)} to the vector δTx ^(α) is calculated. The calculated vector is output to the transmission signal processing unit as a signal for each corrected frequency component.

  In addition, the present invention includes a base station device including a plurality of antenna elements and a plurality of terminal devices that perform radio communication with the base station device, and the base station device and at least two of the terminal devices have the same frequency. Scheduling processing step for selecting a combination of terminal devices that perform spatial multiplexing transmission at the same time on the same frequency component, in a wireless communication method capable of performing spatial multiplexing transmission at the same time on components And, for each of the terminal devices, uplink channel information for acquiring channel information of each frequency component in the uplink between the terminal device and each of the plurality of antenna elements based on a training signal received from the terminal device Each of the terminal devices acquired in the acquiring step and the uplink channel information acquiring step. A reception weight calculating step for calculating a reception weight for spatial multiplexing transmission in each frequency component for each of the plurality of antenna elements based on channel information in the uplink of several components or a physical quantity calculated from the channel information; A reception weight storage step of storing the reception weight calculated in the reception weight calculation step in a reception weight storage unit; the reception weight calculated in the reception weight calculation step; and channel information acquired in the uplink channel information acquisition step; A received signal correction information calculating step for calculating received signal correction information, and a received signal correction information storage step for storing the received signal correction information calculated in the received signal correction information calculating step in a received signal correction information storage unit; ,Previous For each antenna element, the received signal received from the terminal device via the antenna element is separated for each frequency component, and the received signal for each separated frequency component is stored in the received weight storage unit A reception signal processing step of multiplying the reception weights of the corresponding frequency components, and obtaining, for each frequency component, an added synthesized signal obtained by adding and synthesizing the multiplied signals over all or some of the antenna elements. And the added composite signal acquired in the received signal processing step corresponding to the terminal device performing spatial multiplexing transmission at the same time on the same frequency component, based on the received signal correction information stored in the received signal correction information storage unit A received signal correcting step for correcting the received signal and a signal transmitted by the terminal device from the signal corrected in the received signal correcting step. And a signal detection step for outputting the wireless communication method.

  In addition, the present invention includes a base station device including a plurality of antenna elements and a plurality of terminal devices that perform radio communication with the base station device, and the base station device and at least two of the terminal devices have the same frequency. A radio communication system capable of performing spatial multiplexing transmission at the same time on a component, wherein the base station apparatus selects a combination of the terminal apparatuses that perform spatial multiplexing transmission at the same time on the same frequency component An uplink that acquires channel information of each frequency component in the uplink between the terminal device and each of the plurality of antenna elements based on a training signal received from the terminal device for each processing unit and the terminal device A channel information acquisition unit and an uplink of each frequency component for each terminal device acquired by the uplink channel information acquisition unit. A reception weight calculation unit that calculates reception weights for spatial multiplexing transmission at each frequency component for each of the plurality of antenna elements based on channel information or a physical quantity calculated from the channel information; and the reception weight calculation unit The reception signal correction information is calculated based on the reception weight storage unit that stores the reception weight calculated by the reception weight, the reception weight calculated by the reception weight calculation unit, and the channel information acquired by the uplink channel information acquisition unit. A reception signal correction information calculation unit, a reception signal correction information storage unit that stores the reception signal correction information calculated by the reception signal correction information calculation unit, and the antenna element from the terminal device via the antenna element. The received signal is separated for each frequency component, and the received signal is separated for each frequency component. Multiplication signal obtained by multiplying the reception weight of each frequency component corresponding to the terminal device read from the signal weight storage unit, and adding and synthesizing the multiplied signal over all or part of the antenna elements Received signal correction information obtained by the received signal processing unit that acquires the received signal processing unit for each frequency component and the received signal processing unit corresponding to the terminal device that performs spatial multiplexing transmission on the same frequency component at the same time. A reception signal correction unit that corrects based on the reception signal correction information stored in the storage unit, and the reception signal processing unit detects a signal transmitted by the terminal device from a signal corrected by the reception signal correction unit This is a wireless communication system.

According to the present invention, prior to actual data communication, the base station apparatus obtains channel information by receiving a predetermined training signal from the terminal apparatus and calculates the reception weight in advance. Reception processing is performed using the reception weight.
Furthermore, while using this reception weight, for each signal sequence spatially multiplexed on the received signal, reception weights and channel information for two terminal devices included in the overall combination of terminal devices that simultaneously perform spatial multiplexing, By performing the correction using the received signal correction information based on the combination, it is possible to suppress the interference component between the signal sequences and perform the signal detection process with higher accuracy.
In addition, when the reception weight used at this time is a weight that is calculated individually for each terminal device that does not depend on the combination of terminal devices, in comparison with the case where the reception weight that depends on the combination of terminal devices is used, The amount of information to be calculated can be suppressed to greatly reduce the amount of calculation, and the amount of information to store the calculation result can be suppressed to an overwhelmingly small amount. Furthermore, in the scheduling process for selecting terminal devices to be spatially multiplexed at the same time, it is possible to remove restrictions on the combination and deal with any combination of terminal devices.
In addition, the reception weight used at this time is a weight specific to the terminal device that does not depend on the combination of the terminal devices, and is calculated in advance as compared with the case of using the reception weight that depends on the combination of the terminal devices, and the result is The amount of information to be stored can be suppressed to an overwhelmingly small amount. Furthermore, it is possible to cope with any combination of terminal devices in a scheduling process for selecting terminal devices to be spatially multiplexed simultaneously.
As a result, good SIR characteristics can be achieved even if the number of signal systems performing spatial multiplexing simultaneously increases, and the degree of freedom in scheduling can be increased, and frequency utilization efficiency can be improved while performing stable communication.

It is a schematic block diagram which shows the structure of the base station apparatus 200 in this embodiment. It is a figure which shows the outline | summary of the radio relay system in a prior art. It is a figure which shows the outline | summary of the distributed antenna system in a prior art. It is a flowchart which shows the process of the channel feedback in a prior art. It is a figure which shows the principle of a phased array antenna. It is the schematic which shows the structural example of a multiuser MIMO system. It is a flowchart which shows the procedure which calculates the transmission weight matrix W in a multiuser MIMO system. It is a schematic block diagram which shows an example of a structure of the base station apparatus 80 in a multiuser MIMO system. It is a schematic block diagram which shows an example of a structure of the transmission part 81 in the base station apparatus 80 in a multiuser MIMO system. It is a schematic block diagram which shows an example of a structure of the receiving part 85 in the base station apparatus 80 in a multiuser MIMO system. It is a flowchart which shows the transmission process of the base station apparatus 80 in multiuser MIMO. It is a flowchart which shows the reception process of the base station apparatus 80 in multiuser MIMO. It is a figure which shows the example of installation of the base station apparatus with which the radio | wireless communications system which concerns on this invention comprises. It is a figure which shows the operation example of the signal synthesis | combination which the base station apparatus which concerns on this invention performs. It is a figure which shows the example of the training signal in this invention. It is a figure which shows the asymmetry of the channel information of an uplink and a downlink. It is a figure which shows the outline | summary of calibration. It is a schematic block diagram which shows the structure of the base station apparatus 10 in the 1st structural example of related technology. It is a figure which shows an example of a structure of the receiving part 100 with which the base station apparatus 10 in the same structural example is provided. It is a schematic block diagram which shows the structural example of the transmission / reception weight calculation part 120a in the same structural example. It is a figure which shows an example of a structure of the transmission part 140 in the base station apparatus 10 in the same structural example. It is a flowchart which shows the short time averaging process which acquires the channel information of the uplink in the same structural example. It is a flowchart which shows the relative component acquisition process which acquires the relative component of the uplink channel information in the example of a structure. It is a flowchart which shows the long-time averaging process of the uplink channel information in the same configuration example. It is a flowchart which shows the process which acquires the channel information of the downlink in the same structural example. It is a flowchart which shows the process which calculates the transmission weight and the reception weight in the base station apparatus 10 of the same structural example. It is a flowchart which shows the transmission process of the base station apparatus 10 in the same structural example. It is a flowchart which shows the reception process of the base station apparatus 10 in the same structural example. It is a schematic block diagram which shows the structural example of the transmission / reception weight calculation part 120b in the 2nd structural example of related technology. It is a flowchart which shows the process which calculates the transmission weight and the reception weight in the base station apparatus 10 of the same structural example. It is a flowchart which shows the transmission process of the base station apparatus 10 in the same structural example. It is a flowchart which shows the reception process of the base station apparatus 10 in the same structural example. It is a flowchart which shows the other relative component acquisition process which acquires the relative component of the channel information of uplink in each structural example. It is a figure which shows the structure of the received signal processing circuits 109-1-109-L in the 2nd structural example of this invention related technology. It is a schematic block diagram which shows the structure of the received signal processing circuit 209-1 to 209-L and the received signal correction | amendment part 230 in this invention embodiment. It is a figure which shows the structure of the transmission signal processing circuits 140-1 to 140-L in the 2nd structural example of this invention related technology. It is a schematic block diagram which shows the structure of the transmission signal processing circuits 241-1 to 241-L and the transmission signal correction | amendment part 250 in this embodiment. It is a schematic block diagram which shows the structure of the transmission / reception weight calculation part 290 in embodiment of this invention. It is a figure showing the flow of the correction process of the received signal in this invention embodiment. It is a figure showing the flow of the correction process of the transmission signal in this invention embodiment. It is a figure showing the processing flow which concerns on the received signal correction information of the transmission / reception weight calculation part 290 in this invention embodiment. It is a figure showing the processing flow which concerns on the transmission signal correction information of the transmission / reception weight calculation part 290 in this invention embodiment.

[Operational principle of the present invention]
One of the essences of the present invention is that the base station apparatus calculates an estimated value of channel information indicating channel characteristics between a large number of wireless modules provided in the base station apparatus and the terminal apparatus over a long period of time. By using transmission weights and reception weights that are measured and calculated based on the average value of the estimated channel information, it is possible to reduce the effects of channel time fluctuations using multiple wireless modules while coherent using in-phase synthesis. The purpose is to acquire line gain accompanying transmission and to realize high-order spatial multiplexing using a low line gain outside the region where the same phase synthesis is performed at a pinpoint. As for the realization of higher-order spatial multiplexing, more effective operation becomes possible by using in combination with a scheduling technique for managing SIR characteristics, a directivity control technique for improving SIR characteristics, and the like.

(Prerequisite)
In the present invention, it is not always necessary to ensure the visibility of each wireless module and the terminal device, but it is recommended that the wireless module and the terminal device be fixed at a relatively high place. In this case, the transmission path (channel) between each wireless module and the terminal device is a “direct line-of-sight wave”, a “stable reflected wave” by a fixed huge building, etc. ) It can be regarded as a mixture of “multiple reflected waves from objects with movement” such as nearby cars and people. In this case, the “direct line-of-sight wave” and the “stable reflected wave” have a relatively high reception level and smaller time fluctuation than the “multiple reflected wave from the moving object”. On the other hand, “multiple reflected waves from a moving object” has a lower reception level and a large fluctuation with time compared to “direct line-of-sight waves” and “stable reflected waves”.

  A signal for channel estimation (hereinafter referred to as “training signal”) is transmitted continuously or intermittently over a long period of time, and the received signal is averaged over a long period of time. Because of the randomness of the “multiple reflected wave from the object with movement” signal, the average value of the variation of the complex phase and amplitude approaches zero. On the other hand, components related to “direct line-of-sight wave” and “stable reflected wave” converge to a non-zero constant value. As a result, a channel estimation result corresponding to a stable path with a relatively small time variation component is extracted.

In the description of coherent transmission in the prior art, the “wireless module” is “relay station” or “remote base station” in the distributed antenna system. Of course, these are physically located away from the control station or base station in the prior art. Taking a distributed antenna system as an example, a remote base station is located at the center of a plurality of cells, and if it is a relay station using radio, it is far enough to use radio Become. However, in the same phase synthesis (for both transmission and reception) of signals from individual radio modules intended in the present invention, it is not always necessary to separate the radio modules as far as remote base stations and relay stations. Absent.
In addition, if the distance between the antenna elements of each wireless module is smaller than the wavelength of the carrier frequency of communication, the assumed in-phase synthesis of signals may be disturbed due to mutual coupling between the antenna elements. However, this problem can be avoided if an interval of approximately one wavelength or more is ensured between the antenna elements.

That is, the basic configuration of the present invention is that a large number of antenna elements each having an interval of one wavelength or more are secured to one base station apparatus. Of course, since the signals transmitted and received from each antenna element have different transmission and reception weight coefficients, each antenna element has a separate RF (Radio Frequency) circuit in the radio frequency band such as a high power amplifier, low noise amplifier, and filter. And are connected to each other to constitute one wireless module.
In the description so far, each wireless module is physically arranged discretely at a location different from the control station or the like, so the term “wireless module” is intended to be a wireless module that is almost integrated with the antenna element. Although various explanations have been given, in the present invention, a control station and a large number of wireless modules are integrated in one place, and generally, a form of one base station apparatus is natural. The expression “wireless module” may not be appropriate.

  For example, functionally, a function corresponding to a control station such as baseband signal processing and a function of an RF circuit in a radio frequency band such as a plurality of high power amplifiers, low noise amplifiers, filters, etc. are mounted in one casing, If a configuration is assumed in which a casing and a large number of antenna elements are connected by a coaxial cable, it is necessary to consider correction for fluctuations in amplitude / complex phase in amplifiers and filter systems during transmission and reception. In many cases, the expression “channel information between the wireless module and the wireless module” is substantially expressed as “channel information between the antenna element of the terminal device and the antenna element of the wireless module”. Therefore, hereinafter, in the description of the channel, the term “antenna element” will be used instead of the term “wireless module”.

(Examples of wireless communication system installation and basic principles)
FIG. 13 is a diagram illustrating an installation example of the base station apparatus included in the wireless communication system according to the present invention. In the same figure, the code | symbol 11 shows the building in which the base station apparatus is installed, the code | symbols 12-1 to 12-2 shows a terminal device, and the code | symbols 13-1 to 13-4 are equipped with the base station apparatus. An antenna element is shown, the code | symbol 14-1 to 14-3 shows the mobile body on the ground, and the code | symbols 15-1 to 15-2 have shown the large sized building (naturally a stationary state).

  Here, the antenna elements 13-1 to 13-4 included in the base station apparatus are installed in a very high place such as the roof of the building 11. The terminal devices 12-1 to 12-2 may be relatively lower than the antenna elements 13-1 to 13-4 of the base station device such as a telephone pole or a general building rooftop. However, it is installed at a relatively high place. On the other hand, the mobile units 14-1 to 14-3 on the ground are located in places relatively lower than the antenna elements 13-1 to 13-4 and the terminal devices 12-1 to 12-2 of the base station apparatus. In addition to cars, there are many starting points (reflective points) of reflected waves that fluctuate randomly, such as people and trees swaying in the wind.

  For example, the terminal device 12-1 and the antenna elements 13-1 to 13-4 of the base station device are in a line-of-sight environment (direct waves are indicated by thick solid arrows in the figure). On the other hand, although the terminal device 12-2 and the antenna elements 13-1 to 13-4 of the base station device are not in the line-of-sight environment due to the shielding of the large building 15-2, the large building 15-1 or the like And a stable reflected wave (indicated by a thick solid arrow in the figure) has reached.

In addition, for the terminal device 12-1 in the line-of-sight environment, there may be a situation where a stable reflected wave due to a large building exists in addition to the line-of-sight wave, and these signals are always combined to reach the signal. A signal represented by such a thick solid arrow is regarded as a stable incident wave. On the other hand, the reflected waves from the mobile bodies 14-1 to 14-3 on the ground have many signals that arrive as many random multiple reflections, and the level of the signal received is relatively low, and the complex phase component And the amplitude varies randomly with time.
When a number of weak and random waves are combined, the resulting signal has a relatively low signal strength relative to a stable incident wave. Therefore, the “time-varying incident wave” obtained by combining the “stable incident wave” with the “random multiple reflected wave” is a signal in which a minute error is added around the “stable incident wave”. Can see.

Next, signal synthesis performed by the base station apparatus in such a situation will be described.
FIG. 14 is a diagram showing an example of signal combining operation performed by the base station apparatus according to the present invention. Here, as an example, when signals transmitted from terminal apparatus 12-1 in FIG. 13 are received by antenna elements 13-1 to 13-4 of the base station apparatus, they are combined using appropriate reception weights. Shows the case.
The antenna elements 13-1 to 13-4 of the base station apparatus receive the “incident wave that varies with time”. The reception weight used when combining these signals is determined so that the signals at the respective antenna elements are combined in phase with the “stable incident wave” as a reference. The signal indicated by the dotted line in FIG. 14 is a signal in which the “stable incident wave” is multiplied by the reception weight and the phase is aligned in the same phase in each of the antenna elements 13-1 to 13-4.

The actual “time-varying incident wave” multiplied by the reception weight, that is, the “time-varying incident wave” indicated by the thin solid line in FIG. 14 is slightly shifted from the “stable incident wave” indicated by the dotted line. Strictly speaking, each antenna element does not have the same phase composition, but the “time-varying incident wave” behaves close to a “stable incident wave”. When combined using the reception weight set with “stable incident wave” as a reference, a combined signal having a large amplitude indicated by a thick solid line is obtained. In other words, if the number of antenna elements used in the base station apparatus is increased to an enormous number, the “stable incident wave” component of each antenna element is in-phase synthesized as a statistical effect, and the “random multiple reflected wave” is In order to cancel each other, the time-varying component with respect to the “stable incident wave” is relatively suppressed to a very small level.
Here, in the description of FIG. 13 and FIG. 14, the case where the base station apparatus is provided with four antenna elements has been described for the sake of simplicity. A statistical effect can be obtained by providing the antenna element.

  The in-phase synthesis of a statistical signal based on this “stable incident wave” can be used in the same way for both the transmission weight used during transmission and the reception weight used during reception. The transmission / reception weight used in the base station apparatus is calculated based on the channel estimation result. Even if the channel estimation is performed by receiving the training signal transmitted by the base station apparatus by the terminal apparatus, the signal transmitted by the terminal apparatus May be received by the base station apparatus to estimate the channel. In general, downlink and uplink channel information is asymmetric because amplifiers / filters used for transmission / reception are different, but a predetermined conversion formula is used for the uplink channel estimation result and the downlink channel estimation result. If the calibration process described later is used, the training signal transmitted by the terminal device is simultaneously received by all the antenna elements of the base station device, and uplink channel information is obtained by channel estimation using the result, By applying a predetermined conversion formula to this, channel information in the downlink direction can be acquired.

As explained above, by the averaging process over a long time for the channel information between each antenna element of the base station apparatus and the antenna element of each terminal apparatus, random time-varying components are suppressed, It is possible to extract a stable incident wave from no line-of-sight wave or a stable structure. Based on the extracted long-term average channel information, high phase gain can be obtained by performing in-phase synthesis on the target terminal device. In general, when an amplitude when a signal transmitted from one antenna element is received by one antenna element in another location is 1, signals transmitted from N antenna elements are in phase. When combined, the expected value of the amplitude of the received signal becomes N times. Since the received power is proportional to the square of the amplitude, the received power is N ² times. That is, a gain of ₁₀ Log ₁₀ (N ² ) [dB] can be obtained. If N is 100, the line gain associated with coherent transmission by in-phase combining corresponds to 40 [dB].

Next, high-order spatial multiplexing using reduction of given / interfered will be described. The in-phase synthesis described above is performed with a specific terminal device as a target, and a high line gain can be obtained pinpoint only for the terminal device. For example, if in-phase synthesis is performed on the terminal device 12-1 in FIG. 13, random phase synthesis is performed at other points where in-phase synthesis is not performed, such as the terminal device 12-2. Although FIG. 13 shows the case where the antenna elements 13-1 to 13-4 of the base station apparatus are four elements, when the number of antenna elements is N, the received power is increased N times as an expected value as a result of random phase synthesis. Become. In such a situation, when two signal sequences are simultaneously spatially multiplexed, the desired signal having the same phase composition is interfered with the line gain being N ² times per transmission antenna element. For undesired signals, the line gain is N times per transmission antenna element. Therefore, the relative SIR value is ₁₀ Log ₁₀ (N) [dB].

Thus, if the number (N) of transmitting antenna elements is 100, an SIR of 20 [dB] can be earned as an expected value while performing spatial multiplexing. That is, using the two paths indicated by thick solid lines in FIG. 13, it is possible to simultaneously perform communication between both the terminal device 12-1 and the terminal device 12-2 and the base station device in the same frequency band. .
Usually, when the number of antennas is enormous, when performing spatial multiplexing with a general multi-user MIMO technique, the amount of computation of signal processing proportional to the cube of the number of antennas is expected. Operation with a large number of antenna elements has been forced. Particularly important is null control to prevent a signal to a certain terminal device from interfering with other terminal devices. In the present invention, calculation of transmission / reception weights for realizing this null control is performed in real time. The signal processing in communication can be realized by reading the weight calculated in advance. For this reason, it is possible to realize high-order spatial multiplexing with a reduced calculation amount and circuit scale during operation while using a large number of antenna elements. Furthermore, in the present invention, a high SIR can be secured to some extent without performing the above-described null control. For this reason, even if the null control is broken to some extent due to channel fluctuation, the SIR value remains relatively high and stable. It also has the feature of being able to perform spatial multiplexing.

  If the number of transmitting antennas (N) is 100, the interference power will be approximately 10 times relatively large when 10 signals are spatially multiplexed. The expected value is about 10 [dB]. Of course, even if the average SIR value is 10 [dB], there is a certain extent of distribution. Therefore, even when the required SIR value is 10 [dB], in order to perform 10 or more spatial multiplexing, the SIR characteristic is good. It is preferable to combine a scheduling function for combining terminal devices and a directivity control function for performing interference suppression. However, the scheduling function and directivity control function performed here do not need to be premised on real-time control reflecting channel information that varies with time, and the time-varying component is averaged by the long-time averaging described above. Channel information can be used. Therefore, avoiding performing complicated signal processing sequentially every time communication is performed, in the pre-processing performed before the start of communication, by completing the heavy processing, and using the processing results, The load during operation can be reduced. As described above, the present invention constructs a stable condition that excels in SIR characteristics without null control as described above, and further stabilizes higher-order spatial multiplexing by combining a directivity control function for performing interference suppression. To be feasible.

Thus, in the present invention, instead of aiming at strict in-phase synthesis using real-time channel information, transmission / reception that can be suppressed within a certain amount of error even if there is a transmission / reception weight with a slight error from the strict transmission / reception weight. The first important point is to aim at quasi-optimal in-phase synthesis that draws stable and high line gain by a statistical effect by combining weights and using many antenna elements.
Furthermore, in the present invention, since the transmission / reception weight is a fixed value without being conscious of channel time fluctuations, for example, if it is acquired in advance before starting the operation of the communication service, Even if the number of antenna elements used for communication is increased to an enormous number, it is possible to keep the signal processing load low because it is only necessary to read the transmission / reception weights stored in the memory without performing individual calculations. This is the second important point.
Furthermore, in the present invention, channel estimation between individual antenna elements is not performed every time communication is performed, so that it is not necessary to assign the same number of channel estimation preamble signals as the spatial multiplexing number. For this reason, taking the case of using the OFDM modulation method as an example, conventionally, a preamble signal for channel estimation of 10 different symbols is required for 10 spatial multiplexing, but in the present invention, spatial multiplexing is used. A preamble signal for channel estimation of one symbol is sufficient without depending on the number. As a result, it is possible to perform higher-order spatial multiplexing while suppressing a decrease in the efficiency of the MAC layer, which is a third important point.
With respect to the above operating principle, supplementary matters for realizing these will be briefly described below before describing the detailed embodiment.

(About channel estimation averaging processing)
The base station apparatus according to the present invention is characterized by using a transmission / reception weight for performing in-phase synthesis of a statistical signal based on a “stable incident wave”. An overview of the corresponding channel estimation will now be described.
As described above, the base station apparatus extracts a component related to a “stable incident wave” by removing the influence of multiple reflected waves that are reflected on a mobile object and randomly vary. In order to extract a component related to “stable incident wave” in each antenna element as a pre-stage to obtain a statistical effect by a large number of antenna elements, the base station apparatus extracts each antenna element and terminal apparatus of the base station apparatus Channel information corresponding to a “stable incident wave” is acquired by performing channel estimation of individual channels with the antenna element over a long period of time and averaging the results.

  Before describing the specific acquisition method, first, the influence of reflection reflected on the moving bodies 14-1 to 14-3 such as a car in FIG. 13 will be considered. The situation of the reflected waves from these moving bodies does not change so much in a short time when the positions of the moving bodies are not displaced so much, but if these moving bodies move to physically different positions, the influence of the reflected waves is absolutely not. Expect to be different. In other words, even if the channel information is averaged in a short period of time, where the state of multiple multiple reflections reflected on the mobile object does not change so much, the mobile object that is the basis of the random reflected wave moves greatly. Is expected to be in another channel state.

  Next, how to obtain the long-term average channel information will be described focusing on points to be noted. In general, the clock signal of the base station apparatus and the clock signal of the terminal apparatus are not completely synchronized, and there is some frequency error. For example, when performing block transmission such as OFDM modulation scheme or SC-FDE transmission technology, the symbol timing (or block cycle) of one symbol is slightly shifted between the base station apparatus and the terminal apparatus. This symbol timing shift appears as a common complex phase rotation at all frequencies. Note that the clock signal of the base station device and the clock signal of the terminal device are A / D conversion and a clock signal that determines a sampling period when performing D / A conversion.

A similar problem of complex phase rotation is that the local oscillation signal output from the local oscillator used for up-conversion and down-conversion between the baseband signal and the radio frequency signal is not synchronized between the base station device and the terminal device. A frequency error also becomes a problem.
When transmission and reception are asynchronous and accompanied by a frequency error, the channel information measured at different times is complex depending on the time difference and the frequency error even if there is no time variation in the channel information in space. The phase component fluctuates.

  This is because, for example, the initial complex phase of the local oscillation signal input from the local oscillator that is multiplied in the mixer in the down-conversion process on the reception side cannot be matched every time communication is performed. In signal detection processing in communication, when channel estimation is performed with a training signal, channel information as a result including the influence of the initial complex phase is acquired, which causes a problem in signal detection processing of a signal subsequent to the training signal. There is nothing. However, when averaging in discrete time, there is a problem because even if there is no time variation in the channel information, it appears that there is time variation due to the uncertainty of the initial complex phase.

  However, the channel information required to calculate the transmission weight and reception weight for realizing in-phase synthesis at the time of reception is not the absolute value itself including the complex phase of the channel information indicating the characteristics of the transmission path, but the antenna information. It is sufficient to know the relative relationship of complex phases in channel information for each element. Therefore, when averaging channel estimation results measured at discrete times, one reference antenna element is set from a plurality of antenna elements of the base station apparatus, and channel information estimated by the antenna element is set. It is only necessary to add an offset to the complex phase component of the channel information in each antenna element for the complex phase component.

Specifically, when the base station apparatus includes K antenna elements, channel information of the k-th frequency component observed by antenna element #i (i = 1,..., K) is A _i · Exp ( Let φ _i ^(k) j). Here, j represents an imaginary unit, A _i represents the amplitude component of the channel information of the antenna element #i, and φ _i ^(k) represents the complex phase of the channel information of the k-th frequency component of the antenna element #i.
At this time, if the complex phase −φ ₁ ^(k) is added to all the antenna elements using the complex phase φ ₁ ^(k) of the antenna element # 1, the channel information of the antenna element #k corrected by the offset A _i · Exp {(φ _i ^(k) −φ ₁ ^(k) ) j} is obtained. If the spatial channel information is unchanged, the corrected channel information is influenced by the frequency error of the clock signal and the local oscillation signal between the base station device and the terminal device (that is, the uncertainty of the initial phase of the complex phase). Not affected. In the following description, the corrected channel information for removing the uncertainty of the initial phase is referred to as “relative component (of channel information)”. This correction is performed individually for each frequency component.

  Therefore, when channel information is averaged, it is necessary to perform such correction to eliminate the uncertainty of the complex phase component and to perform averaging. In addition, in the area where the line gain shown in (Problem 1) of the problem of the present invention is greatly insufficient in performing this averaging, the channel information obtained by the channel estimation is based on the basis before the averaging. Note that it may be difficult to obtain information. In such a situation, even if any kind of channel estimation training signal is received, it is generally impossible to detect the reception of the signal. Taking the case of the OFDM modulation method as an example, it means that OFDM symbol timing cannot be detected. Of course, if the guard interval cannot be removed, FFT cannot be performed. A channel estimation averaging method in such a low SNR environment and an example of a specific training signal are shown below.

(Example of training signal in the present invention)
FIG. 15 is a diagram illustrating an example of a training signal in the present invention. In the figure, reference numerals 1-1 to 1-3 denote general OFDM symbols, reference numerals 2-1 to 2-3 denote effective signal areas not including guard intervals, and reference numerals 3-1 to 3-3 denote In the present invention, training signals are shown. Reference numerals 4-1 to 4-3 denote end regions of the signals, reference numerals 5-1 to 5-3 denote guard intervals, and reference numerals 6-1 to 6-3 denote actual channels. The signal period used for estimation is shown. Although the OFDM signal includes a plurality of subcarrier components, in the figure, one subcarrier is extracted and illustrated as a sine wave.

  In the case of a conventional OFDM signal, a signal having a period of OFDM symbols (1-1 to 1-3) generates a signal region (2-1 to 2-3) effective as actual data, and the tail region of this signal. (4-1 to 4-3) are copied and pasted as guard intervals (5-1 to 5-3) in the head area of the signal to generate the entire OFDM symbols (1-1 to 1-3). It was. In normal communication, information on the amplitude and complex phase when the timing of the leading portion of the effective signal area (2-1 to 2-3) from which the guard interval is removed is extracted by timing detection and the timing is used as a starting point. Are obtained in the channel estimation.

  However, since it is sufficient to obtain the relative phase relationship of each antenna element in the calculation of the transmission / reception weight of the present invention, it is not necessary to accurately grasp the initial complex phase, and an appropriate value such as the head of the OFDM symbol There is no need to start from a specific timing. Therefore, it is not necessary to be an OFDM signal in which a guard interval is set, and training signals (3-1 to 3-3) which are signals obtained by extracting effective signal regions (2-1 to 2-3) of OFDM symbols and continuing them. ) May be repeated many times. Here, since the sections are continuously connected, the symbol timing does not substantially make sense in the training signal over a plurality of periods. On the receiving side, the received training signals (3-1 to 3-3) are cut out at arbitrary start timings, for example, signal periods (6-1 to 6-3) used for actual channel estimation, and the section 6 -1, section 6-2, and section 6-3 may be added.

(Compensation of local oscillator frequency error between base station and terminal)
In the channel averaging using the training signal, a relatively short time average of a plurality of continuous sections 6-1, 6-2, and 6-3 is performed. The quantitative meaning of “time” is the influence that depends on the frequency error of the clock signal and the local oscillation signal between the base station device and the terminal device (strictly speaking, the residual frequency error remaining after the frequency error compensation processing shown below) Mean in a range that can be ignored.
For example, in a local oscillation signal having a center frequency of 2.4 [GHz], the frequency error of the local oscillator is 1 p. p. m. In this case, the maximum value of the frequency error of the local oscillation signal is 2.4 [kHz]. That is, this is an error that the phase rotates by 2π in 416 μsec. At this time, if it is desired to suppress the rotation of the complex phase accompanying the frequency error within 1/10 of one period (2π) in the time length for averaging, the time length usable for averaging is about 40 μsec. Become.
However, in the WiMAX example where the wide area is the service area, one symbol period for eliminating the influence of the long delay wave is set to about 100 μsec, which is not sufficient as the time for the averaging process. In order to solve these problems, the following correction processing for compensating for the frequency error is performed here.

  In general, frequency error correction can be dealt with by signal processing called AFC (Automatic Frequency Control). If the same signal is repeatedly received like the current training signal, it is generally possible to extract the frequency error component by multiplying the signal shifted by one period. By applying this AFC process, it is possible to extract a frequency error and perform correction to cancel the frequency error. However, when the received signal has a low SNR, even if an attempt is made to extract a frequency error from adjacent symbols by applying AFC processing, there is a possibility that an erroneous frequency error is extracted due to being buried in noise. Therefore, the AFC processing also needs to be performed for a length of time that can improve the SNR of the original signal.

For example, if sampling data represented by a complex number at time t is represented as S (t) and a frequency error is represented as Δf, the amount of rotation of the complex phase at time t is 2πΔf · t. Therefore, when the frequency compensation is ideally performed on the sampling data S (t), the frequency-compensated sampling data is S (t) · Exp (−2πjΔf · t).
If the sampling period is represented by Δt and the period of one symbol is T, the number of data in one period is given by N = T / Δt. At this time, if the time t = m ′ · Δt, and m and M are m = mod (m ′, N) and M = Int (m ′ / N), the sampling data S (t) is discrete. It can be expressed as a numerical sequence {S ^(M) _m } determined by a specific time. Here, the function “mod (x, y)” is a function for obtaining a remainder when x is divided by y. Further, the function “Int (x)” is a function (a function for truncating the decimal part) to obtain the integer part of x.
Furthermore, a sequence in which the sampling data S (t) is ideally frequency-compensated can be expressed as {S ^(M) _m · Exp (−2πjΔf · Δt · [M × N + m]}, where M ₀ symbol period as a whole. Sampling shall be performed.

The frequency compensated number sequence {S ^(M) _m · Exp (−2πjΔf · Δt · [M × N + m]}) obtained by adding a large number of M for each m to S _m is expressed by the following equation (10). Here, in Formula (10), S with “˜ (tilde)” added above is expressed as “˜S.” Also, “Exp (X)” is the base of the natural logarithm e to the Xth power. It is a function which shows.

To improve the SNR by AFC process may be determined to Δf to maximize the amplitude of the to S _m of formula (10). Therefore, an evaluation function G (Δf) expressed by the following equation (11) is determined.

  ^ S (M, M ′) in the equation (11) is expressed by the following equation (12).

  Since Δf that maximizes the evaluation function G (Δf) may be obtained, a conditional expression represented by the following expression (13) is obtained.

  If the real number Δf satisfying the conditional expression (13) is obtained numerically, the frequency error between the base station apparatus and the terminal apparatus is calculated, and using this Δf, the addition for one cycle given by the expression (10) Channel estimation may be performed using averaged sampling data. In the case of the OFDM modulation method, channel information of each subcarrier component is calculated by FFT processing based on the sampling data of one cycle.

Note that even if the equation (13) is not necessarily used, if the possible range of Δf is limited, Δf is set with an appropriate step size within the range, and the equation (11) is set for those Δf. Δf that gives the maximum value may be searched. In this case, the center frequency to be used is 2.4 GHz and the frequency error is 1 p. p. m. In this case, the range of Δf is from −2.4 kHz to +2.4 kHz. The optimum value of the step size varies depending on the required accuracy. For example, if Δf is set in increments of 10 Hz and equation (11) is calculated, true Δf that maximizes equation (11) is obtained. It is possible to search for Δf within the range of residual frequency error within ± 5 Hz. That is, the frequency error is suppressed within 5 Hz, and even if the time length (M ₀ × T) for averaging the M ₀ period is assumed to be about 5 milliseconds, the phase error within the averaging period is It falls within 1/40 of 2π (angle is 9 degrees). Considering that the phase rotates constantly during the averaging period, it is possible to calculate the channel information with an accuracy that does not hinder the operation. In other words, the time length (that is, the value of M ₀ ) during averaging of the M ₀ period is limited within a range in which the phase error within the averaging period can be suppressed to a predetermined value. It will be.

In this way, when performing the averaging process, averaging on a continuous relatively short time scale and averaging on the result of channel estimation in discrete time are performed in two stages. It should be noted that the time length for averaging on a relatively short time scale is subject to the above-mentioned limitations. Also, in the discrete time channel estimation result, by using the complex phase φ ₁ ^(k) of the antenna element # 1 as described above, an offset of the complex phase −φ ₁ ^(k) is added to all the antenna elements. The problem of the uncertainty of the initial complex phase can be avoided.

(Effects of individual amplifier differences (calibration))
In an actual wireless communication device, in signal processing on the transmission side, signal amplification is often performed by a high power amplifier immediately before transmission. In this case, there is an error in the amplification factor due to the individual difference of the high power amplifier, and the complex phase may rotate at a different value for each high power amplifier in the high power amplifier. Similarly, in signal processing on the reception side, signal amplification is often performed by a low noise amplifier immediately after reception. In this case, there is an error in the amplification factor due to individual differences of the low noise amplifiers, and the complex phase may rotate with a different value for each low noise amplifier in the low noise amplifier.

  In particular, the amplification factor and phase rotation amount of the high power amplifier and the low noise amplifier have frequency dependency. If the individual difference in the amplification factor with frequency dependency and the amount of rotation of the complex phase is so large that it cannot be ignored, it is necessary to perform calibration processing when estimating the downlink channel information from the uplink channel information. is there. Since the error of the amplification factor and the amount of phase rotation is almost stable in time, measure the error of the amplification factor and the amount of phase rotation in advance, and use a coefficient to cancel the influence of the error. Converts link channel information to downlink channel information.

  In the base station apparatus in the following embodiment, the transmission weight and the reception weight are calculated using channel information obtained by performing long-time averaging on the uplink channel estimation result. Also in the above description, the amplitude and the complex phase may actually be changed by a high power amplifier or a low noise amplifier (strictly speaking, a transmission system and a reception system circuit including other circuits such as a filter). In this case, it has been described that a calibration coefficient for correcting in accordance with a change in amplitude or complex phase is acquired in advance and used for correction. Although a known technique may be used for the calibration process, an example of the calibration process will be described below.

FIG. 16 is a diagram illustrating asymmetry of channel information between the uplink and the downlink. In the figure, reference numerals 25-1 to 25-3 denote wireless modules, reference numerals 21-1 to 21-3 denote high power amplifiers (HPA), and reference numerals 22-1 to 22-3 denote low noise amplifiers (LNA). Reference numerals 23-1 to 23-3 denote time division switches (TDD-SW), and reference numerals 24-1 to 24-3 denote antenna elements.
Here, since only functions that affect channel information are extracted in the base station apparatus, configurations other than those illustrated are omitted, but the wireless modules 25-1 to 25-3 include other functions. Further, when the signal passes through each of the high power amplifiers 21-1 to 21-3, the amplitude and the complex phase are Z _{HPA # 1} (f _k ), Z _{HPA # 2} (f _k ), Z _{HPA # 3} (f _k ). Further, when the signal passes through each of the low noise amplifiers 22-1 to 22-3, the amplitude and the complex phase are Z _{LNA # 1} (f _k ), Z _{LNA # 2} (f _k ), and Z _{LNA # 3} (f _k ). Here, it is assumed that there is a frequency dependency as a general condition, and the frequency “(f _k )” for the k-th frequency component is described.

Here, for example, channel information when signals are transmitted from the wireless module 25-1 and the wireless module 25-2 to the test wireless module 25-3 will be described. Here, spatial channel information between the antenna element 24-1 of the wireless module 25-1 and the antenna element 24-3 of the wireless module 25-3 is represented by h ₁ (f _k ), and the wireless module 25 Channel information on the space between the antenna element 24-2 of -2 and the antenna element 24-3 of the wireless module 25-3 is represented by h ₂ (f _k ).

At this time, the channel information when the signal is actually transmitted from the wireless module 25-1 to the wireless module 25-3 indicates a change accompanying the passage of the high power amplifier 21-1 at h ₁ (f _k ) in space. A coefficient Z _{HPA # 1} (f _k ) and a coefficient Z _{LNA # 3} (f _k ) indicating a change accompanying the passage through the low noise amplifier 22-3 are observed as multiplied values.
Similarly, the channel information when transmitting a signal from the wireless module 25-2 to the wireless module 25-3 is a coefficient Z indicating a change accompanying passage of the high power amplifier 21-2 at h ₂ (f _k ) in space. _{HPA # 2} (f _k ) and a coefficient Z _{LNA # 3} (f _k ) indicating a change accompanying passage through the low noise amplifier 22-3 are observed as multiplied values.

Therefore, the channel from the wireless module 25-1 to the wireless module 25-3 is represented by Z _{HPA # 1} (f _k ) · h ₁ (f _k ) · Z _{LNA # 3} (f _k ). A channel from the wireless module 25-2 to the wireless module 25-3 is represented by Z _{HPA # 2} (f _k ) · h ₂ (f _k ) · Z _{LNA # 3} (f _k ). Therefore, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), the wireless module 25-1 and the wireless module 25-2 have relatively Z _{HPA # 2} (f _k ) / Z _{HPA # 1} (f _k ) difference occurs.

This situation is the same on the receiving side. When the signal transmitted from the wireless module 25-3 is received by the wireless module 25-1, the channel information is stored in the high power amplifier 21 in h ₁ (f _k ) in space. Observed as a value obtained by multiplying a coefficient Z _{HPA # 3} (f _k ) indicating a change accompanying the passage of −3 and a coefficient Z _{LNA # 1} (f _k ) indicating a change accompanying the passing of the low noise amplifier 22-1 Is done.
Similarly, when the signal transmitted from the wireless module 25-3 is received by the wireless module 25-2, the channel information changes in space h ₂ (f _k ) due to the passage of the high power amplifier 21-3. It is observed as a value obtained by multiplying the indicated coefficient Z _{HPA # 3} (f _k ) by the coefficient Z _{LNA # 2} (f _k ) indicating the change accompanying the passage of the low noise amplifier 22-2.

Therefore, the channel from the wireless module 25-3 to the wireless module 25-1 is represented by Z _{HPA # 3} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ). A channel from the wireless module 25-3 to the wireless module 25-2 is represented by Z _{HPA # 3} (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). Therefore, in addition to the difference between the channel information h ₁ (f _k ) and h ₂ (f _k ), the wireless module 25-1 and the wireless module 25-2 have relatively Z _{LNA # 2} (f _k ) / Z _{LNA # 1} (f _k ) difference occurs.

As described above, the base station apparatus according to the embodiment includes a channel including changes caused by the low noise amplifiers 22-1 to 22-3 connected to each antenna element by taking a long-time average with respect to the received training signal. Information can be acquired on the uplink.
However, the base station apparatus cannot directly obtain channel information in the downlink. Therefore, downlink channel information is obtained by conversion from uplink channel information. For this conversion, the influence of individual differences between the low-noise amplifiers 22-1 to 22-3 and the high-power amplifiers 21-1 to 21-3 connected to the antenna elements 24-1 to 24-3 is canceled. There is a need.

  Therefore, in the manufacturing stage of the base station apparatus, a test radio module 25-3 serving as a reference is prepared, and the antenna terminal of the test radio module 25-3 and the antenna terminals of the radio modules 25-1 and 25-2 are prepared. And channel information including changes by the high power amplifiers 21-1 to 21-3 and the low noise amplifiers 22-1 to 22-3 are measured in an environment where the channel information on the propagation path is a common value. Then, correction is performed using the measured channel information.

FIG. 17 is a diagram showing an outline of calibration. In the figure, reference numerals 26-1 to 26-3 indicate antenna terminals, and reference numeral 27 indicates a coaxial cable. In addition, the same code | symbol is attached | subjected to the same function part as the function part shown in FIG.
FIG. 17A shows a configuration in which the wireless module 25-3 and the wireless module 25-1 are connected by a coaxial cable. FIG. 17B shows a configuration in which the wireless module 25-3 and the wireless module 25-2 are connected by a coaxial cable. FIG. 16 shows a state in which a signal propagates in an actual space, while FIG. 17 shows a state in which a signal propagates on a coaxial cable without passing through an antenna element.

The channel information of the coaxial cable 27 serving as a propagation path connecting the wireless modules 25-1 and 25-2 and the wireless module 25-3 is h ₀ (f _k ).
At this time, the channel information from the wireless module 25-1 to the wireless module 25-3 is expressed by Z _{HPA # 1} (f _k ) · h ₀ (f _k ) · Z _{LNA # 3} (f _k ). The channel information from the wireless module 25-2 to the wireless module 25-3 is expressed by Z _{HPA # 2} (f _k ) · h ₀ (f _k ) · Z _{LNA # 3} (f _k ).
The channel information from the wireless module 25-3 to the wireless module 25-1 is represented by Z _{HPA # 3} (f _k ) · h ₀ (f _k ) · Z _{LNA # 1} (f _k ), and the wireless module 25 -3 to the wireless module 25-2 is represented by Z _{HPA # 3} (f _k ) · h ₀ (f _k ) · Z _{LNA # 2} (f _k ).

Therefore, after measuring these channel information, calibration coefficients C ₁ (f _k ) and C ₂ (f _k ) represented by the following equations (14) and (15) are calculated.

The channel information from the wireless module 25-3 to the wireless module 25-1 is represented by Z _{HPA # 3} (f _k ) · h ₁ (f _k ) · Z _{LNA # 1} (f _k ), and the wireless module 25- It has been described that the channel information from 3 to the wireless module 25-2 is represented by Z _{HPA # 3} · (f _k ) · h ₂ (f _k ) · Z _{LNA # 2} (f _k ). When these are multiplied by the calibration coefficients C ₁ (f _k ) and C ₂ (f _k ) of the equations (14) and (15), the following equations (16) and (17) are obtained.

The right sides of Expression (16) and Expression (17) are the channel information from the wireless module 25-1 to the wireless module 25-3 and the channel information from the wireless module 25-2 to the wireless module 25-3 described above. It matches.
As described above, the calibration coefficients corresponding to the equations (14) and (15) are acquired in the manufacturing stage of the base station device, and stored in the base station device, thereby calibrating them. The downlink channel information can be calculated from the uplink channel information using the coefficient.

  In the following embodiment, these calibration coefficients are acquired in advance, and the description will be made mainly on the case where the values are used in digital signal processing.Naturally, these calibration coefficients are displayed on an analog circuit. If calibration is performed in the base station apparatus so that all values are almost constant (if the complex phase is constant, the absolute value itself may be different), all calibration coefficients are It can also be read as a process regarded as 1. Similarly, even when the complex phases of the uplink and downlink are adjusted to be constant values, as a result, the complex phases of the calibration coefficients expressed by the equations (14) and (15) are all Since the antenna element has a substantially constant value, the same effect can be obtained.

(Residual interference cancellation process using recurrence formula)
In the explanation of the operation principle so far, explanation on the short-time averaging processing of channel information, acquisition of relative components, long-time averaging processing, acquisition of transmission / reception weights, and explanation on calibration and frequency error associated therewith, etc. I went. By using these technologies, it is possible to realize higher-order spatial multiplexing with simple control that can deal with real-time, but generally transmission and reception weights depend on the combination of terminal devices that simultaneously perform spatial multiplexing. First, the transmission / reception weight must be calculated by determining the combination. In the first configuration example of the related art of the present invention described below, a case where an appropriate transmission / reception weight is calculated for the combination is described.

However, considering the case of performing spatial multiplexing of up to 30 terminal devices from 100 terminal devices, for example, the number of combinations is ₁₀₀ C ₃₀ , and astronomical combinations of about 3 × 10 ²⁵ are possible. Exists. For this reason, even when transmission / reception weights are obtained in advance, it is not practical to calculate all transmission / reception weights for this number of combinations. Furthermore, even if it is stored in the memory, the capacity of the memory becomes enormous. Suppose that the number of antenna elements is 100, the number of subcarriers is 1024, the number of spatial multiplexing is 30, and the weight value for one frequency component and antenna element combination is described in 4 bytes. In this case, a memory capacity of 100 × 1024 × 30 × 4 = 12.288 Mbytes is required for all subcarriers for one terminal device combination pattern. Overall, this requires 3 × 10 ²⁵ times the capacity, which far exceeds the practical storage capacity limit. Therefore, it is necessary to limit the combination patterns of terminal devices that can actually be used, and to perform spatial multiplexing by selecting from the patterns.

  On the other hand, in the second configuration example of the related technology of the present invention shown below, a combination of terminal devices that simultaneously perform spatial multiplexing is used to simply set the transmission / reception weights so that signals are in-phase combined with each antenna element. The transmission / reception weight is determined without depending on. For this reason, spatial multiplexing can be performed in any combination without limiting the combination of terminal devices that perform spatial multiplexing simultaneously. However, since this does not suppress mutual interference components by complete null control as shown in the first configuration example of the related art of the present invention, the SIR (or SINR) characteristic is the first of the related art of the present invention. The second configuration example deteriorates more than the first configuration example. As a result, the number of signal systems that can be spatially multiplexed simultaneously within a range that satisfies the desired characteristics decreases, and the frequency utilization efficiency decreases accordingly.

  In the present invention, if attention is paid to the uplink, a signal for each frequency component received at each antenna element is multiplied by a reception weight corresponding to the focused terminal device to obtain a combined signal, and a plurality of signals are simultaneously multiplexed. Based on these received signals corresponding to the terminal device, signal processing for further suppressing interference signals that cannot be removed by the previous reception weight is performed in the base station device on the receiving side. Similarly, in the downlink, the frequency components of transmission signals generated for a plurality of terminal devices that simultaneously perform spatial multiplexing can be eliminated on the terminal device side in the previous transmission weight based on these signals. Signal processing for further suppressing non-interfering signals is performed on the base station apparatus side that is the transmission side.

  In this signal processing, for each combination of two individual terminal devices among a plurality of terminal devices that simultaneously perform spatial multiplexing, calculation is performed based on channel information and transmission / reception weight information corresponding to the terminal device. The correction information of the transmitted / received signal to be transmitted is calculated and stored in advance, and the stored information is read and signal processing is performed. Further, the signal processing is represented by a recurrence formula expressed in a matrix × vector format, and an increase in the entire calculation amount is suppressed by repeating the recurrence formula. In general, multiplication operations between N × N matrices (or matrix inverse matrix operations and singular value decomposition, etc.) require multiplication operations in a number proportional to the cube of N. In the calculation, the number of multiplications can be suppressed to the square of N, and can be realized by a simple routine process as compared with the inverse matrix calculation. For this reason, it is easy to realize hardware as a circuit configuration, which is effective for realizing real-time processing.

  Therefore, in the present invention, the transmission / reception weight shown in the second configuration example of the related technology of the present invention (complete interference suppression by null control is not performed as in the first configuration example of the related technology of the present invention, In other words, the remaining interference by using the transmission / reception weight is estimated by a calculation process using a recurrence formula, and a signal for canceling the residual interference component is added to the remaining transmission / reception weight. Signal processing that suppresses interference components is realized.

[Configuration example of related technology of the present invention]
The present invention is constituted by a combination of a technique related to estimation of channel time variation and communication quality management control and related techniques. Prior to describing a specific embodiment of the present invention, a configuration example of related technology that is the basis of the embodiment will be described first.

(First configuration example of related technology)
In the first configuration example of the related art of the present invention, a radio communication system including a base station apparatus including a plurality of antenna elements and a plurality of terminal apparatuses communicating with the base station apparatus will be described as an example.
FIG. 18 is a schematic block diagram showing the configuration of the base station apparatus 10 in the first configuration example of the technology related to the present invention. As shown in the figure, the base station apparatus 10 includes a reception unit 100, a transmission unit 140, a transmission / reception weight calculation unit 120a, an interface circuit 170, a MAC layer processing circuit 180, a communication control circuit 110, and a storage circuit 115. The MAC layer processing circuit 180 has a scheduling processing circuit 181.
The base station apparatus 10 inputs and outputs data with an external device or a network via the interface circuit 170. The interface circuit 170 detects data to be transferred on the wireless line among the input data, and outputs the detected data to the MAC layer processing circuit 180. The MAC layer processing circuit 180 performs processing related to the MAC layer in accordance with an instruction from the communication control circuit 110 that performs management control of the operation of the entire base station apparatus 10. Here, the processing related to the MAC layer includes conversion of data input / output by the interface circuit 170 and data transmitted / received on the wireless line, addition of MAC layer header information, and the like. In this process, the scheduling processing circuit 181 performs various scheduling processes including a combination of terminal apparatuses that simultaneously perform spatial multiplexing. The scheduling processing circuit 181 outputs the scheduling result to the communication control circuit 110.

In spatial multiplexing transmission, signals are transmitted to a plurality of terminal devices at a time at the time of transmission, so that a plurality of signal sequences are output from the MAC layer processing circuit 180 to the transmission unit 140. In addition, a plurality of signal sequences transmitted from a plurality of terminal devices are output from the receiving unit 100 to the MAC layer processing circuit 180 during reception. The transmission / reception weight calculation unit 120a manages reception weights and transmission weights used when the reception unit 100 and the transmission unit 140 spatially multiplex and transmit / receive data.
The storage circuit 115 stores in advance various information of each terminal device in the wireless communication system (for example, SNR information, information on the degree of time variation, information on the optimum transmission mode, etc.). Information such as the SNR information and optimum transmission mode of the terminal device stored in the storage circuit 115 is used when scheduling is performed. Note that the memory circuit 115 is not necessarily required in both the embodiment of the present invention and related technology, and may be omitted in some cases.
Hereinafter, the configuration (reception unit 100) related to reception (uplink) in the base station apparatus 10 and the configuration (transmission unit 140) related to transmission (downlink) will be described separately.

  FIG. 19 is a schematic block diagram illustrating an example of a configuration of the receiving unit 100 included in the base station device 10 according to the first configuration example of the technology related to the present invention. As shown in the figure, the receiving unit 100 includes antenna elements 101-1 to 101-K, TDD switches 102-1 to 102-K, low noise amplifiers (LNA) 103-1 to 103-K, a local oscillator 104, a mixer. 105-1 to 105-K, filters 106-1 to 106-K, A / D converters 107-1 to 107-K, FFT circuits 108-1 to 108-K, and received signal processing circuits 109-1 to 109 -L is provided. Reception signal processing circuits 109-1 to 109-L and TDD switches 102-1 to 102-K are connected to communication control circuit 110 shown in FIG. The FFT circuits 108-1 to 108-K and the reception signal processing circuits 109-1 to 109-L are connected to the transmission / reception weight calculation unit 120a shown in FIG. The antenna elements 101-1 to 101-K correspond to the antenna elements 13-1 to 13-4 in FIG. In this configuration example, the description is made on the assumption that the TDD system is used, but in principle it can be extended to the FDD system.

  The base station apparatus 10 of this configuration example includes A / D converters 107-1 to 107-K from TDD switches 102-1 to 102-K corresponding to the K antenna elements 101-1 to 101-K, respectively. Are provided in parallel, and the FFT circuits 108-1 to 108-K are connected to the outputs of the A / D converters 107-1 to 107-K. In order to estimate downlink channel information from uplink channel information, the same antenna elements 101-1 to 101-K are used for transmission and reception. The TDD switches 102-1 to 102-K switch the flow of transmission signals and reception signals.

  The TDD switches 102-1 to 102-K output the signals received via the antenna elements 101-1 to 102-K to the low noise amplifiers 103-1 to 103-K. The low noise amplifiers 103-1 to 103-K amplify the signals output from the TDD switches 102-1 to 102-K and output the amplified signals to the mixers 105-1 to 105-K. The local oscillator 104 generates a local oscillation signal having a predetermined frequency, and outputs the generated local oscillation signal to each of the mixers 105-1 to 105-K. Here, the local oscillation signals input to the mixers 105-1 to 105-K are the same signal, and the local oscillation signals having the same frequency and phase are input to the mixers 105-1 to 105-K.

  The mixers 105-1 to 105-K multiply the signals input from the low noise amplifiers 103-1 to 103-K by the local oscillation signal input from the local oscillator 104, down-convert them, and perform filters 106-1. 106-K. Filters 106-1 to 106-K remove signals outside the band of the channel to be received, included in the signals down-converted by mixers 105-1 to 105-K, and A / D converters 107-1 to 107-. Output to K.

  The A / D converters 107-1 to 107-K digitize the baseband signals input from the filters 106-1 to 106-K. If the digital baseband signal input from the A / D converters 107-1 to 107-K is a signal including a normal data communication signal, the FFT circuits 108-1 to 108-K correspond to the digital baseband. The signal is separated into signals for each frequency component. At this time, the FFT circuits 108-1 to 108-K remove the guard interval for each OFDM signal (or block transmission block) for each frequency component signal, and perform FFT processing on the remaining sampling data. The signal on the time axis is converted into a signal on the frequency axis, and the signal is output to the reception signal processing circuits 109-1 to 109-L. Furthermore, if the input digital baseband signal is a training signal for channel estimation different from a normal data communication signal in accordance with the control of the communication control circuit 110, the FFT circuits 108-1 to 108-K The signal is output to the transmission / reception weight calculation unit 120a. The communication control circuit 110 determines whether the digital baseband signal input to the FFT circuits 108-1 to 108-K is a signal including a data communication signal or a different training signal.

  Here, the “training signal for channel estimation different from the normal data communication signal” means the training signals 3-1 to 3-1 shown in FIG. 15 that are used in the channel information estimation process used in the transmission / reception weight calculation in this configuration example. The 3-3 signal is completely different from a radio packet containing user data or various control information in radio communication. In this related technology, except for some configuration examples, it is necessary to perform signal processing (such as short-time averaging processing of channel information shown in FIG. 22 below) different from normal data communication. As described above, since it differs from a conventional OFDM signal or the like, the contents of signal processing are also slightly different. For this reason, in the FFT circuits 108-1 to 108-K, a series of processing associated with FFT is not performed on the training signal for channel estimation different from the normal data communication signal, and the digital baseband signal remains as it is. It is assumed that a function that outputs to the transmission / reception weight calculation unit 120a and performs processing including FFT in the transmission / reception weight calculation unit 120a is implemented. However, although details will be described later, in order to realize the functions described herein, it is of course possible to substitute other functional blocks by performing equivalent processing, and this is also a method for realizing the related technology. Is considered part of

  Reception signal processing circuits 109-1 to 109-L are each associated with a terminal device that transmits data using spatial multiplexing or a spatially multiplexed signal sequence, and detects data of the corresponding terminal device from the received signal. A signal detection process is performed. Specifically, the reception signal processing circuits 109-1 to 109 -L transmit / receive the reception weights corresponding to the transmission source terminal devices assigned to the respective frequency components to the transmission / reception weight calculation unit 120 a. Is multiplied by the reception weight for each frequency component. The reception signal processing circuits 109-1 to 109-L add and synthesize the signals multiplied by the reception weights for each of the antenna elements 101-1 to 101-K for each frequency component, and signal the added and synthesized signals. The detection process is performed, and the obtained data outputs # 1 to #L are output to the MAC layer processing circuit 180.

  Specifically, when the OFDM (A) modulation method is used, reception signal processing circuits 109-1 to 109-L perform demodulation processing for each subcarrier on the added and synthesized signal, and SC- When FDE is used, signal equalization processing on the frequency axis is performed on the frequency component signals added and synthesized, and demodulation processing is performed on the signal synthesized by IFFT processing. The demodulation processing here includes channel estimation for the signal after signal processing such as addition synthesis, and signal detection processing is performed based on the channel information estimated here. Furthermore, it performs decoding processing for error correction as necessary, and outputs data. The signal processing on the MAC layer in the MAC layer processing circuit 180 is the same as the processing using a known technique, and the description is omitted here.

  When the reception signal processing circuits 109-1 to 109-L perform signal processing, it is necessary to use different reception weights for each terminal device as a transmission source. The communication control circuit 110 manages overall control related to a series of communications. In particular, the communication control circuit 110 manages which terminal device receives a signal at which timing and which reception weight is used. Therefore, access control between the base station apparatus 10 and the terminal apparatus in this configuration example is managed by centralized control of the base station apparatus 10.

As a supplement, the communication control circuit 110 uses absolute time / timing synchronization using GPS or the like for timing synchronization between its own device (base station device 10) and the terminal device. May be.
In addition to absolute time synchronization, if a rough distance between the base station apparatus 10 and the terminal apparatus is known, a propagation delay corresponding to the distance is set in the terminal apparatus in advance. The terminal device may start transmission of the uplink signal at a time obtained by subtracting the propagation delay as a predetermined offset from the reception time of the signal serving as a timing reference of the base station device 10.

  Specifically, if an example of access control using time division multiple access (Time Division Multiple Access (TDMA)) is used, the terminal device is based on the frame timing obtained by timing detection such as a preamble at the beginning of the TDMA frame, The contents of slot assignment in the frame are grasped and communication operation is performed. Normally, a signal is transmitted at the timing of the uplink time slot. However, in the so-called time alignment control, the terminal is ahead of the timing that the terminal recognizes by considering the propagation delay. The transmission of the signal is started at the timing of the time, and as a result, the time at which the signal arrives at the base station apparatus 10 is adjusted so as to match the timing recognized by the base station.

  The amount of adjustment required at this time is that the actual signal is reciprocated from the base station device 10 to the terminal device and then to the base station device 10, so the terminal device transmits it ahead of time by twice the propagation delay. Will start. Note that this timing adjustment does not necessarily have to be performed by the terminal device. If the base station device 10 can grasp the distance between the own device and the terminal device or the propagation delay corresponding to the distance, the base station device 10 It is also possible to adjust the timing by adjusting the time at which the signal is received backward by that amount (twice the propagation delay). Alternatively, the base station apparatus may directly instruct the terminal apparatus that the time advanced by that time is the transmission timing.

  In this way, the base station apparatus 10 can start reception processing at a timing grasped by any means such as absolute time synchronization or time alignment control using GPS, and can perform processing with a known symbol timing. It is. The communication control circuit 110 controls and manages all of these timing control, access control, switching of the TDD switches 102-1 to 102-K, provision of terminal device information that is a transmission source when reading the reception weight, and the like. I do.

Next, the configuration of the transmission / reception weight calculation unit 120a will be described.
FIG. 20 is a schematic block diagram showing a configuration example of the transmission / reception weight calculation unit 120a in the first configuration example of the technology related to the present invention. As shown in the figure, the transmission / reception weight calculation unit 120a includes a channel information short-time average circuit 121, a relative component acquisition circuit 122, a channel information long-time average circuit 123, a multiuser MIMO (MU-MIMO) reception weight calculation circuit 124a, Multi-user MIMO (MU-MIMO) reception weight storage circuit 125a, calibration circuit 126, multi-user MIMO (MU-MIMO) transmission weight calculation circuit 127a, multi-user MIMO (MU-MIMO) transmission weight storage circuit 128a, and calibration A coefficient storage circuit 129 is included. Note that channel information, transmission / reception weights, calibration coefficients, and the like in the following description are all different for each frequency component, and are individually calculated, processed, recorded, and managed for each frequency component.

  The channel information short-time averaging circuit 121 performs a short-time averaging process of the training signal on the signals input from the FFT circuits 108-1 to 108-K according to instructions from the communication control circuit 110 (frequency error compensation is performed if necessary). And further separating the signal on the time axis for each frequency component by FFT processing), and for each terminal device, information on the uplink channel between the terminal device and each of the antenna elements 101-1 to 101-K is obtained. Acquired for each frequency component. The relative component acquisition circuit 122 acquires, for example, the relative component of each antenna element 101-1 to 101-K with respect to the antenna element 101-1, using the complex phase of the antenna element 101-1 as a reference. The channel information long-time average circuit 123 determines, for each terminal device, the antenna elements from the relative components for each of the plurality of antenna elements 101-1 to 101-K acquired at discrete times acquired by the relative component acquisition circuit 122. A long-time averaging process for calculating an average value of relative components for each of 101-1 to 101-K is performed, and the calculated average value is output as channel information.

  The multiuser MIMO reception weight calculation circuit 124a calculates a reception weight for each predetermined combination of the terminal devices based on the channel information output from the channel information long-time average circuit 123, and calculates the received weight and the combination. Are associated with each other and output to the multi-user MIMO reception weight storage circuit 125a. The multiuser MIMO reception weight storage circuit 125a stores the reception weight output from the multiuser MIMO reception weight calculation circuit 124a and the combination of terminal devices in association with each other. Here, the combination of terminal devices may be all combinations of a plurality of terminal devices provided in the wireless communication system, or may be a limited combination that is frequently used.

The calibration circuit 126 obtains downlink channel information by multiplying the channel information output from the channel information long-time average circuit 123 by a predetermined calibration coefficient. The multiuser MIMO transmission weight calculation circuit 127a calculates a transmission weight for each predetermined combination of the terminal devices based on the downlink channel information acquired by the calibration circuit 126, and calculates the transmission weight and the combination. Are associated with each other and output to the multi-user MIMO transmission weight storage circuit 128a. The multiuser MIMO transmission weight storage circuit 128a stores the transmission weight output from the multiuser MIMO transmission weight calculation circuit 127a and the combination of terminal devices in association with each other.
The calibration coefficient storage circuit 129 stores in advance the calibration coefficient for each frequency component used when calculating downlink channel information from uplink channel information for each of the antenna elements 101-1 to 101 -K. ing.

Note that a series of processes related to channel information estimation performed by the transmission / reception weight calculation unit 120a and a series of processes such as calculation and storage of subsequent transmission / reception weights are performed for each frequency component. That is, after estimating the frequency error using Equation (11) or Equation (13), the sampling data for each m to S _m after short-time averaging given by Equation (10) corrected for the frequency error On the other hand, after FFT processing is performed and channel information averaged for a short period of time is acquired by separating the frequency components, a series of processing is performed on each frequency component based on the acquired channel information.

  FIG. 21 is a diagram illustrating an example of the configuration of the transmission unit 140 in the base station apparatus 10 in the first configuration example of the technology related to the present invention. As shown in the figure, the transmission unit 140 includes transmission signal processing circuits 141-1 to 141-L, addition synthesis circuits 142-1 to 142-K, IFFT & GI adding circuits 143-1 to 143-K, and D / A conversion. 144-1 to 144-K, a local oscillator 145, mixers 146-1 to 146-K, filters 147-1 to 147-K, and high power amplifiers (HPA) 148-1 to 148-K. . Transmission signal processing circuits 141-1 to 141-L and TDD switches 102-1 to 102-K are connected to communication control circuit 110 shown in FIG. Further, the transmission signal processing circuits 141-1 to 141-L are connected to the transmission / reception weight calculation unit 120a shown in FIG. Here, the antenna elements 101-1 to 101-K and the TDD switches 102-1 to 102-K are used in common with the configuration related to the uplink (reception side). Actually, in the base station apparatus 10, the configuration related to the uplink and the configuration related to the downlink operate integrally, but for the convenience of description, they are described separately.

  Each of the transmission signal processing circuits 141-1 to 141-L is associated with a destination terminal device that transmits data using spatial multiplexing, and performs signal processing on data to be transmitted to the associated terminal device. Do. Specifically, the transmission signal processing circuits 141-1 to 141-L receive the OFDM (A) modulation scheme or SC-FDE when the data inputs # 1 to #L to be transmitted are input from the MAC layer processing circuit 180. The predetermined transmission process in is performed. Data inputs # 1 to #L are data corresponding to each destination terminal device or spatially multiplexed signal sequence, and are input to transmission signal processing circuits 141-1 to 141-L associated with the destination terminal device. The

  When the base station apparatus 10 uses the OFDM (A) modulation scheme, the transmission signal processing circuits 141-1 to 141-L perform signal modulation processing for each subcarrier after error correction coding. The transmission signal processing circuits 141-1 to 141 -L are antenna elements 101-1 to 101-1 corresponding to the destination terminal devices assigned to the respective transmission weights stored in the multiuser MIMO transmission weight storage circuit 128 a. 101-K transmission weights are read from the multi-user MIMO transmission weight storage circuit 128a, and the modulated transmission signal is multiplied by the read transmission weight for each subcarrier.

  In addition, when SC-FDE is used in the base station apparatus 10, the transmission signal processing circuits 141-1 to 141-L each transmit a signal subjected to single carrier modulation processing to each block of the transmission signal by FFT. Separate into components. The transmission signal processing circuits 141-1 to 141-L read the transmission weight corresponding to the destination terminal device out of the transmission weights stored in the multiuser MIMO transmission weight storage circuit 128a, and convert it into a signal separated into frequency components. On the other hand, the read transmission weight is multiplied for each frequency component.

  Further, when SC-FDE is used in the base station apparatus 10, the transmission signal processing circuits 141-1 to 141-L transmit a signal subjected to single carrier modulation processing after error correction coding to a block of the transmission signal. Each frequency component is separated by FFT in units. The transmission signal processing circuits 141-1 to 141-L read the transmission weights corresponding to the combination of the destination terminal devices out of the transmission weights stored in the multiuser MIMO transmission weight storage circuit 128 and separated them into frequency components. The signal is multiplied by the read transmission weight for each frequency component.

  Thereafter, the transmission signal processing circuits 141-1 to 141-L add the signal of each frequency component for each antenna element multiplied by the transmission weight, regardless of which of the OFDM (A) modulation scheme and SC-FDE is used. The data is output to the synthesis circuits 142-1 to 142-K. Addition synthesis circuits 142-1 to 142-K synthesize the signals generated by transmission signal processing circuits 141-1 to 141-L for each frequency component, and output them to IFFT & GI adding circuits 143-1 to 143-K. IFFT & GI adding circuits 143-1 to 143-K perform IFFT processing on the signals synthesized in the adder / synthesizing circuits 142-1 to 142-K, convert the signals from the frequency axis to the signals on the time axis, and further guard intervals , And waveform shaping as necessary to generate a digital baseband signal to be transmitted and output it to the D / A converters 144-1 to 144-K. The digital baseband signal corresponds to each of the antenna elements 101-1 to 101 -K and is individually signal processed.

The D / A converters 144-1 to 144-K convert the signals input from the IFFT & GI adding circuits 143-1 to 143-K into analog signals and output the analog signals to the mixers 146-1 to 146-K.
The local oscillator 145 outputs a local oscillation signal used for up-conversion and having a predetermined frequency to the mixers 146-1 to 146-K.
The mixers 146-1 to 146-K multiply the analog signals input from the D / A converters 144-1 to 144-K by the local oscillation signals input from the local oscillator 145 and up-convert them to radio frequencies. The signal is output to the filters 147-1 to 147-K. The local oscillation signals input to the mixers 146-1 to 146-K are the same signal, and local oscillation signals having the same frequency and phase are input to the mixers 146-1 to 146-K.

Filters 147-1 to 147 -K remove signals outside the band of the channel to be transmitted included in the signals input from mixers 146-1 to 146 -K and provide high-power amplifiers 148-1 to 148 -K. Output.
The high power amplifiers 148-1 to 148-K amplify the signals input from the filters 147-1 to 147-K, and the antenna elements 101-1 to 101-K via the TDD switches 102-1 to 102-K. Send more.
The communication control circuit 110 further controls transmission timing, destination terminal device management, and switching of the TDD switches 102-1 to 102-K.

  In the above description, the case where the signal processing performed in the IFFT & GI adding circuits 143-1 to 143-K is processed in the subsequent stage of the adding / combining circuits 142-1 to 142-K has been described. However, the IFFT & GI adding circuit 143 has been described. The signal processing performed in -1 to 143-K is performed in the transmission signal processing circuits 141-1 to 141-L to obtain digital sampling data on the time axis, and the sampling data at each sampling time is added and synthesized Even if it replaces with the process of carrying out addition synthesis over all the destination terminal stations in 142-1 to 142-K, equivalent signal processing is possible, and either configuration may be selected. However, in this case, the number of times IFFT is performed is larger than that described above, and therefore the configuration of FIG. 21 is preferable from the viewpoint of overall circuit scale suppression.

<Transmission / reception weight calculation process from channel estimation>
Hereinafter, processing from channel estimation to calculation of transmission weights and reception weights in the base station apparatus 10 of this configuration example will be described with reference to FIGS. 22 to 27. These series of processes are basically performed before starting communication with the terminal device, but once these processes are performed, the accuracy of channel information is improved while performing sequential learning, that is, transmission weight and It is also possible to improve the accuracy of the reception weight.
In addition, the base station apparatus 10 is intended for use in a broadband service and is intended for communication with a certain bandwidth. For this reason, assuming that an OFDM (A) modulation method or a communication method such as SC-FDE is used, an explanation is given of performing signal processing by separating each frequency component in units of blocks.

In uplink channel estimation, for example, a training signal as shown in FIG. 15 is continuously transmitted from the terminal device, and is received by the base station device 10 and is averaged in a relatively short time (see FIG. 15). 22) is performed. Further, the base station apparatus 10 acquires a relative component indicating a difference in relative channel information of each of the antenna elements 101-1 to 101-K (FIG. 23), and performs an averaging process for a long time (FIG. 24). Three-stage signal processing is performed.
The uplink channel information thus obtained is multiplied by a calibration coefficient to obtain downlink channel information (FIG. 25), and transmission weights and reception weights are obtained based on the uplink and downlink channel information. Is calculated (FIG. 26).
Hereinafter, each processing will be described.

FIG. 22 is a flowchart showing short-time averaging processing for acquiring uplink channel information in the first configuration example of the technology related to the present invention.
In the base station device 10, the channel information short-time averaging circuit 121 starts receiving a training signal for channel estimation for short-time averaging from the terminal device (step S 101), and sets m and M as sampling counters Reset to zero (step S102). Here, the counter means m and M in Equation (10), and means the m-th sample of the M-th symbol. The channel information short-time averaging circuit 121 samples the training signals input from the FFT circuits 108-1 to 108-K, and sets the sampled signal as S ^(M) _m (step S103).

The channel information short-time averaging circuit 121 adds “1” to the counter m every time the sampling period Δt elapses (step S104), and checks whether the counter m matches the number of data N (m = N). When the determination is made (step S105) and the counter m does not match the number of data N (m ≠ N) (step S105: NO), the process returns to step S103, and steps S103 to S105 are repeated. Here, the data number N is the number of samples per symbol, and is a predetermined value.
On the other hand, when the counter m coincides with the number of data N (step S105: YES), the channel information short-time averaging circuit 121 considers that sampling for one symbol has been completed, and in order to sample the next symbol, the counter m 0 is substituted into “1”, and “1” is added to the counter M (step S106).

The channel information short-time averaging circuit 121 determines whether or not the sampling is finished depending on whether or not the counter M has reached a predetermined value (M _{0 in} Expression (10)) (step S107), and a series of samplings Is not completed (step S107: NO), the process returns to step S103, and the processes of steps S103 to S106 are repeated. Here, the sampling of a series is that the predetermined sampling number of symbols M _0.
On the other hand, when a series of sampling is completed (step S107: YES), the channel information short-time averaging circuit 121 calculates ^ S (M, M ′) using equation (12) (step S108). A frequency error Δf that maximizes the solution of (13) or equation (11) is calculated (step S109).

Channel information short-time average circuit 121, using the calculated frequency error Delta] f, to calculate the sampled data to S _m of the addition averaging over from equation (10) a plurality of cycles (step S110).
Channel information short-time average circuit 121, a short time performs an FFT on averaged sampled data to S _m, calculates information of each frequency component (step S111), and terminates the processing in the short-term average (step S112).
Note that if it is known in advance that the frequency error Δf is negligibly small (when such a setting is used in the design), or the time for performing short-time averaging (T × M ₀ ) is If it is set sufficiently short, it is possible to omit step S108 and step S109 corresponding to the correction of the frequency error Δf and directly execute step S110 with Δf = 0.

  The channel information short-time averaging circuit 121 separates training signals over a plurality of periods for each of the antenna elements 101-1 to 101 -K and synthesizes the separated training signals for a short-time averaging process. I do. Furthermore, each frequency component signal having a different period included in the signal received by each antenna element 101-1 to 101 -K is separated for each frequency component by FFT, and each antenna element is separated from the separated signal for each frequency component. Channel information of each frequency component in the uplink between 101-1 to 101-K and the terminal device is acquired.

FIG. 23 is a flowchart showing a relative component acquisition process for acquiring a relative component of uplink channel information in the first configuration example of the technology related to the present invention. The relative component acquisition circuit 122 completes the short-time averaging process for the signals corresponding to the first antenna element 101-1 to the K-th antenna element 101 -K by the channel information short-time average circuit 121 (step S 1). S121-1 to S121-K), k-th frequency components ^ h ^(k) ₁ ,..., ^ H ^{(k )} in the channel information corresponding to the antenna elements 101-1 to 101-K for which the short-time averaging processing has been completed. ⁾ _K is input from the channel information a short time average circuit 121 (step S122-1~122-K).

The relative component acquisition circuit 122 ^calculates the offset value e ^{−jφ (k} from the channel information (^ h ^(k) ₁ ) in the first antenna element 101-1 and its complex conjugate (^ h ^(k) ₁ ) ^{*. )} (= (^ H ^(k) ₁ ) ^* / ‖ ^ h ^(k) ₁ ‖) is calculated (step S123). Here, “‖x‖” represents the absolute value of x. This offset value is obtained individually for each frequency component.
The relative component acquisition circuit 122 uses the calculated offset value e ^{−jφ (k)} for the kth frequency component as the kth frequency component ^ h ^(k) ₁ ,. multiplying the h ^{_{(k) K} (step} S124-1~S124-K), the channel information ^{to h} _(k) 1 showing the relative complex phase relationships, seek ... ^{~h _{(k) K,}} the processing is terminated (Steps S125-1 to S125-K).

As described above, the relative component acquisition circuit 122 uses the channel information of the first antenna element 101-1 as a reference, and the relative channel information of the antenna elements 101-1 to 101-K to h ^(k) ₁ , ... ~ h ^(k) _K is calculated. The relative component acquisition circuit 122 performs the processing from step S121-1 to step S125-K for all frequency components for each terminal device, and performs a short-time average channel for all frequency components for each terminal device. Relative components of information ~ h ^(k) ₁ , ..., ~ h ^(k) _K are calculated.

  FIG. 24 is a flowchart showing long-time averaging processing of uplink channel information in the first configuration example of the technology related to the present invention. Each of the processes shown in FIGS. 22 and 23 is performed a plurality of times in continuous or discrete time, and long-time averaging is performed in the long-time averaging process based on the short-time average channel information calculated in each process. Calculated channel information is calculated.

When the channel information long-time averaging circuit 123 completes the first to Q-time short-time averaging processing (including relative component acquisition) (steps S131-1 to S131-Q), the short-time average is calculated from the relative component acquisition circuit 122. , H ^(k) ₁ ^[q] ,..., H ^(k) _K ^[q] (q = 1,..., Q) are input (steps S132-1 to S132-Q). . Here, the relative component of short-term average channel information to h ^(k) ₁ ^[q] is the relative component of the channel information with respect to the k-th frequency component of the first antenna element 101-1 calculated for the qth time. . Therefore, steps S132-1 to S132-Q correspond to processing performed at different timings. Note that the number Q of times subjected to long-time averaging processing is determined in advance based on the environment in which the wireless communication system is operated.

Further, the channel information long-time average circuit 123 calculates long-time average channel information h ^(k) _i (i = 1,..., K) using the following equation (18) (step S133). Since steps S132-1 to S132-Q are completed at different timings, the relative component of this channel information is temporarily stored in the memory until step S133, which is a long-time averaging process, is performed. You may memorize | store and implement step S133 at once. Alternatively, the individual summation processing by Σ in step S133 is performed every time the individual processing in steps S132-1 to S132-Q is completed, and is stored in the memory until the next processing. You may read them each time and may implement step S133.

The channel information long-time average circuit 123 calculates long-time average channel information h ^(k) _i ^obtained by averaging the channel information of each frequency component for each antenna element 101-1 to 101 -K. The averaging process ends (step S134). The reception weight calculation described with reference to FIG. 26 described later is performed using the uplink channel information acquired here. However, when these processes are performed, the reception weight is temporarily stored in the memory. You can leave it.

Through the above processing, uplink channel information can be acquired directly. In this configuration example, since the relative component acquisition process (FIG. 23) is performed, the long-time average channel information is obtained without being affected by the phase shift in each short-time average process from the first time to the Q-th time. Can be calculated. Note that the subscript k on the right shoulder of the above-described channel information h ^(k) _i represents a number (subcarrier number) for identifying a frequency component.

  FIG. 25 is a flowchart showing a process of acquiring downlink channel information in the first configuration example of the technology related to the present invention. Since it is difficult for the base station apparatus 10 to obtain channel information directly as in the uplink for the downlink from the base station apparatus 10 to the terminal apparatus, the downlink channel is based on the uplink channel information. Estimate information.

In the base station apparatus 10, the calibration circuit 126 receives the uplink channel information h ^(k) _i from the channel information long-time averaging circuit 123 (step S 142), and the calibration circuit 126 performs the first operation on the input channel information h ^(k) _i . The calibration coefficient C ^(k) _i corresponding to the k-th frequency component in the antenna element 101-i of _i is read from the calibration coefficient storage circuit 129 (step S143).
The calibration circuit 126 multiplies the input channel information h ^(k) _i by the read calibration coefficient C ^(k) _i (step S144), and uses the multiplication result as downlink channel information to complete the process. (Step S145). Also in this case, the calculation of the transmission weight described with reference to FIG. 26 described later is performed using the downlink channel information acquired here, but is temporarily stored in the memory when performing these processes. It does not matter.
The calibration circuit 126 performs the processing from step S142 to step S144 described above for each frequency component for each of the antenna elements 101-1 to 101-K.

  FIG. 26 is a flowchart showing a process of calculating a transmission weight and a reception weight in the base station apparatus 10 according to the first configuration example of the technology related to the present invention. Since the processing for calculating the reception weight for the channel information in the uplink and the processing for calculating the transmission weight for the channel information in the downlink are the same, the processing for calculating the transmission weight in the downlink will be described here. A specific description of the process of calculating the reception weight will be omitted.

  When the multi-user MIMO transmission weight calculation circuit 127a starts processing (step S151), a combination pattern of terminal devices serving as destinations for simultaneously transmitting data using spatial multiplexing is input (step S152). This combination pattern of terminal devices is a pattern determined in advance using some method. For example, all combinations of terminal devices included in a wireless communication system, limited combinations frequently used, predetermined A combination that satisfies the above condition may be used. It is assumed that all terminal devices always belong to one or more combinations. Also, in the case where band allocation by different combinations for each subcarrier is possible as in OFDMA, it is not necessary to prepare the same combination of variations for all frequency components, and different variations may be used for each frequency component. Absent. In this case, if any frequency component belongs to one or more combinations, there is no operational problem even if there is no combination belonging to all frequency components.

The multi-user MIMO transmission weight calculation circuit 127a acquires, from the calibration circuit 126, downlink channel information corresponding to the terminal device included in the combination pattern for each input combination pattern #s (1 ≦ s ≦ S). (Steps S153-1 to S153-S), transmission weights corresponding to the combination patterns are calculated (Steps S154-1 to S154-S). The multiuser MIMO transmission weight calculation circuit 127a stores the calculated transmission weights in the multiuser MIMO transmission weight storage circuit 128a in association with the respective combination patterns (steps S1555-1 to S155-S).
The multiuser MIMO transmission weight calculation circuit 127a performs the processing from steps S153-1 to S153-S to steps S1555-1 to S155-S for all frequency components.

In addition, as a method of determining the combination pattern of terminal devices that perform spatial multiplexing simultaneously, if the number of terminal devices is relatively small, the number of combinations for extracting N terminal devices that are the number of spatial multiplexing from the entire terminal device is prepared. May be. However, the combination of selecting N pieces from the number M of all terminals has only _M C _N, the number of combinations the value of "M" and "N" is large becomes astronomical huge number. In that case, for example, several patterns appropriately divided into N combinations from all terminal devices are carefully selected, and each terminal device always belongs to one of the patterns. As a method of careful selection, for example, a large number of combinations for randomly selecting N may be created, and each terminal device may be selected so as to belong to as many patterns as possible. The operation may be performed using some program, or a combination may be artificially set. In this case, some rules may be set, or the operation may be performed by an appropriate (random) operation. The processing from step S151-1 to S151-S to step S1555-1 to S155-S can be performed without being affected by the method of creating the combination pattern.

  Further, a description of the transmission weight calculation process (steps S154-1 to S154-S) for the combination pattern of terminal devices will be added. The transmission weight here may use the same method as the calculation of the transmission weight in general multiuser MIMO, that is, a known calculation method. For example, the calculation method shown in FIG. 7 may be used. In the method shown in FIG. 7, specific calculations are shown as Expressions (6) to (9). In addition, as a similar calculation method of the transmission weight, there are the following methods, and any method may be used.

First, regarding the downlink transmission weight corresponding to the transmission of the base station apparatus 10, the pseudo inverse matrix represented by the following equation (19) is used for the entire channel matrix H ^[all] shown in equation (1) and the like. May be calculated and used as a transmission weight.

Here, when the number of spatially multiplexed terminal apparatuses is N and the number of antenna elements of the base station apparatus 10 is K (N <K), the size of the channel matrix H ^[all] is N × K (N rows and K columns). ). If the rank of H ^[all] is N, the matrix H ^[all] · H ^{[all] H has} a size of N × N and an inverse matrix exists, and a pseudo inverse matrix can be obtained using equation (19). it can. In general, if the value of K is sufficiently redundant with respect to N, the rank of this N × N matrix is stably N, and the inverse matrix exists stably.

  Similarly, for the uplink reception weight corresponding to reception by the base station apparatus 10, a pseudo inverse matrix expressed by the following equation (20) may be calculated and used as the reception weight.

In the uplink case, the size of the channel matrix H ^[all] is K × N (K rows and N columns), and the size of the matrix H ^{[all] H} · H ^[all] is also N × N and generally has an inverse matrix. However, equation (20) may be used as the reception weight.

In the MMSE weight known as a similar transmission / reception weight, if the noise power is σ ² , the following equations (21-1) and (21-2) are expressed by equations (19) and (20). It may be used instead.

  Further, since the channel matrix is different between the downlink corresponding to transmission from the base station apparatus 10 and the uplink corresponding to reception, it is necessary to separately calculate the transmission weight and the reception weight. Note that “I” in Equation (21-1) and Equation (21-2) is a unit matrix of N × N (N rows and N columns).

As described above, the weights are calculated by the processes in steps S154-1 to S154-S, and the weights calculated by the processes in steps S1555-1 to S155-S are stored in the multiuser MIMO transmission weight storage circuit 128a.
In the transmission / reception weight, the direction indicated by a vector (weight vector) having the weight value for each antenna element as a component of each element has an effective meaning. For this reason, a vector obtained by multiplying a certain weight vector by a predetermined coefficient is the same in direction, so that a weight vector multiplied by a constant coefficient for each antenna element is all equivalent without depending on the coefficient to be multiplied. is there. That is, the weights obtained by multiplying the entire components of each row vector or column vector (transmission / reception weight vector obtained by a conventionally known technique) of the matrix given by Expression (19) to Expression (21-2) and the like by a common coefficient. Are all equivalent to the weights in the technology related to the present invention.

  The above description related to FIG. 26 was related to the calculation of the transmission weight, but the circuit corresponding to the reception weight (for example, the channel information long-time averaging circuit 123, the multi-user MIMO transmission weight calculation circuit with respect to the calibration circuit 126). 127a is replaced with a multiuser MIMO reception weight calculation circuit 124a, and a multiuser MIMO reception weight storage circuit 128a is replaced with a multiuser MIMO reception weight storage circuit 125a), thereby performing reception weight calculation processing. Can be implemented.

  The above-described processing shown in FIGS. 22 to 26 is performed in advance, and the transmission weights and reception weights obtained there are stored in the multiuser MIMO transmission weight storage circuit 128a and the multiuser MIMO reception weight storage circuit 125a. Note that the processes shown in FIGS. 22 and 26 can be executed after temporarily suspending communication at an appropriate cycle even after the start of communication. The processing shown in FIGS. 24 to 26 may be performed using the short-time average channel information obtained there, and the transmission weight and the reception weight may be updated sequentially.

<Transmission process>
Next, signal transmission processing in the base station apparatus 10 will be described with reference to the drawings.
FIG. 27 is a flowchart showing a transmission process of the base station apparatus 10 in the first configuration example of the technology related to the present invention. As described above, here, a case where the OFDM (A) modulation method or SC-FDE is used will be described.
When the base station device 10 starts transmission processing (step S161), the communication control circuit 110 or the scheduling processing circuit 181 selects a terminal device to be spatially multiplexed using a known technique (step S162). It should be noted that here, a detailed description of a method for selecting a terminal device that performs spatial multiplexing simultaneously, that is, a scheduling method is omitted. The transmission signal processing circuits 141-1 to 141-L generate transmission signals of each frequency component from the input data inputs # 1 to #L (step S163).

  The processing performed by the transmission signal processing circuits 141-1 to 141-L in step S163 is necessary for a bit string constituting a wireless packet subjected to MAC layer signal processing, for example, when the OFDM (A) modulation method is used. Depending on the error, encoding processing for error correction, overhead information (preamble signal) including timing detection signals, channel estimation signals, and the like are applied, and bits are divided for each subcarrier and a predetermined modulation scheme (for example, BPSK) , QPSK, 16QAM, etc.).

  In addition, when SC-FDE is used, transmission signal processing circuits 141-1 to 141-L are necessary for a bit string constituting a radio packet subjected to MAC layer signal processing in the same manner as the OFDM (A) modulation method. In accordance with the above, an encoding process for error correction, overhead information (preamble signal) including a timing detection signal, a channel estimation signal, and the like are applied, and a predetermined modulation scheme (for example, BPSK, QPSK, 16QAM, etc.) Single-carrier transmission signal processing such as signal point mapping processing is performed, and FFT is performed in units of blocks in order to perform transmission weight multiplication processing on the frequency axis, thereby generating each frequency component of the transmission signal.

The transmission signal processing circuits 141-1 to 141-L correspond to the terminal devices assigned to the own circuit among the transmission weights corresponding to the combinations of the terminal devices selected by the communication control circuit 110. The transmission weights of the respective frequency components 1-101-K are read out from the multiuser MIMO transmission weight storage circuit 128a (step S164).
The transmission signal processing circuits 141-1 to 141-L respectively transmit the transmission signals separated in the frequency components generated in step S163 for each transmission signal transmitted by the antenna elements 101-1 to 101-K, and step S164. Is multiplied by the transmission weight of each frequency component read out in step S165, and the result is output to the adder / synthesizer circuit 142-1 to 142-K (step S165).

  Each of the adder / synthesizers 142-1 to 142-K adds and synthesizes the signals input from the transmission signal processing circuits 141-1 to 141-L, and the IFFT & GI adding circuits 143-1 to 143-K. A signal on the frequency axis is converted to a signal on the time axis, and a guard interval is given, and a series of processing such as waveform shaping is performed as necessary (steps S166-1 to S166-K). For each of these signals, D / A conversion of sampling data by the D / A converters 144-1 to 144-K, up-conversion to a radio frequency by the mixers 146-1 to 146-K, and filters 147-1 to 147-1 147-K removes the out-of-band signal, performs amplification by the high power amplifiers 148-1 to 148-K, and transmits from each of the antenna elements 101-1 to 101-K (steps S167-1 to S167-K). The transmission process ends (steps S168-1 to S168-K).

These series of processes (from step S163 to steps S167-1 to S167-K), when a wireless packet spans a plurality of symbols or a plurality of blocks, the processing in units of blocks of OFDM symbols or SC-FDEs is the number of symbols. The transmission signal processing of the entire wireless packet is performed by continuing the execution for the number of blocks.
Further, as described with reference to FIG. 26, the transmission weights are not necessarily the combinations of terminal devices managed by the multi-user MIMO transmission weight storage circuit 128a. In this case, in the terminal device combination selection performed by the scheduling processing circuit 181 in step S162, a terminal corresponding to the combination pattern of the terminal devices managed by the multiuser MIMO transmission weight storage circuit 128a is selected.

  A characteristic of the transmission processing of the base station apparatus 10 in this configuration example is that the transmission weight corresponding to the combination of terminal apparatuses stored in the multiuser MIMO transmission weight storage circuit 128a in step S164 is read and used. It is to use transmission weights calculated in advance for each combination of terminal devices without being aware of channel information that changes slightly every moment. Thereby, it is possible to perform transmission processing without calculating a transmission weight each time transmission is performed.

  In addition, the signals transmitted in this manner are substantially the same in phase (strictly) for each frequency component of the signals transmitted from the antenna elements 101-1 to 101-K of the base station apparatus 10 in the antenna elements of each terminal apparatus. Is slightly deviated from perfect in-phase synthesis due to null control for avoiding interference and giving interference). The signal received at each terminal device can be processed as a normal signal that can be received without being conscious of various signal processing performed by the base station device 10 in particular.

  In addition, in the provision of overhead information (preamble signal) such as a channel estimation signal performed by the transmission signal processing circuits 141-1 to 141-L in step S163, signals having a common pattern are used for a plurality of terminal devices. It is possible. This is because transmission by the transmission weight performed in step S165 allows each terminal device to receive a signal destined for another terminal device in a sufficiently suppressed state. Therefore, it is necessary to assign a separate preamble signal to each terminal device. Because there is no. As a result, it is not necessary to assign a preamble signal having the number of symbols depending on the number of spatial multiplexing while performing high-order spatial multiplexing, and it is possible to suppress a decrease in MAC layer efficiency.

<Reception processing>
FIG. 28 is a flowchart showing a reception process of the base station device 10 in the first configuration example of the technology related to the present invention. The signal transmitted by the terminal device is transmitted as a normal signal without being conscious of various signal processing performed by the base station device 10 in this configuration example. Here, although details of a method for selecting a terminal device that performs spatial multiplexing simultaneously, that is, a scheduling method, are omitted, the MAC layer processing circuit 180 selects a terminal device that performs spatial multiplexing and transmits data using a known technique.

  In the base station apparatus 10, when the reception process is started (step S171), the communication control circuit 110 selects a combination of terminal apparatuses that perform spatial multiplexing and transmit data (step S172), and sets scheduling contents related to the uplink. The selected terminal device is notified (step S173). For example, in the case of a system such as WiMAX that employs base station centralized control using a TDMA frame, the notification method here is an UL-MAP (uplink allocation map) at the head of the frame, which is assigned to The carrier number, time slot (OFDM symbol position), and the continuous time (number of OFDM symbols) are notified. Of course, the assignment may be notified to the terminal device by other methods, and step S173 of notifying the terminal device side may be omitted depending on the access control method. Note that the combination of terminal devices in step S172 may be a combination of terminal devices instructed by a higher-level device.

In accordance with the processing of step S173, out of the reception weights corresponding to the combination of the terminal devices of the transmission source, the frequency components of the respective antenna elements 101-1 to 101-K corresponding to the terminal devices assigned to the own circuit. The reception weight is read from the multiuser MIMO reception weight storage circuit 125a (step S176).
In parallel with this, signals are received via the respective antenna elements 101-1 to 101-K from a predetermined timing when the assignment instruction (transmission instruction to the terminal device) is given (steps S174-1 to S174-K). . Here, reception includes processing for performing analog / digital conversion on a received signal or a signal obtained by down-converting the received signal. Thereafter, the FFT circuits 108-1 to 108-K extract signals in symbol units, remove guard intervals, perform FFT processing, and convert signals on the time axis into signals on the frequency axis. The received signal processing is performed (steps S175-1 to S175-K).

  Further, the reception signal processing circuits 109-1 to 109-L multiply the reception weight read from the multiuser MIMO reception weight storage circuit 125a by the reception signal separated for each frequency component (steps S177-1 to S177-). K). The reception signal processing circuits 109-1 to 109-L add and synthesize the multiplication results for each transmission source terminal device (steps S178-1 to S178-L). The processing in steps S177-1 to S177-K and steps S178-1 to S178-L corresponds to the operation of multiplying the reception signal vector by the reception weight matrix as a whole.

Reception signal processing circuits 109-1 to 109-L perform predetermined reception signal processing on each signal series (data output # 1 to #L) thus separated (step S179-1 to S179-1). S179-L), a series of processing ends (steps S180-1 to S180-L).
Here, the predetermined received signal processing is processing after signal separation of a spatially multiplexed signal. Therefore, the signal processing is the same as that of normal SISO communication.

  The received signal processing includes demodulation processing for each subcarrier when the OFDM (A) modulation method is used, and when SC-FDE is used, the frequency axis for the received signal of each frequency component It includes a single carrier demodulation process for a signal obtained by performing the above signal equalization process and synthesizing the signal by IFFT process. Furthermore, a decoding process for error correction may be performed as necessary. Naturally, signal processing such as a MAC layer is also performed after the above processing, but it is omitted here because it is the same as the processing by a known technique.

  Further, as described in the transmission process, the combination pattern of the terminal devices managed by the multiuser MIMO reception weight storage circuit 125a does not necessarily include all combinations of the terminal devices. In the terminal device combination selection performed by the processing circuit 181 in step S172, a terminal corresponding to the terminal device combination pattern managed by the multi-user MIMO reception weight storage circuit 125a is selected.

  Regarding the symbol timing, when the reception level of the reception signal at each of the antenna elements 101-1 to 101-K is very weak, it may be difficult to detect the timing from the reception signal. In this case, for example, in addition to absolute time synchronization using GPS, a subsequent frame using a periodic frame configuration with reference to the timing obtained from the immediately preceding frame timing detection signal, etc. The reception timing and symbol timing of the received signal may be determined using any synchronization means such as estimating the reception timing of the received signal. At this time, when determining the transmission timing, the terminal device may transmit a signal at a predetermined timing in accordance with an instruction from the base station based on the synchronized reception timing.

  As described above, in the wireless communication system of this configuration example, both the base station apparatus 10 and the terminal apparatus are installed at relatively high places, and as a result, stable from a line-of-sight or a fixed huge building or the like. Channel information corresponding to a stable incident wave component given by combining a line-of-sight wave and a stable reflected wave is acquired in an environment where a reflected wave is expected between the base station apparatus and the terminal apparatus. The base station device 10 generates a transmission weight and a reception weight based on the acquired channel information, and realizes in-phase synthesis of signals in the base station device and the terminal device by using the reception weight. Further, by transmitting signals from the plurality of antenna elements 101-1 to 101 -K using the transmission weight generated by the base station apparatus 10, the terminal apparatus aligns the phases of the signals synthesized on the propagation path with high accuracy. Received as a received signal.

Moreover, in the acquisition of channel information in the base station apparatus 10, the estimation accuracy of channel estimation is improved by performing short-time averaging. Further, by combining a plurality of reception signals at discrete times received via the antenna elements 101-1 to 101-K, random signals in the reception signals received via the antenna elements 101-1 to 101-K can be obtained. The relative ratio of the time variation component to the stable component can be statistically suppressed, and the influence of the time variation can be reduced.
As a result, even in an environment where the line gain by each of the antenna elements 101-1 to 101-K is insufficient such that acquisition of channel information between the terminal device and the antenna elements 101-1 to 101-K is difficult. Thus, it is possible to calculate transmission weights in which the signals transmitted from the antenna elements 101-1 to 101-K are combined in phase in the terminal device.

  Further, in the feedback of channel information related to the calculation of transmission weights and reception weights, it is possible to avoid a decrease in transmission efficiency due to overhead for channel estimation, which is a problem when real-time channel information feedback is frequently performed. . Actually, if the average channel information is acquired for a long time before the service is started, it is possible to operate without performing any channel information feedback during the data communication service. Furthermore, since the calculation load associated with the calculation of the transmission weight and the reception weight, which has been performed sequentially in the past, only needs to be calculated once at the start of operation of the wireless communication system, the load during normal operation can be reduced. It is also possible to plan. The calculation of these transmission weights and reception weights is often required to complete arithmetic processing in a short time in the conventional technology premised on real-time processing. For this reason, hardware processing capable of high-speed arithmetic is assumed. There were many. However, although the conventional technique has a problem that the hardware scale increases due to the enormous amount of computation, the base station apparatus 10 is allowed to perform computation processing over time at the start of operation of the wireless communication system. Therefore, even software processing with a slow calculation processing time can be dealt with, and a secondary effect of reducing the overall hardware scale can be obtained.

Thus, by performing transmission / reception using K antenna elements (wireless modules) using the transmission weight and reception weight described above, the maximum transmission power is ₁₀ Log ₁₀ K [dB] at a maximum under a certain condition. Can be obtained. As a result, it is possible to obtain an energy saving effect that reduces the total transmission power, an economic effect that allows an inexpensive amplifier to be used instead of a high output, high linearity, high gain amplifier, and the like. At the same time, by performing spatial multiplexing transmission of L signal series on the same frequency component at the same time, it is possible to increase the transmission capacity, that is, improve the frequency utilization efficiency.

Also, the receiving unit 85 of the base station apparatus 80 shown in FIG. 10 sends a preamble signal or the like used for channel estimation from the A / D converters 856-1 to 856-K to the channel information estimation circuit 861 every time a signal is received. The channel information estimation circuit 861 performs channel estimation of the received signal. The result of channel estimation is input to multi-user MIMO reception weight calculation circuit 862, where reception weights are calculated from a predetermined calculation process for the MIMO channel matrix, and are received by reception signal processing circuits 858-1 to 858-L. The received signal is detected based on the output. In the most typical processing example, an inverse matrix or pseudo inverse matrix of a MIMO channel matrix is calculated as a reception weight, but of course, if the destination terminal device combination is different, the channel information (channel vector) of the terminal device to which attention is paid. Even if the same vector is selected, the row vectors corresponding to the corresponding terminal devices constituting the reception weight matrix obtained by the inverse matrix calculation are different when the other channel vectors are different, and the reception weights are different.
On the other hand, every time a base station apparatus 10 in this configuration example receives a signal, it performs channel estimation for the purpose of acquiring information for generating a reception weight for separating a spatially multiplexed signal sequence, and performs estimation. It is not necessary to calculate the reception weight from the result. This point is essentially different from the receiving unit 85 of the base station apparatus 80 shown in FIG. In particular, the base station apparatus 10 uses reception weights calculated in advance based on channel information corresponding to a large number of antenna elements 101-1 to 101-K subjected to long-time averaging and combinations of terminal apparatuses.

The reception weight is calculated every time a signal is received in the conventional multi-user MIMO technique at a level where the intensity of the interference signal from each of the spatially multiplexed signal sequences (data inputs # 1 to #L) cannot be ignored. This is because, in order to suppress this, it is necessary to calculate a coefficient for canceling a signal of another terminal device at the time of synthesis as null control. The signal processing for signal suppression is the calculation of the (pseudo) inverse matrix shown above, but the calculation load increases in proportion to the cube of the matrix size, and real-time processing tends to be impossible. .
On the other hand, since the base station device 10 in this configuration example only needs to read the reception weight corresponding to the combination and perform the reception signal processing when the combination of the terminal devices is determined, the real-time processing regarding the reception weight calculation calculation (That is, the reception weight need only be a simple read from the memory), and can be realized with a realistic hardware configuration even if a huge number of antenna elements are assumed.

  Further, every time a signal is received in the multi-user MIMO technology, the base station apparatus 80 needs to perform channel estimation individually for each terminal apparatus that is simultaneously spatially multiplexed, and uses the result to calculate the reception weight. It was. For this individual channel estimation, the same number of orthogonal preambles as the spatial multiplexing number or the preamble signal of the channel estimation signal having the same number of symbols as the spatial multiplexing number is required. In this case, overhead corresponding to the number of symbols equal to the number of spatial multiplexing occurs, leading to a decrease in MAC layer efficiency.

  However, in the technology related to the present invention, the reception weight is multiplied by the reception signal in steps S177-1 to S177-K without performing channel estimation for each spatially multiplexed terminal device, and as a result, the transmission source terminal Signal separation is performed for each device. Thereby, in the received signal processing for each signal sequence performed in steps S179-1 to S179-L, a single channel estimation preamble signal (for example, one symbol) is as if the spatial multiplexing number is one. If there is, signal detection processing can be performed. As a result, it is not necessary to assign a preamble signal having the number of symbols depending on the number of spatial multiplexing while performing high-order spatial multiplexing, and it is possible to suppress a decrease in MAC layer efficiency.

Further, when the number of antenna elements 101-1 to 101-K provided in the base station apparatus 10 is a very large number, even if there is a time variation of the channel of each transmitting antenna element versus receiving antenna element, If it changes randomly between a large number of antenna elements, it can be regarded as a statistically averaged state as a whole, and the influence of time fluctuation can be reduced.
Also, by storing transmission weights calculated for each combination of terminal devices in the multi-user MIMO transmission weight storage circuit 128a in advance, when transmitting data in a spatially multiplexed manner, transmission weights corresponding to the combination of terminal devices are stored. Can be transmitted in a spatially multiplexed manner without calculating a transmission weight each time transmission is performed, and a high-order space using a huge number of antenna elements with simple processing. Multiplexing can be realized, and as a result, the frequency utilization efficiency of the downlink can be improved.

(Second configuration example of related technology)
In the first configuration example of the related art described above, the base station apparatus 10 performs directivity control with null control. However, in this configuration example, interference and interference are prevented by directivity control without null control. Repress. In this configuration example, the difference from the first configuration example will be mainly described.
In the description of the first configuration example described above, the configuration examples of the transmission unit 140 and the reception unit 100 related to FIGS. 18 to 19 and FIG. 21 and the flowcharts related to channel information acquisition illustrated in FIGS. 22 to 25 will be described. However, these are the same in the second configuration example, and there is no change. The main difference is that the transmission / reception weight calculation method has been changed to a method that does not depend on the combination of terminal devices that perform spatial multiplexing transmission at the same time on the same frequency component. Relatedly, FIG. 20 showing a configuration example of the transmission / reception weight calculation unit 120a, FIG. 26 showing the calculation processing of the transmission weight and the reception weight, FIG. 27 showing the flowchart of the transmission processing, and FIG. 28 showing the flowchart of the reception processing. However, it has been changed to the corresponding content.

FIG. 29 is a schematic block diagram showing a configuration of the transmission / reception weight calculation unit 120b in the second configuration example of the technology related to the present invention. The base station apparatus in this configuration example has a configuration in which the transmission / reception weight calculation unit 120a is replaced with the transmission / reception weight calculation unit 120b in the base station apparatus 10 (FIGS. 19 and 21) of the first configuration example.
The transmission / reception weight calculation unit 120b shown in the figure differs from the base station apparatus 10 shown in FIG. 19 in that a multiuser MIMO (MU-MIMO) reception weight calculation circuit 124a, a multiuser MIMO (MU-MIMO) reception weight storage circuit. 125a, multi-user MIMO (MU-MIMO) transmission weight calculation circuit 127a, multi-user MIMO (MU-MIMO) transmission weight storage circuit 128a, instead of reception weight calculation circuit 124b, reception weight storage circuit 125b, transmission weight calculation circuit 127b The transmission weight storage circuit 128b is provided.

The reception weight calculation circuit 124b calculates a reception weight for each terminal device based on the channel information output from the channel information long-time averaging circuit 123, and outputs the calculated reception weight to the reception weight storage circuit 125b. The reception weight storage circuit 125b stores the reception weight calculated by the reception weight calculation circuit 124b.
The transmission weight calculation circuit 127b calculates a transmission weight for each terminal device based on the downlink channel information acquired by the calibration circuit 126, and outputs the calculated transmission weight to the transmission weight storage circuit 128b. The transmission weight storage circuit 128b stores the transmission weight calculated by the transmission weight calculation circuit 127b. Note that the transmission weight storage circuit 128b is also a part of the configuration related to the uplink included in the base station apparatus 10.

<Transmission / reception weight calculation process from channel estimation>
The short-time averaging process for acquiring uplink channel information in the base station apparatus 10 of this configuration example is the same as that of the first configuration example shown in FIG.
Also, the relative component acquisition processing for acquiring the relative component of the uplink channel information in the base station apparatus 10 of the present configuration example is the same as that of the first configuration example illustrated in FIG.
Further, the long-time averaging process of uplink channel information in the base station apparatus 10 of this configuration example is the same as the first configuration example shown in FIG.
Further, the process of acquiring downlink channel information in the base station apparatus 10 of the present configuration example is the same as that of the first configuration example shown in FIG.

FIG. 30 is a flowchart showing a process of calculating a transmission weight and a reception weight in the base station apparatus according to the second configuration example of the technology related to the present invention.
When the transmission weight calculation circuit 127b starts processing (step S451), the channel information h ^(k) _i of the k-th frequency component in the i-th antenna element 101-i is input from the calibration circuit 126 (step S452). .

The transmission weight calculation circuit 127b calculates the complex conjugate (h ^(k) _i ) ^* of the channel information h ^(k) _i input from the calibration circuit 126, and calculates the calculated complex conjugate (h ^(k) _i ) ^* . A value obtained by dividing the channel information h ^(k) _{i by} the absolute value is set as a transmission weight w ^(k) _i (step S453). That is, the transmission weight calculation circuit 127b calculates the transmission weight w ^(k) _i using the following equation (22) (step S453).

The transmission weight calculation circuit 127b stores the calculated transmission weight w ^(k) _i in the transmission weight storage circuit 128b (step S454), and ends the process (step S455).
The transmission weight calculation circuit 127b performs the processing from step S452 to step S453 described above for all the terminal devices for each frequency component for each of the antenna elements 101-1 to 101 -K.
The reception weight calculation circuit 124b calculates the reception weight from the channel information h ^(k) _i input from the channel information long-time average circuit 123 by the same operation as the transmission weight calculation circuit 127b, and calculates the calculated reception weight. The data is stored in the reception weight storage circuit 125b.

  In general, as a weight for signal synthesis when signals are received by a plurality of antennas, there may be a large difference in the signal reception level for each antenna due to the influence of fading or the like. In this case, the reception level is low. In order to suppress the influence of noise of the received signal of the antenna element, the maximum ratio combining weight shown below is often used. Therefore, in the present configuration example, the following equation (23) can be used instead of the equation (22).

  The difference between the two weights of Expression (22) and Expression (23) is the difference in whether the coefficient magnitude (absolute value) of the i-th antenna element is slightly different or the same for each antenna element. (23) incorporates an effect of reducing the weight of a signal having a relatively high noise level (ie, a low reception level). However, both are common in that both are adjusted so that the complex phase becomes zero or a constant value after multiplication with channel information averaged for a long time. In a broad sense, equation (23) can also be said to be a kind of weight for in-phase synthesis. In this configuration example, the same effect can be obtained even if other weights are used as long as the complex phase is zero or a constant value after multiplication with the channel information averaged for a long time.

  In general, it is preferable to use the weight of Expression (22) as the transmission weight and the weight of Expression (23) as the reception weight. In this configuration example, it is recommended that the system be installed so that a line of sight can be secured between the base station apparatus and the terminal apparatus. Therefore, unlike a multipath environment where a large number of multiple reflected waves exist, a line-of-sight wave is not generated. Since this is a dominant environment, the difference in reception level for each antenna element is relatively difficult. As a result, the weight obtained by equation (23) is effectively a weight equivalent to equation (22).

  In the transmission / reception weight, the direction indicated by a vector (weight vector) having the weight value for each antenna element as a component of each element has an effective meaning. For this reason, a vector obtained by multiplying a certain weight vector by a predetermined coefficient is the same in direction, so that a weight vector multiplied by a constant coefficient for each antenna element is all equivalent without depending on the coefficient to be multiplied. is there. That is, the weights obtained by multiplying the entire components of the vectors given by the equations (22) and (23) by the common coefficient are all equivalent to the weights in the technology related to the present invention.

  The above description regarding FIG. 30 was related to the calculation of the transmission weight, but the circuit corresponding to the reception weight (for example, the channel information long-time averaging circuit 123 for the calibration circuit 126 and the transmission weight calculation circuit 127b). Thus, the reception weight calculation circuit 124b and the transmission weight storage circuit 128b can be replaced with the reception weight storage circuit 125b), and the reception weight calculation process can be performed by performing the same processing.

  The above-described processing shown in FIG. 22 to FIG. 25 and FIG. 30 is performed in advance, the transmission weight obtained there is stored in the transmission weight storage circuit 128b, and the reception weight is stored in the reception weight storage circuit 125b. . Note that the processing shown in FIGS. 22 and 30 can be executed after temporarily suspending communication at an appropriate cycle even after the communication is started. The processing shown in FIGS. 24 to 25 and FIG. 30 may be performed using the short-time average channel information obtained there, and the transmission weight and the reception weight may be updated sequentially.

<Transmission process>
Next, signal transmission processing in the base station apparatus 10 of the present configuration example will be described with reference to the drawings.
FIG. 31 is a flowchart showing the transmission processing of the base station apparatus 10 in this configuration example. Here, a case where OFDM (OFDMA) modulation or SC-FDE is used will be described. Also, the same numbers are assigned to the processes common to FIG.
Processing other than step S169 is the same as the transmission processing of the first configuration example shown in FIG. The difference between this configuration example and the first configuration example is that the transmission weight read in step S169 does not depend on the combination of terminal devices that are simultaneously spatially multiplexed. A transmission weight specific to each terminal apparatus to be multiplexed is read from the transmission weight storage circuit 128b and used.

  As a feature of the transmission processing of the base station apparatus 10 in this configuration example, in step S169, out of the transmission weights stored in the transmission weight storage circuit 128b, the transmission weight corresponding to the destination terminal apparatus is read and used. In other words, the transmission weight corresponding to each terminal device calculated in advance is used without being conscious of the combination of terminal devices and channel information that changes slightly every moment. That is, the reception weight for spatial multiplexing transmission calculated by the reception weight calculation circuit 124b and the transmission weight for spatial multiplexing transmission calculated by the transmission weight calculation circuit 127b depend on the combination of terminal devices that simultaneously perform spatial multiplexing. It is characterized by a weight that does not.

In the first configuration example, it is necessary to calculate transmission / reception weights for each combination of terminal devices. Here, the combination of selecting N pieces from the number M of all terminals has only _M C _N, the number of the value thereof is large combination of "M" and "N" is become astronomical vast number However, the fact that the transmission weight to be used does not depend on the combination of terminal devices means that all variations of the astronomical number can be dealt with. At the same time, the amount of information to be stored in the transmission weight storage circuit 128b may be only the transmission weight vector of the number of frequency components (for example, the number of subcarriers) × the number of terminal devices accommodated, and even when the number of terminal devices increases, the transmission weight It also has a feature that an increase in the storage capacity of the storage circuit 128b can be suppressed.

  In addition, the signals transmitted in this way are received in the same phase for each frequency component in the antenna elements of each terminal apparatus, and the signals transmitted from the antenna elements 101-1 to 101-K of the base station apparatus 10 are received. Will be. The signal received at each terminal device can be processed as a normal signal that can be received without being conscious of various signal processing performed by the base station device 10 in particular.

<Reception processing>
FIG. 32 is a flowchart showing a reception process of the base station apparatus 10 in the second configuration example of the technology related to the present invention. Also, the same numbers are assigned to the processes common to FIG.
Processing other than step S181 is the same as the reception processing of the first configuration example shown in FIG. The difference between this configuration example and the first configuration example is that the reception weight read in step S181 does not depend on the combination of terminal devices that are simultaneously spatially multiplexed. A transmission weight specific to each terminal apparatus that performs multiplexing is read out from the reception weight storage circuit 125b and used.

  In the process of step S181, the reception weight corresponding to each terminal device may be simply read for each terminal device without being conscious of the combination of the terminal devices. For this reason, it is possible to perform a reception process for any combination of terminal stations, as described in the above transmission process. Even when the number of terminal devices increases, an increase in the storage capacity of the reception weight storage circuit 125b can be suppressed.

(Supplement regarding configuration example)
The above is the description of the configuration example of the technology related to the present invention. The following are supplementary items common to these configuration examples.
In the base station apparatus 10 in the first and second configuration examples, the configuration in which the short-time averaging process illustrated in FIG. 22 is performed when acquiring the channel information in the uplink has been described. However, this was premised on the response to (Requirement 1). For example, although channel estimation is at a level where channel estimation is feasible in terms of channel design, when the base station apparatus 10 is used as a means for further obtaining channel gain for communication at a higher transmission rate, it is not necessarily short. There is no need to perform time averaging. In this case, the short-time averaging process (FIG. 22) for acquiring uplink channel information can be simply replaced with a channel estimation process.

  Similarly, in the base station apparatus 10 in the first and second configuration examples, when calculating the downlink channel information from the uplink channel information, the calibration coefficients shown in the equations (14) and (15) are used. The configuration using is described. However, as described above, the frequencies in the low noise amplifiers 103-1 to 103-K, the filters 106-1 to 106-K, the high power amplifiers 148-1 to 148-K, the filters 147-1 to 147-K, and the like. Adjust by analog signal processing so that the relative value (angle difference of complex phase) between the uplink and downlink of the amount of complex phase rotation for each component becomes a constant value in the circuit corresponding to all antenna elements. (For example, the complex phase of the uplink and downlink may be adjusted to be a constant value), it is not necessary to perform the process using the calibration coefficient. In this case, the process of acquiring the downlink channel information (FIG. 25) can be omitted, and the uplink channel information and the downlink channel information are equivalent, so the transmission weight and the reception weight are common. Value. In this case, the circuit related to the calculation of the transmission weight in the downlink and the circuit related to the calculation of the reception weight in the uplink can be shared.

  In this case, in the base station apparatus according to the first and second configuration examples of the technology related to the present invention, the multiuser MIMO transmission / reception weight calculation circuit 124a and the multiuser MIMO transmission / reception weight storage circuit 125a, or the transmission / reception weight calculation circuit 124b and the transmission / reception. This corresponds to one of the weight storage circuits 125b being configured to calculate and store the transmission weight and the reception weight. However, the present invention is not limited to this, and the first and second configurations of the technology related to the present invention. With the same configuration as the base station apparatus 10 in the example, the transmission weight may be calculated by regarding the calibration coefficient as “1” in all combinations of antenna elements and frequency components.

  Furthermore, in this case, the rotation amount of the complex phase does not necessarily have to be the same (or the calibration coefficient is 1) in all the antenna elements of each frequency component, and this condition is exceptional in some small number of antenna elements. Even in a situation that is not satisfied, as long as at least half of the antenna elements satisfy this condition, the operation intended by the present invention can be realized as a whole.

  Further, in the OFDM modulation scheme, all subcarriers are used for communication with the same terminal apparatus, and therefore, transmission / reception weights for the terminal apparatuses of a combination common to all subcarriers are used. However, in OFDMA, since allocations to different combinations of terminal devices in a patchwork pattern on the time axis and the frequency axis are gathered together, terminals allocated for each time (OFDM symbol) and frequency (subcarrier) It is necessary to use transmission / reception weights for the device. In this case, the reception signal processing circuits 109-1 to 109-L and the transmission signal processing circuits 141-1 to 141-L configured by a plurality of surfaces correspond to the terminal device that is the communication partner regardless of the frequency and time. However, it should be understood that it corresponds to a terminal device that is a communication partner when attention is paid to each frequency component or each time (OFDM symbol). However, except for the difference, OFDM and OFDMA can be processed in exactly the same way. In this specification, the description has been focused on OFDM, but the present invention is also applied to OFDMA. can do.

  In addition, there are various operational variations regarding SC-FDE, but reception signal processing and reception after signals transmitted from each antenna element are combined in space are multiplied by transmission weights on the transmission side. In any of the received signal processings after the reception weights are multiplied on the side and the signals of the respective antenna elements are added and combined, the processing performed in the conventional SC-FDE is applied as it is in each of the above-described configuration examples. Therefore, the present invention is applicable to all variations of SC-FDE.

  Furthermore, when adding and combining signals multiplied by reception weights over multiple antenna elements, it is not always necessary to add and combine over all antenna elements, and add and combine over some antenna elements in the whole. Even if it performs, it is possible to implement | achieve the operation | movement which the technique relevant to this invention intends as a whole, and can obtain the same effect as a result. Similarly, when adding and combining signals for a plurality of terminal devices multiplied by transmission weights for each antenna element, addition combining is not performed for all the antenna elements, and addition and combining is performed for some antenna elements. Even if it performs, it is possible to implement | achieve the operation | movement which the technique relevant to this invention intends as a whole.

  Similarly, in the technology related to the present invention, the preamble signal for channel estimation used for data communication can be a common preamble signal in all terminal devices, but other preambles are used in some terminal devices. It is of course possible to employ a configuration using a signal. If a common preamble is used when transmitting and receiving a signal by spatially multiplexing at least a plurality of terminal devices at the same time, it is intended by the technology related to the present invention. Corresponds to the action.

  Furthermore, in the channel information acquisition process shown in FIGS. 22 to 24 in the technology related to the present invention, what is the exchange process between the base station apparatus and the terminal apparatus for various control information such as instructions for starting these processes? It may be realized by a method. These processes are basically performed before the start of service operation. In this case, since an appropriate transmission / reception weight is unknown at the beginning, a sufficient line gain is secured between the base station apparatus and the terminal apparatus. It is assumed that various controls are performed in a situation where it is impossible. However, before the start of service operation, for example, an operator can manually instruct the start of processing in the installation work of the terminal device, or temporarily control using other wireless standards It doesn't matter. Therefore, it is possible to implement the technology related to the present invention regardless of various control processing methods such as an instruction for starting the acquisition processing of channel information.

Furthermore, the offset value φ ^(k) of the complex phase used when acquiring the relative component can be acquired by a method other than the processing shown in FIG.
FIG. 33 is a flowchart showing another relative component acquisition process for acquiring a relative component of uplink channel information in each configuration example of the technology related to the present invention described above. The difference between the process shown in FIG. 23 and the process shown in FIG. 23 is that the complex phase offset value φ ^(k) used when acquiring the relative component is based on the complex phase of the specific antenna element 101-1. Instead, in step S193, an average value of complex phases (that is, angles represented by 0 to 2π) of all the antenna elements 101-1 to 101-K is used. Based on the k-th frequency components ^ h ^(k) ₁ ,..., ^ H ^(k) _K in the channel information corresponding to the antenna elements 101-1 to 101-K in steps S122-1 to S122-K, Using (24), the average value φ ^(k) of the complex phases of all antennas with respect to the k-th frequency component is obtained, and this is used in steps S124-1 to S124-K, thereby obtaining the relative component. This offset value is obtained individually for each frequency component.
Even when the complex phase component of each of the antenna elements 101-1 to 101-K includes an error, the equation (24) averages the error, and as a result, a highly accurate relative component is obtained. Can do.

  As a method of acquiring downlink channel information, in addition to the method using uplink channel information shown in this specification, as shown in the direct method of FIG. A method is also conceivable in which a training signal is directly transmitted on the downlink and channel information acquired by a terminal device that has received the training signal is fed back to the base station device in a form of feedback. In this case, the training signals shown in FIG. 15 are transmitted one by one from the antenna elements included in the base station device, and the same processing as the processing shown in FIGS. 22 to 24 is performed on the terminal device side. The same configuration may be adopted in which the channel information for each averaged antenna element and frequency component obtained as a result is fed back and set to the base station by some method and the transmission weight is calculated using this on the base station apparatus side. It is possible to obtain the effect. However, even in this case, transmission of a training signal from each terminal device is indispensable for acquiring uplink channel information, and this point is exactly the same as the above-described technology related to the present invention.

Further, for example, formula (24) in is performed a process of extracting the complex phase of the channel information ^ h ^(k) _i, from the ratio of the real and imaginary parts of the channel information ^ h ^(k) _i of complex phase angle It is also possible to acquire information and calculate a value equivalent to the equation (24) based on the angle information. Although this appears to be mathematically different, it is mathematically equivalent, and of course it is possible to substitute processing with such mathematically equivalent alternatives for all arithmetic operations. It is.

  Similarly, the base station apparatus 10 in the second configuration example performs the long-time averaging process shown in FIG. 24 as the process after obtaining the relative component of the uplink channel information shown in FIG. The transmission / reception weight calculation processing shown in FIG. However, since the weights shown in the equations (25) to (26) correspond to simple complex phase component extraction, the weight calculation processing shown in FIG. 31 is performed after the relative components shown in FIG. By replacing steps S131-1 to S131-Q with acquisition of individual reception weights and replacing the long-time averaging target shown in step S133 with this reception weight, an approximately equivalent long-time average reception weight is acquired. It is possible to do. That is, in the first and second configuration examples described above, the physical quantity that is subject to long-term averaging is a relative component of channel information, but the reception weight calculated from the relative component of channel information is subject to long-term averaging. It is also possible to replace it with a physical quantity.

  Furthermore, in the installation example of the base station apparatus included in the wireless communication system according to the present invention illustrated in FIG. 13, the terminal apparatuses 12-1 to 12-2 are illustrated as including one antenna. Even if a plurality of antennas are provided, the same processing can be performed. In principle, if the terminal apparatuses 12-1 to 12-2 are provided with a plurality of antennas, a plurality of signal sequences can be spatially multiplexed in one terminal station. In this case, it is possible to similarly implement the technology related to the present invention by regarding each antenna of the terminal devices 12-1 to 12-2 as an antenna element of each terminal station. However, in the present invention, since it is assumed that the terminal devices 12-1 to 12-2 and the antenna elements 13-1 to 13-4 of the base station device are in a line-of-sight environment, generally, It is often difficult to perform MIMO transmission between one terminal station (after the second eigenvalue approaches zero). Therefore, when the terminal station is provided with a plurality of antennas, it is practical to operate transmission / reception of a single signal sequence as a diversity configuration of a plurality of antennas. In this case, by combining a plurality of antennas with appropriate weights, it can be regarded as a virtual one antenna, and the same processing is realized with this virtual one antenna. Then, it is possible to apply the technology related to the present invention in exactly the same way.

Furthermore, in the above description of the operation principle and the configuration example of the technology related to the present invention, the channel information and transmission / reception weight corresponding to each antenna element have been described. However, the channel information or transmission / reception weight of each antenna element is used as a component. In the constructed vector, the direction indicated by the vector has an effective meaning. For this reason, the vector obtained by multiplying a certain vector by a predetermined coefficient is the same in direction, so that a vector obtained by multiplying a certain coefficient for each antenna element has all equivalent meanings without depending on the coefficient to be multiplied. Will have.
On the other hand, as described in the preconditions of the present invention, since the antennas of the base station device and the terminal device are assumed to be a line-of-sight environment or an environment close to the line-of-sight environment, the intensity and amplitude of the signal received by each antenna element Is expected to be roughly constant. For this reason, for example, the vector of channel information of each antenna element is effectively not so significant in the absolute value of each component of the vector, and the channel information value is normalized (the channel information is the absolute value). (Complex number obtained by dividing by) is significant information. For this reason, the processing in which the “channel information” used in the description of the operation principle and the above is approximately regarded as the “value obtained by normalizing the value of the channel information” is the present invention and related technology of the present invention. In this sense, the above-mentioned “channel information” includes up to “a value obtained by standardizing the value of channel information” in a broad sense.

Furthermore, in the specification of the present invention, for convenience of explanation, “row vector” and “column vector” are treated without being distinguished from each other. For example, the channel information vector h _i in the expression (3) is a row vector, the transmission weight vector w _j is a column vector, and “transpose” or the like if it is a strict mathematical notation that unifies the direction of vector arrangement. Should be written using the symbol etc. However, the information required in the practice of the present invention is the value of each component of the vector, and it does not make much sense whether the vector is a row vector or a column vector. The description does not distinguish between “vector” and “column vector”.

  Still further, the main feature of the related technology described above is that, as pre-processing prior to actual communication, uplink and / or downlink channel information is acquired in advance, and reception weight and / or transmission weight is determined in advance based on that information. In actual communication, transmission / reception signal processing is performed with reference to the reception weight or transmission weight acquired in advance. As a result, even when the number of antenna elements becomes enormous, it is possible to perform real-time processing while reducing the circuit scale without accompanying a very heavy calculation load in the transmission / reception weight acquisition processing. It becomes possible. Therefore, acquisition of relative components for each antenna element of channel information is not necessarily performed, and averaging processing thereof is not performed. Even if transmission / reception weights are acquired using a single channel information acquisition result as preprocessing, signal separation is not performed. Therefore, the accuracy of signal separation due to the transmission / reception weight may be somewhat lowered, but it is also possible to obtain the intended effect of the related technology. In the present invention, the use of this related technique is mainly assumed. However, acquisition of the relative component of the channel information and the averaging process are not necessary as an essential condition for obtaining the effect of the present invention.

Furthermore, regarding the scheduling process shown in step S162 of FIGS. 27 and 32 and step S172 of FIGS. 28 and 33, for example, the SINR (Signal to Interference and Noise Power Ratio) of each terminal device that deteriorates due to the influence of time variation or the like. : Signal-to-interference noise power ratio) value may be determined by determining whether the expected value can satisfy a predetermined condition. For example, the SNR value of the i-th terminal device which is the i-th terminal device is x _i [dB], the SIR value corresponding to the class value of the time variation index is y _i [dB], and the required SINR value of the transmission mode to be applied is z _{It is} assumed that _i [dB]. In this case, in order to satisfy the desired communication quality, it is necessary to satisfy the following equation (25).

Here, the range in which the sum of Σ in the second term on the left side of Equation (25) is taken is that only the i-th terminal subject to quality evaluation of Equation (25) is compared to the terminal number j assumed as the object of spatial multiplexing transmission. The excluded range (j ≠ j). For example, when all of the terminal devices that are assumed to be spatially multiplexed at a certain stage of scheduling are spatially multiplexed, if the equation (25) is not satisfied for a certain number j of terminal devices, the i-th terminal is In addition to being excluded from the target of spatial multiplexing transmission, the terminal number may be excluded from the range in which the sum of Σ in the determination of whether or not the following equation (25) is satisfied is taken. As long as such a condition can be satisfied, the scheduling processing circuit 181 narrows down the terminal devices to be spatially multiplexed. Here, since the required SINR value (z _i [dB]) of the transmission mode to be applied that appears on the right side of the equation (25) changes depending on the transmission mode to be applied, the adaptation of such a transmission mode is applied. It may be dealt with by a change. As described above, various conditions need to be determined and a search process for a combination of terminal devices is required for scheduling. However, the related technology of the present invention can be applied without depending on the scheduling process, and there are particularly limited conditions. It is not necessary.

[Specific Embodiment of the Present Invention Based on Related Technology]
Based on the above principle of operation, a specific embodiment of the present invention will be described below. The embodiment of the present invention is implemented in combination with the second configuration example of the technology related to the present invention. In the following description, for the sake of simplicity, the uplink / downlink, transmission / reception, and frequency components (sub-components) are explicitly specified for channel information vectors, transmission / reception weight vectors (and combinations of them), and the like. The subscript representing the carrier number) is omitted, and for example, the channel information vector of the i-th terminal device of the target subcarrier is denoted by h _i and the weight vector of the i-th terminal device is denoted by w _i . Therefore, in this information, a vector and matrix of different values exist for each frequency component, and a vector and matrix of individual values exist for the uplink / downlink, and the same processing is performed. Further, since they are common in terms of vectors composed of elements corresponding to the respective antenna elements, they are described without distinguishing row vectors or column vectors. In strict mathematical notation, the product of a row vector (size is 1 × N) and a column vector (size is N × 1) is a scalar quantity (size is 1 × 1), a column vector (size is N × 1) and a row The product of vectors (size is 1 × N) represents a matrix of size N × N. Here, the sum of the multiplication results of each component of the vector is simply made without distinguishing the row vector or the column vector. This means the calculation of the vector product obtained by (because each component is a complex number, it is different from the so-called vector inner product).

(Residual interference suppression processing)
<Transmitter signal processing>
Assuming spatial multiplexing for the first to N-th terminal devices, channel information vectors of i-th terminal devices (i = 1, 2,..., N) of subcarriers of interest in the downlink are denoted by h _i (strictly, row vectors). ) For transmission weights w _i (strictly column vectors), the product of channel information matrix H and transmission weight matrix W configured using these vectors for each row and column is given by The sum of the matrix W _d having only the diagonal component expressed by (26) and the matrix W _nd having only the non-diagonal component expressed by the following formula (27), that is, W _d + W _nd , It can be expressed as

Further, if a transmission signal addressed to the i-th terminal apparatus in a certain OFDM symbol is t _i , a transmission signal vector having components t ₁ to t _N corresponding to each terminal apparatus is Tx. Further, a transmission signal correction signal addressed to a virtual k-th order i-th terminal device is δt _i ^(k), and this transmission signal correction signal δt _i ^(k) is an i-th component. Let the signal vector be δTx ^(k) . As an initial condition, δTx ^{(0) is set} to Tx. A signal received by each terminal device when a signal obtained by adding transmission signal correction signals from the 0th order to the kth order is transmitted using the transmission weight matrix W is expressed by the following equation (29).

  On the right side of Expression (29), the i-th diagonal term and the (i-1) -th non-diagonal term are described as a pair. Here, the condition that the paired portion becomes zero is set by the following equation (30).

  When formula (30) is transformed, a recurrence formula of the following formula (31) is obtained.

The matrix ΔW in the equation (31) is given by the following equation (32). Note that the matrix W _d ⁻¹ in Equation (32) is a matrix whose diagonal component is the inverse of the product h _i w _i of the channel information vector h _i and the transmission weight w _i .

  By substituting the condition of Expression (30) into Expression (29), the following Expression (33) is obtained.

Here, since the first term on the right side of Equation (33) is obtained by multiplying the diagonal matrix by the transmission signal vector, there is no mutual interference component between the terminal devices. On the other hand, since the mutual interference component is included in the second term on the right side, interference suppression can be achieved by reducing the second term on the right side.
Further, looking at each element of the matrix ΔW in equation (32), the denominator is multiplied by the transmission weight corresponding to the channel information vector (that is, the vector subscript is common), so that the complex phase is obtained at each antenna element. Each component is added and synthesized under the same conditions, but as the numerator is different from the subscript of the vector, the complex phase is added in a random state instead of in-phase synthesis, so the antenna When the number of elements is very large, the denominator of in-phase synthesis is overwhelmingly larger than the numerator of random phase synthesis. That is, since it is expected with a very high probability that each component of the matrix is a matrix sufficiently smaller than 1, every time ΔW is multiplied, the absolute value of the transmission signal correction signal vector δTx ^(k) Becomes smaller. That is, the interference component can be suppressed by increasing the approximate order k.

Here, the (i, j) component of the matrix ΔW in the equation (32) is −h _i w _j / h _i w _i , and the service operation in which the base station apparatus and the plurality of terminal apparatuses perform data communication. if in advance to _{-h i w j / h i w} i is stored in the calculated memory or the like before the start, at the time of sending the actual signal after the service started, -h _i w which has been stored is calculated in advance _The matrix ΔW may be formed by reading out _j / h _i w _i from a memory or the like and substituting it for each component.
For example, when 100 terminal devices are under the control of a base station device, 100 × 100 = 10000 vector operations may be performed in advance for all combinations of (i, j). In view of the above processing, when the same processing is performed on all frequency components, the amount of computation is not a problem level.

On the other hand, in the recurrence formula of Expression (31), the number of multiplications is N × (N−1) times. If the number of antenna elements is M, the orthogonal transmission weights represented by equations (6) to (9) may be calculated by referring to a table for calculating the square root or the like. The number of multiplications is M × N × (N−1) times or more. Even assuming values of M = 100, N = 20, and k = 3, the computation amount is only about 1/33, and complicated procedures such as Gram Schmidt orthogonalization are not repeated in order. In addition, since only a simple matrix-vector multiplication is required, the number of multiplications is not only small, but the circuit configuration is simple and can be realized with an efficient configuration in which the same circuit can be easily used for similar operations. it can. Therefore, even when the number of antenna elements is large and the spatial multiplexing number N is large, it can be realized with a realistic amount of calculation.
These calculations are performed individually for each frequency component.

<Receiver signal processing>
Similarly, in the case of the uplink, the procedure for interference suppression will be described below. Assuming spatial multiplexing for the first to N-th terminal devices, channel information vectors of the i-th terminal devices (i = 1, 2,..., N) of the frequency component of interest in the uplink are represented as h _i (strictly column vectors). ) receiving weight if w _{i (strictly} row vector) with respect to the product of the receive weight matrix W and the channel information matrix H constructed using these vectors are represented by the following formula (34) The sum of the matrix W _d having only the diagonal component and the matrix W _nd having only the non-diagonal component expressed by the following formula (35), that is, W _d + W _nd can be expressed by the following formula (36). .

  In order to simplify the notation as described at the beginning, the same symbol is used for the notation for transmission and reception (or downlink and uplink), but in reality, a vector of different values (Mathematical Are also different in terms of row vectors and column vectors). In addition, the weight vector and the channel information vector are reversed left and right for the components of each matrix. This is because the received weight is multiplied by the received signal multiplied by the channel matrix and is in this format. It is. However, in this specification, row vectors and column vectors are treated without distinction, so that the weight vector and channel information vector are reversed in the left and right directions in each component of the matrix. Not give.

Next, a received signal from the i-th terminal device obtained as a result of multiplying a signal received by each antenna element in an OFDM symbol by a reception weight is denoted by r _i , and a received signal vector having components r ₁ to r _N as components. Is Rx. Using Expression (36), when a transmission signal vector having a transmission signal from each terminal device as each component is Tx, a signal obtained by multiplying the reception signal by the reception weight is expressed by the following Expression (37). .

Here, since the matrix W _d is a square matrix having only non-zero diagonal components, there is an inverse matrix, and when the inverse matrix of W _d is multiplied from the left in Expression (37), the following Expression (38) is obtained. .

The matrix ΔW in the equation (38) is represented by the following equation (39). The matrix W _d ⁻¹ in Expression (39) is a matrix having a diagonal component that is the inverse of the product w _i h _i of the reception weight w _i and the channel information vector h _i .

Also, the received signal correction signal from the virtual k-th order i-th terminal device is δr _i ^(k), and this received signal correction signal is the k-th order received signal with δr _i ^(k) as the i-th component. Let the correction signal vector be δRx ^(k) . As an initial condition, δRx ⁽⁰⁾ is given by the following equation (40).

  Further, a recurrence formula between the (k−1) th order and the kth order received signal correction signal vector is defined by the following formula (41), and a general solution when sequentially solving the recurrence formula using the initial condition is It is defined by the following formula (42).

  Further, when Expression (40) and Expression (38) are substituted into Expression (42), the following Expression (43) is obtained.

  Using this equation (43), a corrected received signal obtained by adding received signal correction signals from the 0th order to the kth order is expressed by the following equation (44).

  On the right side of the equation (44), the first term in the (i + 1) th order Σ cancels out with the second term in the i-th order Σ, so that what remains finally is the following equation (45) Only the right side of

  Here, the first term on the right side of the equation (45) means that the received signal detection without interference components between the terminal devices can be performed because the i-th component is the transmission signal from the i-th terminal device. Since the second term is a mutual interference component, interference suppression can be achieved by reducing the second term on the right side.

Similarly, when looking at each element of the matrix of equation (39), the denominator is multiplied by the reception weight corresponding to the channel information vector (that is, the vector subscript is common), so that each antenna element has a complex phase. Since each component is added and combined under the same condition, but the numerator is different from the subscript of the vector, the complex phase is added in a random state without being in-phase combined. When the number of antenna elements is very large, the denominator of in-phase synthesis is overwhelmingly larger than the numerator of random phase synthesis. That is, since it is expected with a very high probability that the absolute values of the respective components of the matrix are all sufficiently smaller than 1, each time the matrix ΔW is multiplied, the received signal correction signal vector δRx ^{(k )} Becomes smaller. That is, the interference component can be suppressed by increasing the approximate order k.
The approximate order k may be given by any method. For example, for the downlink, the second term on the right side of Equation (33) is the residual interference and the second term on the right side of Equation (45) is the residual interference for the uplink. The value may be determined as described above, or after the temporary value is once determined, the value of k may be appropriately adjusted in operation.

Here, the matrix ΔW of formula (39) (i, j) component is a _{-w i h j / w i h} i , the _{-w i h j / w i h} i in advance before the start of service operation If it is calculated and stored in a memory or the like, at the time of actual signal reception after the service is started, the previously calculated -w _i h _j / w _i h _i is read from the memory or the like and substituted for each component to obtain the matrix ΔW. What is necessary is just to comprise. Similar to the processing on the transmission side, not only can the number of multiplications be significantly reduced on the reception side, but also the circuit configuration is simple and can be realized with an efficient configuration that makes it easy to divert the same circuit to the same operation. Therefore, even when the number of antenna elements is large and the spatial multiplexing number N is large, it can be realized with a realistic calculation amount.

Note that the transmission / reception weights used in the above processing can be applied to general weights obtained using equation (22), equation (23), etc., and correcting imperfections in these transmission / reception weights. Thus, residual interference due to imperfection can be suppressed.
With regard to the reception weight, formula (22), instead of using such equation (23) may be used to receive weights _{w i,} given by the following equation (46).

  In addition, here, in order to simplify the explanation, unlike the expressions (22) and (23), the subscripts representing the frequency components (subcarriers) are omitted. Of different weights. When this reception weight is used, the following relational expression (47) is obtained.

That is, since the matrix W _d ⁻¹ is a unit matrix, the initial vector, which is the initial value of the above recurrence formula, is equal to the reception signal Rx after δRx ⁽⁰⁾ is multiplied by the reception weight and subjected to addition synthesis. Therefore, the arithmetic processing can be slightly simplified.
These calculations are performed individually for each frequency component.

(About embodiment which concerns on circuit structure)
<About the overall configuration>
An embodiment relating to a device configuration for realizing the signal processing described above will be described below.
FIG. 1 is a schematic block diagram showing a configuration of a base station apparatus 200 in the embodiment of the present invention. The same reference numerals are used for the same functions as in FIG. As shown in the figure, the base station apparatus 200 includes a communication control circuit 110, a storage circuit 115, an interface circuit 170, a MAC layer processing circuit 180, a reception unit 210, a transmission / reception weight calculation unit 290, a transmission unit 240, and a reception signal correction unit. 230 and a transmission signal correction unit 250. The MAC layer processing circuit 180 has a scheduling processing circuit 181.

The difference between the base station apparatus 200 in this embodiment and the base station apparatus 10 shown in FIG. 18 is that a reception signal correction unit 230 and a transmission signal correction unit 250 are added, and a reception unit 210, a transmission unit 240, The transmission / reception weight calculation unit 290 is changed.
The reception signal correction unit 230 is connected to the transmission / reception weight calculation unit 290, and the reception signal correction matrix ΔW given by the equation (39) is input from the transmission / reception weight calculation unit 290. The transmission signal correction unit 250 is also connected to the transmission / reception weight calculation unit 290, and the transmission signal correction matrix ΔW given by Expression (32) is input from the transmission / reception weight calculation unit 290.
Furthermore, the reception signal correction unit 230 is connected to the reception unit 210, receives the reception signal vector Rx multiplied by the reception weight from the reception unit 210, corrects the reception signal, and outputs it to the reception unit 210. The transmission signal correction unit 250 is connected to the transmission unit 240, receives the transmission signal vector Tx from the transmission unit 240, corrects the transmission signal, and outputs the corrected transmission signal to the transmission unit 240.

<Receiver configuration>
FIG. 34 is a diagram showing a configuration of the reception signal processing circuits 109-1 to 109-L in the second configuration example of the technology related to the present invention. In the configuration of the receiving unit 100 shown in FIG. 19, the L-side received signal processing circuits 109-1 to 109-L are provided, but FIG. Is shown.

  In the figure, 120b is a transmission / reception weight calculation unit, 109-i is a reception signal processing circuit corresponding to the i-th terminal device, 201-i-1 to 201-i-K are multiplication circuits, and 202-i is a reception weight vector temporary. A storage unit, 203-i represents an addition / synthesis circuit, and 204-i represents a demodulation processing circuit. The reception signal processing circuit 109-i (i = 1, 2,..., L) includes K multiplication circuits 201-i-1 to 201-i-K, a reception weight vector temporary storage unit 202-i, and addition. A synthesis circuit 203-i and a demodulation processing circuit 204-i are included.

In the base station apparatus 10, signals received by the antenna elements 101-1 to 101-K are input to the reception signal processing circuit 109-i via the FFT circuits 108-1 to 108-K of the reception unit 100. The
In the reception signal processing circuit 109-i, the reception weight vector temporary storage unit 202-i reads the reception weight vector corresponding to the i-th terminal device from the transmission / reception weight calculation unit 120b and stores it. Multiplying circuits 201-i-1 to 201 -i-K are the signals input from the FFT circuits 108-1 to 108 -K for each OFDM symbol and the reception weights stored in the reception weight vector temporary storage unit 202-i. And outputs the multiplication result to the adder / synthesizer circuit 203-i. The adder / synthesizer circuit 203-i adds and synthesizes the signals input from the multiplier circuits 201-i-1 to 201-i-K to obtain an added synthesized signal which is a reception signal after reception weight multiplication. The demodulation processing circuit 204-i performs signal detection processing on the signal transmitted from each terminal device through signal detection processing on the addition combined signal obtained by the addition combining circuit 203-i. When the demodulation processing circuit 204-i detects a signal transmitted from the terminal device, the demodulation processing circuit 204-i outputs the detected signal as data output #i.

  The above is the basic operation, but in a system using OFDM (A) or SC-FDE, it is necessary to perform the same processing for each frequency component. For this reason, in order to carry out the above processing for a plurality of frequency components, for example, in the reception signal processing circuits 109-1 to 109-L on the L surface in the related art, circuits each having a number of surfaces corresponding to the number of subcarriers. It is also possible to perform processing of the entire frequency component by arranging the processing of a plurality of subcarriers in order in time and reusing the same circuit. In addition, a configuration may be adopted in which processing is performed by arranging the same circuits in order in time. Therefore, in the following description, a circuit configuration and processing focusing on a certain frequency component will be described for simplicity, but it should be understood that the same processing can be performed on all frequency components. is there.

  As a supplement, FIG. 19 was originally a schematic block diagram showing the configuration of the receiving unit 100 of the base station apparatus 10 of the first configuration example in the technology related to the present invention. In this configuration example, the transmission / reception weight calculation unit 120a is simply replaced with the transmission / reception weight calculation unit 120b, and the description has been given with reference to this figure.

The above is the processing in the related art so far, but in the embodiment of the present invention, this is changed as follows.
FIG. 35 is a schematic block diagram showing the configuration of the reception signal processing circuits 209-1 to 209-L and the reception signal correction unit 230 in the embodiment of the present invention. The same components as those in FIG. 34 are given the same reference numerals. The reception signal processing circuits 209-1 to 209-L are provided in the reception unit 210, and correspond to the reception signal processing circuits 109-1 to 109-L of the reception unit 100 in the technology related to the present invention.

In the figure, 201-i-1 to 201-i-K are multiplication circuits, 202-i is a reception weight vector temporary storage unit, 203-i is an addition synthesis circuit, 204-i is a demodulation processing circuit, and 209-i is Received signal processing circuit corresponding to the i-th terminal device, 230 is a received signal correction unit, 231 is a received signal correction management circuit, 232 is a received signal vector temporary storage unit, 233 is a matrix / vector multiplier circuit, 234 is a received signal correction matrix A temporary storage circuit 290 represents a transmission / reception weight calculation unit.
As shown in the figure, the reception signal processing circuit 209-i (i = 1, 2,..., L) includes K multiplication circuits 201-i-1 to 201-i-K and a reception weight vector temporary storage. A unit 202-i, an addition / synthesis circuit 203-i, and a demodulation processing circuit 204-i. The reception signal correction unit 230 includes a reception signal correction management circuit 231, a reception signal vector temporary storage unit 232, a matrix / vector multiplication circuit 233, and a reception signal correction matrix temporary storage circuit 234.

  Also, the received signal processing circuits 109-1 to 109-L on the L plane in FIG. 19 are similarly replaced with the received signal processing circuits 209-1 to 209-L on the L plane. In FIG. The reception signal processing circuit 209-i on the surface (corresponding to the i-th terminal device) is shown in the drawing, but similar reception signal processing circuits 209-1 to 209-L exist.

  As a feature of the present embodiment, in the related technology of FIG. 34, the addition combined signal output from the addition combining circuit 203-i is directly input to the demodulation processing circuit 204-i. A signal (added synthesized signal) added and synthesized after reception weight multiplication, which is an output from the addition synthesis circuit 203-i, is input to the received signal correction unit 230. The reception signal correction unit 230 receives this signal from the reception signal processing circuits 209-1 to 209-L corresponding to all the terminal devices, and the reception signal correction management circuit 231 manages this signal. The reception signal correction management circuit 231 manages a series of recurrence calculations, and further receives a reception signal before correction. The reception signal processing circuits 209-1 to 209 -L receive the corrected reception signal as a corrected reception signal. To the demodulation processing circuit 204-i. The correction by the reception signal correction unit 230 is performed for each frequency component. An input signal to the received signal correction management circuit 231 from the adder / synthesizer circuit 203-i and a signal output from the received signal correction management circuit 231 to the received signal processing circuits 209-1 to 209-L are frequencies for the corresponding terminal device. This is a signal for each component, and the same processing is performed for all frequency components.

  The process related to the recurrence formula is as follows: received signal correction management circuit 231 → received signal vector temporary storage unit 232 → matrix / vector multiplication circuit 233 → received signal correction management circuit 231 → received signal vector temporary storage unit 232 → matrix / vector multiplication circuit 233 → Execute the process while transferring the signals in order.

First, the received signal _{r i} from the corresponding terminal apparatus from the received signal processing circuit 209-1~209-L to the received signal correction management circuit 231 is input. The reception signal correction management circuit 231 causes the reception signal vector temporary storage unit 232 to store a reception signal vector Rx = {r ₁ , r ₂ ,..., R _N } that is a combination of all the input reception signals r _i . On the other hand, first, the matrix W _d ⁻¹ is input from the transmission / reception weight calculation unit 290 to the reception signal correction matrix temporary storage circuit 234. Strictly speaking, since this is a diagonal matrix, it suffices if a value of 1 / w ₁ h ₁ to 1 / w _N h _N corresponding to the diagonal component is input.

First, an initial value δRx ⁽⁰⁾ of the received signal correction signal vector is obtained using this matrix. δRx ⁽⁰⁾ is given by equation (40), which can be obtained as (r ₁ / w ₁ h ₁ , r ₂ / w ₂ h ₂ ,..., r _N / w _N h _N ). Instead of the matrix operation, it may be obtained by simply executing the operation (r ₁ / w ₁ h ₁ , r ₂ / w ₂ h ₂ ,..., R _N / w _N h _N ) as it is. The initial value δRx ⁽⁰⁾ obtained in this way is returned from the matrix / vector multiplication circuit 233 to the reception signal correction management circuit 231.
In the case of using the reception weight of the formula (46), the product w _i h _i of the reception weight and channel information for a 1 as shown in equation (47), the matrix W _d ^-1 becomes an identity matrix In this case, δRx ⁽⁰⁾ matches Rx. In this case, the information of the matrix W _d ⁻¹ is not necessary in the process, and the information input of the matrix W _d ⁻¹ from the transmission / reception weight calculation unit 290 to the reception signal correction matrix temporary storage circuit 234 can be omitted.

The reception signal correction management circuit 231 further stores the calculated initial value δRx ⁽⁰⁾ in the reception signal vector temporary storage unit 232. Next, in order to sequentially solve the recurrence formula represented by Expression (41), the matrix ΔW represented by Expression (39) is read from the transmission / reception weight calculation unit 290, and this is received by the received signal correction matrix temporary storage circuit 234. Remember. The matrix / vector multiplication circuit 233 performs multiplication of the matrix stored in the received signal correction matrix temporary storage circuit 234 and the vector stored in the received signal vector temporary storage unit 232, thereby obtaining the order of the recurrence formula j Next (j + 1) next.

Further, the matrix / vector multiplication circuit 233 similarly returns the result to the reception signal correction management circuit 231. The received signal correction management circuit 231 manages whether or not the approximate processing from the number of loop processes to the target order k has been performed, and when the target order is reached, δRx ⁽⁰⁾ to δRx ^{(k )} Are added, and the respective components are output as corrected reception signals for each of the reception signal processing circuits 209-1 to 209-L.

The demodulating circuit 204-i receives the i-th component of the corrected received signal vector ΣδRx ^(j) , which is the sum of j from 0 to k, from the received signal correction unit 230. The demodulation processing circuit 204-i performs signal detection processing using this. Note that this signal is already in a state in which the transmission signal Tx can be estimated, as shown in Expression (45), and processing such as hard decision or soft decision processing by reverse mapping or error correction is performed. It will be.

  The matrix operation in the matrix / vector multiplication circuit 233 is not necessarily performed all at once, and a process of sequentially multiplying the matrix component and the vector component and adding them may be performed. For this reason, the reception signal vector temporary storage unit 232 is provided so that the vector components can be read as appropriate. However, the reception signal vector temporary storage unit 232 is not necessarily provided when performing matrix operation in a lump. The signal correction management circuit 231 may directly input the matrix / vector multiplication circuit 233.

  Furthermore, as mentioned earlier, in a system that employs OFDM (A), SC-FDE, etc., each vector or matrix has a different value for each frequency component, and these processes are individually performed for all frequency components. Need to be done. They may be processed in parallel by individual circuits, or may be implemented serially using the same circuit.

<Configuration on the sending side>
FIG. 36 is a diagram showing an internal configuration of the transmission signal processing circuits 141-1 to 141-L in the second configuration example of the technology related to the present invention. The configuration of the transmission unit 140 illustrated in FIG. 21 includes the L-plane transmission signal processing circuits 141-1 to 141-L, but FIG. 36 illustrates the transmission signal processing circuits 141-1 to 141-L. The configuration is shown.

  In the figure, 120b is a transmission / reception weight calculation unit, 141-i is a transmission signal processing circuit corresponding to the i-th terminal device, 205-i is a modulation processing circuit, 206-i is a signal duplication circuit, and 207-i is a transmission weight vector. Temporary storage units 208-i-1 to 208-i-K represent multiplication circuits. The transmission signal processing circuit 141-i (i = 1, 2,..., L) includes a modulation processing circuit 205-i, a signal duplicating circuit 206-i, a transmission weight vector temporary storage unit 207-i, and K pieces of signals. Multiplier circuits 208-i-1 to 208-i-K.

In the base station apparatus 10, transmission data for each frequency component input to the transmission signal processing circuit 141-i is subjected to a series of modulation processing by the modulation processing circuit 205-i, and this is processed by the signal duplicating circuit 206-i. Copy for each antenna element 101-1 to 101 -K.
In the transmission signal processing circuit 141-i, the transmission weight vector temporary storage unit 207-i reads the transmission weight vector corresponding to the i-th terminal device from the transmission / reception weight calculation unit 120b and stores it. Multiplication circuits 208-i-1 to 208-i-K multiply each component of the transmission weight vector and the signal input from the signal duplication circuit 206-i for each OFDM symbol. Multiplication circuits 208-i-1 to 208 -i-K output the multiplication results to addition synthesis circuits 142-1 to 142 -K in transmission section 140.

  The above is the basic operation, but in a system using OFDM (A) or SC-FDE, it is necessary to perform the same processing for each frequency component. For this reason, in order to perform the above processing for a plurality of frequency components, for example, in the transmission signal processing circuits 141-1 to 141-L on the L plane in the related art, circuits each having a number corresponding to the number of subcarriers. It is also possible to perform processing of the entire frequency component by arranging the processing of a plurality of subcarriers in order in time and reusing the same circuit. In addition, a configuration may be adopted in which processing is performed by arranging the same circuits in order in time. Therefore, in the following description, a circuit configuration and processing focusing on a certain frequency component will be described for simplicity, but it should be understood that the same processing can be performed on all frequency components. is there.

  As a supplement, FIG. 21 was originally a schematic block diagram showing the configuration of the transmission unit 140 of the base station apparatus 200 of the first configuration example according to the technology related to the present invention. In the configuration example of 2, the transmission / reception weight calculation unit 120a is simply replaced with the transmission / reception weight calculation unit 120b, and the description has been given with reference to this figure.

  The above is the processing in the related art so far, but in the embodiment of the present invention, this is changed as follows. FIG. 37 is a schematic block diagram showing configurations of the transmission signal processing circuits 241-1 to 241-L and the transmission signal correction unit 250 in the embodiment of the present invention. The same reference numerals are given to the same components as those in FIG. The transmission signal processing circuits 241-1 to 241-L are provided in the transmission unit 240, and correspond to the transmission signal processing circuits 141-1 to 141-L of the transmission unit 140 in the technology related to the present invention.

  In the figure, 120b is a transmission / reception weight calculation unit, 241-i is a transmission signal processing circuit corresponding to the i-th terminal device, 205-i is a modulation processing circuit, 206-i is a signal duplication circuit, and 207-i is a transmission weight vector. Temporary storage units, 208-i-1 to 208-i-K are multiplication circuits, 250 is a transmission signal correction unit, 251 is a transmission signal correction management unit, 252 is a transmission signal vector temporary storage unit, and 253 is a matrix / vector multiplication circuit. Reference numeral 254 denotes a transmission signal correction matrix temporary storage circuit, and 290 denotes a transmission / reception weight calculation unit.

  As shown in the figure, the transmission signal processing circuit 241-i (i = 1, 2,..., L) includes a modulation processing circuit 205-i, a signal duplicating circuit 206-i, and a transmission weight vector temporary storage unit 207. -I and K multiplication circuits 208-i-1 to 208-i-K. The transmission signal correction unit 250 includes a transmission signal correction management circuit 251, a transmission signal vector temporary storage unit 252, a matrix / vector multiplication circuit 253, and a transmission signal correction matrix temporary storage circuit 254.

  Also, the L-plane transmission signal processing circuits 141-1 to 141-L in FIG. 21 are similarly replaced with L-plane transmission signal processing circuits 241-1 to 241-L. In FIG. Although attention is paid to the transmission signal processing circuit 241-i of the plane (corresponding to the i-th terminal device), similar transmission signal processing circuits 241-1 to 241-L exist.

  As a feature of the present embodiment, in the related technique of FIG. 36, the output from the modulation processing circuit 205-i is directly input to the signal duplicating circuit 206-i, but in this embodiment, the modulation processing circuit 205-i. The output from −i is input to the transmission signal correction unit 250. This signal is input to the transmission signal correction unit 250 from the transmission signal processing circuits 241-1 to 241-L corresponding to all the terminal devices, and this is managed by the transmission signal correction management circuit 251. The transmission signal correction management circuit 251 manages a series of recursive equations, further receives a transmission signal before correction, and sets the transmission signal after correction as a transmission signal after correction as transmission signal processing circuits 241-1 to 241-L. To the signal duplicating circuit 206-i. Note that correction by the transmission signal correction unit 250 is performed for each frequency component. An input signal from the modulation processing circuit 205-i to the transmission signal correction management circuit 251 and a signal output from the transmission signal correction management circuit 251 to the transmission signal processing circuits 241-1 to 241-L are frequencies for the corresponding terminal device. This is a signal for each component, and the same processing is performed for all frequency components.

  The process related to the recurrence formula is as follows: transmission signal correction management circuit 251 → transmission signal vector temporary storage unit 252 → matrix / vector multiplication circuit 253 → transmission signal correction management circuit 251 → transmission signal vector temporary storage unit 252 → matrix / vector multiplication circuit 253 → Execute the process while transferring the signals in order.

First, transmission signals _{t i} destined corresponding terminal device is input from the transmission signal processing circuit 241-1~241-L to the transmission signal correction management circuit 251. Transmission signal correction management circuit 251, the transmitted signal vector _{Tx = {t 1, t 2} , ..., t N} the combined all transmission signals _{t i} inputted initial value δTx transmission signal correction signal vector and ⁽⁰⁾ This is set and stored in the transmission signal vector temporary storage unit 252. Next, in order to sequentially solve the recurrence formulas represented by Expression (31), the matrix ΔW represented by Expression (32) is read from the transmission / reception weight calculation unit 290 and stored in the transmission signal correction matrix temporary storage circuit 254. Remember. In the matrix / vector multiplication circuit 253, the matrix ΔW stored in the transmission signal correction matrix temporary storage circuit 254 is multiplied by the transmission signal correction signal vector stored in the transmission signal vector temporary storage unit 252, thereby refining. The order of the expression is raised from the jth order to (j + 1) th order.

Further, the matrix / vector multiplication circuit 253 similarly returns the result to the transmission signal correction management circuit 251. The transmission signal correction management circuit 251 manages whether or not the approximation process from the number of loop processes to the target order k has been performed, and when the target order is reached, δTx ⁽⁰⁾ to δTx ^{(k )} Are added, and the respective components are output as corrected transmission signals to the transmission signal processing circuits 241-1 to 241-L.

The signal duplication circuit 206-i receives from the transmission signal correction unit 250 the i-th component of the corrected transmission signal vector ΣδTx ^(j) that is the sum of j from 0 to k. The signal duplication circuit 206-i outputs the corrected duplication of the transmission signal to the multiplication circuits 208-i-1 to 208-i-K. Multiplication circuits 208-i-1 to 208-i-K use the transmission weights of the antenna elements 101-1 to 101-K stored in the transmission weight vector temporary storage unit 207-i as corrected transmission signals. Multiply. Multiplication circuits 208-i-1 to 208 -i-K output the multiplication results to addition synthesis circuits 142-1 to 142 -K in transmission section 240.

  The matrix operation in the matrix / vector multiplication circuit 253 is not necessarily performed all at once, and a process of sequentially multiplying the matrix component and the vector component and adding them may be performed. For this reason, the transmission signal vector temporary storage unit 252 is provided as appropriate so that vector components can be read. However, the transmission signal vector temporary storage unit 252 is not necessarily provided when performing matrix operations in a batch. The signal correction management circuit 251 may directly input the matrix / vector multiplication circuit 253.

  Furthermore, as mentioned earlier, in a system that employs OFDM (A), SC-FDE, etc., each vector or matrix has a different value for each frequency component, and these processes are individually performed for all frequency components. Need to be done. They may be processed in parallel by individual circuits, or may be implemented serially using the same circuit.

<Configuration of transmission / reception weight calculation unit>
FIG. 38 is a schematic block diagram showing the configuration of the transmission / reception weight calculation unit 290 in the embodiment of the present invention. The transmission / reception weight calculation unit 290 in this embodiment has a configuration in which a function is added to the transmission / reception weight calculation unit 120b in FIG. Therefore, the same reference numerals are assigned to the same functions.

  As shown in the figure, the transmission / reception weight calculation unit 290 includes a channel information short-time average circuit 121, a relative component acquisition circuit 122, a channel information long-time average circuit 123, a reception weight calculation circuit 124b, a reception weight storage circuit 125b, and a calibration circuit. 126, transmission weight calculation circuit 127b, transmission weight storage circuit 128b, calibration coefficient storage circuit 129, reception signal correction information calculation circuit 291, reception signal correction information storage circuit 292, transmission signal correction information calculation circuit 293, transmission signal correction information storage A circuit 294 is included.

  The transmission / reception weight calculation unit 290 shown in the figure is different from the transmission / reception weight calculation unit 120b of the second configuration example in the related technology shown in FIG. 29 in that a reception signal correction information calculation circuit 291, a reception signal correction information storage circuit 292, A transmission signal correction information calculation circuit 293 and a transmission signal correction information storage circuit 294 are provided. Further, since the difference in signal processing is only processing related to these circuits, only the processing of the difference will be described here.

When the uplink channel information is acquired by the channel information long-time averaging circuit 123, the reception signal correction information calculation circuit 291 receives the input from the reception weight calculation circuit 124b separately from the calculation of the reception weight in the reception weight calculation circuit 124b. -W _i h _j / w corresponding to the (i, j) component of the matrix ΔW of Equation (39) using the received weight and the channel information of the uplink inputted from the channel information long-time averaging circuit 123 Information on _i h _i and information on w _i h _i or 1 / w _i h _i necessary for obtaining δRx ⁽⁰⁾ which is the initial condition of the recurrence formula (41) are calculated and received. It is stored in the received signal correction information storage circuit 292 as signal correction information. The reception signal correction information storage circuit 292 outputs reception signal correction information to the reception signal correction matrix temporary storage circuit 234 in the reception signal correction unit 230 as necessary in accordance with an instruction from the communication control circuit 110. In the description, the reception weight is input from the reception weight calculation circuit 124b in the reception signal correction information calculation circuit 291, but may be configured to be input from the reception weight storage circuit 125b.

In the case of using the reception weight of the formula (46), the product w _i h _i of the reception weight and channel information for a 1 as shown in equation (47), the matrix W _d ^-1 becomes an identity matrix In this case, δRx ⁽⁰⁾ matches Rx. In this case, information of the matrix W _d ^-1 in the process is not necessary, _(a product of the reception weight and channel information that is the same terminal device) product w _{i h} i of the reception weight and channel information in the transceiver weight calculator 290 , Storage of this information in the received signal correction information storage circuit 292, and information on the matrix W _d ⁻¹ from the received signal correction information storage circuit 292 to the received signal correction matrix temporary storage circuit 234 in the received signal correction unit 230 Input is optional.

Similarly, when downlink channel information is acquired by the calibration circuit 126, the transmission signal correction information calculation circuit 293 is input from the transmission weight calculation circuit 127b separately from the calculation of the transmission weight in the transmission weight calculation circuit 127b. -H _i w _j / h _i w _i corresponding to the (i, j) -th component of the matrix ΔW of Expression (32) using the transmission weight of the transmission channel and downlink channel information input from the calibration circuit 126 Is calculated and stored in the transmission signal correction information storage circuit 294 as transmission signal correction information. The transmission signal correction information storage circuit 294 outputs transmission signal correction information to the transmission signal correction matrix temporary storage circuit 254 in the transmission signal correction unit 250 as necessary in accordance with an instruction from the communication control circuit 110. In the description, the transmission weight is input from the transmission weight calculation circuit 127b in the transmission signal correction information calculation circuit 293, but may be configured to be input from the transmission weight storage circuit 128b.

Note that the matrix ΔW given by Equation (32) and Equation (39) is determined by the combination of terminal devices that simultaneously perform spatial multiplexing, but −w _i h _j corresponding to the (i, j) component of the matrix ΔW. / w _i h _i and -h i _w infos j _/ h _i w _i is independent of the combination patterns of all the terminal devices can be calculated corresponding to each pair of two terminal devices . That is, if information on -w _i h _j / w _i h _i and -h _i w _j / h _i w _i for each of these pairs is calculated, the information is appropriately selected and values are obtained for each component of the matrix. Can be used for an arbitrary combination pattern of a large number of terminal devices that simultaneously perform spatial multiplexing. That is, even if the combination pattern of terminal devices becomes an astronomical number as described above, it is possible to cope with the amount of information that can be accommodated with a realistic memory capacity.

<Receiver side signal correction processing flow>
FIG. 39 is a diagram showing a flow of received signal correction processing in the embodiment of the present invention. Basically, the received signal correction matrix ΔW represented by the equation (39) used for the received signal correction processing differs for each combination of terminal devices that simultaneously perform spatial multiplexing, and conversely, in signal processing related to a series of OFDM symbols. If there is no change in the combination of terminal devices that simultaneously perform spatial multiplexing, there is no need to replace the received signal correction matrix or the like. In the following description of the process, a process performed in units for replacing such a received signal correction matrix will be described.

When the processing is started (step S201), the reception signal correction matrix temporary storage circuit 234, according to an instruction from the communication control circuit 110, receives information on each component of the reception signal correction matrix -w _i h _j / w _i h _i and an initial value. reading receive signal correction information on _w i _{h i} or _{1 /} w _{i h} i necessary for obtaining the δRx ⁽⁰⁾ is a condition from the received signal correction information storage circuit 292 of the transmitting and receiving weight calculator 290 (step S202).
Thereafter, the symbol unit processing based on this information is started (step S203). Step S203 has no particular meaning in terms of execution, but is described in order to clearly indicate that processing in symbol units is performed with the subsequent step S211. As this symbol unit processing, the addition synthesis circuit 203-i in the reception signal processing circuit 209-i corresponding to the terminal device #i inputs the addition synthesis signal to the reception signal correction management circuit 231 (step S204).

The matrix / vector multiplication circuit 233 calculates an initial value δRx ⁽⁰⁾ based on the addition synthesis signal input from the addition synthesis circuits 203-1 to 203-L and the matrix W _d ⁻¹ (step S205).
In executing the process of step S205, the reception signal correction management circuit 231 temporarily stores the reception signal vector Rx first input from the reception signal processing circuits 209-1 to 209-L in the reception signal vector temporary storage unit 232. Let Matrix-vector multiplication circuit 233, w _i h _i or 1 / w _i h and the received signal correction information on _i, reception stored in the received signal vector temporary storage unit 232 stores the received signal correction matrix temporary storage circuit 234 Based on the signal vector Rx, the initial value δRx ⁽⁰⁾ is calculated by the equation (40).

In the case of using the reception weight of the formula (46), the product w _i h _i of the reception weight and channel information for a 1 as shown in equation (47), the matrix W _d ^-1 becomes an identity matrix In this case, δRx ⁽⁰⁾ matches Rx. In this case, information of the matrix W _d ⁻¹ is not necessary in the process, and w _i h _i or 1 / w _i h _i (that is, the same terminal device ⁾ necessary for obtaining δRx ⁽⁰⁾ in the process of step S202 Of the received signal correction information relating to the product of the received weight and the channel information) from the received signal correction information storage circuit 292 of the transmission / reception weight calculation unit 290 is omitted, and in the process of step S205, the initial value δRx is omitted. ⁽⁰⁾ is replaced with the processing given by Rx.

Here, the received signal correction management circuit 231 initializes j to manage the order of approximation of the recurrence formula to 0 (step S206), and the matrix / vector multiplication circuit 233 performs the recurrence formula of formula (41). Thus, δRx ^{(j + 1)} , which is a j + 1 order approximate solution, is calculated from δRx ^(j), which is a jth order approximate solution (step S207). Also in this process, the received signal correction signal vector δRx ^(j) managed by the received signal correction management circuit 231 is temporarily stored in the received signal vector temporary storage unit 232 and stored in the received signal correction matrix temporary storage circuit 234. Based on the received signal correction information related to the matrix ΔW and the received signal correction signal vector δRx ^(j) , the matrix / vector multiplication circuit 233 performs a matrix operation to calculate δRx ^{(j + 1)} .

Further, the received signal correction management circuit 231 adds j managing the approximate order of the recurrence formula to j + 1 (step S208), and checks whether the target value k of the approximate order and j match (step S208). S209) If they do not match (step S209: equivalent to NO), the process returns to step S207, and the processes from step S207 to step S209 are repeated up to the approximate target number k times.
On the other hand, when k and j match (corresponding to step S209: YES), the received signal correction management circuit 231 ends the recurrence processing, and adds δRx ^(j) from j to 0 to k, Each component of the added vector is output to the demodulation processing circuit 204-i in the reception signal processing circuits 209-1 to 209-L (step S210).

The above processing from step S203 to step S210 is performed on a symbol-by-symbol basis, but the received signal correction management circuit 231 has the entire range in which the same received signal correction matrix that continues spatial multiplexing with the same combination of terminal devices is reused. It is determined whether or not the symbol processing has been completed (step S211). If it has not been completed (step S211: corresponding to NO), the processing returns to step S203, and the processing from step S203 to step S211 is repeated.
On the other hand, if it is completed (corresponding to step S211: YES), the reception signal correction management circuit 231 ends the process (step S212).

  Note that, for simplicity of explanation, subcarrier number suffixes are omitted for each symbol of a matrix or a vector, but the above processing is performed when OFDM (A) or SC-FDE is used. The same implementation is performed for each subcarrier. In particular, when OFDMA is used, the OFDM symbol that uses the same received signal correction matrix ΔW may be different for each subcarrier. However, each time a condition for using a new received signal correction matrix ΔW is used for each subcarrier, processing is performed. Step S202 will be performed.

<Transmitter signal correction processing flow>
FIG. 40 is a diagram showing a flow of transmission signal correction processing in the embodiment of the present invention. Basically, the transmission signal correction matrix ΔW represented by the equation (32) used for the transmission signal correction processing differs for each combination of terminal devices that perform spatial multiplexing at the same time. In other words, in signal processing related to a series of OFDM symbols. If there is no change in the combination of terminal devices that simultaneously perform spatial multiplexing, there is no need to replace the transmission signal correction matrix or the like. In the following description of the processing, processing performed in units for replacing such a transmission signal correction matrix will be described.

When the processing is started (step S221), the transmission signal correction matrix temporary storage circuit 254, the transmission signal correction each component of the matrix of information _{-h i w j / h i w} i transmission signal correction information for transmission and reception weight calculating section 290 about Reading is performed from the transmission signal correction information storage circuit 294 (step S222).
Thereafter, the symbol unit processing based on this information is started (step S223). Step S223 has no particular meaning in terms of execution, but is described in order to clearly indicate that processing in symbol units is performed with the subsequent step S231. As the processing in units of symbols, the modulation processing circuits 205-1 to 205-L in the transmission signal processing circuits 241-1 to 241-L corresponding to the terminal devices that simultaneously perform spatial multiplexing perform transmission signal correction on the modulation processing results. Input to the management circuit 251 (step S224).

The transmission signal correction management circuit 251 sets the transmission signal vector Tx obtained by bundling the signals input from the modulation processing circuits 205-1 to 205-L to the initial value δTx ⁽⁰⁾ as it is (step S225).
Here, the transmission signal correction management circuit 251 initializes j, which manages the order of approximation of the recurrence formula, to 0 (step S226), and the matrix / vector multiplication circuit 253 performs the recurrence formula of the formula (31). Thus, δTx ^{(j + 1)} as the j + 1-order approximate solution is calculated from δTx ^{(j) as the} j-th order approximate solution (step S227). Also in this process, the transmission signal correction signal vector δTx ^(j) managed by the transmission signal correction management circuit 251 is temporarily stored in the transmission signal vector temporary storage unit 252 and stored in the transmission signal correction matrix temporary storage circuit 254. Based on the transmission signal correction information relating to the matrix ΔW and the transmission signal correction signal vector δTx ^(j) , the matrix / vector multiplication circuit 253 performs a matrix operation to calculate δTx ^{(j + 1)} .

Further, the transmission signal correction management circuit 251 adds j for managing the recursive approximation order to j + 1 (step S228), and checks whether the target value k of the approximation order matches j (step S228). If they do not match (step S229: equivalent to NO), the process returns to step S227, and the processes from step S227 to step S229 are repeated up to the approximate target number k times.
On the other hand, when k and j match (corresponding to step S229: YES), the transmission signal correction management circuit 251 finishes the recurrence processing, adds δTx ^(j) from j to 0 to k, Each component of the added vector is output as a corrected transmission signal to the signal duplicating circuit 206-i in the transmission signal processing circuits 141-1 to 141-L (step S230).

The above processing from step S223 to step S230 is performed on a symbol-by-symbol basis, but the transmission signal correction management circuit 251 uses the same transmission signal correction matrix that uses the same transmission signal correction matrix that continues spatial multiplexing with the same combination of terminal devices. It is determined whether or not the symbol processing has been completed (step S231). If the symbol processing has not been completed (step S231: equivalent to NO), the processing returns to step S223, and the processing from step S223 to step S231 is repeated.
On the other hand, if it is completed (corresponding to YES at step S231), the transmission signal correction management circuit 251 ends the process (step S232).

  Note that, for simplicity of explanation, subcarrier number suffixes are omitted for each symbol of a matrix or a vector, but the above processing is performed when OFDM (A) or SC-FDE is used. The same implementation is performed for each subcarrier. In particular, when OFDMA is used, the OFDM symbol that uses the same received signal correction matrix ΔW may be different for each subcarrier, but the processing step S222 is performed each time a condition for using a new received signal correction matrix ΔW is satisfied. Will do.

<Processing flow of transmission / reception weight calculation unit>
FIG. 41 shows a processing flow related to the received signal correction information of the transmission / reception weight calculation unit 290 in the embodiment of the present invention. This processing flow is performed together with a process of calculating transmission / reception weights performed in advance before the start of service operation. Further, (i ′, j ′) in the notation of the combination of the i′-th terminal device and the j-th terminal device used in the following description is the serial number i ′, j ′ of the terminal device arbitrarily assigned in the system. Means a combination. Note that (i, j) in the description related to the equations (32) and (39) is based on serial numbers sequentially allocated only to the terminal devices in the combination pattern of the terminal devices simultaneously spatially multiplexed. This corresponds to the combination of the i-th and j-th terminal devices to be designated. Therefore, in order to use equations (32) and (39), the components (i, j) of the matrix and (i ′, j ′) given by the original serial number arbitrarily assigned in the system A correspondence is taken, a value corresponding to (i ′, j ′) is read from the received signal correction information storage circuit 292, set in each component of each matrix, and used. In addition, if information corresponding to all the combinations (i ′, j ′) is obtained in advance by performing such management, by using the values, it is possible to arbitrarily select the terminal devices that perform spatial multiplexing simultaneously. It is possible to easily obtain these matrices for the combination patterns.

  When the transmission / reception weight calculation unit 290 starts processing related to the received signal correction information (step S261), the received signal correction information calculation circuit 291 in the series of processing is performed by the channel information long-time averaging circuit 123. Uplink channel information obtained by performing the averaging process for a long time is collected (step S262). The reception signal correction information calculation circuit 291 collects the reception weight information calculated by the reception weight calculation circuit 124b (step S263).

When the reception signal correction information calculation circuit 291 has collected a series of information, 1 / w _{i ′} h _i from the uplink channel information vector h _{i ′} and the reception weight vector w _{i ′} corresponding to the i th terminal device. _' Is calculated (step S264).
Further, the received signal correction information calculation circuit 291 calculates w _{i ′} h _{j ′} for all combinations of (i ′, j ′) of the i ′ ′ terminal device and the j ′ terminal device (step S265). Then, −w _{i ′} h _{j ′} / w _{i ′} h _{i ′} which is each component of the matrix represented by the equation (39) is calculated (step S266).

Thereafter, the reception signal correction information calculation circuit 291 receives the information 1 / w _{i ′} h _{i ′} calculated in step S264 and the information −w _{i ′} h _{j ′} / w _{i ′} h _{i ′} calculated in step S266. The signal correction information storage circuit 292 is stored (step S267), and the process is terminated (step S268).
In addition, although step S265 and step S266 were described separately as what implements step S266 using the result of step S264 and step S265 here, it is good also as step S266 collectively.
Further, when the reception weight represented by Expression (46) is used, the product of the reception weight and the channel information w _{i ′} hi _{i ′} is 1 as shown in Expression (47), and therefore the process of Step S264 can be omitted. It is. In addition, step S266 is replaced with a calculation process of -w _{i ′} h _{j ′} . Further, in the storage process to the received signal correction information storage circuit 292 in step S267, it is not necessary to store information on 1 / w _{i ′} h _{i ′} , and information −w _{i ′} h _{j ′} / w _{i ′} h _i Instead of _' information-w _i' h _{j '} is stored.

FIG. 42 shows a processing flow relating to transmission signal correction information of the transmission / reception weight calculation unit 290 in the embodiment of the present invention. Similar to the above description, this processing flow is processing performed in advance before starting service operation.
When the transmission / reception weight calculation unit 290 starts processing related to transmission signal correction information (step S271), the transmission signal correction information calculation circuit 293 is obtained by the calibration circuit 126 in these series of processing. Downlink channel information is collected (step S272). The transmission signal correction information calculation circuit 293 collects the transmission weight information calculated by the transmission weight calculation circuit 127b (step S273).

When the transmission signal correction information calculation circuit 293 finishes collecting a series of information, 1 / h _{i ′} w _i from the downlink channel information vector h _{i ′} and the transmission weight vector w _{i ′} corresponding to the i th terminal device. _' Is calculated (step S274).
Further, the transmission signal correction information calculation circuit 293 calculates h _{i ′} w _{j ′} for all (i ′, j ′) combinations (step S275), and each of the matrices represented by the equation (32). The component -h _{i '} w _j' / h _{i '} w _i' is calculated (step S276).

Thereafter, the transmission signal correction information calculation circuit 293 stores the information −h _{i ′} w _{j ′} / h _{i ′} w _{i ′} calculated in step S276 in the transmission signal correction information storage circuit 294 (step S277), and ends the processing. (Step S278).
Here, step S275 and step S276 are described separately as performing step S276 using the results of step S274 and step S275, but may be combined into step S276.

  As described earlier, sub-carrier number suffixes are omitted in the matrix and vector symbols for the sake of simplicity, but the above processing is performed by OFDM (A) or SC-FDE. When using, the same processing is performed on each subcarrier in the signal processing of FIGS. 41 and 42 described above.

  The above is the processing for calculating information necessary for correcting the transmission / reception signal. The information obtained as a result of these processes can be used in combination patterns of various terminal devices that perform spatial multiplexing. For example, the (i, j) component of the matrix represented by Equation (32) or Equation (39) is It is possible to read out from the received signal correction information storage circuit 292 or the transmission signal correction information storage circuit 294 by designating a combination of two terminal devices.

  The combination pattern of the terminal devices that perform spatial multiplexing at the same time is determined by the scheduling performed by the scheduling processing circuit 181. Upon receiving the scheduling result, the combination pattern of the terminal devices is transmitted to the transmission / reception weight calculation unit 290 via the communication control circuit 110. When instructed, the transmission / reception weight calculation unit 290 takes a correspondence between (i, j) which is a matrix component and (i ′, j ′) which is a serial number of the corresponding terminal device, and corresponds to the combination pattern. The reception signal correction information or the transmission signal correction information is output to the reception signal correction unit 230 or the transmission signal correction unit 250. The output information is stored in the reception signal correction matrix temporary storage circuit 234 or the transmission signal correction matrix temporary storage circuit 254 and is used for the recurrence calculation. The correspondence between the matrix component (i, j) and the corresponding terminal device serial number (i ′, j ′) may be managed by the communication control circuit 110 and instructed. Is possible.

  In the above description of the embodiment of the present invention, the serial numbers of the terminal devices arbitrarily allocated in the system, the serial numbers of the terminal devices sequentially allocated to the terminal devices that perform spatial multiplexing simultaneously (serial numbers representing the components of the matrix), the repetition row Variables such as i, j, k, i ′, j ′ in the order indicating the intermediate stage of the recurrence formula processing, the target number of repetitions of the recurrence formula, the frequency component number (subcarrier number), etc. However, for convenience of explanation, the same symbol is used as a variable having a different meaning. Therefore, the meaning of these variables should be understood as directed by each individual description.

Prior to actual data communication, the base station apparatus 200 according to the present embodiment acquires channel information by receiving a predetermined training signal from each terminal apparatus, and further acquires multiple times at different timings. The channel information is accurately estimated by averaging the relative components of the received channel information, and the reception weight used when simultaneously multiplexing multiple terminal devices simultaneously in actual data communication using this channel information is calculated in advance. The reception processing is performed using this spatial multiplexing reception weight.
Furthermore, the base station apparatus 200 repeatedly calculates a recurrence formula for each signal sequence spatially multiplexed on the received signal while using reception weights that do not depend on the combination of terminal devices, so that interference components between the signal sequences can be obtained. Can be suppressed, and more accurate signal detection processing can be performed. As a result, good SIR characteristics can be achieved even if the number of signal systems performing spatial multiplexing simultaneously increases, and higher frequency utilization efficiency can be realized.

[Other supplementary items]
Note that a program for realizing the functions of the base station apparatus according to the present invention is recorded on a computer-readable recording medium, and the program recorded on the recording medium is read into a computer system and executed, thereby transmitting weights. In addition, a process of calculating a reception weight and a transmission / reception weight may be performed. Here, the “computer system” includes an OS and hardware such as peripheral devices. The “computer system” includes a WWW system having a homepage providing environment (or display environment). The “computer-readable recording medium” refers to a storage device such as a flexible medium, a magneto-optical disk, a portable medium such as a ROM and a CD-ROM, and a hard disk incorporated in a computer system. Further, the “computer-readable recording medium” refers to a volatile memory (RAM) in a computer system that becomes a server or a client when a program is transmitted via a network such as the Internet or a communication line such as a telephone line. In addition, those holding programs for a certain period of time are also included.

  The program may be transmitted from a computer system storing the program in a storage device or the like to another computer system via a transmission medium or by a transmission wave in the transmission medium. Here, the “transmission medium” for transmitting the program refers to a medium having a function of transmitting information, such as a network (communication network) such as the Internet or a communication line (communication line) such as a telephone line. The program may be for realizing a part of the functions described above. Furthermore, what can implement | achieve the function mentioned above in combination with the program already recorded on the computer system, and what is called a difference file (difference program) may be sufficient.

1-1, 1-2, 1-3 OFDM symbol 2-1, 2-2, 2-3 Effective signal region 3-1, 3-2, 3-3 Training signal 4-1, 4-2, 4 -3 Trailing area 5-1, 5-2, 5-3 Guard interval 6-1, 6-26-3 Signal period 10 Base station apparatus 11 Building 12-1, 12-2 Terminal apparatus 13-1, 13- 2, 13-3, 13-4 Base station equipment antenna elements 14-1, 14-2, 14-3 Ground mobile units 15-1, 15-2, 15-3 Building 16 Long-term average channel information Vectors 17-1, 17-2, 17-3, 17-4 corresponding to short-time average channel information 18 Channel estimation error range 21-1, 21-2, 21-3 High power amplifier ( HPA)
22-1, 22-2, 22-3 Low noise amplifier (LNA)
23-1, 23-2, 23-3 Time division switch (TDD-SW)
24-1, 24-2, 24-3 Antenna elements 25-1, 25-2, 25-3 Radio module 26-1, 26-2, 26-3 Antenna terminal 27 Coaxial cable 80 Base station device 81 Transmitter 85 Receiving unit 87 Interface circuit 88 MAC layer processing circuit 100 Receiving unit 101-1, 101-2, 101-K Antenna element 102-1, 102-2, 102-K TDD switch 103-1, 103-2, 103-K Low noise amplifier (LNA)
104 local oscillators 105-1, 105-2, 105-K mixers 106-1, 106-2, 106-K filters 107-1, 107-2, 107-K A / D converters 108-1, 108-2 108-K FFT circuit 109-1, 109-2, 109-L, 109-i received signal processing circuit (received signal processing unit)
110 Communication control circuit (communication control unit)
115 Storage Circuit 120a, 120b Transmission / Reception Weight Calculation Unit 121 Channel Information Short-Time Average Circuit 122 Relative Component Acquisition Circuit 123 Channel Information Long-Time Average Circuit 124a Multiuser MIMO Reception Weight Calculation Circuit (Reception Weight Calculation Unit)
124b Reception weight calculation circuit (reception weight calculation unit)
125a Multi-user MIMO reception weight storage circuit 125b Reception weight storage circuit (reception weight storage unit)
126 Calibration circuit (downlink channel information calculation unit)
127a Multi-user MIMO transmission weight calculation circuit (transmission weight calculation unit)
127b Transmission weight calculation circuit (transmission weight calculation unit)
128a Multi-user MIMO transmission weight storage circuit 128b Transmission weight storage circuit 129 Calibration coefficient storage circuit (calibration coefficient storage unit)
140 Transmitters 141-1, 141-2, 141-L, 141-i Transmission signal processing circuit (transmission signal processing unit)
142-1, 142-2, 142-K addition synthesis circuit 143-1, 143-2, 143-K IFFT & GI adding circuit 144-1, 144-2, 144-K D / A converter 145 local oscillator 146-1 146-2, 146-K mixers 147-1, 147-2, 147-K filters 148-1, 148-2, 148-K high power amplifiers (HPA)
170 Interface Circuit 180 MAC Layer Processing Circuit 181 Scheduling Processing Circuit (Scheduling Processing Unit)
200 base station apparatus 201-i-1, 201-i-2, 201-i-K multiplication circuit 202-i reception weight vector temporary storage unit 203-i addition synthesis circuit 204-i demodulation processing circuit 205-i modulation processing circuit 206-i Signal Duplicating Circuit 207-i Transmission Weight Vector Temporary Storage Unit 208-i-1, 208-i-2, 208-i-K Multiplication Circuit 209-i Reception Signal Processing Circuit 210 Reception Unit 230 Reception Signal Correction Unit 231 Reception signal correction management circuit 232 Reception signal vector temporary storage unit 233 Matrix / vector multiplication circuit 234 Reception signal correction matrix temporary storage circuit 240 Transmission unit 241-i Transmission signal processing circuit 250 Transmission signal correction unit 251 Transmission signal correction management circuit 252 Transmission signal Vector temporary storage unit 253 Matrix / vector multiplication circuit 254 Transmission signal correction matrix temporary storage circuit 2 0 transceiver weight calculating unit 291 receives the signal correction information calculating circuit (received signal correction information calculator)
292 Received signal correction information storage circuit (received signal correction information storage unit)
293 transmission signal correction information calculation circuit (transmission signal correction information calculation unit)
294 Transmission signal correction information storage circuit (transmission signal correction information storage unit)
801 Base station device 802-1, 802-2, 802-3 Terminal device 811-1, 811-2, 811-L Transmission signal processing circuit 812-1, 812-2, 812-K Addition / synthesis circuit 813-1 813-2, 813 -K IFFT & GI adding circuit 814-1, 814-2, 814 -K D / A converter 815 Local oscillator 816-1, 816-2, 816 -K mixer 817-1, 817-2, 817 -K filters 818-1, 818-2, 818-K high power amplifier (HPA)
819-1, 819-2, 819-K Antenna element 820 Communication control circuit 830 Transmission weight processing unit 831 Channel information acquisition circuit 832 Channel information storage circuit 833 Multi-user MIMO transmission weight calculation circuit 851-1, 851-2, 851- K antenna element 852-1, 852-2, 852-K Low noise amplifier (LNA)
853 Local oscillators 854-1, 854-2, 854-K mixers 855-1, 855-2, 855-K filters 856-1, 856-2, 856-K A / D converters 857-1, 857-2 857-K FFT circuit 858-1, 858-2, 858-L reception signal processing circuit 860 reception weight processing unit 861 channel information estimation circuit 862 multiuser MIMO reception weight calculation circuit 881 scheduling processing circuit 901 transmitting station 902-1 902-N ₁ relay station 903 receiving station 911-1, 911-2, 911-3 cell 912-1, 912-2, 912-3 remote base station 913-1, 913-2, 913-3, 913-4 , 913-5, 913-6 terminal device 914 control station 915 optical fiber 961-1, 961-2, 96 -3,961-4,961-5 antenna element

Claims (7)

A base station apparatus including a plurality of antenna elements; and a plurality of terminal apparatuses that perform wireless communication with the base station apparatus, wherein the base station apparatus and at least two of the terminal apparatuses are at the same time on the same frequency component. A base station apparatus in a wireless communication system capable of performing spatial multiplexing transmission,
A scheduling processing unit for selecting a combination of the terminal devices that perform spatial multiplexing transmission at the same time on the same frequency component;
An uplink channel information acquisition unit that acquires channel information of each frequency component in the uplink between the terminal device and each of the plurality of antenna elements based on a training signal received from the terminal device for each terminal device When,
Based on the channel information in the uplink of each frequency component for each terminal device acquired by the uplink channel information acquisition unit or the physical quantity calculated from the channel information, the space in each frequency component for each of the plurality of antenna elements A reception weight calculation unit for calculating a reception weight for multiplex transmission;
A reception weight storage unit for storing the reception weight calculated by the reception weight calculation unit;
A reception signal correction information calculation unit that calculates reception signal correction information based on the reception weight calculated by the reception weight calculation unit and the channel information acquired by the uplink channel information acquisition unit;
A reception signal correction information storage unit for storing the reception signal correction information calculated by the reception signal correction information calculation unit;
For each antenna element, the received signal received from the terminal device via the antenna element is separated for each frequency component, and the received signal for each separated frequency component is read from the received weight storage unit. A reception signal processing unit that multiplies the reception weight of each frequency component and adds and combines the multiplied signals over all or some of the antenna elements for each frequency component; ,
Based on the received signal correction information stored in the received signal correction information storage unit, the added combined signal acquired by the received signal processing unit corresponding to the terminal device performing spatial multiplexing transmission at the same time on the same frequency component is corrected. A received signal correction unit,
With
The received signal processor is
A base station apparatus, wherein a signal transmitted by the terminal apparatus is detected from a signal corrected by the reception signal correction unit. The base station apparatus according to claim 1,
The i-th terminal device (1 ≦ i ≦ N, i is an integer) and a plurality of devices included in the own device with respect to the terminal device of N (N is a positive integer) that simultaneously performs spatial multiplexing at a predetermined frequency component at a certain time When the channel information vector in the uplink configured by channel information in the predetermined frequency component between the antenna element and h _Ui is set as h _Ui and the reception weight vector corresponding to the channel information vector is set as w _Ui ,
In the received signal correction unit,
The received signal correction information includes a matrix W _Ud ⁻¹ whose components are reciprocals of the product w _i h _i of two vectors corresponding to the i th terminal device, and the j th (1 ≦ j ≦ N, i ≠). j, j are integers) and a value −w _Ui h _Uj / w _Ui h _Ui given by vector calculation of the channel information vector and the reception weight vector corresponding to the i-th terminal device is the (i, j) -th component. And the diagonal component (i = j) is a matrix ΔW _U given by all zeros,
When a signal vector whose component is an added composite signal acquired by the received signal processing unit corresponding to the terminal apparatus that simultaneously performs spatial multiplexing at a predetermined frequency component at a certain time is denoted by Rx,
The 0th-order initial vector δRx ⁽⁰⁾ is given by the following equation (A):

The recurrence formula between the kth order (0 ≦ k, k is an integer) and the (k + 1) th order vector is given by the following formula (B):

After calculating the vector up to the target order α-order (1 ≦ α, α is an integer) by the recurrence formula of Formula (B), the vector sum from vector δRx ^{( 0)} to vector δRx ^(α) is calculated. The base station apparatus characterized in that the calculated vector is output to the reception signal processing unit as a signal for each of the corrected frequency components. The base station apparatus according to claim 1 or 2, wherein
The i-th terminal device (1 ≦ i ≦ N, i is an integer) and a plurality of devices included in the own device with respect to the terminal device of N (N is a positive integer) that simultaneously performs spatial multiplexing at a predetermined frequency component at a certain time When the channel information vector in the uplink configured by channel information in the predetermined frequency component between the antenna element and h _Ui is set as h _Ui and the reception weight vector corresponding to the channel information vector is set as w _Ui ,
In the received signal correction unit,
The received signal correction information is given by vector calculation of the channel information vector and the received weight vector corresponding to the j-th (1 ≦ j ≦ N, i ≠ j, j is an integer) and the i-th terminal device. The value −w _Ui h _Uj is the (i, j) component and the diagonal component (i = j) is a matrix ΔW _U given by all zeros;
When a signal vector whose component is an added composite signal acquired by the received signal processing unit corresponding to the terminal apparatus that simultaneously performs spatial multiplexing at a predetermined frequency component at a certain time is denoted by Rx,
The zeroth-order initial vector δRx ⁽⁰⁾ is given by the following equation (C):

The recurrence formula between the k th order (0 ≦ k, k is an integer) and the (k + 1) th order vector is given by the following formula (D):

After calculating the vector up to the target order α-order (1 ≦ α, α is an integer) by the recurrence formula of Formula (D), the vector sum from vector δRx ^{( 0)} to vector δRx ^(α) is calculated. , Outputting the calculated vector as a signal for each corrected frequency component to the received signal processing unit,
A reception weight w _Ui of a certain frequency component calculated by the reception weight calculation unit with respect to the i-th terminal device is given by the following equation (E).

A base station apparatus. The base station apparatus according to any one of claims 1 to 3,
A calibration coefficient storage that stores, for each frequency element, a calibration coefficient used for calculating downlink channel information from uplink channel information between the antenna element and the terminal device. And
Each uplink channel information of each frequency component for each of the antenna elements calculated by the uplink channel information acquisition unit for each terminal device is multiplied by a calibration coefficient corresponding to the combination of the antenna element and the frequency component to be down. A downlink channel information calculation unit for calculating channel information in the link;
A transmission weight calculation unit for calculating a transmission weight for spatial multiplexing transmission of each frequency component for each antenna element from channel information of each frequency component for each antenna element in the downlink calculated by the downlink channel information calculation unit When,
A transmission weight storage unit for storing the reception weight calculated by the transmission weight calculation unit;
A transmission signal correction information calculation unit for calculating transmission signal correction information based on the transmission weight calculated by the transmission weight calculation unit and the channel information calculated by the downlink channel information calculation unit;
A transmission signal correction information storage unit for storing transmission signal correction information calculated by the transmission signal correction information calculation unit;
For each of the terminal devices selected as data transmission destinations, a transmission signal processing unit that generates a transmission signal to be transmitted to the terminal device and separates it into signals for each frequency component;
Based on transmission signal correction information stored in the transmission signal correction information storage unit, a signal for each frequency component separated by the transmission signal processing unit corresponding to the terminal device performing spatial multiplexing transmission at the same time on the same frequency component. A transmission signal correction unit for correcting
Further comprising
The transmission signal processing unit
A signal for each frequency component obtained by reading out the transmission weight of each frequency component corresponding to the terminal device as a data transmission destination from the transmission weight storage unit, and correcting the transmission weight of each read frequency component by the transmission signal correction unit. The base station apparatus, wherein the base station apparatus multiplies and transmits from the plurality of antenna elements. The base station apparatus according to claim 4, wherein
For the terminal device of N (N is a positive integer) station that performs spatial multiplexing simultaneously at a predetermined frequency component at a certain time, the i-th terminal device (1 ≦ i ≦ N, i is an integer) and the base station device When the channel information vector in the downlink configured by the channel information in the predetermined frequency component between the plurality of antenna elements provided is h _Di and the transmission weight vector corresponding to the channel information vector is w _Di ,
In the transmission signal correction unit,
The transmission signal correction information includes j-th (1 ≦ j ≦ N, i ≠ j, j is an integer) corresponding to the j-th terminal device, the channel information vector corresponding to the i-th terminal device, and The value −h _Di w _Dj / h _Di w _Di given by the vector calculation of the transmission weight vector is the matrix (ΔW _D ) with the (i, j) component and the diagonal components (i = j) all given by zero,
When a signal vector having a signal for each frequency component separated by the transmission signal processing unit corresponding to the terminal device that performs spatial multiplexing simultaneously at a predetermined frequency component at a certain time is set as Tx,
The zeroth-order initial vector δTx ⁽⁰⁾ is given by the following equation (F):

The recurrence formula between the k th order (0 ≦ k, k is an integer) and the (k + 1) th order vector is given by the following formula (G):

After calculating the vector up to the target order α order (1 ≦ α, α is an integer) by the recurrence formula of Expression (G), the vector sum from the vector δTx ^{( 0)} to the vector δTx ^(α) is calculated. The calculated vector is output to the transmission signal processing unit as a signal for each corrected frequency component. A base station apparatus including a plurality of antenna elements; and a plurality of terminal apparatuses that perform wireless communication with the base station apparatus, wherein the base station apparatus and at least two of the terminal apparatuses are at the same time on the same frequency component. A wireless communication method in a wireless communication system capable of performing spatial multiplexing transmission,
A scheduling process step of selecting a combination of the terminal devices performing spatial multiplexing transmission at the same time on the same frequency component;
Uplink channel information acquisition step for acquiring channel information of each frequency component in the uplink between the terminal device and each of the plurality of antenna elements based on a training signal received from the terminal device for each terminal device When,
Based on the channel information in the uplink of each frequency component for each of the terminal devices acquired in the uplink channel information acquisition step or the physical quantity calculated from the channel information, the space in each frequency component for each of the plurality of antenna elements A reception weight calculating step for calculating a reception weight for multiplex transmission;
A reception weight storage step of storing the reception weight calculated in the reception weight calculation step in a reception weight storage unit;
A reception signal correction information calculation step for calculating reception signal correction information based on the reception weight calculated in the reception weight calculation step and the channel information acquired in the uplink channel information acquisition step;
A reception signal correction information storage step for storing the reception signal correction information calculated in the reception signal correction information calculation step in a reception signal correction information storage unit;
For each antenna element, the received signal received from the terminal device via the antenna element is separated for each frequency component, and the received signal for each separated frequency component is stored in the received weight storage unit Reception signal processing for obtaining, for each frequency component, an added synthesized signal obtained by multiplying the received weights of each frequency component corresponding to the frequency component and adding and synthesizing the multiplied signals over all or some of the antenna elements. Steps,
Based on the received signal correction information stored in the received signal correction information storage unit, the added composite signal acquired in the received signal processing step corresponding to the terminal device performing spatial multiplexing transmission at the same time on the same frequency component is corrected. A received signal correction step,
And a signal detection step of detecting a signal transmitted by the terminal device from the signal corrected in the reception signal correction step. A base station apparatus including a plurality of antenna elements; and a plurality of terminal apparatuses that perform wireless communication with the base station apparatus, wherein the base station apparatus and at least two of the terminal apparatuses are at the same time on the same frequency component. A wireless communication system capable of performing spatial multiplexing transmission,
The base station device
A scheduling processing unit for selecting a combination of the terminal devices that perform spatial multiplexing transmission at the same time on the same frequency component;
An uplink channel information acquisition unit that acquires channel information of each frequency component in the uplink between the terminal device and each of the plurality of antenna elements based on a training signal received from the terminal device for each terminal device When,
Based on the channel information in the uplink of each frequency component for each terminal device acquired by the uplink channel information acquisition unit or the physical quantity calculated from the channel information, the space in each frequency component for each of the plurality of antenna elements A reception weight calculation unit for calculating a reception weight for multiplex transmission;
A reception weight storage unit for storing the reception weight calculated by the reception weight calculation unit;
A reception signal correction information calculation unit that calculates reception signal correction information based on the reception weight calculated by the reception weight calculation unit and the channel information acquired by the uplink channel information acquisition unit;
A reception signal correction information storage unit for storing the reception signal correction information calculated by the reception signal correction information calculation unit;
For each antenna element, the received signal received from the terminal device via the antenna element is separated for each frequency component, and the received signal for each separated frequency component is read from the received weight storage unit. A reception signal processing unit that multiplies the reception weight of each frequency component and adds and combines the multiplied signals over all or some of the antenna elements for each frequency component; ,
Based on the received signal correction information stored in the received signal correction information storage unit, the added combined signal acquired by the received signal processing unit corresponding to the terminal device performing spatial multiplexing transmission at the same time on the same frequency component is corrected. A received signal correction unit,
With
The received signal processor is
A radio communication system, wherein a signal transmitted by the terminal device is detected from a signal corrected by the reception signal correction unit.

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