JP5672844B2 - Wireless power transmission system - Google Patents

Description

  The present invention relates to wireless power feeding, and more particularly to power control thereof.

  Wireless power supply technology that supplies power without a power cord is drawing attention. Current wireless power transfer technologies are (A) a type that uses electromagnetic induction (for short distance), (B) a type that uses radio waves (for long distance), and (C) a type that uses magnetic field resonance (medium distance). Can be roughly divided into three types.

  The type (A) using electromagnetic induction is generally used in household appliances such as an electric shaver. However, if the distance is increased, the power transmission efficiency is drastically reduced, so that the short distance is about several centimeters. There is a problem that can only be used in. The type (B) using radio waves can be used at a long distance, but has a problem that power is small. The type (C) using the magnetic field resonance phenomenon is a relatively new technology, and is particularly expected from the fact that high power transmission efficiency can be realized even at a middle distance of about several meters. For example, a proposal has been studied in which a receiving coil is embedded in the lower part of an EV (Electric Vehicle) and electric power is sent in a non-contact manner from a power feeding coil in the ground. Since it is wireless, a completely insulated system configuration is possible, and it is considered to be particularly effective for power supply in rainy weather. Hereinafter, the type (C) is referred to as “magnetic field resonance type”.

  The magnetic resonance type is based on a theory published by Massachusetts Institute of Technology in 2006 (see Patent Document 1). In Patent Document 1, four coils are prepared. These coils are called “exciting coil”, “power feeding coil”, “power receiving coil”, and “load coil” in order from the power feeding side. The exciting coil and the feeding coil face each other at a short distance and are electromagnetically coupled. Similarly, the power receiving coil and the load coil are also faced at a short distance and are electromagnetically coupled. Compared to these distances, the distance from the feeding coil to the receiving coil is a “medium distance”, which is relatively large. The purpose of this system is to wirelessly feed power from the feeding coil to the receiving coil.

  When AC power is supplied to the exciting coil, current also flows through the feeding coil due to the principle of electromagnetic induction. When the power feeding coil generates a magnetic field and the power feeding coil and the power receiving coil resonate magnetically, a large current flows through the power receiving coil. Due to the principle of electromagnetic induction, a current also flows through the load coil, and AC power is extracted from a load connected in series with the load coil. By using the magnetic field resonance phenomenon, high power transmission efficiency can be realized even if the distance from the feeding coil to the receiving coil is large.

US Publication No. 2008/0278264 JP 2006-230032 A International Publication No. 2006/022365 US Publication No. 2009/0072629 US Publication No. 2009/0015075 JP 2008-172872 A JP 2006-74848 A JP 2003-33011 A

  The present inventor considers that in order to expand the availability of wireless power feeding, a mechanism for automatically controlling the power feeding power and stabilizing the output is necessary. The non-contact power transmission device disclosed in Patent Document 7 is of the type (A), but the secondary unit on the power receiving side transmits the magnitude of the output voltage to the primary unit on the power transmitting side. Controls the power supply according to the output voltage. Specifically, a signal indicating the magnitude of the output voltage is transmitted from the coil L4 (secondary unit) to the coil L3 (primary unit).

  The non-contact power transmission device of Patent Document 7 is based on an implicit premise that the resonance frequency of the primary side series resonance circuit or the secondary side series resonance circuit is constant. However, in the case of the magnetic field resonance type, the resonance frequency is likely to change due to the positional relationship between the feeding coil and the power receiving coil, and therefore it is not practical to apply the mechanism of Patent Document 7 as it is to the magnetic field resonance type. In the case of the magnetic field resonance type, it is considered that the magnetic field generated by the power feeding coil and the power receiving coil has a great influence on signal transmission by electromagnetic waves from the coils L4 to L3.

  The main object of the present invention is to efficiently control power supply in a magnetic field resonance type wireless power supply.

  A wireless power feeding apparatus according to the present invention is an apparatus for wirelessly feeding power from a power feeding coil to a power receiving coil. This device includes a power transmission control circuit that supplies AC power to a power supply coil at a driving frequency, a power supply coil circuit that includes a power supply coil and a capacitor and resonates at a resonance frequency of the power reception coil, and a voltage phase and a current phase of the AC power. A phase detection circuit that detects a phase difference and a signal reception circuit that receives an output signal indicating an output by a duty ratio from the AC power receiving side and converts the output signal into a direct current according to the duty ratio. The power transmission control circuit causes the drive frequency to follow the resonance frequency by adjusting the drive frequency so that the phase difference decreases. Further, the phase detection circuit performs post-adjustment on the phase value detected for both or one of the voltage phase and the current phase according to the signal level of the output signal that has been DC converted.

  The drive frequency can be made to follow the resonance frequency by comparing the current phase and the voltage phase of the AC power, detecting the phase difference, and adjusting the drive frequency so that the phase difference is reduced. As a result, even if the resonance frequency changes, the power transmission efficiency can be easily maintained constant. Further, when the output voltage or the like changes, if the voltage phase or the current phase is adjusted afterwards according to the amount of change, the drive frequency changes according to the adjusted phase difference. Since the feed power can be feedback controlled using the drive frequency as a parameter, the output is easily stabilized.

  The phase detection circuit compares the first phase value, which is the timing when the voltage level of the AC power becomes the first reference value, and the second phase value, which is the timing when the current level of the AC power becomes the second reference value. By detecting the phase difference, both or one of the first and second phase values may be post-adjusted by changing both or one of the first and second reference values according to the signal level. . The signal receiving circuit may receive the output signal as an optical signal such as infrared rays.

  The apparatus may further include an exciting coil that is magnetically coupled to the power supply coil and supplies AC power supplied from the power transmission control circuit to the power supply coil. The power transmission control circuit includes first and second current paths, and alternately turns on the first and second switches connected in series to the first and second current paths, respectively, at the drive frequency. AC power may be supplied to the exciting coil.

  The apparatus may further include a detection coil that generates an induced current by a magnetic field generated by AC power. The phase detection circuit may measure the current phase of the AC power from the phase of the induced current flowing through the detection coil. Since the current phase is measured from the induction current of the detection coil, it is difficult to apply a direct measurement load to the feeding coil. The detection coil may generate an induced current by a magnetic field generated by an alternating current flowing through the feeding coil.

  A wireless power receiving device according to the present invention is a device that receives AC power wirelessly fed from the above-described wireless power feeding device using a power receiving coil. This device includes a power receiving coil and a capacitor, and includes a power receiving coil circuit that resonates at the resonance frequency of the power feeding coil, a load coil that is magnetically coupled to the power receiving coil, and receives AC power from the power receiving coil, and supplies power from the load coil. And a signal transmission circuit that transmits an output signal indicating an output voltage applied to a part of the load circuit by a duty ratio to the wireless power feeding apparatus.

  The signal transmission circuit may transmit the output signal as a signal indicating a difference value between the output voltage and the reference voltage by a duty ratio. The reference voltage may be manually adjustable.

  The apparatus further includes a control signal generation circuit that generates a control signal at a predetermined control frequency, and a comparison circuit that generates an effective signal when a predetermined magnitude relationship is established between the signal level of the control signal and the output voltage. You may prepare. The signal transmission circuit may determine the duty ratio of the output signal based on the duty ratio of the effective signal. The “predetermined magnitude relationship” may be, for example, when the level of the control signal is higher than the level of the predetermined voltage indicating the fluctuation of the output voltage, or vice versa.

  The apparatus may further include a reference signal generation circuit that generates a reference signal having a reference frequency higher than the control frequency. The signal transmission circuit may transmit the reference signal as an output signal on condition that the effective signal is being generated.

  A wireless power transmission system according to the present invention is a system for wirelessly feeding power from a feeding coil to a receiving coil. This system includes a power transmission control circuit that supplies AC power to a power feeding coil at a driving frequency, a power feeding coil circuit that includes a power feeding coil and a first capacitor, a power receiving coil circuit that includes a power receiving coil and a second capacitor, A load circuit including a load coil that receives AC power from a receiving coil by magnetic coupling with the coil and a load that is supplied with power from the load coil, and a phase that detects a phase difference between the voltage phase and the current phase of the AC power A detection circuit, a signal transmission circuit for transmitting an output signal indicating a duty ratio of an output voltage applied to a part of the load circuit to the power supply side, an output signal received at the power supply side, and an output signal according to the duty ratio Is provided with a signal receiving circuit for direct-current conversion. The power transmission control circuit adjusts the drive frequency so that the phase difference decreases. In addition, the phase detection circuit performs post-adjustment of the phase value detected for both or one of the voltage phase and the current phase according to the signal level after DC conversion of the output signal.

  It should be noted that any combination of the above-described constituent elements and a representation of the present invention converted between a method, an apparatus, a system, and the like are also effective as an aspect of the present invention.

  ADVANTAGE OF THE INVENTION According to this invention, in magnetic field resonance type wireless electric power feeding, it becomes easy to control electric power feeding efficiently.

1 is a system configuration diagram of a wireless power transmission system. FIG. It is a graph which shows the relationship between load current and load voltage. It is a graph which shows the relationship between the distance between coils, and load voltage. It is a graph which shows the relationship between the impedance of a feeding coil circuit, and a drive frequency. It is a graph which shows the relationship between output power efficiency and a drive frequency. It is a circuit diagram of a control signal generation circuit. It is a time chart which shows the relationship of T0-T2 signal. It is a circuit diagram of a signal transmission circuit. It is a time chart which shows the relationship between T2, T3, and T6 signals. It is a circuit diagram of a current phase detection circuit and a signal reception circuit. It is a time chart which shows the relationship between S1, S3, and T5 signals. It is a time chart which shows the relationship between S1 signal and S2 signal. It is a circuit diagram of the wireless power receiving apparatus in a modification. It is a modification of the system block diagram of a wireless power transmission system.

  Hereinafter, preferred embodiments of the present invention will be described with reference to the accompanying drawings.

  FIG. 1 is a system configuration diagram of a wireless power transmission system 100. The wireless power transmission system 100 includes a wireless power feeder 116 and a wireless power receiver 118. The wireless power feeder 116 includes a power transmission control circuit 200, an exciting circuit 110, a power feeding coil circuit 120, a phase detection circuit 114, and a signal reception circuit 112 as a basic configuration. The wireless power receiving apparatus 118 includes a power receiving coil circuit 130, a load circuit 140, a control signal generation circuit 170, a reference signal generation circuit 172, and a signal transmission circuit 122.

  There is a distance of about 0.2 to 1.0 m (hereinafter referred to as “inter-coil distance”) between the feeding coil L2 included in the feeding coil circuit 120 and the receiving coil L3 included in the receiving coil circuit 130. The main purpose of the wireless power transmission system 100 is to send AC power wirelessly from the feeding coil L2 to the receiving coil L3. In the present embodiment, description will be made assuming that the resonance frequency fr = 100 kHz. Note that the wireless power transmission system according to the present embodiment can be operated in a high frequency band such as an ISM (Industry-Science-Medical) frequency band. The low frequency band has an advantage that the cost and switching loss of a switching transistor (described later) can be easily suppressed, and regulations of the Radio Law are loose.

  The exciting circuit 110 is a circuit in which an exciting coil L1 and a transformer T2 secondary coil Li are connected in series. The transformer T2 secondary coil Li forms a coupling transformer T2 together with the transformer T2 primary coil Lb, and is supplied with AC power from the power transmission control circuit 200 by electromagnetic induction. The number of turns of the exciting coil L1 is one, the conductive wire diameter is 5 mm, and the shape of the exciting coil L1 itself is a square of 210 mm × 210 mm. In FIG. 1, the excite coil L <b> 1 is drawn in a circle for easy understanding. The same applies to the other coils. Each coil shown in FIG. 1 is made of copper. An alternating current I1 flows through the exciting circuit 110.

  The feeding coil circuit 120 is a circuit in which a feeding coil L2 and a capacitor C2 are connected in series. The exciting coil L1 and the feeding coil L2 face each other. The distance between the exciting coil L1 and the feeding coil L2 is relatively close to 10 mm or less. For this reason, the exciting coil L1 and the feeding coil L2 are strongly coupled electromagnetically. The number of turns of the feeding coil L2 is seven, the conductor diameter is 5 mm, and the shape of the feeding coil L2 itself is a square of 280 mm × 280 mm. When an alternating current I1 is passed through the exciting coil L1, an electromotive force is generated in the feeding coil L2 due to the principle of electromagnetic induction, and an alternating current I2 flows in the feeding coil circuit 120. The alternating current I2 is much larger than the alternating current I1. The values of the feeding coil L2 and the capacitor C2 are set so that the resonance frequency fr of the feeding coil circuit 120 is 100 kHz.

  The power receiving coil circuit 130 is a circuit in which a power receiving coil L3 and a capacitor C3 are connected in series. The power feeding coil L2 and the power receiving coil L3 face each other. The number of turns of the power receiving coil L3 is seven, the conductor diameter is 5 mm, and the shape of the power receiving coil L3 itself is a square of 280 mm × 280 mm. The values of the receiving coil L3 and the capacitor C3 are set so that the resonance frequency fr of the receiving coil circuit 130 is also 100 kHz. The feeding coil L2 and the receiving coil L3 do not have to have the same shape. When the feeding coil L2 generates a magnetic field at the resonance frequency fr = 100 kHz, the feeding coil L2 and the receiving coil L3 are magnetically resonated, and a large current I3 also flows through the receiving coil circuit 130.

  The load circuit 140 is a circuit in which the load coil L4 is connected to the load LD via the rectifier circuit 124 and the measurement circuit 126. The power receiving coil L3 and the load coil L4 face each other. The distance between the power receiving coil L3 and the load coil L4 is relatively close to 10 mm or less. For this reason, the receiving coil L3 and the load coil L4 are strongly coupled electromagnetically. The winding number of the load coil L4 is 1, the conductor diameter is 5 mm, and the shape of the load coil L4 itself is 300 mm × 300 mm. When the current I3 flows through the power receiving coil L3, an electromotive force is generated in the load circuit 140, and an alternating current I4 flows through the load circuit 140. The alternating current I4 is rectified into a direct current by the rectifying circuit 124. A part flows through the measurement circuit 126, but most flows through the load LD as a direct current I5. The rectifier circuit 124 is a general circuit including a bridge circuit 128 and a capacitor C5. The measurement circuit 126 will be described later.

  The AC power transmitted from the power feeding coil L2 of the wireless power feeding device 116 is received by the power receiving coil L3 of the wireless power receiving device 118, and is taken out as DC power from the load LD. The voltage applied to the load LD is referred to as “load voltage V5”.

  When the load LD is directly connected to the power receiving coil circuit 130, the Q value of the power receiving coil circuit 130 is deteriorated. For this reason, the receiving coil circuit 130 for receiving power and the load circuit 140 for extracting power are separated. In order to increase the power transmission efficiency, it is preferable to align the center lines of the feeding coil L2, the receiving coil L3, and the load coil L4.

  The measurement circuit 126 includes resistors R1 and R2, a control power supply Vs, and a comparator 132. The load voltage V5 is divided by resistors R1 and R2. The voltage applied across the resistor R2 is referred to as “output voltage”. The potential at the connection point F between the resistors R1 and R2 is input to the negative terminal of the comparator 132 as a “measurement potential”. A control power supply Vs is connected to the positive terminal of the comparator 132. The input potential of the positive terminal by the control power supply Vs is called “reference potential”.

  The comparator 132 amplifies the difference between the measurement potential and the reference potential (hereinafter referred to as “correction voltage”), and outputs the amplified value as the T0 signal. The T0 signal is a DC voltage signal and indicates the magnitude of the correction voltage. In other words, the T0 signal is a signal indicating the amount of change in the load voltage V5. As will be described in detail later, the wireless power transmission system 100 stabilizes the output voltage (load voltage V5) by controlling the feed power so that the correction voltage becomes zero. In this embodiment, the reference potential is set to 2.5 (V). Further, when the load voltage V5 is 24 (V), the resistors R1 and R2 are set so that the measured potential is 2.5 (V) and the correction voltage is 0 (V). The control power source Vs is a variable DC voltage source and can be arbitrarily adjusted in voltage.

  The control signal generation circuit 170 is a circuit that generates an AC voltage signal having a control frequency fc as a T1 signal. The control frequency fc in this embodiment is 1.0 kHz. The control signal generation circuit 170 will be described later with reference to FIG. The comparator 174 compares the T0 signal and the T1 signal, and generates a T2 signal (valid signal: AC voltage signal) that becomes a high level when T1> T0. As will be described in detail later, the duty ratio of the T2 signal changes depending on the correction voltage. The relationship between the T0 and T2 signals will be described later with reference to FIG.

  The reference signal generation circuit 172 generates an AC voltage signal having a reference frequency fs as a T3 signal. The reference frequency fs in this embodiment is 38 kHz. The signal transmission circuit 122 generates a T4 signal of an AC optical signal based on the T2 signal and the T3 signal. The T4 signal is an “output signal” indicating the magnitude of output on the power receiving side, and is received by the signal receiving circuit 112 of the wireless power feeder 116. From the T4 signal, the power feeding side can recognize the magnitude of the correction voltage, in other words, the amount of change in the load voltage V5. The circuit configurations and processing contents of the signal transmission circuit 122 and the reference signal generation circuit 172 will be described later with reference to FIGS.

  Next, the configuration of the power transmission control circuit 200 will be described. First, a VCO (Voltage Controlled Oscillator) 202 is connected to the primary side of the gate driving transformer T1. The VCO 202 functions as an “oscillator” that generates an AC voltage Vo having a drive frequency fo. The waveform of the AC voltage Vo may be a sine wave, but here it will be described as a rectangular wave (digital waveform). Due to the AC voltage Vo, current flows alternately in the positive and negative directions in the transformer T1 primary coil Lh. The transformer T1 primary coil Lh, the transformer T1 secondary coil Lf, and the transformer T1 secondary coil Lg form a coupling transformer T1 for driving a gate. Due to the electromagnetic induction, current flows alternately in both the positive and negative directions in the transformer T1 secondary coil Lf and the transformer T1 secondary coil Lg.

  The VCO 202 in this embodiment uses a built-in unit of Motorola: product number MC14046B. The VCO 202 also has a function of dynamically changing the drive frequency fo based on a phase difference indicating voltage SC (described later) output from the phase comparison circuit 150.

  In the following description, it is assumed that the minimum value of the drive frequency fo is fo1 = 90 kHz and the maximum value is fo2 = 99 kHz. The appropriate range of the phase difference indicating voltage SC is 1.0 to 4.0 (V). The phase difference indicating voltage SC and the drive frequency fo are directly proportional. That is, when the phase difference indicating voltage SC = 1.0 (V), the drive frequency fo = fo1 = 90 kHz, and when SC = 4.0 (V), fo = fo2 = 99 kHz.

  The power sources for the power transmission control circuit 200 are capacitors CA and CB that are charged by the DC power source Vdd. The capacitor CA is provided between the points C and E shown in FIG. 1 and the capacitor CB is provided between the points E and C. When the voltage of the capacitor CA (voltage between CE) is VA and the voltage of the capacitor CB (voltage between ED) is VB, VA + VB (voltage between CD) is the input voltage. Capacitors CA and CB function as a DC voltage source.

  One end of the transformer T1 secondary coil Lf is connected to the gate of the switching transistor Q1, and the other end is connected to the source of the switching transistor Q1. One end of the transformer T1 secondary coil Lg is connected to the gate of another switching transistor Q2, and the other end is connected to the source of the switching transistor Q2. When the VCO 202 generates the AC voltage Vo at the drive frequency fo, the voltage Vx (Vx> 0) is alternately applied to the gates of the switching transistor Q1 and the switching transistor Q2 at the drive frequency fo. For this reason, the switching transistor Q1 and the switching transistor Q2 are alternately turned on and off at the drive frequency fo. The switching transistor Q1 and the switching transistor Q2 are enhancement-type MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) having the same characteristics, but may be other transistors such as bipolar transistors. Other switches such as a relay switch may be used instead of the transistor.

  The drain of the switching transistor Q1 is connected to the positive electrode of the capacitor CA. The negative electrode of the capacitor CA is connected to the source of the switching transistor Q1 via the transformer T2 primary coil Lb. The source of the switching transistor Q2 is connected to the negative electrode of the capacitor CB. The positive electrode of the capacitor CB is connected to the drain of the switching transistor Q2 via the transformer T2 primary coil Lb.

  The voltage between the source and drain of the switching transistor Q1 is referred to as source-drain voltage VDS1, and the voltage between the source and drain of the switching transistor Q2 is referred to as source-drain voltage VDS2. Further, a current flowing between the source and the drain of the switching transistor Q1 is a source / drain current IDS1, and a current flowing between the source and the drain of the switching transistor Q2 is a source / drain current IDS2. For the source / drain currents IDS1 and IDS2, the direction indicated by the arrow in the figure is the positive direction, and the opposite direction is the negative direction.

  When the switching transistor Q1 becomes conductive (ON), the switching transistor Q2 becomes non-conductive (OFF). The main current path at this time (hereinafter referred to as “first current path”) is a path that returns from the positive electrode of the capacitor CA to the negative electrode via the point C, the switching transistor Q1, the transformer T2, the primary coil Lb, and the point E. It becomes. The switching transistor Q1 functions as a switch that controls conduction / non-conduction of the first current path.

  When the switching transistor Q2 is conductive (ON), the switching transistor Q1 is non-conductive (OFF). The main current path at this time (hereinafter referred to as “second current path”) is a path that returns from the positive electrode of the capacitor CB to the negative electrode via the point E, the transformer T2 primary coil Lb, the switching transistor Q2, and the point D. It becomes. The switching transistor Q2 functions as a switch that controls conduction / non-conduction of the second current path.

  In the power transmission control circuit 200, the current flowing through the transformer T2 primary coil Lb is referred to as “current Is”. The current Is is an alternating current, and when flowing through the first current path, it is called a positive direction, and when flowing through the second current path, it is called a negative direction.

  When the VCO 202 supplies the AC voltage Vo at the drive frequency fo, the first current path and the second current path are alternately switched at the drive frequency fo. Since the alternating current Is of the driving frequency fo flows through the transformer T2 primary coil Lb, the alternating current I1 flows through the exciting circuit 110 at the driving frequency fo, and further, the alternating current I2 of the driving frequency fo also flows through the feeding coil circuit 120. . The closer the drive frequency fo is to the resonance frequency fr, the higher the power transmission efficiency. If driving frequency fo = resonance frequency fr, the feeding coil L2 and the capacitor C2 of the feeding coil circuit 120 are in a resonance state. Since the receiving coil circuit 130 is also a resonance circuit having a resonance frequency fr, the feeding coil L2 and the receiving coil L3 magnetically resonate. At this time, the power transmission efficiency is maximized.

  However, in the case of the present embodiment, the resonance frequency fr is not included in the operating range of the drive frequency fo, so that the power transmission efficiency does not become maximum. This is because the stability of the load voltage V5 is given priority over maximizing the power transmission efficiency. Since the change in the load voltage V5 can be detected by the correction voltage, the wireless power feeder 116 automatically adjusts the drive frequency fo so that the correction voltage becomes zero. Details will be described later.

  The resonance frequency fr slightly changes depending on the use state and use environment of the feeding coil circuit 120 and the receiving coil circuit 130. The resonance frequency fr also changes when the power feeding coil circuit 120 and the power receiving coil circuit 130 are replaced. Alternatively, there may be a case where it is desired to positively change the resonance frequency fr by making the capacitances of the capacitors C2 and C3 variable. Further, it has been found from the experiments of the present inventor that the resonance frequency fr begins to decrease when the distance between the feeding coil L2 and the receiving coil L3 is reduced to some extent. When the difference between the resonance frequency fr and the drive frequency fo changes, the power transmission efficiency changes. When the power transmission efficiency changes, the load voltage V5 changes. Therefore, in order to stabilize the load voltage V5, it is necessary to keep the difference between the resonance frequency fr and the drive frequency fr constant even when the resonance frequency fo changes.

  The feeding coil circuit 120 is provided with a detection coil LSS. The detection coil LSS is a coil wound Ns times around a core 154 (toroidal core) having a through hole. The material of the core 154 is a known material such as ferrite, a silicon steel plate, or permalloy. The number of turns Ns of the detection coil LSS in the present embodiment is 100 times.

  A part of the current path of the feeding coil circuit 120 also passes through the through hole of the core 154. This means that the number of turns Np of the feeding coil circuit 120 with respect to the core 154 is one. With such a configuration, the detection coil LSS and the feeding coil L2 form a coupling transformer. An in-phase induction current ISS flows through the detection coil LSS due to an AC magnetic field generated by the AC current I2 of the power supply coil L2. According to the equal ampere-turn law, the magnitude of the induced current ISS is I2 · (Np / Ns).

  A resistor R4 is connected to both ends of the detection coil LSS. One end B of the resistor R4 is grounded, and the potential VSS at the other end A is connected to the current phase detection circuit 144 via the comparator 142.

  The potential VSS is binarized by the comparator 142 and becomes the S0 signal. The comparator 142 outputs a saturation voltage of 3.0 (V) when the potential VSS becomes higher than a predetermined threshold, for example, 0.1 (V). The potential VSS is converted into an S0 signal having a digital waveform by the comparator 142. The current I2 and the induced current ISS are in phase, and the induced current ISS and the potential VSS (S0 signal) are in phase. Moreover, the alternating current Is flowing through the power transmission control circuit 200 is in phase with the current I2. Therefore, the current phase of the alternating current Is can be measured by observing the waveform of the S0 signal.

  When the resonance frequency fr and the drive frequency fo coincide, the current phase and the voltage phase also coincide. The deviation between the resonance frequency fr and the drive frequency fo can be measured from the phase difference between the current phase and the voltage phase. The wireless power transmission system 100 according to the present embodiment automatically follows the drive frequency fo with respect to the change in the resonance frequency fr by measuring the difference between the resonance frequency fr and the drive frequency fo based on this phase difference. .

  Phase detection circuit 114 includes a current phase detection circuit 144, a phase comparison circuit 150, and a low pass filter 152. The low-pass filter 152 is a known circuit and is inserted to cut a high frequency component of the phase difference indicating voltage SC. The phase comparison circuit 150 in the present embodiment uses a built-in unit (Phase Comparator) of Motorola: product number MC14046B, similar to the VCO 202. Therefore, the phase comparison circuit 150 and the VCO 202 can be realized with one chip.

  The current phase detection circuit 144 generates an S1 signal as a signal indicating the current phase. The S1 signal is input to the phase comparison circuit 150. The AC voltage Vo generated by the VCO 202 is input to the phase comparison circuit 150 as an S2 signal indicating the voltage phase. The phase comparison circuit 150 detects a deviation (phase difference) between the current phase and the voltage phase from the S1 and S2 signals, and generates a phase difference indicating voltage SC indicating the magnitude of the phase difference. By detecting the phase difference, the magnitude of the deviation between the resonance frequency fr and the drive frequency fo is detected. By controlling the drive frequency fo according to the phase difference indicating voltage SC, the phase difference between the drive frequency fo and the resonance frequency fr can be kept constant.

  For example, since the phase difference increases when the drive frequency fo and the resonance frequency fr deviate, the phase comparison circuit 150 may generate the phase difference indicating voltage SC so as to reduce this phase difference. Therefore, even if the resonance frequency fr changes, the power transmission efficiency can be kept constant and the load voltage V5 can be stabilized. The circuit configurations of the current phase detection circuit 144 and the signal reception circuit 112 will be described in detail later with reference to FIG. 10, and the relationship between the S1 signal and the S2 signal will be described in detail with reference to FIG.

  A resistor may be connected in parallel to both ends of the transformer T1 primary coil Lh, and the AC voltage Vo may be divided and used as the S2 signal. Even when the AC voltage Vo generated by the VCO 202 is large, the voltage can be reduced to an easy-to-handle voltage. The voltage phase may be measured from the source / drain voltages VDS1, VDS2, the source / gate voltages VGS1, VGS2, and the like.

  Further, even if the resonance frequency fr is constant, the load voltage V5 may change. For example, the load voltage V5 changes when the load LD is a variable resistor or when the load LD itself is replaced. In the present embodiment, the load voltage V5 is stabilized by detecting a change in the load voltage V5 as a correction voltage and automatically adjusting the drive frequency fo so that the correction voltage becomes zero.

  The correction voltage is transmitted from the signal transmission circuit 122 to the signal reception circuit 112 as a T4 signal (AC optical signal). The signal receiving circuit 112 converts the T4 signal that is an AC optical signal into a T5 signal that is a DC voltage signal. The voltage level of the T5 signal is positively correlated with the load voltage V5.

  The current phase detection circuit 144 adjusts the S0 signal (AC voltage signal) indicating the current phase with the T5 signal (DC voltage signal) indicating the correction voltage, and outputs the S1 signal (AC voltage signal) as the correction current phase. When the load voltage V5 is a desired value of 24 (V), the S0 signal becomes the S1 signal as it is. The phase comparison circuit 150 detects the phase difference between the voltage phase and the current phase of the AC power based on the S1 signal and the S2 signal, and outputs the phase difference indicating voltage SC. The VCO 202 adjusts the drive frequency fo based on the phase difference indicating voltage SC. More specifically, the VCO 202 changes the drive frequency fo by changing the pulse width of the AC voltage Vo.

  Even when the correction voltage is not zero, that is, when adjustment is performed by the T5 signal, the phase comparison circuit 150 detects the phase difference between the voltage phase and the current phase of the AC power based on the S1 signal and the S2 signal, and The phase difference indicating voltage SC is output. However, since the S1 signal at this time is a signal obtained by adjusting the S0 signal according to the T5 signal, it is not a signal indicating the actual current phase. The adjustment logic based on the correction voltage will be described in more detail with reference to FIG.

  FIG. 2 is a graph showing the relationship between the load current I5 and the load voltage V5. The horizontal axis indicates the magnitude of the load current I5 (direct current) flowing through the load LD, and the vertical axis indicates the load voltage V5. The non-adjustment characteristic 134 indicates a current / voltage characteristic when the adjustment based on the correction voltage is not performed. In the case of the non-adjustment characteristic 134, when the load LD increases, the load current I5 decreases and the load voltage V5 increases. Conversely, when the load LD becomes smaller, the load current I5 increases and the load voltage V5 decreases. In this way, even if the supplied power is constant, when the load LD is changed, the load voltage V5 also changes.

  The wireless power transmission system 100 according to the present embodiment realizes the current / voltage characteristics indicated by the characteristics 136 at the time of adjustment. Specifically, the power transmission efficiency is changed and the load voltage V5 is stabilized by adjusting the S1 signal based on the correction voltage.

  FIG. 3 is a graph showing the relationship between the inter-coil distance d and the load voltage V5. The horizontal axis represents the inter-coil distance d between the power feeding coil L2 and the power receiving coil L3, and the vertical axis represents the load voltage V5. The non-adjustment characteristic 146 indicates a voltage / distance characteristic when adjustment based on the correction voltage is not performed. As described above, the resonance frequency fr varies depending on the inter-coil distance d. When the resonance frequency fr changes and the difference between the drive frequency fo and the resonance frequency fr changes, the power transmission efficiency changes. Even if the drive frequency fo is made to follow the resonance frequency fr, the load voltage V5 slightly changes depending on the inter-coil distance d.

  The wireless power transmission system 100 according to the present embodiment realizes the voltage / distance characteristic indicated by the adjustment characteristic 148. By adjusting the S1 signal based on the correction voltage, the power transmission efficiency is changed and the load voltage V5 is stabilized.

FIG. 4 is a graph showing the relationship between the impedance Z of the feeding coil circuit 120 and the drive frequency fo. The vertical axis represents the impedance Z of the feeding coil circuit 120 (a series circuit of the capacitor C2 and the feeding coil L2). The horizontal axis indicates the drive frequency fo. The impedance Z becomes the minimum value Z _min during resonance. Ideally, Z _min = 0 at the time of resonance, but since the feed coil circuit 120 includes some resistance component, Z _min is not normally zero.

  When the drive frequency fo = the resonance frequency fr, the impedance Z is the lowest, and the capacitor C2 and the feeding coil L2 are in a resonance state. When the drive frequency fo and the resonance frequency fr are deviated, the capacitive reactance or inductive reactance at the impedance Z becomes dominant, and the impedance Z also increases.

  As the drive frequency fo and the resonance frequency fr deviate, the impedance Z increases and the power transmission efficiency decreases. Therefore, the power transmission efficiency can be changed by changing the difference between the drive frequency fo and the resonance frequency fr.

  FIG. 5 is a graph showing the relationship between output power efficiency and drive frequency fo. The output power efficiency indicates the ratio of the power actually supplied from the feeding coil L2 to the maximum output value. When the drive frequency fo coincides with the resonance frequency fr, the difference between the current phase and the voltage phase becomes zero, and the power transmission efficiency is maximized, so that the output power efficiency = 100 (%). In the wireless power transmission system 100 of the present embodiment, the drive frequency fo is adjusted in a range of fo1 to fo2 that is lower than the resonance frequency fr.

  FIG. 6 is a circuit diagram of the control signal generation circuit 170. The T1 signal (control signal) that is the output of the control signal generation circuit 170 is input to the positive terminal of the comparator 174. The T0 signal output from the measurement circuit 126 is input to the negative terminal of the comparator 174. The T0 signal is a DC voltage signal indicating a correction voltage.

  The control signal generation circuit 170 generates an alternating voltage that changes in a sawtooth waveform at a control frequency = 1.0 kHz as a T1 signal. Control signal generation circuit 170 includes resistors R5 to R7, capacitor C6, and thyristor 138. A gate voltage VG obtained by dividing the power supply voltage of the power supply VCC by resistors R5 and R6 is applied to the gate G of the thyristor 138. The gate voltage VG is a constant value. The anode A of the thyristor 138 is connected to the power supply VCC through the resistor R7 and is connected to the ground through the capacitor C6. The anode potential VA is applied to the thyristor 138 by dropping the power supply voltage with the resistor R7. The T1 signal indicates this anode potential VA.

  When the anode potential VA ≦ the gate potential VG, the anode and cathode of the thyristor 138 are turned off (non-conducting), and the capacitor C6 is charged during this period. When the capacitor C6 is charged, the anode potential VA (T1 signal) increases, and when the anode potential VA> the gate potential VG, the anode and cathode of the thyristor 138 are turned on (conductive). At this time, since the electric charge of the capacitor C6 is discharged through the thyristor 138, the anode potential VA ≦ the gate potential VG is satisfied again. The control signal generation circuit 170 repeats the above process at 1.0 kHz (control frequency fc). As a result, a sawtooth T1 signal is generated as will be described later with reference to FIG. The control frequency fc is determined by the time constant of the capacitor C6 and the resistor R7.

  The comparator 174 outputs a high-level T2 signal (valid signal) when the T1 signal> T0 signal, and outputs a low-level T2 signal otherwise. The period of T1 signal> T0 signal is a valid period, and the other period is an invalid period. The higher the correction potential, the higher the level of the T0 signal and the shorter the effective period.

  FIG. 7 is a time chart showing the relationship between the T0 and T2 signals. In control signal generation circuit 170, charging of capacitor C6 is started at time t0. Since the anode potential VA gradually increases, the T1 signal also gradually increases. At time t1, the anode potential VA> the gate potential VG, and the anode and cathode of the thyristor 138 are turned on (conductive). Since the capacitor C6 is discharged, the anode potential VA (S1 signal) drops rapidly. The period from time t0 to time t1 is referred to as “unit period”. The same applies after time t1. Since the control frequency fc = 1.0 kHz, the length of the unit period is 1.0 (msec).

  The T0 signal is a DC voltage signal whose voltage level changes in accordance with the correction voltage. The comparator 174 compares the T0 signal and the T1 signal, and generates a high level T2 signal when T1> T0 signal and a low level signal when T1 ≦ T0 signal. In the unit period from t0 to t1, the T2 signal is low level from t0 to t4, and the T2 signal is high level from t4 to t1. That is, in the unit period (t0 to t1), the time t0 to the time t4 is the invalid period, and the time t4 to the time t1 is the valid period. When the level of the T0 signal changes according to the correction voltage, the duty ratio between the valid period and the invalid period changes. When the load voltage V5 increases, the correction potential decreases and the duty ratio of the T2 signal increases. On the contrary, when the load voltage V5 decreases, the correction potential increases and the duty ratio of the T2 signal decreases. In the present embodiment, the duty ratio is set to be less than 100% even when the correction potential becomes zero.

  FIG. 8 is a circuit diagram of the signal transmission circuit 122. The signal transmission circuit 122 includes an infrared LED (Light Emitting Diode) 158, a transistor Q3, and a signal control circuit 156. The transistor Q3 is a bipolar transistor whose emitter is grounded, and the base and the emitter are connected via a resistor R9. One end of the infrared LED 158 is connected to the power supply VCC via the resistor R8, and the other end is connected to the base of the transistor Q3. A signal control circuit 156 is also connected to the base of the transistor Q3.

  The signal control circuit 156 is a known circuit, and for example, an IC (Integrated Circuit) of product number UCC37321 manufactured by Texas Instruments may be used. A reference signal generation circuit 172 and a comparator 174 are connected to the signal control circuit 156. The signal control circuit 156 receives the T2 signal and the T3 signal and outputs the T6 signal. The reference signal generation circuit 172 is an oscillator that generates a T3 signal (reference signal) at a predetermined reference frequency fs. In the present embodiment, it is assumed that the reference frequency fs is 38 kHz that is sufficiently larger than the control frequency fc. The T3 signal may be a sine wave, but here it will be described as a rectangular wave (digital waveform).

  The signal control circuit 156 outputs the T3 signal (reference signal) as the T6 signal only during a period in which the T2 signal is at a high level, that is, an effective period. During the invalid period, the T6 signal is fixed at a low level.

  The T6 signal (AC voltage signal) is changed to a T4 signal (AC light signal) by the infrared LED 158. The infrared LED 158 sends a T4 signal (AC light signal) to the signal receiving circuit 112. The T4 signal in the present embodiment is an infrared signal. The infrared wavelength is about 940 nm. Since the T4 signal reaches a distance of about several meters, it can sufficiently cope with a large distance between the coils. Infrared rays are hardly affected by the magnetic field generated by the power feeding coil L2 or the power receiving coil L3, so that there is an advantage that the T4 signal and the power feeding power hardly affect each other.

  FIG. 9 is a time chart showing the relationship between the T2, T3, and T6 signals. As described with reference to FIG. 7, the T2 signal (effective signal) is an AC voltage signal having a control frequency fc = 1.0 kHz with t0 to t1, t1 to t2,. A period in which the T2 signal is at a high level is an effective period, and a period in which the T2 signal is at a low level is an invalid period. The T3 signal is an AC voltage signal having a reference frequency fs = 38 kHz. The signal control circuit 156 outputs the T3 signal as the T6 signal only during the valid period. That is, the logical product of the T2 signal and the T3 signal is the T6 signal.

  The T6 signal that is an AC voltage signal is converted into a T4 signal that is an AC optical signal, and is irradiated toward the signal receiving circuit 112. The infrared LED 158 blinks at the reference frequency fs = 38 kHz during the effective period and is turned off during the invalid period. The blinking period and the extinguishing period are repeated at the control frequency fc. The duty ratio between the blinking period and the extinguishing period varies depending on the correction voltage. The higher the correction voltage, the shorter the blinking period. The ratio of the blinking period in the unit period is called “duty ratio of T4 signal (output signal)”.

  Note that the infrared LED 158 may be turned on by the T2 signal instead of the T6 signal. In this case, the infrared LED 158 continues to be lit during the effective period. If the infrared LED 158 blinks in accordance with the T3 signal during the effective period as in the present embodiment, it is effective in suppressing the power consumption of the infrared LED 158.

  FIG. 10 is a circuit diagram of the current phase detection circuit 144 and the signal reception circuit 112. The current phase detection circuit 144 includes a comparator 166 and a current waveform shaping circuit 168. First, the potential VSS is shaped into a digital waveform S0 signal by the comparator 142 and input to the current waveform shaping circuit 168. The current waveform shaping circuit 168 shapes the digital waveform (rectangular wave) S0 signal into a sawtooth S3 signal. In the current waveform shaping circuit 168, a resistor R10 is inserted in the path of the S0 signal, and a diode D1 is connected in parallel to the resistor R10. The transmission path of the S0 signal is grounded via the capacitor C7. The S3 signal (AC voltage signal) is input to the positive terminal of the comparator 166. The S3 signal is a signal indicating a current phase.

  The signal receiving circuit 112 includes a photodiode 160, a voltage conversion unit 164, and a low pass filter 176. The voltage conversion unit 164 includes a comparator 162 and a resistor R12.

  The photodiode 160 receives a T4 signal that is an optical signal that blinks intermittently. The T4 signal (AC light signal) is converted into a T7 signal (AC voltage signal) by the voltage converter 164. In the voltage converter 164, the resistor R12 is adjusted so that 1 (mV) is output per 1 (lux). Since the brightness of the T4 signal during light reception is about 0 to 2000 (lux), the voltage level of the T7 signal is about 0 to 2.0 (V). The duty ratio of the T7 signal indicates a correction voltage. The T7 signal (AC voltage signal) is converted by the low-pass filter 176 into a constant value T5 signal (DC voltage signal). The low pass filter 176 is a general circuit including a resistor R11 and a capacitor C8. The higher the correction voltage, the lower the voltage level of the T5 signal. The T5 signal (DC voltage signal) is input to the negative terminal of the comparator 166. The T5 signal is a DC voltage signal indicating a correction voltage.

  When S3> T5, the comparator 166 outputs a high level S1 signal, and otherwise outputs a low level S1 signal.

  FIG. 11 is a time chart showing the relationship between the S1, S3, and T5 signals. The S3 signal is an AC voltage signal having a drive frequency fo. The S3 signal indicates the current phase. The S3 signal rises from time t10 and drops sharply at time t11. The period from time t10 to t11 is a unit period of the S3 signal. Since the drive frequency fo is 90 to 99 kHz, the length of the unit period is about 0.01 (msec).

  The T5 signal is a DC voltage signal whose voltage level changes according to the correction voltage. The comparator 166 compares the S3 signal and the T5 signal, and generates a high-level S1 signal when S3> T5 signal and a low level when S3 ≦ T5 signal. Of the unit periods from t10 to t11, the S1 signal is low level from t10 to t14, and the S1 signal is high level from t14 to t11. As the level of the T5 signal changes according to the correction voltage, the duty ratio of the S1 signal changes. As will be described in detail later, as the load voltage V5 increases, the correction potential decreases and the signal level of the T5 signal increases. As a result, the duty ratio of the S1 signal is reduced and the rising time of the S1 signal is later than the rising time of the S3 signal.

  FIG. 12 is a time chart showing the relationship between the S1 signal and the S2 signal. A period from time t20 to time t21 (hereinafter referred to as “first period”) is a period in which the switching transistor Q1 is on and the switching transistor Q2 is off. A period from time t21 to time t22 (hereinafter referred to as “second period”) is a period in which the switching transistor Q1 is turned off and the switching transistor Q2 is turned on, and a period from time t22 to time t23 (hereinafter referred to as “third period”). In the period when the switching transistor Q1 is on and the switching transistor Q2 is off, the period from time t23 to time t24 (hereinafter referred to as “fourth period”), the switching transistor Q1 is off and the switching transistor Q2 is off. It is assumed that the period is on.

  At time t20, the AC voltage Vo (S2 signal) changes from the minimum value to the maximum value. At time t21 when the first period ends, the AC voltage Vo (S2 signal) changes from the maximum value to the minimum value. Hereinafter, the timing at which the S2 signal rises at time t20 is referred to as “voltage phase value”.

  When the drive frequency fo is smaller than the resonance frequency fr, a capacitive reactance component appears in the impedance Z of the feeding coil circuit 120 (LC resonance circuit), and the current phase of the current Is advances with respect to the voltage phase. The S0 signal indicating the current phase rises at time t10 earlier than time t20. Hereinafter, the timing at which the S0 signal rises at time t10 is referred to as “current phase value”. In the case of FIG. 12, t20-t10 shows a phase difference. Since t20-t10> 0, the current phase is advanced with respect to the voltage phase.

  When the S0 signal rises at time t10, the level of the S3 signal also starts to rise. At time t11 when the S0 signal becomes low level, the S3 signal also drops rapidly.

  The T5 signal is a DC voltage signal whose level changes depending on the magnitude of the correction voltage. FIG. 12 shows a state in which the correction voltage is detected and the load voltage V5 is deviated from the desired value.

  The S3 signal and the T5 signal are input to the positive terminal and the negative terminal of the comparator 166, respectively, and the output is the S1 signal. When S3> T5, S1 is at a high level, and when S3 ≦ T5, S1 is at a low level. In FIG. 12, S3> T5 at time t14 after time t10 (hereinafter, such timing is also referred to as “corrected current phase value”). The voltage level of the T5 signal becomes a “reference value” that determines the corrected current phase value.

  The phase comparison circuit 150 compares the rising time t10 of the S2 signal with the rising time t14 of the S1 signal and detects it as the phase difference td. The actual phase difference is t20−t10 (> 0), but the phase difference recognized by the phase comparison circuit 150 is t20−t14 (<0). The phase comparison circuit 150 outputs a phase difference indicating voltage SC corresponding to t20-t14. The VCO 202 determines that the current phase is behind the voltage phase, that is, the drive frequency fo is higher than the resonance frequency fr, and tries to eliminate the phase difference by reducing the drive frequency fo. As a result, the power transmission efficiency is lowered, the load voltage V5 is suppressed, and feedback control is performed so that the correction voltage is eliminated.

  For example, when the resistance value of the load LD increases, the load current I5 decreases and the load voltage V5 increases (see FIG. 2). When the load voltage V5 increases, the measurement potential increases and the correction voltage decreases. As a result, the voltage level of the T0 signal (DC voltage signal) decreases.

  When the voltage level of the T0 signal decreases, the duty ratio of the T2 signal increases (see FIG. 7). As a result, the duty ratio of the T4 signal (output signal) also increases (see FIG. 9). As the duty ratio of the T4 signal increases, the voltage level of the T5 signal (DC voltage signal) increases. As a result, the duty ratio of the S1 signal becomes small. Further, since the rising time of the S1 signal is later than the rising time of the S2 signal, the phase comparison circuit 150 recognizes that the current phase is delayed with respect to the voltage phase. In order to advance the current phase, the phase comparison circuit 150 instructs the VCO 202 to decrease the drive frequency fo by the phase difference indicating voltage SC. Since the difference between the resonance frequency fr and the drive frequency fo is further increased and the power transmission efficiency is reduced (see FIGS. 4 and 5), the load voltage V5 is reduced. By such feedback control, the load voltage V5 can be maintained at a constant value. Similar feedback control is also performed when the load voltage V5 decreases.

  FIG. 13 is a circuit diagram of the wireless power receiving apparatus 118 in the modification. In FIG. 1, the direct current I5 is supplied to the load LD. However, as a modification, the alternating current I4 may be supplied to the load LD as it is. In this case, the rectifier circuit 124 and the measurement circuit 126 may be connected to a part of the load coil L4 to output the T0 signal.

  FIG. 14 is a modification of the system configuration diagram of the wireless power transmission system 100. In the modification, the power transmission control circuit 200 directly drives the feeding coil circuit 120 without passing through the exciting circuit 110. The configurations denoted by the same reference numerals as those in FIG. 1 have the same or similar functions as the configurations described in FIG.

  The feed coil circuit 120 in the modification is a circuit in which a transformer T2 secondary coil Li is connected in series to a feed coil L2 and a capacitor C2. The transformer T2 secondary coil Li forms a coupling transformer T2 together with the transformer T2 primary coil Lb, and is supplied with AC power from the power transmission control circuit 200 by electromagnetic induction. In this way, AC power may be directly supplied from the power transmission control circuit 200 to the feeding coil circuit 120 without using the exciting circuit 110.

The wireless power transmission system 100 has been described above based on the embodiment. In the case of magnetic field resonance type wireless power feeding, the power transmission efficiency can be controlled by the difference between the resonance frequency fr and the drive frequency fo. Even if the resonance frequency fr changes, the drive frequency fo can be automatically followed. Therefore, even if the use conditions change, the power transmission efficiency can be easily maintained constant. Further, even when the load LD or inter-coil distance d is changed, the feedback control based on the correction voltage can be kept the load voltage V ₅ constant. By changing the S1 signal based on the correction voltage, the power transmission efficiency can be adjusted afterwards. According to the experiment by the present inventor, no significant power loss was observed by adjusting the level of the S1 signal.

  The T0 signal that is a DC voltage signal is converted into a T4 signal that is an AC optical signal, and is irradiated from the wireless power receiving device 118 to the wireless power feeding device 116. Since the T4 signal, which is an optical signal, is hardly affected by the magnetic field generated by the feeding coil L2 or the like, there is an advantage that the signal can be transmitted satisfactorily.

  Further, the reference potential may be manually adjusted on the power receiving side. The correction voltage is detected not only when the measurement potential changes but also when the reference voltage changes, and as a result, the power transmission efficiency is adjusted. For example, when the reference potential is lowered, feedback control is performed to lower the measurement potential, and the load voltage V5 is also lowered. That is, the power supply can be controlled on the power receiving side.

  As an application example, the wireless power feeding device 116 may be integrated with a table, and the wireless power receiving device 118 may be built in a desk lamp placed on the table. Since conventional desk lamps are obstructed by the power cord, hanging lamps are often used for dining tables. According to the above application example, the power cord of the desk lamp can be eliminated. This increases the usability of the desk lamp. For example, lighting with a desk lamp may make the dish look delicious. A hanging lamp will fix the lighting location, but the desk lamp can be laid out freely, so a variety of lighting is possible. Moreover, even when a plurality of desk lamps are placed on the table, the brightness of the other desk lamps can be collectively controlled by adjusting the reference potential of one desk lamp.

  The present invention has been described based on the embodiments. It will be understood by those skilled in the art that the embodiments are illustrative, and that various modifications and changes are possible within the scope of the claims of the present invention, and that such modifications and changes are also within the scope of the claims of the present invention. By the way. Accordingly, the description and drawings herein are to be regarded as illustrative rather than restrictive.

  The power transmission control circuit 200 is a half-bridge circuit, but may be formed by a push-pull circuit. The S3 signal generated by the current waveform shaping circuit 168 and the T1 signal generated by the control signal generation circuit 170 are not limited to a sawtooth wave, but may be an AC signal whose voltage value gradually increases or decreases in a predetermined period, such as a triangular wave or a sine wave. . In the present embodiment, the current phase is adjusted, but the voltage phase may be adjusted by the T0 signal. Further, feedback control may be performed based on current, power, and the like without being limited to the output voltage.

  The T4 signal is not limited to an optical signal such as infrared rays but may be a radio signal. In any case, a signal in a frequency band sufficiently separated from a frequency band such as the driving frequency fo and the resonance frequency fr may be used. Since the infrared LED 158 and the photodiode 160 are relatively inexpensive, an optical signal is employed in this embodiment.

  DESCRIPTION OF SYMBOLS 100 Wireless power transmission system, 110 Excite circuit, 112 Signal receiving circuit, 114 Phase detection circuit, 116 Wireless power feeder, 118 Wireless power receiver, 120 Feed coil circuit, 122 Signal transmission circuit, 124 Rectifier circuit, 126 Measurement circuit, 128 Bridge Circuit, 130 Power receiving coil circuit, 132 Comparator, 134 Non-adjustment characteristic, 136 Adjustment-time characteristic, 138 Thyristor, 140 Load circuit, 142 Comparator, 144 Current phase detection circuit, 146 Non-adjustment characteristic, 148 Adjustment-time characteristic, 150 phase Comparison circuit, 152 low-pass filter, 154 core, 156 signal control circuit, 158 infrared LED, 160 photodiode, 162 comparator, 164 voltage converter, 166 comparator, 168 Flow waveform shaping circuit, 170 a control signal generating circuit, 172 a reference signal generating circuit, 174 a comparator, 176 a low-pass filter, 200 power transmission control circuit, 202 VCO, L1 exciting coil, L2 feeding coil, L3 receiving coil, L4 loading coil.

Claims (7)

A system for wireless power feeding from a feeding coil to a receiving coil,
A power transmission control circuit for supplying AC power to the power supply coil at a driving frequency;
A power supply coil circuit including the power supply coil and a first capacitor and resonating at a resonance frequency of the power reception coil;
A power receiving coil circuit including the power receiving coil and a second capacitor and resonating at a resonance frequency of the power feeding coil;
A load circuit including a load coil that receives the AC power from the power receiving coil by magnetic coupling with the power receiving coil, and a load that is supplied with power from the load coil;
A phase detection circuit for detecting a phase difference between a voltage phase and a current phase of the AC power;
A signal transmission circuit that transmits an output signal indicating a duty ratio of an output voltage applied to a part of the load circuit to the power supply side;
A signal receiving circuit that receives an output signal indicating an output by a duty ratio from a power receiving side of the AC power, and converts the output signal into a DC according to the duty ratio; and
The power transmission control circuit causes the drive frequency to follow the resonance frequency by adjusting the drive frequency so that the phase difference decreases,
Further, the phase detection circuit post-adjusts the phase value detected for both or one of the voltage phase and the current phase according to the signal level of the DC-converted output signal ,
The signal transmission circuit transmits the output signal as a signal indicating a difference value between the output voltage and a reference voltage by the duty ratio,
A wireless power transmission system, wherein the reference voltage is adjustable by a variable DC voltage source capable of arbitrarily adjusting the voltage . The phase detection circuit includes a first phase value that is a timing when the voltage level of the AC power becomes a first reference value, and a second phase that is a timing when the current level of the AC power becomes a second reference value. The phase difference is detected by comparing values, and both or one of the first and second phase values is changed by changing both or one of the first and second reference values according to the signal level. The wireless power transmission system according to claim 1, wherein post-adjustment is performed. The wireless power transmission system according to claim 1, wherein the signal receiving circuit receives the output signal as an optical signal. An exciting coil that is magnetically coupled to the power supply coil and supplies AC power supplied from the power transmission control circuit to the power supply coil;
The power transmission control circuit includes first and second current paths, and causes the first and second switches connected in series to the first and second current paths to alternately conduct at the drive frequency. The wireless power transmission system according to claim 1, wherein the AC power is supplied to the exciting coil. A detection coil for generating an induced current by a magnetic field generated by the AC power;
5. The wireless power transmission system according to claim 1, wherein the phase detection circuit measures a current phase of the AC power from a phase of the induced current flowing through the detection coil. 6. A control signal generation circuit for generating a control signal at a predetermined control frequency;
A comparison circuit that generates a valid signal when a predetermined magnitude relationship is established between the signal level of the control signal and the output voltage; and
Said signal transmission circuit, a wireless power transmission system according to any of claims 1 5, characterized in that to determine the duty ratio of the output signal by the duty ratio of the useful signal. A reference signal generation circuit for generating a reference signal having a reference frequency higher than the control frequency, and
The wireless power transmission system according to claim 6 , wherein the signal transmission circuit transmits the reference signal as the output signal on condition that the effective signal is being generated.

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