JP5652024B2 - Switching power supply - Google Patents

Description

  The present invention relates to a step-down chopper type switching power supply device that generates and outputs a predetermined voltage by a switching operation.

  2. Description of the Related Art Conventionally, a switching power supply device that performs on / off control of a switching element to perform output voltage control has been used for OA equipment, consumer equipment, and the like. FIG. 8 is a circuit diagram showing a configuration example of a conventional step-down chopper switching power supply controlled in a current mode. Further, FIG. 9 is a waveform diagram in each part for explaining the operation of the conventional switching power supply device. As shown in FIG. 8, the conventional switching power supply device is a power supply device as a DCDC converter that supplies power to the output load 14 using an input voltage source V1, and includes a control circuit 1b, an inductor 12, a smoothing capacitor 13, And a flywheel diode 15. This switching power supply device supplies stable power to the output load 14 by configuring an LC circuit as an output, and performs PWM control of the switching element 2 based on the current flowing through the switching element 2 in the control circuit 1b. Is.

  The control circuit 1b includes a switching element 2, a current sensing element 3, a current detection circuit 4, a feedback control circuit 5, a current control comparator 6, a delay circuit 7, an AND circuit 8, an oscillator 9, a PWM latch 10, and a gate drive circuit 11. In many cases, the circuit is configured as a one-chip integrated circuit.

  The current sense element 3 is an element that is ON / OFF controlled at the same timing as the switching element 2, and a sense current I 2 that is proportional to the current I 1 that flows through the switching element 2 flows. For example, the sense current I2 is a small current such as one hundredth of the current I1. The current detection circuit 4 detects the sense current I2 flowing through the current sense element 3, and outputs a current detection signal V2 proportional to the current I1 based on the detected sense current I2.

  The feedback control circuit 5 outputs an error amplification signal V3 corresponding to the difference between the internal reference voltage and the output voltage V13. The error amplification signal V3 is a signal having a small voltage value when the output voltage is higher than the reference voltage, and conversely a large voltage value when the output voltage is lower than the reference voltage.

  The oscillator 9 outputs an on-trigger signal that is a pulse signal with a constant period to the S terminal of the PWM latch 10. The PWM latch 10 outputs a drive input signal V15 from the Q terminal based on an on trigger signal input to the S terminal and a first off trigger signal V6 described later input to the R terminal. When an on trigger signal is input by the oscillator 9, the PWM latch 10 generates and outputs a high level drive input signal V 15, and turns on the switching element 2 via the gate drive circuit 11.

  The current control comparator 6 compares the error amplification signal V3 and the current detection signal V2, and generates a high-level first off trigger signal V6 when the current detection signal V2 exceeds the error amplification signal V3. The first off trigger signal V6 generated via the AND circuit 8 is output to the R terminal of the PWM latch 10.

  The PWM latch 10 outputs a low-level drive input signal V15 based on the first off trigger signal V6 output by the current control comparator 6, and turns off the switching element 2 via the gate drive circuit 11.

  That is, the conventional switching power supply device shown in FIG. 8 controls on / off of the switching element 2 connected to the input voltage source V1 based on the on-trigger signal and the first off-trigger signal V6, and outputs load 14 to transmit energy.

  The delay circuit 7 receives the drive input signal V15 output from the PWM latch 10, delays the phase by a predetermined time, and outputs the delayed signal to the AND circuit 8 as the mask signal V4. As a result, while the mask signal V4 output by the delay circuit 7 is low, even if the current detection signal V2 exceeds the error amplification signal V3, the AND circuit 8 does not output the first off trigger signal V6, The switching element 2 is prevented from turning off.

  The setting of the mask period by the delay circuit 7 is to prevent malfunction caused by the spike current of the current detection signal V2. More specifically, at the moment when the switching element 2 is turned on, the recovery current I6 flows from the cathode to the anode in the flywheel diode 15 during the reverse recovery time. Therefore, the drain current I1 is superimposed with the spike-like recovery current I6 and the current flowing through the parasitic capacitance with respect to the inductor current I5, and the spike current component is also superimposed with respect to the current detection signal V2. . Therefore, if the above-described mask period is not provided, the current detection signal V2 on which the spike current component is superimposed at the moment when the switching element 2 is turned on exceeds the error amplification signal V3, and the first off trigger signal V6. May be generated and the switching element 2 may be turned off immediately.

  However, since the switching power supply device shown in FIG. 8 has a mask period T1 of about 100 nS after the switching element 2 is turned on by the delay circuit 7 (see FIG. 9), the switching element 2 is turned off in the mask period. It is possible to prevent malfunction caused by spike current.

  The delay time by the delay circuit 7 is appropriately set so that the mask signal V4 is kept at a low level for a predetermined mask period (a time of about 100 nS) after the switching element 2 is turned on in order to prevent the malfunction described above. It has been adjusted.

  Patent Document 1 describes a switching power supply that suppresses current waveform imbalance and has stable control characteristics. This switching power supply includes a control circuit for current mode control that detects an output voltage and outputs a control signal to a main switch provided on the input side. In this control circuit, an output voltage detection means, an inductor current detection means, First compensation waveform generating means for generating a quadratic curve or a compensation waveform similar to the quadratic curve and means for synthesizing these waveforms are provided. Further, the control circuit combines the signal detected by the output voltage detection means, the signal detected by the inductor current detection means, and the compensation waveform obtained from the compensation waveform generation means to generate two signals, Comparing means for comparing the signals and outputting them to the main switch is provided.

  Therefore, according to this switching power source, even if the phase compensation capacitor of the error amplifier is made small and the frequency at which the loop gain of the error amplifier becomes 0 dB is high enough to approach the switching frequency, low frequency oscillation of the inductor current is unlikely to occur. Thus, by using the compensation waveform, a high-speed response having a loop response frequency close to the switching frequency can be realized, and a small and highly reliable power source can be realized at low cost.

Japanese Patent Laid-Open No. 2005-110390

  Here, the duty ratio of the switching element 2 is determined by the input / output voltage ratio (V13 / V1). That is, in the duty ratio of the switching element 2, the ON width is wide when the input / output voltage difference is small, and the ON width is narrow when the input / output voltage difference is large. Therefore, when the output voltage V13 is kept constant, the switching element 2 is controlled with a wide on-width when the input voltage V1 is low (low input), and narrow on when the input voltage V1 is high (high input). Controlled by width.

  However, as shown in FIG. 9, when the input voltage V1 is extremely high or the switching frequency is high, the ON width is equal to or smaller than the mask period. There is a disadvantage that the switching element 2 cannot be turned off. That is, in the conventional switching power supply device shown in FIG. 8, the provision of the mask period is an obstacle, and the switching element 2 cannot be turned off at an appropriate timing, and is turned off after the mask period ends. It can be considered that the ON width is wider than the original ON width, and the output voltage V13 gradually rises and becomes uncontrollable as shown on the right side of FIG. In the worst case, it is assumed that the output voltage V13 rises beyond a specified value, and the circuit in the output load 14 is destroyed.

  Further, as the input voltage V1 is higher, the peak value of the recovery current I6 that instantaneously flows through the flywheel diode 15 increases and the duration of the surge current also increases, so that appropriate control of the switching element 2 in the current mode method is achieved. There is also a problem that becomes more difficult.

  Note that FIG. 9 does not necessarily indicate that the input voltage V1 is switched from a low input to a high input in the middle, and a case where the input is low and a case where the input is high are shown by a single waveform diagram. Since there is a great need for a switching power supply device with a wide input voltage range that operates properly at both high and low inputs, it is important to solve the above-described problems.

  Furthermore, the switching power supply device needs to ensure a sufficient phase margin so that stable operation can be obtained in consideration of frequency characteristics (gain / phase characteristics).

  An object of the present invention is to solve the above-described problems of the prior art, and to provide a switching power supply device that operates stably even in a region where the ON width of a switching element due to a high input voltage or the like is narrow.

In order to solve the above problems, a switching power supply according to the present invention is a switching power supply that converts a DC voltage input by a switching operation of a main switching element into a predetermined voltage and supplies the voltage to a load. Based on a comparison result between an error amplification signal corresponding to the difference between the output voltage value supplied to the predetermined reference voltage value and a current detection signal corresponding to the current value flowing through the main switching element. A current mode control unit that performs on / off control, a voltage mode control unit that performs on / off control of the main switching element based on a comparison result between a voltage value of the error amplification signal and a ramp signal of a predetermined frequency, and A control mode selection unit that selects either a current mode control unit or the voltage mode control unit to control on / off of the main switching element. , E Bei a phase compensation unit for adjusting the phase compensation amount such that the phase compensation amount in accordance with the selection result by the control mode selecting section, the current-mode control unit detects a current value flowing through the main switching device And a current detection unit that generates a current detection signal according to the current value, and a feedback control unit that generates an error amplification signal according to the difference between the output voltage value supplied to the load and a predetermined reference voltage value And the current detection signal generated by the current detection unit and the error amplification signal generated by the feedback control unit, and the voltage value of the current detection signal exceeds the voltage value of the error amplification signal. A current control comparator that generates a first off-trigger signal for off-controlling the main switching element; and a spy that flows when the main switching element is turned on. A mask signal generating unit that generates a mask signal for preventing the main switching element from being controlled to be turned off by the first off trigger signal according to the current, and the voltage mode control unit includes the control mode. When the voltage mode control unit is selected by the selection unit, the error amplification signal is compared with the ramp signal, and the mask signal is compared when the voltage value of the ramp signal exceeds the voltage value of the error amplification signal. And a voltage control comparator for generating a second off-trigger signal for off-controlling the main switching element regardless of the state .

  ADVANTAGE OF THE INVENTION According to this invention, the switching power supply device which operate | moves stably also in the area | region where the ON width | variety of the switching element resulting from a high input voltage etc. is narrow can be provided.

It is a circuit diagram which shows the structure of the switching power supply device of the form of Example 1 of this invention. It is a wave form diagram of each part which shows operation | movement of the switching power supply device of the form of Example 1 of this invention. It is a circuit diagram which shows the detailed structure of the feedback control circuit of the switching power supply device of the form of Example 1 of this invention. It is a figure which shows the frequency characteristic in an open loop at the time of the current mode of the switching power supply device of the form of Example 1 of this invention. It is a figure which shows the frequency characteristic at the time of assuming that it does not switch phase compensation in the current mode at the time of the voltage mode of the switching power supply device of the form of Example 1 of this invention. It is a figure which shows the frequency characteristic in an open loop at the time of the voltage mode of the switching power supply device of the form of Example 1 of this invention. It is a figure which shows the frequency characteristic at the time of assuming that it does not switch phase compensation in the voltage mode at the time of the current mode of the switching power supply device of the form of Example 1 of this invention. It is a circuit diagram which shows the structure of the conventional switching power supply apparatus. It is a wave form diagram of each part which shows operation | movement of the conventional switching power supply apparatus.

  Embodiments of a switching power supply apparatus according to the present invention will be described below in detail with reference to the drawings.

  Embodiments of the present invention will be described below with reference to the drawings. First, the configuration of the present embodiment will be described. FIG. 1 is a circuit diagram showing a configuration of a switching power supply apparatus according to Embodiment 1 of the present invention. This switching power supply device is a power supply device that converts a DC voltage V1 input by a switching operation of the switching element 2 as a main switching element into a predetermined voltage and supplies the voltage to a load 14, and includes a control circuit 1, an inductor 12, and a smoothing device. The capacitor 13 and the flywheel diode 15 are included. The switching element 2 is, for example, a high side MOSFET. The control circuit 1 includes a switching element 2, a current sense element 3, a current detection circuit 4, a feedback control circuit 5, a current control comparator 6, a delay circuit 7, an AND circuit 8, an oscillator 9, a PWM latch 10, and a gate drive circuit 11. , And a voltage mode control circuit 16.

  That is, the switching power supply device of this embodiment has a configuration including the voltage mode control circuit 16 in addition to the configuration of the conventional switching power supply device described with reference to FIG. 8, and functions as a DCDC converter. In FIG. 1, the same or equivalent components as those in FIG. 8 are denoted by the same reference numerals as those described above, and redundant description is omitted.

  The current detection circuit 4, the feedback control circuit 5, the current control comparator 6, the delay circuit 7, the AND circuit 8, the oscillator 9, the PWM latch 10, and the gate drive circuit 11 correspond to the current mode control unit of the present invention, and the load 14 Based on the comparison result between the error amplification signal V3 corresponding to the difference between the output voltage value supplied to the predetermined reference voltage value and the current detection signal V2 corresponding to the current value flowing through the switching element 2, Perform on / off control. Each configuration of the current mode control unit will be described below.

  The current detection circuit 4 corresponds to the current detection unit of the present invention, detects the current value flowing through the switching element 2, and generates a current detection signal V2 corresponding to the current value. Specifically, the current detection circuit 4 detects a sense current I2 flowing through the current sense element 3, and outputs a current detection signal V2 proportional to the current I1 based on the detected sense current I2.

  The feedback control circuit 5 corresponds to the feedback control unit of the present invention, and generates an error amplification signal V3 corresponding to the difference between the output voltage value supplied to the load 14 and a predetermined reference voltage value. The error amplification signal V3 is a signal having a small voltage value when the output voltage is higher than the reference voltage, and conversely a large voltage value when the output voltage is lower than the reference voltage.

  The feedback control circuit 5 corresponds to the phase compensation unit of the present invention, and adjusts the phase compensation amount so that the phase compensation amount corresponds to the selection result by the control mode selection unit described later. A detailed description of the feedback control circuit 5 as a phase compensation unit will be given later.

  The current control comparator 6 corresponds to the current control comparison unit of the present invention, compares the current detection signal V2 generated by the current detection circuit 4 with the error amplification signal V3 generated by the feedback control unit 5, and compares the current detection signal V2 with the current detection signal V3. When V2 exceeds the error amplification signal V3, the first off trigger signal V6 for turning off the switching element 2 is generated, and the first off trigger signal V6 generated through the AND circuit 8 is PWM latched 10 Output to the R terminal.

  The oscillator 9 outputs an on-trigger signal that is a pulse signal having a constant period to the S terminal of the PWM latch 10, and controls the switching element 2 together with the PWM latch 10. The PWM latch 10 is the same as the PWM latch 10 in the conventional switching power supply device described in FIG.

  The delay circuit 7 corresponds to the mask signal generation unit of the present invention, and prevents the switching element 2 from being turned off by the first off trigger signal V6 corresponding to the spike current that flows when the switching element 2 is turned on. A mask signal is generated for this purpose. Specifically, the delay circuit 7 delays the phase of the drive input signal V15 output from the PWM latch 10 by a predetermined time and outputs the delayed signal to the AND circuit 8 as the mask signal V4. Thus, while the mask signal V4 output by the delay circuit 7 is at a low level, the first off trigger signal V6 is not output from the AND circuit 8 even if the current detection signal V2 exceeds the error amplification signal V3, and switching is performed. Element 2 does not turn off.

  The voltage mode control circuit 16 corresponds to the voltage mode control unit of the present invention, and performs on / off control of the switching element 2 based on a comparison result between the error amplification signal V3 and a ramp signal having a predetermined frequency. Strictly speaking, the voltage mode control circuit 16 performs only the off control of the switching element 2, and the on control is performed by the oscillator 9, the PWM latch 10, and the gate drive circuit 11. Therefore, it can be said that the oscillator 9, the PWM latch 10, and the gate drive circuit 11 are a shared part of the current mode control unit and the voltage mode control unit.

  The voltage mode control circuit 16 includes a constant current source I3, a capacitor 161, a Schmitt circuit 162, an SR flip-flop 163, a switch 164, an inverter 165, a voltage control comparator 166, an AND circuit 167, an OR circuit 168, an AND circuit 169, and an AND circuit 170. , SR flip-flop 171 and 1-shot circuit 172.

  The capacitor 161 is charged by the constant current I3 when the switch 164 is in the open state, generates the ramp signal V10, and outputs the ramp signal V10 to the non-inverting input terminal of the voltage control comparator 166 and the Schmitt circuit 162.

  The SR flip-flop 163 outputs a control signal from the Q terminal to the switch 164 based on the signal input from the Schmitt circuit 162 to the S terminal and the signal input from the one-shot circuit 172 to the R terminal. 164 is controlled.

  When the drive input signal V15 is switched from the low level to the high level, the one-shot circuit 172 outputs a high level reset trigger signal to the R terminal of the SR flip-flop 163 for a certain period, and the SR flip-flop 163 is reset. And the switch 164 is opened.

  The Schmitt circuit 162 outputs the input detection clock V5 to the AND circuit 170 via the inverter 165 when the ramp signal V10 reaches a predetermined threshold value, and sets the SR flip-flop 163 to the set state to switch the Schmitt circuit 162. 164 is made conductive. Thereby, the electric charge of the capacitor 161 is discharged.

  Therefore, the ramp signal V10 and the input detection clock V5 are signals having a predetermined frequency synchronized with the on trigger signal of the oscillator 9. However, an important point for the input detection clock V5 is not to have a high timing, but to maintain a high state for a certain period (for example, 100 nS) from the time when the mask signal V4 is switched from low to high. The timing at which the input detection clock V5 changes from high to low can be adjusted by selecting the Schmitt circuit 162 having an appropriate threshold value.

  The AND circuit 169, the AND circuit 170, and the SR flip-flop 171 correspond to the control mode selection unit of the present invention, and select either the current mode control unit or the voltage mode control unit to turn on / off the switching element 2. Let me control. In the switching power supply of this embodiment, the control mode selection unit is built in the voltage mode control circuit 16, but it is not always necessary to be built in the voltage mode control circuit 16.

  The control mode selection unit selects the voltage mode control unit to perform on / off control of the switching element 2 when the ON width of the switching element 2 is equal to or shorter than the first predetermined period. Further, the control mode selection unit detects that the ON width of the switching element 2 is equal to or less than the first predetermined period based on the timing of the OFF control by the current mode control unit. In this embodiment, the first predetermined period is a period having the same length as the mask period. Further, the off control by the current mode control unit indicates control by the first off trigger signal V6. When both the input detection clock V5 and the first off-trigger signal V6 are in the high state, the switching element 2 is controlled to be turned off immediately after the mask period ends, so that the ON width is equal to or less than the first predetermined period. I know that there is.

  That is, the control mode selection unit determines whether or not both of the input detection clock V5 and the first off-trigger signal V6 are in the high state by the AND circuit 169, so that the ON width of the switching element 2 is the first predetermined value. Detect that it is less than the period.

  Therefore, when the control mode selection unit detects that the ON width of the switching element 2 is equal to or less than the first predetermined period as described above, the AND circuit 169 sets the SR flip-flop 171 in the set state, thereby causing the voltage mode. The control unit is selected to control the switching element 2 on / off. This is realized by enabling the voltage control comparator 166 described later to output the second off trigger signal V11 via the AND circuit 167 when the SR flip-flop 171 is set.

  Further, the control mode selection unit selects the current mode control unit to perform on / off control of the switching element 2 when the ON width of the switching element 2 is equal to or longer than the second predetermined period. Here, the control mode selection unit detects that the ON width of the switching element 2 is equal to or longer than the second predetermined period when the switching element 2 is in the ON state after the mask period by the mask signal V4 ends. That is, the control mode selection unit determines whether or not the mask signal V4, the input detection clock V5, and the drive control signal V15 are all in the high state by the AND circuit 170, so that the ON width of the switching element 2 is increased. It is detected that the period is equal to or longer than the second predetermined period.

  Accordingly, when the control mode selection unit detects that the ON width of the switching element 2 is equal to or longer than the second predetermined period as described above, the AND circuit 170 sets the SR flip-flop 171 to the reset state, thereby causing the current mode. The control unit is selected and the switching element 2 is controlled to be turned on / off. This is because when the SR flip-flop 171 is reset, the voltage control comparator 166 cannot output the second off trigger signal V11 via the AND circuit 167, and the control by the voltage mode control unit becomes invalid. Because it becomes.

  That is, the control mode selection unit sets the SR flip-flop 171 to the reset state when selecting the current mode control unit, and sets the SR flip-flop 171 to the set state when selecting the voltage mode control unit.

  The voltage control comparator 166 corresponds to the voltage control comparison unit of the present invention. When the voltage mode control unit is selected by the control mode selection unit, the voltage control comparator 166 compares the error amplification signal V3 with the ramp signal V10, and the ramp signal V10 is When the error amplification signal V3 is exceeded, a second off trigger signal V11 for controlling the switching element 2 to be turned off is generated regardless of the state of the mask signal V4. Since the second off trigger signal V11 is output to the R terminal of the PWM latch 10 via the AND circuit 167 and the OR circuit 168, it is not affected by the mask signal V4.

  The OR circuit 168 outputs a high level signal to the R of the PWM latch 10 when the high level first off trigger signal V6 or the second off trigger signal V11 is input by either the AND circuit 8 or the AND circuit 167. Output to the terminal.

  Next, the operation of the present embodiment configured as described above will be described. FIG. 2 is a waveform diagram of each part showing the operation of the switching power supply device of the present embodiment. First, the case where the input voltage is low will be described. In this case, the switching power supply device of this embodiment operates with normal current mode control. The current detection circuit 4 detects the sense current I2 flowing through the current sense element 3, generates a current detection signal V2 proportional to the current I1 based on the detected sense current I2, and supplies it to the positive input terminal of the current control comparator 6. Output.

  On the other hand, the feedback control circuit 5 generates an error amplification signal V3 corresponding to the difference between the output voltage value supplied to the load 14 and a predetermined reference voltage value, and the inverting input terminal of the current control comparator 6 and the voltage control comparator 166. Output to the inverting input terminal.

  The oscillator 9 outputs an on-trigger signal that is a pulse signal having a constant period to the S terminal of the PWM latch 10, and controls the switching element 2 together with the PWM latch 10. Further, the delay circuit 7 outputs to the AND circuit 8 a mask signal V4 delayed in phase by a predetermined time (for example, about 100 nS) based on the drive input signal V15 output from the PWM latch 10.

  The current control comparator 6 compares the current detection signal V2 generated by the current detection circuit 4 with the error amplification signal V3 generated by the feedback control unit 5, and when the current detection signal V2 exceeds the error amplification signal V3. The first off trigger signal V6 for controlling the switching element 2 to be turned off is generated.

  The AND circuit 8 sends the first off-trigger signal V6 to the R terminal of the PWM latch 10 via the OR circuit 168 when the mask signal V4 is high and the output signal of the current control comparator 6 is high. Output. At that time, since the gate drive signal V12 is switched from high to low, the switching element 2 is turned off.

  When the mask signal V4 is in the low state, even if the output signal of the current control comparator 6 is in the high state, the AND circuit 8 does not output the first off-trigger signal V6 and does not output the spike current. To prevent malfunctions.

  Next, the operation when the input voltage V1 is increased and the mode is changed from the current mode to the voltage mode will be described. Since the on-duty width of the switching element 2 is determined by the output voltage V13 / the input voltage V1, it operates with a narrow on-duty width at a high input voltage than at a low input voltage. At this time, if the ON width of the switching element 2 is equal to or shorter than the first predetermined period (mask period T2 in this embodiment), the control mode selection unit selects the voltage mode control unit and turns on the switching element 2. Control off / on.

  Specifically, when the ON width of the switching element 2 is equal to or shorter than the mask period T2, the AND circuit 8 is connected to the R terminal of the PWM latch 10 in the period of about 100 nS until the mask period T2 ends. 1 The off trigger signal V6 cannot be output. Therefore, the switching element 2 is not turned off during that time, and is turned off after the mask period T2 ends.

  In this case, the control mode selection unit detects that both the input detection clock V5 and the first off trigger signal V6 are in the high state by the AND circuit 169, and the ON width of the switching element 2 is equal to or less than the first predetermined period. As a result, the high input detection trigger V7 is output to the S terminal of the SR flip-flop 171. As a result, the SR flip-flop 171 enters the set state, and outputs the high state input detection signal V9 to the AND circuit 167, thereby bringing the voltage mode control circuit 16 into the active state. As described above, when the control mode selection unit selects the voltage mode control unit, the current mode is changed to the voltage mode, and the switching power supply device of this embodiment operates in the voltage mode after the next switching cycle.

  When the voltage mode control circuit 16 becomes active, the feedback control circuit 5 reduces the control voltage of the error amplification signal V3 to fall within the amplitude range of the ramp signal V10 in order to keep the output voltage V13 constant.

  The voltage mode control circuit 16 performs on / off control of the switching element 2 based on the comparison result between the error amplification signal V3 and the ramp signal having a predetermined frequency. More specifically, the voltage control comparator 166 in the voltage mode control circuit 16 compares the error amplification signal V3 with the ramp signal V10, and the state of the mask signal V4 when the ramp signal V10 exceeds the error amplification signal V3. Regardless, the second off trigger signal V11 for turning off the switching element 2 is generated.

  The second off trigger signal V11 is input to the R terminal of the PWM latch 10 via the AND circuit 167 and the OR circuit 168, thereby turning off the switching element 2.

  As described above, the voltage mode control circuit 16 can perform on / off control of the switching element 2 by PWM control even within the mask period.

  Next, the operation when the input voltage V1 is changed from the high state to the low state to return from the voltage mode to the current mode will be described. When the input voltage V1 decreases, the charging slope of the inductor current I5 becomes gentle, and the output voltage V13 slightly decreases. Therefore, the feedback control circuit 5 increases the voltage of the error amplification signal V3 in order to keep the output voltage V13 constant. When the voltage of the error amplification signal V3 rises outside the amplitude range of the ramp signal V10, the switching element 2 is controlled to be turned off by the first off trigger signal V6 instead of the second off trigger signal V11. Controlled with a wide on-width.

  Here, the control mode selection unit selects the current mode control unit to perform on / off control of the switching element 2 when the ON width of the switching element 2 is equal to or longer than the second predetermined period. More specifically, the control mode selection unit detects that all of the mask signal V4, the input detection clock V5, and the drive control signal V15 are in the high state by the AND circuit 170, so that the ON width of the switching element 2 is set. Is detected for the second predetermined period or longer. At that time, the AND circuit 170 switches the input detection signal V9 from high to low by outputting the low input detection trigger V8 to the R terminal of the SR flip-flop 171.

  As described above, when the control mode selection unit selects the current mode control unit, the voltage mode is changed to the current mode, and the switching power supply device of this embodiment operates in the current mode after the next switching cycle.

  The switching power supply device according to the present embodiment operates by current mode control when the ON width of the switching element 2 is relatively wide at a low input voltage by the series of operations described above, and the ON width of the switching element 2 at a high input voltage. Is extremely narrow, it is possible to ensure stable operation over a wide input voltage range by automatically switching to voltage mode control.

  Next, the phase compensation operation of the switching power supply device of this embodiment will be described. The switching power supply device of this embodiment switches between current mode control and voltage mode control in order to realize a non-insulated step-down DC-DC converter with a wide input range that operates properly regardless of whether the input is high or low. Since current mode control and voltage mode control have different frequency characteristics (gain / phase characteristics) at a small signal control level, stable operation cannot be obtained simply by switching modes.

  Therefore, the switching power supply device according to the present embodiment has a configuration in which the phase compensation amount in the feedback control circuit 5 is changed according to the mode in order to obtain a stable operation in each mode, and a sufficient phase margin can be secured. Yes. Specifically, the feedback control circuit 5 as the phase compensation unit performs phase compensation so that the phase advances, for example, when the control mode selection unit selects the voltage mode control unit.

  FIG. 3 is a circuit diagram showing a detailed configuration of the feedback control circuit 5 of the switching power supply device of this embodiment. As shown in FIG. 3, the feedback control circuit 5 as a phase compensation unit has a phase compensation capacitor C3 connected in parallel with the resistor R1 and connected in series with the switch 200 at the input stage of the error amplifier 201. ing.

  The feedback control circuit 5 of this embodiment includes an error amplifier 201, a reference voltage 202, resistors R1 and R2 for setting the gain of the error amplifier 201, and phase compensation capacitors C1 to C3. The output voltage V13 is connected to one terminal of a capacitor C3 connected in parallel with the resistor R1, and the other terminal of the resistor R1 is the other terminal of the switch 200, the inverting input terminal of the error amplifier 201, and the phase compensation capacitor C2. And the resistor R2 are connected in parallel to one terminal, the phase compensation capacitor C2, the other terminal in which the resistor R2 is connected in parallel and one terminal of the phase compensation capacitor C1 are connected, and the phase compensation capacitor The other terminal of C1 is connected to the output terminal of the error amplifier 201 and is output as an error amplification signal V3.

  Here, the phase compensation capacitors C1 and C2 act to delay the phase.

  The switch 200 is a switch that closes (turns on) when the input detection signal V9 is in a high state and opens (turns off) when the input detection signal V9 is in a low state. The switch 200 may be of any type as long as it can be electrically turned on / off. The input detection signal V9 is a signal output from the SR flip-flop 171 as described above, and is in a low state during the current mode control and is in a high state during the voltage control mode.

  That is, the feedback control circuit 5 as the phase compensation unit can change the phase compensation according to the mode. Specifically, when the control mode selection unit selects the voltage mode control unit, the switch 200 is turned on. Then, the phase compensation is performed by the phase compensation capacitor C3 so that the phase advances, for example.

  FIG. 4 is a diagram illustrating frequency characteristics (gain / phase characteristics) in an open loop in the current mode of the switching power supply device according to the present embodiment. Since it is in the current mode, the input detection signal V9 is in the low state, and the switch 200 is open (off).

  Ro shown in FIG. 4 is the resistance value of the load 14 shown in FIG. Further, Co shown in FIG. 4 is a capacitance value of the smoothing capacitor 13 shown in FIG. Also, Adc shown in FIG. 4 indicates the open-loop DC gain of the entire control, and Av indicates the voltage gain of the feedback control circuit 5 (error amplifier configuration). Further, “error amplifier” in FIG. 4 indicates the error amplifier 201.

  As shown in FIG. 4, the frequency characteristic (gain / phase characteristic) due to the power stage in the current mode (consisting of the switching element 2, the flywheel diode 15, the inductor 12, the smoothing capacitor 13, and the load 14) is the smoothing capacitor 13. And one pole fp constituted by the load 14.

  In general, the crossover frequency fc of the DC-DC converter is set to 1/5 to 1/20 of the switching frequency by the feedback control circuit 5 (error amplifier), 2 poles (fp1, fp2), 1 zero. Design (fz1). However, the switching frequency in FIGS. 4 to 7 is assumed to be about 100 kHz to 1 MHz.

  Expressions of fp1, fp2, and fz1 shown in FIG. 4 are expressions when the switch 200 is in an off state (during current mode control), and are derived from the detailed circuit of the feedback control circuit 5 of FIG.

  As shown in FIG. 4, in the switching power supply of this embodiment, during current mode control, the phase of the entire control at the crossover frequency fc at which the gain of the entire control is 0 dB is about −135 degrees, and the stable control is achieved. Since there is a sufficient phase margin with respect to less than -180 degrees which is an exponent, stable control can be performed.

  FIG. 5 is a diagram illustrating frequency characteristics (gain / phase characteristics) when it is assumed that the phase compensation is not switched in the current mode in the voltage mode of the switching power supply device of the present embodiment. That is, FIG. 5 shows what happens when the switch 200 in the switching power supply device of this embodiment is turned on in the voltage mode as described above, but remains off.

  Note that L shown in FIG. 5 is the inductance value of the inductor 12 shown in FIG. As shown in FIG. 5, in the voltage mode, unlike the current mode, the power stage becomes a secondary filter of the inductor 12 and the smoothing capacitor 13, and the phase turns around -180 degrees immediately before the resonance frequency fo. The voltage gain of the control circuit is 45 dB when the phase characteristic of the entire control is at a phase of −180 degrees, which is much higher than 0 dB. Therefore, the control becomes unstable.

  On the other hand, FIG. 6 is a diagram showing a feedback control circuit frequency characteristic (gain / phase characteristic) in an open loop in the voltage mode of the switching power supply device of the present embodiment, and shows a case where the switch 200 is in an ON state. . As shown in FIG. 6, in the switching power supply device of this embodiment, when the switch 200 is turned on when switching to the voltage mode control, zero fz2 is added due to the influence of the phase compensation capacitor C3, and the phase is changed. Phase compensation is performed so as to advance. As a result, the maximum phase is improved to about 135 degrees, and a margin of about 45 degrees with respect to the -180 degree phase is formed, so that the control becomes stable.

  FIG. 7 is a diagram illustrating frequency characteristics when it is assumed that the phase compensation is not switched in the voltage mode in the current mode of the switching power supply device of the present embodiment. That is, FIG. 7 shows what happens when the switch 200 in the switching power supply of this embodiment is turned off in the current mode as described above, but remains on.

  As described above, unless the crossover frequency fc of the DC-DC converter is generally set to 1/5 to 1/20 of the switching frequency, the DC-DC converter reacts to its own switching frequency component each time. As a result, control becomes unstable. However, as shown in FIG. 7, if the switch 200 remains on in the current mode (that is, when phase compensation in the voltage mode is applied to the current mode), the crossover frequency is set to 10 MHz or more. It becomes unstable because it picks up high frequency noise such as switching noise and radio noise.

  Therefore, it is necessary to switch the phase compensation according to the control mode, and the switching power supply according to the present embodiment includes the switch 200 so that the phase compensation unit adjusts the phase compensation amount according to the selection result of the control mode selection unit. Therefore, stable operation can be ensured in any control mode.

  As described above, the switching power supply device according to the first embodiment of the present invention can operate stably even in a region where the ON width of the switching element due to a high input voltage or the like is narrow.

  That is, the switching power supply device of the present embodiment includes the control mode selection unit that switches between the current mode control and the voltage mode control according to the ON width of the switching element 2, so that the switching element 2 that is a high-side MOSFET is turned on. When the width is wider than the mask time, it operates in current mode control. When the ON width of the switching element 2 is equal to or narrower than the mask time, it can be automatically switched to voltage mode control that does not require current information. Appropriate PWM control can be performed on the switching element 2 even in a region where the ON width is equal to or shorter than the mask time.

  Thereby, the switching power supply device of the present embodiment appropriately performs PWM control on the switching element 2 even when the input voltage difference is wide and the on-duty width is extremely narrow, or even when the switching frequency is high and the on-width is extremely narrow, It is possible to avoid a trouble that the output voltage V13 rises to a specified value or more and destroys the circuit in the load 14.

  Moreover, since the switching power supply apparatus of the present embodiment includes a phase compensation unit that adjusts the phase compensation amount according to the selection result by the control mode selection unit, the frequency characteristics ( A sufficient phase margin can be secured so that stable operation can be obtained in consideration of the gain / phase characteristics.

  As a modification, it is possible to replace the flywheel diode 15 of FIG. 1 with a power switch and change it to a synchronous rectifier circuit, or to make a system in which the changed power switch is provided inside the control circuit 1 It is. The output voltage V13 may be divided and input to the feedback control circuit 5.

  The switching power supply device according to the present invention can be used for a switching power supply device used in electrical equipment and the like that require stable power supply.

DESCRIPTION OF SYMBOLS 1,1b Control circuit 2 Switching element 3 Current sense element 4 Current detection circuit 5 Feedback control circuit 6 Current control comparator 7 Delay circuit 8 AND circuit 9 Oscillator 10 PWM latch 11 Gate drive circuit 12 Inductor 13 Smoothing capacitor 14 Load 15 Flywheel diode 16 voltage mode control circuit 161 capacitor 162 Schmitt circuit 163 SR flip-flop 164 switch 165 inverter 166 voltage control comparator 167 AND circuit 168 OR circuit 169 AND circuit 170 AND circuit 171 SR flip-flop 172 1-shot circuit 200 switch 201 error amplifier 202 reference voltage C1 to C3 Phase compensation capacitors R1, R2 resistors

Claims (7)

A switching power supply device that converts a DC voltage input by a switching operation of a main switching element into a predetermined voltage and supplies the voltage to a load,
The main switching based on a comparison result between an error amplification signal according to a difference between an output voltage value supplied to the load and a predetermined reference voltage value and a current detection signal according to a current value flowing through the main switching element. A current mode control unit for performing on / off control of the element;
A voltage mode control unit that performs on / off control of the main switching element based on a comparison result between a voltage value of the error amplification signal and a ramp signal of a predetermined frequency;
A control mode selection unit that selects either the current mode control unit or the voltage mode control unit to perform on / off control of the main switching element;
A phase compensation unit that adjusts the phase compensation amount so as to be a phase compensation amount according to a selection result by the control mode selection unit;
Bei to give a,
The current mode controller is
A current detection unit that detects a current value flowing through the main switching element and generates a current detection signal according to the current value;
A feedback control unit that generates an error amplification signal according to a difference between an output voltage value supplied to the load and a predetermined reference voltage value;
The current detection signal generated by the current detection unit is compared with the error amplification signal generated by the feedback control unit, and when the voltage value of the current detection signal exceeds the voltage value of the error amplification signal, the main detection signal is generated. A current control comparator for generating a first off-trigger signal for off-controlling the switching element;
A mask signal generating unit that generates a mask signal for preventing the main switching element from being turned off by the first off-trigger signal according to a spike current that flows when the main switching element is turned on;
Have
The voltage mode control unit compares the error amplification signal and the ramp signal when the voltage mode control unit is selected by the control mode selection unit, and the voltage value of the ramp signal is equal to the error amplification signal. A switching power supply device comprising: a voltage control comparator for generating a second off trigger signal for controlling off the main switching element regardless of the state of the mask signal when a voltage value is exceeded .   The switching power supply according to claim 1, wherein the phase compensation unit performs phase compensation so that the phase advances when the control mode selection unit selects the voltage mode control unit.   The control mode selection unit selects the voltage mode control unit to perform on / off control of the main switching element when an ON width of the main switching element is equal to or less than a first predetermined period. The switching power supply device according to claim 1 or 2.   The said control mode selection part detects that the ON width | variety of the said main switching element is below a 1st predetermined period based on the timing of the OFF control by the said current mode control part. Switching power supply.   The control mode selection unit selects the current mode control unit to perform on / off control of the main switching element when the ON width of the main switching element is equal to or longer than a second predetermined period. The switching power supply device according to any one of claims 1 to 4. The phase compensation unit includes a phase compensation capacitor connected in parallel with a resistor and connected in series with a switch in front of an error amplifier that generates an error amplification signal in the feedback control unit, and the control mode selection unit 2. The switching power supply device according to claim 1, wherein when the voltage mode control unit is selected, the switch is turned on to perform phase compensation so that the phase is advanced by the phase compensation capacitor. The control mode selection unit detects that an ON width of the main switching element is equal to or longer than a second predetermined period when the main switching element is in an ON state after the mask period by the mask signal is ended, and the current mode control 2. The switching power supply device according to claim 1, wherein the main switching element is controlled to be turned on / off by selecting a part.

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