JP5594029B2 - Feedback control circuit, feedback control system - Google Patents

Description

  The present invention relates to a feedback control circuit that processes a signal input from a controlled device to generate an output signal and feeds back the output signal to the controlled device, and feedback control including the feedback control circuit and the controlled device About the system.

  Conventionally, a feedback control circuit for controlling a controlled device such as a sensor for detecting a physical quantity or a servo mechanism is known. This feedback control circuit can improve the characteristics of the signal input from the controlled device, and can stabilize the operation and output of the controlled device.

This type of feedback control circuit is preferably applied, for example, to MEMS (Micro
Electro Mechanical Systems) Inertial force sensor system using electrostatic force feedback of weight, control system for controlling motor position, power supply control system including LC filter, and the like. As a representative inertial force sensor, there is an angular velocity sensor known as a gyroscope or a yaw rate sensor. In these systems, the input-output characteristic of the controlled device may exhibit a second-order LPF (low-pass filter) characteristic or a resonance characteristic (a characteristic that exhibits a high amplitude characteristic at a specific resonance frequency). The controlled device can be stably controlled without abnormal oscillation of the controlled device.

  The angular velocity sensor detects a Coriolis force generated in a weight (vibrator) that is vibrated and driven by a rotational motion, but its output signal is usually a minute signal. For this reason, in the feedback control circuit that controls the angular velocity sensor, the resonance Q value of the weight is used to increase the amplitude in the vibration frequency band to be detected, that is, the sensitivity, thereby improving the detection resolution.

  Also, in a feedback control system using an angular velocity sensor as a controlled device, by feeding back the output signal of the feedback control circuit to the electrostatic force applied to the weight, the sensitivity fluctuation or non-linearity of the angular velocity sensor due to variations in Q value or temperature characteristics Control is performed to stabilize the characteristics.

  Such a control technique is referred to as “electrostatic force feedback”, “force feedback”, “servo”, and the like. Non-Patent Document 1 describes a technique for performing force feedback on an inertial sensor.

  FIG. 1 is a configuration example of a conventional angular velocity sensor system controlled by force feedback. In the figure, characters in the functional block frame indicate a transfer function or the like in Z conversion. In this conventional angular velocity sensor, a controlled device A including a sensor element Aa exhibiting secondary LPF characteristics is controlled by a feedback control circuit B. The feedback control circuit B includes amplifiers Ba, Bb, Bc, integrators Bd, Be, an adder (Summation) Bf, a phase lead compensator Bg, and a quantizer Bh. With such a configuration, the conventional angular velocity sensor system detects the displacement x of the weight when the force Fin as the input is applied to the weight by the detection electrode and the detection circuit, and the force ( -Fin) is applied as a voltage from the feedback electrode and balanced with the input Fin, and this balanced force Fin, that is, the applied voltage is used as the detection output.

  As a result, the characteristics such as the sensitivity of the angular velocity sensor can be determined by the characteristics of the feedback electrode (electrostatic force-applied voltage characteristics), and the influence of the mechanical characteristics of the weight (spring constant, damping coefficient, sensitivity nonlinearity, etc.) Can be suppressed. In general, the characteristic of the feedback electrode is more stable than the mechanical characteristic of the weight, so that the detection characteristic of the angular velocity sensor can be stabilized as a result of force feedback. Specifically, sensitivity fluctuation and drift can be suppressed, and the frequency band (Bandwidth), input range (Dynamic Range), and linearity can be improved (see Non-Patent Document 2, page 372, line 8).

  By the way, when the input-output characteristic shows a second-order LPF characteristic like the MEMS weight in the angular velocity sensor, the phase delay of 90 degrees in phase is higher than the cutoff frequency when the vibration frequency is equal to the cutoff frequency. In the case of the frequency, a phase delay of 90 degrees to 180 degrees occurs. When attempting to perform negative feedback control (Negative Feedback Control) of such a controlled device, the entire control loop transfer characteristic including the controlled device and the feedback control circuit exceeds 180 degrees by exceeding a certain frequency. If the phase is delayed and the loop gain at this time is 1 or more, the originally intended negative feedback control becomes positive feedback control. As a result, the controlled device may be in an abnormal oscillation state and the control system may become unstable. Such a concern is common to analog feedback circuits and digital feedback circuits.

  Although the said nonpatent literature 1 has proposed the distributed feedback type | mold (Distributed feedback architecture) and the feedforward addition type | mold (Architecture with feedforward summation) as a structure for implement | achieving the digital feedback system by delta-sigma modulation, Even in the configuration, since a signal path cannot be provided in the sensor element portion, it is described that a phase lead compensator is necessary to compensate for this and stabilize oscillation. FIG. 1 shows a feedforward addition type configuration example.

  Non-Patent Document 3 describes a distributed feedback type acceleration sensor. In this document, a phase lead compensator is used for the feedback output.

  Non-Patent Document 4 describes the relationship between phase delay and vibration frequency in a feedback control system having a sensor with a high Q value.

Petkov and Boser, “A Fourth-Order ΣΔ Interface for Micromachined Inertial Sensors”, IEEE Journal of Solid-State Circuits, Vol.40, No.8, Aug 2005. Boser and Howe, “Surface micromachined accelerometers”, IEEE Journal of Solid-State Circuits, Vol. 31, No. 3, March 1996. Kajita, Moon and Temes, “A two-chip interface for a MEMS accelerometer”, IEEE Transactions on Instrumentation and Measurement, Vol.51, No.4, August 2002. Ezekwe and Boser, “Robust compensation of a force-balanced high-Q gyroscope”, IEEE Sensors 2008 Conference.

  However, when a phase lead compensator is used as in the above non-patent document, the circuit scale is increased thereby, which may not be suitable for mounting on a mobile body such as an automobile or a mobile phone.

  In addition, since the phase lead compensator is connected in series to the signal transmission path, the signal delay from the input sampling to the feedback output increases due to the circuit delay of the phase lead compensator itself, which stabilizes the oscillation of the feedback system. Sex margin is reduced.

  In particular, in the distributed feedback type, since the input signal passes through one or more integrators and then a feedback output is made via a comparator or the like, a signal transmission delay corresponding to the settling time of the integrator is inevitably required. Arise. Assuming that one clock period, that is, one sampling period Ts (= 1 / fs; fs is a sampling frequency) is assigned to the settling time, the delay of the one sampling period Ts is a phase angle as the oscillation frequency increases. A signal delay of 10 degrees or more is caused. For example, a delay of one sampling period Ts causes a non-negligible phase delay of 45 degrees at the vibration frequency f = fs / 8 and 90 degrees at the f = fs / 4. This phase delay reduces the oscillation stability margin of the negative feedback control system.

  On the other hand, in the feed-forward addition type, the sampled and amplified signal is bypass-transmitted via a feed-forward path (in FIG. 1, a path via amplifiers Bb and Bc), and this is added to the signal output via the integrator. Thereafter, the signal is output through a phase lead compensator and a comparator. In the feedforward addition type, the signal transmission delay can be made smaller than that in the distributed feedback type because the signal is transmitted without going through the integrator, but the signal transmission delay due to the presence of the phase advance compensator still exists.

  In addition, in order to reduce the signal transmission delay as described above, the first stage sampling / signal amplifying circuit, the addition operation operational amplifier, the phase advance compensation operational amplifier, etc. need high speed and large output load drive capability, which is costly and consuming. Electric power may increase.

  For example, in the conventional configuration shown in FIG. 1, the adder Bf and the phase lead compensator Bg are connected in series. Therefore, in order to ensure a certain accuracy, an operational amplifier is connected to one or both of these arithmetic units. Must be used. This is because when only a voltage dividing circuit using a passive element such as a capacitor is connected in series without using an operational amplifier, an error due to a parasitic capacitance such as a capacitor or wiring increases, and high-accuracy calculation is generally difficult. On the other hand, when an operational amplifier is used, the output signal is output after all of the circuits including the operational amplifier have completed settling, so that the signal transmission delay is inevitably increased. In order to drive the circuit configuration at a time to increase the speed, the first stage sampling and signal amplification circuit, the phase lead compensator or the operational amplifier which is an adder requires a high speed and a large output load driving capability.

  Note that suppression of signal transmission delay is particularly important when the non-control device is a resonant element having a high Q value. A resonance element that exhibits a high Q value by vacuum sealing exhibits a high peak gain in the vicinity of the resonance frequency f0 in amplitude characteristics. On the other hand, the phase characteristics of a resonant element exhibiting a high Q value are such that the phase delay angle is suddenly 0 degrees (f <f0) → 90 degrees (f = f0) → 180 degrees (f> f0) around the resonant frequency f0. Change. As a result, since the phase lag angle is 180 degrees in the region where the vibration frequency f is slightly higher than the resonance frequency f0, oscillation is likely to occur in the region near the resonance frequency f0. In order to avoid this, it is necessary to ensure a sufficient amount of phase advance on the feedback control circuit side.

  Furthermore, a resonance element that exhibits a high Q value by vacuum sealing may exhibit a plurality of gain peaks due to higher-order resonance in a frequency region higher than the main resonance frequency f0 (see Non-Patent Document 4 above). In order to stably perform feedback control of such a resonant element, not only the resonant frequency f0 but also a frequency that causes higher-order resonance and further a wide frequency range up to a frequency range where the transmission specification of the resonant element is sufficiently attenuated. It is necessary to ensure a sufficient phase margin in the entire control loop in which the resonance element and the feedback control circuit are combined so that the phase advance characteristic can be exhibited over the entire period.

  Accordingly, it is necessary to set the signal transmission delay between the input and output sufficiently small and set the sampling frequency fs sufficiently high. However, in the configuration using the phase lead compensator, as described above, the circuit scale and cost are reduced. As a result, problems such as power consumption occur, making it difficult to design a fast and stable feedback control circuit.

  The present invention has been made to solve the above-mentioned problems, and a feedback control circuit capable of suppressing a signal transmission delay while reducing circuit scale, cost, and power consumption, and feedback including a part thereof. The main purpose is to provide a control system.

In order to achieve the above object, the first aspect of the present invention provides:
A feedback control circuit that generates an output signal based on an input signal input from a controlled device and performs feedback control by negatively feeding back the output signal to the controlled device;
Amplifying means for amplifying the input signal;
Three or more stages of integration means for performing an integration operation on the output signal of the amplification means;
One or more feed-forward paths for transmitting signals bypassing at least a portion of the integrating means;
Adding means for adding a signal input from the integrating means and a signal input via the feed-feedforward path;
Quantization means for quantizing the output of the adding means to generate an output signal;
A first feedback path for transmitting an output signal generated by the quantizing means to an input side of a last-stage integrating means among the three or more integrating means;
A second feedback path for transmitting the output of the last stage integrating means among the three or more stages of integrating means to the input side of the second and subsequent stages of integrating means of the three or more stages;
Is a feedback control circuit.

  According to the first aspect of the present invention, signal transmission delay can be suppressed while reducing circuit scale, cost, and power consumption.

The second aspect of the present invention is:
A feedback control circuit according to the first aspect of the present invention;
The controlled device that is feedback controlled by the feedback control circuit;
Is a feedback control system.

According to the second aspect of the present invention, signal transmission delay can be suppressed while reducing circuit scale, cost, and power consumption.

In addition, among the three or more stages of integration means, the DC (frequency 0 [Hz]) is obtained by the first stage integration means not included in the feedback path that transmits the output of the last stage integration means to the input side of the second and subsequent integration means. ) The zero point of the noise transfer function can be arranged in the vicinity, and the zero point of the noise transfer function can be arranged in a higher frequency band by the integration means and the food back path after the second stage. Therefore, the detection resolution can be improved in both the frequency band near DC and the higher frequency band.

In a second aspect of the invention,
The controlled device is a device having, for example, a second-order low-pass filter characteristic.

in this case,
The controlled device is, for example, an inertial force sensor using electrostatic force feedback of a MEMS weight.

The inertial force sensor may be a sensor that detects both the yaw rate and the acceleration.

The third aspect of the present invention is:
A controlled device having secondary low-pass filter characteristics;
A feedback control circuit for generating an output signal based on an input signal input from the controlled device and performing feedback control by negatively feeding back the output signal to the controlled device;
A feedback control system comprising:
The feedback control circuit includes:
Amplifying means for amplifying the input signal;
Two-stage integration means for performing an integration operation on the output signal of the amplification means;
A first feedforward path having a first amplifying unit that bypasses both of the two-stage integrating means and amplifies the signal at a first amplification factor;
A second feedforward path having a second amplification section that bypasses only the second-stage integration means of the two-stage integration means and amplifies the signal at a second amplification factor;
Adding means for adding the signal input from the integrating means and the signals input via the first and second feedforward paths;
Quantization means for quantizing the output of the adding means to generate an output signal;
A feedback path for transmitting the output signal generated by the quantization means to the input side of the last-stage integration means of the two-stage integration means, and
A value obtained by dividing the sampling frequency of the feedback control circuit by a value obtained by multiplying the square root of the difference between the first amplification factor and the second amplification factor by 2π is a cut in the secondary low-pass filter characteristic of the controlled device. It is not more than 2.5 times the off frequency,
Feedback control system.

  According to this third aspect of the present invention, the phase delay of the transfer function of the control loop in the entire feedback control system including the controlled device and the feedback control circuit can be suppressed to less than 180 degrees. As a result, the phase lead compensator is not necessary, and the signal transmission delay can be suppressed while reducing the circuit scale, cost, and power consumption.

  ADVANTAGE OF THE INVENTION According to this invention, the feedback control circuit which can suppress a signal transmission delay, reducing a circuit scale, cost, and power consumption, and the feedback control system which includes this in part can be provided.

It is a structural example of the conventional angular velocity sensor system controlled by force feedback. It is a functional block diagram which shows the concept common to each Example of this invention. It is a functional block diagram showing the feedback control system 1 concerning the 1st example of the present invention by discrete time expression. It is a functional block diagram of the feedback control circuit 10 according to the first embodiment. It is another example of the functional block diagram of the feedback control circuit 10 which concerns on 1st Example. It is the functional block diagram which represented the feedback control system 2 which concerns on 2nd Example of this invention by discrete time expression. It is the functional block diagram which represented the feedback control system 3 which concerns on 3rd Example of this invention by discrete time expression. It is a figure which shows the feedback control system which concerns on the other Example of this invention. It is a structural example at the time of applying this invention to a servo mechanism.

  DESCRIPTION OF EMBODIMENTS Hereinafter, embodiments for carrying out the present invention will be described with reference to the accompanying drawings.

<Basic concept>
Embodiments of the present invention will be described below with reference to the drawings. First, the basic concept common to the first to third embodiments will be described. FIG. 2 is a functional block diagram showing a concept common to the embodiments of the present invention. The details of each embodiment will be described in discrete time representation with reference to FIG.

  The feedback control system according to the present invention includes a controlled device 5 to be controlled and a feedback control circuit 10.

  The controlled device 5 is, for example, a MEMS weight having a secondary LPF characteristic, an LC filter, a servo motor, or the like (in the drawing, simply indicated as “LPF”). In the following embodiments, it is assumed that the MEMS weight is driven by electrostatic force, and the angular velocity sensor detects the angular velocity by detecting the Coriolis force. The controlled device 5 is applied with electrostatic force Fin and force feedback force (−Ffb) as inputs such as Coriolis force and inertial force, and outputs displacement x.

  The feedback control circuit 10 according to the present invention includes an amplifier that amplifies an input signal Vx, one or more integrators that integrate the output of the amplifier, and the output of the final stage of the one or more integrators and the integrator of the final stage. There is an adder that adds the outputs of the feedforward paths that are not, and an output circuit. The output of the output circuit is output to the outside and negatively fed back to the controlled device 5 and further fed back to the input side of the final stage integrator.

  This will be described more specifically. The feedback control circuit 10 includes amplifiers 12 (amplification factor a0), 14 (amplification factor a12), 16 (amplification factor a2), 18 (amplification factor b1), adders 20 and 22, an integrator 30, and an output circuit. 32 and a computing unit 34.

  Note that the calculator 34 is an integrator, an amplifier, or the like, and may not exist (H1 = 1). Further, the feedforward path FF2 passing through the amplifier 16 can be omitted. Further, the feedback path FB2 passing through the amplifier 18 exists, but the amplifier 18 does not exist (b1 = 1). In this <basic concept>, description will be made assuming that the amplifier 18 exists but the arithmetic unit 34 does not exist, and the feedforward path passing through the amplifier 16 is omitted.

  In the feedback control circuit 10, first, an input signal Vx input as a minute signal is amplified by the amplifier 12 and output to the arithmetic unit 34 and the amplifier 14. The signals a 0 and Vx output to the calculator 34 are output to the adder 22 via the integrator 30 after the output b 1 and vfb of the amplifier 18 are subtracted by the adder 20.

  The output a0 · Vx of the amplifier 12 is output to the adder 22 via the amplifier 14 (a0 · a12 · Vx; feedforward path FF1). The adder 22 adds the signals input from the integrator 30 and the amplifier 14 and outputs the result to the output circuit 32. The output circuit 32 is a quantizer typified by a comparator (comparator), for example, and generates and outputs a feedback output Vfb.

  The feedback output Vfb is output to the outside as the output of the feedback control system 1, that is, the angular velocity sensor system, or is V / F converted and negatively fed back to the controlled device 5 (feedback path FB1). Thereby, characteristics such as sensitivity of the angular velocity sensor can be determined by the characteristics of the feedback electrode, and the influence of the mechanical characteristics of the weight can be suppressed. In general, the characteristic of the feedback electrode is more stable than the mechanical characteristic of the weight, so that the detection characteristic of the angular velocity sensor can be stabilized as a result of force feedback.

  The feedback output Vfb is fed back to the adder 20 via the amplifier 18 (feedback path FB2). That is, the feedback output Vfb of the output circuit 32 that is a quantizer typified by a comparator (comparator) is fed back to the input side of the integrator 30 that is the final stage integrator. As a result, as will be described later, it is possible to suppress signal transmission delay without providing a phase advance compensator.

<First embodiment>
Hereinafter, a feedback control circuit 10 according to a first embodiment of the present invention and a feedback control system 1 including a part thereof will be described with reference to the drawings.

  FIG. 3 is a functional block diagram showing the feedback control system 1 according to the first embodiment of the present invention in discrete time representation. In the figure, characters in the functional block frame indicate a transfer function or the like in Z conversion. Moreover, the same code | symbol is attached | subjected about the component which is common in FIG.

  The amplifier 12 receives an input signal Vx sampled at the sampling period Ts. A feedback output Vy (= Vfb) is input to the amplifier 18 via the delay element 36. Note that the delay element provided by the delay element 36 may be included in the integrator 30, as will be described later with reference to FIG.

  In this embodiment, it is assumed that there is a feedforward path FF2 passing through the amplifier 16. That is, the signal output from the calculator 34 is output to the adder 22 via the amplifier 16 (feed forward addition; a0 · a2 · H1 · Vx). Therefore, two feedforward paths exist together with the feedforward path FF1 passing through the amplifier 14.

  Regarding the other components, the description is omitted by referring to the description of FIG.

[Transfer function characteristics]
Hereinafter, the transfer function and characteristics of the feedback control circuit 10 will be considered. FIG. 4 is a functional block diagram of the feedback control circuit 10 according to the first embodiment. Here, the quantization error due to the output circuit 32 is e, the Z conversion is E, the input of the feedback control circuit 10 is Vx, and the output is Vy. From FIG. 3, the following equation (2) is established.

[Conventional configuration]
First, consider a case where there is no feedback path FB2 passing through the amplifier 18 (that is, when b1 = 0). In this case, the following expression (3) is established. Here, if H1 = z ⁻¹ / (1−z ⁻¹ ), this is the same as the conventional configuration shown in FIG. 1 with the phase lead compensator (1−α · z ⁻¹ ) omitted. In the following equation (3), STF (Signal Transfer Function) is a transfer function for the input signal Vx, and NTF (Noise Transfer Function) is a transfer function for the quantization noise E.

  The signal transfer function Hxy0 between the input Vx and the output Vy corresponds to this STF and is expressed by the following equation (4).

  Here, in order to evaluate the stability of the feedback system in this case, the frequency characteristic of the feedback control circuit in which the transfer function is expressed by the above equation (4) is obtained. Such evaluation can be performed by substituting the expression represented by the following expression (5) into the variable z of the Z conversion. In the equation, j is an imaginary unit, f is a frequency, ω = 2πf is an angular frequency, T is a sampling period, and fs = 1 / T is a sampling frequency. Further, it is assumed that a0, a12, and a2 are constants and have no frequency characteristics or phase delay.

The integral term {1 / (1−z ⁻¹ )} in parentheses in the above equation (5) has a phase lag characteristic in which the phase lag is about 90 degrees at a low frequency (z≈1). Hxy0 has a phase delay characteristic if H1 does not have a phase advance characteristic (that is, if it does not have a differentiation circuit characteristic). In the conventional configuration shown in FIG. 1, since H1 is a delay integrator {z ⁻¹ / (1−z ⁻¹ )}, it has a phase delay characteristic instead of a phase advance characteristic. For this reason, when the feedback control circuit having the conventional configuration shown in FIG. 1 is used to perform feedback control of the controlled device exhibiting the second-order LPF characteristic, the phase delay becomes 90 degrees to 180 degrees when the frequency f exceeds the cutoff frequency f0. If a phase delay of the feedback control circuit is added to this, the entire control loop has a phase delay characteristic exceeding 180 degrees. Therefore, if the phase advance compensator is not provided, the controlled device may oscillate abnormally and control may become unstable.

[In the case of this example]
Next, consider the case where the feedback path FB2 passing through the amplifier 18 exists (b1 ≠ 0) as in this embodiment. Here, b1 = 1 is set for simplification.

  In the above equation (2), when b1 = 1, the following equations (6) and (7) are established.

A signal transfer function Hxy1 between the input Vx and the output Vy corresponds to STF and is expressed by the following equation (8). NTF is (1−z ⁻¹ ).

Comparing equation (8) with equation (4) of the conventional configuration, equation (4) includes an integral term {1 / (1−z ⁻¹ )}, whereas in equation (8), H1 The integral term disappears except inside. Furthermore, in the formula (8), the differential term a12 · (1−z ⁻¹ ) appears in the first term in parentheses. This is because the signal based on the input signal Vx is added to the input of the integrator 30 from the output circuit 32 and the feedforward path FF1 where the signal is added to the output of the integrator 30 as the final stage integrator without passing through the integrator. This shows that a phase lead term appears in the signal transfer function Hxy1 by the feedback path FB2 fed back.

  Furthermore, even when H1 includes an integrator and has a phase delay characteristic, the signal transfer function Hxy1 can have a phase advance characteristic in a certain frequency range depending on the setting of a12, a2, and H1.

For example, as in the conventional configuration shown in FIG. 1, assuming that H1 = z ⁻¹ / (1−z ⁻¹ ), the signal transfer function Hxy1 is expressed by the following equation (9).

The first term (a12−a2) · (1−z ⁻¹ ) in parentheses in the formula (9) is a differential term and has a phase advance characteristic, but the third term z ⁻¹ / (1−z ⁻¹ ) Is an integral term showing a large gain at a low frequency and has a phase delay characteristic. Here, in the low frequency region of ωT << 1, that is, f << (fs / 2π), the differential (1−z ⁻¹ ) is j2πf / fs and the integral z ⁻¹ / (1−z ⁻¹ ) is -J / (2πf / fs) can be approximated respectively. The imaginary unit j in the differentiation represents a phase advance of 90 degrees, and the negative imaginary unit -j in the integration represents a phase delay of 90 degrees.

  Accordingly, when the amplification factors of the amplifiers 14 and 16 are set so that a12> a2, the frequency at which (a12−a2) · 2πf / fs <(2πf / fs), that is, f <fs / {2π√ (a12− a2)} has a phase lag characteristic because the integral term (third term) is the dominant term, and in the high frequency region where f> fs / {2π√ (a12−a2)} Since the term (first term) becomes the dominant term, it has phase lead characteristics.

  Here, when the input-output characteristic of the controlled device 5 exhibits the second-order LPF characteristic as described above, the phase delay of 90 degrees is higher than the cutoff frequency f0 when the frequency f is equal to the cutoff frequency f0. Then, the phase delay is 90 to 180 degrees. Therefore, if the transfer function characteristic of the feedback control circuit 10 is changed to a phase advance at a frequency lower than the frequency at which the phase delay of the controlled device approaches 180 degrees, the controlled device 5 and the feedback control circuit 10 are combined. In addition, the phase delay of the transfer function of the control loop in the entire feedback control system 1 can be suppressed to less than 180 degrees.

  For example, in order to set the transfer function characteristic of the feedback control circuit 10 to transition to phase advance at 2.5 times or less of the cut-off frequency f0 of the controlled device 5, the amplification of the amplifiers 14 and 16 is expressed by the following equation (10). What is necessary is just to set a rate.

  Thus, in the feedback control circuit 10 of the present embodiment, the feedforward path FF1 in which the signal based on the input signal Vx is added to the output of the integrator 30 as the final stage integrator without passing through the integrator, Since the feedback path FB2 fed back from the output circuit 32 to the input side of the integrator 30 is provided, the transfer function can have a phase advance characteristic.

  For this reason, even if the controlled device 5 exhibits the second-order LPF characteristics and exhibits a phase delay of 90 to 180 degrees in a high frequency region higher than the cutoff frequency f0, it is an essential structure in the conventional structure. The phase delay of the transfer function in the control loop of the entire feedback control system 1 can be suppressed to less than 180 degrees without providing the phase lead compensator. As a result, a stable feedback control system 1 can be realized.

  As described above, in the configuration including the phase lead compensator, the circuit scale increases accordingly. In addition, since the phase lead compensator is connected in series to the signal transmission path, it is necessary to consider the circuit delay of the phase lead compensator itself, and the first stage sampling / signal amplification circuit and the addition operation operational amplifier, A phase lead compensation operational amplifier or the like requires high-speed and large output load driving capability, which may increase cost and power consumption.

  On the other hand, in the feedback control circuit 10 of the present embodiment, it is not necessary to provide a phase lead compensator with the above-described characteristic configuration. As a result, the circuit scale, cost, and power consumption can be reduced. Further, after the input signal is sampled, it is possible to perform a comparison output operation by performing an addition operation of a feedforward signal or the like only with a passive element such as a capacitor voltage division without using an operational amplifier or an integration operation. -The delay time between outputs can be shortened, and the stability of control can be improved.

  According to the feedback control circuit 10 of the present embodiment described above and the feedback control system 1 including this part, the signal transmission delay can be suppressed while reducing the circuit scale, cost, and power consumption.

  In the digital feedback system expressed in discrete time, the quantization noise E is added to the output Vy, so strictly speaking, not only the transfer function STF for the signal but also the influence of the quantization noise is included in the stabilization determination. Must be considered. However, when the controlled device 5 exhibits the secondary LPF characteristic, if the sampling frequency fs is set sufficiently higher than the cut-off frequency f0 in the secondary LPF characteristic, the system stability of the quantization noise can be improved. The effect on sex can be reduced. The reason will be described below.

As described above, the transfer function NTF for the quantization noise E is represented by (1−z ⁻¹ ). This indicates that the first-order noise shape is applied to the quantization noise E and transmitted to the output Vy, and the noise is low in the low frequency region and the noise is high in the high frequency region. On the other hand, when the controlled device 5 has a secondary LPF characteristic, a component having a frequency higher than the cutoff frequency f0 is attenuated by the controlled device 5 and transmitted to the feedback control circuit 10 as a loop. Therefore, if the sampling frequency fs is set sufficiently higher than the cut-off frequency f0 in the second-order LPF characteristic (for example, about fs / f0> 200), the low-frequency component of the quantization noise depends on the noise shape. Attenuating and the high frequency component is attenuated by the secondary LPF component of the controlled device 5, and therefore, it is considered that the influence on the system stability is small.

  The applicant of the present application sets the cutoff frequency f0 of the controlled device 5 to 10 [kHz], varies the Q value within a certain range, and sets the sampling frequency of the feedback control circuit 10 to fs = 2.56 [ MHz], the amplification factor of the feedback control circuit 10 is set as a0 = 1, a12 = 256, a2 = 16, b1 = 1, and the input Fin of the controlled device 5 is 85% of the maximum output range of the force feedback output Ffb. The simulation is performed under the condition that the range is changed (that is, the range of the input Fin is changed within a range of ± 0.85 when the output range of the force feedback output Ffb is ± 1). It has been confirmed that stable feedback control can be performed without abnormal oscillation under such conditions.

[Modification]
The feedback control circuit 10 according to the first embodiment can be modified as follows. FIG. 5 is another example of a functional block diagram of the feedback control circuit 10 according to the first embodiment.

As shown in the figure, the feedback control circuit 10 according to the first embodiment changes the integrator 30 to an integrator with a delay element (z ⁻¹ / (1−z ⁻¹ )) 30 _ ^{1, and} replaces the calculator 34 with an integrator. (1 / (1−z ⁻¹ )), the feed forward path FF2 passing through the amplifier 16 bypassing the integrator 30 is fed to bypass the calculator 34 via the amplifier 38 (amplification factor a1). It can be changed to the forward path FF2_1.

  That is, in the feedback control circuit 10 shown in FIG. 5, the input signal Vx is amplified by the amplifier 12 and output to the computing unit 34, the amplifier 14, and the amplifier 38. The signal output to the computing unit 34 is integrated by the computing unit 34, and the adder 20 subtracts the outputs b1 and vfb of the amplifier 18 and adds the outputs a0, a1 and Vx of the amplifier 38, thereby integrating the integrator. It is output to the adder 22 via 30_1.

  The output a0 · Vx of the amplifier 12 is output to the adder 22 via the amplifier 14 (a0 · a12 · Vx). The adder 22 adds the signals input from the integrator 30 </ b> _ <b> 1 and the amplifier 14 and outputs the result to the output circuit 32. The amplifier 18 receives the feedback output Vy.

  In such an aspect, the delay element 36 in the feedback path via the amplifier 18 is not necessary. When the integrator of the feedback control circuit 10 is second-order or higher, a local resonance feedback (γ) may be provided to form a bandpass filter.

<Second embodiment>
Hereinafter, a feedback control circuit 10 according to a second embodiment of the present invention and a feedback control system 2 including a part thereof will be described with reference to the drawings.

  FIG. 6 is a functional block diagram showing the feedback control system 2 according to the second embodiment of the present invention in discrete time expression. In the figure, characters in the functional block frame indicate a transfer function or the like in Z conversion. Moreover, the same code | symbol is attached | subjected about the component which is common in 1st Example.

As shown in the figure, the feedback control circuit according to the present embodiment is composed of a primary integrator, and includes an arithmetic unit 34_1 that is an amplifier (amplification factor a1) and an integrator (z ⁻¹ / (1−z) with a delay element. ^-1 )) 30_1, compared to the first embodiment, the feedforward path FF2 (or FF2_1) that bypasses the calculator 34 and the integrator 30_1 with a delay element is omitted.

  That is, in the feedback control circuit 10 according to the present embodiment, the input signal Vx is amplified by the amplifier 12 and is output to the arithmetic unit 34_1 and the amplifier 14. The adder 20 subtracts the output b1 · vfb of the amplifier 18 from the signal output to the calculator 34_1 and outputs the signal to the adder 22 via the integrator 30_1.

  The output a0 · Vx of the amplifier 12 is a0 · a12 · Vx output to the adder 22 via the amplifier 14. The adder 22 adds the signals input from the integrator 30 </ b> _ <b> 1 and the amplifier 14 and outputs the result to the output circuit 32. The amplifier 18 receives the feedback output Vy and outputs the amplified signal to the adder 20 (feedback path FB2).

  The feedback path FB1 is the same as in the first embodiment.

[Transfer function characteristics]
The signal transfer function Hxy2 between the input Vx and the output Vy in this embodiment is expressed by the following equation (11) by setting H1 = a1 · z ⁻¹ and a2 = 0 in the above equation (4).

In the above equation (11), if (a12−a1) · 2πf0 / fs >> a1, that is, a1 << (2f0 / fs) · a12, the first term in parentheses (a12−a1) · ( 1−z ⁻¹ ) has a differential characteristic and is a phase advance. Since the second term a2 is a constant term, an integral term that causes a phase delay does not appear. If a1 = 0, a differential characteristic having a phase advance of 90 degrees in the low frequency region can be obtained.

  Therefore, the signal transfer function Hxy2 of the feedback control circuit 10 according to the present embodiment has a phase advance characteristic over a wide range of frequencies from a low frequency to a high frequency. This eliminates the need for a phase advance compensator, thereby reducing circuit scale, cost, and power consumption. Further, after the input signal is sampled, it is possible to perform a comparison output operation by performing an addition operation of a feedforward signal or the like only with a passive element such as a capacitor voltage division without using an operational amplifier or an integration operation. -The delay time between outputs can be shortened, and the stability of control can be improved.

  In practice, it is desirable to set the gain in the low frequency region (z≈1) that is the signal band, that is, a0 · a1 sufficiently high in order to make the feedback control work effectively.

  According to the feedback control circuit 10 of the present embodiment described above and the feedback control system 2 including a part thereof, the signal transmission delay can be suppressed while reducing the circuit scale, cost, and power consumption.

<Third embodiment>
Hereinafter, a feedback control circuit 10 according to a third embodiment of the present invention and a feedback control system 3 including a part thereof will be described with reference to the drawings.

  FIG. 7 is a functional block diagram showing the feedback control system 3 according to the third embodiment of the present invention in discrete time expression. In the figure, characters in the functional block frame indicate a transfer function or the like in Z conversion. Moreover, the same code | symbol is attached | subjected about the component which is common in 1st Example.

  As shown in the figure, the feedback control circuit according to this embodiment includes a third-order integrator, and has a local feedback path FB3 that feeds back from the last-stage integrator to the previous-stage integrator.

  The feedback control circuit 10 according to this embodiment includes amplifiers 12 (amplification factor a0), 18 (amplification factor b1), 40 (amplification factor a13), 42 (amplification factor a23), 44 (amplification factor a3), and 70 (amplification factor). Rate γ), integrators 50, 52, and 54, adders 60, 62, and 64, an output circuit 32, and delay elements 36 and 72.

  In the feedback control circuit 10 according to the present embodiment, the input signal Vx is amplified by the amplifier 12 and output to the integrator 50 with a delay element and the amplifier 40. The output of the amplifier 70 is subtracted by the adder 60 from the signal integrated by the integrator 50 and output to the integrator 52. The integrator 52 outputs the signal obtained by the integration calculation to the adder 62 and the amplifier 44. The adder 62 subtracts the output of the amplifier 18 from the output of the integrator 52 and outputs the result to the integrator 54.

  The signal integrated by the integrator 54 is output to the adder 64. The adder 64 receives a feedforward signal that is input via a feedforward path FF1 that passes through the amplifier 40, a feedforward path FF2 that passes through the amplifier 42, and a feedforward path FF3 that passes through the amplifier 44. Is added to the output of the integrator 54 and output to the output circuit 32. The output of the output circuit 32 is fed back to the amplifier 18 via the delay element 36, and the amplified signal is output to the adder 62 (feedback path FB2). The amplifier 70 is fed back with the output of the integrator 54 via the delay element 72, and outputs the amplified signal to the adder 60 (feedback path FB3).

  The feedback path FB1 is the same as in the first embodiment.

Here, a conventional technique corresponding to the basic technique of the third embodiment will be described. Non-Patent Document 1 described above proposes a feedback control circuit configured by a secondary integrator and having a local feedback path via an amplifier having an amplification factor γ. This local feedback path creates a zero point (NTFzero, also called Notch) at a specific frequency fr = (√γ / 2π) · fs of the quantization noise frequency distribution, and the signal detection resolution at this frequency fr It is known that can be improved. For example, the yaw rate sensor includes a vibration element in which the sensor element vibrates at a specific frequency fosc, and the vibration component of the output signal is concentrated on the frequency fosc. Therefore, by setting γ = (ωosc · T) ² = (2π · fosc / fs) ² , a high-resolution yaw rate signal can be obtained by extracting only signal components in the frequency band near the signal band fosc.

  However, when a secondary integrator is used and a notch is arranged at the specific frequency fr by local feedback, the quantization noise in the low frequency region near DC (frequency 0 [Hz]) increases conversely. Therefore, it cannot be suitably applied to a sensor that detects both yaw rate and acceleration. This is because in a sensor that detects both the yaw rate and the acceleration, the yaw rate signal is in the vicinity of the vibration frequency fosc of the sensor element, but the acceleration signal is in the low frequency region near DC. Therefore, when local feedback is added to the secondary integrator, there is a drawback that it is impossible to accurately detect both the yaw rate signal and the acceleration signal at the same time.

On the other hand, in the feedback control circuit 10 of the present embodiment, the zero point (NTFzero) of the noise transfer function is arranged in DC by the first-stage integrator 50 that does not include local feedback, and the second and subsequent stages of local feedback are provided. With FB2, the zero point (NTFzero) of the noise transfer function can be arranged at the frequency fr = (√γ / 2π) · fs. In order to realize this, γ = (ωosc · Ts) ² = (2π · fosc / fs) ² may be set. Thereby, a plurality of zero points can be set, quantization noise in both frequency bands near DC and fr can be suppressed, and detection resolution can be improved. Therefore, it is possible to realize a highly accurate multi-function sensor that detects both yaw rate and acceleration.

  According to the feedback control circuit 10 of the present embodiment described above and the feedback control system 3 including a part thereof, the phase lead compensator is not necessary as in the first and second embodiments, so that the circuit scale is increased. In addition, signal transmission delay can be suppressed while reducing cost and power consumption.

<Deformation, application example>
Further, since the zero point of the noise transfer function can be arranged in a plurality of frequency bands, for example, a highly accurate sensor that detects both the yaw rate and the acceleration can be realized.

  The best mode for carrying out the present invention has been described above with reference to the embodiments. However, the present invention is not limited to these embodiments, and various modifications can be made without departing from the scope of the present invention. And substitutions can be added.

  For example, in each embodiment, processing such as computation is performed on an analog signal in the feedback control circuit 10, but the A / D converter 80 is input to the feedback control circuit 10 as shown in FIG. It is good also as what arrange | positions in the side and processes a calculation etc. with respect to a digital signal. Such a modification can be made for each of the above embodiments. FIG. 8 is a diagram illustrating a feedback control system according to another embodiment of the present invention. The A / D converter 80 encodes (numerizes) the analog signal that has been sampled and amplified by the amplifier 12, and outputs the encoded analog signal to the subsequent components. As a result, the constituent elements in the subsequent stages can execute the processes as in the above embodiments by digital calculation.

  Further, the present invention can be applied to a servo mechanism that controls the controlled device 5 so as to follow the target value. FIG. 9 shows a configuration example when the present invention is applied to a servo mechanism. In this case, as shown in the figure, the target value is negatively input as a digital value, and the difference from the value obtained by converting the output signal of the controlled device 5 into a digital value by the A / D converter 80 is determined by the feedback control circuit 10 Process by force feedback loop. Thus, so-called servo control is realized. In this configuration, the target value can be input as a digital value by the digital signal conversion by the A / D converter 80. Further, the target value can be set as an offset adjustment value, and the offset adjustment can be easily realized by digital addition.

1, 2, 3 Feedback control system 5 Controlled device 10 Feedback control circuit 12, 14, 16, 18, ... Amplifier 20, 22, ... Adder 30, ... Integrator 32 Output circuit 34 Calculator 80 A / D converter

Claims (6)

A feedback control circuit that generates an output signal based on an input signal input from a controlled device and performs feedback control by negatively feeding back the output signal to the controlled device;
Amplifying means for amplifying the input signal;
Three or more stages of integration means for performing an integration operation on the output signal of the amplification means;
One or more feed-forward paths for transmitting signals bypassing at least a portion of the integrating means;
Adding means for adding a signal input from the integrating means and a signal input via the feed-feedforward path;
Quantization means for quantizing the output of the adding means to generate an output signal;
A first feedback path for transmitting an output signal generated by the quantizing means to an input side of a last-stage integrating means among the three or more integrating means;
A second feedback path for transmitting the output of the last stage integrating means among the three or more stages of integrating means to the input side of the second and subsequent stages of integrating means of the three or more stages;
A feedback control circuit comprising: A feedback control circuit according to claim 1 ;
The controlled device that is feedback controlled by the feedback control circuit;
A feedback control system comprising: The feedback control system according to claim 2 ,
The controlled device is a device having a secondary low-pass filter characteristic.
Feedback control system. A feedback control system according to claim 3 ,
The controlled device is an inertial force sensor using electrostatic force feedback of a MEMS weight.
Feedback control system. The feedback control system according to claim 4,
The inertial force sensor is a sensor that detects both yaw rate and acceleration.
Feedback control system. A controlled device having secondary low-pass filter characteristics;
A feedback control circuit for generating an output signal based on an input signal input from the controlled device and performing feedback control by negatively feeding back the output signal to the controlled device;
A feedback control system comprising:
The feedback control circuit includes:
Amplifying means for amplifying the input signal;
Two-stage integration means for performing an integration operation on the output signal of the amplification means;
A first feedforward path having a first amplifying unit that bypasses both of the two-stage integrating means and amplifies the signal at a first amplification factor;
A second feedforward path having a second amplification section that bypasses only the second-stage integration means of the two-stage integration means and amplifies the signal at a second amplification factor;
Adding means for adding the signal input from the integrating means and the signals input via the first and second feedforward paths;
Quantization means for quantizing the output of the adding means to generate an output signal;
A feedback path for transmitting the output signal generated by the quantization means to the input side of the last-stage integration means of the two-stage integration means, and
A value obtained by dividing the sampling frequency of the feedback control circuit by a value obtained by multiplying the square root of the difference between the first amplification factor and the second amplification factor by 2π is a cut in the secondary low-pass filter characteristic of the controlled device. It is not more than 2.5 times the off frequency,
Feedback control system.

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