JP5554178B2 - Transmitting apparatus, receiving apparatus, and communication system - Google Patents

Description

  The present invention relates to a transmission device, a reception device, and a communication system that transmit and receive a signal including a known sequence.

  With recent downsizing and higher functionality of communication devices, the problem (self-poisoning phenomenon) that weak electromagnetic noise generated in circuits inside the device wraps around the reception circuit and degrades reception performance has become serious. This is often caused by a clock signal used in a digital circuit and its multiplied / divided frequency.

In general, a radio receiver or an optical receiver converts a high frequency band (Radio Frequency: RF) signal that is actually transmitted and received into a baseband (BB) and performs reception signal processing. Depending on the receiver, the RF signal is once converted into an intermediate frequency (IF) signal and then converted back into the BB signal. If the clock signal of the digital circuit, its harmonics or frequency division leaks into the receiving circuit in any frequency band of BB, IF, and RF, they become interference signals for the received signal. Such a problem may occur not only in wireless communication but also in wired communication. In the following description, the clock signal (frequency is the signal bandwidth) of the digital circuit block that digitally processes the received signal, or the ²ⁿ times (n is an integer) multiple wave / frequency division of this clock signal frequency. Interference waves caused by the are treated as electromagnetic noise (Electro-Magnetic Interference: EMI). The EMI is a standing wave having a periodicity comparable to the signal bandwidth (below the signal bandwidth frequency and more than 1/1000 of the signal bandwidth) when the received signal is digitally processed.

  In a reception environment where EMI exists (hereinafter referred to as an EMI environment), the data performance of the received signal is subject to interference, so that the reception performance deteriorates. An example of reception performance in such an EMI environment is disclosed in Non-Patent Document 1 below. Here, an example of the reception performance described in Non-Patent Document 1 is a characteristic when transmission path estimation is ideally performed. However, in the EMI environment, there is a problem that a correct estimated value cannot be obtained in an operation of estimating a transmission path necessary for demodulating data performed before data demodulation.

  In response to this problem, Non-Patent Document 2 below discloses a technique for improving the reception performance by reducing the influence of EMI. In the technique described in Non-Patent Document 2, the reception performance is improved by reducing the demodulation level of the EMI subcarrier (subcarrier affected by EMI).

  Non-Patent Document 3 below describes conventional EMI countermeasure technology. Specifically, a physical measure such as predicting a route through which EMI leaks and arranging a metal shield or a radio wave absorbing sheet in the predicted result, a measure in circuit board design, and the like are described.

Hiroshi Nishimoto, Hiroaki Hirai, Akihiro Shibuya, "OFDM transmission characteristics evaluation in electromagnetic noise environment," IEICE Society Conference, B-5-67, Sept. 2009. Hiroshi Nishimoto, Hiroaki Hirai, Akinori Hira, Hiroki Kubo, Toshihiko Nishimura, Takeo Ogane, Yasutaka Ogawa, "Spectrum reduction method in multicarrier transmission under electromagnetic noise environment," IEICE General Conference, B-5-69, March 2010. Nikkei Electronics, August 24, 2009, No.1011

  However, the conventional technique has the following problems. That is, in order to reduce the influence of EMI and improve the reception performance by applying the technique described in Non-Patent Document 2, it is necessary to detect subcarriers affected by EMI. Non-Patent Document 2 shows that subcarriers affected by EMI are detected using no-communication time or the like, but in this case, there is a problem in that transmission efficiency is reduced by providing a no-communication section. there were. Further, the EMI countermeasure technique described in Non-Patent Document 3 requires a physical area and volume, and there is a problem that it is difficult to reduce the size and increase the integration of the communication device.

  The present invention has been made in view of the above, and in a EMI environment, a transmission apparatus and a reception apparatus capable of improving reception performance without taking the above-described physical countermeasures and countermeasures in circuit board design. And to obtain a communication system. It is another object of the present invention to provide a transmission device, a reception device, and a communication system that can detect an EMI component from a reception signal by reception signal processing on the reception side without providing a no-communication section in an EMI environment.

  In order to solve the above-described problems and achieve the object, the present invention configures a communication device on the transmission side of a communication system and receives a training signal sequence for channel estimation composed of two or more identical training signals. A transmission device for transmitting to a communication device on the side, a transmission phase rotation means for rotating the phase of each of the training signals constituting the training signal sequence by a different rotation amount, and a training signal after the phase rotation Signal transmission means for performing predetermined signal processing on the sequence and transmitting the sequence.

  According to the present invention, the EMI component included in the training signal can be removed on the signal receiving side, and the transmission path estimation value based on the training signal from which the influence of EMI has been removed can be calculated. As a result, the reception performance can be improved and the communication quality can be improved.

FIG. 1 is a diagram of a configuration example of a transmission apparatus according to the first embodiment. FIG. 2A is a diagram illustrating an example of the EMI phase rotation amount and the transmission phase rotation amount. FIG. 2B is a diagram of an example of the EMI phase rotation amount and the transmission phase rotation amount. FIG. 3 is a diagram of a configuration example of the receiving apparatus according to the first embodiment. FIG. 4 is a diagram illustrating an example of a transmission phase rotation, a reception phase rotation, and a synthesis result. FIG. 5 is a diagram illustrating an example of a transmission path estimation result by the receiving apparatus according to the first embodiment. FIG. 6 is a diagram of a configuration example of the receiving apparatus according to the second embodiment. FIG. 7 is a diagram illustrating an example of an EMI estimation result by the receiving apparatus according to the second embodiment. FIG. 8 is a diagram illustrating a configuration example of a transmission processing unit included in the transmission apparatus according to the third embodiment. FIG. 9 is a diagram illustrating a configuration example of a reception processing unit included in the reception device of the third embodiment. FIG. 10 is a diagram illustrating a configuration example of a transmission apparatus to which the OFDM scheme is applied. FIG. 11 is a diagram illustrating a configuration example of a receiving apparatus to which the OFDM scheme is applied. FIG. 12 is a diagram illustrating an example of a time waveform of EMI caused by the OFDM analog BB signal and the divided clock signal. FIG. 13 is a diagram illustrating a result of EMI observed in the frequency domain. FIG. 14 is a diagram illustrating a relationship between a subcarrier signal and EMI in the time domain. FIG. 15 is a diagram illustrating an example of a relationship between a subcarrier signal and EMI. FIG. 16 is a diagram illustrating an example of observation signals and desired signal points. FIG. 17 is a diagram illustrating an example of a transmission path estimation result under an EMI environment by a conventional receiving apparatus.

  Hereinafter, embodiments of a transmission device, a reception device, and a communication system according to the present invention will be described in detail with reference to the drawings. Note that the present invention is not limited to the embodiments.

  First, the reception performance in the EMI environment when the present invention is not applied will be described as a prerequisite technique. Specifically, using an Orthogonal Frequency Division Multiplexing (OFDM) transmission system, an EMI component observed in a receiver (receiving device) under an EMI environment, and a transmission path estimation error in that case, This will be described with reference to the drawings. In the description, an OFDM transmission system based on the IEEE 802.11a wireless LAN specification is assumed, the frequency bandwidth to be used is 20 MHz, the fast Fourier transform (FFT), and the fast inverse Fourier transform (IFFT). The number of points is 64, the number of subcarriers is 64, and the cyclic prefix (CP) length is 16 points (1/4 of the FFT length).

  FIG. 10 is a diagram illustrating a configuration example of a transmission apparatus (transmitter that performs signal transmission in a transmission-side communication apparatus) that configures a communication system to which the OFDM scheme is applied. The transmission apparatus illustrated in FIG. 10 includes a transmission port (transmission antenna) 101, a BB / RF conversion unit 110, and a transmission processing unit 120. The transmission processing unit 120 includes a primary modulation unit 125, a sequence switching / multiplexing unit 124, an IFFT unit 123, a CP addition unit 122, and a digital / analog (Digital / Analog: D / A) conversion unit 121. It is configured.

  A signal transmission operation will be described. In the transmission apparatus configured as described above, a transmission bit sequence s130 that is an information data sequence is input to the primary modulation unit 125, and the primary modulation unit 125 performs phase shift keying (PSK) on the transmission bit sequence s130. And primary modulation mapping such as quadrature amplitude modulation (QAM). The resulting signal is output to sequence switching / multiplexing section 124 as primary modulation signal s125. On the other hand, the training sequence s 131 is also input to the sequence switching / multiplexing unit 124. Note that the training sequence may be called a known signal, a unique word, a learning signal, or the like. The training sequence s131 is a known signal sequence also in the receiving device (receiving-side communication device), and is a fixed signal sequence after primary modulation such as BPSK (Binary PSK) is performed.

  For example, the sequence switching / multiplexing unit 124 selects the training sequence s131 when generating a preamble estimation preamble symbol attached to the head of an OFDM frame or the like, and generates a data symbol other than that. Primary modulation signal s125 is selected, and the selected signal is output to IFFT section 123 as subcarrier signal s124. Alternatively, sequence switching / multiplexing section 124 selects training sequence s131 for the pilot subcarrier in the data symbol, selects primary modulation signal s125 for the other data subcarriers, and IFFT section as subcarrier signal s124. To 123. However, when the training sequence s131 is output as the subcarrier signal s124, the same signal is output at least twice.

  The IFFT unit 123 performs IFFT on the input subcarrier signal s124, which is a frequency domain signal, to convert it into a time domain signal, and outputs the time domain signal s123 to the CP adding unit 122. CP adding section 122 adds a CP to time domain signal s123, and outputs the result as digital BB signal s122 to D / A converting section 121. The D / A converter 121 converts the digital BB signal s122 into an analog BB signal s121 and outputs the analog BB signal s121 to the BB / RF converter 110. The BB / RF conversion unit 110 converts the input analog BB signal s121 into an RF signal s110. The RF signal s110 obtained in this way is transmitted from the transmission port 101 (for example, a transmission antenna).

  FIG. 11 is a diagram illustrating a configuration example of a reception device (a receiver that receives a signal in a reception-side communication device) that configures a communication system to which the OFDM scheme is applied. The receiving apparatus includes a receiving port (for example, a receiving antenna) 201, an RF / BB converter 210, a reception processor 220, a clock generator 241, and a clock divider 242. The reception processing unit 220 includes an analog / digital (A / D) conversion unit 221, a CP removal unit 222, an FFT unit 223, a transmission line equalization unit 224, a primary demodulation unit 225, a transmission line, And an estimation unit 230. The transmission path estimation unit 230 includes a combining unit 231. The reception processing unit 220 is driven at the timing of the clock signal s241 input from the outside and the divided clock signal s242. The clock signal s241 is generated by the clock generator 241. The divided clock signal s242 is generated when the clock divider 242 divides the clock signal s241. Here, it is assumed that the reception sampling rate is 20 MHz, and the frequency of the clock signal s241 is 20 MHz. The divided clock signal s242 assumes a divided signal of 32 of the clock signal s241, that is, a clock frequency of 0.625 MHz. It is assumed that the duty ratio of the frequency-divided clock signal s242 (the ratio of the pulse ON time to the clock cycle time) is 0.25. Note that the clock generation unit 241 and the clock divider 242 may be inside the reception processing unit 220.

  A signal receiving operation will be described. In the receiving apparatus configured as described above, the RF signal s201 received at the reception port 201 is input to the RF / BB converter 210, and the RF / BB converter 210 converts the input RF signal into an analog BB signal s210. . The RF / BB conversion unit 210 outputs the converted reception signal (analog BB signal s210) to the reception processing unit 220. Here, the EMIs 250 having a frequency of 0.625 MHz leaking from the clock frequency divider 242 is added to the analog BB signal s210 in the adder (EMI adding unit) 211, and the reception processing unit 220 is converted into the analog BB signal s211 including EMI. Shall be entered. However, the EMI adding unit 211 is shown for convenience in order to model EMI addition, and does not exist in an actual receiving circuit.

  In the reception processing unit 220, the input analog BB signal s 211 is input to the A / D conversion unit 221. The A / D conversion unit 221 converts the input analog BB signal s211 into a digital BB signal s221 and outputs it to the CP removal unit 222. CP removing section 222 removes CP from digital BB signal s221 and outputs the result to FFT section 223 as time domain signal s222. The FFT unit 223 performs FFT on the time domain signal s222 and converts it into a subcarrier signal s223. The subcarrier signal s223 is input to the transmission path equalization unit 224 and the transmission path estimation unit 230. When the input subcarrier signal s223 is a preamble symbol or a pilot subcarrier, the combining unit 231 of the transmission path estimation unit 230 estimates a transmission path between transmission and reception (transmission path from the transmitter to the receiver) and transmits A path estimation value s230 is obtained. When there is one preamble symbol or pilot subcarrier, channel estimation is performed from this signal to obtain a channel estimation value s230. When there are a plurality of preamble symbols or pilot subcarriers, they are first combined, and transmission path estimation is performed from the resulting signal (subcarrier signal after combining) to obtain a transmission path estimation value s230. The transmission path equalization unit 224 equalizes the subcarrier signal s223 using the transmission path estimation value s230, and obtains an equalized signal s224. The primary demodulation unit 225 demaps the equalized signal s224 and outputs the received bit sequence s225 obtained as a result.

FIG. 12 is a diagram showing an example of a time waveform of an OFDM analog BB signal (corresponding to the above s210) and an EMI (corresponding to the above s250) caused by the divided clock signal. In the example of the EMI environment, the signal s211 input to the reception processing unit 220 is a combination of these two signals. As shown in FIG. 12, the clock signal indicated by the broken line is generally a square wave, and in the case of 1 / ^{2n of} the sampling rate (where n is a positive integer), the clock signal has cyclicity within the FFT interval, It is expressed by the sum of carrier components, that is, the sum of a plurality of sine waves. The result of observing this EMI in the frequency domain is as shown in FIG. In FIG. 13, the subcarrier number shown on the horizontal axis represents a physical frequency number, not a logical subcarrier number. That is, the component of subcarrier number 0 is a direct current component, and the higher the absolute value of the subcarrier number, the higher the frequency component. As can be seen from FIG. 13, a waveform that is a square wave in the time domain is observed as a plurality of subcarrier components in the frequency domain. The clock signal is a time-continuous signal, and the same spectral component is observed at any time when the cyclicity is satisfied within the FFT interval. Note that E _emi / N ₀ described in FIG. 13 is the EMI energy (E _emi ) versus thermal noise power spectral density (N ₀ ) in the FFT interval, and is defined in Non-Patent Documents 1 and 2 above. EMI level. Further, the SNR described in FIG. 13 indicates a signal-to-noise power ratio.

  Here, focusing on a certain subcarrier component, an OFDM signal waveform and an EMI waveform are considered. As an example, consider a component with subcarrier number f = 1, that is, a component having one cycle within the FFT interval. It is assumed that EMI exists in the subcarrier. At this time, the relationship between the subcarrier signal and the EMI in the time domain is, for example, as shown in FIG. For simplification, the subcarrier signal is assumed to be a BPSK-modulated training signal here, and is the same signal in OFDM symbols # 1 to # 4. Since the subcarrier signal and the EMI are the same subcarrier component, the frequency is the same. However, focusing on the FFT interval of each OFDM symbol, the subcarrier signal has the same waveform in every OFDM symbol, whereas EMI has a phase shift for each OFDM symbol because a CP is inserted in each OFDM symbol. doing. In this example, since the CP length is ¼ of the FFT interval, the EMI phase is shifted by π / 2 for each OFDM symbol. This is shown in FIG. 15 on the In-phase channel (Ich) / Quadrature-phase channel (Qch) plane, that is, on the constellation. In FIG. 15, a subcarrier signal is a subcarrier signal in which no EMI component is superimposed, and an observation signal is a subcarrier signal in which an EMI component is superimposed (the subcarrier signal shown in the figure). EMI is synthesized). The subcarrier signal has the same point in every OFDM symbol, whereas EMI rotates by π / 2 around the origin for each OFDM symbol. Actually, a signal obtained by combining the subcarrier signal and the EMI is observed like the observed signal shown in the figure. At this time, the observation signal rotates around the desired subcarrier signal point for each OFDM symbol on the circumference of the radius corresponding to the EMI amplitude.

  In such a situation, for example, when channel estimation is performed using two consecutive OFDM symbols, the observation signal of the second symbol is π around the desired signal point with respect to the first symbol, as shown in FIG. Since the rotation is performed by / 2, the result of combining the two symbols in the combining unit 231 is greatly deviated from the desired signal.

  FIG. 17 is a diagram showing an example of a transmission path estimation result under an EMI environment by a conventional receiving apparatus, and shows a transmission path estimation result when an EMI spectrum as shown in FIG. 13 exists. As shown in the figure, when the EMI spectrum exists, the transmission path estimation result according to the conventional method is distorted due to the influence of EMI, so that a correct estimated value cannot be obtained.

When the EMI phase rotation amount is formulated, when the subcarrier f has an EMI spectrum component, the EMI phase rotation amount θ _{m, f} in the subcarrier f m symbols after the reference OFDM symbol is expressed by the following equation (1). The

Here, ρ _cp is the CP ratio, and represents the ratio of the CP length to the FFT interval length. In the above example, ρ _cp = 1/4. As can be seen from this equation (1), the EMI phase rotation amount θ _{m, f} differs depending on the OFDM symbol number m and the subcarrier f (determined by m and f).

  The present invention is configured to improve reception quality by utilizing the above-described properties. Hereinafter, Embodiments 1 to 3 of the present invention will be described.

Embodiment 1 FIG.
In the present embodiment, an OFDM transmission system based on the IEEE802.11a specification is assumed in the same manner as the base technology described above. Further, it is assumed that the frequency bandwidth (original clock frequency) to be used is 20 MHz, the number of FFT / IFFT points is 64, the number of subcarriers is 64, the CP length is 16 points, and ρ _cp = _¼ . Further, it is assumed that the receiving apparatus is in an EMI environment, and the same EMI (clock signal having a 32 frequency division and a duty ratio of 0.25) similar to the above-described prerequisite technology is superimposed on the received analog BB signal. Here, for simplification of explanation, it is assumed that the receiving apparatus performs transmission path estimation in the preamble section of the OFDM frame. However, the present invention is not limited to this case. It is also possible to perform transmission path estimation with pilot subcarriers according to a similar procedure. A continuous M symbol (two or more symbols) is used as a preamble of the OFDM frame, and the M channel is linearly synthesized on the receiving side to perform transmission path estimation. The training sequence is assumed to use the same sequence for each OFDM preamble symbol.

First, the transmission apparatus of the communication system according to the present embodiment will be described. FIG. 1 is a diagram of a configuration example of a transmission apparatus according to the first embodiment. It should be noted that the same components as those in the transmission apparatus of FIG. As illustrated, the transmission apparatus according to the present embodiment is obtained by replacing the transmission processing unit 120 of the transmission apparatus illustrated in FIG. The transmission processing unit 12 is obtained by adding a transmission phase rotation unit 126 to the transmission processing unit 120 of the transmission apparatus illustrated in FIG. The transmission phase rotation unit 126 receives the training sequence s131, applies an appropriate phase rotation to the training sequence s131, and outputs the training sequence s126 obtained as a result of the phase rotation to the sequence switching / multiplexing unit 124. The transmission phase rotation unit 126 performs phase rotation on the training signal of the subcarrier f in the m-th OFDM symbol (m = 0,..., M−1) according to the phase rotation amount φ _{m, f} shown in the following equation (2). give.

When the original training signal to send d _f subcarrier f (s131), the signal s _m after phase rotation sent on subcarriers _{f, f} (s126) is expressed by the following equation (3).

Examples of the EMI phase rotation amount θ _{m, f} and the transmission phase rotation amount φ _{m, f} are shown in FIGS. FIG. 2-1 shows an example when M = 2, and FIG. 2-2 shows an example when M = 3.

  By transmitting the training signal with phase rotation as described above over a plurality of symbol times, the transmitting apparatus removes the influence of EMI from the received training signal in the opposite apparatus (receiving apparatus) as will be described later. Communication quality can be improved.

  Next, the receiving apparatus of the communication system according to the present embodiment will be described. FIG. 3 is a diagram of a configuration example of the receiving apparatus according to the first embodiment. It should be noted that the same components as those in the receiving apparatus of FIG. As shown in the figure, the receiving apparatus of this embodiment is obtained by replacing the receiving processing unit 220 of the receiving apparatus shown in FIG. The reception processing unit 22 is obtained by replacing the transmission path estimation unit 230 provided in the reception processing unit 220 of the reception apparatus illustrated in FIG. The transmission path estimation unit 23 is configured by adding a reception phase rotation unit 232 to the transmission path estimation unit 230.

In the transmission path estimation unit 23, the reception phase rotation unit 232 gives an appropriate phase rotation to the input subcarrier signal s 223 and outputs the subcarrier signal s 232 after the phase rotation to the synthesis unit 231. In reception phase rotation section 232, with respect to the signal of subcarrier f in the m-th OFDM symbol, rotation opposite to the transmission phase rotation (phase rotation given on the signal transmission side), that is, the phase of −φ _{m, f} Give rotation.

The received signal r _{m, f} (s223) in the subcarrier f is expressed by the following equation (4) using the transmission path response h _f and the EMI component _{nm, f} .

Here, ρ _f is the amplitude value of the EMI component, and η _f is the phase offset amount of EMI. For simplification of explanation, the thermal noise component added at the reception point is omitted here. When the received M symbols are combined after giving a phase rotation of −φ _{m, f} , the EMI component is removed as shown in the following equation (5).

Dividing the combined signal shown in Equation (5) in the original training signal d _f, represented by the following formula (6), s23 (h _f denoted by ^) channel estimation value without the EMI component is obtained It is done.

  As described above, in this example, it is assumed that there is no thermal noise in order to simplify the description.

The above-described transmission path estimation method (transmission path estimation operation in the transmission path estimation unit 23) will be specifically described with reference to the drawings. FIG. 4 is a diagram illustrating an example of transmission phase rotation, reception phase rotation, and a combination result with subcarrier f = 1 when M = 2. For simplification, the case where h _f = 1 and η _f = 0 is shown. 4, (a) and (b) are input signals rm _{, f} (s223) to the reception phase rotating unit 232, (c) and (d) are input signals s232 to the combining unit 231, and (e) are The received signal (signal shown by the above formula (5)) synthesized by the synthesis unit 231 is shown. As can be seen from FIG. 4, the phase rotation amount θ _{m, f} of EMI shown in the above equation (1) is determined by ρ _cp , subcarrier f, and OFDM symbol number m. If transmission phase rotation and reception phase rotation are applied so as to cancel each other, the combined received signal is not affected by EMI.

  An example of a transmission path estimation result by the receiving apparatus of this embodiment is shown in FIG. Here, the evaluation conditions are the same as when the estimation result of FIG. 17 used in the description of the base technology is obtained. As is clear from FIG. 5, unlike the estimation result of FIG. 17 (estimation result by the conventional receiving apparatus that does not rotate the phase), the transmission path estimation result according to the first embodiment is not affected by EMI and is ideal. It can be seen that the estimation result is almost the same as the value.

In the above description of the embodiment, the case where transmission path estimation is performed using continuous M symbols in the EMI environment has been described. However, the present invention is not limited thereto, and transmission path estimation is performed even when M symbols are temporally dispersed. Is possible. The EMI phase rotation amount θ ′ _{m, f in} this case can be formulated as follows.

Here, τ _m indicates the symbol time of the m-th symbol with respect to the reference symbol (m = 0), and τ ₀ = 0. Similarly, the transmission phase rotation amount φ ′ _{m, f} in this case can be formulated as the following equation (8).

  The transmission phase rotation amount may be given by the following equation (9).

  In the present embodiment, the procedure has been described for the case where the transmission path is estimated using the preamble of the OFDM frame in the EMI environment. However, the present invention is not limited to this, and the transmission path estimation procedure described above can be applied to pilot subcarriers in data symbols.

  Further, in the present embodiment, the case where transmission path estimation is performed in an EMI environment is illustrated. However, the present invention is not limited to this, and by performing the transmission phase rotation and the reception phase rotation according to the present embodiment, the signal synthesized by the synthesis unit 231 of the reception device (the signal expressed by the above equation (5)) is not EMI. The reception apparatus may be configured to estimate other parameters such as carrier frequency deviation from the synthesized signal by utilizing the fact that no component is included.

  In this embodiment, the transmission port 101 in FIG. 1 and the reception port 201 in FIG. 3 are used as antennas for wireless communication. However, the present invention is not limited to this, and wired communication is performed between transmission / reception devices. The present invention can be applied to a system that has been made. In the case of wired communication, the BB / RF conversion unit 110 and the RF / BB conversion unit 210 may be omitted.

  In this embodiment, transmission path estimation in an OFDM transmission system has been described. However, the present invention is not limited thereto, and the present invention can be applied to any system that performs multicarrier communication.

Further, in the present embodiment, description has been made assuming that a CP is added to each OFDM symbol. However, the present invention is not limited to this, and an OFDM system to which no CP is added may be used. In this case, ρ _cp = 0 in each of the above equations.

  The configurations of the transmission device and the reception device described in the present embodiment are merely examples, and any configuration may be used as long as the above-described training signal phase rotation processing can be performed.

  As described above, in the communication system according to the present embodiment, the transmission apparatus responds to a plurality of training signals having the same content according to the frequency (subcarrier) and the transmission time (symbol number corresponding to the symbol transmission timing). The receiving device applies the rotation opposite to the rotation given by the transmitting device to each training signal, and synthesizes the training signal after the phase rotation obtained as a result. did. As a result, a training signal from which the influence of EMI has been removed can be obtained on the receiving side, and a transmission path estimation value based on the training signal from which the influence of EMI has been removed can be obtained. That is, it is possible to realize a receiver that demodulates a received signal by performing transmission path estimation with high accuracy even in an EMI environment.

Embodiment 2. FIG.
Next, the second embodiment will be described. It is assumed that the transmission device described in the first embodiment is used as the transmission device, and that the transmission environment and conditions are the same as those in the first embodiment.

  FIG. 6 is a diagram of a configuration example of the receiving apparatus according to the second embodiment. Note that the same components as those of the receiving apparatus (see FIG. 3) described in Embodiment 1 are denoted by the same reference numerals. As illustrated, the receiving apparatus according to the present embodiment is obtained by replacing the receiving processing unit 22 of the receiving apparatus described in the first embodiment (the receiving apparatus illustrated in FIG. 3) with a receiving processing unit 22a. The reception processing unit 22a is obtained by replacing the transmission path estimation unit 23 of the reception processing unit 22 with a transmission path estimation unit 23a. The transmission path estimation unit 23a is combined with the transmission path estimation unit 23 by a combining unit 231a. In addition, a reception phase rotation unit 232a is added.

  In the transmission path estimation unit 23a, the reception phase rotation unit 232a gives the input subcarrier signal s223 a phase rotation different from that of the reception phase rotation unit 232, and the subcarrier signal s232a after the phase rotation is supplied to the synthesis unit 231a. Output. The combining unit 231a combines the input subcarrier signals s232a (M symbol subcarrier signals), and outputs the combined signal as an EMI estimated value s23a.

Below, the procedure of EMI estimation is demonstrated. It is assumed that the transmitted training signal is subjected to transmission phase rotation processing by the phase rotation amount φ _{m, f} given by the equation (2) shown in the first embodiment. Reception phase rotation section 232a applies a rotation opposite to the EMI phase rotation, that is, a phase rotation of −θ _{m, f to} the signal of subcarrier f in the m-th OFDM symbol. When the signal (OFDM symbol) after this phase rotation is given is synthesized by the synthesis unit 231a, an EMI estimated value s23a is obtained as shown in the following equation (10).

The received signal rm _{, f} is assumed to be given by the equation (4) shown above.

  An example of the EMI estimation result by the receiving apparatus of this embodiment is shown in FIG. Here, the evaluation conditions are the same as in the examples of FIGS. As shown in FIG. 7, by performing EMI estimation by the receiving apparatus according to the present embodiment, an estimation result close to actual EMI can be obtained.

  When the EMI estimation operation in the receiving apparatus according to the present embodiment is applied, the technique described in Non-Patent Document 2 (the technique for improving the reception performance by reducing the demodulation level of the EMI subcarrier) can be applied. become. As a result, the reception performance can be further improved. In addition, since no special control such as providing a non-communication section is required, transmission efficiency is not reduced by the EMI estimation operation.

  In the present embodiment, a case has been described in which EMI estimation is performed using consecutive M symbols in an EMI environment. However, the present invention is not limited to this, and even when M symbols are dispersed in time, the phase rotation amount can be formulated as shown in equations (7) to (9) in the first embodiment, and EMI estimation is also performed. It is possible as well.

  Further, in the present embodiment, the procedure has been described for the case where EMI estimation is performed using the preamble of the OFDM frame in the EMI environment. However, the present invention is not limited to this, and it is also possible to apply the above-described EMI estimation procedure to pilot subcarriers in data symbols.

  Further, in the present embodiment, the case where the reception port 201 of FIG. 6 is an antenna for wireless communication has been described. However, the present invention is not limited to this, and for a system configured to perform wired communication between transmission / reception devices. However, the present invention is applicable. In the case of wired communication, the RF / BB conversion unit 210 may not be provided.

  In this embodiment, transmission path estimation in an OFDM transmission system has been described. However, the present invention is not limited thereto, and the present invention can be applied to any system that performs multicarrier communication.

  The configurations of the transmission device and the reception device described in the present embodiment are merely examples, and any configuration may be used as long as the above-described training signal phase rotation processing can be performed.

  As described above, the receiving apparatus according to the present embodiment provides each training signal with a rotation opposite to the EMI phase rotation, and synthesizes the training signal after the phase rotation obtained as a result to calculate the EMI estimated value. It was. As a result, it is possible to detect the EMI component from the received signal by the received signal processing without providing a non-communication section, and the technique described in Non-Patent Document 2 can be realized without reducing the transmission efficiency. .

Embodiment 3 FIG.
As described in the first and second embodiments, a plurality of training signals are subjected to phase rotation on the transmission side, and an appropriate phase rotation is applied to them on the reception side, and then combined, so that reception that does not include the influence of EMI A training signal or EMI estimate is obtained. In the first and second embodiments, the case where the phase rotation applied to the transmission signal and the reception signal is performed in the frequency domain has been described. In contrast, in the present embodiment, the phase rotation given by Equation (2), Equation (8), or Equation (9) shown above can be realized in the time domain. A communication system (transmitting device, receiving device) that obtains the same effects as those of Embodiments 1 and 2 will be described.

As is clear from the description of the first and second embodiments, whether transmission path estimation or EMI estimation is performed can be switched only by the phase rotation amount on the reception side. The description will be made without limiting the target to the transmission path or EMI. In the following description, it is assumed that the communication system is the same as in the first and second embodiments, and the description will focus on the transmission processing unit of the transmission device and the reception processing unit of the reception device. However, unlike Embodiments 1 and 2, this embodiment is also applicable to single carrier transmission with CP insertion. In the following description, it is assumed that the phase rotation of the phase rotation amount φ ′ _{m, f} shown in Expression (8) is performed in the transmission processing unit of the transmission device included in the transmission-side communication device, and the communication on the reception side It is assumed that the phase rotation of −φ ′ _{m, f} is performed in the reception processing unit of the receiving device included in the device. However, as described above, it is clear that EMI estimation can be realized by performing phase rotation of −θ ′ _{m, f} in the reception processing unit, and this case is also included in this embodiment.

  FIG. 8 is a diagram illustrating a configuration example of a transmission processing unit included in the transmission apparatus according to the third embodiment. In addition, the same code | symbol is attached | subjected to the component similar to the transmitter (refer FIG. 1 etc.) demonstrated in previous embodiment. The illustrated transmission processing unit 13 includes a D / A conversion unit 121, a CP addition unit 122, a sequence switching unit 133, an overall phase rotation unit 134, and a time shift unit 135. Although not shown, the data signal time series s137 and the training signal time series s136 are assumed to have been subjected to appropriate encoding and primary modulation if necessary, and these may be multicarrier signals. It may be a signal.

An operation of the transmission processing unit 13 illustrated in FIG. 8 will be described. In the transmission processing unit 13, the data signal time series s 137 is input to the series switching unit 133. On the other hand, the training signal time series s136 for the training section is input to the time shift unit 135, and the time shift unit 135 performs a cyclic shift for each block in which 1 CP is inserted into the training signal time series s136. When the block length (the length of the block after the CP is inserted) is Nb and the block time for the reference block at the block number m is τ _m , the time cyclic shift amount Δ _m for the block number _m is expressed by the following equation (11). It is represented by

When the time shift unit 135 is observed in the frequency domain the signal s135 obtained by executing the time cyclic shift by delta _m, so that the phase rotation shown in the following equation in the frequency point f (12) is performed.

  Next, the entire phase rotation unit 134 uniformly applies the phase rotation represented by the following equation (13) to the entire block with respect to the signal s135 after the time cyclic shift.

  Since the phase rotation shown in Expression (13) is uniformly applied to the entire block, the phase rotation is the same in both the time domain and the frequency domain. As a result, the output signal s134 of the overall phase rotation unit 134 is a signal that has been subjected to the phase rotation represented by the following equation (14) in the frequency domain.

  The sequence switching unit 133 outputs the input signal s134 as the time domain signal s133 at the time of training block transmission, and outputs the input signal s137 as the time domain signal s133 at the time of data transmission. Subsequent signal processing is the same as that of the transmission apparatus of Embodiment 1 (see FIG. 1). Therefore, the description is omitted. Also, the description of the circuits and signal lines denoted by the same reference numerals as those of the transmission apparatus shown in FIG.

  FIG. 9 is a diagram illustrating a configuration example of a reception processing unit included in the reception device of the third embodiment. Note that the same components as those of the receiving apparatus (see FIG. 3 or the like) described in the previous embodiment are denoted by the same reference numerals. The illustrated reception processing unit 26 includes an A / D conversion unit 221, a CP removal unit 222, an overall phase reverse rotation unit 261, a time reverse shift unit 262, and a synthesis unit 263. Although not shown in the figure, the received data signal time series s222 and the estimated signal s263 are subjected to appropriate primary demodulation and decoding at a later stage if necessary. These may be multicarrier signals or single carrier signals.

An operation of the reception processing unit 26 illustrated in FIG. 9 will be described. In the reception processing unit 26, the time signal s222 after CP removal is input to the overall phase reverse rotation unit 261 for the block signal in the training section. The overall phase reverse rotation unit 261 performs −β _m phase rotation on the entire block with respect to the input signal. That is, a phase rotation opposite to the phase rotation given by the overall phase rotation unit 134 on the transmission side (transmission device) is given. A signal obtained by performing this process is output to the time reverse shift unit 262 as a block signal s261 after the overall phase reverse rotation. Note that the overall phase reverse rotation unit 261 does not perform phase rotation during EMI estimation. At time reverse shift unit 262, to block signal s261 after total phase reverse rotation input, run time cyclic shift of - [delta _m, the combining unit 263 and the resulting signal as a block signal s262 after inverse temporal shift Output to. As can be understood from the meaning of the time cyclic shift and the entire phase rotation process in the transmission device described above, the processing in the entire phase reverse rotation unit 261 and the time reverse shift unit 262 of the reception device is transmitted when observed in the frequency domain. This is equivalent to performing phase rotation of −φ ′ _{m, f} at the time of path estimation, and equivalent to performing phase rotation of −θ ′ _{m, f} at the time of EMI estimation. The synthesis unit 263 synthesizes all the M block time cyclic shift of - [delta _m is applied to obtain a proper estimation signal S263. Note that the description of the circuits and signal lines denoted by the same reference numerals as those of the receiving apparatus illustrated in FIG. 3 is omitted.

  In the present embodiment, the case where all the phase rotation processes are executed in the time domain has been described. However, the present invention is not limited to this, and the entire phase rotation processing in the transmission device or the entire phase reverse rotation processing in the reception device may be performed in the frequency domain.

  In addition, the configuration of the transmission device and the configuration of the reception device described in this embodiment are merely examples, and any configuration is possible as long as the above-described overall phase rotation processing and time shift processing can be performed.

  In this embodiment, the entire phase rotation process is performed after the time shift process is performed in the transmission apparatus. However, the present invention is not limited to this, and the time shift process may be performed after the entire phase rotation process is performed.

  In the present embodiment, the time reverse shift process is performed after the overall phase reverse rotation process is performed in the receiver. However, the present invention is not limited to this, and the entire phase reverse rotation process may be performed after the time reverse shift process is performed.

  Thus, in the present embodiment, it has been shown that the phase rotation process in the frequency domain described in the first and second embodiments can be performed in the time domain. When the configuration of the present embodiment is applied, it is not necessary to convert the signal to the frequency domain at the time of phase rotation, and it is easy to calculate a transmission path estimation value from which the influence of EMI has been removed and to perform highly accurate EMI estimation. Can be realized.

  The present invention has been described with reference to the first to third embodiments. However, these embodiments are exemplifications, and it is possible to add various modifications to the combinations of the respective constituent elements and the respective processing processes, and that such modifications are also within the scope of the present invention. It can be easily understood by a trader.

  As described above, the transmission device, the reception device, and the communication system according to the present invention are useful for improving reception performance when reception performance deterioration occurs due to the influence of EMI.

12, 13, 120 Transmission processing unit 22, 22a, 26, 220 Reception processing unit 23, 23a, 230 Transmission path estimation unit 101 Transmission port 110 BB / RF conversion unit 121 D / A conversion unit 122 CP addition unit 123 IFFT unit 124 Sequence switching / multiplexing unit 125 Primary modulation unit 126 Transmission phase rotation unit 133 Sequence switching unit 134 Overall phase rotation unit 135 Time shift unit 201 Reception port 210 RF / BB conversion unit 211 Adder 221 A / D conversion unit 222 CP removal unit 223 FFT unit 224 Transmission path equalization unit 225 Primary demodulation unit 231, 231a, 263 Combining unit 232, 232a Reception phase rotation unit 241 Clock generation unit 242 Clock divider 261 Overall phase reverse rotation unit 262 Time reverse shift unit

Claims (13)

A transmission device that constitutes a communication device on the transmission side of a communication system and transmits a training signal sequence for channel estimation composed of two or more identical training signals to a communication device on the reception side,
Transmission phase rotation means for rotating the phase of each of the training signals constituting the training signal series by different rotation amounts;
A signal transmission means for performing predetermined signal processing and transmitting the training signal sequence after phase rotation;
With
The transmission phase rotation means is configured to perform phase rotation based on the number of training signals in one frame, the transmission time and frequency of each training signal, and the ratio between the processing unit length of the predetermined signal processing and the length of the cyclic prefix. Phase rotate each training signal by an amount ,
A transmission apparatus characterized by the above. The communication system is a multi-carrier communication system;
M is the number of symbols included in the training signal transmission section in one frame, m is the symbol number (m = 0, 1, 2,..., M−1) in the training signal transmission section, and τ _m is m = 0. The symbol time of the m-th symbol with respect to the reference symbol when the 0th symbol is the reference symbol, f is the subcarrier number, and ρ _cp is the ratio of the processing unit length of the predetermined signal processing to the length of the cyclic prefix If
The transmission phase rotation means, the rotation amount phi _'m, the rotation amount phi shown in _f or the following formula (2)' shown by the following formula (1) _'m, at _f, that is the phase rotation of each training signal The transmission device according to claim 1 , wherein

A transmission device that constitutes a communication device on the transmission side of a communication system and transmits a training signal sequence for channel estimation composed of two or more identical training signals to a communication device on the reception side,
A time shift means for cyclically shifting each of the training signals constituting the training signal sequence by a different shift amount;
Phase rotation means for rotating the phase of each of the training signals after the time cyclic shift by a different rotation amount;
A signal transmission means for performing predetermined signal processing and transmitting the training signal sequence after phase rotation;
With
The time shift means includes the transmission time of each training signal, the sum of the processing unit length of the predetermined signal processing and the length of the cyclic prefix, and the processing unit length of the predetermined signal processing and the length of the cyclic prefix. Each training signal is cyclically shifted by a shift amount based on the ratio of
The phase rotation means rotates each training signal by a phase rotation amount based on the number of training signals in one frame and the transmission time of each training signal .
A transmission apparatus characterized by the above. A transmission device that constitutes a communication device on the transmission side of a communication system and transmits a training signal sequence for channel estimation composed of two or more identical training signals to a communication device on the reception side,
Phase rotation means for rotating the phase of each of the training signals constituting the training signal series by different rotation amounts;
Time shifting means for cyclically shifting each of the training signals after the phase rotation by different shift amounts;
Signal transmitting means for performing predetermined signal processing on the training signal sequence after the time cyclic shift and transmitting the training signal sequence;
With
It said phase rotation means, 1 and training the number of signals in a frame, each training signal is much phase rotation in phase rotation amount based on the transmission time of each training signal,
The time shift means includes the transmission time of each training signal, the sum of the processing unit length of the predetermined signal processing and the length of the cyclic prefix, and the processing unit length of the predetermined signal processing and the length of the cyclic prefix. Each training signal is cyclically shifted by a shift amount based on the ratio of
A transmission apparatus characterized by the above. M is the number of symbols included in the training signal transmission section in one frame, m is the symbol number (m = 0, 1, 2,..., M−1) in the training signal transmission section, and τ _m is the 0th of m = 0. The symbol time of the m-th symbol with respect to the reference symbol when the symbol is the reference symbol, N _b is the sum of the processing unit length of the predetermined signal processing and the length of the cyclic prefix, and ρ _cp is the processing of the predetermined signal processing If the ratio between the unit length and the cyclic prefix length,
It said time shifting means, the shift amount delta _m shown by the following formula (3), the transmission device according to claim 3 or 4, characterized in that to time cyclic shift of each training signal.

When M is the number of symbols included in the training signal transmission section in one frame, m is the symbol number (m = 0, 1, 2,..., M-1) in the training signal transmission section,
It said phase rotation means, the rotation amount beta _m shown by the following formula (4) or-beta _m is the reverse phase of the rotation amount, according to claim, characterized in that for each training signal phase rotation 3,4 Or the transmitting apparatus of 5 .

Two or more identical training signals that constitute a communication device on the reception side of the communication system and transmitted by the communication device on the transmission side, the number of training signals in one frame, the transmission time and frequency of each training signal, , channel estimation using a training No. signal phase rotation is performed in phase rotation amount based on the ratio of the length of the processing unit length and cyclic prefix of a predetermined signal processing in the transmitting communication device A receiving device for performing
Receiving phase rotation means for performing processing opposite to the phase rotation performed in the transmission side communication device for each of the training signals included in the received signal;
Combining means for combining the training signals after the reverse processing is performed in the reception phase rotating means, and calculating a transmission path estimation value based on the combined training signals;
A receiving apparatus comprising: The communication system is a multi-carrier communication system;
M is the number of symbols included in the training signal transmission section in one frame, m is the symbol number (m = 0, 1, 2,..., M−1) in the training signal transmission section, and τ _m is m = 0. Symbol time of the m-th symbol with respect to the reference symbol when the 0th symbol is the reference symbol, f is a subcarrier number, ρ _cp is a predetermined signal that the transmission side communication device performs on the training signal after phase rotation When the processing unit length of processing and the ratio of the cyclic prefix length,
The receiving apparatus according to claim 7 , wherein the reception phase rotating means rotates each training signal by a rotation amount θ ′ _{m, f} expressed by the following equation (5).

Two or more identical training signals that constitute a receiving side communication device of the communication system and are transmitted by the transmitting side communication device , based on the number of training signals in one frame and the transmission time of each training signal The phase rotation by the amount of phase rotation , the transmission time of each training signal, the sum of the processing unit length of the predetermined signal processing and the length of the cyclic prefix in the communication device on the transmission side, and the processing unit length, a receiver that performs channel estimation using training No. signal which cyclic prefix length ratio and shift amount in a time cyclic shift based on is performed,
Time reverse shift means for performing processing opposite to the time cyclic shift performed in the transmission side communication device for each of the training signals included in the received signal;
Phase reverse rotation means for performing processing reverse to the phase rotation performed in the transmission side communication device for each of the training signals after the reverse time cyclic shift is performed in the time reverse shift means,
Combining means for combining the training signals after the reverse processing is performed in the phase reverse rotation means, and calculating a transmission path estimation value based on the combined training signals;
A receiving apparatus comprising: Two or more identical training signals that constitute the receiving side communication device of the communication system and are transmitted by the transmitting side communication device , the transmission time of each training signal, and a predetermined signal in the transmitting side communication device The number of training signals in one frame is cyclically shifted by a shift amount based on the sum of the processing unit length of processing and the length of the cyclic prefix and the ratio of the processing unit length and the length of the cyclic prefix. When, a receiving apparatus for performing transmission path estimation using a training No. signal phase rotation is performed in phase rotation amount based on the transmission time of each training signal,
Phase reverse rotation means for performing processing opposite to the phase rotation performed in the transmission side communication device for each of the training signals included in the received signal;
A time reverse shift means for performing processing opposite to the time cyclic shift performed in the communication device on the transmission side for each of the training signals after the reverse phase rotation is performed in the phase reverse rotation means;
Combining means for combining the training signals after the reverse processing is performed in the time reverse shift means, and calculating a transmission path estimation value based on the combined training signals;
A receiving apparatus comprising: Two or more identical training signals that constitute the receiving side communication device of the communication system and are transmitted by the transmitting side communication device , the transmission time of each training signal, and a predetermined signal in the transmitting side communication device The number of training signals in one frame is cyclically shifted by a shift amount based on the sum of the processing unit length of processing and the length of the cyclic prefix and the ratio of the processing unit length and the length of the cyclic prefix. When, a receiving apparatus for performing transmission path estimation using a training No. signal phase rotation is performed in phase rotation amount based on the transmission time of each training signal,
Time reverse shift means for performing processing opposite to the time cyclic shift performed in the transmission side communication device for each of the training signals included in the received signal;
Combining means for combining the training signals after the reverse processing is performed in the time reverse shift means, and calculating a transmission path estimation value based on the combined training signals;
A receiving apparatus comprising: The transmission device according to claim 1 or 2 ,
A receiving device according to claim 7 or 8 ,
A communication system comprising: The transmission device according to any one of claims 3 to 6 ,
The receiving device according to any one of claims 9 to 11 ,
A communication system comprising:

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