JP5549691B2 - Filter device and electronic musical instrument - Google Patents

Description

  The present invention relates to a filter device and an electronic musical instrument equipped with the filter device.

  Conventionally, an electronic musical instrument is equipped with a digital filter in order to change the tone of a musical tone according to the strength of performance and the passage of time. By using the digital filter, the timbre is changed by blocking or enhancing a desired frequency component of the original waveform, for example, the PCM waveform. In a conventional digital filter, a timbre change is generally realized by controlling parameters such as a cutoff frequency fc and a bandwidth Q. Such control has been inherited since the analog filter was controlled in the analog synthesizer.

  For example, a timbre change of a natural musical instrument is caused by a change in the amplitude ratio of the harmonic overtone. In particular, in the timbre change due to the strength of performance, the amplitude ratio of the higher harmonics increases as the performance is stronger (for example, the key is strongly pressed on a piano). On the other hand, the amplitude ratio of the higher harmonics decreases as the performance is weaker (the key is weaker on the piano).

  16 (a) and 16 (b) are graphs showing the waveform of an acoustic piano. FIG. 16 (a) is a waveform when a key is played with fortissimo, and FIG. 16 (b) is a waveform when a key is played with a meso piano. is there. FIGS. 17A and 17B are graphs showing spectra calculated from the waveforms of FIGS. 16A and 16B. In FIGS. 17A and 17B, the horizontal axis (frequency axis) is linear, and the vertical axis is dB. These are calculated based on the 4096 samples from the vicinity of the start of sound generation of the waveforms shown in FIGS. 16A and 16B, which are cut out by the Blackmann window function.

  FIG. 18 is a diagram showing a spectrum envelope extracted from the spectra shown in FIGS. 17 (a) and 17 (b). As shown in FIG. 18, it can be seen that the spectrum envelope has a substantially constant slope from the fundamental wave to the harmonic, and the slope varies uniformly with the strength of the performance. In the example of FIG. 18, the gradient increases (that is, approaches 0) as the performance strength increases. Therefore, even when the timbre is changed by the filter circuit in the electronic musical instrument, it is desirable that the filter circuit has a filter characteristic having a spectrum envelope as shown in FIG.

JP-A-4-78213

  The transfer function of the conventional filter is shown in equation (1).

なお、ω_０＝２πｆｃ／ｆｓ （０＜ω_０＜１）
ここに、ｆｃはカットオフ周波数、ｆｓはサンプリング周波数、ω_０はカットオフ角周波数、Ｑは選択度を表す。

Ω ₀ = 2πfc / fs (0 <ω ₀ <1)
Here, fc is the cutoff frequency, fs is the sampling frequency, ω ₀ is the cutoff angular frequency, and Q is the selectivity.

In the conventional filter circuit, since the filter characteristic is changed by ω ₀ (or fc) and Q, the filter circuit accepts ω ₀ (or fc) and Q as parameters, and changes the characteristic based on this. It is configured to let you.

  FIG. 19 is a graph showing the filter characteristics when fc is changed as a parameter in the secondary IIR filter having the characteristics shown in the equation (1). Also in FIG. 19, the horizontal axis (frequency axis) is linear (0 to 10 kHz), and the vertical axis is dB. As shown in FIG. 19, even if fc is changed, the changed filter characteristics are far from the characteristics shown in FIGS. 17 (a), (b) and FIG. 18, and therefore the cutoff frequency is controlled. Therefore, there is a problem that it is very difficult to realize a change equivalent to a change in piano tone.

  Currently, many electronic musical instruments adopt the PCM method. In this PCM method, the waveform at the time of the occurrence of a musical sound (at the time of attack) and the waveform data of the waveform for repetition thereafter are stored in the waveform memory, and after the time of attack, the waveform of the waveform for repetition is stored. By repeatedly reading the waveform data, the capacity of the waveform memory is reduced. However, since the timbre is fixed and monotonous only by repeating the same waveform, the timbre is changed over time by the filter circuit. In particular, an acoustic piano attenuates faster with higher harmonics over time. Such a change cannot be realized in a conventional filter circuit.

  For example, Patent Document 1 proposes a filter that uses, as parameters, a specific frequency f0 at which the transfer characteristic starts to change, and its rate of change, instead of the cutoff frequency fc. However, even if the filter disclosed in Patent Document 1 is used, it has been particularly difficult to realize a change equivalent to a change in piano timbre. In particular, there is a problem in that a frequency (hereinafter referred to as “transition frequency” in this specification) having a predetermined ratio from the maximum level of attenuation or enhancement at a predetermined frequency moves due to a change in parameters.

  Furthermore, in an electronic musical instrument, it is necessary to generate not only a natural instrument sound such as a piano, but also a tone color that cannot be produced by a natural instrument with a so-called synthesizer. Therefore, if the digital filter of the electronic musical instrument has only the frequency characteristics suitable for the timbre change of the natural musical instrument sound, there is a problem that it cannot cope with various music genres.

  An object of the present invention is to provide a filter device and an electronic musical instrument that can realize not only a timbre change equivalent to a timbre change of a natural musical instrument but also a timbre change unique to a so-called synthesizer.

前記フィルタ係数出力手段が、
サンプリング周波数ｆｓの１／２での最大ゲインをＡとすると、当該最大ゲインの１／２のゲインとなる周波数をｆ０とした場合、前記フィルタ係数ｂ及びｃを、

の演算により算出して前記フィルタ手段に出力することを特徴とするフィルタ装置により達成される。 An object of the present invention is to provide filter coefficient output means for outputting filter coefficients b and c , filter coefficients b output from the filter coefficient output means for a musical sound signal having a predetermined frequency supplied from the outside, and a filter device having a transfer function H (z) below in order to perform a filtering process based on a filter characteristic defined by c ,

The filter coefficient output means comprises:
Assuming that the maximum gain at 1/2 of the sampling frequency fs is A, when the frequency that is 1/2 of the maximum gain is f0, the filter coefficients b and c are

This is achieved by a filter device that calculates and outputs to the filter means .

In a preferred embodiment, the maximum gain A and the frequency f0 are determined corresponding to the frequency and intensity of the supplied musical sound signal.

In another preferred embodiment, the filter coefficient output means further includes an envelope generating means for changing the maximum gain A with time.

であり、
かつ前記フィルタ係数生成手段は、前記第２のパラメータをサンプリング周波数ｆｓの１／２での最大ゲインをＡとし、当該最大ゲインの１／２のゲインとなる周波数ｆ0を前記第１のパラメータとした場合に、前記フィルタ係数ｂ及びｃを、

の演算により算出して前記フィルタ手段に出力することを特徴とする電子楽器により達成される。

Furthermore, an object of the present invention is to provide sound generation instruction means for instructing the frequency and intensity of a musical sound to be generated by an external operation,
Control means for generating a first parameter based on the frequency and calculating a second parameter based on the intensity;
Filter coefficient generation means for extracting from the parameter table a set of filter coefficients corresponding to the frequency and level determined based on the first and second parameters calculated by the control means;
Filter means for performing a filtering process based on a filter characteristic defined by a set of filter coefficients from the filter coefficient generation means and outputting the musical sound signal supplied ;
In an electronic musical instrument with
The filter means has a transfer function H (z) where the filter coefficients are b and c.

And
The filter coefficient generation means uses the second parameter as the maximum gain at 1/2 of the sampling frequency fs as A, and the frequency f0 at which the gain as 1/2 as the maximum gain as the first parameter. The filter coefficients b and c are

This is achieved by an electronic musical instrument characterized in that it is calculated by the above calculation and output to the filter means.

In a preferred embodiment, the filter coefficient generating means further includes an envelope generating means for changing the maximum gain A over time.

  According to the present invention, it is possible to provide a filter device and an electronic musical instrument that can realize not only a timbre change equivalent to a timbre change of a natural musical instrument but also a timbre change unique to a so-called synthesizer.

FIG. 1 is a block diagram showing an outline of an electronic musical instrument according to an embodiment of the present invention. FIG. 2 is a block diagram showing the musical tone generating circuit according to the present embodiment in more detail. FIG. 3 is a diagram for explaining ideal filter characteristics for realizing a timbre filter of an acoustic piano depending on performance strength. FIG. 4 shows the characteristics of the second-order IIR filter when the gain is changed at a certain transition frequency. FIG. 5 is a graph showing second-order filter coefficients for realizing the filter characteristics shown in FIG. FIG. 6 is a diagram for explaining filter characteristics and a parameter table in the present embodiment. FIG. 7 is a graph showing respective filter characteristics (frequency characteristics) when the cut-off fc is changed in a certain bandwidth (Q = 2). FIG. 8 is a graph showing the filter coefficients b to d in each state of the cutoff fc. FIG. 9 is a block diagram showing the configuration of the filter coefficient calculation circuit according to the present embodiment. FIG. 10 is a diagram illustrating an example of a parameter table according to the present embodiment. FIG. 11 is a block diagram showing the interpolation circuit according to this embodiment in more detail. FIG. 12 is a block diagram showing an outline of the filter circuit according to the present embodiment. FIG. 13 is a diagram for explaining two-dimensional interpolation according to the present embodiment. FIG. 14 is a timing chart when calculating the filter coefficient according to the present embodiment. FIG. 15 is a diagram illustrating an example of a parameter table according to the second embodiment. FIGS. 16A and 16B are graphs showing the waveform of an acoustic piano. FIGS. 17A and 17B are graphs showing spectra calculated from the waveforms of FIGS. 16A and 16B. FIG. 18 is a diagram showing a spectrum envelope extracted from the spectra shown in FIGS. 17 (a) and 17 (b). FIG. 19 is a graph showing filter characteristics when fc is changed as a parameter in a conventional feedback secondary filter.

  Embodiments of the present invention will be described below with reference to the accompanying drawings. FIG. 1 is a block diagram showing an outline of an electronic musical instrument according to an embodiment of the present invention. As shown in FIG. 1, the electronic musical instrument includes a microcomputer 1, a ROM (Read Only Memory) 2, a RAM (Random Access Memory) 3, switches 4, a touch detection circuit 5, a keyboard 6, a tone generation circuit 7, and a waveform ROM 8. , A D / A converter (DAC) 9, an amplifier circuit 10, and speakers 11 and 12. The microcomputer 1, the ROM 2, the RAM 3 and the tone generation circuit 7 are connected to the data bus 13, and the touch detection circuit 5 is connected to the microcomputer 1.

  The microcomputer 1 controls the entire electronic musical instrument, reads the program and data from the ROM 2 storing the program and data, and executes the program. Data generated by executing the program is stored in the RAM 3 which is a work area. The switches 4 are arranged on the console panel of the electronic musical instrument. The microcomputer 1 detects the operation of the switches 4 by the performer. The touch detection circuit 5 sends a scanning signal to the keyboard 6 at a predetermined timing, and in response to turning on the two switches arranged on the keys of each keyboard 6, performance operation data (pitch and touch response data). ) And output to the microcomputer 1. In the present embodiment, the keyboard 6 has 88 keys, and each key is provided with two switches in the longitudinal direction. When the key is pressed, the first switch is first turned on. When pressed, the second switch is turned on.

  The microcomputer 1 controls the musical tone generation circuit 7 based on the tone color designated by the operation of the switches 4 and the performance operation data including the touch response data and the pitch output from the touch detection circuit 5. A predetermined musical tone is generated. The musical tone generation circuit 7 reads the waveform data of the tone color designated from the waveform ROM 8, generates a musical tone having a pitch and volume (velocity) according to the performance operation data, and outputs it to the DAC 9. The DAC 9 converts the digital data output from the tone generation circuit 7 into an analog signal. The analog signal is emitted from the speakers 11 and 12 via the amplifier circuit 10.

  FIG. 2 is a block diagram showing the musical tone generating circuit according to the present embodiment in more detail. As shown in FIG. 2, the tone generation circuit 7 includes an interface 20, a waveform generation circuit 21, a filter circuit 22, a multiplication circuit 23, a mixer 24, a first envelope generation circuit 25, a filter coefficient calculation circuit 26, and a second envelope. A generation circuit 27 is provided. The interface 20, waveform generation circuit 21, filter circuit 22, mixer 24, first envelope generation circuit 25, filter coefficient calculation circuit 26 and second envelope generation circuit 27 are connected to the internal bus 28.

  The interface 20 is connected to the data bus 13 shown in FIG. 1, and writes setting data and the like to each arithmetic unit such as the waveform generation circuit 21 and the filter circuit 22 in the musical tone generation circuit 7 via the internal bus 28. The waveform generation circuit 21 is connected to the waveform ROM 8, reads PCM data from a predetermined address of the waveform ROM 8, and generates musical tone waveform data having a frequency corresponding to the pitch in the performance operation data.

  The filter coefficient circuit 26 is a filter coefficient according to a parameter based on a performance operation signal given from the microcomputer 1 via the interface 20 and a first envelope signal output from the first envelope generation circuit 25 and time-varying. Is calculated. The frequency characteristic is changed by the first envelope signal, and the filter coefficient changes with time. The filter circuit 22 performs a filter process on the musical sound waveform data according to the filter coefficient.

  The multiplier 23 multiplies the musical sound waveform data and the second envelope signal that is output from the second envelope generation circuit 27 and changes with time. Thereby, the rising and falling of the musical sound according to the on / off of the key of the keyboard 6 and the volume according to the touch response data are controlled. Note that the waveform generation circuit 21, the filter circuit 22, the first envelope generation circuit 25, the filter coefficient calculation circuit 26, the multiplier 23, and the second envelope generation circuit 27 perform processing of the maximum simultaneous sound generation time-divisionally. Thus, it is possible to generate musical sounds with a sufficient number of channels for keyboard performance. The mixer 24 accumulates the generated musical tones of the maximum number of simultaneous pronunciations with predetermined weights, respectively, and finally distributes the musical tones of the two left and right channels. The output of the mixer 24 is output to the DAC 9.

  The outline of the filter circuit 22 and the filter coefficient calculation circuit 26 according to the present embodiment will be described below. In the present embodiment, both of the acoustic instrument filter characteristic mainly corresponding to the filter characteristic suitable for the timbre change of the natural musical instrument and the synthesizer filter characteristic mainly corresponding to the filter characteristic suitable for the timbre change of the so-called synthesizer. Filter characteristics can be provided.

  The acoustic instrument filter characteristics will be described below. FIG. 3 is a diagram for explaining ideal filter characteristics for realizing a timbre filter of an acoustic piano depending on performance strength. In FIG. 3, the vertical axis represents gain, and the horizontal axis represents linear frequency.

  As shown in FIG. 3, the ideal filter characteristics are inferred from the spectrum envelope of the acoustic piano shown in FIG. 17, and the harmonic component of the fundamental wave is almost linear from a certain point based on the fundamental wave. It is like decaying. In addition, the inclination is small when the volume is small (that is, the degree of right-down is large: the inclination is small), and increases as the volume increases (that is, the degree of right-down is small and becomes more horizontal). : Increasing the inclination).

  Hereinafter, a point (frequency) at which linear attenuation or enhancement starts based on the fundamental wave is set as a reference frequency, an attenuation (dB) at a frequency half of the sampling frequency fs is set as a predetermined ratio with respect to gain and gain. Such a frequency (in this embodiment, a frequency that is ½ of the attenuation at fs / 2) is referred to as a transition frequency. The degree of linear attenuation or enhancement from the reference frequency is referred to as inclination. For the slope, the frequency axis is the x-axis and the attenuation is the y-axis. Therefore, the larger the negative attenuation amount, the smaller the slope in the frequency characteristic. The closer the negative attenuation amount is to 0, the larger the slope becomes and approaches 0. The greater the positive current guess, the greater the slope in the frequency characteristic.

  In the present embodiment, the reference frequency changes as the fundamental wave changes. In addition, the transition frequency changes according to a certain rule as the reference frequency changes.

  In this embodiment, as will be described later, the filter circuit is controlled by a transition frequency that changes based on the fundamental wave and a gain that indicates the depth of the filter. This realizes control of harmonic components with respect to the fundamental wave, which is almost the same as the actual timbre change of an acoustic piano.

  In the present embodiment, it is considered that not only the harmonic components are attenuated but also enhanced. This is to change the timbre from the PCM data of the waveform ROM 8 over a wider range. An FIR (Finite Impulse Response) filter is suitable for realizing such characteristics. However, the FIR filter has a large calculation amount and requires large-scale hardware. Therefore, in the present embodiment, a characteristic approximate to the ideal filter characteristic shown in FIG. 3 is realized by a low-order IIR (Infinite Impulse Resonance) filter, and the filter coefficient is expressed as the ideal filter characteristic. It is controlled so that it is almost the same.

  In order to obtain a filter having the above filter characteristics, various methods are conceivable. In order to minimize the amount of calculation, an approximation with a first-order IIR filter is attempted, and the ideal is obtained by extending the approximation. A frequency specification close to is obtained. First, the transfer function of the first-order IIR filter is considered as equation (2).

（２）式において、最大ゲイン（つまり、サンプリング周波数ｆｓの１／２の周波数での最大ゲインをＡ）とすると、Ａは、（３）式のように表すことができる。

In equation (2), if the maximum gain (that is, the maximum gain at a frequency that is ½ of the sampling frequency fs) is A, A can be expressed as equation (3).

また、遷移周波数ｆ０のときに、最大ゲインの１／２のゲインとなるとすると、（４）式が導き出される。

If the gain is ½ of the maximum gain at the transition frequency f0, the equation (4) is derived.

上記（３）式および（４）式を解くと、係数ｂ、ｃは（５）式および（６）式に示すようになる。

When the above equations (3) and (4) are solved, the coefficients b and c are as shown in equations (5) and (6).

したがって、（２）式に示す一次ＩＩＲフィルタの係数ｂ、ｃは、最大ゲインＡおよび遷移周波数ｆ０とから、（５）式および（６）式に示すように求めることができる。なお、上記（５）式からフィルタ係数には２通りのセットが存在するが、その係数範囲から制御しやすい方を選択すればよい。

Therefore, the coefficients b and c of the first-order IIR filter shown in the equation (2) can be obtained from the maximum gain A and the transition frequency f0 as shown in the equations (5) and (6). Although there are two sets of filter coefficients from the above equation (5), it is only necessary to select one that is easier to control from the coefficient range.

  The first-order IIR filter has too slow attenuation characteristics. Therefore, in the present embodiment, two primary filters IIR having different transition frequencies are connected in series to form a secondary IIR filter, thereby obtaining a filter characteristic more approximate to an ideal filter. Equation (7) is a diagram showing a transfer function of a secondary IIR filter in which primary IIR filters are connected in series.

図４は、ある遷移周波数で、ゲインを変更したときの二次ＩＩＲフィルタの特性を示す。図４に示す特性は、図３に示す理想的な特性に近似していることが理解できる。なお、本実施の形態においては、二次ＩＩＲフィルタにより理想フィルタを近似しているが、フィルタの次数を上げることにより、その特性を、さらに理想フィルタの特性と近似させることが可能である。

FIG. 4 shows the characteristics of the second-order IIR filter when the gain is changed at a certain transition frequency. It can be understood that the characteristics shown in FIG. 4 approximate the ideal characteristics shown in FIG. In the present embodiment, the ideal filter is approximated by a secondary IIR filter. However, by increasing the order of the filter, the characteristic can be further approximated to the characteristic of the ideal filter.

  FIG. 5 is a graph showing second-order filter coefficients for realizing the filter characteristics shown in FIG. The horizontal axis represents the negative gain level. Level 1 represents a gain having the largest negative level (that is, the slope is minimum), and level 0 represents a gain having the smallest negative level (that is, the slope is 0). Thus, in the present embodiment, the filter coefficients b, c, d, e, and f are determined at levels 1 to 9, respectively.

  The filter coefficient for the positive gain can be calculated by exchanging the denominator and numerator of the above equation (7) (see equation (8)).

したがって、負のゲインの係数のｂ〜ｆをそれぞれ、
ｂ：ｅ／ｄ、ｃ：ｆ／ｄ、ｄ：１／ｄ、ｅ：ｂ／ｄ、ｆ：ｃ／ｄ
と入れ替えることにより、正のゲイン特性のフィルタ係数を得ることができる。

Therefore, the negative gain coefficients b to f are
b: e / d, c: f / d, d: 1 / d, e: b / d, f: c / d
Can be obtained to obtain a filter coefficient having a positive gain characteristic.

  In this embodiment, this filter coefficient is not calculated by calculation, but a table (parameter table) with transition frequency and gain as addresses is provided in the filter coefficient calculation circuit 26, and a data value is extracted from the table. An appropriate filter coefficient is calculated by interpolating the data value.

  FIG. 6 is a diagram for explaining a parameter table of acoustic filter characteristics and acoustic filter characteristics in the present embodiment. In the acoustic filter characteristics, the transition frequency is the first parameter, and the gain level is the second parameter.

FIG. 6A is a diagram illustrating a filter characteristic at the transition frequency F ₁ based on a certain reference frequency f ₁ . As described above, the reference frequency f ₁ is determined according to the frequency of the fundamental wave. For example, the reference frequency may be substantially the same as the fundamental wave, or may be a frequency that is a predetermined multiple of the fundamental wave. As shown in FIG. 6 (a), the single transition frequencies F _1, is considered a filter characteristic having a plurality of tilt. In the example of FIG. _6A , gain levels G ₁₁ , G ₁₂ ,... G _1i (= attenuation “0”) from the smallest negative characteristic (attenuating characteristic), and G ₁ as the positive characteristic. _{(I + 1)} ... G1 _(2i-1) is shown. Further, considering the case where the reference frequency is changed from f ₁ to f ₂ (f ₁ <f ₂ ) due to the difference in the fundamental wave, as shown in FIG. 6B, the reference frequency is changed. Te, the transition frequency _{F 2} is changed. In addition, gain levels G ₂₁ , G ₂₂ ,..., G _2i ,..., G _{2 (2i−1)} can also be changed.

For example, in FIG. 6 (a), the the single transition frequencies _{F 1,} 2i-1 single level exists. Therefore, in this example, there is one set of filter coefficients (b, c, d, e, f) for a combination of the transition frequency F _{1 and a} certain level. That is, if there are 2i-1 levels, there are 2i-1 filter coefficient pairs. The same applies to FIG. 6B. Therefore, if there are n types of transition frequencies, there may be n × (number of levels ((2i−1) in the example of FIG. 6)) filter coefficient sets.

  If a set of filter coefficients for each gain is held for each transition frequency, it is necessary to prepare a table of enormous size (number of transition frequency settings × number of gain settings × type of filter coefficient (5)). Therefore, by limiting the number of transition frequency settings and the number of gain settings, prepare only a table that stores a series of filter coefficients according to these combinations, and interpolate both the transition frequency and gain (two-dimensional interpolation). By realizing the above, the size of the table is reduced and the filter coefficient is optimized.

  Next, synthesizer filter characteristics will be described. In the synthesizer filter characteristics, the bandwidth Q is the first parameter and the cut-off fc is the second parameter, as in the conventional synthesizer filter.

  FIG. 7 shows the respective filter characteristics (frequency characteristics) when the cut-off fc is exponentially changed to 3000 Hz, 1765 Hz, 1038 Hz, 611 Hz, 359 Hz, 211 Hz, 124 Hz, 73 Hz, and 43 Hz at Q = 2. It is a graph to show. FIG. 8 is a graph showing filter coefficients b to d in each state of the cutoff fc. Note that the transfer function of the conventional second-order IIR filter is expressed by equation (9).

H (z) = f ² / (1 + (− 2 + f ² + f / Q) z ⁻¹ + (1−f / Q) z ⁻² )
... (9)
Therefore, the filter coefficients e and f are fixed at e = 0 and f = 0.

  Hereinafter, the configurations of the filter coefficient calculation circuit 26 and the filter circuit 22 according to the present embodiment will be described in more detail.

  FIG. 9 is a block diagram showing the configuration of the filter coefficient calculation circuit according to this embodiment. As shown in FIG. 9, the filter coefficient calculation circuit 26 includes an interface 31, adders 32 to 34, a parameter table 35, an interpolation circuit 36, and registers 37 to 41. Although not shown in FIG. 9, the filter coefficient calculation circuit 26 outputs the address signal increment signals X1 and Y1 to the parameter table 35 and the selection signal SEL of the parameter table 35 at a predetermined timing. It has a circuit.

  The interface 31 is connected to the internal bus 28 of the tone generation circuit 7, receives and holds two types of parameters transmitted from the microcomputer 1 via the interface 20, and outputs them to the filter calculation circuit 26. In the present embodiment, a set of parameters including transition frequency data F [15: 0], which is a first parameter related to acoustic instrument filter characteristics, and gain data G [15: 0], which is a second parameter. And a set of parameters including the bandwidth data Q [15: 0] as the first parameter relating to the filter characteristics for the synthesizer and the cut-off data fc [15: 0] as the second parameter. 1 to the interface.

  In the present embodiment, the interface 31 receives the filter mode selection signal A / S for selecting the acoustic instrument filter characteristic or the synthesizer filter characteristic from the microcomputer 1, and receives the filter mode selection signal A / S as Output to the parameter table 35. In the present embodiment, when the filter mode selection signal A / S is “0”, the filter coefficient calculation circuit 26 operates based on the acoustic instrument filter characteristics. Therefore, the transition frequency data F is given from the microcomputer 1 to the interface 31 as the first parameter P1, and the gain data G is given as the second parameter P2.

  On the other hand, when the filter mode selection signal A / S is “1”, the filter coefficient calculation circuit 26 operates based on the synthesizer filter characteristics. Therefore, the bandwidth data Q is given from the microcomputer 1 to the interface 31 as the first parameter P1, and the cut-off data fc is given as the second parameter P2.

  Further, in the present embodiment, the interface 31 receives data, addresses and various control signals for rewriting the parameter table 35 from the microcomputer 1 and outputs them to the parameter table 35. In FIG. 9, WRB is a write enable signal for the parameter table, and S is an address switching signal. In FIG. 9, D is the data for writing in the parameter table 35, and AD is the address for writing in the parameter table 35.

  As shown in FIG. 9, the second parameter P <b> 2 [15: 0] is given to the adder 32. The adder 32 is supplied with the first envelope signal that changes over time from the first envelope generation circuit 25 as the other input. Therefore, the value of the second parameter P2 [15: 0] output from the adder changes with time based on the first envelope signal.

  Among the second parameters P2 [15: 0], P2 [15:12] is given to the adder 33. Further, P1 [15:13] of the first parameter P1 [15: 0] is given to the adder 34. In the adders 33 and 34, the increment signal “1” is added at a predetermined timing as necessary, and is output as an address of the parameter table 35. The adder 33 can output a continuous address (P2 [15:12] +1) by adding the increment signal X1 at the timing next to the output of P2 [15:12]. The adder 34 can also output a continuous address (P1 [15:13] +1) by adding the increment signal Y1 at the timing next to the output of P1 [15:13].

  A signal from the adder 33 is given to the lower address A [4: 0] of the parameter table 35, and a signal from the adder 34 is given to the next higher address A [8: 5]. Further, a selection signal SEL from a control circuit (not shown) is given to the highest address A [11: 9].

  FIG. 10 is a diagram illustrating an example of a parameter table according to the present embodiment. As shown in FIG. 10, the parameter table 35 includes a selector 101, OR circuits 102 and 104, an inverter 103, memories 105 and 106, and a selector 107. In the present embodiment, the memory 105 (RAMA) stores file coefficients based on the first parameter (transition frequency data) and the second parameter (gain data) of the acoustic filter characteristics. The memory 106 (RAMB) stores filter coefficients based on the first parameter (bandwidth data) and the second parameter (cutoff data) of the synthesizer filter characteristics.

  In the present embodiment, for example, the memory 105 includes 9 types of transition frequencies × 17 types of gains (8 types of negative gain, gain “0”, and 8 types of positive gains) × 5 types of filters. Store the coefficients. Therefore, the memory 105 stores the number of filters of 9 × 17 × 5 = 765 words.

  Similarly, the memory 105 stores five types of filter coefficients for each of nine types of bandwidth × 8 types of cutoffs.

  For example, for the transition frequency, nine types of addresses are prepared in order to interpolate the upper 3 bits (8 types) with the lower 13 bits. As for the gain, 17 kinds of addresses are prepared in order to complement the upper 4 bits (16 kinds) with the lower 12 bits. The selection signal SEL is used to select one of the five types of coefficients b to f.

  As shown in FIG. 10, the WRB signal and the A / S signal are supplied to the OR circuit 102, and the output of the OR circuit 102 is input as a write enable of the memory 105. On the other hand, the WRB signal and the inverted signal of the A / S signal by the inverter 103 are supplied to the OR circuit 104, and the output of the OR circuit 104 is input as a write enable of the memory 106. This write enable makes the memory writable when “0” is input.

  The selector 101 selects the write address AD [11: 0] given from the interface 31 when the S signal is “0”, and the parameter given from the interface 31 when the S signal is “1”. Select A [11: 0].

  When writing data to the memories 105 and 106, the microcomputer 1 outputs predetermined control signals (WRB signal, A / S signal, S signal) to the interface 31, and outputs an address AD and data D.

  Since the WRB signal “0”, the A / S signal “0”, and the S signal “0” are given from the microcomputer 1 via the interface 31, they are given from the microcomputer 1 via the interface 31 in the memory 105. The data D [19: 0], which is a filter coefficient based on the acoustic filter characteristics, is written to the write address AD [11: 0].

  Further, the WRB signal “0”, the A / S signal “1”, and the S signal “0” are given from the microcomputer 1 via the interface 31, so that in the memory 106 from the microcomputer 1 via the interface 31. Data D [19: 0], which is a filter coefficient based on the synthesizer filter characteristics, is written to the given write address AD [11: 0].

  When the WRB signal is “1”, the memories 105 and 106 are for reading. The microcomputer 1 reads the filter coefficient from the memory 105 if the timbre specified by the switch 4 belongs to the timbre of the acoustic instrument, and from the memory 106 if it belongs to the so-called synthesizer timbre. A control signal and a parameter are output to the parameter table 35 so as to read out the filter coefficient.

  When the WRB signal “1”, the A / S signal “0”, and the S signal “1” are given from the microcomputer 1 via the interface 31, the memory 105 outputs the SEL signal as A [11: 9], Data is read with the predetermined upper address P1 [15:13] of the first parameter as A [8: 5] and the predetermined upper address P2 [15:12] of the second parameter as A [4: 0]. . Since the A / S signal “0” is also given to the selector 107, the output from the memory 105 is selected. Therefore, the filter coefficient before interpolation based on the transition frequency and the gain is output from the parameter table 35.

  When the WRB signal “1”, the A / S signal “1”, and the S signal “1” are given from the microcomputer 1 via the interface 31, the memory 106 transmits the SEL signal as A [11: 9], Data is read with the predetermined upper address P1 [15:13] of the first parameter as A [8: 5] and the predetermined upper address P2 [15:12] of the second parameter as A [4: 0]. . Since the selector 107 is also given the A / S signal “1”, the output from the memory 106 is selected. Therefore, the filter coefficient before interpolation based on the bandwidth and the cutoff is output from the parameter table 35.

  Hereinafter, the interpolation circuit 36 according to the present embodiment will be described. For convenience of explanation, the case where the filter coefficient before interpolation based on the transition frequency and the gain is output from the parameter table 35 will be described. However, it goes without saying that the same interpolation calculation is also performed when filter coefficients before interpolation based on the bandwidth and the cutoff are output.

  The output of the parameter table 35, that is, the set of filter coefficients before interpolation is given to the interpolation circuit 36. Further, the lower data F [12: 0] of transition frequency data (P1 [12: 0] in FIG. 9) and the lower data G [11: 0] of gain data (P2 [12: 0] in FIG. 9) are also interpolated. The circuit 36 is provided. In the interpolation circuit 36, five types of post-interpolation filter coefficients b to f are calculated and output. The registers 37 to 41 hold the filter coefficients b to f after interpolation that are sequentially output, respectively.

  FIG. 11 is a block diagram showing the interpolation circuit according to this embodiment in more detail. As shown in FIG. 11, the interpolation circuit 36 includes registers 51 to 54, a subtractor 55, a multiplier 56, an adder 57, a register 58, a subtractor 65, a multiplier 66, an adder 67, a register 68, and a subtractor 70. , A multiplier 71, and an adder 72.

  The subtractor 55 calculates a difference value obtained by subtracting the output of the register 51 from the output of the register 51, and the difference value and the lower data G [11: 0] (P2 [11: 0]) of the gain data are obtained. Multiplication is performed in the multiplier 56, and the multiplied value is output to the adder 57. The adder 57 adds the multiplication value and the output from the register 51, and the added value is stored in the register 58.

  Similarly, the subtractor 65 calculates a difference value obtained by subtracting the output of the register 54 from the output of the register 53, and this difference value and the lower data G [11: 0] (P2 [11: 0] of the gain data). ) Are multiplied by the multiplier 66, and the multiplied value is output to the adder 67. The adder 67 adds the multiplication value and the output from the register 53, and the added value is stored in the register 68.

  Further, in the subtractor 70, a difference value obtained by subtracting the output of the register 68 from the output of the register 58 is calculated, and this difference value and the lower-order data F [12: 0] (P1 [11: 0] of the transition frequency data) ) And the multiplication value is output to the adder 72. The adder 72 adds the multiplication value and the output from the register 58. The output from the adder 72 becomes the interpolated filter coefficient.

  FIG. 12 is a block diagram showing an outline of the filter circuit according to the present embodiment. As shown in FIG. 12, the filter circuit 22 includes adders 80 and 88, multipliers 81, 82, 85, 86 and 87, and delay circuits 83 and 84. Filter coefficients b, c, d, e, and f calculated by the filter coefficient calculation circuit 22 are given to the multipliers 81, 82, 85, 86, and 87, respectively, and applied to respective inputs of the multipliers. The signal and the filter coefficient are multiplied.

  Hereinafter, two-dimensional interpolation according to the present embodiment will be described. FIG. 13 is a diagram for explaining two-dimensional interpolation according to the present embodiment. For convenience of explanation, it is assumed that x = G [15:12], dx = G [11: 0], y = F [15:13], and dy = F [12: 0]. The following values are stored in the registers 51 to 54 of FIG.

Register 51: o [x, y] (where o [x, y] represents the output of the parameter table 35 at the address xy)
Register 52: o [x + 1, y]
Register 53: o [x, y + 1]
Register 54: o [x + 1, y + 1]
The subtracter 55 subtracts the output o [x, y] of the register 51 from the output o [x + 1, y] of the register 52, and the multiplier 56 multiplies the subtracted value by dx. Thereafter, the adder 57 adds the multiplication value and o [x, y] and stores them in the register 58. Therefore, the register 58 stores the following values.

Register 58: o [x, y] + (o [x + 1, y] −o [x, y]) * dx
= O [x + dx, y]
As illustrated in FIG. 13, this corresponds to linear interpolation between o [x, y] and o [x + 1, y] based on dx.

  The subtracter 65 subtracts the output o [x, y + 1] of the register 53 from the output o [x + 1, y + 1] of the register 54, and the multiplier 66 multiplies the subtracted value by dx. Thereafter, the adder 67 adds the multiplication value and o [x, y + 1] and stores them in the register 58. Therefore, the register 68 stores the following values.

Register 68: o [x, y + 1] +
(O [x + 1, y + 1] −o [x, y + 1]) * dx
= O [x + dx, y + 1]
As shown in FIG. 13, this corresponds to linear interpolation between o [x, y + 1] and o [x + 1, y + 1] based on dx.

  Further, the subtracter 70 subtracts the output o [x + dx, y] of the register 58 from the output o [x + dx, y + 1] of the register 68, and the multiplier 71 multiplies the subtracted value by dy. Thereafter, the adder 72 adds the multiplication value and o [x + dx, y] and outputs the result. Therefore, the output value is as follows.

Output value: o [x + dx, y] +
(O [x + dx, y + 1] −o [x + dx, y]) * dy
= O [x + dx, y + dy]
As shown in FIG. 13, the output value corresponds to an output value obtained by interpolating between o [x + dx, y] and o [x + dx, y + 1] based on dy. Therefore, the obtained output value is a two-dimensional interpolation value based on dx and dy.

  In this embodiment, by sequentially changing the selection signal SEL and outputting the pre-interpolated values of the filter coefficients b, c, d, e, and f from the parameter table 35, the interpolation circuit 36 Interpolated values of c, d, e, and f are generated and output.

  FIG. 14 is a timing chart for calculating the filter coefficient. 14, R0 to R5 correspond to the registers 51 to 54, 58, and 68 in FIG. 11, respectively, and B to F correspond to the registers 37 to 41 in FIG. 9, respectively.

  In FIG. 14, the address A [4: 0] is switched to “x” or “x + 1” from the signal X1, and the address A [8: 5] is changed to “y” or “y + 1” by the signal Y1. Can be switched to. Further, the address A [9:11] is changed by the selection signal SEL. In one cycle of the first Y1 (see reference numeral 1401), the selection signal SEL indicates “b”. Therefore, the filter coefficient b is determined by the four types of addresses given to the parameter table 35 in the cycle of the first Y1. O [x, y], o [x + 1, y], o [x, y + 1] and o [x + 1, y + 1] are output and stored in R0 to R3.

  Based on o [x, y] and o [x + 1, y] stored in R0 and R1, respectively, o [x + dx, y] is calculated and stored in R4. Further, o [x + dx, y + 1] is calculated based on o [x, y + 1] and o [x + 1, y + 1] stored in R2 and R3, respectively, and stored in R5. Based on o [x + dx, y] and o [x + dx, y + 1] stored in R4 and R5, an interpolated value b [x + dx, y + dy] for the filter coefficient b is calculated and stored in B. .

  Similarly, when the selection signal SEL indicates “c”, “d”, “e”, and “f”, finally, the interpolated values c [x + dx, y + dy] of the filter coefficients c to f are finally obtained. ], D [x + dx, y + dy], e [x + dx, y + dy] and f [x + dx, y + dy] are calculated, and the calculated values are stored in C to F, respectively.

  According to the present embodiment, the first memory that outputs the filter coefficient based on the transition frequency and the gain, and the second memory that outputs the filter coefficient based on the bandwidth and the cutoff are provided and designated. Filter coefficients are output from one of the memories according to the tone color.

  The transition frequency is based on the frequency of the fundamental wave of the musical sound to be generated, and the gain indicates the degree of attenuation (or the degree of enhancement). Therefore, it is possible to realize a digital filter having a filter characteristic very close to the filter characteristic of an acoustic instrument, particularly an acoustic piano, with parameters that are easy to understand intuitively and without performing complicated calculations. It is possible to output a musical tone with an equivalent tone.

  On the other hand, a digital filter having a conventional filter characteristic is realized by outputting a filter coefficient based on the conventional bandwidth and cut-off using the second memory, and outputs a musical tone tone unique to a so-called synthesizer. It is also possible.

  Further, according to the present embodiment, the envelope data that changes with time is output from the first envelope generation circuit, and this envelope data is added to the gain data or the cut-off by the filter coefficient calculation circuit. Accordingly, it is possible to realize a desired timbre change by realizing a time change of gain attenuation (or enhancement) by a filter or a time change of attenuation by a cut-off change with a simple circuit configuration. Become.

  Furthermore, according to the present embodiment, the upper bits of the first parameter (transition frequency data or bandwidth data) and the upper bits of the second parameter (gain data or cut-off data) are used as an address, and the parameter table Outputs a coarse filter coefficient, and two-dimensionally corrects the coarse filter coefficient with the low-order bit of the first parameter and the low-order bit of the second parameter to obtain a corrected filter coefficient. As a result, it is possible to obtain an appropriate filter coefficient with high accuracy without increasing the number of data (coefficients) stored in the parameter table.

  Next, a second embodiment of the present invention will be described. In the first embodiment, two memories (RAMA and RAMB) are provided, filter coefficients based on the acoustic instrument filter characteristics are stored in one memory (RAMA), and the filter is performed using transition frequency data and gain data as addresses. The coefficient is output, the filter coefficient based on the synthesizer filter characteristics is stored in the other memory (RAMB), and the filter coefficient is output using the bandwidth data and the cut-off data as addresses. Of course, this does not have to be physically two memories, it divides the address space of a single memory into two, one storing the filter coefficients based on acoustic instrument filter characteristics as RAMA and the other The case where the filter coefficient based on the filter characteristic for synthesizers is stored as RAMB is also included.

  On the other hand, in the second embodiment, a filter coefficient based on the acoustic instrument filter characteristics or the synthesizer filter characteristics is controlled by the microcomputer 1 using a single rewritable memory. Store any of the filter coefficients based on it.

  FIG. 15 is a block diagram showing an example of a parameter table according to the embodiment of the present invention. As illustrated in FIG. 15, the parameter table includes a selector 111 and a memory 112. A WRB signal and an S signal are given as control signals from the interface 31 to the parameter table. Since the memory does not need to be switched as in the first embodiment, the A / S signal is omitted. Other addresses and data are the same as those in the first embodiment.

  When writing data to the memory 112, the microcomputer 1 outputs predetermined control signals (WRB signal, S signal) to the interface 31 and outputs addresses AD and D.

  Since the WRB signal “0” and the S signal “0” are given from the microcomputer 1 via the interface 31, the write address AD [11: given by the microcomputer 1 via the interface 31 is given in the memory 112. 0] is written with data D [19: 0] corresponding to the filter coefficient.

  For example, if the tone color designated by the switch 4 belongs to the tone color of the acoustic instrument, a filter coefficient based on the acoustic instrument filter characteristics is written in the memory 112 as data D. After that, the microcomputer 1 gives the WRB signal “1” and the S signal “1” via the interface 31, whereby the memory 112 sends the address A [11: 0] based on the transition frequency data and the gain data. Corresponding filter coefficients are output.

  If the tone color designated by the switch 4 belongs to a so-called synthesizer tone color, a filter coefficient based on the synthesizer filter characteristics is written in the memory 112 as data D. Thereafter, the microcomputer 1 gives the WRB signal “1” and the S signal “1” via the interface 31, whereby the address A [11: 0] based on the bandwidth data and the cutoff data is received from the memory 112. The filter coefficient corresponding to is output.

  The present invention is not limited to the above embodiments, and various modifications can be made within the scope of the invention described in the claims, and these are also included in the scope of the present invention. Needless to say.

1 Microcomputer 2 ROM
3 RAM
4 Switches 5 Touch detection circuit 6 Keyboard 7 Musical sound generation circuit 8 Waveform ROM
9 DAC
DESCRIPTION OF SYMBOLS 10 Amplifier circuit 11, 12 Speaker 20 Interface 21 Waveform generation circuit 22 Filter circuit 23 Multiplier 24 Mixer 25 1st envelope generation circuit 26 Filter coefficient calculation circuit 27 2nd envelope generation circuit 35 Parameter table 36 Interpolation circuit

Claims (5)

Filter coefficient output means for outputting the filter coefficients b and c;
In order to perform a filtering process based on the filter characteristics defined by the filter coefficients b and c output from the filter coefficient output means on a musical sound signal having a predetermined frequency supplied from the outside, the following transfer function A filter device comprising H (z),

The filter coefficient output means comprises:
Assuming that the maximum gain at 1/2 of the sampling frequency fs is A, when the frequency at which the gain is 1/2 of the maximum gain is f0, the filter coefficients b and c are

A filter device characterized in that the filter device calculates and outputs to the filter means.   2. The filter device according to claim 1, wherein a maximum gain A and a frequency f0 are determined corresponding to the frequency and intensity of the supplied musical sound signal.   The filter device according to claim 1, wherein the filter coefficient output unit further includes an envelope generation unit that changes the maximum gain A over time. Pronunciation instruction means for instructing the frequency and intensity of a musical tone to be generated by an external operation;
Control means for generating a first parameter based on the frequency and calculating a second parameter based on the intensity;
Filter coefficient generation means for extracting from the parameter table a set of filter coefficients corresponding to the frequency and level determined based on the first and second parameters calculated by the control means;
Filter means for performing a filtering process based on a filter characteristic defined by a set of filter coefficients from the filter coefficient generation means and outputting the musical sound signal supplied ;
In an electronic musical instrument with
The filter means has a transfer function H (z) where the filter coefficients are b and c.

And
The filter coefficient generation means uses the second parameter as the maximum gain at 1/2 of the sampling frequency fs as A, and the frequency f0 at which the gain as 1/2 as the maximum gain as the first parameter. The filter coefficients b and c are

An electronic musical instrument characterized in that the electronic musical instrument is calculated and output to the filter means.   5. The electronic musical instrument according to claim 4, wherein the filter coefficient generation unit further includes an envelope generation unit that changes the maximum gain A with time.

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