JP5512458B2 - Precoding method, precoding device, transmission method, transmission device, reception method, and reception device - Google Patents

Description

  The present invention relates to a precoding method, a precoding device, a transmission method, a transmission device, a reception method, and a reception device that perform communication using a multi-antenna in particular.

  Conventionally, as a communication method using multiple antennas, for example, there is a communication method called MIMO (Multiple-Input Multiple-Output). In multi-antenna communication typified by MIMO, data transmission speed is increased by modulating transmission data of a plurality of sequences and transmitting each modulated signal simultaneously from different antennas.

  FIG. 28 shows an example of the configuration of the transmission / reception apparatus when the number of transmission antennas is 2, the number of reception antennas is 2, and the number of transmission modulation signals (transmission streams) is 2. In the transmission apparatus, the encoded data is interleaved, the interleaved data is modulated, frequency conversion or the like is performed to generate a transmission signal, and the transmission signal is transmitted from the antenna. At this time, a scheme in which different modulation signals are transmitted from the transmission antenna to the same frequency at the same time is the spatial multiplexing MIMO scheme.

At this time, Patent Document 1 proposes a transmission apparatus having a different interleave pattern for each transmission antenna. That is, in the transmission apparatus of FIG. 28, two interleaves (πa, πb) have different interleave patterns. In the receiving apparatus, as shown in Non-Patent Document 1 and Non-Patent Document 2, reception quality is improved by repeatedly performing a detection method using a soft value (MIMO detector in FIG. 28). Will do.
By the way, there are an NLOS (non-line of sight) environment represented by a Rayleigh fading environment and an LOS (line of sight) environment represented by a rice fading environment as models of an actual propagation environment in wireless communication. In the case of transmitting a single modulated signal in a transmitting device, performing maximum ratio combining on signals received by a plurality of antennas in a receiving device, and performing demodulation and decoding on the signal after maximum ratio combining, a LOS environment, In particular, good reception quality can be obtained in an environment where the rice factor indicating the magnitude of the direct reception power relative to the reception power of the scattered wave is large. However, for example, in the spatial multiplexing MIMO transmission system, there is a problem that reception quality deteriorates when the rice factor increases. (See Non-Patent Document 3)
29A and 29B show 2 × 2 LDPC (low-density parity-check) encoded data in a Rayleigh fading environment and a rice fading environment with Rise factors K = 3, 10, and 16 dB. (2 antenna transmission, 2 antenna reception) An example of a simulation result of BER (Bit Error Rate) characteristics (vertical axis: BER, horizontal axis: signal-to-noise power ratio (SNR)) when spatial multiplexing MIMO transmission is performed ing. FIG. 29A shows the BER characteristics of Max-log-APP (see Non-Patent Document 1 and Non-Patent Document 2) (APP: a posteriprovability) without iterative detection, and FIG. The BER characteristic of Max-log-APP (refer nonpatent literature 1 and nonpatent literature 2) (number of repetitions 5 times) which performed the detection is shown. As can be seen from FIGS. 29A and 29B, regardless of whether or not iterative detection is performed, in the spatial multiplexing MIMO system, it can be confirmed that reception quality deteriorates as the rice factor increases. Therefore, the spatial multiplexing MIMO system has a problem inherent to the spatial multiplexing MIMO system, which is not found in a conventional system that transmits a single modulation signal, such as “the reception quality deteriorates when the propagation environment becomes stable”. Recognize.

Broadcasting and multicast communication are services for users who are in line of sight, and the radio wave propagation environment between a receiver and a broadcasting station owned by the user is often a LOS environment. When a spatial multiplexing MIMO system with the above-mentioned problems is used for broadcasting or multicast communication, a phenomenon occurs in which the receiver receives a service due to a deterioration in reception quality although the received electric field strength of radio waves is high. there is a possibility. In other words, in order to use the spatial multiplexing MIMO system in broadcasting or multicast communication, it is desired to develop a MIMO transmission system that can obtain a certain level of reception quality in both NLOS and LOS environments.
Non-Patent Document 8 describes a method of selecting a codebook (precoding matrix) to be used for precoding from feedback information from a communication partner, but as described above, a communication partner, such as broadcast or multicast communication, is described. In the situation where the feedback information from cannot be obtained, there is no description about a method for performing precoding.

  On the other hand, Non-Patent Document 4 describes a method of switching a precoding matrix with time, which can be applied even when there is no feedback information. In this document, it is described that a unitary matrix is used as a matrix used for precoding and that the unitary matrix is randomly switched. However, as to the application method for the degradation of reception quality in the LOS environment described above, It is not described at all, and only switching at random is described. As a matter of course, there is no description about a precoding method for improving degradation of reception quality in a LOS environment and a method for constructing a precoding matrix.

International Publication No. 2005/050885

“Achieving near-capacity on a multiple-antenna channel” IEEE Transaction on communications, vol. 51, no. 3, pp. 389-399, March 2003. “Performance analysis and design optimization of LDPC-coded MIMO OFDM systems” IEEE Trans. Signal Processing. , Vol. 52, no. 2, pp. 348-361, Feb. 2004. “BER performance evaluation in 2 × 2 MIMO spatial multiplexing systems under Rician fading channels,” IEICE Trans. Fundamentals, vol. E91-A, no. 10, pp. 2798-2807, Oct. 2008. “Turbo space-time codes with time varying linear transformations,” IEEE Trans. Wireless communications, vol. 6, no. 2, pp. 486-493, Feb. 2007. “Likelihood function for QR-MLD suite for soft-decision turbo decoding and its performance,” IEICE Trans. Commun. , Vol. E88-B, no. 1, pp. 47-57, Jan. 2004. “Signpost to Shannon limit:“ Parallel concordated (Turbo) coding ”,“ Turbo (iterative) decoding ”and its surroundings” The Institute of Electronics, Information and Communication Engineers, IEICE IT98-51 “Advanced signal processing for PLCs: Wavelet-OFDM,” Proc. of IEEE International symposium on ISPLC 2008, pp. 187-192, 2008. D. J. et al. Love, and R.M. W. heat, Jr. "Limited feedback unitary precoding for spatial multiplexing systems," IEEE Trans. Inf. Theory, vol. 51, no. 8, pp. 2967-1976, Aug. 2005. DVB Document A122, Framing structure, channel coding and modulation for a second generation digital terrestrial broadcasting system, m (DVJ-Tun. L. Vangelista, N.A. Benvenuto, and S.M. Thomasin, “Key technologies for next-generation terrestrial digital television standard DVB-T2,” IEEE Commun. Magazine, vo. 47, no. 10, pp. 146-153, Oct. 2009. T.A. Ohgane, T .; Nishimura, and Y. Ogawa, “Application of space division multiplexing and this performance in a MIMO channel,” IEICE Trans. Commun. , Vo. 88-B, no. 5, pp. 1843-1851, May 2005.

  An object of the present invention is to provide a MIMO system capable of improving reception quality in a LOS environment.

  In order to solve such a problem, a precoding method according to one aspect of the present invention is simultaneously transmitted from a signal based on a plurality of selected modulation schemes represented by an in-phase component and a quadrature component to the same frequency band, respectively. A precoding method for generating a plurality of precoded signals, wherein one precoding weight matrix is selected from among a plurality of precoding weight matrices while switching regularly, and the selected precoding weight matrix is selected. The plurality of precoded signals are generated by multiplying signals based on the plurality of selected modulation schemes.

  In addition, a precoding device according to an aspect of the present invention includes a plurality of precoding signals simultaneously transmitted in the same frequency band from signals based on a plurality of selected modulation schemes represented by in-phase components and quadrature components, respectively. A precoding apparatus for generating a received signal, wherein a precoding weight generation unit that selects one precoding weight matrix from among a plurality of precoding weight matrices and regularly selects the precoding weight matrix; And a weighting / synthesizing unit that generates the plurality of precoded signals by weighting and combining signals based on the plurality of selected modulation schemes.

  In addition, the transmission method according to one aspect of the present invention modulates data to be transmitted based on a selected modulation scheme, thereby causing signals based on a plurality of selected modulation schemes respectively represented by an in-phase component and a quadrature component. And selecting one precoding weight matrix from among a plurality of precoding weight matrices while switching regularly, and multiplying the selected precoding weight matrix by a signal based on the plurality of selected modulation schemes Thus, the plurality of precoded signals are generated, and the plurality of precoded signals are simultaneously transmitted to the same frequency band.

  In addition, the transmission device according to one embodiment of the present invention modulates data to be transmitted based on a selected modulation scheme, thereby causing signals based on a plurality of selected modulation schemes respectively represented by an in-phase component and a quadrature component. A precoding weight generation unit that selects one precoding weight matrix by switching regularly among a plurality of precoding weight matrices, and the plurality of precoding weight matrices according to the selected precoding weight matrix A weighting / synthesizing unit for generating the plurality of precoded signals by weighting and synthesizing signals based on the selected modulation scheme, and a transmitting unit for simultaneously transmitting the plurality of precoded signals to the same frequency band And comprising.

  The reception method according to one aspect of the present invention receives a plurality of precoded signals simultaneously transmitted from the transmission apparatus in the same frequency band, and the plurality of precoded signals are in-phase component and quadrature, respectively. Generated by precoding the signal based on the selected modulation method represented by the component, and the signal based on the selected modulation method is a signal obtained by modulating transmission data based on the selected modulation method. The received signal is demodulated by a demodulation method corresponding to the selected modulation method to reproduce the transmission data, and the plurality of precoded signals are regularly selected from a plurality of precoding weight matrices. A precoding weight matrix selected while switching to a signal based on the plurality of selected modulation schemes. It has been produced.

  In addition, the receiving device which is one embodiment of the present invention receives a plurality of precoded signals transmitted simultaneously from the transmitting device in the same frequency band, and the plurality of precoded signals are in-phase component and quadrature, respectively. Generated by precoding the signal based on the selected modulation method represented by the component, and the signal based on the selected modulation method is a signal obtained by modulating transmission data based on the selected modulation method. A reception unit; and a demodulation unit that demodulates the received signal by a demodulation method according to the selected modulation method and reproduces the transmission data, and the plurality of precoded signals are a plurality of signals. One precoding weight matrix selected by regularly switching from among the precoding weight matrices is selected from the plurality of precoding weight matrices. It is generated by multiplying the signal based on the modulation scheme.

  According to each aspect of the present invention described above, a signal precoded by one precoding weight matrix selected while switching regularly among a plurality of precoding weight matrices is used for precoding. Since the precoding weight matrix is one of a plurality of precoding weight matrices determined in advance, the reception quality in the LOS environment can be improved according to the design of the plurality of precoding weight matrices.

  As described above, according to the present invention, it is possible to provide a precoding method, a precoding device, a transmission method, a reception method, a transmission device, and a reception device that improve degradation of reception quality in the LOS environment. Therefore, it is possible to provide a high-quality service to a user who is in the line of sight.

Example of configuration of transmission / reception device in spatial multiplexing MIMO transmission system Example of frame configuration Example of transmitter configuration when applying precoding weight switching method Example of transmitter configuration when applying precoding weight switching method Frame configuration example Example of precoding weight switching method Example of receiver configuration Configuration example of signal processing unit of receiving apparatus Configuration example of signal processing unit of receiving apparatus Decryption processing method Example of reception status Example of BER characteristics Example of transmitter configuration when applying precoding weight switching method Example of transmitter configuration when applying precoding weight switching method Frame configuration example Frame configuration example Frame configuration example Frame configuration example Frame configuration example Position of poor reception quality Position of poor reception quality Example of frame configuration Example of frame configuration Example of mapping method Example of mapping method Example of weighted composition unit configuration Example of how symbols are rearranged Example of configuration of transmission / reception device in spatial multiplexing MIMO transmission system Example of BER characteristics Example of spatially multiplexed 2x2 MIMO system model Position of reception poor point Position of reception poor point Position of reception poor point Position of reception poor point Position of reception poor point Example of characteristics of minimum distance in complex plane of reception poor point Example of characteristics of minimum distance in complex plane of reception poor point Position of reception poor point Position of reception poor point Example of configuration of transmitting apparatus according to Embodiment 7 Example of frame structure of modulated signal transmitted by transmitting apparatus Position of reception poor point Position of reception poor point Position of reception poor point Position of reception poor point Position of reception poor point Example of frame structure on time-frequency axis Example of frame structure on time-frequency axis Signal processing method Configuration of modulated signal using space-time block code Example of frame configuration details on the time-frequency axis Example of transmission device configuration An example of the configuration of the modulation signal generators # 1 to #M in FIG. The figure which shows the structure of the OFDM system related process part (5207_1 and 5207_2) in FIG. Example of frame configuration details on the time-frequency axis Example of receiver configuration The figure which shows the structure of the OFDM system related process part (5600_X, 5600_Y) in FIG. Example of frame configuration details on the time-frequency axis An example of a broadcasting system Position of reception poor point

Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.
(Embodiment 1)
The transmission method, transmission device, reception method, and reception device of this embodiment will be described in detail.

Prior to this description, an outline of a transmission method and a decoding method in a spatial multiplexing MIMO transmission system which is a conventional system will be described.
The configuration of the N _t × N _r spatial multiplexing MIMO system is shown in FIG. The information vector z is encoded and interleaved. As a result of interleaving, a vector u = (u ₁ ,..., U _Nt ) of encoded bits is obtained. However, u _i = (u _i1 ,..., U _iM ) (M: number of transmission bits per symbol). When the transmission vector s = (s ₁ ,..., S _Nt ) ^T , the transmission signal s _i = map (u _i ) is represented from the transmission antenna #i, and when transmission energy is normalized, E {| s _i | ² } = Es / Nt (E _s : total energy per channel). Then, if the received vector is y = (y ₁ ,..., Y _Nr ) ^T , it is expressed as in equation (1).

At this _{time, H NtNr} channel _{matrix, n = (n 1, ...} , n Nr) T is the noise vector, _{n i} is zero mean, variance sigma ² of i. i. d. Complex Gaussian noise. From the relationship between the transmission symbol and the reception symbol introduced by the receiver, the probability regarding the reception vector can be given by a multidimensional Gaussian distribution as shown in Equation (2).

Here, consider a receiver that performs iterative decoding as shown in FIG. 1 comprising an outer soft-in / soft-out decoder and MIMO detection. The log-likelihood ratio vector (L-value) in FIG. 1 is expressed as in equations (3)-(5).

<Repetitive detection method>
Here, iterative detection of a MIMO signal in an N _t × N _r spatial multiplexing MIMO system will be described.
The log likelihood ratio of x _mn is defined as in Equation (6).

From Bayes' theorem, equation (6) can be expressed as equation (7).

However, U _{mn, ± 1} = {u | u _mn = ± 1}. When approximated by lnΣa _{j to} max ln a _j , Expression (7) can be approximated as Expression (8). Note that the symbol “˜” above represents approximation.

P (u | u _mn ) and ln P (u | u _mn ) in Expression (8) are expressed as follows.

By the way, the logarithmic probability of the equation defined by equation (2) is expressed as in equation (12).

Therefore, from the formulas (7) and (13), the posterior L-value is expressed as follows in MAP or APP (a posteriori probability).

Hereinafter, it is called iterative APP decoding. In addition, from the equations (8) and (12), in the log likelihood ratio (Max-Log APP) based on the Max-Log approximation, the posterior L-value is expressed as follows.

Hereinafter, it is called iterative Max-log APP decoding. The external information required in the iterative decoding system can be obtained by subtracting the prior input from the equation (13) or (14).
<System model>
FIG. 28 shows a basic configuration of a system that leads to the following description. Here, a 2 × 2 spatial multiplexing MIMO system is used, and streams A and B each have an outer encoder, and the two outer encoders are encoders of the same LDPC code (in this case, an LDPC code encoder is used as the outer encoder). However, the error correction code used by the outer encoder is not limited to the LDPC code, and other error correction codes such as a turbo code, a convolutional code, and an LDPC convolutional code are used in the same manner. In addition, the outer encoder is configured to be provided for each transmission antenna, but the configuration is not limited thereto, and there may be a plurality of transmission antennas or a single outer encoder. Has more outer encoders than antennas Even if it is.). In streams A and B, there are interleavers (π _a , π _b ), respectively. Here, the modulation scheme is 2 ^h -QAM (h bits are transmitted in one symbol).
In the receiver, it is assumed that the MIMO signal is iteratively detected (iterative APP (or Max-log APP) decoding). As the decoding of the LDPC code, for example, sum-product decoding is performed.
FIG. 2 shows a frame structure and describes the order of symbols after interleaving. At this time, it is assumed that (i _a , j _a ) and (i _b , j _b ) are expressed as in the following equations.

At this time, i _a , i _b : order of symbols after interleaving, j _a , j _b : bit positions (j _a , j _b = 1,..., H) in modulation scheme, π _a , π _b : stream A and B interleavers, Ω ^a _{ia, ja} , Ω ^b _{ib, jb} : The order of data before interleaving of streams A and B is shown. However, FIG. 2 shows a frame configuration when i _a = i _b .
<Iterative decoding>
Here, the sum-product decoding and the iterative detection algorithm of the MIMO signal used in decoding of the LDPC code in the receiver will be described in detail.

Sum-product decoding Let a binary MxN matrix H = {H _mn } be a parity check matrix of an LDPC code that is to be decoded. Subsets A (m) and B (n) of the set [1, N] = {1, 2,..., N} are defined as follows:

At this time, A (m) means a set of column indexes that are 1 in the m-th row of the check matrix H, and B (n) is a set of row indexes that are 1 in the n-th row of the check matrix H. . The sum-product decoding algorithm is as follows.
Step A · 1 (initialization): Prior-value log ratio β _mn = 0 for all pairs (m, n) satisfying H _mn = 1. A loop variable (number of iterations) is set to l _sum = 1, and the maximum number of loops is set to l _{sum, max} .
Step A · 2 (row processing): logarithm of external values using the following update formula for all pairs (m, n) satisfying H _mn = 1 in the order of m = 1, 2,... Update the ratio α _mn .

At this time, f is a Gallager function. The method for obtaining λ _n will be described in detail later.
Step A · 3 (column processing): logarithm of external values using the following update formula for all pairs (m, n) satisfying H _mn = 1 in the order of n = 1, 2,... Update the ratio β _mn .

Step A · 4 (calculation of log-likelihood ratio): The log-likelihood ratio L _n is obtained as follows for n∈ [1, N].

Step A · 5 (counting the number of iterations): If l _sum <l _{sum, max} , increment l _sum and return to step A · 2. In the case of l _sum = l _{sum, max} , this round of sum-product decoding ends.

The above is one sum-product decoding operation. Thereafter, iterative detection of the MIMO signal is performed. Variable m used in the description of the operation of the aforementioned sum-product _{decoding, n, α mn, β mn} , at λ _{n, L n,} the variables in the stream _{A m a, n a, α} a mana, β a mana, λ _{na, L na,} variables _m b in the stream ^{_{B, n b, α b mbnb}} , β b mbnb, λ nb, shall be represented by _{L nb.}
<Repeated detection of MIMO signal>
Here, a method of obtaining λ _n in the MIMO signal iterative detection will be described in detail.

    From the expression (1), the following expression is established.

From the frame configuration of FIG. 2, the following relational expressions are established from the equations (16) and (17).

At this time, n _a , n _b ε [1, N]. Hereinafter, λ _na , L _na , λ _nb , and L _nb at the iteration number k of the MIMO signal iterative detection are represented as λ _{k, na} , L _{k, na} , λ _{k, nb} , L _{k, nb} , respectively. And

Step B · 1 (initial detection; k = 0): At the time of initial detection, λ _{0, na} , λ _{0, nb} is obtained as follows.
For iterative APP decoding:

For iterative Max-log APP decoding:

However, X = a, b. Then, the number of iterations of MIMO signal iterative detection is set to l _mimo = 0, and the maximum number of iterations is set to l _{mimo, max} .
Step B · 2 (iterative detection; number of iterations k): λ _{k, na} , λ _{k, nb} when the number of iterations is _k is obtained from equations (11), (13)-(15), (16), and (17). 31)-(34). However, (X, Y) = (a, b) (b, a).
For iterative APP decoding:

For iterative Max-log APP decoding:

Step B · 3 (count of iterations, codeword estimation): If l _mimo <l _{mimo, max} , increment l _mimo and return to step B · 2. In the case of l _mimo = l _{mimo, max} , the estimated codeword is stopped as follows.

However, X = a, b.
FIG. 3 is an example of a configuration of transmitting apparatus 300 in the present embodiment. The encoding unit 302A receives the information (data) 301A and the frame configuration signal 313 as input, and the frame configuration signal 313 (the error correction method used by the encoding unit 302A for error correction encoding of data, the encoding rate, the block length, etc.) For example, a convolutional code, an LDPC code, a turbo code, or the like may be used in accordance with a method specified by the frame configuration signal 313. Error correction encoding is performed, and encoded data 303A is output.

The interleaver 304A receives the encoded data 303A and the frame configuration signal 313, performs interleaving, that is, rearranges the order, and outputs the interleaved data 305A. (The interleaving method may be switched based on the frame configuration signal 313.)
The mapping unit 306A receives the interleaved data 305A and the frame configuration signal 313, and performs QPSK (Quadrature Phase Shift Keying), 16QAM (16 Quadrature Amplitude Modulation), 64QAM (64 Quadrature Amplitude Modulation), and so on. The signal 307A is output. (The modulation method may be switched based on the frame configuration signal 313.)
FIG. 24 shows an example of a mapping method on the IQ plane of the in-phase component I and the quadrature component Q constituting the baseband signal in QPSK modulation. For example, as shown in FIG. 24A, when input data is “00”, I = 1.0 and Q = 1.0 are output. Similarly, when input data is “01”, I = -1.0, Q = 1.0 is output, and so on. FIG. 24B is an example of a mapping method on the IQ plane of QPSK modulation different from that in FIG. 24A. FIG. 24B is different from FIG. 24A in that FIG. The signal point in FIG. 24B can be obtained by rotating the signal point around the origin. Such constellation rotation methods are shown in Non-Patent Document 9 and Non-Patent Document 10, and even if the Cyclic Q Delay shown in Non-Patent Document 9 and Non-Patent Document 10 is applied. Good. As an example different from FIG. 24, FIG. 25 shows signal point arrangement on the IQ plane at 16QAM, and an example corresponding to FIG. 24A is FIG. 25A, and FIG. An example corresponding to is shown in FIG.

  Encoding section 302B receives information (data) 301B and frame configuration signal 313 as input, and includes frame configuration signal 313 (including information such as an error correction method to be used, a coding rate, and a block length). In addition, the error correction method may be switched, for example, error correction coding such as convolutional code, LDPC code, turbo code, etc., and the encoded data. 303B is output.

The interleaver 304B receives the encoded data 303B and the frame configuration signal 313, performs interleaving, that is, rearranges the order, and outputs the interleaved data 305B. (The interleaving method may be switched based on the frame configuration signal 313.)
The mapping unit 306B receives the interleaved data 305B and the frame configuration signal 313, and performs QPSK (Quadrature Phase Shift Keying), 16QAM (16 Quadrature Amplitude Modulation), 64QAM (64 Quadrature Amplitude Modulation), and so on. The signal 307B is output. (The modulation method may be switched based on the frame configuration signal 313.)
The weighted synthesis information generation unit 314 receives the frame configuration signal 313 and outputs information 315 related to the weighting synthesis method based on the frame configuration signal 313. Note that the weighting synthesis method is characterized in that the weighting synthesis method is regularly switched.

  The weighting combining unit 308A receives the baseband signal 307A, the baseband signal 307B, and the information 315 related to the weighting combining method as inputs, and weights and combines the baseband signal 307A and the baseband signal 307B based on the information 315 related to the weighting combining method. The signal 309A after the weighted synthesis is output. Note that. Details of the weighting method will be described later.

  Radio section 310A receives signal 309A after weighted synthesis, performs processing such as quadrature modulation, band limitation, frequency conversion, and amplification, and outputs transmission signal 311A. Transmission signal 511A is output as a radio wave from antenna 312A. The

  The weighting synthesis unit 308B receives the baseband signal 307A, the baseband signal 307B, and the information 315 regarding the weighting synthesis method as inputs, and weights and synthesizes the baseband signal 307A and the baseband signal 307B based on the information 315 regarding the weighting synthesis method. The signal 309B after the weighted synthesis is output.

FIG. 26 shows the configuration of the weighting synthesis unit. The baseband signal 307A is multiplied by w11 (t) to generate w11 (t) s1 (t), and is multiplied by w21 (t) to generate w21 (t) s1 (t). Similarly, the baseband signal 307B is multiplied by w12 (t) to generate w12 (t) s2 (t) and is multiplied by w22 (t) to generate w22 (t) s2 (t). Next, z1 (t) = w11 (t) s1 (t) + w12 (t) s2 (t) and z2 (t) = w21 (t) s1 (t) + w22 (t) s2 (t) are obtained.
Note that. Details of the weighting method will be described later.

  Radio section 310B receives weighted signal 309B as input, performs processing such as quadrature modulation, band limitation, frequency conversion, and amplification, and outputs transmission signal 311B. Transmission signal 511B is output as a radio wave from antenna 312B. The

FIG. 4 shows a configuration example of a transmission apparatus 400 different from that in FIG. In FIG. 4, a different part from FIG. 3 is demonstrated.
Encoding section 402 receives information (data) 401 and frame configuration signal 313 as input, performs error correction encoding based on frame configuration signal 313, and outputs encoded data 402.

  The distribution unit 404 receives and distributes the encoded data 403, and outputs data 405A and data 405B. In FIG. 4, the case where there is one encoding unit is described. However, the present invention is not limited to this, and the encoding unit is m (m is an integer of 1 or more), and the codes created by each encoding unit The present invention can also be implemented in the same way when the distribution unit outputs divided data into two systems of data.

  FIG. 5 shows an example of a frame configuration on the time axis of the transmission apparatus according to the present embodiment. Symbol 500_1 is a symbol for notifying the receiving apparatus of the transmission method. For example, an error correction method used for transmitting a data symbol, information on its coding rate, and a modulation method used for transmitting a data symbol The information etc. is transmitted.

  Symbol 501_1 is a symbol for estimating channel fluctuation of modulated signal z1 (t) {where t is time} transmitted by the transmission apparatus. Symbol 502_1 is a data symbol transmitted by modulated signal z1 (t) to symbol number u (on the time axis), and symbol 503_1 is a data symbol transmitted by modulated signal z1 (t) to symbol number u + 1.

  Symbol 501_2 is a symbol for estimating channel fluctuation of modulated signal z2 (t) {where t is time} transmitted by the transmission apparatus. Symbol 502_2 is a data symbol transmitted from modulated signal z2 (t) to symbol number u, and symbol 503_2 is a data symbol transmitted from modulated signal z2 (t) to symbol number u + 1.

The relationship between the modulation signal z1 (t) and the modulation signal z2 (t) transmitted by the transmission device and the reception signals r1 (t) and r2 (t) in the reception device will be described.
In FIG. 5, 504 # 1 and 504 # 2 indicate transmission antennas in the transmission apparatus, 505 # 1 and 505 # 2 indicate reception antennas in the reception apparatus, and the transmission apparatus transmits the modulated signal z1 (t) to the transmission antenna 504. # 1, modulated signal z2 (t) is transmitted from transmitting antenna 504 # 2. At this time, it is assumed that the modulation signal z1 (t) and the modulation signal z2 (t) occupy the same (common) frequency (band). The channel fluctuation of each transmission antenna of the transmission device and each antenna of the reception device is set to h11 (t), h12 (t), h21 (t), and h22 (t), respectively, and reception received by the reception antenna 505 # 1 of the reception device. When the signal is r1 (t) and the received signal received by the receiving antenna 505 # 2 of the receiving device is r2 (t), the following relational expression is established.

FIG. 6 is a diagram related to the weighting method (precoding method) in the present embodiment. The weighting synthesis unit 600 is a weighting synthesis unit that integrates both the weighting synthesis units 308A and 308B of FIG. is there. As shown in FIG. 6, the stream s1 (t) and the stream s2 (t) correspond to the baseband signals 307A and 307B of FIG. 3, that is, the base according to the mapping of modulation schemes such as QPSK, 16QAM, and 64QAM. Band signal in-phase I and quadrature Q components. 6, the stream s1 (t) represents the signal with symbol number u as s1 (u), the signal with symbol number u + 1 as s1 (u + 1), and so on. Similarly, in the stream s2 (t), a signal with a symbol number u is represented as s2 (u), a signal with a symbol number u + 1 is represented as s2 (u + 1), and so on. Then, the weighting synthesis unit 600 receives the baseband signals 307A (s1 (t)) and 307B (s2 (t)) in FIG. 3 and the information 315 related to the weighting information, and performs a weighting method according to the information 315 related to the weighting information. Then, the signals 309A (z1 (t)) and 309B (z2 (t)) after the weighted synthesis in FIG. 3 are output. At this time, z1 (t) and z2 (t) are expressed as follows.
For symbol number 4i (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number 4i + 1:

For symbol number 4i + 2:

For symbol number 4i + 3:

As described above, the weighting / synthesizing unit in FIG. 6 regularly switches the precoding weights in a 4-slot period. (However, here, the precoding weight is regularly switched in 4 slots, but the number of regularly switched slots is not limited to 4 slots.)
Incidentally, Non-Patent Document 4 describes switching precoding weights for each slot, and Non-Patent Document 4 is characterized by switching precoding weights at random. On the other hand, the present embodiment is characterized in that a predetermined period is provided and the precoding weights are switched regularly. In a 2 × 2 precoding weight matrix composed of four precoding weights, 4 The absolute values of the two precoding weights are equal (1 / sqrt (2)), and the precoding weight matrix having this characteristic is switched regularly.

  In a LOS environment, reception quality may be greatly improved by using a special precoding matrix, but the special precoding matrix differs depending on the situation of the direct wave. However, there is a certain rule in the LOS environment. If a special precoding matrix is regularly switched according to this rule, the data reception quality is greatly improved. On the other hand, if the precoding matrix is switched at random, there may be precoding matrices other than the special precoding matrix described above, and only the precoding matrix that is not suitable for the LOS environment is offset. There is also the possibility of performing precoding, and this does not always provide good reception quality in a LOS environment. Therefore, it is necessary to realize a precoding switching method suitable for the LOS environment, and the present invention proposes a precoding method related thereto.

FIG. 7 shows an example of the configuration of receiving apparatus 700 in the present embodiment. Radio section 703_X receives reception signal 702_X received by antenna 701_X, performs processing such as frequency conversion and orthogonal demodulation, and outputs baseband signal 704_X.
Channel fluctuation estimation section 705_1 in modulated signal z1 transmitted by the transmission apparatus receives baseband signal 704_X, extracts channel estimation reference symbol 501_1 in FIG. 5, and obtains a value corresponding to h11 in equation (36). The channel estimation signal 706_1 is output.

  Channel fluctuation estimation section 705_2 in modulated signal z2 transmitted by the transmission apparatus receives baseband signal 704_X, extracts channel estimation reference symbol 501_2 in FIG. 5, and obtains a value corresponding to h12 in equation (36). The channel estimation signal 706_2 is output.

Radio section 703_Y receives reception signal 702_Y received by antenna 701_Y, performs processing such as frequency conversion and orthogonal demodulation, and outputs baseband signal 704_Y.
Channel fluctuation estimation section 707_1 in modulated signal z1 transmitted from the transmission apparatus receives baseband signal 704_Y as input, extracts channel estimation reference symbol 501_1 in FIG. 5, and obtains a value corresponding to h21 in equation (36). The channel estimation signal 708_1 is output.

  Channel fluctuation estimation section 707_2 in modulated signal z2 transmitted from the transmission apparatus receives baseband signal 704_Y as input, extracts channel estimation reference symbol 501_2 in FIG. The channel estimation signal 708_2 is output.

  Control information decoding section 709 receives baseband signals 704_X and 704_Y, detects symbol 500_1 for notifying the transmission method of FIG. 5, and outputs a signal 710 related to the transmission method information notified by the transmission apparatus.

  The signal processing unit 711 receives the baseband signals 704_X and 704_Y, the channel estimation signals 706_1, 706_2, 708_1, and 708_2, and the signal 710 related to the transmission method notified by the transmission apparatus, performs detection and decoding, and performs reception data 712_1 and 712_2 are output.

Next, the operation of the signal processing unit 711 in FIG. 7 will be described in detail. FIG. 8 shows an example of the configuration of the signal processing unit 711 in the present embodiment. FIG. 8 mainly includes an INNER MIMO detection unit, a soft-in / soft-out decoder, and a weighting coefficient generation unit. The details of the iterative decoding method in this configuration are described in Non-Patent Document 2 and Non-Patent Document 3, but the MIMO transmission system described in Non-Patent 2 and Non-Patent Document 3 is a spatial multiplexing MIMO transmission system. However, the transmission method in the present embodiment is a MIMO transmission method in which the precoding weight is changed over time, which is different from Non-Patent Document 2 and Non-Patent Document 3. In Equation (36), the (channel) matrix is H (t), the precoding weight matrix in FIG. 6 is W (t) (where the precoding weight matrix changes depending on t), and the received vector is R (t) = When (r1 (t), r2 (t)) ^T and stream vector S (t) = (s1 (t), s2 (t)) ^T , the following relational expression is established.

At this time, the receiving apparatus considers H (t) W (t) as a channel matrix, and can apply the decoding methods of Non-Patent Document 2 and Non-Patent Document 3 to R (t) as a received vector. it can.
Therefore, the weighting coefficient generation unit 819 in FIG. 8 receives a signal 818 (corresponding to 710 in FIG. 7) related to the transmission method information notified by the transmission apparatus, and outputs a signal 820 related to the weighting coefficient information.

  The INNNER MIMO detection unit 803 receives a signal 820 related to the weighting coefficient information, and uses this signal to perform the calculation of Expression (41). Then, iterative detection and decoding will be performed, and the operation will be described.

  The signal processing unit in FIG. 8 needs to perform a processing method as shown in FIG. 10 in order to perform iterative decoding (iterative detection). First, one codeword (or one frame) of the modulation signal (stream) s1 and one codeword (or one frame) of the modulation signal (stream) s2 are decoded. As a result, from the soft-in / soft-out decoder, one codeword (or one frame) of the modulation signal (stream) s1 and one codeword (or one frame) of the modulation signal (stream) s2 A log-likelihood ratio (LLR) is obtained. Then, detection and decoding are performed again using the LLR. This operation is performed a plurality of times (this operation is called iterative decoding (iterative detection)). In the following, the description will focus on a method for creating a log likelihood ratio (LLR) of a symbol at a specific time in one frame.

  8, the storage unit 815 has a baseband signal 801X (corresponding to the baseband signal 704_X in FIG. 7), a channel estimation signal group 802X (corresponding to the channel estimation signals 706_1 and 706_2 in FIG. 7), and a baseband. The signal 801Y (corresponding to the baseband signal 704_Y in FIG. 7) and the channel estimation signal group 802Y (corresponding to the channel estimation signals 708_1 and 708_2 in FIG. 7) are input to realize iterative decoding (iterative detection). Then, H (t) W (t) in equation (41) is executed (calculated), and the calculated matrix is stored as a modified channel signal group. Then, the storage unit 815 outputs the above signals as a baseband signal 816X, a modified channel estimation signal group 817X, a baseband signal 816Y, and a modified channel estimation signal group 817Y when necessary.

Subsequent operations will be described separately for the case of initial detection and the case of iterative decoding (iterative detection).
<In case of initial detection>
The INNER MIMO detection unit 803 receives the baseband signal 801X, the channel estimation signal group 802X, the baseband signal 801Y, and the channel estimation signal group 802Y. Here, the modulation scheme of the modulation signal (stream) s1 and the modulation signal (stream) s2 will be described as 16QAM.

The INNER MIMO detection unit 803 first executes H (t) W (t) from the channel estimation signal group 802X and the channel estimation signal group 802Y, and obtains candidate signal points corresponding to the baseband signal 801X. The state at that time is shown in FIG. In FIG. 11, ● (black circle) is a candidate signal point on the IQ plane. Since the modulation method is 16QAM, there are 256 candidate signal points. (However, since the image diagram is shown in FIG. 11, 256 candidate signal points are not shown.) Here, 4 bits transmitted by the modulation signal s1 are b0, b1, b2, b3, and the modulation signal s2. If the 4 bits to be transmitted are b4, b5, b6, and b7, there are candidate signal points corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) in FIG. Then, the squared Euclidean distance between the reception signal point 1101 (corresponding to the baseband signal 801X) and each candidate signal point is obtained. Then, each square Euclidean distance is divided by the noise variance σ ² . Therefore, a value obtained by dividing the candidate signal points and a received signal point square Euclidean distances corresponding with the variance of noise (b0, b1, b2, b3 , b4, b5, b6, b7) E X (b0, b1, b2 , B3, b4, b5, b6, b7).

Similarly, H (t) W (t) is executed from channel estimation signal group 802X and channel estimation signal group 802Y, a candidate signal point corresponding to baseband signal 801Y is obtained, and a received signal point (corresponding to baseband signal 801Y) And the square Euclidean distance is divided by the noise variance σ ² . Therefore, a value obtained by dividing the candidate signal point corresponding to (b0, b1, b2, b3, b4, b5, b6, b7) and the received signal point squared Euclidean distance by the variance of noise is represented by E _Y (b0, b1, b2 , B3, b4, b5, b6, b7).

Then, E _X (b0, b1, b2, b3, b4, b5, b6, b7) + E _Y (b0, b1, b2, b3, b4, b5, b6, b7) = E (b0, b1, b2, b3) , B4, b5, b6, b7).

The INNER MIMO detection unit 803 outputs E (b0, b1, b2, b3, b4, b5, b6, b7) as a signal 804.
Log likelihood calculation section 805A receives signal 804 as input, calculates log likelihood for bits b0 and b1, and b2 and b3, and outputs log likelihood signal 806A. However, in the calculation of the log likelihood, the log likelihood when “1” and the log likelihood when “0” are calculated. The calculation method is as shown in Expression (28), Expression (29), and Expression (30), and details are shown in Non-Patent Document 2 and Non-Patent Document 3.

Similarly, log likelihood calculation section 805B receives signal 804 as input, calculates log likelihood of bits b4 and b5 and b6 and b7, and outputs log likelihood signal 806B.
The deinterleaver (807A) receives the log likelihood signal 806A, performs deinterleaving corresponding to the interleaver (interleaver (304A in FIG. 3)), and outputs a log likelihood signal 808A after deinterleaving.

  Similarly, the deinterleaver (807B) receives the log likelihood signal 806B as input, performs deinterleaving corresponding to the interleaver (interleaver (304B) in FIG. 3), and outputs a log likelihood signal 808B after deinterleaving.

  Log likelihood ratio calculation section 809A receives log likelihood signal 808A after deinterleaving, and calculates a log likelihood ratio (LLR: Log-Likelihood Ratio) of bits encoded by encoder 302A in FIG. The log likelihood ratio signal 810A is output.

  Similarly, log-likelihood ratio calculation section 809B receives log-likelihood signal 808B after deinterleaving and inputs the log-likelihood ratio (LLR: Log-Likelihood Ratio) of bits encoded by encoder 302B in FIG. ) And a log likelihood ratio signal 810B is output.

The soft-in / soft-out decoder 811A receives the log likelihood ratio signal 810A, performs decoding, and outputs a log likelihood ratio 812A after decoding.
Similarly, Soft-in / soft-out decoder 811B receives log-likelihood ratio signal 810B as input, performs decoding, and outputs log-likelihood ratio 812B after decoding.

<In the case of iterative decoding (iterative detection), the number of iterations k>
The interleaver (813A) receives the log likelihood ratio 812A after decoding obtained in the (k-1) th soft-in / soft-out decoding, performs interleaving, and outputs a log likelihood ratio 814A after interleaving. . At this time, the interleave pattern of the interleaver (813A) is the same as the interleave pattern of the interleaver (304A) of FIG.

  The interleaver (813B) receives the log likelihood ratio 812B after decoding obtained in the (k-1) th soft-in / soft-out decoding, performs interleaving, and outputs the log likelihood ratio 814B after interleaving. . At this time, the interleave pattern of the interleaver (813B) is the same as the interleave pattern of the interleaver (304B) of FIG.

  The INNER MIMO detection unit 803 inputs a baseband signal 816X, a modified channel estimation signal group 817X, a baseband signal 816Y, a modified channel estimation signal group 817Y, an interleaved log likelihood ratio 814A, and an interleaved log likelihood ratio 814B. And Here, not the baseband signal 801X, the channel estimation signal group 802X, the baseband signal 801Y, and the channel estimation signal group 802Y, but the baseband signal 816X, the modified channel estimation signal group 817X, the baseband signal 816Y, and the modified channel estimation signal group 817Y. Is used because of a delay time due to iterative decoding.

  The difference between the operation at the time of iterative decoding and the operation at the time of initial detection of the INNER MIMO detection unit 803 is that the log likelihood ratio 814A after interleaving and the log likelihood ratio 814B after interleaving are used in signal processing. It is. The INNNER MIMO detection unit 803 first obtains E (b0, b1, b2, b3, b4, b5, b6, b7) as in the case of initial detection. In addition, coefficients corresponding to Equation (11) and Equation (32) are obtained from the log likelihood ratio 814A after interleaving and the log likelihood ratio 914B after interleaving. Then, the value of E (b0, b1, b2, b3, b4, b5, b6, b7) is corrected using the obtained coefficient, and the value is changed to E ′ (b0, b1, b2, b3, b4, b5). , B6, b7) and output as a signal 804.

  Log likelihood calculation section 805A receives signal 804 as input, calculates log likelihood for bits b0 and b1, and b2 and b3, and outputs log likelihood signal 806A. However, in the calculation of the log likelihood, the log likelihood when “1” and the log likelihood when “0” are calculated. The calculation method is as shown in Expression (31), Expression (Formula 32), Expression (33), Expression (34), and Expression (35), and is shown in Non-Patent Document 2 and Non-Patent Document 3. Yes.

  Similarly, log likelihood calculation section 805B receives signal 804 as input, calculates log likelihood of bits b4 and b5 and b6 and b7, and outputs log likelihood signal 806B. The operation after the deinterleaver is the same as the initial detection.

FIG. 8 shows the configuration of the signal processing unit in the case of performing iterative detection. However, iterative detection is not necessarily an essential configuration for obtaining good reception quality, and is a component required only for iterative detection. The interleaver 813A or 813B may be omitted. At this time, the INNNER MIMO detection unit 803 does not perform repetitive detection.
An important part of the present embodiment is to calculate H (t) W (t). In addition, as shown in Non-Patent Document 5 and the like, initial detection and iterative detection may be performed using QR decomposition.
Further, as shown in Non-Patent Document 11, linear calculation of MMSE (Minimum Mean Square Error) and ZF (Zero Forcing) is performed based on H (t) W (t), and initial detection is performed. Good.

  FIG. 9 shows a configuration of a signal processing unit different from that in FIG. 8, and is a signal processing unit for a modulated signal transmitted by the transmission apparatus in FIG. The difference from FIG. 8 is the number of soft-in / soft-out decoders. The soft-in / soft-out decoder 901 receives log likelihood ratio signals 810A and 810B as inputs and performs decoding. A log likelihood ratio 902 is output. The distribution unit 903 receives the log likelihood ratio 902 after decoding as input, and performs distribution. The other parts are the same as those in FIG.

  FIG. 12 shows BER characteristics when the transmission method is the transmission method using the precoding weight of the present embodiment under the same conditions as in FIG. 12A shows the BER characteristic of Max-log-APP (Non-patent Document 1 and Non-patent Document 2) (APP: a posteriprobability) in which iterative detection is not performed, and FIG. The BER characteristic of Max-log-APP (refer nonpatent literature 1 and nonpatent literature 2) (number of repetitions 5 times) which performed the detection is shown. Comparing FIG. 12 and FIG. 29, it can be seen that, when the transmission method of the present embodiment is used, the BER characteristic when the rice factor is large is greatly improved compared to the BER characteristic when using spatial multiplexing MIMO transmission. Thus, the effectiveness of the system of this embodiment can be confirmed.

  As described above, when the transmission apparatus of the MIMO transmission system transmits a plurality of modulated signals from a plurality of antennas as in the present embodiment, the precoding weight is switched over time and the switching is performed regularly. In the LOS environment where the direct wave is dominant, it is possible to obtain an effect that the transmission quality is improved as compared with the case of using the conventional spatial multiplexing MIMO transmission.

  In this embodiment, the operation of the receiving apparatus is described with the number of antennas being limited, but it can be similarly implemented even when the number of antennas is increased. That is, the number of antennas in the receiving apparatus does not affect the operation and effect of the present embodiment. In this embodiment, the LDPC code has been particularly described as an example. However, the present invention is not limited to this, and the decoding method is limited to the sum-product decoding as an example of a soft-in / soft-out decoder. However, there are other soft-in / soft-out decoding methods such as BCJR algorithm, SOVA algorithm, Msx-log-MAP algorithm, and the like. Details are described in Non-Patent Document 6.

  In the present embodiment, the single carrier scheme has been described as an example. However, the present invention is not limited to this, and the present invention can be similarly implemented even when multicarrier transmission is performed. Therefore, for example, spread spectrum communication system, OFDM (Orthogonal Frequency-Division Multiplexing) system, SC-FDMA (Single Carrier Frequency Multiple Access), SC-OFDM (Single Carrier Multiple Access), SC-OFDM (Single Carrier Multiple Access). The same can be applied to the case of using the wavelet OFDM method shown in FIG. In the present embodiment, symbols other than data symbols, for example, pilot symbols (preamble, unique word, etc.), control information transmission symbols, and the like may be arranged in any manner.

Hereinafter, an example in which the OFDM method is used as an example of the multicarrier method will be described.
FIG. 13 shows a configuration of a transmission apparatus when the OFDM method is used. In FIG. 13, the same reference numerals are given to those that operate in the same manner as in FIG. 3.

  The OFDM scheme-related processing unit 1301A receives the weighted signal 309A, performs OFDM scheme-related processing, and outputs a transmission signal 1302A. Similarly, OFDM scheme related processing section 1301B receives weighted signal 309B and outputs transmission signal 1302B.

  FIG. 14 shows an example of the configuration after the OFDM scheme-related processing units 1301A and 1301B in FIG. 13, and portions related to 1301A to 312A in FIG. Are 1401B to 1410B.

  The serial / parallel converter 1402A performs serial / parallel conversion on the weighted signal 1401A (corresponding to the weighted signal 309A in FIG. 13), and outputs a parallel signal 1403A.

Rearrangement section 1404A receives parallel signal 1403A as input, performs rearrangement, and outputs rearranged signal 1405A. The rearrangement will be described in detail later.
The inverse fast Fourier transform unit 1406A receives the rearranged signal 1405A, performs an inverse fast Fourier transform, and outputs a signal 1407A after the inverse Fourier transform.

Radio section 1408A receives signal 1407A after inverse Fourier transform as input, performs processing such as frequency conversion and amplification, outputs modulated signal 1409A, and modulated signal 1409A is output from antenna 1410A as a radio wave.
The serial / parallel converter 1402B performs serial / parallel conversion on the weighted signal 1401B (corresponding to the weighted signal 309B in FIG. 13), and outputs a parallel signal 1403B.

Rearranger 1404B receives parallel signal 1403B as input, performs rearrangement, and outputs rearranged signal 1405B. The rearrangement will be described in detail later.
The inverse fast Fourier transform unit 1406B receives the rearranged signal 1405B as input, performs inverse fast Fourier transform, and outputs a signal 1407B after inverse Fourier transform.

  The radio unit 1408B receives the signal 1407B after the inverse Fourier transform, performs frequency conversion, amplification, and the like, outputs a modulation signal 1409B, and the modulation signal 1409B is output as a radio wave from the antenna 1410B.

  Since the transmission apparatus of FIG. 3 is not a transmission method using multicarriers, precoding is switched so as to have four periods as shown in FIG. 6, and symbols after precoding are arranged in the time axis direction. When a multi-carrier transmission scheme such as the OFDM scheme shown in FIG. 13 is used, naturally, symbols after precoding are arranged in the time axis direction as shown in FIG. In the case of the multi-carrier transmission method, a method of arranging using the frequency axis direction or both the frequency axis and the time axis can be considered. Hereinafter, this point will be described.

  FIG. 15 shows an example of the symbol rearrangement method in the rearrangement units 1401A and 1401B in FIG. 14 at the horizontal axis frequency and the vertical axis time, and the frequency axis ranges from (sub) carrier 0 to (sub) carrier 9 The modulation signals z1 and z2 use the same frequency band at the same time (time), and FIG. 15A shows a symbol rearrangement method of the modulation signal z1, and FIG. Indicates a rearrangement method of symbols of the modulation signal z2. Numbers such as # 1, # 2, # 3, # 4,... Are sequentially assigned to the symbols of the weighted signal 1401A input to the serial / parallel converter 1402A. At this time, as shown in FIG. 15A, symbols # 1, # 2, # 3, # 4,... Are arranged in order from carrier 0, and symbols # 1 to # 9 are arranged at time $ 1. Thereafter, symbols # 10 to # 19 are regularly arranged such that they are arranged at time $ 2.

  Similarly, # 1, # 2, # 3, # 4,... Are sequentially assigned to the symbols of the weighted signal 1401B input to the serial / parallel converter 1402B. At this time, as shown in FIG. 15B, symbols # 1, # 2, # 3, # 4,... Are arranged in order from carrier 0, and symbols # 1 to # 9 are arranged at time $ 1. Thereafter, symbols # 10 to # 19 are regularly arranged such that they are arranged at time $ 2.

  A symbol group 1501 and a symbol group 1502 shown in FIG. 15 are symbols for one period when the precoding weight switching method shown in FIG. 6 is used, and symbol # 0 is a precoding weight of slot 4i in FIG. Symbol # 1 is a symbol when the precoding weight of slot 4i + 1 in FIG. 6 is used, and symbol # 2 is a symbol when the precoding weight of slot 4i + 2 in FIG. 6 is used. Yes, symbol # 3 is a symbol when the precoding weight of slot 4i + 3 in FIG. 6 is used. Therefore, in symbol #x, when x mod 4 is 0, symbol #x is a symbol when the precoding weight of slot 4i in FIG. 6 is used, and when x mod 4 is 1, symbol #x is 6 is a symbol when the precoding weight of slot 4i + 1 is used, and when x mod 4 is 2, symbol #x is a symbol when the precoding weight of slot 4i + 2 of FIG. 6 is used, and x mod 4 Is # 3, symbol #x is a symbol when the precoding weight of slot 4i + 3 in FIG. 6 is used.

  As described above, when a multi-carrier transmission scheme such as the OFDM scheme is used, unlike single carrier transmission, symbols can be arranged in the frequency axis direction. The way of arranging the symbols is not limited to the arrangement as shown in FIG. Another example will be described with reference to FIGS.

  FIG. 16 shows an example of the symbol rearrangement method in the rearrangement units 1401A and 1401B in FIG. 14 at the horizontal axis frequency and the vertical axis time different from FIG. 15, and FIG. 16 (A) shows the modulation signal z1. FIG. 16B shows a symbol rearrangement method of the modulation signal z2. FIGS. 16A and 16B differ from FIG. 15 in that the symbol rearrangement method of the modulation signal z1 and the symbol rearrangement method of the modulation signal z2 are different. In FIG. 0 to # 5 are allocated from carrier 4 to carrier 9, symbols # 6 to # 9 are allocated to carriers 0 to 3, and then symbols # 10 to # 19 are allocated to each carrier according to the same rule. At this time, similarly to FIG. 15, symbol group 1601 and symbol group 1602 shown in FIG. 16 are symbols for one period when the precoding weight switching method shown in FIG. 6 is used.

  FIG. 17 shows an example of the symbol rearrangement method in the rearrangement units 1401A and 1401B in FIG. 14 at the horizontal axis frequency and the vertical axis time different from FIG. 15, and FIG. 17 (A) shows the modulation signal z1. A symbol rearrangement method, and FIG. 17B shows a symbol rearrangement method of the modulation signal z2. 17A and 17B differ from FIG. 15 in that symbols are arranged in order on the carrier in FIG. 15, whereas symbols are not arranged in order on the carrier in FIG. Is a point. Of course, in FIG. 17, the rearrangement method of the modulation signal z1 and the rearrangement method of the modulation signal z2 may be different as in FIG.

  FIG. 18 shows an example of a symbol rearrangement method in rearrangement sections 1401A and 1401B in FIG. 14 at a horizontal axis frequency and a vertical axis time different from those in FIGS. 18 and 15 to 17, and FIG. The symbol rearrangement method of z1 and FIG. 18B show the symbol rearrangement method of the modulation signal z2. 15 to 17, symbols are arranged in the frequency axis direction, but in FIG. 18, symbols are arranged using both the frequency and the time axis.

  In FIG. 6, an example in which the switching of precoding weight is switched in 4 slots has been described, but here, a case in which switching is performed in 8 slots will be described as an example. Symbol group 1801 and symbol group 1802 shown in FIG. 18 are symbols for one period when using the precoding weight switching method (and therefore, eight symbols), and symbol # 0 uses the precoding weight of slot 8i. Symbol # 1 is a symbol when the precoding weight of slot 8i + 1 is used, symbol # 2 is a symbol when the precoding weight of slot 8i + 2 is used, and symbol # 3 is slot 8i + 3 The symbol # 4 is a symbol when the precoding weight of the slot 8i + 4 is used, and the symbol # 5 is a symbol when the precoding weight of the slot 8i + 5 is used. , Symbol # 6 is the symbol when using the precoding weight of slot 8i + 6 symbols # 7 is a symbol when using the precoding weight of slot 8i + 7. Therefore, in symbol #x, when x mod 8 is 0, symbol #x is a symbol when the precoding weight of slot 8i is used, and when x mod 8 is 1, symbol #x is a pre-slot of slot 8i + 1. Symbol when using coding weight, when x mod 8 is 2, symbol #x is a symbol when using precoding weight of slot 8i + 2, and when symbol x mod 8 is 3, symbol #x is Symbol when using precoding weight of slot 8i + 3, when x mod 8 is 4, symbol #x is a symbol when using precoding weight of slot 8i + 4, and when x mod 8 is 5, Symbol #x is the precoding weight of slot 8i + 5 When x mod 8 is 6, symbol #x is a symbol when the precoding weight of slot 8i + 6 is used, and when x mod 8 is 7, symbol #x is a pre-slot of slot 8i + 7. This is a symbol when coding weight is used. In the arrangement of symbols in FIG. 18, symbols for one period are arranged using 4 slots in the time axis direction and 2 slots in the frequency axis direction in total, 4 × 2 = 8 slots. The number of symbols per minute is m × n symbols (that is, there are m × n types of precoding weights). The slot (number of carriers) in the frequency axis direction used for arranging symbols for one period is n, time When the slot used in the axial direction is m, it is preferable that m> n. This is because the phase of the direct wave is more gradual in fluctuation in the time axis direction than in the frequency axis direction. Therefore, since the precoding weight change of this embodiment is performed in order to reduce the influence of the stationary direct wave, it is desired to reduce the fluctuation of the direct wave in the period in which the precoding weight is changed. Therefore, m> n is preferable. Further, considering the above points, it is more direct wave to perform rearrangement using both the frequency axis and the time axis as shown in FIG. 18 than to rearrange symbols only in the frequency axis direction or only in the time axis direction. Is likely to be stationary, and the effect of easily obtaining the effects of the present invention can be obtained. However, when arranged in the frequency axis direction, since the fluctuation of the frequency axis is steep, there is a possibility that diversity gain can be obtained, so the method of rearranging using both the frequency axis and the time axis is not necessarily optimal This is not always the case.

  FIG. 19 shows an example of the symbol rearrangement method in the rearrangement units 1401A and 1401B in FIG. 14 at the horizontal axis frequency and the vertical axis time, which is different from FIG. 18, and FIG. 19A shows the modulation signal z1. FIG. 19B shows a symbol rearrangement method of the modulation signal z2. In FIG. 19, symbols are arranged using both the frequency and the time axis as in FIG. 18, but the difference from FIG. 18 is that in FIG. 18, the frequency direction is given priority, and then in the time axis direction. In contrast to symbols being arranged, in FIG. 19, the time axis direction is prioritized, and thereafter symbols are arranged in the time axis direction. In FIG. 19, a symbol group 1901 and a symbol group 1902 are symbols for one period when the precoding switching method is used.

  In FIG. 18 and FIG. 19, similar to FIG. 16, even if the symbol arrangement method of the modulation signal z1 and the symbol arrangement method of the modulation signal z2 are arranged differently, it can be implemented in the same manner. An effect that reception quality can be obtained can be obtained. 18 and 19, even if symbols are not arranged sequentially as shown in FIG. 17, it can be carried out in the same manner, and an effect that high reception quality can be obtained can be obtained. it can.

  FIG. 27 shows an example of the symbol rearrangement method in the rearrangement units 1401A and 140B in FIG. 14 at the horizontal axis frequency and the vertical axis time different from the above. Consider a case where the precoding matrix is switched regularly using four slots such as Expression (37) to Expression (40). In FIG. 27, the characteristic point is that symbols are arranged in order in the frequency axis direction, but when proceeding in the time axis direction, cyclic shift is performed cyclically by n (n = 1 in the example of FIG. 27). It is. In the four symbols shown in the symbol group 2710 in the frequency axis direction in FIG. 27, the precoding matrixes of Expressions (37) to (40) are switched.

  At this time, precoding using the precoding matrix of Expression (37) is used for the symbol # 0, precoding using the precoding matrix of Expression (38) is used for # 1, and precoding matrix of Expression (39) is used for # 2. In # 3, precoding using the precoding matrix of Equation (40) is performed.

  Similarly, for the symbol group 2720 in the frequency axis direction, precoding using the precoding matrix of Expression (37) is used for the symbol # 4, precoding using the precoding matrix of Expression (38) is used for # 5, # 6 Then, precoding using the precoding matrix of Expression (39) is performed, and precoding using the precoding matrix of Expression (40) is performed in # 7.

  Although the precoding matrix is switched as described above for the symbol of time $ 1, since the cyclic shift is performed in the time axis direction, the symbol groups 2701, 2702, 2703, and 2704 are pre-coded as follows. The coding matrix is switched.

  In the symbol group 2701 in the time axis direction, precoding using the precoding matrix of Expression (37) is performed for the symbol # 0, precoding using the precoding matrix of Expression (38) is used for # 9, and Expression ( It is assumed that precoding using the precoding matrix of 39) and precoding using the precoding matrix of Equation (40) are performed in # 27.

  In the symbol group 2702 in the time axis direction, precoding using the precoding matrix of Expression (37) is used for the symbol # 28, precoding using the precoding matrix of Expression (38) is used for # 1, and expression ( It is assumed that precoding using the precoding matrix of 39) and precoding using the precoding matrix of Equation (40) are performed in # 19.

  In the symbol group 2703 in the time axis direction, precoding using the precoding matrix of Expression (37) is performed for the symbol # 20, precoding using the precoding matrix of Expression (38) is performed for # 29, and Expression ( It is assumed that precoding using the precoding matrix of 39) and precoding using the precoding matrix of Equation (40) are performed in # 10.

  In the symbol group 2704 in the time axis direction, precoding using the precoding matrix of Expression (37) is used for the symbol # 12, precoding using the precoding matrix of Expression (38) is used for # 21, and Expression ( It is assumed that precoding using the precoding matrix of 39) and precoding using the precoding matrix of Equation (40) are performed in # 3.

  The feature in FIG. 27 is that, for example, when attention is paid to the symbol # 11, both adjacent symbols (# 10 and # 12) in the frequency axis direction at the same time are both precoded using a precoding matrix different from # 11. And the symbols (# 2 and # 20) on both sides in the time axis direction of the same carrier of the symbol # 11 are both precoded using a precoding matrix different from # 11. is there. This is not limited to the # 11 symbol, and all symbols having symbols on both sides in the frequency axis direction and the time axis direction have the same characteristics as the # 11 symbol. As a result, the precoding matrix is effectively switched and the influence of the direct wave on the steady state is less likely to occur, so that there is a high possibility that the data reception quality is improved.

  In FIG. 27, the case where n = 1 is described. However, the present invention is not limited to this, and the same can be implemented when n = 3. Also, in FIG. 27, when the symbols are arranged on the frequency axis and the time advances in the axial direction, the above feature is realized by providing the feature of cyclically shifting the order of symbol arrangement. There is also a method for realizing the above characteristics by arranging them in a regular manner (which may be regular).

(Embodiment 2)
In the first embodiment, the case where the precoding weights are switched regularly as shown in FIG. 6 has been described, but in this embodiment, a specific precoding weight design method different from the precoding weights in FIG. Will be described.

In FIG. 6, the method of switching the precoding weights of Expression (37) to Expression (40) has been described. When this is generalized, the precoding weight can be changed as follows. (However, the precoding weight switching cycle is 4, and the same description as in equations (37) to (40) is given.)
For symbol number 4i (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number 4i + 1:

For symbol number 4i + 2:

For symbol number 4i + 3:

From the equations (36) and (41), the received vector R (t) = (r1 (t), r2 (t)) ^T can be expressed as follows.
For symbol number 4i:

For symbol number 4i + 1:

For symbol number 4i + 2:

For symbol number 4i + 3:

At this time, it is assumed that only direct wave components exist in the channel elements h ₁₁ (t), h ₁₂ (t), h ₂₁ (t), and h ₂₂ (t), and the amplitude components of the direct wave components are Assume that all are equal and that there is no variation in time. Then, Formula (46)-Formula (49) can be expressed as follows.
For symbol number 4i:

For symbol number 4i + 1:

For symbol number 4i + 2:

For symbol number 4i + 3:

However, in Formula (50)-Formula (53), A shall be a positive real number and q shall be a complex number. The values of A and q are determined according to the positional relationship between the transmission device and the reception device. Formulas (50) to (53) are expressed as follows.
For symbol number 4i:

For symbol number 4i + 1:

For symbol number 4i + 2:

For symbol number 4i + 3:

Then, when q is expressed as follows, a signal component based on one of s1 and s2 is not included in r1 and r2, and thus it is impossible to obtain a signal of either s1 or s2.
For symbol number 4i:

For symbol number 4i + 1:

For symbol number 4i + 2:

For symbol number 4i + 3:

At this time, if q has the same solution in symbol numbers 4i, 4i + 1, 4i + 2, and 4i + 3, the channel component of the direct wave does not vary greatly. Since the apparatus cannot obtain good reception quality at any symbol number, it is difficult to obtain error correction capability even if an error correction code is introduced. Therefore, in order for q not to have the same solution, focusing on the solution that does not include δ among the two solutions of q, the following conditions are necessary from the equations (58) to (61). It becomes.

(X is 0, 1, 2, 3, y is 0, 1, 2, 3, and x ≠ y.)

As an example that satisfies condition # 1,
(Example # 1)
<1> θ11 (4i) = θ11 (4i + 1) = θ11 (4i + 2) = θ11 (4i + 3) = 0 radians,
<2> θ21 (4i) = 0 radians <3> θ21 (4i + 1) = π / 2 radians <4> θ21 (4i + 2) = π radians <5> θ21 (4i + 3) = 3π / 2 radians is considered It is done. (The above is an example, and in the set of (θ21 (4i), θ21 (4i + 1), θ21 (4i + 2), θ21 (4i + 3)), 0 radians, π / 2 radians, π radians, and 3π / 2 radians are one. At this time, especially if there is a condition <1>, it is not necessary to apply signal processing (rotation processing) to the baseband signal S1 (t), so the circuit scale can be reduced. There is an advantage of being able to plan. As another example,
(Example # 2)
<6> θ11 (4i) = 0 radians <7> θ11 (4i + 1) = π / 2 radians <8> θ11 (4i + 2) = π radians <9> θ11 (4i + 3) = 3π / 2 radians,
<10> A method of setting θ21 (4i) = θ21 (4i + 1) = θ21 (4i + 2) = θ21 (4i + 3) = 0 radians is also conceivable. (The above is an example, and in the set of (θ11 (4i), θ11 (4i + 1), θ11 (4i + 2), θ11 (4i + 3)), 0 radians, π / 2 radians, π radians, and 3π / 2 radians are one. At this time, especially if there is a condition <6>, it is not necessary to apply signal processing (rotation processing) to the baseband signal S2 (t). There is an advantage of being able to plan. Another example is as follows.
(Example # 3)
<11> θ11 (4i) = θ11 (4i + 1) = θ11 (4i + 2) = θ11 (4i + 3) = 0 radians,
<12> θ21 (4i) = 0 radians <13> θ21 (4i + 1) = π / 4 radians <14> θ21 (4i + 2) = π / 2 radians <15> θ21 (4i + 3) = 3π / 4 radians In the set of (θ21 (4i), θ21 (4i + 1), θ21 (4i + 2), θ21 (4i + 3)), 0 radians, π / 4 radians, π / 2 radians, and 3π / 4 radians one by one It only has to exist.)
(Example # 4)
<16> θ11 (4i) = 0 radians <17> θ11 (4i + 1) = π / 4 radians <18> θ11 (4i + 2) = π / 2 radians <19> θ11 (4i + 3) = 3π / 4 radians
<20> θ21 (4i) = θ21 (4i + 1) = θ21 (4i + 2) = θ21 (4i + 3) = 0 radians (The above is an example, (θ11 (4i), θ11 (4i + 1), θ11 (4i + 2), θ11 ( In the set of 4i + 3), it is only necessary to have 0 radians, π / 4 radians, π / 2 radians, and 3π / 4 radians one by one.)
Although four examples have been given, the method satisfying the condition # 1 is not limited to this.

  Next, not only θ11 and θ12 but also design requirements for λ and δ will be described. It is only necessary to set a certain value for λ. As a requirement, it is necessary to give a requirement for δ. Therefore, a method for setting δ when λ is 0 radians will be described.

In this case, when π / 2 radians ≦ | δ | ≦ π radians with respect to δ, it is possible to obtain good reception quality particularly in the LOS environment.
By the way, in symbol numbers 4i, 4i + 1, 4i + 2, and 4i + 3, there are two points q each having bad reception quality. Therefore, there are 2 × 4 = 8 points. In order to prevent the reception quality from deteriorating at a specific receiving terminal in the LOS environment, these 8 points may be all different solutions. In this case, in addition to <condition # 1>, the condition <condition # 2> is required.

In addition, it is preferable that the phases of these eight points exist uniformly. In the following, a method for setting δ that satisfies this requirement will be described (because it is highly likely that the phase of the direct wave has a uniform distribution).

In the case of (Example # 1) and (Example # 2), by setting δ to be ± 3π / 4 radians, the point where the reception quality is poor is uniformly present. For example, if (example # 1) and δ is 3π / 4 radians, there is a point that the reception quality deteriorates once every four slots as shown in FIG. 20 (A is a positive real number). (Example # 3) In the case of (Example # 4), by setting δ to ± π radians, the phase with poor reception quality is present uniformly. For example, when (example # 3) is assumed and δ is π radians, there is a point that the reception quality deteriorates once every four slots as shown in FIG. (If the element q in the channel matrix H is present at the points shown in FIGS. 20 and 21, the reception quality is degraded.)
As described above, good reception quality can be obtained in the LOS environment. In the above description, the example in which the precoding weight is changed at a 4-slot period has been described. However, the case where the precoding weight is changed at an N-slot period will be described below. When considered in the same manner as in the first embodiment and the above description, the following processing is performed after the symbol number.
For symbol number Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

Therefore, r1 and r2 are expressed as follows.
For symbol number Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

At this time, it is assumed that only direct wave components exist in the channel elements h ₁₁ (t), h ₁₂ (t), h ₂₁ (t), and h ₂₂ (t), and the amplitude components of the direct wave components are Assume that all are equal and that there is no variation in time. Then, Formula (66)-Formula (69) can be expressed as follows.
For symbol number Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

However, in Formula (70)-Formula (73), A shall be a real number and q shall be a complex number. The values of A and q are determined according to the positional relationship between the transmission device and the reception device. Formulas (70) to (73) are expressed as follows.
For symbol number Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

Then, when q is expressed as follows, a signal component based on one of s1 and s2 is not included in r1 and r2, and thus it is impossible to obtain a signal of either s1 or s2.
For symbol number Ni (i is an integer greater than or equal to 0):

For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

At this time, in the symbol numbers N to Ni + N−1, if q has the same solution, the channel component of the direct wave does not change greatly. Even in the case of numbers, it is difficult to obtain good reception quality, so even if an error correction code is introduced, it is difficult to obtain error correction capability. Therefore, in order for q not to have the same solution, focusing on the solution that does not include δ among the two solutions of q, the following conditions are necessary from equations (78) to (81): It becomes.

(X is 0, 1, 2,..., N−2, N−1, y is 0, 1, 2,..., N−2, N−1, and x ≠ y. .)

Next, not only θ11 and θ12 but also design requirements for λ and δ will be described. It is only necessary to set a certain value for λ. As a requirement, it is necessary to give a requirement for δ. Therefore, a method for setting δ when λ is 0 radians will be described.

  In this case, as in the case of the method of changing the precoding weight in the 4-slot period, if π / 2 radians ≦ | δ | ≦ π radians with respect to δ, particularly in the LOS environment, good reception quality can be obtained. Can be obtained.

  In symbol numbers Ni to Ni + N−1, there are two points q each having bad reception quality, and therefore there are 2N points. In order to obtain good characteristics in the LOS environment, these 2N points are all preferably different solutions. In this case, in addition to <condition # 3>, the condition <condition # 4> is required.

In addition, the phase of these 2N points should exist uniformly. (Because the direct wave phase in each receiver is likely to have a uniform distribution)
As described above, when the transmission apparatus of the MIMO transmission system transmits a plurality of modulated signals from a plurality of antennas, the precoding weight is switched over time, and the switching is regularly performed, so that a direct wave is dominant in the LOS environment. As compared with the conventional spatial multiplexing MIMO transmission, the effect of improving the transmission quality can be obtained.

  In this embodiment, the configuration of the receiving apparatus is as described in Embodiment 1. In particular, the configuration of the receiving apparatus has been described by limiting the number of antennas, but the number of antennas has increased. Can also be implemented in the same manner. That is, the number of antennas in the receiving apparatus does not affect the operation and effect of the present embodiment. In the present embodiment, the error correction code is not limited as in the first embodiment.

  Also, in this embodiment, the method of changing the precoding weight on the time axis has been described in comparison with the first embodiment. However, as described in the first embodiment, the multi-carrier transmission scheme is used, and the frequency axis and frequency -By arranging symbols on the time axis, the same method can be implemented even if the precoding weight is changed. In the present embodiment, symbols other than data symbols, for example, pilot symbols (preamble, unique word, etc.), control information symbols, etc. may be arranged in any manner.

(Embodiment 3)
In the first and second embodiments, the case where the amplitudes of the elements of the precoding weight matrix are equal in the method of regularly switching the precoding weights has been described. However, this embodiment satisfies this condition. An example that does not exist will be described.
For comparison with the second embodiment, a case where the precoding weight is changed in an N slot period will be described. When considered in the same manner as in the first embodiment and the second embodiment, the following processing is performed on the symbol number. However, β is a positive real number, and β ≠ 1.
For symbol number Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

Therefore, r1 and r2 are expressed as follows.
For symbol number Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

At this time, it is assumed that only direct wave components exist in the channel elements h ₁₁ (t), h ₁₂ (t), h ₂₁ (t), and h ₂₂ (t), and the amplitude components of the direct wave components are Assume that all are equal and that there is no variation in time. Then, Formula (86)-Formula (89) can be expressed as follows.
For symbol number Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

However, in Formula (90)-Formula (93), A shall be a real number and q shall be a complex number. Formulas (90) to (93) are expressed as follows.
For symbol number Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

Then, when q is expressed as follows, it becomes impossible to obtain any signal of s1 and s2.
For symbol number Ni (i is an integer greater than or equal to 0):

For symbol number Ni + 1:

・
・
・
When symbol number Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number Ni + N-1:

At this time, if q has the same solution in symbol numbers N to Ni + N−1, the channel component of the direct wave does not vary greatly, so that it is impossible to obtain good reception quality at any symbol number. Therefore, it is difficult to obtain error correction capability even if an error correction code is introduced. Therefore, in order for q not to have the same solution, focusing on the solution that does not include δ among the two solutions of q, the following conditions are necessary from equations (98) to (101): It becomes.

(X is 0, 1, 2,..., N−2, N−1, y is 0, 1, 2,..., N−2, N−1, and x ≠ y. .)

Next, not only θ11 and θ12 but also design requirements for λ and δ will be described. It is only necessary to set a certain value for λ. As a requirement, it is necessary to give a requirement for δ. Therefore, a method for setting δ when λ is 0 radians will be described.

  In this case, as in the case of the method of changing the precoding weight in the 4-slot period, if π / 2 radians ≦ | δ | ≦ π radians with respect to δ, particularly in the LOS environment, good reception quality can be obtained. Can be obtained.

  In symbol numbers Ni to Ni + N−1, there are two points q each having bad reception quality, and therefore there are 2N points. In order to obtain good characteristics in the LOS environment, these 2N points are all preferably different solutions. In this case, in addition to <Condition # 5>, β is a positive real number, and considering that β ≠ 1, the condition <Condition # 6> is required.

  As described above, when the transmission apparatus of the MIMO transmission system transmits a plurality of modulated signals from a plurality of antennas, the precoding weight is switched over time, and the switching is regularly performed, so that the LOS environment in which the direct wave is dominant. In this case, it is possible to obtain an effect that the transmission quality is improved as compared with the conventional spatial multiplexing MIMO transmission.

  In this embodiment, the configuration of the receiving apparatus is as described in Embodiment 1. In particular, the configuration of the receiving apparatus has been described by limiting the number of antennas, but the number of antennas has increased. Can also be implemented in the same manner. That is, the number of antennas in the receiving apparatus does not affect the operation and effect of the present embodiment. In the present embodiment, the error correction code is not limited as in the first embodiment.

  Also, in this embodiment, the method of changing the precoding weight on the time axis has been described in comparison with the first embodiment. However, as described in the first embodiment, the multi-carrier transmission scheme is used, and the frequency axis and frequency -By arranging symbols on the time axis, the same method can be implemented even if the precoding weight is changed. In the present embodiment, symbols other than data symbols, for example, pilot symbols (preamble, unique word, etc.), control information symbols, etc. may be arranged in any manner.

(Embodiment 4)
In Embodiment 3, in the method of switching precoding weights regularly, the amplitude of each element of the precoding weight matrix is set to 1 and β
The two cases have been described as an example.

  Here,

Is ignored.

Subsequently, an example in which the value of β is switched in a slot will be described.
For comparison with the third embodiment, a case where the precoding weight is changed in a 2 × N slot period will be described.
When considered in the same manner as in the first embodiment, the second embodiment, and the third embodiment, the following processing is performed on the symbol number. However, β is a positive real number, and β ≠ 1. Α is a positive real number, and α ≠ β.
For symbol number 2Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number 2Ni + 1:

・
・
・
When symbol number 2Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + N-1:

When the symbol number is 2Ni + N (i is an integer of 0 or more):

However, j is an imaginary unit.
For symbol number 2Ni + N + 1:

・
・
・
When symbol number 2Ni + N + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + 2N-1:

Therefore, r1 and r2 are expressed as follows.
For symbol number 2Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number 2Ni + 1:

・
・
・
When symbol number 2Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + N-1:

When the symbol number is 2Ni + N (i is an integer of 0 or more):

However, j is an imaginary unit.
For symbol number 2Ni + N + 1:

・
・
・
When symbol number 2Ni + N + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + 2N-1:

At this time, it is assumed that only direct wave components exist in the channel elements h ₁₁ (t), h ₁₂ (t), h ₂₁ (t), and h ₂₂ (t), and the amplitude components of the direct wave components are Assume that all are equal and that there is no variation in time. Then, Formula (110)-Formula (117) can be expressed as follows.
For symbol number 2Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number 2Ni + 1:

・
・
・
When symbol number 2Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + N-1:

When the symbol number is 2Ni + N (i is an integer of 0 or more):

However, j is an imaginary unit.
For symbol number 2Ni + N + 1:

・
・
・
When symbol number 2Ni + N + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + 2N-1:

However, in Formula (118)-Formula (125), A shall be a real number and q shall be a complex number. Formulas (118) to (125) are expressed as follows.
For symbol number 2Ni (i is an integer greater than or equal to 0):

However, j is an imaginary unit.
For symbol number 2Ni + 1:

・
・
・
When symbol number 2Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + N-1:

When the symbol number is 2Ni + N (i is an integer of 0 or more):

However, j is an imaginary unit.
For symbol number 2Ni + N + 1:

・
・
・
When symbol number 2Ni + N + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + 2N-1:

Then, when q is expressed as follows, it becomes impossible to obtain any signal of s1 and s2.
For symbol number 2Ni (i is an integer greater than or equal to 0):

For symbol number 2Ni + 1:

・
・
・
When symbol number 2Ni + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + N-1:

When the symbol number is 2Ni + N (i is an integer of 0 or more):

For symbol number 2Ni + N + 1:

・
・
・
When symbol number 2Ni + N + k (k = 0, 1,..., N−1):

・
・
・
For symbol number 2Ni + 2N-1:

At this time, if q has the same solution in symbol numbers 2N to 2Ni + N−1, the channel component of the direct wave does not change greatly, so that it is impossible to obtain good reception quality at any symbol number. Therefore, it is difficult to obtain error correction capability even if an error correction code is introduced. Therefore, in order for q not to have the same solution, focusing on the solution that does not include δ among the two solutions of q, from Equation (134) to Equation (141) and α ≠ β, <Condition # 7> or <Condition # 8> is required.

At this time, <Condition # 8> is the same as the condition described in Embodiments 1 to 3, but <Condition # 7> is set to 2 of q because α ≠ β. Of the two solutions, the solution that does not include δ will have a different solution.

  Next, not only θ11 and θ12 but also design requirements for λ and δ will be described. It is only necessary to set a certain value for λ. As a requirement, it is necessary to give a requirement for δ. Therefore, a method for setting δ when λ is 0 radians will be described.

  In this case, as in the case of the method of changing the precoding weight in the 4-slot period, if π / 2 radians ≦ | δ | ≦ π radians with respect to δ, particularly in the LOS environment, good reception quality can be obtained. Can be obtained.

  In the symbol numbers 2Ni to 2Ni + 2N−1, there are two points q each having bad reception quality, and therefore there are 4N points. In order to obtain good characteristics in the LOS environment, all these 4N points should be different solutions. At this time, focusing on the amplitude, since α ≠ β with respect to <Condition # 7> or <Condition # 8>, the following condition is necessary.

  As described above, when the transmission apparatus of the MIMO transmission system transmits a plurality of modulated signals from a plurality of antennas, the precoding weight is switched over time, and the switching is regularly performed, so that the LOS environment in which the direct wave is dominant. In this case, it is possible to obtain an effect that the transmission quality is improved as compared with the conventional spatial multiplexing MIMO transmission.

  In this embodiment, the configuration of the receiving apparatus is as described in Embodiment 1. In particular, the configuration of the receiving apparatus has been described by limiting the number of antennas, but the number of antennas has increased. Can also be implemented in the same manner. That is, the number of antennas in the receiving apparatus does not affect the operation and effect of the present embodiment. In the present embodiment, the error correction code is not limited as in the first embodiment.

  Also, in this embodiment, the method of changing the precoding weight on the time axis has been described in comparison with the first embodiment. However, as described in the first embodiment, the multi-carrier transmission scheme is used, and the frequency axis and frequency -By arranging symbols on the time axis, the same method can be implemented even if the precoding weight is changed. In the present embodiment, symbols other than data symbols, for example, pilot symbols (preamble, unique word, etc.), control information symbols, etc. may be arranged in any manner.

(Embodiment 5)
In Embodiments 1 to 4, the method of switching precoding weights regularly has been described, but in the present embodiment, a modified example thereof will be described.

  In Embodiments 1 to 4, the method of switching the precoding weight regularly as shown in FIG. 6 has been described. In the present embodiment, a method for regularly switching precoding weights different from that in FIG. 6 will be described.

Similar to FIG. 6, FIG. 22 shows a diagram relating to a switching method different from that in FIG. 6 in a method of switching four different precoding weights (matrixes). In FIG. 22, four different precoding weights (matrixes) are represented as W ₁ , W ₂ , W ₃ , and W ₄ . (For example, W ₁ is a precoding weight (matrix) in equation (37), W ₂ is a precoding weight (matrix) in equation (38), W ₃ is a precoding weight (matrix) in equation (39), W ₄ Is a precoding weight (matrix) in the equation (40).) The same reference numerals are given to components that operate in the same manner as in FIGS. In FIG. 22, the unique part is
The first cycle 2201, the second cycle 2202, the third cycle 2203,... Are each composed of 4 slots.
In 4 slots, a different precoding weight matrix for each slot, that is, W ₁ , W ₂ , W ₃ , and W ₄ is used once.
In the first period 2201, the second period 2202, the third period 2203,..., The order of W ₁ , W ₂ , W ₃ , W ₄ is not necessarily the same.
It is. In order to realize this, precoding weight matrix generation section 2200 receives as input a signal related to a weighting method, and outputs information 2210 related to precoding weights according to the order in each period. The weighting / synthesizing unit 600 receives the signal and s1 (t) and s2 (t) as inputs, performs weighting / synthesizing, and outputs z1 (t) and z2 (t).

  FIG. 23 shows a weighted synthesis method with respect to the above-described precoding method. 23 differs from FIG. 22 in that a rearrangement unit is arranged after the weighting synthesis unit and signals are rearranged to realize the same method as in FIG.

23, a precoding weight generation unit 2200 receives information 315 on the weighting method as an input, and precoding weights W ₁ , W ₂ , W ₃ , W _4, W ₁ , W ₂ , W ₃ , W ₄ ,. Output precoding weight information 2210 in the order of Therefore, the weighting synthesis unit 600 uses precoding weights in the order of precoding weights W ₁ , W ₂ , W ₃ , W _4, W ₁ , W ₂ , W ₃ , W ₄ ,. Signals 2300A and 2300B are output.

Reordering section 2300 receives precoded signals 2300A and 2300B as input, and outputs the signals after precoding so that the first cycle 2201, the second cycle 2202, and the third cycle 2203 in FIG. Rearrangement is performed for 2300A and 2300B, and z1 (t) and z2 (t) are output.
In the above description, the switching cycle of the precoding weight has been described as 4 in order to compare with FIG. 6. However, as in Embodiments 1 to 4, the switching can be performed in the same way even when the cycle is not 4. Is possible.
In Embodiments 1 to 4 and the above-described precoding method, the values of δ and β are the same for each slot within the period. However, the values of δ and β for each slot are described. May be switched.

  As described above, when the transmission apparatus of the MIMO transmission system transmits a plurality of modulated signals from a plurality of antennas, the precoding weight is switched over time, and the switching is regularly performed, so that the LOS environment in which the direct wave is dominant. In this case, it is possible to obtain an effect that the transmission quality is improved as compared with the conventional spatial multiplexing MIMO transmission.

  In this embodiment, the configuration of the receiving apparatus is as described in Embodiment 1. In particular, the configuration of the receiving apparatus has been described by limiting the number of antennas, but the number of antennas has increased. Can also be implemented in the same manner. That is, the number of antennas in the receiving apparatus does not affect the operation and effect of the present embodiment. In the present embodiment, the error correction code is not limited as in the first embodiment.

  Also, in this embodiment, the method of changing the precoding weight on the time axis has been described in comparison with the first embodiment. However, as described in the first embodiment, the multi-carrier transmission scheme is used, and the frequency axis and frequency -By arranging symbols on the time axis, the same method can be implemented even if the precoding weight is changed. In the present embodiment, symbols other than data symbols, for example, pilot symbols (preamble, unique word, etc.), control information symbols, etc. may be arranged in any manner.

(Embodiment 6)
In Embodiments 1 to 4, the method for switching the precoding weights regularly has been described. In this embodiment, the precoding weights are regularly changed again, including the contents described in Embodiments 1 to 4. A method of switching will be described.

  Here, first, a method for designing a precoding matrix of a spatial multiplexing type 2x2 MIMO system to which precoding without feedback from a communication partner is applied in consideration of the LOS environment will be described.

FIG. 30 shows a spatial multiplexing type 2 × 2 MIMO system model to which precoding without feedback from a communication partner is applied. The information vector z is encoded and interleaved. Then, an encoded bit vector u (p) = (u ₁ (p), u ₂ (p)) is obtained as an interleave output (p is a slot time). Here, u _i (p) = (u _i1 (p)..., U _ih (p)) (h: number of transmission bits per symbol). If the signal after modulation (after mapping) is s (p) = (s ₁ (p), s ₂ (p)) ^T , and the precoding matrix is F (p), the signal after precoding x (p) = (x ₁ (p), x ₂ (p)) ^T is expressed by the following equation.

Therefore, if the received vector is y (p) = (y ₁ (p), y ₂ (p)) ^T , the following expression is obtained.

At this time, H (p) is a channel matrix, n (p) = (n ₁ (p), n ₂ (p)) ^T is a noise vector, n _i (p) is an average value 0, and variance σ ² iid complex Gaussian noise. And when the rice factor is K, the above equation can be expressed as follows.

At this time, H _d (p) is a channel matrix of the direct wave component, and H _s (p) is a channel matrix of the scattered wave component. Therefore, the channel matrix H (p) is expressed as follows.

In Expression (145), it is assumed that the environment of the direct wave is uniquely determined by the positional relationship between the communication devices, and the channel matrix H _d (p) of the direct wave component does not vary with time. In addition, the channel matrix H _d (p) of the direct wave component is a channel matrix regular matrix of the direct wave component because there is a high possibility that the distance between the transmitter and receiver is sufficiently long compared to the transmission antenna interval. And Therefore, the channel matrix H _d (p) is represented as follows.

  Here, A is a positive real number, and q is a complex number. The following describes a precoding matrix design method for a spatially multiplexed 2x2 MIMO system that applies precoding that does not have feedback from the communication partner in consideration of the LOS environment.

  From the equations (144) and (145), it is difficult to analyze in a state including a scattered wave. Therefore, it is difficult to obtain an appropriate precoding matrix without feedback in a state including a scattered wave. In addition, in the NLOS environment, data reception quality is less degraded than in the LOS environment. Therefore, an appropriate precoding matrix design method without feedback in the LOS environment (precoding matrix of a precoding method for switching a precoding matrix with time) will be described.

  As described above, since it is difficult to analyze in a state including a scattered wave from equations (144) and (145), an appropriate precoding matrix is obtained in a channel matrix including a component of only a direct wave. To. Therefore, consider a case in which the channel matrix includes a component of only a direct wave in Expression (144). Therefore, from the formula (146), it can be expressed as follows.

  Here, a unitary matrix is used as the precoding matrix. Therefore, the precoding matrix is expressed as follows.

  At this time, λ is a fixed value. Therefore, Formula (147) can be expressed as follows.

As can be seen from equation (149), when the receiver performs a linear operation such as ZF (zero forcing) or MMSE (minimum mean squared error), the bit transmitted by s ₁ (p), s ₂ (p) is determined. I can't do it. From this, iterative APP (or iterative Max-log APP) or APP (or Max-log APP) as described in the first embodiment is performed (hereinafter referred to as ML (Maximum Likelihood) operation), and s The log likelihood ratio of each bit transmitted in ₁ (p), s ₂ (p) is obtained, and decoding in the error correction code is performed. Therefore, a method for designing a precoding matrix without an appropriate feedback in an LOS environment for a receiver that performs ML calculation will be described.

Consider precoding in equation (149). The right side and the left side of the first row are multiplied by e ^−jΨ , and similarly, the right side and the left side of the second row are multiplied by e ^−jΨ . Then, it is expressed as the following formula.

e ^-jΨ y ₁ (p), e ^-jΨ y ₂ (p), e ^-jΨ q are ^redefined as y ₁ (p), y ₂ (p), q, respectively, and e ^-jΨ n (p ) = (e ^-jΨ n ₁ (p), e ^-jΨ n ₂ (p)) ^T , and e ^-jΨ n ₁ (p), e ^-jΨ n ₂ (p) is the mean of 0 and the variance σ ² Since it becomes iid (independent identically distributed) complex Gaussian noise, e ^−jΨ n (p) is ^{redefined as} n (p). Then, even if Expression (150) is changed to Expression (151), generality is not lost.

  Next, the equation (151) is transformed into the equation (152) so that the equation (151) can be easily understood.

At this time, when the minimum value of the Euclidean distance between the reception signal point and the reception candidate signal point is d _min ² , d _min ² is a bad point that takes a minimum value of zero and is transmitted at s ₁ (p) There are two qs that are in a bad state where all bits or all bits transmitted in s ₂ (p) are lost.

In formula (152), s ₁ (p) does not exist:

In formula (152), s ₂ (p) does not exist:

(Hereinafter, q satisfying equations (153) and (154) will be referred to as “reception bad points of s ₁ and s ₂ ”, respectively)
When the expression (153) is satisfied, all the bits transmitted by s ₁ (p) are lost, and therefore the reception log likelihood ratio of all the bits transmitted by s ₁ (p) cannot be obtained. When all of the bits transmitted by s ₂ (p) are lost, the received log likelihood ratio of all the bits transmitted by s ₂ (p) cannot be obtained.

  Here, consider a broadcast / multicast communication system in which the precoding matrix is not switched. At this time, consider a system model in which there is a base station that transmits a modulated signal using a precoding scheme that does not switch the precoding matrix, and there are a plurality (Γ) of terminals that receive the modulated signal transmitted by the base station.

  The state of direct waves between the base station and the terminal is considered to change little over time. Then, from the equations (153) and (154), a terminal that is in a position that satisfies the condition of the equation (155) or the equation (156) and that is in an LOS environment with a large Rice factor is said to have deteriorated data reception quality. There is a possibility of falling into a phenomenon. Therefore, in order to improve this problem, it is necessary to switch the precoding matrix temporally.

Accordingly, a method of switching the precoding matrix regularly with a time period of N slots (hereinafter referred to as a precoding hopping method) is considered.
For the time period N slots, N types of precoding matrices F [i] based on the equation (148) are prepared (i = 0, 1,..., N−1). At this time, the precoding matrix F [i] is expressed as follows.

Here, it is assumed that α does not change with time, and λ also does not change with time (may be changed).
Then, as in the first embodiment, the signal after precoding in the equation (142) of time (time) N × k + i (k is an integer equal to or greater than 0, i = 0, 1,..., N−1) The precoding matrix used to obtain x (p = N × k + i) is F [i]. The same applies to the following.

  At this time, based on the equations (153) and (154), the following precoding hopping precoding matrix design conditions are important.

According to <Condition # 10>, in all the Γ terminals, the number of slots having the poor reception of s ₁ is 1 slot or less at N in the time period. Therefore, the log likelihood ratio of bits transmitted in s ₁ (p) over N−1 slots can be obtained. Similarly, the <Condition # 11>, in all Γ number of terminals, in N time period, the slot taking the reception poor point of s ₂ is less than or equal to 1 slot. Therefore, the log likelihood ratio of bits transmitted in s ₂ (p) over N−1 slots can be obtained.

In this way, by giving the precoding matrix design criteria of <Condition # 10> and <Condition # 11>, the number of bits from which the log likelihood ratio of the bits transmitted in s ₁ (p) can be obtained, and s _{2 By} guaranteeing the number of bits that can obtain the log-likelihood ratio of the bits transmitted in (p) to a certain number or more in all Γ terminals, data in the LOS environment with a large Rice factor in all Γ terminals Consider improving the degradation of reception quality.

Hereinafter, an example of a precoding matrix in the precoding hopping method will be described.
The probability density distribution of the phase of the direct wave can be considered as a uniform distribution of [0 2π]. Therefore, the probability density distribution of the phase of q in the equations (151) and (152) can also be considered to be a uniform distribution of [0 2π]. Therefore, in the same LOS environment in which only the phase of q is different, the following conditions are given as conditions for giving the Γ terminals as fair a data reception quality as possible.
<Condition # 12>
When the precoding hopping method of time period N slots is used, the reception bad points of s ₁ are arranged so as to have a uniform distribution with respect to the phase in N within the time period, and the reception bad points of s ₂ are It arrange | positions so that it may become uniform distribution with respect to a phase.

Therefore, an example of a precoding matrix in the precoding hopping method based on <condition # 10> to <condition # 12> will be described. Let α = 1.0 of the precoding matrix in equation (157).
(Example # 5)
In order to satisfy <Condition # 10> to <Condition # 12> with a time period N = 8, a precoding matrix in a precoding hopping method with a time period N = 8 as shown in the following equation is given.

However, j is an imaginary unit, and i = 0, 1,. Expression (161) may be given instead of expression (160) (λ, θ ₁₁ [i] shall not change over time (may change)).

Therefore, the poor reception points of s ₁ and s ₂ are as shown in FIGS. 31 (a) and 31 (b). (In FIG. 31, the horizontal axis is the real axis and the vertical axis is the imaginary axis.) Also, instead of the equations (160) and (161), the equations (162) and (163) may be given (i = 0, 1,..., 7) (λ and θ ₁₁ [i] shall not change over time (may change)).

Next, in the same LOS environment where only the phase of q is different from the condition 12, the following is given as a condition for giving the data reception quality as fair as possible to the Γ terminals.
<Condition # 13>
When using the precoding hopping method of time period N slots,

In addition, in N within the time period, the reception bad points of s ₁ are arranged in a uniform distribution with respect to the phase and the bad reception points of s ₂ with respect to the phase.
Therefore, an example of a precoding matrix in the precoding hopping method based on <Condition # 10>, <Condition # 11>, and <Condition # 13> will be described. Let α = 1.0 of the precoding matrix in equation (157).
(Example # 6)
A precoding matrix in a precoding hopping method with a time period N = 4 and a time period N = 4 as shown in the following equation is given.

However, j is an imaginary unit and i = 0, 1, 2, and 3. Instead of the equation (165), the equation (166) may be given (λ, θ ₁₁ [i] does not change with time (may change)).

Therefore, the poor reception points of s ₁ and s ₂ are as shown in FIG. (In FIG. 32, the horizontal axis is the real axis, and the vertical axis is the imaginary axis.) Also, instead of the expressions (165) and (166), the expressions (167) and (168) may be given (i = 0,1,2,3) (λ and θ ₁₁ [i] shall not change over time (may change)).

Next, a precoding hopping method using a non-unitary matrix will be described.
Based on equation (148), the precoding matrix treated in this study is expressed as follows.

  Then, the expressions corresponding to the expressions (151) and (152) are expressed as the following expressions.

At this time, there are two qs where the minimum value d _min ² of the Euclidean distance between the reception signal point and the reception candidate signal point is zero.
In formula (171), s ₁ (p) does not exist:

In formula (171), s ₂ (p) does not exist:

  In the precoding hopping method of time period N, N types of precoding matrices F [i] are expressed as follows with reference to Equation (169).

  Here, α and δ are assumed not to change with time. At this time, the following precoding hopping precoding matrix design conditions are given based on equations (34) and (35).

(Example # 7)
It is assumed that α of the precoding matrix in Expression (174) is 1.0. In order to satisfy <Condition # 12>, <Condition # 14>, and <Condition # 15> with a time period N = 16, a precoding matrix in a precoding hopping method with a time period N = 8 as shown in the following equation: give.

  When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Further, as a precoding matrix different from the equations (177) and (178), it can be given as follows.
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Therefore, the poor reception points of s ₁ and s ₂ are as shown in FIGS.
(In FIG. 33, the horizontal axis is the real axis and the vertical axis is the imaginary axis.) Also, instead of the equations (177), (178), (179), and (180), precoding is performed as follows. A matrix may be given.

  When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Or
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

(In formulas (177) to (184), 7π / 8 may be set to −7π / 8.)
Next, in the same LOS environment where only the phase of q is different from <Condition # 12>, the following is given as a condition for giving the Γ number of terminals as fair a data reception quality as possible: .
<Condition # 16>
When using the precoding hopping method of time period N slots,

In addition, in N within the time period, the reception bad points of s ₁ are arranged in a uniform distribution with respect to the phase and the bad reception points of s ₂ with respect to the phase.
Therefore, an example of a precoding matrix in the precoding hopping method based on <Condition # 14>, <Condition # 15>, and <Condition # 16> will be described. It is assumed that α of the precoding matrix in Expression (174) is 1.0.
(Example # 8)
A precoding matrix in a precoding hopping method with a time period N = 8 and a time period N = 8 as shown in the following equation is given.

However, i = 0,1, ..., 7.
Further, as a precoding matrix different from the equation (186), it can be given as follows (i = 0, 1,..., 7) (λ, θ ₁₁ [i] is assumed not to change with time ( May change)).

Therefore, the poor reception points of s ₁ and s ₂ are as shown in FIG. Further, instead of the equations (186) and (187), a precoding matrix may be given as follows (i = 0, 1,..., 7) (λ, θ ₁₁ [i] changes with time. (Do not change.)

  Or

(Also, in Formula (186) to Formula (189), 7π / 8 may be set to −7π / 8.)
Next, a precoding hopping method different from (Example # 7) and (Example # 8) in which α ≠ 1 in the precoding matrix of Expression (174) and considering the distance point in the complex plane between the reception poor points Think.

Here, the precoding hopping method of time period N in Expression (174) is handled. At this time, according to <Condition # 14>, reception of s ₁ is received at N in the time period by all Γ terminals. Slots that take inferiority will be 1 slot or less. Therefore, the log likelihood ratio of bits transmitted in s ₁ (p) over N−1 slots can be obtained. Similarly, according to <Condition # 15>, in all Γ terminals, the number of slots having the reception poor point of s ₂ is 1 slot or less at N in the time period. Therefore, the log likelihood ratio of bits transmitted in s ₂ (p) over N−1 slots can be obtained.

Therefore, it can be seen that the larger the time period N, the larger the number of slots in which the log likelihood ratio can be obtained.
By the way, since the actual channel model is affected by the scattered wave component, when the time period N is fixed, the reception quality of data is improved when the minimum distance on the complex plane of the reception poor point is as large as possible. There seems to be a possibility. Therefore, consider a precoding hopping method in which α ≠ 1 in (Example # 7) and (Example # 8) and (Example # 7) and (Example # 8) are improved. First, a precoding method improved from (Example # 8) that facilitates understanding will be described.
(Example # 9)
From the equation (186), the precoding matrix in the precoding hopping method with the time period N = 8 improved from (Example # 7) is given by the following equation.

However, i = 0,1, ..., 7. Further, as a precoding matrix different from the equation (190), it can be given as follows (i = 0, 1,..., 7) (λ, θ ₁₁ [i] shall not change with time ( May change)).

  Or

  Or

  Or

  Or

  Or

  Or

Accordingly, the poor reception points of s ₁ and s ₂ are shown in FIG. 35 (a) when α <1.0 and as shown in FIG. 35 (b) when α> 1.0.
(i) When α <1.0 When α <1.0, the minimum distance in the complex plane of the reception poor point is the distance between the reception poor points # 1 and # 2 (d _{# 1, # 2} ) and the reception poor point # 1. And # 3 (d _{# 1, # 3} ), min {d _{# 1, # 2} , d _{# 1, # 3} } is expressed. At this time, the relationship between α and d _{# 1, # 2} and d _{# 1, # 3} is shown in FIG. And α that maximizes min {d _{# 1, # 2} , d _{# 1, # 3} } is

It becomes. Min {d _{# 1, # 2} , d _{# 1, # 3} } at this time is

It becomes. Therefore, a precoding method in which α is given by equation (198) in equations (190) to (197) is effective. However, setting the value of α as equation (198) is one suitable method for obtaining good data reception quality. However, even if α is set to take a value close to Equation (198), there is a possibility that good data reception quality can be obtained in the same manner. Therefore, the set value of α is not limited to the equation (198).

(ii) When α> 1.0 When α> 1.0, the minimum distance in the complex plane of the reception inferior point is the distance between the reception inferior points # 4 and # 5 (d _{# 4, # 5} ) and the reception inferior point # 4. And # 6 (d _{# 4, # 6} ), min {d _{# 4, # 5} , d _{# 4, # 6} } is expressed. At this time, the relationship between α and d _{# 4, # 5} and d _{# 4, # 6} is shown in FIG. And α that maximizes min {d _{# 4, # 5} , d _{# 4, # 6} } is

It becomes. Min {d _{# 4, # 5} , d _{# 4, # 6} } at this time is

It becomes. Therefore, the precoding method in which α is given by equation (200) in equations (190) to (197) is effective. However, setting the value of α as equation (200) is one suitable method for obtaining good data reception quality. However, even if α is set to take a value close to Equation (200), there is a possibility that good data reception quality can be obtained similarly. Therefore, the set value of α is not limited to the formula (200).
(Example # 10)
The precoding matrix in the precoding hopping method with improved time period N = 16 from the examination of (Example # 9) can be given by the following equation (λ, θ ₁₁ [i] is temporally (It may change).

  When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Or
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Or
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Or
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Or
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Or
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Or
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

Or
When i = 0,1, ..., 7:

  When i = 8,9, ..., 15:

However, α is suitable for obtaining good data reception quality when the equation (198) or the equation (200) is satisfied. At this time, the poor reception point of s ₁ is shown in FIGS. 38A and 38B when α <1.0, and FIGS. 39A and 39B when α> 1.0.

  In the present embodiment, the configuration method of N different precoding matrices for the precoding hopping method of time period N has been described. At this time, F [0], F [1], F [2],..., F [N-2], F [N-1] are prepared as N different precoding matrices. However, since the present embodiment is described by taking the case of the single carrier transmission method as an example, F [0], F [1], F [2],... In the time axis (or frequency axis) direction. , F [N-2], F [N-1] are described in this order. However, the present invention is not limited to this, and N different precoding matrices F [0] generated in the present embodiment are not necessarily limited to this. , F [1], F [2],..., F [N-2], F [N-1] can be applied to a multicarrier transmission scheme such as an OFDM transmission scheme. As for the application method in this case, as in the first embodiment, the precoding weight can be changed by arranging symbols on the frequency axis and the frequency-time axis. Although described as a precoding hopping method of time period N, the same effect can be obtained even when N different precoding matrices are used at random, that is, a regular period is not necessarily used. It is not necessary to use N different precoding matrices to have.

Example # 5 to Example # 10 are shown based on <Condition # 10> to <Condition # 16>. In order to increase the switching cycle of the precoding matrix, for example, from Example # 5 to Example # 10, a plurality of examples An example may be selected, and a long-period precoding matrix switching method may be realized using the precoding matrix shown in the selected example. For example, using the precoding matrix shown in Example # 7 and the precoding matrix shown in Example # 10, a long-period precoding matrix switching method is realized. In this case, it does not necessarily follow from <Condition # 10> to <Condition # 16>. (<Condition # 10> Formula (158), <Condition # 11> Formula (159), <Condition # 13> Formula (164), <Condition # 14> Formula (175), <Condition # 15> In the equation (176), the condition that “all x, all y” is “existence x, existence y” is important to give good reception quality. From another viewpoint, in the precoding matrix switching method of period N (N is a large natural number), it is preferable that any of the precoding matrices of Example # 5 to Example # 10 is included. There is a high possibility of giving a good reception quality.
(Embodiment 7)
In this embodiment, a configuration of a receiving apparatus that receives a modulated signal transmitted by the transmission method that regularly switches the precoding matrix described in Embodiments 1 to 6 will be described.

  In the first embodiment, a transmission apparatus that transmits a modulated signal using a transmission method that regularly switches a precoding matrix transmits information on the precoding matrix, and a reception apparatus uses it for a transmission frame based on the information. A method has been described in which regular precoding matrix switching information is obtained, decoding of precoding and detection are performed, a log likelihood ratio of transmission bits is obtained, and then error correction decoding is performed.

In the present embodiment, a configuration of a receiving apparatus different from the above and a precoding matrix switching method will be described.
FIG. 40 shows an example of the configuration of the transmission apparatus according to the present embodiment, and components that operate in the same manner as in FIG. The encoder group (4002) receives the transmission bit (4001). At this time, as described in Embodiment 1, the encoder group (4002) holds a plurality of error correction code encoding units, and, for example, one encoding is performed based on the frame configuration signal 313. Any number of encoders of two encoders and four encoders will operate.

  When one encoder operates, the transmission bit (4001) is encoded to obtain a transmission bit after encoding, and the transmission bit after encoding is distributed to two systems and distributed. The encoder group (4002) outputs the bits (4003A) and the distributed bits (4003B).

  When two encoders operate, the transmission bit (4001) is divided into two (named as divided bits A and B), and the first encoder receives the divided bits A and performs encoding. The encoded bits are output as distributed bits (4003A). The second encoder receives the divided bits B as input, performs encoding, and outputs the encoded bits as distributed bits (4003B).

  When four encoders operate, the transmission bit (4001) is divided into four (named as divided bits A, B, C, and D), and the first encoder receives the divided bits A as input. Encoding is performed, and the encoded bit A is output. The second encoder receives the divided bits B, performs encoding, and outputs the encoded bits B. The third encoder receives the divided bits C, performs encoding, and outputs the encoded bits C. The fourth encoder receives the divided bits D, performs encoding, and outputs the encoded bits D. Then, the encoded bits A, B, C, and D are divided into distributed bits (4003A) and distributed bits (4003B).

  As an example, the transmission apparatus supports a transmission method as shown in Table 1 (Table 1A and Table 1B) below.

  As shown in Table 1, as the number of transmission signals (the number of transmission antennas), transmission of 1 stream signal and transmission of 2 stream signals are supported. The modulation scheme supports QPSK, 16QAM, 64QAM, 256QAM, and 1024QAM. In particular, when the number of transmission signals is 2, stream # 1 and stream # 2 can set modulation schemes separately. For example, in Table 1, “# 1: 256QAM, # 2: 1024QAM” is “ This indicates that the modulation method of stream # 1 is 256QAM, and the modulation method of stream # 2 is 1024QAM (others are also expressed in the same manner). Assume that three types of error correction coding schemes A, B, and C are supported. At this time, A, B, and C may all be different codes, A, B, and C may be different coding rates, and A, B, and C are coding methods having different block sizes. It may be.

In the transmission information of Table 1, each transmission information is assigned to each mode in which “number of transmission signals”, “modulation method”, “number of encoders”, and “error correction coding method” are defined. Therefore, for example, in the case of “the number of transmission signals: 2”, “modulation method: # 1: 1024QAM, # 2: 1024QAM”, “number of encoders: 4”, and “error correction coding method: C”, the transmission information is 01001101. Set. Then, the transmission device transmits transmission information and transmission data in the frame. When transmitting transmission data, especially when the “number of transmission signals” is 2, the “precoding matrix switching method” is used according to Table 1. In Table 1, five types of D, E, F, G, and H are prepared as “precoding matrix switching methods”, and any one of these five types is set according to Table 1. . At this time, as five different implementation methods,
-Prepare and implement 5 types of different precoding matrices.
This is realized by setting five different periods, for example, the period of D is 4, the period of E is 8, and so on.
Realize by using both different precoding matrices and different periods.
Etc. are considered.

  FIG. 41 shows an example of the frame configuration of the modulation signal transmitted by the transmission apparatus of FIG. 40. The transmission apparatus sets a mode for transmitting two modulation signals z1 (t) and z2 (t). It is also possible to set both modes for transmitting one modulated signal.

In FIG. 41, a symbol (4100) is a symbol for transmitting “transmission information” shown in Table 1. Symbols (4101_1 and 4101_2) are reference (pilot) symbols for channel estimation. Symbol (4102_1, 4103_1) is a symbol for data transmission transmitted with modulated signal z1 (t), symbol (4102_2, 4103_2) is a symbol for data transmission transmitted with modulated signal z2 (t), and symbol ( 4102_1) and symbols (4102_2) are transmitted using the same (common) frequency at the same time, and symbols (4103_1) and symbols (4103_2) are transmitted using the same (common) frequency at the same time. The symbols (4102_1, 4103_1) and symbols (4102_2, 4103_2) are pre-coded when the method of switching the precoding matrix regularly described in Embodiments 1 to 4 and Embodiment 6 is used. The symbols are obtained after the coding matrix calculation (therefore, as described in Embodiment 1, the structures of the streams s1 (t) and s2 (t) are as shown in FIG. 6).
Furthermore, in FIG. 41, a symbol (4104) is a symbol for transmitting the “transmission information” shown in Table 1. Symbol (4105) is a reference (pilot) symbol for channel estimation. Symbols (4106, 4107) are data transmission symbols transmitted with the modulated signal z1 (t). At this time, the number of transmission signals is one for the data transmission symbol transmitted with the modulated signal z1 (t). This means that precoding is not performed.

  Therefore, the transmission apparatus of FIG. 40 generates and transmits a modulated signal according to the frame configuration of FIG. 41 and Table 1. In FIG. 40, the frame configuration signal 313 includes information related to “number of transmission signals”, “modulation method”, “number of encoders”, and “error correction coding method” set based on Table 1. The encoding unit (4002), the mapping units 306A and B, and the weighting synthesis units 308A and B receive the frame configuration signal as input and set the “number of transmission signals”, “modulation scheme”, and “encoder” set based on Table 1 The operation based on “number” and “error correction coding method” is performed. Also, “transmission information” corresponding to the set “number of transmission signals”, “modulation scheme”, “number of encoders”, and “error correction coding method” is transmitted to the receiving apparatus.

  The configuration of the receiving apparatus can be represented in FIG. 7 as in the first embodiment. The difference from the first embodiment is that since the information in Table 1 is shared in advance by the transmission / reception device, the “transmission signal number” even if the transmission device does not transmit the information of the precoding matrix to be switched regularly. The transmission apparatus transmits “transmission information” corresponding to “modulation method”, “number of encoders”, and “error correction coding method”, and the reception apparatus obtains this information, so that the pre-switching regularly switched from Table 1. It is that the information of a coding matrix can be obtained. Therefore, in the receiving apparatus of FIG. 7, the control information decoding unit 709 obtains “transmission information” transmitted by the transmitting apparatus of FIG. 40, so that information on the precoding matrix that is regularly switched from the information corresponding to Table 1 is obtained. It is possible to obtain a signal 710 relating to information on the transmission method notified by the transmission apparatus including Therefore, when the number of transmission signals is 2, the signal processing unit 711 can perform detection based on the switching pattern of the precoding matrix, and can obtain a reception log likelihood ratio.

  In the above description, as shown in Table 1, “transmission information” is set for “number of transmission signals”, “modulation method”, “number of encoders”, and “error correction coding method”, and precoding is performed for this. Although the matrix switching method is set, it is not always necessary to set “transmission information” for “number of transmission signals”, “modulation method”, “number of encoders”, and “error correction coding method”. As shown in Table 2, “transmission information” may be set for “number of transmission signals” and “modulation scheme”, and a precoding matrix switching method may be set for this.

  Here, the “transmission information” and the setting method of the precoding matrix switching method are not limited to those in Tables 1 and 2, but the precoding matrix switching method includes “number of transmission signals”, “modulation scheme”, “sign” If the rules are determined in advance so as to switch based on the transmission parameters such as “number of encoders” and “error correction encoding method” (if the rules determined in advance by the transmitting device and the receiving device are shared) (That is, if the precoding matrix switching method is switched according to one of the transmission parameters (or any one of a plurality of transmission parameters)), the transmission apparatus can obtain information on the precoding matrix switching method. There is no need to transmit, and the receiving device determines the transmission parameter information, thereby switching the precoding matrix used by the transmitting device. Since the method can be determined, it is possible to perform accurate decoding, the detection. In Tables 1 and 2, a transmission method that regularly switches the precoding matrix when the number of transmission modulation signals is 2 is used. However, if the number of transmission modulation signals is 2 or more, the transmission method is regularly updated. A transmission method for switching a coding matrix can be applied.

  Therefore, if the transmission / reception apparatus shares a table regarding transmission parameters including information on the precoding switching method, the transmission apparatus does not transmit information on the precoding switching method and does not include information on the precoding switching method. Information is transmitted, and the receiving apparatus obtains this control information, so that the precoding switching method can be estimated.

  As described above, in the present embodiment, the transmitting apparatus does not transmit the direct information regarding the method for switching the precoding matrix regularly, and the receiving apparatus uses the “regular precoding matrix used by the transmitting apparatus”. The method of estimating information related to precoding in “Method of switching” has been described. As a result, the transmission apparatus does not transmit the direct information regarding the method of switching the precoding matrix regularly, so that it is possible to obtain the effect that the data transmission efficiency is improved accordingly.

  In the present embodiment, the embodiment for changing the precoding weight in the time axis has been described. However, as described in the first embodiment, the present embodiment can be used even when a multicarrier transmission scheme such as OFDM transmission is used. Embodiments can be similarly implemented.

  In particular, when the precoding switching method is changed only by the number of transmission signals, the reception device can perform the precoding switching method by obtaining information on the number of transmission signals transmitted by the transmission device. .

In this specification, it is conceivable that the transmission device is equipped with a communication / broadcasting device such as a broadcasting station, a base station, an access point, a terminal, a mobile phone, and the like. It is conceivable that the receiving device is equipped with a communication device such as a television, a radio, a terminal, a personal computer, a mobile phone, an access point, and a base station. In addition, the transmission device and the reception device in the present invention are devices having a communication function, and the devices provide some interface to a device for executing an application such as a television, a radio, a personal computer, or a mobile phone. It is also conceivable that the connection is possible.
In the present embodiment, symbols other than data symbols, for example, pilot symbols (preamble, unique word, postamble, reference symbol, etc.), control information symbols, etc., are arranged in any manner. Good. Here, the pilot symbol and the control information symbol are named, but any naming method may be used, and the function itself is important.

  The pilot symbol is, for example, a known symbol modulated by using PSK modulation in a transmitter / receiver (or the receiver may know the symbol transmitted by the transmitter by synchronizing the receiver). .), And the receiver uses this symbol to perform frequency synchronization, time synchronization, channel estimation (for each modulated signal) (estimation of CSI (Channel State Information)), signal detection, and the like. Become.

  In addition, the control information symbol is information (for example, a modulation method, an error correction coding method used for communication, a communication information symbol) that needs to be transmitted to a communication partner in order to realize communication other than data (such as an application). This is a symbol for transmitting an error correction coding method coding rate, setting information in an upper layer, and the like.

  The present invention is not limited to Embodiments 1 to 5 described above, and can be implemented with various modifications. For example, in the above embodiment, the case of performing as a communication device has been described. However, the present invention is not limited to this, and this communication method can also be performed as software.

  In the above description, the precoding switching method in the method of transmitting two modulated signals from two antennas has been described. However, the present invention is not limited to this, and precoding is performed on four mapped signals. In a method of generating one modulated signal and transmitting from four antennas, that is, a method of generating N modulated signals by performing precoding on N mapped signals and transmitting from N antennas Similarly, a precoding switching method for changing precoding weights (matrixes) can be similarly implemented.

  In the present specification, terms such as “precoding” and “precoding weight” are used, but any name may be used. In the present invention, the signal processing itself is important.

Different data may be transmitted by the streams s1 (t) and s2 (t), or the same data may be transmitted.
Both the transmitting antenna of the transmitting device and the receiving antenna of the receiving device may be configured by a plurality of antennas.

  For example, a program for executing the communication method may be stored in a ROM (Read Only Memory) in advance, and the program may be operated by a CPU (Central Processor Unit).

  In addition, a program for executing the communication method is stored in a computer-readable storage medium, the program stored in the storage medium is recorded in a RAM (Random Access Memory) of the computer, and the computer is operated according to the program. You may do it.

  Each configuration such as the above-described embodiments may be typically realized as an LSI (Large Scale Integration) which is an integrated circuit. These may be individually made into one chip, or may be made into one chip so as to include all or part of the configurations of the respective embodiments. Although referred to here as LSI, depending on the degree of integration, it may also be called IC (Integrated Circuit), system LSI, super LSI, or ultra LSI. Further, the method of circuit integration is not limited to LSI's, and implementation using dedicated circuitry or general purpose processors is also possible. An FPGA (Field Programmable Gate Array) that can be programmed after manufacturing the LSI, or a reconfigurable processor that can reconfigure the connection and setting of circuit cells inside the LSI may be used.

  Further, if integrated circuit technology comes out to replace LSI's as a result of the advancement of semiconductor technology or a derivative other technology, it is naturally also possible to carry out function block integration using this technology. There is a possibility of adaptation of biotechnology.

(Embodiment 8)
In this embodiment, an application example of the method for regularly switching the precoding weights described in Embodiments 1 to 4 and Embodiment 6 will be described here.

  FIG. 6 is a diagram related to the weighting method (precoding method) in the present embodiment. The weighting synthesis unit 600 is a weighting synthesis unit that integrates both the weighting synthesis units 308A and 308B of FIG. is there. As shown in FIG. 6, the stream s1 (t) and the stream s2 (t) correspond to the baseband signals 307A and 307B of FIG. 3, that is, the base according to the mapping of modulation schemes such as QPSK, 16QAM, and 64QAM. Band signal in-phase I and quadrature Q components. 6, the stream s1 (t) represents the signal with symbol number u as s1 (u), the signal with symbol number u + 1 as s1 (u + 1), and so on. Similarly, in the stream s2 (t), a signal with a symbol number u is represented as s2 (u), a signal with a symbol number u + 1 is represented as s2 (u + 1), and so on. Then, the weighting synthesis unit 600 receives the baseband signals 307A (s1 (t)) and 307B (s2 (t)) in FIG. 3 and the information 315 related to the weighting information, and performs a weighting method according to the information 315 related to the weighting information. Then, the signals 309A (z1 (t)) and 309B (z2 (t)) after the weighted synthesis in FIG. 3 are output.

At this time, for example, when the precoding matrix switching method of period N = 8 in Example 8 in Embodiment 6 is used, z1 (t) and z2 (t) are expressed as follows.
For symbol number 8i (i is an integer greater than or equal to 0):

However, j is an imaginary unit and k = 0.
For symbol number 8i + 1:

However, k = 1.
For symbol number 8i + 2:

However, k = 2.
For symbol number 8i + 3:

However, k = 3.
For symbol number 8i + 4:

However, k = 4.
For symbol number 8i + 5:

However, k = 5.
For symbol number 8i + 6:

However, k = 6.
For symbol number 8i + 7:

However, k = 7.
Here, the symbol number is described, but the symbol number may be considered as time (time). As described in other embodiments, for example, in Expression (225), z1 (8i + 7) and z2 (8i + 7) at time 8i + 7 are signals at the same time, and z1 (8i + 7) and z2 (8i + 7) Are transmitted by the transmitter using the same (common) frequency. That is, if the signal at time T is s1 (T), s2 (T), z1 (T), z2 (T), from some precoding matrix and s1 (T) and s2 (T), z1 (T) and z2 (T) is obtained, and z1 (T) and z2 (T) are transmitted by the transmission device using the same (common) frequency (at the same time (time)). When a multi-carrier transmission scheme such as OFDM is used, signals corresponding to (sub) carrier L and s1, s2, z1, and z2 at time T are s1 (T, L), s2 (T, L), and z1. If (T, L) and z2 (T, L), z1 (T, L) and z2 (T, L) are obtained from some precoding matrix and s1 (T, L) and s2 (T, L). , Z1 (T, L) and z2 (T, L) are transmitted by the transmitting device using the same (common) frequency (at the same time (time)).

At this time, as an appropriate value of α, there is formula (198) or formula (200).
In the present embodiment, a precoding switching method for increasing the period based on the precoding matrix of Equation (190) described above will be described.

  When the period of the precoding switching matrix is 8M, 8M different precoding matrices are represented as follows.

At this time, i = 0,1,2,3,4,5,6,7, k = 0,1,..., M-2, M-1.
For example, assuming that M = 2 and α <1, the reception poor point (◯) of s1 and the reception bad point (□) of s2 when k = 0 are as shown in FIG. Appears. Similarly, the reception poor point (◯) of s1 when k = 1 and the reception poor point (□) of s2 are expressed as shown in FIG. Thus, based on the precoding matrix of Expression (190), the reception inferior point is as shown in FIG. 42A, and each element in the second row of the matrix on the right side of Expression (190) is set to e ^jX Is used as a precoding matrix (see equation (226)) so that the reception poor point has a rotated reception bad point with respect to FIG. 42 (a) (see FIG. 42 (b)). . (However, the poor reception points in FIGS. 42A and ^42B do not overlap. In this way, even if e ^jX is multiplied, the bad reception points should not be overlapped. instead of multiplying e ^jX to each element of the second row of the right side of the matrix 190), a matrix obtained by multiplying the e ^jX to each element of the first row of the matrix of the right side of the equation (190) as a pre-coding matrix At this time, the precoding matrices F [0] to F [15] are expressed by the following equations.

However, i = 0,1,2,3,4,5,6,7 and k = 0,1.
Then, when M = 2, precoding matrices F [0] to F [15] are generated (the precoding matrices F [0] to F [15] are arranged in any order). In addition, the matrices F [0] to F [15] may be different from each other. Then, for example, precoding is performed using F [0] when the symbol number is 16i, precoding is performed using F [1] when the symbol number is 16i + 1, and F [h when the symbol number is 16i + h. ] To perform precoding (h = 0, 1, 2,..., 14, 15). (Here, as described in the previous embodiment, it is not always necessary to switch the precoding matrix regularly.)
Summarizing the above, the precoding matrix of period N is expressed by the following equation with reference to equations (82) to (85).

  At this time, since the cycle is N, i = 0, 1, 2,..., N-2, N-1. A precoding matrix having a period N × M based on the equation (228) is expressed by the following equation.

At this time, i = 0, 1, 2,..., N-2, N-1, k = 0, 1,.
Then, precoding matrices F [0] to F [N × M-1] are generated (the precoding matrices F [0] to F [N × M-1] have a period N × M. You can use them in any order.) For example, when symbol number N × M × i, precoding is performed using F [0], and when symbol number N × M × i + 1 is performed, precoding is performed using F [1],... When the symbol number is N × M × i + h, precoding is performed using F [h] (h = 0, 1, 2,..., N × M−2, N × M−1). (Here, as described in the previous embodiment, it is not always necessary to switch the precoding matrix regularly.)
When the precoding matrix is generated in this way, a switching method of a precoding matrix having a large period can be realized, and the position of the reception poor point can be easily changed, which improves the reception quality of data. May lead to In addition, although the precoding matrix of the cycle N × M is expressed by the equation (229), as described above, the precoding matrix of the cycle N × M may be expressed by the following equation.

At this time, i = 0, 1, 2,..., N-2, N-1, k = 0, 1,.

In Equations (229) and (230), when 0 radians ≦ δ <2π radians, a unitary matrix is obtained when δ = π radians, and a non-unitary matrix is obtained when δ ≠ π radians. In this method, one characteristic configuration is a non-unitary matrix of π / 2 radians ≦ | δ | <π radians (the condition of δ is the same as in other embodiments). Therefore, good data reception quality can be obtained. As another configuration, a unitary matrix may be used. As will be described in detail in the tenth embodiment and the sixteenth embodiment, if N is an odd number in the equations (229) and (230), good data reception is possible. The possibility of obtaining quality increases.

(Embodiment 9)
In the present embodiment, a method for regularly switching a precoding matrix using a unitary matrix will be described.

  As described in the eighth embodiment, in the method of switching the precoding matrix regularly in the cycle N, the precoding matrix prepared for the cycle N with reference to the equations (82) to (85) is represented by the following equation: It expresses.

At this time, i = 0, 1, 2,..., N-2, N-1. (It is assumed that α> 0.) In this embodiment, since a unitary matrix is handled, the precoding matrix of Equation (231) can be expressed by the following equation.

At this time, i = 0, 1, 2,..., N-2, N-1. (It is assumed that α> 0.) At this time, from the condition 5 of (Equation 106) and the condition 6 of (Equation 107) in the third embodiment, the following conditions give good data reception quality. It is important to get.

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

In the description of the sixth embodiment, the distance between the reception poor points has been described. However, in order to increase the distance between the reception bad points, it is important that the period N is an odd number of 3 or more. This point will be described below.

As described in the sixth embodiment, <condition 19> or <condition 20> is given in order to arrange the reception poor points so as to have a uniform distribution with respect to the phase on the complex plane.

That is, <Condition 19> means that the phase difference is 2π / N radians. <Condition 20> means that the phase difference is −2π / N radians.


Then, when θ ₁₁ (0) −θ ₂₁ (0) = 0 radians and α <1, on the complex plane of the reception poor point of s1 and the bad reception point of s2 when the period N = 3 FIG. 43 (a) shows the arrangement in FIG. 43, and FIG. 43 (b) shows the arrangement on the complex plane of the bad reception point of s1 and the bad reception point of s2 when the period N = 4. Further, when θ ₁₁ (0) −θ ₂₁ (0) = 0 radians and α> 1, when the period N = 3, the reception poor point of s1 and the reception bad point of s2 are on the complex plane. FIG. 44A shows the arrangement in FIG. 44A, and FIG. 44B shows the arrangement on the complex plane of the reception poor point of s1 and the bad reception point of s2 when the period N = 4.

  At this time, when considering the phase formed by the line segment formed by the poor reception point and the origin and the half line of Real ≧ 0 in the Real axis (see FIG. 43A), α> 1, In any case of α <1, when N = 4, there is always a case where the above-described phase at the reception poor point regarding s1 and the above-described phase at the reception poor point regarding s2 have the same value. (Refer to 4301 and 4302 in FIG. 43 and 4401 and 4402 in FIG. 44) At this time, the distance between the reception inferior points becomes small in the complex plane. On the other hand, when N = 3, the above-described phase at the reception poor point for s1 and the above-mentioned phase at the reception poor point for s2 do not occur in the same value.

  From the above, when the period N is an even number, considering that the above-described phase at the reception poor point for s1 and the above-mentioned phase at the reception poor point for s2 always have the same value, the period N is an odd number. When compared to when the period N is an even number, there is a high possibility that the distance between the reception inferior points will increase in the complex plane. However, when the period N is a small value, for example, N ≦ 16 or less, the minimum distance of reception poor points in the complex plane can be secured to some extent because the number of reception bad points is small. Therefore, when N ≦ 16, there is a possibility that the reception quality of data can be ensured even if the number is even.

  Therefore, in the method of regularly switching the precoding matrix based on the equation (232), if the period N is an odd number, there is a high possibility that the data reception quality can be improved. Note that the precoding matrices F [0] to F [N-1] are generated based on the equation (232) (the precoding matrices F [0] to F [N-1] have a period N). Can be used in any order.) Then, for example, precoding is performed using F [0] when the symbol number is Ni, precoding is performed using F [1] when the symbol number is Ni + 1, and F is performed when the symbol number is N × i + h. Precoding is performed using [h] (h = 0, 1, 2,..., N−2, N−1). (Here, as described in the previous embodiment, it is not always necessary to switch the precoding matrix regularly.) When both of the modulation schemes of s1 and s2 are 16QAM, α is

Then, there is a possibility that an effect that the minimum distance between 16 × 16 = 256 signal points on the IQ plane can be increased in a specific LOS environment may be obtained.
In the present embodiment, the configuration method of N different precoding matrices for the precoding hopping method of time period N has been described. At this time, F [0], F [1], F [2],..., F [N-2], F [N-1] are prepared as N different precoding matrices. However, since the present embodiment is described by taking the case of the single carrier transmission method as an example, F [0], F [1], F [2],... In the time axis (or frequency axis) direction. , F [N-2], F [N-1] are described in this order. However, the present invention is not limited to this, and N different precoding matrices F [0] generated in the present embodiment are not necessarily limited to this. , F [1], F [2],..., F [N-2], F [N-1] can be applied to a multicarrier transmission scheme such as an OFDM transmission scheme. As for the application method in this case, as in the first embodiment, the precoding weight can be changed by arranging symbols on the frequency axis and the frequency-time axis. Although described as a precoding hopping method of time period N, the same effect can be obtained even when N different precoding matrices are used at random, that is, a regular period is not necessarily used. It is not necessary to use N different precoding matrices to have.

  In addition, in the precoding matrix switching method of period H (where H is a period N in which the precoding matrix is regularly switched) is a larger natural number, N different precoding matrices in the present embodiment are included. If it is, the possibility of giving good reception quality increases. At this time, <condition # 17> and <condition # 18> can be replaced with the following conditions. (Consider the period as N.)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)
(Embodiment 10)
In the present embodiment, an example different from that in Embodiment 9 will be described regarding a method for regularly switching a precoding matrix using a unitary matrix.

  In the method of switching the precoding matrix regularly with the period 2N, the precoding matrix prepared for the period 2N is expressed by the following equation.

Let α> 0 and assume a fixed value (regardless of i).

Let α> 0 and assume a fixed value (regardless of i). (It is assumed that α in equation (234) and α in equation (235) have the same value.)
At this time, from the condition 5 in (Equation 106) and the condition 6 in (Equation 107) of the third embodiment, the following condition is obtained for the equation (234) in order to obtain good data reception quality: It becomes important.

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

Consider adding the following conditions.

  Next, as described in the sixth embodiment, <condition # 24> or <condition # 25> is set in order to arrange the reception inferior points so as to have a uniform distribution with respect to the phase on the complex plane. give.

In other words, <Condition 24> means that the phase difference is 2π / N radians. <Condition 25> means that the phase difference is −2π / N radians.

And when θ ₁₁ (0) −θ ₂₁ (0) = 0 radians and α> 1, when N = 4, the reception poor point of s1 and the reception bad point of s2 on the complex plane The arrangement is shown in FIGS. 45 (a) and 45 (b). As can be seen from FIGS. 45 (a) and 45 (b), in the complex plane, the minimum distance of the reception poor point of s1 is kept large, and similarly, the minimum distance of the reception bad point of s2 can be kept large. Yes. The same state is obtained when α <1. Considering the same as in the ninth embodiment, when N is an odd number, it is more likely that the distance between the reception inferior points is larger in the complex plane than when N is an even number. However, when N is a small value, for example, N ≦ 16 or less, the minimum distance of reception poor points in the complex plane can be ensured to some extent because the number of reception bad points is small. Therefore, when N ≦ 16, there is a possibility that the reception quality of data can be ensured even if the number is even.

  Therefore, in the method of regularly switching the precoding matrix based on the equations (234) and (235), if N is an odd number, there is a high possibility that the data reception quality can be improved. Note that the precoding matrices F [0] to F [2N-1] are generated based on the equations (234) and (235) (the precoding matrices F [0] to F [2N-1]). Can be used in any order for period 2N.) For example, precoding is performed using F [0] when the symbol number is 2Ni, precoding is performed using F [1] when the symbol number is 2Ni + 1, and F is performed when the symbol number is 2N × i + h. Precoding is performed using [h] (h = 0, 1, 2,..., 2N−2, 2N−1). (Here, as described in the previous embodiment, it is not always necessary to switch the precoding matrix regularly.) Also, when both the modulation schemes of s1 and s2 are 16QAM, α is expressed by the equation (233). ), It may be possible to obtain an effect that the minimum distance between 16 × 16 = 256 signal points in the IQ plane can be increased in a specific LOS environment.

  Further, the following conditions are considered as conditions different from <Condition # 23>.

(X is N, N + 1, N + 2, ..., 2N-2,2N-1 and y is N, N + 1, N + 2, ..., 2N-2,2N-1 And x ≠ y.)

(X is N, N + 1, N + 2, ..., 2N-2,2N-1 and y is N, N + 1, N + 2, ..., 2N-2,2N-1 And x ≠ y.)
At this time, by satisfying <Condition # 21>, <Condition # 22>, <Condition # 26>, and <Condition # 27>, the distance between the poor reception points of s1 in the complex plane is increased, and between s2 Since the distance between the poor reception points can be increased, good data reception quality can be obtained.

  In the present embodiment, the configuration method of 2N different precoding matrices for the precoding hopping method of time period 2N has been described. At this time, F [0], F [1], F [2], ..., F [2N-2], F [2N-1] are prepared as 2N different precoding matrices. However, since the present embodiment is described by taking the case of the single carrier transmission method as an example, F [0], F [1], F [2],... In the time axis (or frequency axis) direction. , F [2N-2], and F [2N-1] are described in this order, but the present invention is not necessarily limited to this, and 2N different precoding matrices F [0] generated in the present embodiment , F [1], F [2],..., F [2N-2], F [2N-1] can be applied to a multicarrier transmission scheme such as an OFDM transmission scheme. As for the application method in this case, as in the first embodiment, the precoding weight can be changed by arranging symbols on the frequency axis and the frequency-time axis. Although described as a precoding hopping method with a time period of 2N, the same effect can be obtained even when 2N different precoding matrices are used at random, that is, a regular period is not necessarily used. It is not necessary to use 2N different precoding matrices to have.

In addition, in the precoding matrix switching method of period H (where H is the above-described regular switching precoding matrix period 2N is a larger natural number), 2N different precoding matrices in this embodiment are included. If it is, the possibility of giving good reception quality increases.
(Embodiment 11)
In the present embodiment, a method for regularly switching a precoding matrix using a non-unitary matrix will be described.

  In the method of switching the precoding matrix regularly with the period 2N, the precoding matrix prepared for the period 2N is expressed by the following equation.

Let α> 0 and assume a fixed value (regardless of i). Further, δ ≠ π radians.

Let α> 0 and assume a fixed value (regardless of i). (It is assumed that α in equation (236) and α in equation (237) have the same value.)
At this time, from the condition 5 in (Equation 106) and the condition 6 in (Equation 107) of the third embodiment, the following condition is obtained for the equation (236) in order to obtain good data reception quality: It becomes important.

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

Consider adding the following conditions.

Note that a precoding matrix of the following equation may be given instead of equation (237).

Let α> 0 and assume a fixed value (regardless of i). (It is assumed that α in equation (236) and α in equation (238) have the same value.)
As an example, as described in the sixth embodiment, <condition # 31> or <condition # 32> is set in order to arrange the reception poor points so as to have a uniform distribution with respect to the phase on the complex plane. give.

That is, <Condition 31> means that the phase difference is 2π / N radians. <Condition 32> means that the phase difference is −2π / N radians.
Then, when θ ₁₁ (0) −θ ₂₁ (0) = 0 radians, α> 1, and δ = (3π) / 4 radians, the reception poor point of s1 and s2 when N = 4 46 (a) and 46 (b) show the arrangement of reception poor points on the complex plane. In this way, the period for switching the pull coding matrix can be increased, and the minimum distance of the reception poor point of s1 can be kept large in the complex plane. Since the minimum distance between points can be kept large, good reception quality can be obtained. Here, the case where α> 1, δ = (3π) / 4 radians, and N = 4 has been described as an example. However, the present invention is not limited to this, and π / 2 radians ≦ | δ | <π radians and α If> 0 and α ≠ 1, the same effect can be obtained.

  The following conditions are considered as conditions different from <Condition # 30>.

(X is N, N + 1, N + 2, ..., 2N-2,2N-1 and y is N, N + 1, N + 2, ..., 2N-2,2N-1 And x ≠ y.)

(X is N, N + 1, N + 2, ..., 2N-2,2N-1 and y is N, N + 1, N + 2, ..., 2N-2,2N-1 And x ≠ y.)
At this time, by satisfying <Condition # 28>, <Condition # 29>, <Condition # 33>, and <Condition # 34>, the distance between the reception inferior points of s1 in the complex plane is increased, and between s2 Since the distance between the poor reception points can be increased, good data reception quality can be obtained.

  In the present embodiment, the configuration method of 2N different precoding matrices for the precoding hopping method of time period 2N has been described. At this time, F [0], F [1], F [2], ..., F [2N-2], F [2N-1] are prepared as 2N different precoding matrices. However, since the present embodiment is described by taking the case of the single carrier transmission method as an example, F [0], F [1], F [2],... In the time axis (or frequency axis) direction. , F [2N-2], and F [2N-1] are described in this order, but the present invention is not necessarily limited to this, and 2N different precoding matrices F [0] generated in the present embodiment , F [1], F [2],..., F [2N-2], F [2N-1] can be applied to a multicarrier transmission scheme such as an OFDM transmission scheme. As for the application method in this case, as in the first embodiment, the precoding weight can be changed by arranging symbols on the frequency axis and the frequency-time axis. Although described as a precoding hopping method with a time period of 2N, the same effect can be obtained even when 2N different precoding matrices are used at random, that is, a regular period is not necessarily used. It is not necessary to use 2N different precoding matrices to have.

In addition, in the precoding matrix switching method of period H (where H is the above-described regular switching precoding matrix period 2N is a larger natural number), 2N different precoding matrices in this embodiment are included. If it is, the possibility of giving good reception quality increases.
(Embodiment 12)
In the present embodiment, a method for regularly switching a precoding matrix using a non-unitary matrix will be described.
In the method of switching the precoding matrix regularly with period N, the precoding matrix prepared for period N is expressed by the following equation.

Let α> 0 and assume a fixed value (regardless of i). Further, δ ≠ π radians (a fixed value regardless of i), i = 0, 1, 2,..., N−2, N−1.
At this time, from the condition 5 in (Equation 106) and the condition 6 in (Equation 107) of the third embodiment, the following condition is obtained for the equation (239) in order to obtain good data reception quality: It becomes important.

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)
As an example, as described in the sixth embodiment, <condition # 37> or <condition # 38> is set in order to arrange the reception poor points so as to have a uniform distribution with respect to the phase on the complex plane. give.

That is, <Condition 37> means that the phase difference is 2π / N radians. <Condition 38> means that the phase difference is −2π / N radians.
At this time, if π / 2 radians ≦ | δ | <π radians, α> 0, and α ≠ 1, the distance between the reception inferior points of s1 in the complex plane is increased, and reception of s2 is performed. Since the distance between the bad points can be increased, good data reception quality can be obtained. <Condition # 37> and <Condition # 38> are not necessarily required conditions.

  In the present embodiment, the configuration method of N different precoding matrices for the precoding hopping method of time period N has been described. At this time, F [0], F [1], F [2],..., F [N-2], F [N-1] are prepared as N different precoding matrices. However, since the present embodiment is described by taking the case of the single carrier transmission method as an example, F [0], F [1], F [2],... In the time axis (or frequency axis) direction. , F [N-2], F [N-1] are described in this order, but the present invention is not necessarily limited to this, and 2N different precoding matrices F [0] generated in the present embodiment , F [1], F [2],..., F [N-2], F [N-1] can be applied to a multicarrier transmission scheme such as an OFDM transmission scheme. As for the application method in this case, as in the first embodiment, the precoding weight can be changed by arranging symbols on the frequency axis and the frequency-time axis. Although described as a precoding hopping method of time period N, the same effect can be obtained even when N different precoding matrices are used at random, that is, a regular period is not necessarily used. It is not necessary to use N different precoding matrices to have.

  In addition, in the precoding matrix switching method of period H (where H is a period N in which the precoding matrix is regularly switched) is a larger natural number, N different precoding matrices in the present embodiment are included. If it is, the possibility of giving good reception quality increases. At this time, <condition # 35> <condition # 36> can be replaced with the following conditions. (Consider the period as N.)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(Embodiment 13)
In the present embodiment, another example of the eighth embodiment will be described.

  In the method of switching the precoding matrix regularly with the period 2N, the precoding matrix prepared for the period 2N is expressed by the following equation.

Let α> 0 and assume a fixed value (regardless of i). Further, δ ≠ π radians.

Let α> 0 and assume a fixed value (regardless of i). (It is assumed that α in equation (240) and α in equation (241) have the same value.)
A precoding matrix having a period of 2 × N × M based on the equations (240) and (241) is expressed by the following equation.

At this time, k = 0, 1,..., M-2, M-1.

At this time, k = 0, 1,..., M-2, M-1. Further, Xk = Yk may be satisfied, or Xk ≠ Yk may be satisfied.
Then, a precoding matrix of F [0] to F [2 × N × M-1] is generated (the precoding matrix of F [0] to F [2 × N × M-1] is Cycle 2 x N x M may be used in any order.) For example, when symbol number 2 × N × M × i, precoding is performed using F [0], and when symbol number 2 × N × M × i + 1 is performed, precoding is performed using F [1]. ..., precoding is performed using F [h] when symbol number 2 × N × M × i + h (h = 0, 1, 2,..., 2 × N × M-2, 2 × N × M-1) (Here, as described in the previous embodiment, it is not always necessary to switch the precoding matrix regularly.)
When the precoding matrix is generated in this way, a switching method of a precoding matrix having a large period can be realized, and the position of the reception poor point can be easily changed, which improves the reception quality of data. May lead to

Note that the formula (242) of the precoding matrix having a period of 2 × N × M may be changed to the following formula.

At this time, k = 0, 1,..., M-2, M-1.
Also, the equation (243) of the precoding matrix having a period of 2 × N × M may be any one of the equations (245) to (247).

At this time, k = 0, 1,..., M-2, M-1.

At this time, k = 0, 1,..., M-2, M-1.

At this time, k = 0, 1,..., M-2, M-1.
When attention is paid to the reception inferior point, in the equations (242) to (247),

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

If all of the above are satisfied, good data reception quality can be obtained. In the eighth embodiment, <condition # 39> and <condition # 40> may be satisfied.
When attention is paid to Xk and Yk in the equations (242) to (247),

(A is 0,1,2, ..., M-2, M-1 and b is 0,1,2, ..., M-2, M-1 and a ≠ b .)
However, s is an integer.

(A is 0,1,2, ..., M-2, M-1 and b is 0,1,2, ..., M-2, M-1 and a ≠ b .)
However, u is an integer.
If these two conditions are satisfied, good data reception quality can be obtained. In the eighth embodiment, it is preferable to satisfy <condition 42>.

  In Expressions (242) and (247), when 0 radians ≦ δ <2π radians, a unitary matrix is obtained when δ = π radians, and a non-unitary matrix is obtained when δ ≠ π radians. In this method, a non-unitary matrix with π / 2 radians ≦ | δ | <π radians is one characteristic configuration, and good data reception quality can be obtained. As another configuration, a unitary matrix may be used. However, as will be described in detail in the tenth and sixteenth embodiments, good data reception is possible when N is an odd number in the equations (242) to (247). The possibility of obtaining quality increases.

(Embodiment 14)
In the present embodiment, an example of proper use when a unitary matrix is used as a precoding matrix and a non-unitary matrix is used as a precoding matrix in a method of switching a precoding matrix regularly.

For example, when a 2 × 2 precoding matrix (each element is composed of complex numbers) is used, that is, two modulation signals (s1 (t) and s2 (t )), A case where precoding is performed and two signals after precoding are transmitted from two antennas will be described.
When data is transmitted using a method of regularly switching the precoding matrix, the mapping unit 306A, 306B switches the modulation scheme in the transmission apparatus of FIG. At this time, the relationship between the modulation multi-level number of the modulation scheme (number of modulation multi-level: the number of signal points of the modulation scheme on the IQ plane) and the precoding matrix will be described.

The advantage of the method of switching the precoding matrix regularly is that a good data reception quality can be obtained in the LOS environment as described in the sixth embodiment. When APP based on computation (or Max-log APP) is applied, the effect is great. By the way, the ML operation greatly affects the circuit scale (computation scale) with the modulation multi-level number of the modulation method. For example, when two signals after precoding are transmitted from two antennas and two modulation signals (signals based on the modulation method before precoding) both use the same modulation method, the modulation method Is QPSK, the number of candidate signal points (received signal points 1101 in FIG. 11) on the IQ plane is 4 × 4 = 16, 16 × 16 = 256 for 16QAM, 64 × 64 = 4096 for 64QAM, 256 × 256 = 65536 for 256QAM and 1024 × 1024 = 1048576 for 1024QAM. To reduce the computation scale of the receiver to a certain circuit scale, receive when the modulation method is QPSK, 16QAM, 64QAM. In the apparatus, ML calculation ((Max-log) APP based on ML calculation) is used, and in the case of 256QAM and 1024QAM, detection using linear calculation such as MMSE and ZF is used. (In some cases, ML operation may be used for 256QAM.)
Assuming such a receiving device, considering the signal-to-noise power ratio (SNR) after demultiplexing multiple signals, if the receiving device uses linear operations such as MMSE and ZF, precoding When a unitary matrix is suitable as a matrix and ML operation is used, either a unitary matrix or a non-unitary matrix may be used as a precoding matrix. Considering the description of any of the above-described embodiments, two signals after precoding are transmitted from two antennas, and the two modulation signals (signals based on the modulation scheme before precoding) are both the same modulation. Assuming that the method is used, when the modulation multi-level number of the modulation method is 64 values or less (or 256 values or less), the precoding matrix when the method of switching the precoding matrix regularly is used. If a unitary matrix is used and the unitary matrix is larger than 64 values (or larger than 256 values), if the unitary matrix is used, the circuit scale of the receiving apparatus in any modulation system supported by the communication system. There is a high possibility that it is possible to obtain the effect of obtaining good data reception quality while reducing the signal size.

  In addition, there is a possibility that it is better to use the unitary matrix even when the modulation multi-level number of the modulation scheme is 64 or less (or 256 or less). In consideration of this, if multiple modulation schemes with a modulation multi-level number of 64 or less (or 256 or less) are supported, multiple supported modulation schemes of 64 or less are supported. It is important that there is a case where a non-unitary matrix is used as a precoding matrix when a method of regularly switching the precoding matrix is used in any of the modulation methods.

In the above description, as an example, the case where two signals after precoding are transmitted from two antennas has been described. However, the present invention is not limited to this, and N signals after precoding are transmitted from N antennas. , N modulation signals (signals based on the modulation method before precoding) all use the same modulation method, a threshold value β _N is provided for the modulation multi-level number of the modulation method, and the modulation method If multiple modulation schemes with a modulation multi-level number of β _N or less are supported, a method of switching the precoding matrix regularly with any one of the supported modulation schemes of β _N or less is supported. In some cases, a non-unitary matrix is used as a precoding matrix, and in the case of a modulation scheme in which the modulation multi-level number of the modulation scheme is larger than β _N , if a unitary matrix is used, a communication system is used. In any modulation system supported by the system, there is a possibility that the effect of obtaining good data reception quality while reducing the circuit scale of the receiving apparatus can be obtained in any modulation system. Get higher. (When the modulation multi-level number of the modulation scheme is equal to or less than β _N , a non-unitary matrix may always be used as a precoding matrix when a scheme that regularly switches the precoding matrix is used.)
In the above description, the modulation scheme of N modulation signals transmitted simultaneously is described as being the same modulation scheme. However, in the following, two or more modulation schemes are used in N modulation signals transmitted simultaneously. The case where it exists is demonstrated.

As an example, a case where two signals after precoding are transmitted from two antennas will be described. When two modulation signals (signals based on a modulation scheme before precoding) are both the same modulation scheme or different modulation schemes, a modulation scheme having a modulation multi-level number of 2 ^a1 and a modulation multi-level It is assumed that the number ^2a2 modulation method is used. At this time, when ML operation ((Max-log) APP based on ML operation) is used in the receiving apparatus, the number of candidate signal points (reception signal points 1101 in FIG. 11) on the IQ plane is 2 ^a1 × 2 ^a2 = 2 There is a candidate signal point of ^{a1 + a2} . At this time, as described above, in order to be able to obtain reception quality of good data while reducing the circuit scale of the receiving apparatus, the threshold of relative ^{2 a1 + a2} 2 β provided, ^{2 a1 + a2} ≦ 2 β In this case, it is preferable to use a non-unitary matrix as a precoding matrix when a method of regularly switching the precoding matrix is used, and use a unitary matrix when 2 ^{a1 + a2} > 2 ^β .

Even in the case of 2 ^{a1 + a2} ≦ ^2β , it may be better to use the unitary matrix. In view of such fact, 2 ^{a1 + a2} ≦ 2 if it supports a combination of a plurality of modulation schemes ^beta, a combination of a plurality of modulation schemes 2 ^{a1 + a2} ≦ 2 ^β supporting either modulation method It is important that there is a case where a non-unitary matrix is used as a precoding matrix when a method of switching precoding matrices regularly in combination is used.

In the above description, the case where two signals after precoding are transmitted from two antennas has been described as an example. However, the present invention is not limited to this. For example, when all of N modulation signals (signals based on a modulation scheme before precoding) have the same modulation scheme or different modulation schemes, the modulation multilevel of the modulation scheme of the i-th modulation signal Let the number be 2 ^ai (i = 1, 2,..., N−1, N).

At this time, when ML operation ((Max-log) APP based on ML operation) is used in the receiving apparatus, the number of candidate signal points (reception signal points 1101 in FIG. 11) on the IQ plane is 2 ^a1 × 2 ^a2 X ... x2 ^ai x ... x2 ^aN = 2 ^{a1 + a2 + ... + ai + ... + aN} candidate signal points exist. At this time, as described above, in order to be able to obtain reception quality of good data while reducing the circuit scale of the receiving ^{apparatus, 2 a1 + a2 + ··· +} ai + ··· + threshold of 2 ^beta to ^aN Provided,

When a combination of a plurality of modulation schemes satisfying <Condition # 44> is supported, a combination of modulation schemes of a plurality of modulation schemes satisfying <Condition # 44> is pre-ordered regularly. There are cases where a non-unitary matrix is used as a precoding matrix when a coding matrix switching method is used,

In the case of a combination of all modulation schemes that satisfy <Condition # 45>, if a unitary matrix is used, the circuit scale of the receiving apparatus is the same for all modulation schemes supported by the communication system in any modulation scheme combination. There is a high possibility that it is possible to obtain the effect of obtaining good data reception quality while reducing the signal size. (A non-unitary matrix may be used as a precoding matrix when a method of regularly switching precoding matrices is used in all combinations of a plurality of modulation schemes that satisfy <condition # 44> that are supported.)
(Embodiment 15)
In this embodiment, a system example of a system that regularly switches a precoding matrix using a multicarrier transmission system such as OFDM will be described.

  FIG. 47 shows a time-frequency of a transmission signal transmitted by a broadcast station (base station) in a system that regularly switches a precoding matrix using a multicarrier transmission method such as OFDM in this embodiment. An example of a frame configuration on the shaft is shown. (The frame configuration from time $ 1 to time $ T is assumed.) FIG. 47A shows the frame configuration on the time-frequency axis of the stream s1 described in Embodiment 1 and the like, and FIG. The frame configuration on the time-frequency axis of the stream s2 described in the first embodiment is shown. The symbols of the same (sub) carrier in the stream s1 and the stream s2 are transmitted at the same time and the same frequency using a plurality of antennas.

47 (A) and 47 (B), (sub) carriers used when OFDM is used are carrier group #A composed of (sub) carrier a to (sub) carrier a + Na, and (sub) carrier b. Carrier group #B composed of (sub) carrier b + Nb, carrier group #C composed of (sub) carrier c to (sub) carrier c + Nc, (sub) carrier d to (sub) carrier d + Nd It shall be divided by carrier group #D,. Each subcarrier group supports a plurality of transmission methods. Here, by supporting a plurality of transmission methods, it is possible to effectively utilize the advantages of each transmission method. For example, in FIGS. 47A and 47B, the carrier group #A uses a spatial multiplexing MIMO transmission scheme or a MIMO transmission scheme with a fixed precoding matrix, and the carrier group #B is regularly precoded. It is assumed that a MIMO transmission method for switching a matrix is used, carrier group #C transmits only stream s1, and carrier group #D transmits using a space-time block code.
FIG. 48 shows a time-frequency of a transmission signal transmitted by a broadcast station (base station) in a system that regularly switches a precoding matrix using a multi-carrier transmission method such as OFDM in the present embodiment. 48 shows an example of a frame configuration on the axis, and shows a frame configuration from time $ X to time $ X + T ′, which is different from FIG. In FIG. 48, as in FIG. 47, (sub) carriers used when OFDM is used are a carrier group #A composed of (sub) carrier a to (sub) carrier a + Na, and (sub) carrier b. Carrier group #B composed of (sub) carrier b + Nb, carrier group #C composed of (sub) carrier c to (sub) carrier c + Nc, (sub) carrier d to (sub) carrier d + Nd It shall be divided by carrier group #D,. 48 differs from FIG. 47 in that there are carrier groups in which the communication method used in FIG. 47 differs from the communication method used in FIG. In FIG. 48, in (A) and (B), carrier group #A is transmitted using a space-time block code, and carrier group #B is assumed to use a MIMO transmission scheme that regularly switches a precoding matrix. Suppose that the carrier group #C uses a MIMO transmission system that regularly switches the precoding matrix, and the carrier group #D transmits only the stream s1.

Next, a supported transmission method will be described.
FIG. 49 shows a signal processing method when a spatial multiplexing MIMO transmission scheme or a MIMO transmission scheme with a fixed precoding matrix is used, and the same numbers as those in FIG. 6 are given.
The weighting synthesizer 600, which is a baseband signal according to a certain modulation method, receives the stream s1 (t) (307A), the stream s2 (t) (307B), and information 315 on the weighting method as input, Modulation signal z1 (t) (309A) and weighted modulation signal z2 (t) (309B) are output. Here, when the information 315 regarding the weighting method indicates the spatial multiplexing MIMO transmission method, the signal processing of the method # 1 in FIG. 49 is performed. That is, the following processing is performed.

  However, when a method for transmitting one modulated signal is supported, Expression (250) may be expressed as Expression (251) from the viewpoint of transmission power.

  Then, when the information 315 on the weighting method indicates a MIMO transmission method in which the precoding matrix is fixed, for example, signal processing of method # 2 in FIG. 49 is performed. That is, the following processing is performed.

Here, θ11, θ12, λ, and δ are fixed values.
FIG. 50 shows the structure of a modulation signal when a space-time block code is used. The space-time block encoding unit (5002) in FIG. 50 receives a baseband signal based on a certain modulation signal. For example, the space-time block encoding unit (5002) receives the symbols s1, s2,. Then, as shown in FIG. 50, space-time block coding is performed, and z1 (5003A) is “s1 as symbol # 0”, “−s2 ^* as symbol # 1,” “s3 as symbol # 2,” “symbol # 3”. -S4 ^* "..., And z2 (5003B) is" s2 as symbol # 0 "," s1 ^* as symbol # 1, "" s4 as symbol # 2, "" s3 ^* as symbol # 3, "and so on. Become. At this time, symbol #X in z1 and symbol #X in z2 are transmitted from the antenna at the same time and at the same frequency.

47 and 48, only symbols for transmitting data are described, but actually, it is necessary to transmit information such as a transmission method, a modulation method, and an error correction method. For example, as shown in FIG. 51, if these pieces of information are periodically transmitted using only one modulation signal z1, these pieces of information can be transmitted to the communication partner. In addition, it is necessary to transmit transmission path fluctuations, that is, symbols for the receiver to estimate channel fluctuations (for example, pilot symbols, reference symbols, preambles, and known (PSK: Phase Shift Keying) symbols). . In FIG. 47 and FIG. 48, these symbols are omitted and described, but actually, symbols for estimating channel fluctuation are included in the frame configuration of the time-frequency axis. Therefore, each carrier group is not composed only of symbols for transmitting data. (This also applies to the first embodiment.)
FIG. 52 illustrates an example of a configuration of a transmission device of a broadcast station (base station) in the present embodiment. The transmission method determination unit (5205) determines the number of carriers in each carrier group, the modulation method, the error correction method, the coding rate of the error correction code, the transmission method, and the like, and outputs it as a control signal (5205).
Modulation signal generation unit # 1 (5201_1) receives information (5200_1) and control signal (5205) as inputs, and modulates carrier group #A in FIGS. 47 and 48 based on the communication method information of control signal (5205). The signal z1 (5202_1) and the modulation signal z2 (5203_1) are output.

  Similarly, modulation signal generation section # 2 (5201_2) receives information (5200_2) and control signal (5205) as input, and based on the communication method information of control signal (5205), carrier group # in FIG. 47 and FIG. B modulation signal z1 (5202_2) and modulation signal z2 (5203_2) are output.

  Similarly, modulation signal generation section # 3 (5201_3) receives information (5200_3) and control signal (5205) as input, and based on information on the communication method of control signal (5205), carrier group # in FIG. 47 and FIG. The C modulation signal z1 (5202_3) and the modulation signal z2 (5203_3) are output.

  Similarly, modulation signal generation section # 4 (5201_4) receives information (5200_4) and control signal (5205) as input, and based on the communication method information of control signal (5205), carrier group # in FIG. 47 and FIG. D modulation signal z1 (5202_4) and modulation signal z2 (5203_4) are output.

・
・
・
Similarly, the modulation signal generation unit #M (5201_M) receives the information (5200_M) and the control signal (5205) as inputs, and based on the communication method information of the control signal (5205), the modulation signal z1 (5202_M) of a certain carrier group. ) And the modulated signal z2 (5203_M).

  The OFDM-related processing unit (5207_1) includes a modulation signal z1 (5202_1) of carrier group #A, a modulation signal z1 (5202_2) of carrier group #B, a modulation signal z1 (5202_3) of carrier group #C, and a carrier group #D. Modulation signal z1 (5202_4),..., And a modulation signal z1 (5202_M) of a certain carrier group and a control signal (5206) are input, and processing such as rearrangement, inverse Fourier transform, frequency conversion, and amplification is performed. The transmission signal (5208_1) is output, and the transmission signal (5208_1) is output as a radio wave from the antenna (5209_1).

  Similarly, the OFDM system-related processing unit (5207_2) includes a modulation signal z1 (5203_1) of carrier group #A, a modulation signal z2 (5203_2) of carrier group #B, a modulation signal z2 (5203_3) of carrier group #C, and a carrier. The modulation signal z2 (5203_4) of the group #D,..., The modulation signal z2 (5203_M) of a certain carrier group, and the control signal (5206) are input, and rearrangement, inverse Fourier transform, frequency conversion, amplification, etc. Processing is performed and a transmission signal (5208_2) is output, and the transmission signal (5208_2) is output as a radio wave from the antenna (5209_2).

FIG. 53 shows an example of the configuration of the modulation signal generators # 1 to #M in FIG. The error correction coding unit (5302) receives the information (5300) and the control signal (5301) as input, and sets the error correction coding method and the coding rate of the error correction coding according to the control signal (5301). Then, error correction encoding is performed, and data (5303) after error correction encoding is output. (By setting the error correction coding method and the error correction coding rate, for example, when using LDPC code, turbo code, convolutional code, etc., depending on the coding rate, puncturing is performed to realize the coding rate. May be.)
The interleaving unit (5304) receives the data (5303) after error correction coding and the control signal (5301) as input, and the data after error correction coding (5303) according to the information of the interleaving method included in the control signal (5301). ) And rearranged data (5305) is output.

  The mapping unit (5306_1) receives the interleaved data (5305) and the control signal (5301) as input, performs mapping processing according to the modulation scheme information included in the control signal (5301), and generates the baseband signal (5307_1). Output.

  Similarly, mapping section (5306_2) receives interleaved data (5305) and control signal (5301) as input, performs mapping processing according to the modulation scheme information included in control signal (5301), and performs baseband signal ( 5307_2) is output.

  The signal processing unit (5308) receives the baseband signal (5307_1), the baseband signal (5307_2), and the control signal (5301) as inputs, and a transmission method (here, for example, spatial multiplexing MIMO) included in the control signal (5301). Signal processing based on information of a transmission method, a MIMO method using a fixed precoding matrix, a MIMO method that regularly switches the precoding matrix, a space-time block coding, and a transmission method that transmits only the stream s1) The signal z1 (5309_1) after processing and z2 (5309_2) after signal processing are output. When the transmission method for transmitting only the stream s1 is selected, the signal processing unit (5308) may not output z2 (5309_2) after the signal processing. FIG. 53 shows the configuration in the case where there is one error correction encoding unit. However, the configuration is not limited to this. For example, as shown in FIG. 3, a plurality of encoders may be provided. .

  FIG. 54 shows an example of the configuration of the OFDM scheme-related processing units (5207_1 and 5207_2) in FIG. 52, and the same reference numerals are given to those that operate in the same way as in FIG. Rearranger (5402A) modulates modulated signal z1 (5400_1) of carrier group #A, modulated signal z1 (5400_2) of carrier group #B, modulated signal z1 (5400_3) of carrier group #C, and modulated of carrier group #D. The signal z1 (5400_4),..., The modulation signal z1 (5400_M) of a certain carrier group and the control signal (5403) are input, rearrangement is performed, and rearranged signals 1405A and 1405B are output. 47, FIG. 48, and FIG. 51, the carrier group allocation is described as an example of the configuration of aggregated subcarriers. However, the present invention is not limited to this. You may comprise a carrier group. 47, FIG. 48, and FIG. 51, the number of carriers in the carrier group is described as an example that does not change in time, but is not limited thereto. This point will be described later separately.

  FIG. 55 shows an example of the details of the frame configuration on the time-frequency axis in the method of setting the transmission method for each carrier group as shown in FIG. 47, FIG. 48, and FIG. In FIG. 55, control information symbols are indicated by 5500, individual control information symbols by 5501, data symbols by 5502, and pilot symbols by 5503. FIG. 55A shows the frame configuration of the stream s1 on the time-frequency axis, and FIG. 55B shows the frame configuration of the stream s2 on the time-frequency axis.

  The control information symbol is a symbol for transmitting control information common to the carrier group, and includes a symbol for the transceiver to perform frequency and time synchronization, information on (sub) carrier allocation, and the like. It is assumed that the control control symbol is transmitted only from stream s1 at time $ 1.

  The individual control information symbol is a symbol for transmitting control information for each subcarrier group, and the data symbol transmission scheme, modulation scheme, error correction coding scheme, error correction coding rate, and error correction code. Block size information, pilot symbol insertion method information, pilot symbol transmission power information, and the like. It is assumed that the individual control information symbol is transmitted only from the stream s1 at time $ 1.

  The data symbol is a symbol for transmitting data (information). As described with reference to FIGS. 47 to 50, for example, a spatial multiplexing MIMO transmission scheme, a MIMO scheme using a fixed precoding matrix, a rule This symbol is a symbol of any one of the MIMO scheme that switches the precoding matrix, the space-time block coding, and the transmission scheme that transmits only the stream s1. In addition, in the carrier group #A, the carrier group #B, the carrier group #C, and the carrier group #D, the data symbol is described so as to exist in the stream s2, but a transmission scheme that transmits only the stream s1 is used. In some cases, there may be no data symbol in the stream s2.

  The pilot symbol is a symbol used by the receiving apparatus for channel estimation, that is, estimation of fluctuations corresponding to h11 (t), h12 (t), h21 (t), and h22 (t) in Expression (36). (Here, since a multi-carrier transmission scheme such as the OFDM scheme is used, fluctuations corresponding to h11 (t), h12 (t), h21 (t), and h22 (t) are estimated for each subcarrier. Therefore, the pilot symbol uses, for example, the PSK transmission method and is configured to have a known pattern in the transceiver. In addition, the receiving apparatus may use the pilot symbols for frequency offset estimation, phase distortion estimation, and time synchronization.

FIG. 56 shows an example of the configuration of a receiving apparatus for receiving the modulated signal transmitted by the transmitting apparatus of FIG. 52, and the same reference numerals are given to those operating in the same manner as in FIG.
In FIG. 56, the OFDM scheme-related processing unit (5600_X) receives the received signal 702_X, performs predetermined processing, and outputs a signal 704_X after signal processing. Similarly, the OFDM scheme-related processing unit (5600_Y) receives the received signal 702_Y, performs predetermined processing, and outputs a signal 704_Y after signal processing.

  Control information decoding section 709 in FIG. 56 receives signal processed signal 704_X and signal processed signal 704_Y as input, extracts control information symbols and individual control information symbols in FIG. 55, and transmits the control information using these symbols. And a control signal 710 containing this information is output.

  The channel fluctuation estimation unit 705_1 of the modulated signal z1 receives the signal 704_X after signal processing and the control signal 710 as input, performs channel estimation in a carrier group (desired carrier group) required by this receiving apparatus, and performs channel estimation The signal 706_1 is output.

  Similarly, channel fluctuation estimation section 705_2 of modulated signal z2 receives signal processed signal 704_X and control signal 710 as input, and performs channel estimation in a carrier group (desired carrier group) required by this receiving apparatus. The channel estimation signal 706_2 is output.

  Similarly, channel fluctuation estimation section 705_1 of modulated signal z1 receives signal processed signal 704_Y and control signal 710 as input, and performs channel estimation in a carrier group (desired carrier group) required by this receiving apparatus. The channel estimation signal 708_1 is output.

  Similarly, channel fluctuation estimation section 705_2 of modulated signal z2 receives signal processed signal 704_Y and control signal 710 as input, and performs channel estimation in a carrier group (desired carrier group) required by this receiving apparatus. The channel estimation signal 708_2 is output.

  Then, the signal processing unit 711 receives the signals 706_1, 706_2, 708_1, 708_2, 704_X, 704_Y, and the control signal 710, and transmits the data symbols transmitted in a desired carrier group included in the control signal 710. Based on information such as the system, modulation system, error correction coding system, coding rate of error correction coding, block size of error correction code, etc., demodulation and decoding are performed, and received data 712 is output.

  FIG. 57 shows the configuration of the OFDM system-related processing units (5600_X, 5600_Y) in FIG. 56. The frequency conversion unit (5701) receives the received signal (5700) as an input, performs frequency conversion, and performs frequency conversion. The signal (5702) is output.

The Fourier transform unit (5703) receives the signal (5702) after frequency conversion, performs Fourier transform, and outputs a signal (5704) after Fourier transform.
As described above, when a multi-carrier transmission scheme such as the OFDM scheme is used, by dividing into a plurality of carrier groups and setting a transmission scheme for each carrier group, the reception quality and transmission for each carrier group are set. Since the speed can be set, an effect that a flexible system can be constructed can be obtained. At this time, by making it possible to select a method for switching the precoding matrix regularly as described in the other embodiments, it is possible to obtain high reception quality and a high transmission rate for the LOS environment. The advantage that can be obtained can be obtained. In this embodiment, as a transmission scheme in which a carrier group can be set, a “spatial multiplexing MIMO transmission scheme, a MIMO scheme using a fixed precoding matrix, a MIMO scheme that regularly switches a precoding matrix, a space-time block, Although the encoding method and the transmission method for transmitting only the stream s1 have been described, the present invention is not limited to this. At this time, the method of FIG. 50 has been described as the space-time code, but the method is not limited to this, and is not fixed. The MIMO scheme using a typical precoding matrix is not limited to scheme # 2 in FIG. 49, but may be composed of a fixed precoding matrix. Further, in the present embodiment, the case where the number of antennas of the transmission apparatus is two has been described. However, the present invention is not limited to this, and even when the number of antennas is larger than two, “spatial multiplexing MIMO transmission scheme, fixed If the transmission method can be selected from the MIMO method using a precoding matrix, the MIMO method that regularly switches the precoding matrix, the space-time block coding, and the transmission method that transmits only the stream s1, the same effect can be obtained. Can be obtained.

  FIG. 58 shows a carrier group allocation method different from that in FIGS. 47, 48, and 51. 47, FIG. 48, FIG. 51, and FIG. 55 illustrate an example in which the allocation of carrier groups is configured with aggregated subcarriers, but in FIG. 58, carriers in the carrier groups are discretely arranged. It is a feature. FIG. 58 shows an example of a frame configuration on the time-frequency axis that is different from FIGS. 47, 48, 51, and 55. In FIG. 58, carrier 1 to carrier H, time $ 1 to time $ K The same structure as that of FIG. 55 is denoted by the same reference numeral. In the data symbols of FIG. 58, a symbol described as “A” is a symbol of carrier group A, a symbol described as “B” is a symbol of carrier group B, and is described as “C”. Indicates that the symbol is a symbol of the carrier group C, and a symbol described as “D” is a symbol of the carrier group D. As described above, the carrier groups can be similarly implemented even if they are arranged discretely in the (sub) carrier direction, and it is not always necessary to use the same carrier in the time axis direction. By performing such an arrangement, it is possible to obtain an effect that time and frequency diversity gain can be obtained.

  47, 48, 51, and 58, the control information symbols and the unique control information symbols are arranged at the same time for each carrier group, but may be arranged at different times. Further, the number of (sub) carriers used by the carrier group may be changed with time.

(Embodiment 16)
In the present embodiment, as in Embodiment 10, a method for regularly switching a precoding matrix using a unitary matrix will be described in the case where N is an odd number.

  In the method of switching the precoding matrix regularly with the period 2N, the precoding matrix prepared for the period 2N is expressed by the following equation.

Let α> 0 and assume a fixed value (regardless of i).

Let α> 0 and assume a fixed value (regardless of i). (It is assumed that α in equation (253) and α in equation (254) have the same value.)
At this time, from the condition 5 in (Equation 106) and the condition 6 in (Equation 107) of the third embodiment, the following condition is obtained for the equation (253) in order to obtain good data reception quality: It becomes important.

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

(X is 0,1,2, ..., N-2, N-1, y is 0,1,2, ..., N-2, N-1 and x ≠ y .)

Consider adding the following conditions.

  Next, as described in the sixth embodiment, <condition # 49> or <condition # 50> is set in order to arrange the reception inferior points so as to have a uniform distribution with respect to the phase on the complex plane. give.

That is, <Condition 49> means that the phase difference is 2π / N radians. <Condition 50> means that the phase difference is −2π / N radians.
Then, when θ ₁₁ (0) −θ ₂₁ (0) = 0 radians and α> 1, when N = 3, the reception bad point of s1 and the bad reception point of s2 on the complex plane The arrangement is shown in FIGS. 60 (a) and 60 (b). As can be seen from FIGS. 60A and 60B, in the complex plane, the minimum distance of the reception poor point of s1 is kept large, and similarly, the minimum distance of the reception bad point of s2 can be kept large. Yes. The same state is obtained when α <1. Compared with FIG. 45 of the tenth embodiment, when considered in the same manner as in the ninth embodiment, when N is an odd number, the distance between reception poor points in the complex plane is larger than when N is an even number. Is likely to grow. However, when N is a small value, for example, N ≦ 16 or less, the minimum distance of reception poor points in the complex plane can be ensured to some extent because the number of reception bad points is small. Therefore, when N ≦ 16, there is a possibility that the reception quality of data can be ensured even if the number is even.

  Therefore, in the method of regularly switching the precoding matrix based on the equations (253) and (254), if N is an odd number, there is a high possibility that the data reception quality can be improved. Note that the precoding matrices F [0] to F [2N-1] are generated based on the expressions (253) and (254) (the precoding matrices F [0] to F [2N-1]) Can be used in any order for period 2N.) For example, precoding is performed using F [0] when the symbol number is 2Ni, precoding is performed using F [1] when the symbol number is 2Ni + 1, and F is performed when the symbol number is 2N × i + h. Precoding is performed using [h] (h = 0, 1, 2,..., 2N−2, 2N−1). (Here, as described in the previous embodiment, it is not always necessary to switch the precoding matrix regularly.) Also, when both the modulation schemes of s1 and s2 are 16QAM, α is expressed by the equation (233). ), It may be possible to obtain an effect that the minimum distance between 16 × 16 = 256 signal points in the IQ plane can be increased in a specific LOS environment.

  Further, the following conditions are considered as conditions different from <Condition # 48>.

(X is N, N + 1, N + 2, ..., 2N-2,2N-1 and y is N, N + 1, N + 2, ..., 2N-2,2N-1 And x ≠ y.)

(X is N, N + 1, N + 2, ..., 2N-2,2N-1 and y is N, N + 1, N + 2, ..., 2N-2,2N-1 And x ≠ y.)
At this time, by satisfying <Condition # 46>, <Condition # 47>, <Condition # 51>, and <Condition # 52>, the distance between the poor reception points of s1 in the complex plane is increased, and between s2 Since the distance between the poor reception points can be increased, good data reception quality can be obtained.

  In the present embodiment, the configuration method of 2N different precoding matrices for the precoding hopping method of time period 2N has been described. At this time, F [0], F [1], F [2], ..., F [2N-2], F [2N-1] are prepared as 2N different precoding matrices. However, since the present embodiment is described by taking the case of the single carrier transmission method as an example, F [0], F [1], F [2],... In the time axis (or frequency axis) direction. , F [2N-2], and F [2N-1] are described in this order, but the present invention is not necessarily limited to this, and 2N different precoding matrices F [0] generated in the present embodiment , F [1], F [2],..., F [2N-2], F [2N-1] can be applied to a multicarrier transmission scheme such as an OFDM transmission scheme. As for the application method in this case, as in the first embodiment, the precoding weight can be changed by arranging symbols on the frequency axis and the frequency-time axis. Although described as a precoding hopping method with a time period of 2N, the same effect can be obtained even when 2N different precoding matrices are used at random, that is, a regular period is not necessarily used. It is not necessary to use 2N different precoding matrices to have.

  In addition, in the precoding matrix switching method of period H (where H is the above-described regular switching precoding matrix period 2N is a larger natural number), 2N different precoding matrices in this embodiment are included. If it is, the possibility of giving good reception quality increases.

(Other supplements)
In this specification, it is conceivable that the transmission device is equipped with a communication / broadcasting device such as a broadcasting station, a base station, an access point, a terminal, a mobile phone, and the like. It is conceivable that the receiving device is equipped with a communication device such as a television, a radio, a terminal, a personal computer, a mobile phone, an access point, and a base station. In addition, the transmission device and the reception device in the present invention are devices having a communication function, and the devices provide some interface to a device for executing an application such as a television, a radio, a personal computer, or a mobile phone. It is also conceivable that the connection can be made via a network.

  In the present embodiment, symbols other than data symbols, for example, pilot symbols (preamble, unique word, postamble, reference symbol, etc.), control information symbols, etc., are arranged in any manner. Good. Here, the pilot symbol and the control information symbol are named, but any naming method may be used, and the function itself is important.

  The pilot symbol is, for example, a known symbol modulated by using PSK modulation in a transmitter / receiver (or the receiver may know the symbol transmitted by the transmitter by synchronizing the receiver). .), And the receiver uses this symbol to perform frequency synchronization, time synchronization, channel estimation (for each modulated signal) (estimation of CSI (Channel State Information)), signal detection, and the like. Become.

  In addition, the control information symbol is information (for example, a modulation method, an error correction coding method used for communication, a communication information symbol) that needs to be transmitted to a communication partner in order to realize communication other than data (such as an application). This is a symbol for transmitting an error correction coding method coding rate, setting information in an upper layer, and the like.

  The present invention is not limited to Embodiments 1 to 5 described above, and can be implemented with various modifications. For example, in the above embodiment, the case of performing as a communication device has been described. However, the present invention is not limited to this, and this communication method can also be performed as software.

  In the above description, the precoding switching method in the method of transmitting two modulated signals from two antennas has been described. However, the present invention is not limited to this, and precoding is performed on four mapped signals. In a method of generating one modulated signal and transmitting from four antennas, that is, a method of generating N modulated signals by performing precoding on N mapped signals and transmitting from N antennas Similarly, a precoding switching method for changing precoding weights (matrixes) can be similarly implemented.

  In the present specification, terms such as “precoding” and “precoding weight” are used, but any name may be used. In the present invention, the signal processing itself is important.

Different data may be transmitted by the streams s1 (t) and s2 (t), or the same data may be transmitted.
Both the transmitting antenna of the transmitting device and the receiving antenna of the receiving device may be configured by a plurality of antennas.

In this specification, “∀” represents a universal quantifier, and “∃” represents an existent quantifier.

Further, in this specification, the unit of phase, such as declination, in the complex plane is “radian”.
Using a complex plane, it can be displayed in polar form as a display of complex polar coordinates. When a complex number z = a + jb (a and b are both real numbers and j is an imaginary unit) and a point (a, b) on the complex plane is made to correspond, this point is expressed in polar coordinates [r, θ ]
a = r × cos θ,
b = r × sin θ

Where r is the absolute value of z (r = | z |) and θ is the argument. Z = a + jb is expressed as re ^jθ .
FIG. 59 shows an example of a broadcasting system using the method for switching the precoding matrix regularly described in this specification. In FIG. 59, a video encoding unit 5901 receives video as input, performs video encoding, and outputs video 5902 after video encoding. The speech encoding unit 5903 receives speech as input, performs speech encoding, and outputs speech encoded data 5904. The data encoding unit 5905 receives data, performs data encoding (for example, data compression), and outputs data 5906 after data encoding. These are collectively referred to as an information source encoding unit 5900.

  The transmission unit 5907 receives the data 5902 after video encoding, the data 5904 after audio encoding, and the data 5906 after data encoding as one of these data or all of these data as transmission data. Processing such as error correction coding, modulation, and precoding (for example, signal processing in the transmission apparatus in FIG. 3) is performed, and transmission signals 5908_1 to 5908_N are output. Transmission signals 5908_1 to 5908_N are transmitted as radio waves by antennas 5909_1 to 5909_N, respectively.

  The receiving unit 5912 receives the received signals 5911_1 to 5911_M received by the antennas 5910_1 to 5910_M, and performs processing such as frequency conversion, precoding decoding, log likelihood ratio calculation, error correction decoding, etc. (for example, in the receiving apparatus of FIG. 7). Process) and output received data 5913, 5915, 5917. The information source decoding unit 5919 receives the received data 5913, 5915, and 5917, and the video decoding unit 5914 receives the received data 5913, decodes the video, and outputs a video signal. Appears on the display. The voice decoding unit 5916 receives the received data 5915 as an input. Audio decoding is performed and an audio signal is output, and the audio flows from the speaker. The data decoding unit 5918 receives the received data 5917, performs data decoding, and outputs data information.

For example, a program for executing the communication method may be stored in a ROM (Read Only Memory) in advance, and the program may be operated by a CPU (Central Processor Unit).

  In addition, a program for executing the communication method is stored in a computer-readable storage medium, the program stored in the storage medium is recorded in a RAM (Random Access Memory) of the computer, and the computer is operated according to the program. You may do it.

  Each configuration such as the above-described embodiments may be typically realized as an LSI (Large Scale Integration) which is an integrated circuit. These may be individually made into one chip, or may be made into one chip so as to include all or part of the configurations of the respective embodiments. Although referred to here as LSI, depending on the degree of integration, it may also be called IC (Integrated Circuit), system LSI, super LSI, or ultra LSI. Further, the method of circuit integration is not limited to LSI's, and implementation using dedicated circuitry or general purpose processors is also possible. An FPGA (Field Programmable Gate Array) that can be programmed after manufacturing the LSI, or a reconfigurable processor that can reconfigure the connection and setting of circuit cells inside the LSI may be used.

  Further, if integrated circuit technology comes out to replace LSI's as a result of the advancement of semiconductor technology or a derivative other technology, it is naturally also possible to carry out function block integration using this technology. There is a possibility of adaptation of biotechnology.

  The present invention can be widely applied to wireless systems that transmit different modulated signals from a plurality of antennas, and is suitable for application to, for example, an OFDM-MIMO communication system. The present invention also applies to a case where MIMO transmission is performed in a wired communication system having a plurality of transmission points (for example, a PLC (Power Line Communication) system, an optical communication system, a DSL (Digital Subscriber Line) system). At this time, a plurality of modulated signals as described in the present invention are transmitted using a plurality of transmission locations. The modulated signal may be transmitted from a plurality of transmission locations.

302A, 302B Encoder 304A, 304B Interleaver 306A, 306B Mapping unit 314 Weighted synthesis information generation unit 308A, 308B Weighting synthesis unit 310A, 310B Radio unit 312A, 312B Antenna 402 Encoder 404 Distribution unit 504 # 1, 504 # 2 Transmitting antenna 505 # 1, 505 # 2 receiving antenna 600 weighting combining section 703_X radio section 701_X antenna 705_1 channel fluctuation estimating section 705_2 channel fluctuation estimating section 707_1 channel fluctuation estimating section 707_2 channel fluctuation estimating section 709 control information decoding section 711 signal processing section 803 INNER MIMO detection unit 805A, 805B log likelihood calculation unit 807A, 807B deinterleaver 809A, 809B log likelihood ratio calculation unit 811A, 811B Soft- n / soft-out decoders 813A and 813B Interleaver 815 Storage unit 819 Weighting coefficient generation unit 901 Soft-in / soft-out decoder 903 Distributor 1301A and 1301B OFDM system related processing units 1402A and 1402A Serial parallel conversion units 1404A and 1404B Rearrangement 1406A, 1406B Inverse fast Fourier transform units 1408A, 1408B Radio unit 2200 Precoding weight matrix generation unit 2300 Rearrangement unit 4002 Encoder group

Claims (4)

A transmission method,
N matrices F [i] (where i is an integer from 0 to N-1 and N is an integer of 3 or more) defining precoding processing to be applied to a plurality of modulated signals. Select one of the matrix by switching for each slot,
For the first modulated signal s1 generated from the first bit group and the second modulated signal s2 generated from the second bit group for each slot, the selected matrix F [i] By performing a corresponding precoding process, a first transmission signal z1 and a second transmission signal z2 are generated,
Transmitting the first transmission signal z1 and the second transmission signal z2 simultaneously from the first antenna and the second antenna at the same frequency, respectively;
The first transmission signal z1 and the second transmission signal z2 satisfy (z1, z2) ^T = F [i] (s1, s2) ^T ,
The N matrices F [i] are

Where λ is an arbitrary angle, α is a positive real number excluding 1,
θ ₁₁ (i) and θ ₂₁ (i) are

Or

The filling,
Each of the N matrices is selected at least once within a predetermined number of slots.
Transmission method. A transmitting device,
Among N matrices F [i] (where i is an integer from 0 to N-1 and N is an integer of 3 or more) that defines precoding processing to be applied to a plurality of modulated signals. A weighted synthesis information generation unit that selects and selects one matrix from slot to slot;
For the first modulated signal s1 generated from the first bit group and the second modulated signal s2 generated from the second bit group for each slot, the selected matrix F [i] A weighting and synthesizing unit that generates a first transmission signal z1 and a second transmission signal z2 by performing a corresponding precoding process;
A transmission unit for transmitting the first transmission signal z1 and the second transmission signal z2 simultaneously from the first antenna and the second antenna at the same frequency, respectively,
The first transmission signal z1 and the second transmission signal z2 satisfy (z1, z2) ^T = F [i] (s1, s2) ^T ,
The N matrices F [i] are

Where λ is an arbitrary angle, α is a positive real number excluding 1,
θ ₁₁ (i) and θ ₂₁ (i) are

Or

The filling,
Each of the N matrices is selected at least once within a predetermined number of slots.
Transmitter device. A receiving method,
Obtaining a reception signal that is a signal obtained by receiving the first transmission signal z1 and the second transmission signal z2 transmitted simultaneously at the same frequency from the first antenna and the second antenna, respectively;
The first transmission signal z1 and the second transmission signal z2 are the first modulation signal s1 generated from the first bit group and the second modulation signal generated from the second bit group for each slot. s2 is generated by performing precoding processing, and the precoding processing is performed by N matrices F [i] (where i is an integer of 0 to N−1 and N is an integer of 3 or more). Is a process corresponding to one matrix selected by switching from slot to slot,
The first transmission signal z1 and the second transmission signal z2 satisfy the relationship of (z1, z2) ^T = F [i] (s1, s2) ^T ,
The N matrices F [i] are

Where λ is an arbitrary angle, α is a positive real number excluding 1,
θ ₁₁ (i) and θ ₂₁ (i) are

Or

The filling,
Each of the N matrices is selected at least once within a predetermined number of slots;
Furthermore, the reception method includes a step of generating reception data by demodulating the reception signal in accordance with a matrix selected for each slot on the transmission side. A receiving device,
Receiving a received signal, which is a signal obtained by receiving the first transmission signal z1 and the second transmission signal z2 that are simultaneously transmitted from the first antenna and the second antenna at the same frequency, respectively. A signal acquisition unit,
The first transmission signal z1 and the second transmission signal z2 are the first modulation signal s1 generated from the first bit group and the second modulation signal generated from the second bit group for each slot. s2 is generated by performing precoding processing, and the precoding processing is performed by N matrices F [i] (where i is an integer of 0 to N−1 and N is an integer of 3 or more). Is a process corresponding to one matrix selected by switching from slot to slot,
The first precoded signal z1 and the second precoded signal z2 satisfy the relationship (z1, z2) ^T = F [i] (s1, s2) ^T
The N matrices F [i] are

Where λ is an arbitrary angle, α is a positive real number excluding 1,
θ ₁₁ (i) and θ ₂₁ (i) are

Or
The filling,
Each of the N matrices is selected at least once within a predetermined number of slots;
Furthermore, the receiving apparatus includes a signal processing unit that generates received data by demodulating the received signal according to a matrix selected for each slot on the transmitting side.