JP5492953B2 - High frequency circuit - Google Patents

Description

  The present invention relates to a high-frequency circuit that takes into account the influence of surrounding dielectrics.

  An antenna that transmits and receives radio signals in wireless communication generally has a frequency band in which radiation (including reception) efficiency is higher than the frequency of radio waves that are transmitted and received. Therefore, the antenna is designed according to the frequency of wireless communication. In the conventional antenna technology, studies have been made to expand the frequency band of the antenna in order to improve the versatility of the antenna, and an antenna having a wide frequency band is called a broadband antenna.

  The frequency used for radio tags such as RFID tags is regulated by the radio law of each country. For example, in the UHF band, it is around 867 MHz in Europe, around 915 MHz in the United States, and around 953 MHz in Japan. There is a difference of about 10% in (wavelength). For this reason, studies are also being conducted to expand the frequency band of RFID antennas.

  On the other hand, since the RFID tag is generally attached to some object and used to track the location of the object, the characteristics of the antenna change due to the influence of the relative dielectric constant of the object. This problem is solved by designing the RFID antenna according to the object to be attached. However, since a large variety of RFID antennas are produced in small quantities, the cost of the RFID tag is increased. For this reason, there is a need for a general-purpose RFID tag that does not select an object to be attached.

Since one end of the influence of relative permittivity appears as a change in transmission wavelength, a method of adopting a broadband antenna as an RFID antenna has been proposed as a countermeasure against the influence of relative permittivity. (For example, Patent Document 1)
However, in this proposal, only the change in the wavelength of the radio wave is considered for the influence of the dielectric on the antenna. This is because the change of only the wavelength (change of the frequency) can be seen at a glance with a Smith chart or the like, and the antenna widening technology accumulated in the past can be applied. In other words, this is because the measurement / evaluation means and design method of the broadband antenna are enhanced to some extent.

  On the other hand, it is not only the wavelength of radio waves that changes due to the influence of the dielectric. The wave impedance of the space where the dielectric affects is also changed. However, the actual situation is that no consideration has been given to the effect of changes in wave impedance.

  In addition, conventionally proposed broadband antennas are generally antennas that match in a wide band when the imaginary part of the impedance of the element connected to the feeding point is zero. On the other hand, since the RFID tag chip is required to be inexpensive, it generally does not have a matching circuit. Therefore, the impedance of the RFID tag chip has a negative imaginary part whose absolute value is larger than the absolute value of its real part (for example, Non-Patent Document 1).

JP 2006-235825 A

2005 Energy Use Rationalization Electronic Tag System Development Survey Project (Development of UHF Band Electronic Tag Manufacturing Technology and Packaging Technology) Report Volume 1 http://www.meti.go.jp/policy/it_policy/tag/tagtyousakenkyuu (hibiki) .htm http://www.meti.go.jp/policy/it_policy/tag/tag_pdf/hibiki1.pdf (88 pages)

  In order for the RFID tag to have good communication characteristics without selecting an attachment target, it is necessary to match the antenna impedance and the impedance of the RFID chip without selecting the attachment target. Conventionally, nothing has been considered about this point.

  The present invention has been made in consideration of the above circumstances, and an object thereof is to provide a high-frequency circuit capable of obtaining good communication characteristics without selecting an attachment target by removing the influence from surrounding dielectrics.

The high-frequency circuit of the present invention is a high-frequency circuit in which a first element and a second element are connected by a transmission line, and is provided between the connection of the first element and the second element. Matching means for canceling the change in the transmission wavelength shortening rate on the circuit due to the relative permittivity by the change in the characteristic impedance of the transmission line due to the effective relative permittivity; When the power loss is Ploss and the real part of the impedance Zchip of the first element is Re [Zchip], the real part of the impedance Zacl of the feeding point viewed from the first element where the power loss is within Ploss. The range is a range between an upper limit value and a lower limit value represented by the following Expression 25.

The figure which shows the fundamental structure of the RFID tag in one Embodiment of this invention. The figure which shows the equivalent circuit of FIG. The figure which shows the dielectric constant concerning one Embodiment as a reference according to a substance. The figure which shows the structure of the A-type antenna of the specific example of one Embodiment. The figure which shows the structure of the planar antenna of the specific example of one Embodiment. The figure which shows the change of the real part of the admittance Ya of the antenna radiation | emission part of FIG. The figure which shows the change of the imaginary part of the admittance Ya of the antenna radiation | emission part of FIG. The figure which shows the structure of the RFID antenna of the specific example of one Embodiment. The figure which shows the cross section which follows the AA line of FIG. The figure which shows the structure where the line | wire space | interval of the RFID antenna of FIG. 8 expanded. The figure which shows the antenna characteristic simulation result in the RFID antenna of FIG. The figure which shows the structure of the RFID antenna of the modification of one Embodiment. Smith chart related to the modification. The figure which shows the structure which the space | interval of the open end of the open stub of the RFID antenna of FIG. 12 expanded. The figure which shows another structure with which the space | interval of the open end of the open stub of the RFID antenna of FIG. 12 expanded. The antenna characteristic simulation result in a modification is shown. The figure which shows the fundamental structure in the case of applying this invention to antennas, such as a portable terminal.

Hereinafter, an embodiment of the present invention will be described with reference to the drawings.
[1] Basic configuration
A basic configuration of a high-frequency circuit such as an RFID tag (wireless tag) is shown in FIG. An antenna radiating section (including reception; also referred to as a radio wave radiating section) 3 as a second element for transmission / reception is connected to an RFID tag chip (wireless tag chip) 1 as a first element via transmission lines 2a and 2b. Further, a closed stub 4 is connected to a connection point between the transmission lines 2 a and 2 b and the antenna radiating unit 3.

  The transmission lines 2 a and 2 b and the close stub 4 function as a matching circuit (matching means) between the RFID tag chip 1 and the antenna radiating unit 3.

  An equivalent circuit of this RFID tag is shown in FIG. That is, the parallel circuit of the antenna radiating unit 3 and the closed stub 4 is connected to the RFID tag chip 1 via the transmission lines 2a and 2b, and the impedance of the antenna radiating unit 3 is Za and the impedance of the closed stub 4 is Zc. The impedance of the RFID tag chip 1 is represented by Zp, and the length corresponding to the transmission lines 2a and 2b is represented by Lo.

  In order for the RFID tag chip 1 to match the antenna radiating unit 3, the complex conjugate of Zp and the feed point need to be close to each other.

  Generally, the directivity required for an antenna is determined depending on the use application, and the shape for realizing the directivity is limited. As a result, the impedance Za of the antenna radiating portion 3 is also largely determined by the intended use. Therefore, here, the impedance Za of the antenna radiating portion 3 is assumed to be known. The impedance Zp of the RFID tag chip 1 is also known.

  In the present embodiment, a concept of designing an RFID antenna that is hardly affected by surrounding dielectric materials when Za and Zp are known will be described.

Here, if the impedance of the connection point between the closed stub 4 and the antenna radiating unit 3 is Zac, and the impedance of the feeding point when the antenna is viewed from the RFID tag chip 1 side is Zacl, Zacl is a characteristic of Zac and transmission lines 2a and 2b. It is expressed as Equation 1 by impedance Zo.

Here, assuming that the complex number Zac is RZac + j × IZac, the real part of Zacl is expressed by Formula 2 and the imaginary part is expressed by Formula 3.

Here, when tan βLo is around “0” to “1” and Zo is sufficiently larger than | Zac |, Equation 2 can be approximated by Equation 4, and Equation 3 can be approximated by Equation 5.

  Note that tan βLo is about “0” to about “1” when the line length Lo is about 1/8 or less of the wavelength in the corresponding line.

Here, the influence of the dielectric around the RFID tag on the RFID antenna is represented by an effective relative dielectric constant εeff on the antenna element. In this case, the wavelength shortening rate on the antenna element is expressed by Equation 6.

  For reference, the relative permittivity of main substances is shown in FIG. The ratio ε / εo between the dielectric constant ε and the relative dielectric constant εo of vacuum is called the relative dielectric constant. The relative dielectric constant is a dimensionless quantity and takes a constant value regardless of the unit system used.

Further, the characteristic impedance value affected by only the transmission wavelength in the transmission lines 2a and 2b is expressed by Equation 7, and the change in Zac affected by only the transmission wavelength is expressed by Equation 8.

In this case, the change of Zo and Zac with respect to the effective relative dielectric constant εeff is expressed as Equation 9 and Equation 10 using Equation 7 and Equation 8 in this embodiment.

  In addition, the change of the value in the parenthesis in Formula 9 and Formula 10 has shown the influence of the transmission wavelength shortening rate by the change of an effective relative dielectric constant. In addition, the denominators in Equations 9 and 10 indicate the influence of the change in the wave impedance around the antenna space due to the change in the effective relative permittivity.

  The relationship between Zo and effective relative permittivity εeff shown in Equation 9 is known as a transmission line characteristic, and the relationship between Zac and effective relative permittivity εeff shown in Equation 10 is the result of actual measurement or calculation by an electromagnetic simulator. Etc.

  As described above, the impedance Zp of the RFID tag chip 1 has a negative imaginary part having a large absolute value compared to the absolute value of the real part. Therefore, when considering the matching of Zp and Zacl, it is first considered to match the imaginary part of the complex conjugate of Zp with Equation 5, which is an approximate expression of the imaginary part of Zacl.

Here, the imaginary part of Zacl when the effective relative permittivity εeff changes can be expressed by substituting Equation 6 into β of Equation 5, Substituting Equation 9 into Zo, and Equation 10 into Zac. it can. Here, when ZoW is sufficiently larger than | ZacW |, the value in parentheses in Equation 5 is almost “1”. Therefore, the imaginary part of Zacl when the effective relative permittivity εeff changes is as shown in Equation 11. become.

Here, tan (βoLo√εeff) is assumed to be around 1 from zero,

When approximated as follows, Formula 11 is approximated as Formula 13.

  From Equation 13, it can be seen that the effective relative permittivity εeff has only a change in the transmission wavelength with respect to the characteristic impedance ZoW on the imaginary part of Zacl. This is a result of canceling the influence of the change of tan (βoLo√εeff) on the effective relative dielectric constant εeff in Equation 11 and the change of the wave impedance expressed by the denominator √εeff.

  ZoW is the characteristic impedance of the transmission lines 2a and 2b. For example, when the transmission lines 2a and 2b are coplanar strip lines, the characteristic impedances of the transmission lines 2a and 2b are constant even if the transmission wavelength changes. That is, ZoW (λo / √εeff) = ZoW. Therefore, when tan (βoLo√εeff) is about “0” to “1” and ZoW is sufficiently larger than | ZacW |, Equation 13 is established, and the imaginary part of Zacl is determined by the effective relative permittivity εeff. It becomes constant.

  If the ZoW transmission lines 2a and 2b have a line cross section that is robust to changes in transmission wavelength, tan (βoLo√εeff) is about “0” to “1”, When ZoW is sufficiently larger than | ZacW |, the imaginary part of Zacl has characteristics that are robust to changes in the effective relative permittivity εeff.

As described above, in order to match the antenna radiating unit 3 and the RFID tag chip 1, it is necessary to match Zacl with the complex conjugate of the impedance Zp of the RFID tag chip 1. That is, −Im [Zp] is made to agree with Expression 11. From this, the length Lo of the transmission lines 2a and 2b connecting the connection point of the antenna radiating unit 3 and the close stub 4 to the feeding point of the RFID tag chip 1 can be obtained by Expression 14.

  The inverse function of tan has a plurality of values, but here, Lo is obtained from the smallest value of “0” or more among the plurality of values.

  Here, assuming that the structure of the line cross section for determining the characteristic impedance ZoW is known and the range of the effective relative permittivity to be considered in the design of the RFID antenna is ε1 to ε2, Zp and the above two types of εeff are substituted into Equation 14. Then, two types of line lengths Lo1 and Lo2 are obtained. The actual line length may be determined within the range of these two values. The larger the characteristic impedance ZoW is, the smaller the general angle of tan in Equation 11 becomes, and Equation 12 holds. As a result, the change in Im [Zacl] between ε1 and ε2 is reduced, and the difference between the line lengths Lo1 and Lo2 is also reduced. For this reason, it is desirable that the ZoW is as large as possible within a range that does not affect antenna characteristics such as antenna directivity.

  On the other hand, it can be seen from Equation 14 that the line lengths Lo1 and Lo2 become smaller as the characteristic impedance ZoW increases. In general, in order to give a large characteristic impedance to the transmission lines 2a and 2b, a large line spacing is required. If the value of the line length Lo1 is smaller than the line spacing between the transmission lines 2a and 2b necessary for realizing ZoW, the characteristics of the transmission lines 2a and 2b are not stable. Therefore, when considering the stability of the characteristics of the transmission lines 2a and 2b, it is desirable that Lo be a value sufficiently larger than the line interval between the transmission lines 2a and 2b. From the above, it can be seen that the upper limit of the ZoW is limited by the line spacing and length of the transmission lines 2a and 2b.

Here, for reference, the condition that the imaginary part of Zacl is robust against changes in frequency will be considered using Equation 5. The change of the imaginary part of Zacl with respect to the frequency f is expressed as Equation 5, Equation 9, and Equation 10 to Equation 15.

Here, εeff is “1”, and ZoW (λ) is constant with respect to the change of the transmission wavelength as described above. Therefore, in order to reduce the change in the imaginary part of Zacl with respect to the frequency f (that is, with respect to the transmission wavelength λ), it is necessary to cancel the change in “tanβo (λo / λ) Lo” with the change in the value in the parentheses in Equation 15. is there. Therefore, | ZacW | needs to be a certain size relative to ZoW. For example,
When “tanβo (λo / λ) Lo” changes from “0.5” to “1”, | ZacW | / ZoW needs to change from “0” to about “0.707”. When | ZacW | / ZoW is “0.707”, it cannot be said that ZoW is sufficiently larger than | ZacW |.

That is, the condition that the imaginary part of Zacl is robust against the change in effective relative permittivity εeff
It can be seen that ZoW is sufficiently larger than | ZacW | and a condition that is robust against changes in frequency is in a trade-off relationship.

  Note that the difference between a condition that is robust with respect to the change in effective relative permittivity εeff and a condition that is robust with respect to a change in frequency naturally manifests itself in the shape of the antenna. In an antenna that is robust to changes in frequency, that is, a wideband antenna for RFID, | ZacW | needs to be somewhat larger than ZoW as described above. When the characteristic impedance ZoWc of the closed stub of the RFID broadband antenna and the antenna that is robust with respect to the effective relative dielectric constant εeff is the same, the line length Lc of the closed stub 4 needs to be increased in order to increase | ZacW |. From the above, the wideband antenna for RFID has a longer line length of the closed stub 4 than an antenna robust to the effective relative dielectric constant εeff.

From the above, it was found that ZoW and Lo are determined from the imaginary part of Zp. At this point, the only unknown in FIG. 2 is Zc of the closed stub 4. Zc may be obtained from the condition for matching the real part of Zp with Formula 2 or Formula 4. In the present embodiment, a procedure for determining Zc from Equation 4 and the real part of Zp is shown. The condition for matching the real parts of Zacl and Zp is expressed by Equation 16 from Equation 4.

Here, when the reciprocal (admittance) of Zac is Yac, Equation 17 is obtained.

  Since the closed stub 4 and the antenna radiating unit 3 are connected in parallel, Yac is represented by the addition of the admittance Yc of the closed stub 4 and the admittance Ya of the antenna radiating unit 3. Here, when Expression 6 is substituted for β in Expression 17, the condition for matching the real parts of Zacl and Zp is Expression 19.

When the line length of the closed stub 4 is Lc, the change of Yc with respect to the effective relative dielectric constant εeff is expressed by Equation 18.

Here, as described above, when the effective relative permittivity range corresponding to the RFID tag to be designed is ε1 to ε2, the radiation part admittance when the effective relative permittivity on the antenna element is ε1 and ε2. Measure Ya1 and Ya2, or calculate by electromagnetic field simulation. Substituting Ya1 into Ya in Equation 19 and substituting ε1 and Lo into Equation 19 yields Yc. Then, if YoWc (λo / √ε1) and ε1 are substituted into Equation 18, Lc1 is obtained. Similarly, Lc2 can be obtained by substituting each parameter in the case of ε2 into Equation 19. From the above, the line length Lc of the closed stub 4 may be determined within the range of the above two values.

  Note that ZoWc in Equation 18 is the characteristic impedance of the line of the closed stub 4 when εeff = 1. A connection point between the antenna radiating unit 3 and the close stub 4 is connected to transmission lines 2 a and 2 b that reach the feeding point of the RFID tag chip 1. Therefore, here, the transmission lines 2a and 2b and the closed stub 4 have the same cross-sectional shape. That is, ZoW = ZoWc.

  In addition, since the right side of Equation 19 is square, two types of Yc are obtained when the effective relative permittivity on the antenna element is ε1, but the value Lc in Equation 18 that is the minimum when Lc is zero or more is the value of Lc1. To do. The same applies to Lc (Lc2) when the effective relative permittivity on the antenna element is ε2.

  So far, the RFID tag antenna that is not easily affected by the dielectric when the impedance Zp of the RFID tag chip 1 does not change when the effective relative permittivity is ε1 and ε2 has been described. However, since the RFID tag chip 1 is also an electric circuit, the value of Zp changes under the influence of a dielectric around the chip. Thereafter, various antenna parameters when the impedance of the RFID tag chip 1 mounted at the feeding point of the antenna element when the effective relative permittivity on the antenna element is ε1 is Zp1, and similarly when Zp2 is ε2 Describe.

First, parameters ZoW and Lo of the transmission lines 2a and 2b connecting the connection point between the antenna radiating unit 3 and the close stub 4 to the feeding point of the RFID tag chip 1 will be considered. The ZoW of the transmission lines 2a and 2b is considered to be constant with respect to changes in only the transmission wavelength. That is,
Let “ZoW (λo / √εeff) = ZoW”.

  When the equation obtained by substituting Zp1 and ε1 for Zp and εeff in Equation 14 and the equation obtained by substituting Zp2 and ε2 are combined, and Lo is eliminated, Equation 20 is obtained. On the other hand, when ZoW is deleted, Equation 21 is obtained. ZoW may be a value that satisfies Equation 20, and Lo may be a guideline for adjusting the value that satisfies Equation 21 later.

In addition, the rough indication of Lo may be obtained by substituting ZoW, ε1, Zp1 in which Formula 20 is satisfied (or ZoW, ε2, Zp2 in which Formula 20 is satisfied) into Formula 14.

In addition, when calculating | requiring Lo previously, Lo is taken as Lo in which Numerical formula 21 is materialized. As for ZoW, a value obtained by substituting Lo, ε1, and Zp1 in which Equation 21 is established into Equation 22 may be used as a guideline for subsequent adjustment.

  From the above, if the antenna is designed with ZoW and Lo satisfying Equations 20 and 21, Im [Zacl] and -Im [Zp] are almost the same even if the effective relative permittivity changes. As can be seen from Equation 9, ZoW calculated by Equation 20 is the same as Zo when the effective relative dielectric constant is “1”.

When “βoLo√ε2” is 90 °, Im [Zacl] according to Equation 11 becomes infinite. Assuming that −Im [Zp] is never infinite, the value that Lo can take on the antenna is limited by Equation 23.

Within this limit, Equation 24 holds between Zp1 and Zp2.

  Equation 24 is for a case where ε2 is larger than ε1. When ε2 is smaller than ε1, the relationship between Zp1 and Zp2 in Equation 24 is switched.

  From the above, when the formula 24 is not established between Zp1 and Zp2 or the ZoW calculated by the formula 20 (or formula 22) is very large or very small, the formula 20 ( Alternatively, there are cases where the ZoW calculated by Equation 22) cannot be reproduced. In such a case, the cross-sectional shape of the transmission lines 2a and 2b having the value closest to the calculated value in the characteristic impedance that can be configured by the matching circuit is set as the cross-sectional shape of the new transmission lines 2a and 2b. Then, a new “ZoW (λo / √εeff)” is obtained from the cross-sectional shape. Substituting ε1 and Zp1 into Formula 14, and further substituting a new “ZoW (λo / √εeff)”, finds Lo1. Substituting ε2 and Zp2 into Equation 14 and further substituting new “ZoW (λo / √εeff)”, finds Lo2. Then, the newly calculated value within the range of Lo1 and Lo2 can be used as an indication of Lo when adjusting later.

  The case where Zp does not change when the effective relative dielectric constant is ε1 and ε2 can be considered as the case where Zp1 and Zp2 are equal.

  Next, the line length of the closed stub 4 when the value of Zp changes under the influence of the change in the effective relative permittivity on the antenna element is obtained from Expression 18 and Expression 19. That is, Yc is obtained by substituting Zp1 for Zp in Equation 19, substituting Ya1 for Ya, substituting ε1 for εeff, and substituting Lo. Substituting Yc, ε1, and “YoWc (λo / √ε1)” into Equation 18, Lc1 is obtained. Similarly, if each parameter in the case of ε2 is substituted into Equation 18 and Equation 19, Lc2 is also obtained. From the above, the line length Lc of the closed stub 4 may be set within the range of the above two values.

The relationship between the cross-sectional shape of ZoW and the transmission line can be calculated on the following website.
http://www1.sphere.ne.jp/i-lab/ilab/
http://www1.sphere.ne.jp/i-lab/ilab/tool/cps.htm
[2] Specific examples
Next, a specific example will be described. Here, an RFID label antenna (RFID antenna) used for physical distribution management will be described as an example. The RFID antenna is defined as a portion obtained by removing the RFID tag chip from the RFID tag.

  An RFID label for logistics management is affixed to a corrugated cardboard and used for storage management. At this time, the RFID antenna is affected by the contents of the cardboard. Therefore, the RFID reading area changes depending on the contents of the cardboard. For example, when reading RFID label information with a handheld RFID reader, the user of the handheld RFID reader needs to grasp the RFID label reading area that differs depending on the contents of the cardboard. Furthermore, the handy RFID reader must be moved to the reading area for each cardboard. Due to the above factors, the RFID label for logistics management needs to have stable characteristics regardless of the contents of the cardboard.

  Next, the directivity required for the RFID label is considered. In entry / exit management, it is necessary to read the RFID label stably, and at the same time not to read the RFID label unintended by the user of the handy type RFID reader. This is because if the handy type RFID reader reads the label information other than the RFID label of the article that is actually entering / exiting, inconsistency occurs in the entry / exit management data. Therefore, an RFID label reading area must be known at a glance for an operator who reads label information. For example, when an RFID label is affixed to the upper surface or the lower surface of the cardboard, it is difficult to visually confirm the direction of the label, and the RFID label reading area is unclear. From the viewpoint of clarifying the reading area of the RFID label for the user of the handy type RFID reader, it is preferable that the reading area of the RFID label is only in the direction perpendicular to the label surface. In this case, it becomes clear that the RFID label that is horizontal when viewed from the user of the handheld RFID reader cannot be read from the position of the user.

  Therefore, the RFID label antenna is an antenna whose maximum gain direction is perpendicular to the label surface regardless of the contents of the cardboard.

  The effect of the contents of the cardboard can be expressed by a change in the effective relative permittivity on the antenna element. As described above, both the change in the transmission wavelength and the change in the wave impedance due to the change in the effective relative permittivity have a great influence on the antenna impedance. On the other hand, the antenna directivity is dominated by the change in the transmission wavelength. Therefore, if the antenna is stable in a wide band with respect to directivity, the directivity is stable even when the effective relative permittivity changes.

  An A-type antenna is known as a broadband antenna having a maximum gain direction in the axial direction of the antenna surface. As shown in FIG. 4, the A-type antenna includes a ground plane, a feeding point existing on the ground plane, and a right-angled triangular antenna radiating portion 3 that rises from the feeding point and is in contact with the ground plane. If it is considered that a line through which an image current flows is provided on the opposite side of the ground plane instead of the ground plane, it can be considered as a planar antenna having a symmetrical antenna radiation portion 3 as shown in FIG. Therefore, here, it is considered that the antenna radiating portion 3 of the antenna as shown in FIG.

  The matching frequency f is 915 MHz, which is an intermediate frequency among the RFID frequencies in each country. In this case, the wavelength λo when the effective relative dielectric constant is “1” is about 328 mm.

  Next, effective relative dielectric constants ε1 and ε2 to be considered in the design of the RFID antenna are determined. Conventionally proposed broadband antennas have many antennas that match in a wide band when the imaginary part of the impedance (admittance) of the element connected to the feeding point is close to “0”. 3 is the same. In this case, within the effective relative permittivity range in which the directivity of the antenna is stable, the real part and the imaginary part of the feed point impedance (admittance) repeat increase and decrease periodically as the effective relative permittivity increases. Further, when the real part becomes an extreme value in the cycle, the imaginary part corresponding to the real part takes a value close to “0”. 6 and 7 are graphs showing changes in the real part and the imaginary part of the admittance Ya of the antenna radiating part 3 in FIG. 5 in accordance with the change in the effective relative permittivity. The periodicity of the feeding point admittance described above can be seen from FIGS.

  Here, paying attention to Equation 19, it can be seen that Yc satisfying Equation 19 is also affected by the period of the real part and the imaginary part of Ya. Therefore, Yc also has periodicity with respect to changes in the effective relative dielectric constant. In general, the real part Re [Zp] of the chip impedance is a small value. From Equation 16, RZac needs to be smaller than Re [Zp]. In order to reduce Rzac, it is necessary to reduce the impedance Zc of the closed stub 4. As a result, the admittance Yc of the close stub 4 is a very large value. Since the absolute value of the imaginary part of Ya is smaller than the absolute value of the imaginary part of Yc, it can be seen that it is the real part of Ya that has a dominant influence on the periodicity of Yc that satisfies Equation 19. When Ya becomes the maximum value or the minimum value in an arbitrary range εarea of the effective relative permittivity, it is considered that Yc corresponding to the value is close to the upper limit and the lower limit in the range εarea. Therefore, the effective relative permittivity range to be considered in the antenna design of this embodiment is determined from the real part of Ya.

In this embodiment, the effective relative dielectric constant at which the real part of Ya in the range εarea is an extreme value is ε1,
After ε1, let ε2 be the effective relative dielectric constant of the next extreme value that the Ya real part reaches. The larger the difference between ε1 and ε2, the more difficult it is to design an antenna that is robust against changes in the effective dielectric constant. In that case, the upper limit εmax of the range εarea is made smaller than ε2 or the lower limit εmin is made larger than ε1. Then, new ε1 and ε2 may be determined from the minimum and maximum values of Ya in the new range εarea2 and the value of the effective relative permittivity that provides them.

  In the present embodiment, ε1 is the relative dielectric constant (approximately 1) of the atmosphere. FIG. 6 shows that when the effective relative permittivity is 1, the Ya real part has a maximum value. In FIG. 6, the peak of the minimum value after the effective relative permittivity of 1 is around the effective relative permittivity “4”. Therefore, in this embodiment, ε2 is an effective relative dielectric constant “4”.

  Next, the impedances Zp1 and Zp2 of the RFID tag chip 1 when the effective relative dielectric constant values ε1 and ε2 determined as described above are obtained. The impedance of the RFID tag chip 1 changes due to the influence of the bonding material of the chip and the antenna element, the mounting method, etc. in addition to the effective relative dielectric constant. In the present embodiment, Zp1 and Zp2 are the impedances on the chip side when the influence of the bonding material and the mounting method is taken into consideration.

  For example, the antenna shape is adjusted so that the reading distance is maximized in a state where the RFID antenna on which the chip is mounted is buried in an object having a relative dielectric constant of ε1. Since this antenna shape is for obtaining Zp1 and Zp2, a simple configuration in which a chip is mounted on a dipole antenna and a close stub is connected may be used. In this case, the adjustment parameters of the RFID antenna are the length of the dipole antenna and the line length of the closed stub.

  An antenna with the same shape as that of the antenna after adjustment and without a chip is prepared, and the feed point impedance is measured in a state where the antenna is buried in an object having a relative dielectric constant of ε1. This measured value is the complex conjugate of Zp1.

  Alternatively, the antenna shape after adjustment is input to an electromagnetic field simulator or the like, and the environment where the reading distance is measured is input, and then the feeding point impedance is calculated. This calculated value is the complex conjugate of Zp1.

  In the same way, the complex conjugate of Zp2 when the effective relative permittivity is ε2 can also be obtained.

  Further, the antenna gain also affects the RFID reading distance. In some cases, the antenna gain varies greatly with antenna adjustment. In this case, the antenna shape in which the value obtained by subtracting the influence of the antenna gain from the reading distance is specified is specified. Then, the feeding point impedance of the specified antenna shape is measured or calculated.

  Note that objects having effective relative dielectric constants ε1 and ε2 are objects whose sizes are larger than the near field of the RFID antenna.

  In the RFID tag chip 1 in this embodiment, for example, Zp1 is “10-j140” and Zp2 is “10-j190”.

  When Zp1, Zp2, ε1, and ε2 are substituted into Equation 20, ZoW is about 273Ω. Next, when Zp1, Zp2, ε1, and ε2 are substituted into Equation 21, Lo becomes approximately 24.7 mm. Here, transmission lines 2a and 2b connecting the connection point between the antenna radiating unit 3 and the close stub 4 to the feeding point of the RFID tag chip 1 are assumed to be corporate strip lines. In this case, ZoW is constant regardless of the frequency (wavelength). Then, by substituting ZoW and ε1 or ε2 into Equation 9, the characteristic impedance Zo of the transmission lines 2a and 2b at the time of ε1 or ε2 can be calculated. In this embodiment, since ε1 is “1”, the characteristic impedance when ε1 is about 273Ω, and the characteristic impedance when ε2 = “4” is about 136.5Ω.

  Next, it is confirmed whether a corporate strip line with a characteristic impedance of 273Ω when the effective relative dielectric constant is “1” is practically possible. The corporate strip line has a characteristic impedance of about 295Ω when the line spacing is 4 mm, the line width is 2 mm, and the effective relative permittivity is “1”. When performing fine adjustment of the antenna shape, which will be described later, the amount of change in Zacl is adjusted with respect to Zp that changes depending on the effective relative dielectric constant. At that time, it is easier to match the change of Zp when the change of Zacl is smaller than when the change of Zacl is large. For this reason, it is desirable that ZoW is slightly larger than the calculation result of Expression 20. In addition, when ZoW is large, Lo becomes smaller than the calculation result of Expression 14. Therefore, 295Ω is ZoW of this embodiment, and 22 mm is a guide for the length of Lo.

  The change in Zp is determined when the RFID tag chip 1 is selected, whereas the change in Zacl can be controlled by the antenna designer. Therefore, in order to make it easy for the antenna designer to match Zp and Zacl, the characteristics of the transmission lines 2a and 2b connecting the connection point of the antenna radiating unit 3 and the close stub 4 to the feeding point of the RFID tag chip 1 as described above. Determine impedance ZoW and line length Lo.

  Next, Lc is obtained from Equation 18 and Equation 19. Ya1 and Ya2 to be substituted for Ya in Expression 19 are assumed to be Ya1 = 0.00735 + j × 0.00159 and Ya2 = 0.00169 + j × 0.00215 from FIG. Further, it is assumed that the line cross-sectional shape of the closed stub 4 is the same as that of the transmission lines 2a and 2b. In this case, YoWc is the reciprocal of ZoW. Substituting the parameters at ε1 into Equations 18 and 19, Lc1 is about 5.8 mm. Substituting the parameters at ε2 into Equations 18 and 19, Lc2 is about 7.9 mm. In actual antenna manufacturing, it is difficult to control the antenna shape in units of 0.1 mm. Moreover, the ratio of the line | wire space | interval (4mm) with respect to each Lc is large. Therefore, in this embodiment, 8 mm is used as a guide for the length of Lc.

FIG. 8 shows a configuration of an RFID antenna in which a matching circuit is added to the planar antenna of FIG.
That is, a notch with a predetermined interval is formed in the approximate center of the bottom part of the isosceles triangle antenna radiating portion 3, and a pair of transmission lines 2 a is formed in a substantially vertical direction from both ends of the notch toward the inside of the antenna radiating portion 3. , 2b rise, and the transmission lines 2a, 2b are bent so that the front ends face each other and are connected to both ends of the RFID tag chip 1, respectively. The transmission lines 2a and 2b have the same line width, and the transmission line interval between the transmission lines 2a and 2b is wider than the transmission line width. Each of the transmission lines 2a and 2b has a line width of, for example, 2 mm and a line interval of, for example, 4 mm, and is a printed wiring on the pet material 10 as shown in FIG. 9 along the line AA. It is installed with an interval between tracks. The width of the RFID tag chip 1 is, for example, about 1 mm.

  Furthermore, a U-shaped closed stub 4 is connected to both ends of the notch so as to slightly protrude in the direction opposite to the rising direction of the transmission lines 2a and 2b. In the closed stub 4, the distance between both leg portions extending vertically from both ends of the notch is 4 mm, for example, which is the same as the distance between the transmission lines 2a and 2b, and the line width is 2 mm, for example, the same as the transmission lines 2a and 2b.

  In the RFID antenna of FIG. 8, the length Lo of the transmission lines 2a and 2b corresponds to the length of the dotted line path from the α point to the γ point, and the length Lc of the close stub 4 is the dotted line from the α point to the β point. Corresponds to the length of the path. In this case, the distance between the leg portions of the closed stub 4 which is a portion protruding outward from the notch portion of the isosceles triangle antenna radiating portion 3 is, for example, 4 mm, and therefore the length Lc of the closed stub 4 is Lc2 having a large value. Is rounded up. However, even for a length of 8 mm, the proportion of leg spacing is relatively large. Therefore, the leg interval of the closed stub 4 may be reduced in order to reduce the ratio of the size of the leg interval to Lc. For example, if the leg spacing of the closed stub 4 is 2 mm and the line width is 2 mm, “YoWc” is about 0.004.

  In general, when the line interval between the transmission lines 2a and 2b is decreased, the value of “YoWc” increases. Substituting the parameters at ε1 into Equations 18 and 19, Lc is about 7.1 mm. Substituting the parameters at ε2 into Equations 18 and 19, Lc is about 9.6 mm. From the above, the standard of Lc may be 9 mm.

  In this case, since the distance between the transmission lines 2a and 2b is different from the distance between the legs of the close stub 4, the shape of the RFID antenna is as shown in FIG. That is, a notch with a predetermined interval is formed at substantially the center of the bottom of the isosceles triangle antenna radiating portion 3, and a pair of transmission lines 2 a and 2 b are temporarily provided from both ends of the notch toward the inside of the antenna radiating portion 3. It stands up in an expanding state and extends in a direction substantially perpendicular to the bottom side of the antenna radiating portion 3. The front ends of the transmission lines 2a and 2b are bent so as to face each other and connected to both ends of the RFID tag chip 1, respectively. Each of the transmission lines 2a and 2b has a line width of, for example, 2 mm and a line interval of, for example, 4 mm, and the RFID tag chip 1 has a width of, for example, about 1 mm. Furthermore, a U-shaped closed stub 4 is connected to both ends of the notch so as to slightly protrude in the direction opposite to the rising direction of the transmission lines 2a and 2b. In the closed stub 4, the distance between both leg portions extending vertically from both ends of the notch is 2 mm, for example, which is narrower than the distance between the transmission lines 2 a and 2 b, and the line width is 2 mm which is the same as the transmission lines 2 a and 2 b, for example.

  FIG. 11 shows a simulation result of the gain in the direction perpendicular to the plane formed by the real part and the imaginary part of the feed point impedance Zacl of the RFID antenna of FIG. Due to factors such as approximation error, measurement error, and simulation error when calculating each parameter, each value in FIG. 11 deviates from the complex conjugate of the impedance Zp of the RFID tag chip 1. Therefore, in designing an actual RFID antenna, an RFID in which chips are mounted on a plurality of antennas having slightly different shape parameters is prepared, and the RFID reading distance in each antenna shape is measured to determine the actual antenna shape. .

  For example, in the case of the RFID antenna of FIG. 8, the line widths and line intervals of the transmission lines 2a and 2b can be changed by scraping the transmission lines 2a and 2b. Therefore, using a Lo value calculated as described above as a guide, a plurality of RFID antennas having slightly different Los are prepared, and the change of the line width and line spacing of the transmission lines 2a and 2b of each RFID is dealt with by scraping. .

[3] Modification
As a modification, there is an RFID antenna having a shape shown in FIG. That is, the open stubs 5a and 5b are connected to the bent portions of the transmission lines 2a and 2b toward the RFID tag chip 1 side. The open stubs 5a and 5b are composed of a pair of lines extending straight in the rising direction of the transmission lines 2a and 2b, and function as a matching circuit (matching means) together with the transmission lines 2a and 2b and the closed stub 4.

  In this case, the line lengths of the transmission lines 2a and 2b are indicated by Lo1, and the line lengths of the open stubs 5a and 5b are indicated by Lo2. The line length Lo2 of the open stubs 5a and 5b can be adjusted by scraping the open end.

  The larger the line length Lo2 of the open stubs 5a and 5b, the larger the imaginary part of the feeding point impedance. Therefore, when the feeding point impedance Zacl2 is matched with Zp, the length of Lo1 is equal to or less than Lo of the antenna shape of FIG. Therefore, the line lengths of the open stubs 5a and 5b are preferably longer than the line lengths of the transmission lines 2a and 2b.

  If the total length of Lo1 and Lo2 in the RFID antenna of FIG. 12 is substantially the same as Lo of the RFID antenna of FIG. 8, the feeding point impedance of the RFID antenna of FIG. Compared to that, the real part becomes larger and the imaginary part becomes smaller. In addition, the larger the ratio of Lo2 compared to Lo1, the greater the difference in feed point impedance. This can be understood by referring to the Smith chart of FIG.

  In the Smith chart of FIG. 13, if the point X indicates the feed point impedance when Lo of the RFID antenna of FIG. 8 is the same as Lo1, the Lo of the RFID antenna of FIG. 8 is “Lo1 + Lo2”. The feeding point impedance in a certain case exists at the Y point in the Smith chart. The Y point and the X point are on concentric circles in the Smith chart. In addition, the feeding point impedance of the RFID antenna shown in FIG. 12 is at, for example, the Z point on the locus where the admittance real part is constant. It is assumed that the origin of this Smith chart is the characteristic impedance of the transmission lines 2a and 2b.

  On the other hand, as shown in FIG. 14, the open stubs 5a and 5b may be shaped to expand in a direction perpendicular to the transmission lines 2a and 2b.

  FIG. 14 shows a shape in which the mutual distance between the open ends of the open stubs 5a and 5b is maximized, and FIG. 15 shows an example in which the mutual distance between the open ends is narrower than that. That is, the interval between the base ends connected to the transmission lines 2a and 2b of the open stubs 5a and 5b shown in FIG. 15 is narrower than the interval between the open ends of the open stubs 5a and 5b.

  The antenna impedance of the RFID antenna shown in FIG. 8 has a larger imaginary part than the complex conjugate of Zp. Therefore, it is considered that the open stubs 5a and 5b are provided and the antenna shape is adjusted by the line length Lo2 of the open stubs 5a and 5b.

As described above, the line length Lo2 of the open stubs 5a and 5b can be adjusted by scraping the open ends of the open stubs 5a and 5b. Therefore, the values of the line length Lo1 of the transmission lines 2a and 2b are different, and the line length Lo2 of the open stubs 5a and 5b is sufficiently large (e.g., 28 mm) to the extent that the line length Lo2 does not come into contact with the antenna radiating unit 3 Prepare multiple antennas. The value of the line length Lo1 of the transmission lines 2a and 2b of the plurality of RFID antennas to be prepared is shorter than the value of the line length Lo of the transmission lines 2a and 2b of the RFID antenna of FIG. And The RFID tag chip 1 is mounted on each RFID antenna thus prepared, and the line length Lo2 of the open stubs 5a and 5b is slightly cut and changed. For each change, under an environment with an effective dielectric constant between ε1 and ε2, for example εeff = 1,
The RFID reading distance is measured at εeff = 2, εeff = 3, and εeff = 4, and the values of the line length Lo1 of the transmission lines 2a and 2b and the line length Lo2 of the open stubs 5a and 5b are determined.

  Although a plurality of RFID antennas differing only in the value of the line length Lo1 of the transmission lines 2a and 2b are prepared, ZoW and YoWc, that is, the line width and line spacing of the transmission lines 2a and 2b, or the length Lc of the closed stub 4 A plurality of RFID antennas having different values may be prepared.

  As a specific example of the RFID antenna in this modification, the real part of the feeding point impedance Zacl when the line length Lo1 of the transmission lines 2a and 2b is 13.5 mm and the line length Lo2 of the open stubs 5a and 5b is 10 mm FIG. 16 shows a simulation result of the gain in the direction perpendicular to the plane formed by the imaginary part and the antenna radiating part 3.

  Thus, the RFID antenna provided with the open stub has an advantage that it can be easily adjusted as compared with the case of only the closed stub.

  In the above example, the RFID chip has Zp1 of “10-j140” and Zp2 of “10-j190”, and the case where the impedance Zacl of the feeding point viewed from the chip side is matched is described. On the other hand, since RFID is mass-produced to reduce costs, it is necessary to take into account chip variations and antenna shape variations due to mass production. For example, assuming that the imaginary part of each impedance is matched and the real part of the impedance of the RFID chip is “10”, the impedance of the feeding point as seen from the chip side where the power loss is within 1/4 When the range of the real part of Zacl is calculated, it is generally “3.4 ≦ Re [Zacl] ≦ 30”. Thus, the difference between the real part of the impedance and the power loss has a non-linear relationship.

  In order to design the antenna in consideration of the mass production variation in such a case, the design target value of the real part of the impedance Zacl of the feeding point viewed from the chip side is set larger than the real part of the impedance of the RFID chip. That is, the real part of Zp1 and Zp2 in the above example is set to be larger than “10”, and the calculation as described above is performed with the new values of Zp1 and Zp2, and the standard of each dimension of the antenna shape is calculated. This is the same when Zp1 and Zp2 are the same value.

  For example, if the real part of the impedance of the RFID chip is “10” and the power loss is within 1/4, the intermediate value “16.7” between the lower limit “3.4” and the upper limit “30” is new. The real part of Zp1 and Zp2 may be used. Note that the imaginary parts of Zp1 and Zp2 are substantially the same as the impedance Zchip of the first element when the effective relative permittivity corresponding to each of them is set.

Assuming that the imaginary part of each impedance is matched, when the real part of the impedance of the RFID chip is Re [Zchip] and the power loss is Ploss, the upper limit of the power loss within the Ploss range And the lower limit value can be calculated by Equation 25.

  Next, the case where the present invention is applied to an antenna such as a portable terminal will be described. This is an example of a matching circuit using a shielded wire for the first element and an antenna radiating unit for the second element. Unlike an RFID antenna, a general communication device has an imaginary connection terminal for the antenna. Is matched to be “0”. Therefore, a matching circuit with a shield wire connected to the feeding point when Zacl is “10 + j140” when the effective relative permittivity on the antenna element is ε1 and when Zacl is “10 + j190” when ε2 is described. .

  The antenna terminal generally has a characteristic impedance of 50Ω. Further, when the matching circuit is configured by the shield line, the shield line is hardly affected by the outside world, and therefore is not affected by the dielectric around the antenna. Therefore, in this example, the intermediate value “10 + j165” between “10 + j140” and “10 + j190” is matched to 50Ω by the matching circuit using the shield wire.

  For example, consider a single stub matching using a coaxial line with a characteristic impedance of 50Ω as the shield line. When the single stub is configured as shown in FIG. 17 and the transmission wavelength of the coaxial line is λs, the first stub point in the figure is obtained when Ls1 in the figure is about 0.276λs and Ls2 is about 0.021λs. 171 matches at about 50Ω. In the figure, Zos is the characteristic impedance of the coaxial line, and the second stub point 172 is short-circuited as shown.

  In the case of the mobile terminal as described above, a human hand can be considered as a dielectric that affects the antenna on the mobile terminal side. When the portable terminal is a portable RFID reader, it is conceivable that the transmission output is small and the reading distance is short. In this case, the antenna on the mobile terminal side is affected by the RFID sticking target for ID reading. Therefore, when designing the antenna on the portable terminal side, ε1 is set in consideration of the influence of the human hand, and ε2 is set in consideration of the influence of both the human hand and the RFID application target.

  As described above, in the present invention, the matching circuit that cancels the transmission wavelength shortening rate and the surrounding wave impedance change on the RFID tag is provided between the RFID tag chip 1 and the antenna radiating unit 3. It is possible to obtain good communication characteristics without removing the influence from the dielectric material and selecting an attachment object.

  In the above-described embodiment, the case where the antenna 10 is standing upright has been described as an example. However, when the antenna 10 is laid down in parallel with the floor surface, when the antenna 10 is inclined at a predetermined angle, the antenna 10 is suspended from the ceiling surface. The case of lowering can be similarly implemented.

  In addition, the said embodiment is shown as an example and is not intending limiting the range of invention. The novel embodiment can be implemented in various other forms, and various omissions, rewrites, and changes can be made without departing from the spirit of the invention. In these embodiments and modifications, the scope of the invention is included in the gist, and is included in the invention described in the claims and the equivalents thereof.

  DESCRIPTION OF SYMBOLS 1 ... RFID tag chip (1st element), 2a, 2b ... Transmission line, 3 ... Antenna radiation | emission part (2nd element), 4 ... Close stub, 5a, 5b ... Open stub

Claims (9)

In the high-frequency circuit formed by connecting the first element and the second element by a transmission line,
The change in the transmission wavelength shortening rate on the circuit due to the effective relative permittivity of the surrounding dielectric provided between the first element and the second element is changed to the characteristic impedance of the transmission line by the effective relative permittivity. Alignment means to cancel out by changes in
With
When the power loss is Ploss and the real part of the impedance Zchip of the first element is Re [Zchip], the range of the real part of the impedance Zacl at the feeding point viewed from the first element where the power loss is within Ploss is , Which is the range of the upper limit value and the lower limit value represented by the following formula 25

A high-frequency circuit characterized by that. The range of the real part of the impedance Zacl is an intermediate value between the upper limit value and the lower limit value represented by the mathematical formula 25.
The high-frequency circuit according to claim 1. When the characteristic impedance of the transmission line is ZoW, the characteristic impedance ZoW is larger than the absolute value | ZacW | of the impedance ZacW of the second element,
The effective relative permittivity of the surrounding dielectric is εeff, and the effective relative permittivity εeff is a value in the range of ε1 to ε2,
The length of the transmission line is Lo when tanβoLo√εeff in the following Equation 11 is from zero to around 1.

The length Lo of the transmission line is a value larger than the line interval of the transmission line,
The high-frequency circuit according to claim 1. The alignment means includes
Including the transmission line,
When the effective dielectric constant of the surrounding dielectric is ε1, the impedance Zchip of the first element is Zp1, and when the effective dielectric constant of the surrounding dielectric is ε2, the impedance Zchip of the first element is Zp2. When ZoW is a value satisfying the following Equation 20,

The characteristic impedance of the transmission line when the effective relative permittivity of the surrounding dielectric is “1” is equal to or higher than the ZoW.
The high-frequency circuit according to claim 1 or claim 3, wherein The alignment means includes
Including the transmission line,
When the effective dielectric constant of the surrounding dielectric is ε1, the impedance Zchip of the first element is Zp1, and when the effective dielectric constant of the surrounding dielectric is ε2, the impedance Zchip of the first element is Zp2. When Lo is a value satisfying the following Equation 21,

The length of the transmission line is not more than the Lo,
The high-frequency circuit according to claim 1 or claim 3, wherein The alignment means includes
Including the transmission line and the closed stub,
The effective relative permittivity of the surrounding dielectric is εeff, the effective relative permittivity εeff is a value in the range of ε1 to ε2, the admittance of the antenna radiation portion of the second element is Ya, and the first element When the real part of the impedance Zchip is Re [Zchip], the range of Re [Zp] is an intermediate value between the upper limit value and the lower limit value represented by Equation 25.

The admittance of the closed stub is the same as Yc that satisfies the mathematical formula 19.
6. The high frequency circuit according to claim 3, wherein the high frequency circuit is provided. The power loss Ploss is within 1/4.
The high-frequency circuit according to claim 1 or 6, characterized in that The alignment means includes
Including the transmission line,
When the effective dielectric constant of the surrounding dielectric is εeff, the impedance Zchip of the first element is Zp, and Lo and “ZoW (λo / √εeff)” satisfy the following formulas:

The length of the transmission line is not less than the line width of the transmission line and not more than Lo.
The characteristic impedance of the transmission line is not less than the “ZoW (λo / √εeff) / √εeff”.
The high-frequency circuit according to claim 1 or claim 3, wherein When the effective relative permittivity of the surrounding dielectric is εeff and the range of change of the effective relative permittivity εeff is εarea, the admittance Ya of the antenna radiation portion of the second element in the εarea The real part of is the maximum or minimum value,
9. The high-frequency circuit according to claim 3, wherein the high-frequency circuit is any one of claims 3 to 6.

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