JP5032626B2 - Amplifier circuit, gyrator circuit, filter device and method for amplifying a signal - Google Patents

Description

  The invention relates to an amplifier circuit comprising at least one transconductor device connected to at least one phase shifter whose phase shift is adjustable and whose impedance depends at least in part on the frequency of the input signal.

  The invention further relates to circuits and devices comprising at least such an amplifier circuit, for example gyrator devices and filter circuits.

The amplifier can be realized, for example, by an integrator and a capacitor used in an integrated IF filter device. For example, “Analog circuit design, low-power low-voltage, integrated filters, smart power” by Rudy J. van de Plassche, Willy MC Sansen, Johan H. Hujising ) ", 1994, ISBN 0-7923-9513-1, it is known to implement a transconductance amplifier device and capacitor in order to obtain the desired transfer characteristics of the filter device. The transconductance amplification device can be used, for example, to achieve the termination impedance of the filter device. In addition, the two transconductance amplification devices can be used as gyrators, ie electronic devices with an input impedance proportional to the inverse of the load impedance. Therefore, when the output of the gyrator is connected to a capacitor, that is, when the load impedance is a capacitance, the input impedance behaves like an inductance. For this reason, such gyrator-capacitor circuits have inductor characteristics and have frequency dependent impedances. The use of this feature is particularly useful in integrated circuits where physical inductors are difficult and expensive to implement.
"Analog circuit design, low-power low-voltage, integrated filters, smart power" by Rudy J. van de Plassche, Willy MC Sansen, Johan H. Hujising 1994, ISBN 0-7923-9513-1

  However, the problem is that the transconductance amplification device exhibits frequency dependent transfer characteristics. This can lead to deviations from the ideal characteristics of gyrator and filter devices in which such transconductance amplifier devices are used. Also, in a filter device, the frequency dependence of the performance of the amplifying device can affect the attenuation of frequencies outside the desired frequency passband of the filter and even make the filter system unstable.

  Moreover, in the polyphase filter, the image removal characteristic of the polyphase IF filter is finally determined by the matching characteristics between circuits in the branches of different phases. This matching may be obtained by using transistors that occupy as much area as possible in the form of complementary metal oxide semiconductors (CMOS). Since the transconductance cut-off frequency is inversely proportional to the length of the square transistor, the frequency dependence of the transconductance amplification of the transistor depends greatly on their length. Thus, between the image rejection characteristics (set by the matching characteristics) and the passband slope and / or stability (determined by the phase shift exceeding the transconductance stage) of the gyrator device or filter device. A compromise is required.

  Similar problems exist for transconductance amplification devices in the bipolar form. In bipolar transistors, the non-zero resistance of the base causes unnecessary frequency dependence of the amplifier. The presence of the base resistance leads to a frequency dependent behavior between the external base-emitter voltage and the internal base-emitter voltage, and the transistor gain is proportional. As long as matching is relevant for bipolar transistors, the matching characteristics for large transistor regions will be good. A time constant equal to the product of the base resistance and the expected value of the input capacitance is generally independent of scaling, but if the transistor is scaled larger, the base-emitter connection for a given collector current The effect of capacity increases. As a result, this results in a larger excess phase shift in the transconductance gain for a given frequency. Therefore, for bipolar devices, a compromise must be found between high frequency behavior and matching characteristics.

  The present invention seeks to solve or at least mitigate the above-mentioned problems. SUMMARY OF THE INVENTION Accordingly, an object of the present invention is to provide an amplifier circuit with improved frequency-independent transfer characteristics. Thus, according to the present invention, an amplification device as described above is adjusted so that, in use, the adjustable phase shift is the opposite of the amount of phase shift of the transconductor device. It is characterized by.

  With an adjustable phase shift, the integrator phase shift can be set to compensate for the frequency shift of the transconductor device. This allows the amplification of the amplifier to be substantially frequency independent and the filter passband slope to be compensated.

  In addition, compensating for the adjustable phase shift can also eliminate stability problems because it prevents the shift of the filter pole to the right half plane (RHP) caused by excessive phase shift. Also, there is almost no trade-off between image removal requirements and filter stability requirements because larger size transistors can be used for good matching characteristics without compromising filter stability. Furthermore, if an automatic tuning system (ATS) is used, the time constant tuning system may be combined with a frequency tuning system of filter devices that are already available. Since the power consumption required for this compensation is very small, good performance can be obtained with the same level of power consumption.

FIG. 3 is a block diagram illustrating an example embodiment of an amplifier circuit according to the present invention in which an amplifier is used as an adjustable resistor. , , , , FIG. 3 is a block diagram illustrating an example of a capacitor-resistance circuit in which a transistor is used as an adjustable resistor. FIG. 6 is an electrical circuit diagram illustrating an example embodiment of a gyrator device according to the present invention. FIG. 3 is an electric circuit diagram showing an example of an embodiment of a polyphase filter according to the present invention. FIG. 3 is a block diagram of a phase-locked loop with an example of an automatic tuning system for an amplifier circuit according to the present invention.

  Further details and embodiments of the present invention will be described below with reference to the accompanying drawings.

  What is expressed here as a resistor and a capacitor includes any device or element that has at least primarily resistive or capacitive characteristics. FIG. 1 shows an example electrical circuit of an embodiment of an amplifier circuit A1 according to the invention. The amplifier circuit comprises a transconductor device 1 having two input contacts Uin and two output contacts Uout. Generally, a transconductor device outputs a current to an output contact based on the voltage of an input signal. The output contact Uout of the transconductor device 1 is connected to the phase shifter unit 2. The phase shifter device 2 is connected in parallel to the output contact Uout of the transconductor device 1 and includes an adjustable resistor R and a capacitor C connected in series.

The adjustable resistance R is an amplification device R that can be adjusted to a constant resistance value. The amplifying device R has one input contact Rin and two output contacts Rout1 and Rout2. The output contacts, I _1, the current I _1, I ₂ in the direction indicated by arrow I ₂ are supplied. Usually, the currents I ₁ and I ₂ are equal. By applying a constant input signal to the input contact Rin, the output signals at the output contacts Rout1 and Rout2 are controlled, and as a result, the impedance between the output contacts becomes constant. Thus, the impedance can be changed by changing the input signal.

The transconductor device 1 outputs an output current based on the voltage difference between the input contacts. The output current is the formula:
I _u = g _m · U _i (1)
Can be approximated by

In this equation, I _u represents the output current in the direction indicated by the arrow I _u in FIG. 1, U _i represents the voltage applied on the two input contacts Uin, and g _m represents the transconductor The gain of device 1 is represented.

で表されるであろう。 The gain of the transconductance amplifier generally depends on the frequency of the signal appearing at the input contact and is expressed as g _m = g ₀ / (1 + jwτ), where g ₀ is the transconductor for a signal with a frequency of substantially 0 Hz. Represents the transconductance of the device, w represents the frequency of the input signal multiplied by 2π, j is the square root of −1, and τ is the time constant of the transconductor. The voltage transition of the amplifier circuit shown in FIG.

It will be represented by

In Equation (2), U _i and U _o represent potential differences between the input contacts and the output contacts of the transconductor device, R represents the resistance value of the amplification device R, and C represents the capacitance of the capacitor C in FIG. Represents.

で表されるであろう。 The phase shifter device 2 has a phase shift Δφ between the voltage on the first shifter contact 21 and the second shifter contact 22 and the current flowing through these shifter contacts. The relationship between the voltage U _o and the current I _u on the shifter contacts 21, 22 is mathematically:

It will be represented by

In this equation, C represents the capacitance of capacitance C, j is the square root of −1, w represents frequency, I _u is defined by the direction indicated by arrow I _u in FIG. Represents the resistance value.

  As explained, the resistance value of the amplifying device on the output contacts Rout1, Rout2 can be controlled by a signal applied to the input contact Rin. Thereby, the phase shift of the phase shifter unit 2 can also be adjusted. The capacitor C may also be adjustable, for example a varactor device. With an adjustable resistance value and / or capacitance, the phase shift can be set to compensate for the gain phase shift and / or frequency dependence of the transconductor device 1.

で表されるであろう。 Accordingly, the time constant of the phase shifter device, that is, the product of the resistance value and the capacitance can be easily adjusted so that the time constant R · C is substantially equal to the time constant τ of the transconductor device 1. . As a result, the total transfer characteristic of the amplifier circuit A1 is given by the formula:

It will be represented by

  This is the ideal integrator behavior.

  The input of the amplifier may be connected to a control device in which the characteristics of the transconductor device are modeled. The control device then provides a signal to the amplification device so that the amplifier compensates for changes in transconductance, as described above.

  The amplification device may be substantially similar or equivalent to the transconductor device. Thereby, the characteristics of the amplifying device and the transconductor device will have substantially the same dependence. Thus, no external means are required to compensate for changes due to temperature changes, for example.

  The amplifying device in the resistor may be a transistor device such as a field effect transistor (FET), for example. In this application, the term “amplifier” should be understood to include all transistors. If the adjustable resistor is a transistor device, the amplifier circuit may be implemented as a single integrated circuit. This is particularly suitable when the transistor is of the same type as other transistors in the integrated circuit, such as transistors in transconductor devices. In this case, the processing of the integrated circuit is hardly complicated because the devices in the integrated circuit require substantially the same processing.

  2 to 6 show an example of a capacitor C connected to a transistor R used as an adjustable resistance device. In these figures, field effect transistors (FETs) are used as adjustable resistors, but other types of transistors may be used. Resistor contacts Rout1, Rout2 are the drain and source of the FET. The resistance value of the FET may be adjusted by changing the voltage applied to the gates Rin, Rin1, Rin2 of the FET with respect to the substrate of the device. As a result, the conduction channel in the FET between the source and the drain is enlarged or reduced, and as a result, the resistance value of the FET between the source and the drain is lowered or increased. Since the drain-source connection is in series with the capacitor, no direct current flows through the resistor and power consumption is substantially zero or at least very low. In FIGS. 2-6, the voltage across the source, drain and gate of the FET is set so that the transistor operates in the triode region.

  In FIGS. 2 and 6, an n-type FET is used as a resistor, and in FIGS. 3 and 4, a p-type FET is used. In FIG. 5, a complementary FET is used. The complementary FET includes a p-type FET RF1 and an n-type FET RF2 connected non-parallel to each other. In FIG. 5, the source and drain of each FET are connected to a capacitor having a capacitance 2C. 2 to 6, the FET may be replaced with another amplifying device, for example. The resistance value of the amplification device can be adjusted by changing the voltage or current applied to the input of the amplification device.

FIG. 7 shows a gyrator device 10 including an amplifier circuit according to the present invention. The gyrator device 10 has an input U _in and an output U _out . The gyrator includes two transconductance amplifiers 101 and 102 whose front ends and rear ends are connected to each other. The gain direction of the amplifier 102 is opposite to the gain of the amplifier 101. That is, as shown in FIG. 7, when an input signal is input to the input terminal U _i , the amplifier 102 outputs a current indicated by an arrow I _u in a direction opposite to the direction of the output current of the amplifier 101.

  In the gyrator, the amplifiers 101 and 102 are connected to a phase shifter device including a capacitor C1 and a resistor R1 connected in series. The resistor R1 is an adjustable resistor, as described above with reference to FIGS. 1 to 6, and may be, for example, an FET or amplifier connected to a capacitor.

When the resistance value of the resistor R1 is set so that the time constant R1 · C1 is substantially equal to the time constant τ of the two combined transconductors 101-102, the impedance of the gyrator is the impedance of the ideal inductor. Equal to the value multiplied by the gain of the amplifier. The gains of the two amplifiers may be opposite to each other. In this case, amplifier 102 outputs a current having a substantially equal magnitude that is greater than or equal to the magnitude of the output current of amplifier 101, where the gyrator has the same impedance as the ideal inductor. Connected to the input contact U _i of the gyrator 102 is an RC circuit including a resistor R2 and a capacitor C2 connected in series. Thus, the impedance of the circuit of FIG. 7 viewed from the input contact U _i is substantially equal to the impedance of the ideal inductor and ideal capacitor in parallel with each other.

  FIG. 8 shows a polyphase IF filter. Currently, it is becoming common to implement a channel-selective (low) IF filter or IF anti-aliasing filter on a chip. In practice, selecting a non-zero IF frequency with a quadrature mixer and polyphase IF filter essentially suppresses the image frequency, compared to a conventional single mixer and IF filter combination, Bring great benefits. This form of complex frequency processing avoids the need for a complex image removal filter at the RF frequency before the mixer. The level of image frequency suppression ultimately depends on the accuracy of the quadrature transmitter and the matching between the in-phase and quadrature IF branches. It should be noted that the filter of FIG. 8 is an example of a polyphase filter, and the present invention can be applied to other types of polyface filters.

The filter device of FIG. 8 has two branches i and q having different phases. In the illustrated example, the phase difference is assumed to be 90 degrees. The illustrated filter device includes two in-phase input contacts I _{in_i} supplied with one signal, and in this example, two phase-shifted input contacts I _{in_q} supplied with the same signal with a phase shifted by ± 90 degrees. have. An output signal appears at the in-phase output contact _{Uout_i} . The same output signal whose phase is shifted by ± 90 degrees appears at the phase shift output contact _{Uout_q} . Corresponding contacts are connected to each other via resistors R5 to R8, respectively. The two in-phase input contacts I in — _i are connected to the in-phase output contact U out — _i via the gyrator device 12 and the two phase shifter devices F1, F2. The gyrator device 12 may be similar to the gyrator device of FIG. Both the input contact and the output contact of the gyrator device 12 are connected in parallel with phase shifter devices F1 and F2, respectively, including capacitors and resistors connected in series. The phase shift input contact I _{in_q} and the phase shift output contact U _{out_q} are connected to the gyrator device 14 and the phase shifters F3 and F4 in a similar manner. In-phase branch i and the phase shift branches q is input contacts _I in_i, I in_q and output contacts U _{OUT_I,} through a gyrator 11 and 13 respectively connected to the U _{OUT_Q,} are connected to each other. Due to the phase compensation of the adjustable phase shifters F1-F4, the characteristics of the filter are improved because the gyrator device with phase shifter behaves closer to the ideal inductor.

  Since the only necessary components are transistors, resistors and capacitors, the illustrated filter device is particularly suitable to be implemented in a single integrated circuit. In addition, because the inductor is simulated using transistors and capacitors, the integrated circuit is relatively small and consumes less power. Therefore, this circuit is particularly suitable for applications where power is limited, such as mobile phones and Bluetooth devices.

  In practice, the frequency dependence of the gain of the transconductor device is not known accurately enough, for example, because the behavior cannot be modeled well by electronic circuit simulators. Also, variations in device characteristics after processing electronic circuits are usually large, so that the behavior of a particular device is difficult to predict. In the circuit of FIG. 9, an automatic tuning system (ATS) is shown to match the time constant of the adjustable resistor-capacitor circuit to the time constant τ of the transconductor device, so that the resistor-capacitor circuit and the transconductor An accurate correspondence between the time constants of each device can be obtained. Elements constituting the ATS are indicated by broken lines. It should be noted that the ATS shown is merely an example of a control system, and other control systems may be used as well to control the resistance value of the adjustable resistor of the amplifier circuit according to the present invention.

The ATS of FIG. 9 is partially integrated into a phase locked loop (PLL) 200, as is well known in the art. The PLL 200 has a PLL input 201 and PLL outputs 202 and 203. The PLL 200 includes a phase detector 204, a low-pass filter 205, a voltage controlled oscillator (VCO) 206, a limiter device 208, and a frequency divider 207. The PLL input 201 will be supplied with an input signal of a reference frequency (f _ref ). In this case, the PLL outputs a VCO signal having an output frequency (f _out ) to the PLL outputs 202 and 203. The VCO signal is supplied to the PLL outputs 202 and 203, but there is a phase difference between the two outputs. The VCO signal is generated by the VCO 206 based on the voltage of the VCO input signal. When the PLL 200 is locked, the output frequency f _out is a value obtained by multiplying the reference frequency f _ref by the division factor N:
f _out = f _ref · N
It becomes.

The frequency f _out of the VCO output signal is divided by the frequency divider 207 at the division ratio N. The frequency f _{div of} the signal divided by this is:
f _div = f _out / N
Is equal to

The divided signal of frequency f _div is compared with the input signal of reference frequency f _ref by phase detector 204. The phase detector 204 outputs a differential signal based on the phase difference between the divided frequency f _div and the reference frequency f _ref . This differential signal is low-pass filtered by the filter 205 and used as a VCO input signal for controlling the oscillation of the VCO 206.

  The ATS of FIG. 9 includes a PLL voltage controlled oscillator device (VCO) 206. In general, a VCO generates a constant frequency signal that depends on the voltage applied to the input of the VCO. In FIG. 9, the VCO is a copy of the gyrator device as used in the filter, and may be, for example, a gyrator device as shown in FIG. The amplifier circuit used in the VCO gyrator may be similar to the transconductor in the amplifier circuit to be controlled by the ATS.

  In the example of FIG. 9, the output of the filter 205 is connected to a transconductor device (not shown) in the amplification device 8 according to the invention. Thus, the output of filter 205 controls the transconductance gain of the transconductor device. The output of the filter 205 also controls the oscillation frequency of the VCO, for example, by controlling the gain of the transconductor device in the gyrator of the VCO that includes the gyrator circuit according to the present invention. The output of the VCO 206 is supplied to a phase compensation processing (PC) device 71 that provides an output signal that can be used to control the resistance value of the adjustable resistor, for example, by controlling the gain of the amplifier in the adjustable resistor. It is connected. In the illustrated example, the output signal of the PC device 71 is also fed back to the VCO 206 to control the gain of the adjustable resistor amplifier in the gyrator operating as a VCO.

  Instead of the amplifier circuit 8, a gyrator device or polyphase filter according to the invention may be controlled by the ATS. Moreover, (part of) ATS may be combined with (part of) the tuning loop of the filter, usually as already present in the filter device. As a result, additionally required power consumption is minimized. The ATS may be a stand-alone that is not partially incorporated into the PLL.

  The amplifier circuit according to the present invention or the IF filter according to the present invention can be suitably used in an electronic device for receiving radio waves with a limited power source. These electronic devices may be, for example, wireless communication devices powered by batteries, mobile phones or devices that communicate via the Bluetooth protocol, such as local devices via a Bluetooth link. There are laptop computers that communicate with area networks, and personal digital assistants (PDAs) that communicate with computers via a Bluetooth link.

Claims (6)

A first transconductor device having a first input for inputting a signal and a first output for outputting a current amplified from the signal;
  A second transconductor device having a second input connected to the first output for inputting the amplified current and a second output connected to the first input for outputting a current obtained by further converting the current; ,
  A first phase shifter connected between two terminals constituting the first output;
  A second phase shifter connected between two terminals constituting the second output;
With
  The first phase shifter unit includes a first capacitor device and a first variable resistance device in a series configuration, and the second phase shifter unit includes a second capacitor device and a second variable resistance device in a series configuration,
  The first phase shifter unit and the second phase shifter unit have a phase shift with respect to a resonance frequency of the gyrator of the current output from the first output with respect to a signal input to the first input, In order to make the time constant of the output load of the first output equal to the product of the capacitance value of the first capacitor device and the resistance value of the first variable resistance device, the first variable resistance of the first phase shifter unit A gyrator circuit that adjusts both resistance values of a device and a second variable resistance device of the second phase shifter unit. The gyrator circuit according to claim 1, wherein the variable resistance device is a transistor device. The gyrator circuit according to claim 2, wherein the transistor device is a field effect transistor. A method of constructing a gyrator including a first transconductor device and a second transconductor device, comprising:
  Providing the first transconductor device with an input signal having a resonance frequency of a gyrator to the first input, and outputting a current amplified with respect to the input signal from the first output;
  Providing the amplified current to a second input of the second transconductor device to output a further converted current of the amplified current to a second output of the second transconductor device;
  Connecting a first phase shifter section between two terminals constituting the first output of the first transconductor device;
  Connecting a second phase shifter portion between two terminals constituting the second output of the second transconductor device;
Including
  The first phase shifter unit includes a first capacitor device and a first variable resistance device in a series configuration, and the second phase shifter unit includes a second capacitor device and a second variable resistance device in a series configuration,
  The first phase shifter unit and the second phase shifter unit have a phase shift with respect to a resonance frequency of the gyrator of the current output from the first output with respect to a signal input to the first input, In order to make the time constant of the output load of the first output equal to the product of the capacitance value of the first capacitor device and the resistance value of the first variable resistance device, the first variable resistance of the first phase shifter unit Adjusting both resistance values of the device and the second variable resistance device of the second phase shifter unit. The method of claim 4, wherein the variable resistance device is a transistor device. The method of claim 5, wherein the transistor device is a field effect transistor.

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