JP4766072B2 - Communication device - Google Patents

Description

  The present invention relates to a communication system in which a plurality of wireless stations communicate with each other, and more particularly, in a wireless communication system employing an OFDM (Orthogonal Frequency Division Multiplexing) modulation scheme such as IEEE802.11a or HIPERLAN. The present invention relates to a communication device that transmits and receives.

  By connecting multiple computers to form a LAN (Local Area Network), information such as files and data can be shared, peripheral devices such as printers can be shared, e-mails, data and content can be transferred, etc. Exchange information.

  Recently, wireless LAN has attracted attention. According to this type of wireless LAN, since most of the wired cables can be omitted in a work space such as an office, a communication terminal such as a personal computer (PC) can be moved relatively easily. In addition, the demand for wireless LAN systems has increased significantly as the speed and price of wireless LAN systems have decreased. In particular, introduction of a personal area network (PAN) is being studied in order to establish a small-scale wireless network between a plurality of electronic devices existing around a person and perform information communication.

  As wireless LAN standards, for example, IEEE802.11b and IEEE802.11a are widely known in the industry (for example, see Non-Patent Document 1). In the IEEE802.11b standard, wireless communication up to a maximum of 11 Mbps is possible using the CCK (Complementary Code Keying) method in the 2.4 GHz band. In the IEEE802.11a standard, the OFDM scheme is used in the 5 GHz band, and a maximum transmission speed of 54 Mbps is enabled. Here, OFDM (Orthogonal Frequency Division Multiplexing) is a type of multi-carrier transmission method in which the frequency of each carrier is set so that each carrier is orthogonal to each other within a symbol interval. Has been.

  Another standard that is becoming increasingly important in the wireless LAN field is HIPERLAN (High Performance Radio Local Area Network: high performance wireless local area network).

  HiperLAN provides network access via fixed base stations and access points, and supports specialized computing for multimedia systems that do not require specific standards, licenses, or unified infrastructure. In addition, it has sufficient performance to support the latest real-time digital audio / video standards from 6 Mbps to 54 Mbps including MPEG (Moving Picture Experts Group).

  Among them, HIPERLAN2 (hereinafter referred to as “H / 2”) is a wireless LAN system that performs transmission by OFDM modulation in the 5 GHz band (see, for example, Patent Document 2), and the physical layer parameters are IEEE802. There are many similarities to .11a.

IEEE Std 802.11a Part 11: Wireless LAN Medium Access Control (MAC) and Physical Layer (PHY) specialties: high-speed Physical Layer the Hz ETSI TS 101 475, Broadband Radio Access Networks (BRAN); HIPERLAN Type 2; Physical (PHY) Layer

  An object of the present invention is to provide an excellent communication apparatus that employs an OFDM (Orthogonal Frequency Division Multiplexing) modulation scheme such as IEEE802.11a or HIPERLAN.

The present invention has been made in consideration of the above-mentioned problems. The first aspect of the present invention is a communication apparatus that performs transmission / reception in the OFDM modulation system and operates under a communication system in which a plurality of transmission rate modes are defined. And
Amplifying means for amplifying the transmission signal or attenuation means for attenuating the transmission signal;
A switching means for switching the input to the transmission amplifier according to the transmission rate mode to a transmission signal, a signal obtained by amplifying the transmission signal by the amplifying means, a signal attenuated by the attenuation means, or a signal;
It is a communication apparatus characterized by comprising.

  A vector error may occur due to saturation of the amplitude value of the output signal. In the case of OFDM, since the amplitude dynamic range is large, the use of a non-linear region leads to characteristic degradation. For this reason, in order not to cause a transmission symbol error, usually, the input signal amplitude is given a margin and the signal amplitude is performed in a section where the linearity of the amplifier is ensured, that is, the backoff is not performed. Taking is done. However, as the margin is increased, the power supply efficiency tends to deteriorate, and a problem arises from the viewpoint of power saving.

  Therefore, in the first aspect of the present invention, a switch is provided in front of the transmission power amplifier, and the input signal to the power amplifier is changed in accordance with the transmission rate mode, so that the necessary amount for each transmission rate mode is obtained. It is possible to provide power supply efficiency with only back-off. As a result, power saving can be achieved, and transmission power can be increased in the low transmission rate mode.

A second aspect of the present invention is a communication device that operates under a communication system that performs transmission and reception of signals in bursts,
An AD converter for converting a received signal into a digital signal;
The absolute value amplitude of the received signal output from the AD converter is evaluated, and when the absolute value amplitude is not close to the maximum value of the AD converter, the received signal input from the AD converter is output as it is, Received signal evaluation means for outputting a predetermined amplitude value several times the absolute value amplitude maximum value of the AD converter when the absolute value amplitude is very close to the maximum value of the AD converter;
Received power calculating means for calculating received power based on the evaluation result by the received signal evaluating means;
And a gain control of the AGC amplifier based on the received power.

  AGC is used to absorb the dynamic range of received power. In H / 2 and IEEE802.11a, since signals are received in bursts, it is necessary to determine the degree of amplification of the AGC amplifier for each burst, and this processing is performed in a short time after receiving the signals. It is necessary to adjust to the signal range of the converter.

  According to the second aspect of the present invention, a selector is provided before the reception power value is calculated from the reception amplitude, and if the reception amplitude is the maximum value of the absolute value amplitude of the AD converter, the amplitude is several times that By calculating the power value based on the value, it is possible to set the reception signal level in the AGC amplifier at a higher speed when a signal is rapidly received with a high reception power. In addition, by configuring the loop filter in decibel notation, it is possible to increase the amplification level sharply even when the received power is rapidly decreased.

The third aspect of the present invention is a communication device that operates under a communication system that transmits and receives signals in bursts, and a preamble is added to the head of the burst signal,
A multi-stage autocorrelation calculator;
A normalization unit that normalizes the correlation value calculated in each stage of the autocorrelation calculation unit with the received power value;
An autocorrelation measurement unit that measures the autocorrelation of the preamble by adding normalized correlation values over a plurality of stages;
A packet recognition unit for recognizing the presence of a received packet based on the measured autocorrelation of the preamble part;
It is a communication apparatus characterized by comprising.

  According to the third aspect of the present invention, the presence of the received packet is recognized by measuring the autocorrelation of the preamble part, but the autocorrelation calculation block is configured in a plurality of stages and is calculated in each stage. The autocorrelation value independent of the received power is calculated by normalizing the correlation value with the received power value and adding the normalized correlation values calculated in a number of stages to measure the autocorrelation of the preamble. can do. In addition, it is possible to prevent the packet reception time from being mistaken due to a sharp increase in reception power. Further, by configuring the peak estimation process in the same manner, it is possible to extract the maximum value of the autocorrelation value.

  Here, in order to search for the maximum value of the autocorrelation value of the preamble part, after confirming that the correlation value exceeding the predetermined threshold is not updated as the maximum value for a certain period of time, the signal reception detection and the reception time report are performed. You may make it perform.

The fourth aspect of the present invention is a communication device that operates under a communication system that transmits and receives signals in bursts, and a preamble is added to the head of the burst signal,
When a cross-correlation value with a known signal sequence is calculated for a certain time window, and a value that is about 60% to 80% of the maximum value is set as a threshold, and a correlation value exceeding the threshold exists before the maximum value appears. Is extracted as the reception timing the earliest time when the correlation value exceeding the threshold appears.
It is a communication apparatus characterized by this.

  According to the fourth aspect of the present invention, the head position of the received packet can be extracted more accurately even in an environment where delay dispersion exists, and the influence of intersymbol interference can be minimized. . In addition, by correcting the timing detection input signal with the provisional frequency offset value obtained by packet detection, the timing detection process is suitably performed even when a signal having a large frequency offset is received. In addition, since the provisional frequency offset value is used for correction, there is almost no need to consider processing delay.

  Here, the detailed reception timing may be detected by calculating a cross-correlation value between the received signal corrected by the provisional frequency offset value already calculated and the known signal sequence.

A fifth aspect of the present invention is a communication apparatus that operates under a communication system in which signals are transmitted and received in bursts using the OFDM modulation method, and the burst signal is filled with power for each of a plurality of subcarriers. And a preamble in which all subcarriers are filled with power,
Means for calculating a rough frequency offset value as macro information by a preamble filled with power for each of a plurality of subcarriers;
Means for calculating a detailed frequency offset value as micro information by a preamble in which power is charged in all subcarriers;
By correcting the frequency offset value obtained from the micro information based on the value obtained from the macro information and the amplitude value obtained from the micro information, a final detected value of the frequency offset value is obtained. Means for extracting,
Means for driving the oscillator based on the detected value of the final frequency offset value, and correcting the frequency offset by multiplying the output of the oscillator and the received signal once;
It is a communication apparatus characterized by this.

  According to the fifth aspect of the present invention, the frequency offset can be corrected by multiplying the output of the oscillator and the received signal once. In the methods known in the past, it is necessary to perform a primary correction using a provisional frequency offset value and a secondary correction using a more detailed frequency offset value. Multiplication with oscillator output) is reduced by half. Further, by configuring the Arctan circuit, it is possible to obtain an Arctan value that does not depend on the amplitude value of the received signal with a simple configuration.

A sixth aspect of the present invention is a communication device that operates under a communication system that transmits and receives signals in bursts,
AGC by analog amplifier that locks the degree of amplification after a specified time from the start of signal reception;
In the digital domain, a digital gain control unit that calculates and normalizes the power of the received signal;
It is a communication apparatus characterized by comprising.

  According to the sixth aspect of the present invention, by adding a digital gain control process, the fluctuation of the received signal amplitude that cannot be corrected by the AGC by the analog amplifier is absorbed, and the word length in the subsequent processing block is absorbed. It is possible to limit the characteristics without sacrificing the characteristics.

According to a seventh aspect of the present invention, there is provided a communication apparatus that operates under a communication system in which signals are transmitted and received in bursts using the OFDM modulation method. A preamble having the same waveform is added to the beginning of a burst signal over a plurality of symbols. Has been
A plurality of preamble symbols are added in the time axis signal state after correcting the frequency offset, and the signal after the addition is fed to the FFT,
It is a communication apparatus characterized by this.

  According to the seventh aspect of the present invention, it is only possible to increase the accuracy of the channel vector by adding the Pre-3 part and Pre-4 part of the preamble to the time axis signal and outputting them. Instead, it is possible to reduce the number of FFT executions.

Further, an eighth aspect of the present invention is a communication device that operates under a communication system in which signals are transmitted and received in bursts using the OFDM modulation method, and a preamble is added to the head of the burst signal,
A channel vector is generated by weighted addition of adjacent subcarriers of the preamble,
It is a communication apparatus characterized by this.

  According to the eighth aspect of the present invention, two systems of channel vectors with weighted addition of adjacent subcarriers and channel vectors without weighted addition of adjacent subcarriers are generated, and whichever is selected according to the transmission rate mode. Therefore, it is possible not only to improve the accuracy of the channel vector but also to both H / 2 and IEEE802.11a. It is possible.

A ninth aspect of the present invention is a communication device that operates under a communication system in which signals are transmitted and received in bursts using the OFDM modulation method. A preamble is added to the head of the burst signal, and the information signal A known pilot subcarrier is inserted into some subcarriers,
Means for adding pilot subcarriers after transmission path estimation as long as they are inserted;
Means for estimating a frequency offset value from the signal points of the added pilot subcarriers;
Means for updating the channel vector to follow the frequency offset by a loop filter based on the estimated frequency offset value;
A communication device comprising:

  Here, in order to follow the frequency offset, a plurality of loop filters having different loop gains are provided, and it is determined whether to use the calculation result of any of the above loop filters for correction according to the number of times the symbol is received. You may make it do.

A tenth aspect of the present invention is a communication device that operates under a communication system in which signals are transmitted and received in bursts using the OFDM modulation method.
A channel vector estimate is calculated by tentatively determining the received subcarriers of the information symbol, and the reception timing drift is estimated from the difference between the channel vector stored inside and the channel vector estimate, and a loop filter To correct,
It is a communication apparatus characterized by this.

  According to the tenth aspect of the present invention, a channel vector estimated value is calculated by provisional determination of received subcarriers of information symbols, and received from a difference between the channel vector stored therein and the channel vector estimated value. Since the timing drift is estimated and the residual frequency offset and the reception timing drift are corrected by the loop filter, the correction process can be performed with a simple configuration and high accuracy.

  Here, the subcarriers used for estimating the reception timing drift may be applied only to subcarriers far from the center frequency.

An eleventh aspect of the present invention is a communication device that operates under a communication system in which signals are transmitted and received in bursts using the OFDM modulation method,
Each time an information symbol is received, the channel vector is estimated from the signal point of the received symbol, and the channel vector stored in the receiving device is updated.
It is a communication apparatus characterized by this.

  According to the eleventh aspect of the present invention, every time an information symbol (subcarrier) is received, the channel vector is estimated from the signal point of the received symbol, and the channel vector stored in the receiving apparatus is updated. Therefore, it is possible to suitably perform the reception process even in an environment where the reference phase and the reference amplitude of the received signal fluctuate due to fading fluctuations in the signal reception process. Further, it can be realized with a simple circuit configuration.

  According to the present invention, a switch is provided in front of the transmission power amplifier, and the input signal to the power amplifier is changed according to the transmission rate mode, so that the back-off corresponding to each transmission rate mode can be achieved. Power supply efficiency can be provided. As a result, power saving can be achieved, and transmission power can be increased in the low transmission rate mode.

  According to the present invention, a selector is provided before the reception power value is calculated from the reception amplitude, and when the reception amplitude is the maximum value of the absolute value amplitude of the A / D converter, the amplitude value is several times that value. By calculating the power value based on the above, it is possible to set the reception signal level in the AGC amplifier at a higher speed when a signal is rapidly received with a high reception power. In addition, by configuring the loop filter in decibel notation, it is possible to increase the amplification level sharply even when the received power is rapidly decreased.

  Further, according to the present invention, the presence of the received packet is recognized by measuring the autocorrelation of the preamble part, but the autocorrelation calculation block is configured in a plurality of stages, and the correlation calculated in each stage To calculate the autocorrelation value independent of the received power by normalizing the value with the received power value and adding the normalized correlation value calculated at each stage over multiple stages to measure the autocorrelation of the preamble Can do. In addition, it is possible to prevent the packet reception time from being mistaken due to a sharp increase in reception power. Further, by configuring the Peak Examination process in the same manner, it is possible to extract the maximum value of the autocorrelation value.

  Further, according to the present invention, the detailed reception timing detection is obtained from the known signal sequence and the cross-correlation value. The cross-correlation value is calculated for a certain time window, and the maximum value of 60 is obtained. If the correlation value exceeding this threshold exists before the maximum value appears, the earliest time at which the correlation value exceeding the threshold appears is extracted as the reception timing. Even in an environment where dispersion exists, the head position of a received packet can be extracted more accurately, and the influence of intersymbol interference can be minimized. In addition, by correcting the timing detection input signal with the provisional frequency offset value obtained by packet detection, the timing detection process is suitably performed even when a signal having a large frequency offset is received. In addition, since the provisional frequency offset value is used for correction, there is almost no need to consider processing delay.

  In addition, according to the present invention, a rough frequency offset value is calculated as macro information by a preamble in which power is filled for each of a plurality of subcarriers, and a detailed frequency offset is determined by a preamble in which power is filled in all subcarriers. By calculating the value as micro information and correcting the frequency offset value obtained with micro information based on the value obtained with macro information and the amplitude value obtained with micro information, The detected frequency offset value is extracted, and the oscillator is driven based on the final detected frequency offset value obtained by such a procedure, and the output of the oscillator and the received signal are multiplied once. By doing so, the frequency offset can be corrected. In the methods known in the past, it is necessary to perform a primary correction using a provisional frequency offset value and a secondary correction using a more detailed frequency offset value. Multiplication with oscillator output) is reduced by half. Further, by configuring the Arctan circuit, it is possible to obtain an Arctan value that does not depend on the amplitude value of the received signal with a simple configuration.

  In addition, according to the present invention, by adding a digital gain control process, the fluctuation of the received signal amplitude that cannot be corrected by the AGC by the analog amplifier is absorbed, and the word length in the subsequent processing block is limited. However, it is possible not to sacrifice the characteristics.

  In addition, according to the present invention, by adding the Pre-3 part and Pre-4 part of the preamble to the time axis signal and then outputting them, not only can the accuracy of the channel vector be improved, but also the FFT. It is possible to reduce the number of executions of.

  In addition, according to the present invention, two channel vectors for which weighted addition of adjacent subcarriers is added and channel vector for which weighting addition of adjacent subcarriers is not performed are generated. Since it is determined whether the transmission path estimation processing is performed using a vector, not only can the accuracy of the channel vector be improved, but it is also possible to support both H / 2 and IEEE802.11a. is there.

  In addition, according to the present invention, a channel vector estimated value is calculated by provisionally determining the reception subcarrier of the information symbol, and the reception timing drift is calculated from the difference between the channel vector stored inside and the channel vector estimated value. Since the condition is estimated and the residual frequency offset is corrected and the reception timing drift is corrected by the loop filter, correction processing can be performed with a simple configuration and high accuracy.

  Further, according to the present invention, every time an information symbol (subcarrier) is received, a channel vector is estimated from the signal point of the received symbol, and the channel vector stored in the receiving apparatus is updated. Even in an environment where the reference phase and reference amplitude of the received signal fluctuate due to fading fluctuations in the reception process, it is possible to perform the receiving process suitably. Further, it can be realized with a simple circuit configuration.

  Other objects, features, and advantages of the present invention will become apparent from more detailed description based on embodiments of the present invention described later and the accompanying drawings.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

A. The physical layer signal HIPERLAN2 (hereinafter referred to as H / 2) of HIPERLAN2 and IEEE802.11a is a wireless LAN system that performs transmission by OFDM modulation in the 5 GHz band, and the parameters of the physical layer are similar to those of IEEE802.11a. There are many (H / 2 performs multiple access using the TDMA system and IEEE 802.11a uses the CSMA / CA system).

  The OFDM scheme is a type of multicarrier transmission scheme, and the frequency of each carrier is set so that the carriers are orthogonal to each other within a symbol interval. As an example of information transmission, a plurality of data output by serial / parallel conversion of information sent serially for each symbol period slower than the information transmission rate is assigned to each carrier and modulated for each carrier. By performing inverse FFT on the carrier, it is converted into a signal on the time axis and transmitted while maintaining the orthogonality of each carrier on the frequency axis.

  FIG. 1 shows a time view of an OFDM symbol. As shown in the figure, one OFDM symbol is composed of 3.2 microseconds, and a guard time of 0.8 microseconds (or 0.4 microseconds) is added thereto to form a transmission unit of one OFDM symbol.

  FIG. 2 shows a frequency view of the OFDM symbol. As shown in the figure, the subcarrier interval is 312.5 kHz, and the baseband 0 Hz subcarrier is NULL, and transmission / reception is performed using a total of 53 subcarriers. In addition, pilot subcarriers are arranged at −21 × 312.5 kHz, −7 × 312.5 kHz, + 7 × 312.5 kHz, and + 21 × 312.5 kHz in the OFDM symbols in which information is modulated. The complex amplitude of the pilot subcarrier is a known value in transmission and reception.

  In HIPERLAN2 and IEEE802.11a, a plurality of transmission rate modes are defined, and transmission / reception of information is performed by selecting a transmission rate mode according to the SNR (signal-to-noise ratio) in the receiver. Specifically, a high transmission rate mode with a high required SNR is selected when the SNR on the receiving side is high, and a low transmission rate mode with a low required SNR is selected when the SNR on the receiving side is low.

  The table below shows combinations of transmission rate modes, modulation schemes and coding rates defined in HIPERLAN2 and IEEE802.11a.

  In either H / 2 or IEEE 802.11a, signals in the physical layer are transmitted in packet units. Each packet is transmitted / received with a signal called a preamble added to its head.

  In the case of H / 2, several preamble patterns are defined, and the preamble pattern is selected according to the attribute of the packet (burst) to be transmitted / received. On the other hand, in the case of 802.11a, a unique preamble pattern is defined regardless of the attribute of the transmitted burst.

  Hereinafter, a preamble pattern used in these systems will be described.

  FIG. 3 shows the configuration of the preamble at the time of broadcast burst in H / 2. In H / 2, this broadcast burst is sent periodically from the access point (AP) at 2 millisecond intervals. By receiving this signal, each mobile terminal (MT) performs initial synchronization and receives broadcast burst information. More specifically, the BCH / FCH / ACH information in the broadcast burst is decoded to extract the designated transmission time of the downlink burst (DL Burst) and the uplink burst (UL Burst). That is, the specification is such that all communications start when the MT receives a broadcast burst.

  The Pre-1 part and Pre-2 part in FIG. 3 are known pattern signals in which energy is filled for every four subcarriers. Further, the Pre-3 part and the Pre-4 part are known pattern signals in which 52 subcarriers are filled with energy.

  FIG. 4 shows the configuration of the preamble at the time of H / 2 downlink burst. This is a burst used for downlink information channel transmission in an H / 2 system. In this burst, only the Pre-3 part and the Pre-4 part in the broadcast burst are used as preambles.

  Further, FIG. 5 shows the configuration of the preamble at the time of H / 2 uplink burst. This is a burst used in uplink transmission applications in H / 2 systems. As shown, it is similar to a broadcast burst, but the pattern of the Pre-1 part is different. In H / 2, in addition to the downlink and uplink, what is called a direct link for direct transmission and reception of information between MTs is defined. The direct link burst preamble also uses the same pattern as the uplink burst preamble.

  FIG. 6 shows the configuration of an IEEE802.11a packet preamble. As shown, this is almost the same as the uplink burst in the H / 2 system, but the polarity of the last 0.8 microseconds of the Pre-2 part is different.

  In H / 2, as in the cellular system, the AP and the MT are placed in a master-slave relationship, and each MT connected to the AP is synchronized with the timing managed by the AP. The AP transmits a transmission unit called a “MAC frame” at intervals of 2 milliseconds.

  FIG. 7 shows the configuration of the MAC frame. As shown in the figure, it basically comprises a BC phase, a DL phase, a UL phase, and an RA phase. In H / 2, a MAC frame format for an AP in which a transmission / reception antenna is sectorized is defined, but in this specification, only the case of an omnicell AP is taken up for the sake of simplicity.

BC phase:
A broadcast burst corresponding to CCH (Common Control Channel) in the cellular manner is transmitted. The AP is used for the purpose of delivering control information to each MT. Each MAC frame is periodically transmitted at the beginning of the frame.

DL phase:
In the downlink phase, the section in which the information channel transmitted from the AP to the MT is accommodated is called in this way.
The AP autonomously determines at what timing and how much information is transmitted to a desired MT, and schedules allocation slots. A procedure in which the information in the MAC frame is transmitted at the timing in the present broadcast burst in the signal in the previous broadcast burst and the corresponding information is transmitted in the downlink phase is taken without exception. A burst transmitted from the AP to the MT in the downlink phase is called a downlink burst (DL burst).

UL phase:
In the uplink phase, the section in which the information channel transmitted from each MT to the AP is accommodated is called in this way.
By receiving the broadcast burst, the MT receives information on which timing in this MAC frame may be transmitted, and transmits an uplink signal according to this instruction. A burst transmitted from the MT to the AP in the uplink phase is called an uplink burst (UL burst).

RA phase:
Random access phase (Random Access Channel) When the MT first requests a connection to the current AP, the random access channel (Random Access Channel) in the RA phase is used to convey this fact. Use to send.

  In addition, there may be a phase called Direct Link Phase (DiL Phase) in which signals are directly transmitted and received between MTs, but the Direct Link Phase is optional in the H / 2 specification. ing.

B. Overall Configuration of Transmitter / Receiver (Modem Unit) FIGS. 8 and 9 schematically show functional configurations of the transmitter and the receiver according to the present embodiment.

  First, the overall operation of the transmitter will be described with reference to FIG.

  The modem unit performs digital signal processing, such as a channel coding unit (Channel Coding), an inverse Fourier transform unit (IFFT), a preamble generation unit (Preamble Creation), a guard time insertion unit (Guard Insertion), a digital low-pass filter (LPF). ) And subsequent blocks that perform analog signal processing, such as a DA converter, an analog low-pass filter (LPF), a quadrature modulator (IQ Modulator), and a power amplifier (Power Amplifier).

  Each block for performing digital signal processing is controlled by a time base controller (hereinafter referred to as TBC), and the time at which processing should be started, the time at which processing is to be completed, or parameters required for processing are TBC. Is input to each block in a timely manner. Although there are processing blocks (not shown) for the analog signal processing unit, they are not directly related to the present invention and will not be described here.

  Next, a procedure for transmitting a signal in the illustrated transmitter will be described.

  When a bit string to be transmitted from the upper layer of the communication protocol is input to the channel coding unit, encoding such as convolutional encoding, puncturing, and interleaving is executed in the block according to parameters input from the TBC. Is done. Further, symbol mapping for converting the interleaved encoded bit string into transmission symbols is performed, and a transmission symbol sequence expressed by two-dimensional amplitude of IQ is output.

  The transmission symbol sequence generated in this way is input to IFFT. In IFFT, inverse discrete Fourier transform is performed. Thereby, a time axis signal sequence obtained when the transmission symbol sequences are arranged on the frequency axis is generated.

  The time axis signal sequence is input to the guard time insertion unit of the next block, and a redundant time axis signal of about 25%, for example, is generated by a cyclic extension. The signal generated by the above procedure is transmitted as an information part of a transmission signal.

  In the communication system targeted by the present invention, as already explained in the section A, an information signal is transmitted after a known pattern signal called a preamble is added to the head of the transmission signal.

  This preamble signal sequence is generated by the illustrated preamble generation unit or stored in advance in a preamble storage unit such as a ROM. When transmitting a signal, the TBC first issues a command to output a preamble signal sequence, and then issues a command to output the information part of the transmission signal to the guard time insertion unit. As a result, a transmission signal as shown in the term A is generated. Further, the transmission signal is oversampled and band-limited by a digital filter as necessary.

  The transmission signal generated by such a procedure is fed to the DA converter, and the DA converter converts the digital signal into an analog signal. Further, after band limiting by an analog LPF or the like, the center frequency of the transmission signal is converted to a desired frequency channel by an orthogonal modulator. Finally, after the signal amplitude is amplified by the power amplifier, the transmission signal is sent out on the transmission line via a transmission antenna (not shown).

  Next, the overall configuration of the receiver will be described with reference to FIG.

  The modem unit performs digital signal processing, and includes a channel decoder unit (Channel Decoding), a transmission channel estimation unit (Channel Estimate), a Fourier transform unit (FFT), a synchronization processing unit (Synchronization), and a digital low-pass filter (LPF). Each block, and an analog signal processing preceding this, such as an AD converter (ADC), an analog low-pass filter (LPF), an AGC amplifier, an orthogonal demodulator (IQ demodulator), a low noise amplifier (LNA), etc. Consists of.

  Each of the above blocks for performing digital signal processing is controlled by a time base controller (hereinafter referred to as TBC), and the time at which processing should be started, the time at which processing is to be completed, or parameters required for processing, etc. Is input to each block in a timely manner. Although there are processing blocks (not shown) for the analog signal processing unit, they are not directly related to the gist of the present invention, and thus description thereof is omitted in this specification.

  Next, a procedure for receiving a signal in the illustrated receiver will be described.

  When the received signal is received via a receiving antenna (not shown) (shared with the transmitting antenna in this embodiment), linear amplification is performed by the LNA, and then the center frequency of the received signal is based on the orthogonal demodulator. Convert to band. Further, the received power is roughly normalized by the AGC amplifier, the dynamic range of the received signal power is narrowed, and the signal amplitude is adjusted so that the reception level is appropriate so as to be within the dynamic range of the AD converter. This signal is further input to an analog LPF to remove signal components outside the signal band of the received signal. The signal thus obtained is input to an AD converter and converted into a digital signal.

  If the AD converter has a sample rate more than twice the signal bandwidth, the digital LPF further limits the bandwidth. The reception signal thus obtained is fed to the synchronization processing unit. In the synchronization processing unit, if the presence of the signal is confirmed by detecting the preamble, the precise reception timing is confirmed, the frequency offset is removed, and the received signal power is normalized as necessary. The effective symbol part of the OFDM symbol And the received signal is fed to the FFT.

  In FFT, discrete Fourier transform is performed to extract a received symbol sequence arranged on the frequency axis from the received signal waveform. The received symbol sequence is input to the transmission path estimation unit for each OFDM symbol. The transmission path estimation unit generates a channel vector from the input preamble part, and extracts an interleaved encoded bit string based on the channel vector and the input information symbol. This encoded bit string is input to the channel decoder unit, and a received bit sequence is estimated by performing maximum likelihood sequence estimation processing by deinterleaving processing, Viterbi decoding, and the like.

  The received bit estimation sequence obtained by the above-described procedure is transferred to the upper layer of the communication protocol as a received bit sequence. Here, each block of the digital signal processing unit of the receiver is controlled by the TBC, and the TBC performs processing start and end times, and further processing for each block according to the reception timing transmitted from the synchronization processing unit. The necessary parameters are supplied.

C. Since the time axis waveform of the OFDM signal is Gaussian in the power amplifier on the transmission side, it is known that the PAPR (Peak-to-Average Power Ratio) is higher than that of normal complex PAM transmission.

  An input / output characteristic of an amplifier used for a normal transmission power amplifier is generally as shown in FIG. 10, for example. As shown in the figure, there is a phase characteristic corresponding to the input amplitude, and the output amplitude is saturated so as to gradually approach a certain value.

  A vector error may occur due to saturation of the amplitude value of the output signal. In the case of OFDM, since the amplitude dynamic range is large, the use of a non-linear region leads to characteristic degradation. For this reason, in order not to cause a transmission symbol error, the input signal amplitude is usually given a margin and the signal amplitude is performed in a section in which the linearity of the amplifier is ensured. In general, giving a margin is called “back off”.

  However, as the margin is increased, the power supply efficiency (= average output voltage / power supply voltage) tends to deteriorate, and a problem arises from the viewpoint of power saving.

  Thus, the problem is how much back-off the power amplifier has, but the required back-off value differs depending on the transmission rate mode described above. This is because a lower vector error is required in the high transmission rate mode. For example, when the output back-off is taken and a margin of 3 dB is taken, the relationship between the required SNR (Es / No) and the BER (bit error ratio) in each transmission rate mode is as shown in FIG.

  FIG. 11 shows a case where the thin line is amplified by a complete linear amplifier and the dark line is amplified by a non-linear amplifier in each transmission rate mode. As is apparent from the figure, although the characteristics do not deteriorate when the transmission rate is up to 18 Mbps, it can be seen that the deterioration is severe when the transmission rate is as high as 54 Mbps.

  Therefore, in this embodiment, the back-off value given to the transmission amplifier is changed according to the transmission rate mode. FIG. 12 shows the configuration of a transmitter (power amplifier portion) that can switch the back-off value given to the transmission amplifier according to the transmission rate mode.

  In the input signal line to the power amplifier, an attenuator (Attenuator) and a switch (Switch) are inserted before the power amplifier. The output of the switch is input to the power amplifier.

  An input signal (Tx.Signal) is branched into a plurality of signal lines, a line 01 inputted to a switch through an amplifier (Amplifier), a line 02 inputted as it is, a line inputted after being attenuated by an attenuator. 03 is input to each switch. In the switch, the data rate mode of the signal to be transmitted is input, and in response to this, which input signal is output is determined.

  Next, a specific operation of the illustrated transmitter will be described. For example, when transmitting at a transmission rate of 6 Mbps to 18 Mbps, the switch outputs the signal line 01, and when transmitting at a transmission rate of 24 Mbps to 36 Mbps, the switch outputs the signal line 02, and 48 Mbps to 54 Mbps. The switch outputs a signal line 03 when transmission is performed at the transmission rate. This makes it possible to set the backoff level of the transmission power amplifier in accordance with the transmission rate mode.

  It should be noted that the above-described signal lines 01/02/03 are only examples, and the effects of the present invention can be obtained even if the illustrated circuit is configured by only the signal lines 01 and 02 or 02 and 03. The same effect can be expected.

  In addition, an amplifier is inserted in the signal line 01. When the signal line 01 is not output in the switch, if the procedure of turning off the power of the amplifier is used together, the power saving effect and other aspects Is more preferable.

D. Receiver AGC
In AGC (Auto Gain Control) on the receiving side, there is an AGC amplifier used to absorb the dynamic range of received power.

  In H / 2 and IEEE802.11a, since signals are received in bursts, it is necessary to determine the amplification level of the AGC amplifier for each burst. Further, this processing needs to be adjusted to the signal range of the AD converter in a short time after receiving the signal. Specifically, the Pre-3 and Pre-4 parts of the preamble are used as channel vectors in the transmission path estimation unit (described later), so the signal amplitude normalization is completed in the Pre-1 and Pre-2 parts. There is a need to.

  When controlling the AGC amplifier in the digital domain, for example, a control loop as shown in FIG. 13 is formed. In the example shown in the figure, the AGC amplifier output is passed through an analog low-pass filter (LPF) and then digitally converted by an AD converter before gain control is performed. Then, after returning to an analog signal by the DA converter and passing through an analog low-pass filter (LPF), it is fed back to the AGC amplifier. In this case, since a filter is interposed in the loop, a delay occurs in control, and a very high speed (high frequency) loop gain cannot be applied.

  Due to this, there is a possibility that a burst is received while the received power is large, and there are some points to consider in the AGC processing.

  In the AGC amplifier shown in FIG. 13, the degree of amplification is voltage-controlled, but the voltage control signal is calculated by the digital unit and obtained by analogizing it with a DA converter.

  FIG. 14 shows the internal configuration of the gain control unit in FIG. Hereinafter, the processing content executed in the gain control unit will be described with reference to FIG.

  The gain control unit receives an I-axis amplitude and a Q-axis amplitude of a reception signal output from the AD converter, and a command signal Lock Flag that indicates whether or not to perform gain control.

  When the Lock Flag signal is on, the gain control unit stops all processing and holds the output signal for gain control output last time (the holding mechanism is not particularly shown). On the other hand, when the Lock Flag progression is off, the gain control unit calculates the amplification degree in the AGC amplifier based on the received signal amplitude.

  The input received signal, that is, the I-axis amplitude and Q-axis amplitude of the received signal output from the AD converter are first input to each selector (Sel). The selector evaluates the absolute amplitude within the gain control unit, and outputs the received signal that has been input if it is not the maximum value of the AD converter or a value very close to the maximum value. On the other hand, when the absolute value amplitude value of the input signal is the maximum value of the AD converter or a value very close to the maximum value, the Max Value stored therein is output instead of the input signal.

  This Max Value is set to a value that is about 2 to 8 times the absolute value maximum amplitude value of the AD converter (preferably a value that is about 4 times). Thereby, even when the received signal is larger than the dynamic range of the AD converter, it is possible to prevent the received signal power from being saturated.

  Each signal output from the selector is input to the square calculation circuit, the square of the I-axis amplitude and the Q-axis amplitude is taken, and the sum of these is calculated by the adder, and the received power is calculated from the received amplitude It is a mechanism to do.

  The received power value thus obtained is converted into a value expressed in decibels by a linear-to-logarithmic conversion circuit (Log ()). This conversion circuit is generally constituted by a look-up table.

  The received power value that is a value expressed in decibels is input to the first-order loop filter. When the received symbol is input at 40 MHz to the gain control unit, α times the loop gain is added at any time to a value obtained by multiplying the received power value expressed in decibels by 0.0078125 times by a 7-bit shift.

  The output of the loop filter becomes a power normalization coefficient for normalizing the received signal amplitude by converting the polarity. Then, the power normalization coefficient is converted into an amplitude value corresponding to the power normalization coefficient by a logarithm → linear conversion circuit (Linear ()), and then this value is output as a signal indicating the amplification degree in the AGC amplifier. To do. The logarithmic → linear conversion circuit is generally configured by a lookup table.

E. In the communication system based on the synchronization processing unit H / 2, the processes required for the synchronization system of the AP (that is, for uplink reception) and the synchronization system of the communication system based on IEEE802.11a are defined as follows.

(1) Acquisition of a PDU (Protocol Data Unit) train Discovers the existence of a PDU train (packet) transmitted in bursts.
(2) Detection and correction of frequency offset A frequency offset of 200 kHz or higher is detected and corrected, and transferred to a subsequent processing block.
(3) Capture of reception timing Detects an OFDM symbol break and transfers it to a subsequent processing block.

  On the other hand, the processing required for the H / 2 MT synchronization system (that is, for downlink reception) is defined as follows.

(1) Search for AP and capture of MAC frame A broadcast burst transmitted by the AP every 2 milliseconds is detected, and it is recognized that there is an AP to be connected in this carrier.
(2) Detection and correction of frequency offset A frequency offset of 200 kHz or higher is detected and corrected, and transferred to a subsequent processing block.
(3) Capture of reception timing Detects an OFDM symbol break and transfers it to a subsequent processing block.

  Hereinafter, a processing block for performing these processes is referred to as a “synchronization processing unit”, and the configuration of the synchronization processing unit will be described.

E-1. Although the processing contents differ between the synchronization processing for the control information AP and the synchronization processing for the MT from the controller to the synchronization processing section , each link can be received by a substantially similar processing. Therefore, the synchronization processing unit has decided to be able to perform either processing in accordance with a command from the controller. In addition, since the IEEE802.11a synchronization process can be executed in substantially the same manner as the H / 2 uplink, this is also supported.

  That is, the synchronization processing unit includes a mechanism that performs corresponding processing in accordance with a command given from the controller. The synchronization processing unit recognizes that a command is given from the controller to the synchronization processing unit by setting the numerical values shown in the following table.

  Hereinafter, the contents of the message indicated by each numerical value will be described.

E-1-1. OPERATION_MODE
OPERATION_MODE is an environment variable for instructing which system to perform synchronous processing on the synchronous processing unit. The numerical values to be set in OPERATION_MODE are the three types shown in the table below, and are composed of 2-bit information.

  This environment variable (OPERATION_MODE) is set after Power On Reset and is not changed during operation.

E-1-2. iRxOrder
iRxOrder is a command for instructing the synchronization processing unit to perform a specific process or stop the process. There are six commands defined in the present table as shown in the table below, which are composed of 3-bit information.

  This command (iRxOrder) serves as a trigger for causing the synchronization processing unit to perform a desired process from the controller, and upon receiving the command, the synchronization processing unit immediately starts corresponding processing. Some commands require you to set accompanying parameters. These parameter values are described below.

(1) When iRxOrder is SYNC_TBC_PARA_START_11A_SEARCH, SYNC_TBC_PARA_START_H2UL_SEARCH, or SYNC_TBC_PARA_START_H2DL_SEARCH:

(2) When iRxOrder is SYNC_TBC_PARA_SCAN:

E-2. How and at what timing each command described in each command issue timing E-1 should be set will be described with an example for each environment.

E-2-1. SYNC_TBC_PARA_BE_IDLE
The command SYNC_TBC_PARA_BE_IDLE is always issued when all actions are finished. By receiving this command, the synchronization processing unit immediately terminates all operations currently being processed and enters an idle state. The idle state is a state where no processing is performed. A specific issue example will be described later.

E-2-2. SYNC_TBC_PARA_SCAN
The command SYNC_TBC_PARA_SCAN is a command for performing carrier sense. It is assumed to be issued when checking whether there is an AP communicating with the carrier in the vicinity. After turning on the MT, this command is used in all frequency channels to Make sense. FIG. 15 shows an example of issuing this command. Upon receiving this command, the synchronization processing unit continuously performs packet detection processing (no reception processing is performed) for the time specified by the accompanying parameter iRxOrderParam1, and outputs the maximum peak value calculated by the detection processing. .

  The MT will perform processing by the command SYNC_TBC_PARA_SCAN in each frequency channel, and will attempt transmission / reception in the frequency channel in which the maximum peak value is observed. Of course, the command SYNC_TBC_PARA_SCAN can also be used for cell search purposes, and is also effective when the state of other frequency channels can be viewed quickly after starting transmission / reception of signals with a certain AP.

  After the processing by this command is completed, it is necessary to issue a command SYNC_TBC_PARA_BE_IDLE and initialize the synchronization processing unit.

E-2-3. SYNC_TBC_PARA_START_11A_SEARCH
The command SYNC_TBC_PARA_START_11A_SEARCH is issued when an IEEE802.11a packet is found and reception is attempted. In the time zone in which access is performed by CSMA, the controller must always issue this command to search for IEEE802.11a packets. A specific example of issuing this command will be described later.

E-2-4. SYNC_TBC_PARA_START_H2UL_SEARCH
The command SYNC_TBC_PARA_START_H2UL_SEARCH is issued when attempting to receive an H / 2 uplink burst. In the case of H / 2, the reception timing is known in advance, but there is also a problem of frequency offset, etc., and the preamble head part is also processed in anticipation of LongPreamble. Therefore, the processing content is almost the same as when the command SYNC_TBC_PARA_START_11A_SEARCH is issued.

  Processing when SYNC_TBC_PARA_START_11A_SEARCH and SYNC_TBC_PARA_START_H2UL_SEARCH are issued will be described with reference to FIG.

  When the command SYNC_TBC_PARA_START_11A_SEARCH is issued, the synchronization processing unit starts searching for a packet, and automatically proceeds with reception processing when it is determined that the packet has been received. Signals corrected for reception timing, frequency offset, and reception gain are successively fed to the FFT until a command SYNC_TBC_PARA_BE_IDLE is issued.

  Since the synchronization processing unit does not hold information on the length of the received packet, the controller needs to issue a command SYNC_TBC_PARA_BE_IDLE in order to end the reception process when reception of the entire packet is completed.

  In SYNC_TBC_PARA_START_11A_SEARCH and SYNC_TBC_PARA_START_H2UL_SEARCH, it is necessary to specify the accompanying parameters iRxOrderParam1 and iRxOrderParam2. The controller can set these parameters according to each situation and control the synchronization processing unit.

  Although details will be given later, it should be noted that the command SYNC_TBC_PARA_START_H2UL_SEARCH needs to be issued even in the H / 2 downlink.

E-2-5. SYNC_TBC_PARA_START_H2DL_SEARCH
The command SYNC_TBC_PARA_START_H2DL_SEARCH is issued when H / 2 broadcast burst reception is attempted. The H / 2 broadcast burst is always transmitted at the head of the MAC frame that circulates every 2 milliseconds, and the gain value of the MT AGC amplifier is the value used when the previous MAC frame is received. Can be used continuously. Therefore, when this command is issued, the synchronization processing unit assumes that a signal is input with a decent amplitude value from the head of the broadcast burst preamble.

  In addition, when a broadcast burst is received for the first time, since the gain of the AGC amplifier is not set, it may be assumed that a signal is not input with a proper amplitude value from the head of the preamble of the broadcast burst. Therefore, when a broadcast burst is received for the first time, a command SYNC_TBC_PARA_START_H2UL_SEARCH is issued instead of this command. When the presence of the broadcast burst is confirmed by this command, the controller issues a command SYNC_TBC_PARA_START_H2DL_SEARCH two milliseconds later, and the procedure is to start the broadcast burst reception process in earnest.

E-2-6. SYNC_TBC_PARA_START_H2DL_RECEIVE
The command SYNC_TBC_PARA_START_H2DL_RECIVEIVE is supposed to be issued when attempting to receive an H / 2 downlink burst. When receiving a downlink burst, it is assumed that a broadcast burst has already been received in this MAC frame, and information on the gain of the AGC amplifier and the reception timing of the downlink burst is known. (However, detailed reception timing control is performed.)

  FIG. 17 shows the H / 2 downlink reception processing procedure when the broadcast burst can be received for one time. FIG. 18 shows a H / 2 downlink reception processing procedure when an uplink burst from another MT is mistaken for a broadcast burst.

E-3. In the internal processing procedure synchronization processing unit, there are a plurality of functional unit modules, and each module operates according to its purpose. The functional unit modules existing in the synchronization processing unit are shown below.

(1) Packet Detector (packet detection)
(2) Timing Detector (timing detection)
(3) Frequency Offset Measurement Block (frequency offset measurement)
(4) Digital Gain Control Block (Digital Gain Control)
(5) Preamble Extractor (preamble extraction)
(6) Symbol Extractor (symbol extraction)

  A collaborative operation performed by each of these modules in response to a command input from the controller will be described below.

E-3-1. Internal processing procedure (Case 1)
First, processing when the commands SYNC_TBC_PARA_START_11A_SEARCH and SYNC_TBC_PARA_START_H2UL_SEARCH are issued will be described with reference to FIG.

  When SYNC_TBC_PARA_START_11A_SEARCH and SYNC_TBC_PARA_START_H2UL_SEARCH are issued, the operation of the packet detection unit is started immediately and attempts to find the preamble head (Pre-1 and Pre-2 in FIG. 19). A coarse frequency offset value is also measured at the same time.

  The packet detection unit (Packet Detector) continues searching for preambles Pre-1 and Pre-2 using the value specified by iRxOrderParam2 as the threshold value over the time specified by the accompanying parameter iRxOrderParam1. FIG. 19 shows an example in which the output of the packet detection unit exceeding the threshold value is observed within the time.

  When the preamble Pre-2 part has been received, the output value of the packet detection part should output the maximum value, but after a while has passed since the maximum value was output for the purpose of extracting the maximum value, The correct reception timing and frequency offset value are determined.

  Other functional unit modules in the synchronization processing unit start the operation triggered by the output of the packet detection unit.

  The timing detector starts the operation when the output value of the packet detector exceeds the threshold value, and outputs the result around the start of reception of the preamble Pre-4 part. Information indicating which symbol to which symbol should be fed to the FFT is extracted from the output of the timing detection unit.

  The frequency offset measurement unit (Frequency Offset Measurement Block) starts a counter and starts an operation when the packet detection unit determines the reception timing and the frequency offset value. After that, the result is output around the time when the preamble Pre-4 part is received.

  A frequency offset correction value is determined based on the output value of the frequency offset measurement unit and the output value of the packet detection unit. Based on the determined frequency offset value, an oscillator for correcting this is operated, and the output of the oscillator is multiplied by a symbol fed to the FFT at the final stage of the synchronization processing unit.

  The digital gain control unit (Digital Gain Control Block) starts its operation as soon as the packet detection unit determines the reception timing and the frequency offset value. Thereafter, the result is output around the start of reception of the preamble Pre-4 part. The output of the digital gain control unit is a coefficient for absorbing the fluctuation of the signal power output from the A / D converter, and is multiplied by the symbol fed to the FFT at the final stage of the synchronization processing unit. .

  The preamble extractor starts the operation when all the outputs of the timing detection unit, the no-frequency offset measurement unit, and the digital gain control unit are ready, and stores them in the sequential synchronization processing unit. The corresponding symbol is fed to the FFT after a corresponding calculation is performed on the received symbol. When the preamplifier extraction unit finishes outputting the preamble Pre-3 and the Pre-4 part, the process ends.

  The symbol extractor starts the operation triggered by the completion of the process of the preamble extractor, performs a corresponding operation on the received symbol stored in the synchronization processor, and then performs a symbol calculation on the FFT. To feed. The symbol extraction unit continues processing until the command SYNC_TBC_PARA_BE_IDLE is issued from the controller.

E-3-2. Internal processing procedure (Case 2)
Next, processing when SYNC_TBC_PARA_START_H2DL_SEARCH and SYNC_TBC_PARA_START_H2DL_RECIVE are issued will be described with reference to FIG.

  These are commands issued only to MT of H / 2. Since it is impossible to receive a download burst suddenly in a certain MAC frame, the command SYNC_TBC_PARA_START_H2DL_SEARCH is always issued before the command SYNC_TBC_PARA_START_H2DL_RECEIVE is issued and the download burst is attempted to be received. It is a spear. Hereinafter, these sequences will be described later.

  As soon as SYNC_TBC_PARA_START_H2DL_SEARCH is issued, the operation of the packet detection unit is started and an attempt is made to find the preamble head part (Pre-1 and Pre-2 part in FIG. 20) of the broadcast burst. The packet detection unit continues the search process of the preamble Pre-1 and Pre-2 parts using the value specified by iRxOrderParam2 as the threshold value for the time specified by the accompanying parameter iRxOrderParam1.

  It is assumed that the SYNC_TBC_PARA_START_H2DL_SEARCH is issued at a timing when the controller determines that 2 milliseconds (MAC frame) time has elapsed since the reception of the previous broadcast burst.

  When the preamble Pre-2 part has been received, the output value of the packet detector should output the maximum value, but for the purpose of extracting the maximum value, a rough reception will occur after a while from the output of the maximum value. Determine timing and frequency offset values.

  Other functional unit modules in the synchronization processing unit start the operation triggered by the output of the packet detection unit.

  The timing detector starts its operation when the output value of the packet detector exceeds the threshold value, and outputs the result around the start of reception of the preamble Pre-4. Information indicating which symbol to which symbol should be fed to the FFT is extracted by the output of the timing detection unit.

  The frequency offset measurement unit starts a counter and starts an operation when the packet detection unit determines the reception timing and the frequency offset value. After that, the result is output around the time when the preamble Pre-4 part is received. A frequency offset correction value is determined based on the output value of the frequency offset measurement unit. In the H / 2 MT, since the frequency offset value with the AP is already known by the past burst reception, the frequency offset correction value calculated here is used for the purpose of refining the frequency offset value with the AP. . Based on the refined frequency offset value, an oscillator for correcting the frequency offset value is operated, and the output of the oscillator is multiplied by a symbol fed to the FFT at the final stage of the synchronization processing unit.

  The digital gain controller starts its operation as soon as the packet detector determines the reception timing and frequency offset value. Thereafter, the result is output around the start of reception of the preamble Pre-4 part. The output of the digital gain control unit is a coefficient for absorbing fluctuations in signal power output from the A / D converter, and is multiplied by a symbol fed to the FFT at the final stage of the synchronization processing unit.

  The preamble extraction unit starts the operation when all the outputs of the timing detection unit, frequency offset measurement unit, and digital gain control unit are ready, and receives the received symbols stored in the sequential synchronization processing unit. After performing the corresponding calculation, the symbol is fed to the FFT. When the preamble extraction unit finishes outputting the preamble Pre-3 and Pre-4 parts, it ends the processing.

  The symbol extraction unit starts an operation triggered by the completion of the processing of the preamble extraction unit, performs a corresponding operation on the received symbol stored in the synchronization processing unit, and then feeds the symbol to the FFT. The symbol extraction unit continues processing until the command SYNC_TBC_PARA_BE_IDLE is issued from the controller.

  Thus, the reception of the broadcast burst is completed.

  Thereafter, by decoding the signal in the broadcast burst, information such as the reception timing of the download burst transmitted in the MAC frame is extracted. Based on this information, the controller issues a command SYNC_TBC_PARA_START_H2DL_RECEIVE at the appropriate timing.

  The timing detection unit starts the counter with the issue of this command as a trigger, and starts the operation. The result is output around the start of reception of the preamble Pre-4 part. Information indicating which symbol to which symbol should be fed to the FFT is extracted by the output of the timing detection unit.

  The frequency offset measurement unit also starts operation by using the issuance of this command as a trigger. After that, the result is output around the time when the preamble Pre-4 part is received. The output value (frequency offset correction value) of the frequency offset measuring unit is used for the purpose of refining the frequency offset value with the AP. Based on the refined frequency offset value, an oscillator for correcting the frequency offset value is operated, and the output of the oscillator is multiplied by a symbol fed to the FFT at the final stage of the synchronization processing unit.

  The digital gain control unit also starts operation by issuing this command as a trigger and starts operation. Thereafter, the result is output around the start of reception of the preamble Pre-4 part. The output of the digital gain control unit is a coefficient for absorbing fluctuations in signal power output from the A / D converter, and is multiplied by a symbol fed to the FFT at the final stage of the synchronization processing unit.

  The preamble extraction unit starts the operation when all the outputs of the timing detection unit, frequency offset measurement unit, and digital gain control unit are ready, and receives the received symbols stored in the sequential synchronization processing unit. After performing the corresponding calculation, the symbol is fed to the FFT. When the preamble extraction unit finishes outputting the preamble Pre-3 and Pre-4 parts, it ends the processing.

  The symbol extraction unit starts an operation triggered by the completion of the processing of the preamble extraction unit, performs a corresponding operation on the received symbol stored in the synchronization processing unit, and then feeds the symbol to the FFT. The symbol extraction unit continues processing until the command SYNC_TBC_PARA_BE_IDLE is issued from the controller.

  This completes the reception of the download burst.

  In the above description, the command SYNC_TBC_PARA_START_H2DL_SEARCH is used for receiving the broadcast burst. The command SYNC_TBC_PARA_START_H2DL_SEARCH can be used for broadcast burst reception.

E-4. Internal Configuration of Synchronization Processing Unit The internal structure of the synchronization processing unit is illustrated by function unit as shown in FIG.

  Some functional units are determined not to operate at the same time, and some functional units include similar processing contents. When a circuit is formed, these similar processes can be performed using the same circuit and must be shared. Therefore, FIG. 22 shows an overall block diagram of the synchronization processing unit expressed on a circuit basis.

  In FIG. 22, since the synchronization processing unit is configured to operate in any of H / 2 AP, H / 2 MT, and IEEE 802.11a, the processing flow is difficult to understand. FIG. 23 shows the configuration of the synchronization processing unit when operating as H / 2 AP and IEEE 802.11a, and FIG. 24 shows the configuration of the synchronization processing unit when operating as H / 2 MT. . Even in the H / 2 MT, when the first broadcast burst is received, processing similar to the uplink bus and reception processing in the H / 2 AP is required.

  The major difference between FIG. 23 and FIG. 24 is where the frequency offset is corrected. In the case of H / 2 MT, the received signal is corrected with the frequency offset value obtained from the past reception result, regardless of presence or absence.

  Hereinafter, the configuration and processing contents inside each functional unit in the synchronization processing unit will be described in detail.

E-4-1. Packet Detection Unit FIG. 25 shows the internal configuration of the packet detection unit. The operation frequency inside the packet detector is assumed to be 20 MHz, and a signal transmitted from the A / D converter through the digital filter every 50 nanoseconds is input.

  The input signal to the packet detection unit is a complex conjugate multiplication of the output signal of the digital filter (Straight Input), the signal received 0.8 microseconds ago (Delayed Input), and the signal received 0.8 microseconds ago The result (Correlation Input) and the power of the received signal (Power Input).

  For the signal input from the complex conjugate result, the autocorrelation is calculated by performing cumulative addition over the past 0.8 microseconds.

  The signal input from the power of the received signal is subjected to cumulative addition of the signal input over the past 1.6 microseconds.

  The cumulative addition result (autocorrelation value) of the complex conjugate input is normalized by the cumulative addition result of the power of the received signal (this is performed in the Norm block in FIG. 25, but details will be given later). At this time, if the amplitude of the input signal is extremely small, the normalized value is set to a small value, so that the cumulative addition result of the received signal power is smaller than a certain threshold value. Let it bottom out at that threshold. Thus, the “autocorrelation value normalized by power” is generated.

  On the other hand, using the input S1 of the output signal of the digital filter and the signal input S2 received 0.8 microseconds earlier, the calculation shown in the following equation is performed to generate S3, and S3 is further applied over the past 0.8 microseconds Cumulative addition. Thus, the “cumulative addition value of S3” is generated.

  “Power-normalized autocorrelation values” generated as described above are stored in FIFO buffers that store signals for 0.8 microseconds. The FIFO buffer is configured in several stages in a chain shape as shown in the figure. The output of each FIFO buffer is added after the polarity is inverted in accordance with which device of which system in which system the packet detector is currently processing, and the addition result Sy is input to the comparator. .

  On the other hand, the “cumulative addition value of S3” is also stored in the FIFO buffer storing signals for 0.8 microseconds. As shown in the figure, the FIFO buffer is configured in several stages in a chain shape. The output of each FIFO buffer is added after the polarity is inverted depending on which system in which system the synchronization processing unit of the packet detection unit is currently processing (that is, depending on which command is received from the controller). The addition result Sx is input to the comparator.

  Inside the comparator, the signal detection unit outputs a signal Sy or a zero signal depending on which system in which system the synchronization processing unit is currently processing (that is, depending on which command is received from the controller). . The contents of polarity inversion processing and comparator internal processing in each case are shown below.

(1) When the command SYNC_TBC_PARA_START_H2UL_SEARCH or SYNC_TBC_PARA_SCAN is received while operating as an H / 2 receiver:
In the polarity inversion, the polarities of the “autocorrelation value normalized by power” calculated most recently and the “cumulative addition value of S3” calculated most recently are inverted.
In the comparator, a zero signal is output when the value of Sx is a negative value, and Sy is output otherwise. When receiving the preamble Pre-1 part and Pre-2 part of the uplink burst, since the value of Sx should be a positive value, the uplink is performed when the value of Sx becomes a negative value.・ The probability of receiving the preamble of the burst is regarded as zero, and a zero signal is output. This process has the effect of reducing the probability of erroneous detection.

(2) When SYNC_TBC_PARA_START_H2DL_SEARCH is received:
In the polarity inversion, the polarities of the “autocorrelation value normalized by power” calculated most recently and the “cumulative addition value of S3” calculated most recently are inverted.
The comparator outputs a zero signal if the value of Sx is a positive numerical value while receiving the Pre-1 part, and outputs Sy otherwise. On the other hand, while the Pre-2 part is being received, a zero signal is output if the value of Sx is a positive numerical value, and Sy is output otherwise. When the broadcast burst preamble Pre-1 part is received, the value of Sx should be a negative value, and therefore when the value of Sx becomes a positive value, the preamble Pre-1 of the broadcast burst. The probability of receiving a part is regarded as zero, and a zero signal is output. Also, when the broadcast burst preamble Pre-2 part is received, the value of Sx should be a positive value, so when the value of Sx becomes a negative value, the preamble Preb of the broadcast burst. The probability of receiving -2 part is regarded as zero and a zero signal is output. By this processing, the Pre-1 part and the Pre-2 part are identified, and erroneous detection is prevented.

(3) When the command SYNC_TBC_PARA_START_11A_SEARCH or SYNC_TBC_PARA_SCAN is received while operating as an IEEE802.11a receiver:
In polarity inversion, polarity inversion is not performed.
The comparator outputs Sy regardless of the value of Sx. That is, in this case, it is not necessary to perform signal processing on the output signal of the digital filter and the signals received 0.8 microseconds before.

  An amplitude value is calculated from the IQ pair based on the signal output from the comparator after the processing is performed according to the procedure described above, and the IQ pair and the amplitude value are passed to the peak estimation process.

  Next, peak estimation processing will be described. A command parameter from the controller is input to the peak estimation block, and information on how long the packet search is performed and a threshold value for determining whether or not the packet is received are held. Under this assumption, the following processing is performed.

(1) The input amplitude value is compared with the threshold value, and when the amplitude value exceeds the threshold value, this time, the IQ signal pair, and the amplitude value are stored.
(2) After an amplitude value exceeding the threshold value appears, if an amplitude value exceeding the threshold value appears again within a set period (specifically, 0.8 to 2.4 microseconds), these are coupled. Only those that show high amplitude values are stored.
(3) When a time period has elapsed since the last stored maximum amplitude value has been stored, it is considered that the maximum value of the amplitude value has been extracted, the stored time is recognized as the provisional reception timing, and The stored I-Q signal pair Arg is recognized as a temporary frequency offset value. Further, a trigger signal indicating that the preamble of the packet has been found is generated.

  26 and 27 show a flow of a series of processes so far. However, FIG. 26 shows timing detection unit internal signals and states at the time of H / 2 uplink or IEEE802.11a reception, and FIG. 27 shows timing detection unit internal signals and states at the time of H / 2 downlink broadcast burst reception. Show.

  As shown in each figure, when receiving an H / 2 uplink or IEEE802.11a, packet reception is recognized at one peak, but when receiving an H / 2 downlink broadcast burst, two peaks are received. Recognizing packet reception by extracting. In particular, at the time of initial synchronization, it is necessary to determine uplink bursts and broadcast bursts, and therefore recognition of reception of H / 2 downlink broadcast bursts is strictly set.

  Next, an internal configuration and operation of a normalization circuit (Norm. Block) configured in the packet detection unit will be described with reference to FIG. In this block, a process of normalizing the signal of the first input with the signal value of the second input is performed.

  First, the position of the most significant bit where 1 is set in the value of the second input is extracted, and the “position of the most significant bit where 1 is set” is the most significant bit in both the first input and the second input. Bit shift to Further, table lookup is performed with the value of the second input after the shift, and an inverse value component of this value is output. A value obtained by adding a constant to this value is a coefficient for normalization, and a normalized signal is output by multiplying the first input by this coefficient.

E-4-2. The operating frequency inside the timing detector timing detector is assumed to be 40 MHz. The main reason is that the time axis resolution is judged to be insufficient for 20 MHz processing. In the AWGN channel, there is no significant difference even with 20 MHz processing. However, in the propagation path model where each elementary wave arrives every 10 nanoseconds, such as the BRAN channel, the power of each delayed elementary wave is accurately calculated when the resolution is 20 MHz. Can not. Due to this, the time window of the signal fed to the FFT shifts, but it has been found that this influence cannot be ignored in a channel having a reasonably large delay dispersion such as a BRAN-C channel.

  FIG. 29 shows an internal configuration of the timing detection unit. An input signal to the timing detection unit is a reception signal with a corrected frequency offset, and operates as a synchronization processing unit of H / 2AP or IEEE802.11a. In this case, the input of the timing detection unit is obtained by correcting the input signal using the temporary frequency offset value output by the packet detection unit. Further, when operating as H / 2MT, the timing detection unit inputs the one corrected using the frequency offset value measured in the past.

  The timing detection process is basically performed by calculating a cross-correlation between the received preamble Pre-3 part and a replica of the Pre-3 part held in the timing detection part. The cross correlation is calculated over the effective symbol length of the Pre-3 part (specifically, 3.2 microseconds). The cross-correlation value is calculated in the time window range of 1.6 microseconds to 3.2 microseconds centering on the temporary reception timing extracted by the packet detection unit described above, and the timing having the highest cross-correlation value is the reception timing. It is a spear.

  However, in a channel with a delay path, it is not a good idea to simply recognize the timing with the highest cross-correlation value as the reception timing.

  For example, consider a case where signals are received via two-wave channels Path # 1 and Path # 2 as shown in FIG. In this case, the symbol is received as shown in FIG. 31, and when the timing detection unit recognizes the timing having the highest cross-correlation value as the reception timing, the signal in the section indicated by the illustrated FFT window is fed to the FFT. Resulting in. As a result, the energy of the next symbol is mixed, resulting in intersymbol interference.

  In order to prevent such a situation, the timing detector should determine the reception timing in the delay path channel with reference to the first wave that has arrived, as shown in FIG.

  In order to realize this, the first wave that has arrived first is searched for in the following procedure within the timing detection unit. The cross-correlation values at each timing are calculated, and all of them are input to the maximum peak extraction block. Inside this block, first, the timing at which the cross-correlation value is maximum is extracted, and the forefront timing at which the correlation value exceeding 60% (or 80%) of this correlation value appears is extracted as the reception timing. . FIG. 33 shows how reception timing is extracted based on the correlation value.

  Actually, in order to provide a further margin, a signal from the front is fed to the FFT somewhat (for example, for 200 nanoseconds) from the reception timing extracted by the timing detection unit. The smaller this margin is, the more likely it is to receive interference from the next symbol, and the more this margin is, the more likely it is to receive interference from the previous symbol (that is, it is potentially vulnerable to delayed waves). ).

E-4-3. Frequency offset measuring unit The operating frequency inside the frequency offset measuring unit is assumed to be 20 MHz.

  FIG. 34 shows the internal structure of the frequency offset measurement unit. The input signal of the frequency offset measuring unit is a complex conjugate multiplication result with the signal received 3.2 microseconds before. The circuit that performs complex conjugate multiplication can be shared with the circuit used to generate the input signal of the packet detector. When the frequency offset measurement unit is activated, the circuit that performs complex conjugate multiplication The received signal and the signal received 3.2 microseconds ago are switched to feed.

  In the frequency offset measurement unit, only the cumulative addition of the input signals is performed, and this process is performed from the start of reception of the preamble Pre-4 part to just before the reception of the Pre-4 part. .

  The frequency offset measurement unit outputs the frequency offset value as an IQ vector, similar to the temporary frequency offset value output from the packet detection unit. In order to convert this IQ vector into angle information, a block for calculating arctan () is required. This processing is performed by the circuit shown in FIG.

  arctan () receives the I-axis signal and the Q-axis signal and outputs angle information [radian]. First, the input I-axis signal is normalized by the Q-axis signal, and angle information is extracted by referring to the table with the normalized signal value. The normalization method is exactly the same as that of the normalization circuit (normalization) block that is a component of the packet detection unit.

  The arctan circuit operates using the output of the packet detection unit or the output of the frequency offset measurement unit (both IQ signals) as an input signal. Which is input is switched depending on whether the packet detection unit or the frequency offset measurement unit is currently active.

  Basically, it operates only twice per packet reception, so there is no restriction on the operating frequency, but it should be operated at a frequency that can minimize the processing delay.

  The frequency offset value generated by the frequency offset measurement unit and arctan () has a scope of ± 156.25 kHz. However, in practice, a signal having a frequency offset value exceeding this value may be input. Therefore, the final frequency offset value is determined with reference to the signal output from the packet detection unit in addition to the frequency offset value generated by the frequency offset measurement unit and arctan (). The provisional frequency offset value output from the packet detector has a scope up to ± 625.0 kHz, and it is possible to determine a rough frequency offset range from the constellation of the IQ signal output from the packet detector. The precision frequency offset calculation (Precision Frequency Offset Calculation) circuit shown in FIG. 36 performs the processing described above.

  The precision frequency offset calculation circuit is executed once per reception of one packet. As shown in FIG. 36, the precise frequency offset calculation circuit uses the output of the packet detection unit, the output of the frequency offset measurement unit, and the output of the arctan () circuit as input signals. A correction process is performed on the output value of the arctan () circuit in accordance with the condition determined from the output of the packet detection unit and the output of the frequency offset measurement unit. Specifically, the judgment condition and the correction are as shown in the following equation.

Where S _a . i, S _a . q is the output value of the frequency offset measurement unit, S _b . i, S _b . q is an output value of the packet detection unit, and Arg is an output of the arctan () circuit calculated based on the output of the frequency offset measurement unit.

  The frequency offset value generated in this way is input to the oscillator, and the output of the oscillator and the received signal are multiplied to complete the frequency offset correction process. Since an oscillator having a general circuit configuration can be applied, description thereof is omitted in this specification.

E-4-4. Digital Gain Control Unit The operating frequency inside the digital gain control unit is assumed to be 20 MHz.

  FIG. 37 shows the internal configuration of the digital gain control unit. The input signal of the digital gain control unit is the power value of the received signal, and cumulative addition of each symbol (sample) is performed internally to calculate the average power. Based on this average power value, a coefficient to be multiplied with the received signal is extracted by table lookup for the purpose of stabilizing the received signal level.

  The control signal to the AGC amplifier is locked when the packet detection unit finds and confirms the packet, but a certain delay occurs until a symbol amplified by the locked control signal is input ( Filter delay is the main cause). The digital gain control unit determines that the packet detection unit has found the packet, waits for this delay, and then starts the accumulation of power values, and performs the accumulation addition over 3.2 microseconds. .

  In the lookup table (Norm. Factor Table), conversion of power value → amplitude coefficient is performed.

E-4-5. Preamble extraction unit The preamble extraction unit extracts the Pre-3 part and Pre-4 part of the preamble stored in the memory according to the reception timing information output from the timing detection part, and extracts the extracted Pre-3 part and Pre- This block performs frequency offset correction for the four parts, and further feeds a signal to the FFT after performing amplitude correction.

  The preamble extraction unit is activated when all the processing in the timing detection unit, frequency offset measurement unit, and digital gain control unit is completed.

  Note that the preamble Pre-1 part and Pre-2 part are not output from the synchronization processing part. The preamble Pre-3 part and Pre-4 part are used for the purpose of generating a reference channel vector in the transmission path estimation part in the subsequent stage. Since the same signal circulates and transmits in the Pre-3 part and the Pre-4 part, if the same phase is added to generate a reference channel vector, the accuracy of the channel vector is improved by 3 dB in SNR conversion. To do.

  On the other hand, when the preamble extractor starts to operate, the preamble Pre-3 and Pre-4 have already been received, and there is a time lag until the signal is passed to the subsequent block. This time lag becomes a critical problem particularly in an IEEE802.11a receiver (since there is a case where an ACK has to be returned within 16 microseconds after receiving a signal, there is no time lag as much as possible. Need to be finished).

  Therefore, in the preamble extraction unit, the Pre-3 part and the Pre-4 part are added in phase at the time of the time axis signal, and the result of the addition is fed to the FFT, thereby completing the FFT execution of the preamble part in one time. I have to.

E-4-6. Symbol Extraction Unit The symbol extraction unit is activated after the preamble extraction unit feeds the preamble Pre-3 part and Pre-4 part to the FFT, and then feeds the signal to the FFT one after another every time an OFDM symbol is received. . Since the basic processing content is almost the same as that of the preamble extraction unit, the circuit is completely realized by the same circuit. In the symbol extraction unit, only the in-phase addition processing of the Pre-3 part and the Pre-4 part is not performed.

  The symbol extraction unit repeats the process endlessly until it receives a SYNC_TBC_PARA_BE_IDLE command from the controller.

F. Transmission path estimator The processing required for the transmission path estimator is defined as follows.

(1) Channel estimation Channel vector extraction using preamble (known sequence) and channel equalization by complex conjugate multiplication of symbols (2) Residual frequency offset correction Detection and correction of residual frequency offset using pilot subcarriers (3) Countermeasure for drift in sample timing Correction that reference phase is swung on the frequency axis due to drift in reception timing caused by mismatch in relative clock frequency between transmission and reception. (4) Channel vector update After provisional judgment Update the reference using symbols and follow the fading slowly

  Hereinafter, the configuration and operation of the transmission path estimation unit will be described in detail.

F-1. The main input signal to the control information transmission path estimation section from the controller to the transmission path estimation section is the output of the FFT (that is, the subcarrier signal). It is assumed that a signal indicating whether such processing should be performed is input from the controller. The instruction signals from the controller are assumed to be four shown in the table below.

  Hereinafter, the contents of the message indicated by each numerical value will be described.

F-1-1. OPERATION_MODE
OPERATION_MODE is an environment variable for instructing which system to perform synchronous processing on the synchronous processing unit.

  The numerical values that should be set in OPERATION_MODE are the three types shown in the table below, which is 2-bit information. However, the transmission path estimation unit does not change the MT reception process and AP reception process depending on the value of OPERATION_MODE. Therefore, it is sufficient to provide 1-bit information indicating whether IEEE802.11a reception processing or H / 2 reception processing is performed.

F-1-2. iSymbolIndex
It is assumed that the calculation result of FFT is input to the transmission path estimation unit in units of subcarriers. iSymbolIndex is a signal indicating in which OFDM symbol of the packet (PDU train) the currently input subcarrier is stored. It is assumed that the image is input as shown in FIG.

  However, the current counter value 48 may be saturated to the maximum value. It should be composed of about 6 bits.

  If there is experience in receiving a large number of packets from the same AP at the time of H / 2 downlink reception (MT reception processing), if an iSymbolIndex value is input as shown in FIG. Improved characteristics can be expected. Expected to improve the transmission characteristics by narrowing the scope of the residual frequency offset value assumed in the transmission path estimation unit based on the fact that the frequency offset value calculated with the AP in the synchronization processing unit is sufficiently refined It is.

F-1-3. iModMode
iModMode is information indicating the modulation method of the currently input signal, and four types of BPSK, QPSK, 16QAM, and 64QAM are assumed. The minimum is 2-bit information. The transmission path estimation unit is insensitive to the coding rate.

  There are cases where symbols of different modulation schemes exist in one packet. In this case, the value of iModMode is changed in a timely manner.

F-1-4. iFreqCoherent
iFreqCoherent is a flag that indicates whether information on adjacent subcarriers may be referred to when calculating a channel vector of a transmission path. For the time being, PHY_MODE in the portion that provides the highest bit rate in all symbols in the packet is one of the following, and 1 may be input otherwise.

BPSK coding rate R = 1/2
BPSK coding rate R = 3/4
QPSK coding rate R = 1/2

  When receiving IEEE802.11a, it should be noted that PHY_MODE is not clarified unless it is in the stage of receiving the second symbol of the payload part.

  In the above, iFreqCoherent has been uniquely determined by PHY_MODE. However, when the synchronization processing unit extracts information similar to the delay dispersion of the transmission path, this information may be fed to the transmission path estimation unit. Conceivable.

F-2. Control signal input timing The timing at which each signal from the controller should be input differs slightly depending on whether it operates as a H / 2 receiver or as an IEEE 802.11a receiver.

F-2-1. When operating as an H / 2 receiver In the case of H / 2, the number of LCH / SCHs constituting a PDU train and PHY_MODE in each transport channel are determined before signals are transmitted and received (MT Was extracted by decoding the information stored in the broadcast burst).

  Accordingly, iModMode (uniquely determined from PHY_MODE) and iFreqCoherent (currently uniquely determined from PHY_MODE) can be input simultaneously with the start of the PDU train reception process. The input timing of the control signal in this case is as shown in FIG.

F-2-2. When operating as an IEEE802.11a receiver In the case of IEEE802.11a, the number of symbols constituting a packet and information corresponding to PHY_MODE are stored in the first symbol of the payload portion. Therefore, neither iModMode nor iFreqCoherent can be set without decoding. The input timing of the control signal in this case is as shown in FIG.

  In the case of IEEE802.11a, a PLCP header indicating the length, modulation class, etc. of the packet is stored at the beginning of the packet (the first symbol of the payload). Since the modulation method of the information part after the second symbol in the payload part cannot be extracted unless this part is decoded, the value of iFreqCoherent is also input at the timing of iSymbolIndex = 2.

F-3. Internal Processing Procedure As described above, the processing contents in the transmission path estimation unit are as follows.

(1) Complex conjugate multiplication with channel vector (2) Detection and correction of residual frequency offset (3) Countermeasure for reception timing drift (4) Channel vector update

  Hereinafter, the procedure of each process will be described in detail.

F-3-1. When a complex conjugate multiplication packet with a channel vector is received, first, the preamble Pre-3 part / Pre-4 part is input from the FFT. The transmission path estimation unit handles the complex amplitude value obtained from the Pre-3 part / Pre-4 part as the initial value of the channel vector. This value is held internally, and complex conjugate multiplication is performed with the received symbol (difference is taken).

  Since the subcarriers are arranged over approximately 16 MHz, it is assumed that Coherency on the frequency axis is hardly expected. Therefore, a channel vector is generated for each subcarrier, and when a subcarrier whose information is modulated is input, complex conjugate multiplication is performed with the channel vector arranged at the same frequency.

  That is, when the input signal is Pre-3 part / Pre-4 part, the input symbol (subcarrier) is stored as a channel vector, and when the input signal is an information symbol, it is stored. Complex conjugate multiplication is performed between the channel vector and the input symbol. The complex conjugate multiplication is performed once every time one symbol (subcarrier) is input.

F-3-2. A signal whose frequency offset has been corrected by the synchronization processing unit is input to the residual frequency offset detection and correction FFT, but there is actually a residual frequency offset that could not be corrected by the synchronization processing unit. Therefore, a process for tracking the residual frequency offset is performed in the transmission path estimation unit.

  Since four pilot subcarriers, which are known patterns, are inserted in each OFDM symbol, the frequency offset value is sequentially updated each time one OFDM symbol is received.

  That is, the residual frequency offset value is detected once every time one OFDM symbol is received, and the detection result is reflected when the next OFDM symbol is received. Since the reflection of the residual frequency offset value (complex conjugate multiplication) is performed by rotating the phase of the channel vector, it is performed once every time one symbol (subcarrier) is input.

F-3-3. Countermeasure for reception timing drift When the master clock is shifted between transmission and reception, the reception timing drifts. As a result, the reference phase value for each subcarrier turns on the frequency axis, and the reference phase of the subcarrier transmitted and received at a frequency far from the center frequency is greatly rotated.

  In order to correct this, the reception timing drift is detected by referring to the signal point constellation after transmission path estimation of several subcarriers transmitted and received at a frequency far from the center frequency, and correction is applied.

  Similar to the detection of the residual frequency offset value, the detection of the drift state is performed once every time one OFDM symbol is received, and this detection result is reflected when the next OFDM symbol is received. Since reception timing drift information is reflected (complex conjugate multiplication) by rotating the phase of the channel vector, it is performed once every time one symbol (subcarrier) is input.

F-3-4. Channel vector update The channel vector obtains an initial value by holding the preamble Pre-3 / Pre-4 part, but is updated for the purpose of refining this vector each time an information symbol is received. Since the channel vector refinement is performed independently for each subcarrier of each frequency, it is performed once every time one symbol (subcarrier) is input.

F-4. Internal structure of transmission path estimation unit FIG. 42 shows the internal structure of the transmission path estimation unit for each functional unit. FIG. 43 shows the entire block of the transmission path estimation unit on a circuit basis. Hereinafter, the processing contents in each functional block of the transmission path estimation unit will be described in detail with reference to the drawings.

F-4-1. A preamble (Pre-3 part / Pre4 part) and a payload part are sequentially input to the channel vector and complex conjugate multiplication transmission path estimation part. While the preamble part is being input, the input symbol is channel- Stored as a vector in the FIFO buffer. This is shown in FIG.

  Two types of channel vectors are prepared. One stores the input preamble signal with its polarity inverted according to the known subcarrier pattern as it is, and the other stores it after weighted addition with adjacent subcarriers. In the weighted addition, the value obtained by dividing the amplitude of both subcarriers by 0.5 is added to the value obtained by multiplying the amplitude of the subcarrier by 1.0, and the value obtained by dividing the addition result by 2 is stored. In addition, the numerical value described in the symbol in FIG. 44 is a subcarrier number, P has shown that it is a Pad symbol.

  The reason why two channel vectors are prepared is to use them properly depending on whether or not coherency on the frequency axis may be expected. Whether it can be expected is determined by 1-bit information of iFreqCoherent input from the controller, and the output selector of the channel vector buffer is switched according to this value. The operation of the channel vector buffer output selector differs between the case of the H / 2 receiver and the case of the IEEE 802.11a receiver (described above).

  Next, the operation when the payload portion is input will be described with reference to FIG. Each time subcarriers (symbols) are sequentially loaded from the FFT, channel vectors corresponding to the same frequency carrier as the loaded subcarrier are sequentially output from the storage unit. FIG. 45 shows a case where 1 is input as iFreqCoherent and a channel vector referring to adjacent subcarrier information is used. In this case, of the two channel vectors stored in FIG. 44, the channel vector stored in the lower FIFO buffer is selected by the selector.

F-4-2. Residual frequency offset detection / correction and reception timing drift countermeasure For residual frequency offset, the residual frequency offset value is estimated using the pilot subcarriers inserted in the OFDM symbol and corrected by the second-order loop filter. multiply.

  Regarding reception timing drift, the reception timing drift is detected by referring to the signal point arrangement after transmission path equalization in several subcarriers transmitted and received at a frequency far from the center frequency, and correction is performed. Call. Similarly, the correction of the rotation of the reference phase caused by the reception timing drift is performed using a secondary loop filter.

  However, with regard to measures against drift in reception timing, it has been found that transmission can be performed without significant deterioration of characteristics without correction during BPSK and QPSK modulation, so 16QAM and 64QAM are modulated. The reception timing drift countermeasure is taken only when there is a difference (reversely, the characteristic deteriorates if correction is performed).

  FIG. 46 illustrates a processing operation for performing these corrections. The input value to each loop filter is an argument of IQ symbols as a result of adding the same phase for four pilot subcarriers after transmission path equalization. If there is a frequency offset, the same reference phase shift should occur for all subcarriers, and noise immunity is enhanced by adding the four pilot subcarriers in the same phase. Since it is cumbersome to obtain the argument, a Q-axis signal is used instead.

  Also, an estimated reference phase offset value is output from this system, and the output value is added to the reference phase value generated by the reception timing drift countermeasure block and then converted to an IQ vector representation in the oscillator block. . The IQ vector is multiplied by the channel vector output from the FIFO buffer to follow the residual frequency offset of the channel vector and take measures against reception timing drift.

On the other hand, information for several subcarriers far from the center frequency is used as a countermeasure for reception timing drift. When the reception timing drifts by Δt seconds, the reference phase offset value θ _f [radian] in a subcarrier separated by f [Hz] from the center frequency is expressed by the following equation.

  That is, the subcarriers arranged on the outer side have higher sensibility for the reception timing drift. This is the reason why information for several subcarriers far from the center frequency is used.

  The input to the reception timing drift detection corresponds to a signal obtained by subtracting the estimated signal constellation component (the output of the Dec block in FIG. 46) from the signal after transmission path equalization. The estimated signal constellation is extracted by making a tentative decision in the Dec block. If this estimated signal constellation is accurate and there is no frequency offset and no reception timing drift, the variation imposed on the received symbols should be input to the reception timing drift detection. . Therefore, the calculation shown in the following equation is performed on the subcarrier symbol S, and the variation is extracted as P.

  Since the subtraction is performed using the High Band symbol and the Low Band symbol, the effect of the residual frequency offset does not appear in P if it is assumed that there is no frequency characteristic in the transmission path.

  The value of the Q axis of the symbol P calculated by the above formula is input to the loop filter, and the turning amount on the frequency axis of the reference phase caused by the reception timing drift is calculated.

  In order to correct the turning of the reference phase on the frequency axis due to the reception timing drift, it is necessary to calculate a different reference phase value for each subcarrier. Performed in blocks.

F-4-3. Channel vector update The channel vector is updated each time a symbol is received. The update is performed by generating Vn according to the following equation, where Vn is a channel vector in the next state, Vc is a channel vector in the current state, and Vu is an update channel vector.

  Here, α is the Forget Factor, which is a numerical value around 0.95.

  FIG. 47 illustrates the processing operation of channel vector update. Vu is obtained by subtracting the “estimated signal constellation” component output from the Dec block from the received subcarrier (symbol). If the above-mentioned “estimated signal constellation” is correctly determined, Vu becomes the latest channel vector. Therefore, updating Vc with this information makes it possible to follow fading that fluctuates slowly. We are considering.

F-4-4. Temporary decision block In the temporary decision block (Dec. block in FIG. 46), a symbol and channel vector after transmission path equalization are input, and a modulation signal constellation in the symbol is estimated. In short, a process for temporarily determining which signal point is transmitted on the transmission side is performed, and the result is output. Further, a process equivalent to a Demapper that extracts soft decision bits from the input modulation symbol is also performed. This output is also the output of the transmission path estimation unit, and is fed to the deinterleaver.

  In addition to the symbols and channel vectors after transmission path equalization, information (iModMode) indicating which modulation scheme is currently being transmitted / received is also input to the provisional decision, and provisional decision processing and Demap processing are performed according to the modulation scheme. Is performed adaptively. FIG. 48 illustrates the structure of the provisional determination block.

F-4-5. In various setting parameter transmission path estimation units, loop filters are used in various places. The table below shows the various loop gain values that are determined to be as good as they are today.

  It is assumed that the input value to the loop filter of the reception timing drift countermeasure unit is a value normalized to the amount of phase rotation per subcarrier.

  The present invention has been described in detail above with reference to specific embodiments. However, it is obvious that those skilled in the art can make modifications and substitutions of the embodiment without departing from the gist of the present invention. That is, the present invention has been disclosed in the form of exemplification, and the contents described in the present specification should not be interpreted in a limited manner. In order to determine the gist of the present invention, the claims section described at the beginning should be considered.

It is the figure which showed the time view of the OFDM symbol. It is the figure which showed the frequency view of the OFDM symbol. It is the figure which showed the structure of the preamble at the time of the broadcast (broadcast communication) and burst in H / 2. It is the figure which showed the structure of the preamble at the time of H / 2 downlink burst. It is the figure which showed the structure of the preamble at the time of H / 2 uplink burst. It is the figure which showed the structure of the packet preamble of IEEE802.11a. It is the figure which showed the structure of the MAC frame. It is the figure which showed typically the structure of the transmitting apparatus which concerns on one Embodiment of this invention. It is the figure which showed typically the structure of the receiver which concerns on one Embodiment of this invention. It is the figure which showed the input / output characteristic of the amplifier used for a normal transmission power amplifier. It is the figure which showed the relationship between required SNR and BER in each transmission rate mode when the output back-off is taken and a margin of 3 dB is taken. It is the figure which showed the structure (power amplifier part) of the transmitter which can switch the back-off value given to transmission amplifier according to a transmission rate mode. It is the figure which showed the structure of the control loop which controls AGC amplifier by digital processing. It is the figure which showed the internal structure of the gain control part in FIG. It is the figure which showed the example of issue of command SYNC_TBC_PARA_SCAN. It is a figure for demonstrating the process when SYNC_TBC_PARA_START_11A_SEARCH and SYNC_TBC_PARA_START_H2UL_SEARCH are issued. It is the figure which showed the H / 2 downlink reception process procedure when the broadcast burst can be received by one luckily. It is the figure which showed the H / 2 downlink reception process procedure when the uplink burst from other MT is mistaken with the broadcast burst. It is a figure for demonstrating a process when command SYNC_TBC_PARA_START_11A_SEARCH and SYNC_TBC_PARA_START_H2UL_SEARCH are issued. It is a figure for demonstrating a process when SYNC_TBC_PARA_START_H2DL_SEARCH and SYNC_TBC_PARA_START_H2DL_RECEIVE are issued. It is the figure which showed the internal structure of the synchronous process part according to the functional unit. It is a whole block diagram of the synchronous processing part described with the circuit base. It is the figure which showed the structure of the synchronous process part in the case of operate | moving as H / 2 AP and IEEE802.11a. It is the figure which showed the structure of the synchronous process part in the case of operate | moving as H / 2 MT. It is the figure which showed the internal structure of the packet detection part. It is the figure which showed the timing detection part internal signal and state at the time of H / 2 uplink or IEEE802.11a reception. It is the figure which showed the timing detection part internal signal and state at the time of H / 2 downlink broadcast burst reception. It is the figure which showed the internal structure of the normalization circuit comprised inside a packet detection part. It is the figure which showed the internal structure of the timing detection part. It is the figure which showed a mode that the signal was received by solving a 2 path | pass model. It is the figure which showed the mode of the symbol reception (intersymbol interference) at the time of a signal being received via 2 wave channels as shown in FIG. In the delay path channel, as shown in FIG. 32, it is a diagram showing a state in which inter-symbol interference is avoided by determining the reception timing based on the previously arrived elementary wave. It is the figure which showed a mode that reception timing was extracted based on a correlation value. It is the figure which showed the internal structure of the frequency offset measurement part. It is the figure which showed the internal structure of the arctan () circuit. It is the figure which showed the internal structure of the precision frequency offset calculation (Precision Frequency Offset Calculation) circuit. It is the figure which showed the internal structure of the digital gain control part. It is the figure which showed the input image of iSymbolIndex. It is a figure showing signs that iSymbolIndex value is inputted. It is the figure which showed the input timing (in the case of H / 2 receiver) of each control signal. It is the figure which showed the input timing (in the case of an IEEE802.11a receiver) of each control signal. It is the figure which showed the internal structure of the transmission-line estimation part according to the functional unit. It is the figure which showed the whole block of the transmission-line estimation part by the circuit base. It is the figure which showed a mode that the input symbol was accumulate | stored in a FIFO buffer as a channel vector. It is a figure for demonstrating the operation | movement at the time of the payload part being input. It is a figure for demonstrating the operation | movement of residual frequency offset correction | amendment and a reception timing drift countermeasure. It is a figure for demonstrating the processing operation | movement of a channel vector update. It is the figure which showed the structure of the temporary determination block.

Claims (2)

A communication apparatus that operates under a communication system in which signals are transmitted and received in bursts using an OFDM modulation method, in which a preamble is added to the head of a burst signal, and a known pilot sub signal is added to some subcarriers of an information signal. The carrier is inserted,
Means for adding pilot subcarriers after transmission path estimation as long as they are inserted;
Means for estimating a frequency offset value from the signal points of the added pilot subcarriers;
Means for updating the channel vector to follow the frequency offset by a loop filter based on the estimated frequency offset value;
Multiple loop filters with different loop gains to track the frequency offset;
Equipped with,
Deciding whether to use the calculation result of any of the loop filters for correction according to the number of times of receiving the symbol,
A communication device. A communication apparatus that operates under a communication system in which signals are transmitted and received in bursts using an OFDM modulation method, in which a preamble is added to the head of a burst signal, and a known pilot sub signal is added to some subcarriers of an information signal. The carrier is inserted,
Means for adding pilot subcarriers after transmission path estimation as long as they are inserted;
Means for estimating a frequency offset value from the signal points of the added pilot subcarriers;
Means for updating the channel vector to follow the frequency offset by a loop filter based on the estimated frequency offset value;
A channel vector estimate is calculated by tentatively determining the received subcarriers of the information symbol, and the reception timing drift is estimated from the difference between the channel vector stored inside and the channel vector estimate, and a loop filter Correction means for correcting by
Comprising
Adapt the subcarrier used to estimate the reception timing drift only for subcarriers far from the center frequency.
A communication device.