JP4583431B2 - Modulator and modulation method - Google Patents

Description

The present invention relates to a modulator and a modulation method used when transmission data is encoded and transmitted using a block code such as an LDPC (Low Density Parity Check) code.

  In wireless communication, transmission data is generally encoded and transmitted in order to improve error correction capability. As one of the encodings, there is an LDPC code as described in Non-Patent Document 1. Since this LDPC code can perform error correction in a very large block unit (constraint length), it is considered to be resistant to burst errors and suitable for communication in a fading environment.

  As a technique for increasing the data transmission speed, a multi-antenna transmission apparatus that transmits OFDM signals from a plurality of antennas as described in Non-Patent Document 2 is known. In this type of multi-antenna transmission apparatus, as one method for suppressing burst errors due to frequency selective fading, it has been proposed to interleave data in the frequency direction (subcarrier direction).

FIG. 28 shows a frame configuration example of a transmission signal by this type of multi-antenna transmission apparatus. In FIG. 28, a preamble for estimating distortion due to fading fluctuation, that is, channel estimation and a frequency offset between the transmitter and the receiver is arranged at the head of the frame, followed by a data symbol. Also, pilot symbols for estimating a frequency offset that varies with time are arranged in carrier Y. One square in the figure represents one symbol. That is, in the example of FIG. 28, at each time i, i + 1,..., One OFDM symbol consisting of a total of 7 symbols including data symbols and pilots is transmitted. At this time, the data is interleaved within one OFDM symbol, and is arranged in the order of (1) (2) (3) ... (11) (12).
"Low-density parity check code and its decoding method LDPC (Low Density Parity) code / sum-product decoding method" Trikeps 2002 "High Speed Physical Layer (PHY) in 5GHzband" IEEE802.11a 1999.

  By the way, when a block code such as LDPC is used, for example, as the modulation multi-level number is increased, the number of symbols for transmitting one encoded block is reduced, and one encoded block is transmitted in a shorter time. Will be. As a result, when there is a notch due to fading in the transmission period, a burst error is likely to occur. That is, as the modulation multi-level number increases, the probability of burst error increases.

  Here, the block code of LDPC or the like can change the block size. The larger the block size (that is, the longer the constraint length), the smaller the probability of burst errors due to fading notches and the like. Therefore, when the modulation multilevel number is changed as in adaptive modulation, it is considered that the burst error can be suppressed by increasing the encoding block size as the modulation multilevel number is increased.

  However, it is not desirable to design the encoder so as to change the block size each time the modulation multi-level number is changed, because the configuration of the encoder becomes complicated.

  Furthermore, in multi-antenna transmission such as MIMO (Multiple-Input Multiple-Output), high separation accuracy can be secured for the data symbol immediately after the preamble arranged at the head of the frame, so that a high SNR is secured as a received signal. However, there is a problem that the SNR of the received signal also decreases because the separation accuracy decreases as the distance from the preamble increases.

  The present invention has been made in view of such points, and a transmission apparatus capable of suppressing burst errors with a relatively simple configuration without changing the block size of an encoded block even when the modulation multilevel number is increased, and An object of the present invention is to provide a multi-antenna transmission apparatus capable of suppressing deterioration of error rate characteristics due to a distance from a preamble.

In order to solve such a problem, one aspect of the modulator of the present invention is to perform error correction coding processing on transmission data composed of a plurality of bits, and to perform a plurality of errors each composed of a plurality of bits. encoding means for forming a correction coded data, one symbol, as configured to collect bits belonging to one of the plurality of the error correction coded data, the plurality of error correction coded data Rearrangement means for rearranging belonging bits, and modulation means for outputting a baseband signal corresponding to the symbol from the rearranged error correction encoded data, and arbitrarily selecting from a plurality of bits constituting the symbol two bits to be extracted, a bit belonging to the erroneous Ri correction coded data that different from each other, said permutation means is a plurality of mis-Ri correction coded data The generated data of one system is input, and the bits included in the data of the one system are rearranged. In adjacent symbols, the same bit number among the bit numbers indicating the signal point arrangement of the symbols is set. reordering to assign bits belonging to the erroneous Ri correction coded data that different from one another, one error correction coded data and the number of bits constituting the single error correction coded data symbol number and the same used to send the The structure which is is taken .

Also, one aspect of the modulation method of the present invention is to perform error correction encoding processing on transmission data composed of a plurality of bits and to provide a plurality of error correction encoded data each composed of a plurality of bits. And rearranging the bits belonging to the plurality of error correction encoded data so that one symbol collects bits belonging to any of the plurality of error correction encoded data, and rearranging A modulation method for outputting a baseband signal corresponding to the symbol from the error correction coded data, wherein two bits arbitrarily extracted from a plurality of bits constituting the symbol are different from each other in error correction coding Bits belonging to data, and the rearrangement of the bits is performed by inputting one system of data composed of the plurality of error correction encoded data, The bits included in the data are rearranged. In adjacent symbols, rearrangement is performed so that bits belonging to different error correction encoded data are assigned to the same bit number indicating the signal point arrangement of the symbol. The number of bits constituting one error correction encoded data is the same as the number of symbols used for transmitting the one error correction encoded data .

  According to the present invention, since the data in each coding block is discretely arranged in a plurality of symbols, burst errors can be suppressed even when the number of modulation multi-values is increased. A transmission apparatus that can suppress deterioration of error rate characteristics due to fading notches or the like can be realized with a relatively simple configuration without changing the size.

  In addition, since the distance from the preamble between the respective encoded blocks can be substantially uniformed, a multi-antenna transmission apparatus that can suppress deterioration of error rate characteristics due to the distance from the preamble can be realized.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

(Embodiment 1)
FIG. 1 shows the configuration of the transmission apparatus according to Embodiment 1 of the present invention. The transmission device 10 inputs the transmission data S1 to the encoding unit 11. The encoding unit 11 performs block encoding processing on the transmission data S1, and sends the block encoded data S2 obtained thereby to the rearrangement unit 12. In the case of this embodiment, the encoding unit 11 performs LDPC encoding processing.

  The rearrangement unit 12 rearranges the block encoded data S2 and supplies it to the modulation unit 15 so that the intra-block encoded data of the encoded blocks having different one data symbols are collected. Specifically, the block coded data S2 is input to the selector 13, and the selector 13 sends the block coded data S2 to the memories 14-1 to 14-3 or the modulation unit 15 in bit units. The memories 14-1 to 14-2 function as buffer memories, and send the temporarily stored bits to the modulation unit 15 at the same timing. For example, when QPSK is performed by the modulation unit 15, the memory 14-1 is used, and the bits stored in the memory 14-1 are output in synchronization with the bits sent directly from the selector 13 to the modulation unit 15. . As a result, the modulation unit 15 forms one QPSK symbol using a total of two bits, the bit input from the memory 14-1 and the bit input directly from the selector 13. In addition, when 16QAM is performed by the modulation unit 15, the memories 14-1 to 14-3 are used, and the bits stored in the memories 14-1 to 14-3 are directly transmitted from the selector 13 to the modulation unit 15. Output at the same timing. As a result, the modulation unit 15 forms one symbol of 16QAM by using a total of 4 bits including the bits input from the memories 14-1 to 14-3 and the bits directly input from the selector 13.

  For simplicity, FIG. 1 shows only three memories 14-1 to 14-3, but in the case where 64QAM is performed by the modulation unit 15, five memories are provided, 15, one symbol of 64QAM may be formed using a total of 6 bits, that is, an input from each memory and a bit directly input from the selector 13.

  Note that the configuration of the rearrangement unit 12 shown in FIG. 1 is an example. In short, the block encoded data S2 is rearranged and the modulation unit 15 so that one intra-block encoded data is allocated to a plurality of data symbols. Any configuration may be adopted as long as the configuration is supplied to the printer.

  The modulation unit 15 performs adaptive modulation based on the control signal S10. That is, the modulation unit 15 changes which modulation process is performed among BPSK, QPSK, 16QAM, and 64QAM based on the control signal S10. The control signal S10 is also input to the selector 13 of the rearrangement unit 13, and the selector 13 changes the bit rearrangement rule depending on which modulation processing is performed in the modulation unit 15. Details thereof will be described later.

  The transmission symbol S3 obtained by the modulation unit 15 is input to the radio unit 16. The radio unit 16 performs predetermined modulation processing such as digital-analog conversion and up-conversion on the modulation symbol S3, and supplies the RF signal S4 obtained thereby to the antenna 17.

  An LDPC code generation process performed by the encoding unit 11 according to the present embodiment will be described with reference to FIG. An encoding unit (LDPC encoder) 11 receives transmission data S1 (that is, data before LDPC encoding) as input, and outputs block encoded data S2 (that is, data after LDPC encoding) by LDPC encoding this. For example, if the data before LDPC encoding is (m1a, m2a,..., M490a) and the parity check matrix is G, (C1a, C2a,..., C980a) is output as data after LDPC encoding. The That is, a post-encoding block # 1 consisting of 980 bits is formed from a pre-encoding block # 1 consisting of 490 bits.

  The modulation process by the modulation unit 15 will be described with reference to FIG. Since this modulation processing is a well-known technique, it will be briefly described. FIG. 3A shows the signal point arrangement of BPSK, and one bit, that is, b1 is transmitted in one symbol. FIG. 3B shows the signal point arrangement of QPSK, and 2 bits, that is, (b1, b2) are transmitted in one symbol. FIG. 3C shows a 16QAM signal point arrangement, and 4 bits, that is, (b1, b2, b3, b4) are transmitted in one symbol. FIG. 3D shows a 64QAM signal point arrangement, and 4 bits, that is, (b1, b2, b3, b4, b5, b6) are transmitted in one symbol.

  Next, the rearrangement process by the rearrangement unit 12 that is a feature of the present embodiment will be described with reference to FIGS. 4 to 7 show to which symbol after modulation the bits in each coding block subjected to LDPC coding are assigned. Specifically, this indicates which symbol is allocated to one intra-block encoded data (data after LDPC encoding) composed of 980 bits. The horizontal axis indicates the temporal arrangement of symbols. The vertical axis represents the bit number constituting one symbol, that is, b1 for BPSK, b1, b2 for QPSK, b1, b2, b3, b4 for 16QAM, b1, b2, b3, b4, b5, b6 for 64QAM. Is shown.

  Furthermore, #XY in the figure indicates the Yth bit (Yth bit among 980 bits) of the Xth coded block. For example, # 1-1 indicates the first bit of the first encoded block. Similarly, for example, # 3-979 indicates the 979th bit of the third encoded block.

  FIG. 4A shows bit allocation to each symbol when the modulation method is BPSK. When the modulation method is BPSK, one symbol (b1) is transmitted in one symbol, so only one 980-bit encoded block is transmitted in 980 symbols.

  FIG. 4B shows bit allocation to each symbol when the modulation scheme is QPSK. When the modulation method is QPSK, two bits (b1, b2) are transmitted in one symbol, so two 980-bit encoded blocks can be transmitted in 980 symbols. As is apparent from the figure, each symbol here is configured by collecting intra-block encoded data of different encoded blocks. Specifically, bits # 1-1 to # 1-980 of post-encoding block # 1 are assigned to bit b1 of 980 symbols of QPSK, and data # 2- of post-encoding block # 2 is assigned to bit b2 of 980 symbols. 1 to # 2-980 are assigned. As a result, the bits (data) in each coding block can be distributed in time over the same number of symbols as BPSK, so that the quality of the data in the coding block as a whole is improved by notching due to fading. You can avoid being depressed. That is, since the probability that most data in the coding block is erroneous in a burst manner is low, the error rate characteristic can be improved.

  FIG. 4C shows bit assignment to each symbol when the modulation scheme is 16QAM. When the modulation method is 16QAM, four bits (b1, b2, b3, b4) are transmitted in one symbol, so that four 980-bit encoded blocks can be transmitted with 980 symbols. The feature of bit assignment to each symbol here is that one piece of intra-block encoded data is assigned to a plurality of symbols, as in QPSK. Specifically, data # 1-1 to # 1-980 of block # 1 after encoding are assigned to bit b1 of 980 symbols of 16QAM, and data # 2- of block # 2 after encoding are assigned to bits b2 of 980 symbols. 1 to # 2-980, data # 3-1 to # 3-980 of block # 3 after encoding are assigned to bit b3 of 980 symbols, and data # of block # 4 after encoding are assigned to bit b4 of 980 symbols 4-1 to # 4-980 are allocated. As a result, the bits (data) in each coding block can be distributed in time over the same number of symbols as BPSK, so that the quality of the data in the coding block as a whole is improved by notching due to fading. You can avoid being depressed. That is, since the probability that most data in the coding block is erroneous in a burst manner is low, the error rate characteristic can be improved.

  FIG. 4D shows bit allocation to each symbol when the modulation scheme is 64QAM. When the modulation method is 64QAM, 6 bits (b1, b2, b3, b4, b5, b6) are transmitted in one symbol, so that 6 980-bit encoded blocks can be transmitted with 980 symbols. The feature of bit assignment to each symbol here is that one piece of intra-block encoded data is assigned to a plurality of symbols, as in QPSK and 16QAM. Specifically, data # 1-1 to # 1-980 of block # 1 after encoding are assigned to bit b1 of 980 symbols of 64QAM, and data # 2- of block # 2 after encoding are assigned to bits b2 of 980 symbols. 1 to # 2-980, data # 3-1 to # 3-980 of block # 3 after encoding are assigned to bit b3 of 980 symbols, and data # of block # 4 after encoding are assigned to bit b4 of 980 symbols 4-1 to # 4-980 are allocated, data # 5-1 to # 5-980 of post-encoding block # 5 are allocated to bit b5 of 980 symbols, and post-encoding block # 6 is allocated to bit b6 of 980 symbols Data # 6-1 to # 6-980 are allocated.

  As a result, the bits (data) in each coding block can be distributed in time over the same number of symbols as BPSK, so that the quality of the data in the coding block as a whole is improved by notching due to fading. You can avoid being depressed. That is, since the probability that most data in the coding block is erroneous in a burst manner is low, the error rate characteristic can be improved.

  Next, a second example of the rearrangement process of the rearrangement unit 12 of this embodiment will be described with reference to FIG. 5 is the same as FIG. 4 in that one intra-block encoded data is assigned to a plurality of symbols, and the same effect as when rearranged as shown in FIG. 4 can be obtained. FIG. 5 differs from FIG. 4 in that, in QPSK, 16QAM, and 64QAM, one encoded block is not allocated to fixed bits (for example, only b1), but all bits (for example, b1, b2 in the case of 16QAM). , B3, b4). Specifically, for example, when the modulation scheme is 16QAM, for the post-encoding block # 1, data # 1-1 is assigned to b1, # 1-2 is assigned to b2, # 1-3 is assigned to b3, A feature is that data of block # 1 is transmitted using b1, b2, b3, and b4 such that # 1-4 is assigned to b4.

  The reason why such an allocation method is adopted will be described. There are differences in 16QAM b1 reception quality, b2 reception quality, b3 reception quality, and b4 reception quality. At this time, it is assumed that the reception quality of b1 is the worst. Then, for example, when block # 1 is transmitted using only b1, block # 1 becomes a block with poor reception quality. When packet communication is performed, the packet error is affected by the reception quality of the block with the worst reception quality. Therefore, in this case, it is better to make the reception quality of the blocks # 1 to # 4 as uniform as possible. This can be realized by assigning as shown in FIG. Further, preferably, the number of times assigned to b1, b2, b3, and b4 for block # 1 to block # 4 should be as uniform as possible. Note that the difference in the number of allocations is preferably at most one. Incidentally, the difference of one time is not necessarily a multiple of 4 (bits) (the number of bits that 16QAM can be transmitted in one symbol), so it will occur regardless of how it is assigned. Sometimes.

  Although the case of 16QAM has been described here as an example, the same effect can be obtained by performing the same processing for 64QAM. However, in the case of QPSK, since there is no difference in reception quality between b1 and b2, the same effect cannot always be obtained. However, there is a possibility that an effect can be obtained because the possibility of a difference in reception quality due to distortion generated by the transmission device and the reception device cannot be denied.

  Next, FIG. 6 shows a third example of the rearrangement process of the rearrangement unit 12 of this embodiment. 6 is the same as FIG. 4 in that one intra-block encoded data is assigned to a plurality of symbols, and the same effect as when rearranged as shown in FIG. 4 can be obtained. FIG. 5 differs from FIG. 4 in that the same block data is transmitted in the same symbol. However, in QPSK, the data in block # 1 and the data block in block # 2 are alternately transmitted in 16QAM. In the order of block # 1, block # 2, block # 3, 64QAM is transmitted in the order of block # 1, block # 2, block # 3, block # 4, block # 5, block # 6. is there. That is, instead of assigning block data to consecutive symbols as shown in FIG. 4, it may be assigned to spaced symbols. However, the allocation method as shown in FIGS. 4 and 5 can distribute the intra-block data to a larger number of symbols, so that the effect of improving the reception quality is higher.

  Next, the 4th example of the rearrangement process of the rearrangement part 12 of this Embodiment is shown in FIG. 7 is the same as FIG. 4 in that one intra-block encoded data is assigned to a plurality of symbols, and the same effect as when rearranged as shown in FIG. 4 can be obtained. FIG. 7 shows an example in which the ideas of FIGS. 5 and 6 are combined. In FIG. 7, the assigned symbols are changed in units of 2 bits. As a result, the same effect as in FIG. 4 and FIG. 5 can be obtained, but the allocation method as in FIG. 4 and FIG. 5 can distribute the intra-block data to more symbols, so that the reception quality is improved. The effect is high.

  As described above, according to the present embodiment, the encoding unit 11 that performs block encoding processing on transmission data to form block encoded data, and the modulation that modulates block encoded data to form data symbols. By providing the reordering unit 12 and the reordering unit 12 for reordering the block coded data and supplying the data to the modulating unit 15 so that the intra-block coded data of coding blocks having different one data symbols are assembled. Even when the modulation multilevel number is increased, it is possible to realize the transmission apparatus 10 capable of suppressing burst errors with a relatively simple configuration without changing the block size of the encoded block.

  Incidentally, the processing of the rearrangement unit 12 rearranges the block encoded data so that the greater the number of modulation multi-values in the modulation unit 15 is, the more one block contains block encoded data of more blocks. It can also be said.

(Embodiment 2)
FIG. 8 shows the configuration of the multi-antenna transmission apparatus according to Embodiment 2 of the present invention.

  The multi-antenna transmission apparatus 100 is a transmission apparatus that performs so-called OFDM-MIMO communication, and transmits different modulation signals from two antennas. Specifically, modulated signal A is transmitted from antenna 114A, and modulated signal B is transmitted from antenna 114B. Here, in FIG. 8, since the signal processing system for the modulation signal A and the signal processing system for the modulation signal B have substantially the same configuration, “A” is added after the code for the processing system of the modulation signal A. The processing system of the modulation signal B corresponding thereto is indicated by adding “B” after the symbol.

  The frame configuration signal generator 115 of the multi-antenna transmission apparatus 100 outputs a control signal 116 such as information on the frame configuration, information on the encoding method, and information on the modulation scheme. Encoding section 102A receives data 101A of modulated signal A and control signal 116 as input, performs encoding based on control signal 116, and outputs encoded data 103A.

  Rearranger 104A receives encoded data 103A and control signal 116 as input, rearranges encoded data 103A based on control signal 116, and outputs rearranged data 105A.

  The modulation unit 106A receives the rearranged data 105A and the control signal 116, modulates one of BPSK, QPSK, 16QAM, and 64QAM based on the control signal 116, and outputs a baseband signal 107A.

  The serial / parallel converter (S / P) 108A receives the baseband signal 107A, performs serial / parallel conversion, and outputs a parallel signal 109A. The inverse Fourier transform unit (ift) 110A receives the parallel signal 109A, performs a Fourier transform, and outputs a Fourier-transformed signal 111A, that is, an OFDM signal. The radio unit 112A receives the signal 111A after Fourier transform, and performs predetermined radio processing such as frequency conversion and amplification to form a transmission signal 113A of the modulation signal A. The transmission signal A is output as a radio wave from the antenna 114A.

  The same processing is performed on the modulation signal B by the encoding unit 102B, the rearrangement unit 104B, the modulation unit 106B, the serial / parallel conversion unit (S / P) 108B, the inverse Fourier transform unit (ift) 110B, and the radio unit 112B. The transmission signal 113B of the modulation signal B is output as a radio wave from the antenna 114B.

  FIG. 9 shows a frame configuration example of the modulated signal A and the modulated signal B transmitted from the antennas 114A and 114B of the multi-antenna transmission apparatus 100. FIG. 9A shows the frame configuration of the modulated signal A transmitted from the antenna 114A, and FIG. 9B shows the frame configuration of the modulated signal B transmitted from the antenna 114B. In the case of this embodiment, spatially multiplexed MIMO (Multi-Input Multi-Output) transmission is used as a communication method, so that symbols of modulated signal A and modulated signal B at the same carrier and at the same time are transmitted simultaneously from different antennas. And multiplexed in space.

  The preamble arranged at the head of the frame is for estimating the channel fluctuation. The receiver estimates the channel fluctuation using the preamble, and performs ZF (Zero Forcing) and MMSE (Minimum Mean Square Error). By applying, modulated signal A and modulated signal B can be separated.

  The pilot symbols arranged in the time direction of the carrier Y are used as symbols for estimating and removing frequency offsets that could not be removed by the preamble and distortion (amplitude / phase) due to device characteristics in the receiving apparatus. Symbol.

  The data symbol is a symbol for transmitting data, and is transmitted following the preamble.

  FIG. 10 shows a configuration of a multi-antenna reception apparatus that receives and demodulates a signal transmitted from multi-antenna transmission apparatus 100.

  Radio section 303_1 of multi-antenna receiving apparatus 300 receives reception signal 302_1 received by antenna 301_1, performs amplification, frequency conversion, etc., and outputs baseband signal 304_1. The Fourier transform unit (fft) 305_1 receives the baseband signal 304_1, performs a Fourier transform, and outputs a signal 306_1 after the Fourier transform.

  The channel fluctuation estimation unit 307_1 of the modulated signal A receives the signal 306_1 after Fourier transform, extracts the preamble of the modulated signal A shown in FIG. 9A, and calculates the channel fluctuation of the modulated signal A based on this preamble. The channel fluctuation estimation signal 308_1 of the modulated signal A is output.

  The channel fluctuation estimation unit 309_1 of the modulated signal B receives the signal 306_1 after Fourier transform, extracts the preamble of the modulated signal B shown in FIG. 9B, and calculates the channel fluctuation of the modulated signal A based on this preamble. Then, the channel fluctuation estimation signal 310_1 of the modulation signal B is output.

  Radio section 303_2, Fourier transform section 305_2, channel fluctuation estimation section 307_2 for modulated signal A, and channel fluctuation estimation section 309_2 for modulated signal B operate in the same manner as described above.

  The signal processing unit 311 receives the signals 306_1 and 306_2 after Fourier transform, the channel fluctuation estimation signals 308_1 and 308_2 of the modulation signal A, and the channel fluctuation estimation signals 310_1 and 310_2 of the modulation signal B, and inputs ZF (Zero Forcing), MMSE ( The received data 312A of the modulated signal A and the received data 312B of the modulated signal B are obtained by performing processing such as (Minimum Mean Square Error) and decoding. Detailed operation of the signal processing unit 311 will be described later with reference to FIG.

  FIG. 11 shows a communication model between the multi-antenna transmission apparatus and the multi-antenna reception apparatus. The modulated signal transmitted from the antenna 409A is Txa (t), and the modulated signal transmitted from the antenna 409B is Txb (t) (t: time). Further, channel fluctuations between the transmitting and receiving antennas are h11 (t), h12 (t), h21 (t), and h22 (t), respectively, and the received signal received by the antenna 410_1 is received by the Rx1 (t) and the antenna 410_2. If the received signal is Rx2 (t), the following relational expression is established.

図１２に、マルチアンテナ受信装置３００の信号処理部３１１の構成を示す。分離・周波数オフセット推定・補償部４０１は、フーリエ変換後の信号３０６＿１、３０６＿２、変調信号Ａのチャネル変動推定信号３０８＿１、３０８＿２、変調信号Ｂのチャネル変動推定信号３１０＿１、３１０＿２を入力とし、（１）式についての逆行列演算（ＺＦ）を行うことで変調信号Ａと変調信号とを分離する。また分離・周波数オフセット推定・補償部４０１は、図９に示したパイロットシンボルを用いて、周波数オフセットや、デバイスの特性による歪み（振幅・位相）を推定し、推定結果に基づいて、これを補償することで、変調信号Ａの補償後のベースバンド信号４０２Ａ、変調信号Ｂの補償後のベースバンド信号４０２Ｂを得る。

FIG. 12 shows the configuration of the signal processing unit 311 of the multi-antenna receiving apparatus 300. The separation / frequency offset estimation / compensation unit 401 receives the signals 306_1 and 306_2 after Fourier transform, the channel fluctuation estimation signals 308_1 and 308_2 of the modulation signal A, and the channel fluctuation estimation signals 310_1 and 310_2 of the modulation signal B, and (1) The modulation signal A and the modulation signal are separated by performing an inverse matrix operation (ZF) on the equation. The separation / frequency offset estimation / compensation unit 401 estimates the frequency offset and distortion (amplitude / phase) due to device characteristics using the pilot symbols shown in FIG. 9, and compensates based on the estimation result. Thus, the baseband signal 402A after compensation of the modulation signal A and the baseband signal 402B after compensation of the modulation signal B are obtained.

  Soft decision calculation section 403A receives baseband signal 402A after compensation of modulated signal A and calculates a branch metric to obtain soft decision value 404A. Deinterleaving section 405A receives soft decision value 404A as input, and performs deinterleaving (the reverse process of reordering section 104A) to obtain soft decision value 406A after deinterleaving. Decoder 407A receives soft-decision value 406A after deinterleaving, and decodes this to obtain received data 408A of modulated signal A.

  Soft decision calculation section 403B, deinterleave section 405B, and decoder 407B operate in the same manner as described above to obtain received data 408B of modulated signal B.

  FIG. 13 shows the signal power-to-noise power ratio (SNR) of carriers 1 to 6 at times i, i + 1, i + 2, i + 3, i + 4, and i + 5 obtained in the receiving apparatus when the modulated signal is transmitted in the frame configuration of FIG. ) Shows an example of the relationship. As shown in FIG. 13, the SNR of the data symbol decreases as the distance from the preamble increases. This is because the frequency estimation error in the receiving apparatus and the distortion (amplitude / phase) estimation error due to device characteristics increase with time from the preamble.

  For example, as shown in FIG. 28, when interleaving is performed in one OFDM symbol and deinterleaving is performed in the receiving apparatus, data belonging to an OFDM symbol that is temporally separated from the preamble, such as times i + 4 and i + 5, is deinterleaved. However, in consideration of the phenomenon of FIG. 13, since it is composed of only data symbols having a poor SNR, it is difficult to obtain a coding gain even if error correction is performed, and the error rate characteristics deteriorate. It will be.

  In a conventional system in which each transmitting / receiving device has only one antenna, it is very easy to solve this problem. What is necessary is just to insert the symbol for frequency offset and distortion estimation, for example, a pilot symbol. At this time, since it is not necessary to increase the frequency of pilot symbol insertion so much, a decrease in transmission rate due to the insertion of the pilot symbol is small, and even if the pilot symbol is inserted, the demerit as a system is not so great.

  On the other hand, in a multi-antenna system such as a MIMO system using spatial multiplexing, a separation symbol (preamble in FIG. 9) for separating each modulation signal mixed on the transmission path is indispensable. Further, channel fluctuations h11 to h22 are estimated using the separation symbols. Factors that degrade the estimation accuracy of the channel fluctuations h11 to h22 include temporal variations in frequency offset and distortion. However, the above-described decrease in SNR cannot be prevented only by inserting pilot symbols and estimating temporal variations in frequency offset and distortion estimation. To the last, unless the estimation accuracy of the channel fluctuations h11 to h22 is ensured, the above-described decrease in SNR cannot be prevented. In order to realize this, a method of increasing the insertion frequency of the separation symbol can be considered. That is, even if the frequency of pilot symbol insertion is increased, the solution is difficult. However, since the separating symbols need to be arranged over all carriers, there is a problem that the transmission speed is significantly reduced when the frequency of inserting the separating symbols is increased. Therefore, it is important to improve the SNR while keeping the insertion frequency of the separation symbols as low as possible.

  The present embodiment proposes a multi-antenna transmission apparatus that can suppress the deterioration of error rate characteristics of data assigned to symbols at positions far away from the preamble without increasing the frequency of preamble insertion.

  In the present embodiment, the above-described problem is solved by devising the rearrangement processing of the rearrangement units 104A and 104B provided between the encoding units 102A and 102B and the modulation units 106A and 106B. The details will be described below.

  Here, rearrangement sections 104A and 104B arrange the input m-th data in the data symbol at the position of carrier p (m) on the frequency axis and the data symbol at the position of time q (m) on the time axis. Shall be rearranged. This rearrangement process is represented by π (m) = (p (m), q (m)).

  14 and 15 show an example of data rearrangement processing after encoding by the rearrangement units 104A and 104B. 14 and 15 show an example in which data is rearranged within 6 OFDM symbols as an example. Note that the preamble is omitted. 14 and 15, (1), (2), (3),... Indicate the order of data arrangement. For example, the first input data is used as the data symbol of (1). This means that the second input data is arranged in the data symbol (2).

  What is important in the rearrangement of FIGS. 14 and 15 is that the first data and the second data are arranged at data symbol positions at different times. For example, when the encoding units 102A and 102B perform block encoding processing with a block size of 6, the rearrangement units 104A and 104B convert the 6 data in the encoded block into symbols at different positions in time. assign. That is, for example, q (1) ≠ q (2) ≠ q (3) ≠ q (4) ≠ q (5) ≠ q (6), q (7) ≠ q (8) ≠ q (9) ≠ q (10) The data after block coding is assigned to symbols so that q (11) ≠ q (12).

  As a result, in the data sequence subjected to deinterleaving by the receiving device, data with a poor SNR does not continue, so that the coding gain can be obtained by performing error correction, and the error rate characteristic Deterioration can be suppressed.

  Incidentally, in consideration of SNR correlation in the frequency axis direction (SNR correlation is high between nearby carriers), in addition to the above conditions, p (1) ≠ p (2) ≠ p (3) ≠ p ( 4) The encoded data is set such that p (5) ≠ p (6), p (7) ≠ p (8) ≠ p (9) ≠ p (10) ≠ p (11) ≠ p (12). When rearranged, the deterioration of the error rate characteristic can be further suppressed.

  Thus, according to the present embodiment, rearrangement sections 104A and 104B are provided that rearrange encoded data so that encoded data in the same encoded block is allocated to a plurality of data symbols in the time direction. Thus, it can be avoided that all the data in the coding block is assigned to the data symbol located far away from the preamble. In other words, since the distance from the preamble between the respective encoded blocks can be made substantially uniform, it is possible to realize the multi-antenna transmission apparatus 100 that can suppress the deterioration of the error rate characteristics due to the distance from the preamble. In addition, the influence of notches due to fading can be reduced.

  In the present embodiment, as illustrated in FIG. 9, a frame configuration including only a preamble, a data symbol, and a pilot symbol has been described as an example. However, the present embodiment is not limited to this, and for example, control information is transmitted. May be included. In short, it is suitable to be widely applied to those in which a preamble is arranged before a data symbol.

  Further, in the configuration example illustrated in FIG. 8, the configuration in which the encoding units 102A and 102B are provided for the respective modulation signals A and B has been described, but the encoding process for both the modulation signals A and B is performed by one encoding unit. It can also be applied to a configuration that performs the above.

  FIG. 16 shows an example of the configuration. In FIG. 16, in which the same reference numerals are assigned to the parts corresponding to FIG. 8, the multi-antenna transmission apparatus 500 is different from the multi-antenna transmission apparatus 100 in that there is only one encoding unit 102 and rearrangement unit 104. is there.

  Encoding section 102 receives data 101 and control signal 116 as input, and performs encoding based on control signal 116 to obtain encoded data 103. The rearrangement unit 104 receives the encoded data 103 and the control signal 116 as input, rearranges the encoded data 103 based on the frame configuration information included in the control signal 116, and outputs the rearranged data 105A, 105B is supplied to modulators 106A and 106B, respectively.

  17, 18, and 19 show examples of data rearrangement processing after encoding by the rearrangement unit 104.

  In FIG. 17, first, the encoded 6-bit data is assigned to data symbols at different times of the modulated signal A ((1), (2), (3), (4), (5), FIG. 17). (Corresponds to (6)). Further, the encoded 6-bit data is assigned to data symbols at different times of the modulated signal B (corresponding to (7), (8), (9), (10), and (11) in FIG. 17). Next, the encoded 6-bit data is assigned to the modulation signal A. In this way, the encoded data is assigned to data symbols at different times and is alternately assigned to modulated signal A and modulated signal B. In this way, in addition to obtaining the same effect as the allocation example shown in FIGS. 14 and 15, since the modulation signal A and modulation signal B are alternately allocated, spatial diversity is achieved. A further effect that a gain can be obtained can be obtained.

  In FIG. 18, data is alternately allocated to the modulation signal A and the modulation signal B. At that time, 6-bit data obtained by extracting only odd-numbered data or 6-bit data obtained by extracting only even-numbered data is arranged in symbols at different times. For example, it is clear from the data symbols (1), (3), (5), (7), (9), and (11) of the modulation signal A. In this way, in addition to obtaining the same effect as the allocation example shown in FIGS. 14 and 15, since the modulation signal A and modulation signal B are alternately allocated, spatial diversity is achieved. A further effect that a gain can be obtained can be obtained.

  In FIG. 19, first, data is assigned to modulated signal A, and then assigned to modulated signal B. Then, with 6 bits after encoding as a unit, these are arranged in symbols at different times. By doing in this way, the effect similar to the example of allocation shown in FIG. 14, FIG. 15 can be acquired.

  FIG. 20 shows a configuration of a signal processing unit of a multi-antenna receiving apparatus that receives and demodulates a signal transmitted from multi-antenna transmitting apparatus 500 configured as shown in FIG. Here, the overall configuration of the multi-antenna receiving apparatus may be configured as shown in FIG. 10, and the signal processing unit 311 may be configured as shown in FIG.

  The signal processing unit 311 in FIG. 20, in which parts corresponding to those in FIG. 12 are denoted by the same reference numerals, is the signal processing unit in FIG. 12 except that only one deinterleave unit 405 and decoder unit 407 are provided. The configuration is the same as that of the unit 311. Deinterleaving section 405 receives soft decision value 404A of modulated signal A and soft decision value 404B of modulated signal B as input, and performs deinterleaving according to the frame configuration (reverse processing to rearrangement section 104 in FIG. 16 is performed). ) To obtain a soft decision value 406 after deinterleaving. Decoder 407 receives soft decision value 406 after deinterleaving as input, and obtains received data 408 by decoding this.

(Embodiment 3)
In the present embodiment, a specific form in the case of performing LDPC encoding in a multi-antenna transmission apparatus will be described. Furthermore, a specific form when adaptive modulation is performed will be described.

  FIG. 21 is a diagram illustrating data encoded by the rearrangement units 104A and 104B into data symbols when LDPC encoding is performed in which the block size after encoding is 980 bits by the encoding units 102A and 102B in FIG. An assignment example is shown. 980 bits in one encoded block are allocated to 980 symbols of A (1), A (2),..., A (980) of the modulated signal A. Here, (1), (2),..., (980) indicate the order of data. Similarly, 980 bits in one encoded block are assigned to 980 symbols of B (1), B (2),..., B (980) of the modulated signal B. In this way, data (bits) in one encoded block is assigned to a plurality of data symbols. Thereby, a burst error can be suppressed compared with the case where the data in one encoding block are allocated to few data symbols.

  FIG. 22 shows an example of assignment of encoded data to data symbols by the reordering unit 104 when LDPC coding with a block size of 980 bits is performed by the coding unit 102 of FIG. 980 bits in one coding block are assigned to 980 symbols of modulated signal A and modulated signal B. Here, (1), (2),..., (980) indicate the order of data. As described above, data (bits) in one encoded block is allocated to a plurality of data symbols and a plurality of antennas, so that a burst error is compared with a case where data in one encoded block is allocated to a smaller number of data symbols. Can be suppressed, and a further effect that a spatial diversity gain can be obtained can be obtained.

  Next, a mode when the present invention is applied to a multi-antenna transmission apparatus that performs adaptive modulation (that is, switches a modulation scheme) according to communication conditions will be described.

  FIG. 23 shows a configuration of a multi-antenna transmission apparatus that performs adaptive modulation. A multi-antenna transmission apparatus 600 shown in FIG. 23, in which parts corresponding to those in FIG. 8 are assigned the same reference numerals, is provided, for example, in a base station. The reception device 2303 receives the reception signal 2302 received by the antenna 2301 and performs reception processing, thereby performing communication processing information transmitted by the communication partner terminal, such as a bit error rate, packet error rate, frame error rate, and reception. Information such as the electric field strength and the multipath status is obtained, the modulation method is determined from this information, and this is output as control information 2304. The frame configuration signal generation unit 115 receives the control information 2304, determines a modulation scheme and a frame configuration based on the control information 2304, adds these as the frame configuration signal 116 to the modulation units 106A and 106B, the encoding unit 102A, 102B and the sorting units 104A and 104B.

  Reordering sections 104A and 104B change the reordering according to the modulation method, as described in the first embodiment.

  FIG. 24 shows a configuration example of a communication partner terminal that communicates with multi-antenna transmission apparatus 600. The transmission apparatus 2403 of the multi-antenna reception apparatus 700 shown in FIG. 24 with the same reference numerals assigned to the parts corresponding to FIG. 10 receives the transmission data 2402, the baseband signals 304_1 and 304_2, and the reception data 312A and 312B. The reception field strength is estimated from the band signals 304_1 and 304_2, the bit error rate, the packet error rate, and the frame error rate are obtained from the reception data 312A and 312B, and the transmission signal 2404 including these information and transmission data is formed. Is output from the antenna 2405 as a radio wave. As a result, the modulation scheme of the base station (multi-antenna transmission apparatus 600) is changed.

  Note that the method of changing the modulation method is not limited to this, and the modulation method desired by the communication partner terminal may be specified, and the base station receives the modulation signal transmitted by the communication partner terminal. However, the present invention can be similarly implemented by determining the modulation method of the modulation signal to be transmitted based on the reception state.

(Embodiment 4)
In the present embodiment, a device for assigning the last block data after LDPC encoding will be described. In FIG. 25, the vertical axis indicates the frequency, and data is transmitted using carriers 1 to n. The horizontal axis indicates time.

  In FIG. 25, it is assumed that one packet of data is initially transmitted using 16QAM. Therefore, four post-coding blocks # 1 to # 4 are transmitted with 980 symbols. Assuming that the data amount of one packet is variable, the data amount to be transmitted last does not necessarily satisfy the four encoded blocks with 16QAM.

  Therefore, in the present embodiment, when the number of encoding blocks to be transmitted last is one, as shown in FIG. 25A, BPSK is selected as the modulation scheme of the last block, and one encoding is performed with 980 symbols. Send only block # 1.

  If the number of encoding blocks to be transmitted at the end is more than one and not more than two, QPSK is selected as the modulation scheme of the last block as shown in FIG. 25 (b), and two encodings are performed with 980 symbols. Blocks # 1 and # 2 are transmitted. In this case, the rearrangement as described with reference to FIG. 4B, FIG. 5B, FIG. 6B, or FIG.

  When the number of encoded blocks to be transmitted at the end is more than two, 16QAM is selected as the modulation scheme of the last block as shown in FIG. 25C, and for example, four encoded blocks # 1 with 980 symbols are selected. Send ~ # 4. In this case, the rearrangement as described in FIG. 4C, FIG. 5C, FIG. 6C, or FIG. 7C may be performed.

  By transmitting in this way, one encoded block data is always transmitted by 980 symbols, so that the influence of fading notches can be reduced and reception quality is improved.

  As another allocation method, 16QAM may be selected regardless of the number of encoded blocks, and dummy data of “0”, for example, may be transmitted for all missing data. Even if transmission is performed in this way, since one encoded block data is always transmitted with 980 symbols, the influence of fading notches can be reduced, and reception quality can be improved.

  The above operation is very important in order to make the reception quality as uniform as possible when performing packet communication. That is, if the data of the last coding block is transmitted with a smaller number of symbols than 980 symbols, the error rate characteristic of the last coding block is degraded, and the probability of occurrence of a packet error increases. According to the method of the present embodiment, this can be effectively avoided.

(Comparative example)
Here, with reference to FIG. 26, a conventional allocation method and its drawbacks will be described as a comparative example of the method for uniformly allocating encoded block data to a plurality of symbols according to the present invention.

  FIG. 26A shows the state of the received electric field strength in the 980 symbol section as an example of the relationship between the time and the received electric field strength as the communication status.

  FIG. 26B shows a frame configuration example when the modulation method is BPSK. In FIG. 26B, for example, a case of a multi-carrier transmission scheme using carrier 1 to carrier n such as OFDM is shown as an example. Therefore, the vertical axis is the frequency axis, and carrier 1 to carrier n exist. When the modulation method is BPSK, 980 symbols are required to transmit one post-encoding block (block # 1) as shown in FIG.

  On the other hand, when the modulation scheme is 16QAM, 4 bits can be transmitted in one symbol in 16QAM, so 245 symbols are required to transmit one post-encoding block. Therefore, if 980 symbols are used, four blocks of block # 1, block # 2, block # 3, and block # 4 can be transmitted.

  Then, conventionally, as shown in FIG. 26 (c), as in BPSK, the block # 1 symbol, the block # 2 symbol, the block # 3 symbol, and the block # 4 symbol are assigned in this order in the time direction. It is common.

  At this time, when BPSK is used as shown in FIG. 26 (b), even when the communication situation is as shown in FIG. However, if decoding is performed in units of coding blocks, there is a high possibility that errors will be corrected due to the influence of data with good reception field strength.

  In contrast, when 16QAM is used as shown in FIG. 26 (c), block # 1 and block # 3 have good reception quality because the reception field strength is good, but block # 2 and block # 3. No. 4 has poor reception quality due to poor reception field strength. As described above, as the number of modulation multi-levels of the modulation scheme increases, the number of symbols required for one coding block decreases, and therefore, it is easily affected by the notch of the received electric field strength due to fading. That is, it becomes easy for the reception quality to deteriorate due to the notch.

  The transmission apparatus of the present invention effectively solves such a problem without changing the code length (block size) as described in the above embodiment.

(Other embodiments)
In the first embodiment described above, the description is given on the assumption that one encoding unit 11 is used. As another embodiment, the system has an encoding rate of R = 1/2, 2/3, and a block. Even when a code having a length of 980 bits is supported, the above-described embodiment can be similarly implemented if the codes are implemented separately at coding rates R = 1/2 and 2/3. In addition, even when the system supports codes with a coding rate R = 1/2, 2/3, a block length of 980, and 1960 bits, in each case, if the implementation is performed separately, the above implementation It can be implemented in the same way as the form.

  In the above-described Embodiments 2 to 4, the case of a MIMO system using spatial multiplexing in which the multi-antenna transmission apparatus and the multi-antenna reception apparatus each have two antennas has been described. However, the present invention is not limited to this. However, the same can be applied to the case where the number of antennas increases and the number of modulated signals to be transmitted increases. Even when applied to a system using a spread spectrum communication system, the same effect can be obtained.

  Moreover, the multi-antenna transmission apparatus of the present invention is not limited to the configuration shown in Embodiment 2, and can be applied to, for example, a MIMO system using an eigenmode. The eigenmode communication method will be described in detail with reference to FIG.

  In a MIMO system, when channel state information (CSI) is known not only at the receiving station but also at the transmitting station, the transmitting station is vectorized using a channel signature vector of transmission. The received signal is transmitted from the transmitting array antenna to the receiving station, and the receiving station detects the transmitted signal using the received channel signature vector associated with the transmitted channel signature vector from the received signal of the receiving array antenna. A demodulating communication method can be realized.

  In particular, as a communication mode in which a plurality of channels are formed in a communication space and a signal is multiplexed and transmitted, there is an eigenmode (eigenmode) using a singular vector or eigenvector of a channel matrix. This eigenmode is a method of using these singular vectors and eigenvectors as the aforementioned channel signature vector. Here, the channel matrix is a matrix whose elements are complex channel coefficients of a combination of all or a part of each antenna element of the transmitting array antenna and each antenna element of the receiving array antenna.

  As a method for the transmitting station to obtain downlink channel state information, in TDD using the same frequency carrier in the uplink and downlink of the radio channel, the uplink from the receiving station is used due to channel reciprocity. It is possible to estimate or measure channel state information at the transmitting station. On the other hand, in FDD that uses different frequency carriers for uplink and downlink, downlink channel state information is estimated or measured at the receiving station, and the result is reported to the transmitting station. Accurate CSI can be obtained.

  The eigenmode is characterized in that the channel capacity of the MIMO system can be maximized, particularly when the radio channel of the MIMO system can be handled as a narrow-band flat fading process. For example, in a radio communication system adopting OFDM, it is common to design such that each OFDM subcarrier is in a flat fading process by inserting a guard interval in order to remove intersymbol interference caused by multipath delay waves. . Therefore, when transmitting an OFDM signal in a MIMO system, it is possible to spatially multiplex and transmit a plurality of signals on each subcarrier, for example, by using an eigenmode.

  As a communication method using the MIMO system, there are several methods for making the channel state information of the radio channel known only at the receiving station, compared to the eigenmode in which the downlink channel state information is known at the transmitting station and the receiving station. Proposed. For example, BLAST is known as a method for spatially multiplexing and transmitting signals, which has the same purpose as the eigenmode. As a method for obtaining the space diversity effect of the antenna, not at the sacrifice of signal multiplicity, that is, to increase the capacity, for example, transmission diversity using a space-time code is known. The eigenmode is a beam space mode in which a signal is vectorized by a transmission array antenna and transmitted, in other words, a signal is mapped to a beam space and then transmitted. On the other hand, BLAST and transmission diversity transmit a signal to an antenna. Since it is mapped to an element (antenna element), it is considered to be an antenna element mode.

  FIG. 27 is an example of the configuration of a transmitter / receiver for eigenmode communication. A transmission channel analysis unit 2607 calculates channel signature vectors of a plurality of transmissions to form a multiplexed channel based on channel state information that is an estimation result of a propagation channel between the transmission station and the reception station. The channel matrix formed by the state information is based on SVD (SVD: Singular Value Decomposition), eigenvalues (for example, λA, λB, λC,..., ΛX), and eigenpaths (for example, path A, path B, Path C,..., Path X) is obtained and output as control information 2608.

  The transmission station receives a transmission digital signal and control information 2608 as input from the multiplex frame generation unit 2601, generates a plurality of transmission frames for mapping to the multiplexed channel, and transmits a transmission digital signal 2602 A for channel A and a transmission digital for channel B. Signal 2602B,..., Channel X transmission digital signal 2602X is output.

  Encoding / reordering / modulating section 2603A receives channel A transmission digital signal 2602A and control information 2608 as input, determines the coding rate and modulation method based on control information 2608, and outputs channel A baseband signal 2604A. To do. The same operation is performed for channel B to channel X, and baseband signal 2604X of channel X is obtained from baseband signal 2604B of channel B. In FIG. 27, in order to simplify the drawing, the encoding / reordering / modulating unit is shown as one block. However, in actuality, the configuration is the same as in the first to third embodiments. The block encoded data is rearranged and supplied to the modulation unit so that one intra-block encoded data is assigned to a plurality of data symbols by the rearrangement unit.

  Vector multiplexing section 2605 receives aceband signals 2604A to 2604X from channel A to channel X and control information 2608 as input, and multiplies channel signature vectors 2604A to 2604X by channel signature vectors individually and combines them. After that, the data is transmitted from the transmission array antenna 2606 to the receiving station.

  In the receiving station, the receiving channel analyzing unit 2615 uses a plurality of receiving channels in order to separate multiplexed transmission signals based on channel state information that is an estimation result of a propagation channel between the transmitting station and the receiving station in advance. A signature vector is calculated. Multiplex separation section 2610 receives a reception signal received by reception array antenna 2609 and receives a plurality of reception signals obtained by multiplying each channel signature vector, that is, channel A reception signal 2611A to channel X reception 2611X. Generate.

  Decoding section 2612A receives channel A reception signal 2611A and transmission method information 2618 as input, performs decoding based on transmission method information 2618 (modulation scheme and coding rate information), and outputs channel A digital signal 2613A. . The same operation is performed for channel B to channel X, and channel X digital signal 2613X is obtained from channel B digital signal 2613B.

  Transmission method information detection section 2617 receives channel A digital signal 2613A as input, extracts the modulation signal transmission method of each channel, for example, modulation scheme and coding rate information, and outputs transmission method information 2618.

  Received data combining section 2614 receives channel A to channel X digital signals 2613A to 2613X and transmission method information 2618 as input, and generates a received digital signal.

  The present invention has an effect that a burst error can be suppressed with a relatively simple configuration without changing the block size of the encoded block even when the modulation multi-level number is increased. For example, a block code such as an LDPC code is used. Thus, the present invention can be widely applied to a transmission apparatus and a multi-antenna transmission apparatus that encode and transmit transmission data.

The block diagram which shows the structure of the transmitter which concerns on Embodiment 1 of this invention. The figure which uses for description of the LDPC encoding process by an encoding part Diagram for explaining each modulation method The figure which shows the allocation to each symbol of the LDPC coding data by a rearrangement part The figure which shows the allocation to each symbol of the LDPC coding data by a rearrangement part The figure which shows the allocation to each symbol of the LDPC coding data by a rearrangement part The figure which shows the allocation to each symbol of the LDPC coding data by a rearrangement part Block diagram showing a configuration of a multi-antenna transmission apparatus according to a second embodiment The figure which shows the example of a frame structure of the modulation signal transmitted from each antenna of a multi-antenna transmission apparatus FIG. 3 is a block diagram illustrating a configuration of a multi-antenna reception apparatus according to a second embodiment. The figure which shows the communication model between a multi-antenna transmission apparatus and a multi-antenna reception apparatus Block diagram showing the configuration of the signal processing unit of the multi-antenna receiving apparatus The figure which shows the relationship of the SNR characteristic of the signal of each time in a receiver. The figure which shows the rearrangement process example of the data after encoding The figure which shows the rearrangement process example of the data after encoding FIG. 9 is a block diagram illustrating another configuration example of the multi-antenna transmission apparatus according to the second embodiment. The figure which shows the rearrangement process example of the data after encoding The figure which shows the rearrangement process example of the data after encoding The figure which shows the rearrangement process example of the data after encoding Block diagram showing the configuration of the signal processor The figure which shows the rearrangement process example of LDPC encoding data The figure which shows the rearrangement process example of LDPC encoding data Block diagram showing the configuration of a multi-antenna transmission apparatus that performs adaptive modulation The block diagram which shows the structure of the multi-antenna receiver which receives an adaptive modulation signal The figure which serves for description of Embodiment 4 As a comparative example, a diagram for explaining degradation of reception quality characteristics due to communication conditions when a conventional coding block allocation method is applied The block diagram which shows the structural example in the case of applying this invention to the system using an eigenmode The figure which shows the example of a frame structure of the transmission signal by the conventional multi-antenna transmission apparatus

Explanation of symbols

DESCRIPTION OF SYMBOLS 10 Transmitting device 11, 102A, 102B, 102 Encoding unit 12, 104A, 104B, 104 Rearrangement unit 15, 106A, 106B Modulating unit 100, 500, 600 Multi-antenna transmitting device

Claims (6)

Encoding means for performing error correction encoding processing on transmission data composed of a plurality of bits to form a plurality of error correction encoded data each composed of a plurality of bits;
One symbol is to consist collects bits belonging to one of the plurality of the error correction coded data, and rearranging means for rearranging the bits belonging to said plurality of error correction coded data,
Modulation means for outputting a baseband signal corresponding to the symbol from the rearranged error correction encoded data;
Comprising
A plurality of bits constituting the symbol, the two bits to be extracted arbitrarily, a bit belonging to the erroneous Ri correction coded data that different from each other,
The permutation means inputs the data of one line constituted by the plurality of mis-Ri correction coded data, which rearranges the bits included in the data of one line, the adjacent symbol, the symbol the same bit number of the bit number indicating the signal point arrangement, reordering to allocate bits belonging to mis Ri correction coded data that different from each other, said one error and the number of bits constituting one error correction coded data A modulator characterized in that the number of symbols used to transmit correction encoded data is the same . The modulation means can use a plurality of modulation schemes, and in any of the modulation schemes, two bits arbitrarily extracted from a plurality of bits constituting a formed symbol are different from each other in the error correction. The modulator according to claim 1, wherein the modulator belongs to encoded data. The modulator according to claim 1, wherein the baseband signal is a signal formed using an OFDM system. Applying error correction coding processing to transmission data composed of a plurality of bits to form a plurality of error correction coding data each composed of a plurality of bits,
One symbol is to consist collects bits belonging to one of the plurality of the error correction coded data, reordering the bits belonging to said plurality of error correction coded data,
A modulation method for outputting a baseband signal corresponding to the symbol from the rearranged error correction encoded data,
A plurality of bits constituting the symbol, the two bits to be extracted arbitrarily, a bit belonging to the erroneous Ri correction coded data that different from each other,
Reordering the bits, said plurality of mis-Ri data for one line constituted by correction coded data as input, which rearranges the bits included in the data of one line, the adjacent symbol, the symbol the same bit number, different that erroneous Ri rearranged to assign bits belonging to the correction coded data, the number of bits constituting one error correction coded data and the one mutually among the signal point bit number showing the arrangement of A modulation method, wherein the number of symbols used to transmit error correction encoded data is the same . A plurality of modulation schemes can be used, and in any of the modulation schemes, two bits arbitrarily extracted from a plurality of bits constituting a formed symbol belong to different error correction encoded data The modulation method according to claim 4, wherein the modulation method is a bit. The modulation method according to claim 4 or 5, wherein the baseband signal is a signal formed using an OFDM system.

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