JP4510870B2 - Wireless communication method and wireless communication device - Google Patents

Description

  The present invention relates to a wireless communication method and a wireless communication apparatus.

  Conventionally, as in a communication method called MIMO (Multiple-Input Multiple-Output), a plurality of series of transmission data is modulated, and each modulation data is simultaneously transmitted from a plurality of antennas, thereby increasing the data communication speed. There is what I did. On the receiving side, transmission signals from a plurality of antennas are received by a plurality of antennas.

  Here, since the received signal obtained by each receiving antenna is a mixture of a plurality of modulated signals in the propagation space, in order to restore the data corresponding to each modulated signal, the propagation path of each modulated signal is used. Must be estimated (hereinafter referred to as channel fluctuation). For this reason, the transmitting apparatus inserts a known signal such as a pilot symbol in the modulated signal in advance, and the receiving apparatus performs channel fluctuation in the propagation space between each transmitting antenna and each receiving antenna based on the known signal inserted in the modulated signal. presume. Then, each modulation signal is demodulated using this channel fluctuation estimated value.

  As one of the methods, there is a method of separating each modulated signal by performing an inverse matrix operation of a matrix having a channel fluctuation estimated value as an element. As another method, a candidate signal point position is obtained using a channel fluctuation estimated value, and maximum likelihood determination (MLD: Maximum Likelihood Detection) is performed between the candidate signal point position and the received signal point position. There is a method for restoring data transmitted by a modulated signal.

  A communication technique using such a multi-antenna is disclosed in Non-Patent Document 1, for example. The contents disclosed in Non-Patent Document 1 will be briefly described below with reference to FIG. The multi-antenna transmission apparatus 1 inputs the transmission signal A and the transmission signal B to the modulation signal generation unit 3. The modulation signal generation unit 3 performs digital modulation processing such as QPSK (Quadrature Phase Shift Keying) and 16QAM (Quadrature Amplitude Modulation) on the transmission signals A and B, and the baseband signals 4 and 5 obtained thereby are transmitted to the radio unit. 6 to send. The radio unit 6 performs radio processing such as up-conversion and amplification on the baseband signals 4 and 5, and sends the modulated signals 7 and 8 obtained thereby to the antennas 9 and 10. In this way, the multi-antenna transmission apparatus 1 transmits the modulated signal 7 of the transmission signal A from the antenna 9 and simultaneously transmits the modulated signal 8 of the transmission signal B from the antenna 10.

  The multi-antenna reception apparatus 2 inputs the reception signal 12 received by the antenna 11 to the radio unit 13 and inputs the reception signal 16 received by the antenna 15 to the radio unit 17. Radio units 13 and 17 perform radio processing such as down-conversion on received signals 12 and 16, and send baseband signals 14 and 18 obtained thereby to demodulation unit 19.

  The demodulator 19 detects the baseband signals 14 and 18 to obtain a reception digital signal 20 of the transmission signal A and a reception digital signal 21 of the transmission signal B. At this time, in Non-Patent Document 1, the demodulator 19 performs inverse matrix calculation of the channel estimation matrix to obtain the received digital signals 20 and 21, and performs maximum likelihood determination (MLD) to obtain the received digital signals 20 and 21. The method of obtaining is described.

  Conventionally, as a transmission method using a plurality of antennas, a space-time block code (STBC) is transmitted as disclosed in Non-Patent Document 2, so that the quality (error rate characteristic Techniques for realizing (good) data transmission are known. The contents disclosed in Non-Patent Document 2 will be described below with reference to the drawings.

  As shown in FIG. 103, the transmission apparatus has a plurality of antennas AN1 and AN2, and transmits signals simultaneously from the antennas AN1 and AN2. The receiving device receives a plurality of signals transmitted at the same time by the antenna AN3.

  FIG. 104 shows a frame configuration of signals transmitted from the antennas AN1 and AN2. The transmission signal A is transmitted from the antenna AN1, and at the same time, the transmission signal B is transmitted from the antenna AN2. The transmission signal A and the transmission signal B are composed of symbol blocks in which the same symbols are arranged a plurality of times so that a coding gain and a diversity gain can be obtained.

This will be described in more detail. In FIG. 104, S1 and S2 indicate different symbols, and the complex conjugate is indicated by “*”. In space-time block coding, at time point i, the symbol S1 is transmitted from the first antenna AN1 and at the same time the symbol S2 is transmitted from the second antenna AN2, and at the subsequent time point i + 1, the symbol -S2 ^{* is} transmitted from the first antenna AN1 ^. At the same time as transmitting the symbol S1 ^* from the second antenna AN2.

  In the antenna AN3 of the receiving apparatus, a transmission signal A that has received the transmission path fluctuation h1 (t) between the antenna AN1 and the antenna AN3, and a transmission signal B that has received the transmission path fluctuation h2 (t) between the antenna AN2 and the antenna AN3. Are combined.

  The receiving apparatus estimates the transmission path fluctuations h1 (t) and h2 (t), and uses the estimated values to separate the original transmission signal A and the transmission signal B from the combined reception signal, and then each symbol. Is to be demodulated.

At this time, when a space-time block coded signal as shown in FIG. 104 is used, the symbols S1 and S2 can be combined at the maximum ratio regardless of the transmission path fluctuations h1 (t) and h2 (t) during signal separation. As a result, a large coding gain and diversity gain can be obtained. As a result, reception quality, that is, error rate characteristics can be improved.
“Multiple-antenna diversity techniques for transmission over fading channels” IEEE WCNC 1999, pp.1038-1042, Sep. 1999. “Space-Time Block Codes from Orthogonal Design” IEEE Transactions on Information Theory, pp.1456-1467, vol.45, no.5, July 1999

  By the way, in the system using the multi-antenna as in Non-Patent Document 2, although the data communication speed is increased, there is a problem that the configuration of the receiving apparatus is particularly complicated. In particular, in the method of obtaining data corresponding to each modulation signal by performing maximum likelihood determination (MLD), the number of operations required for maximum likelihood determination between candidate signal points and reception points increases, so that the circuit scale increases. End up.

  Specifically, considering the case where the number of transmitting antennas is 2 and the number of receiving antennas is 2, when 4Q4 = 16 candidate signal points exist when a modulated signal subjected to QPSK is transmitted from each antenna become. Further, when a modulated signal subjected to 16QAM is transmitted from each antenna, there are 16 × 16 = 256 candidate signal points. When maximum likelihood determination (MLD) is performed, it is necessary to calculate the distance between an actual reception point and all these candidate signals, which requires enormous calculation and leads to an increase in circuit scale.

  On the other hand, in the method of performing determination after separating each modulated signal from the received signal using the inverse matrix of the channel estimation matrix, the number of operations is smaller than the method of performing maximum likelihood determination (MLD). Although the circuit scale is small, the error rate characteristic is lowered depending on the radio wave propagation environment. As a result, there is a drawback that the error rate characteristic of the received data is deteriorated. When the error rate characteristic is lowered, the substantial data communication speed is lowered.

Further, when a space-time block coded signal as in Non-Patent Document 2 is used, although reception quality (error rate characteristics) is improved, there is a drawback that transmission efficiency is lowered. That is, S1 ^* and -S2 ^* transmitted at time point i + 1 are demodulated as S1 and S2 in the receiving apparatus, and thus are substantially the same information as S1 and S2 transmitted at time point i. For this reason, the same information is transmitted twice, and the data transmission efficiency is reduced accordingly.

  For example, in a general multi-antenna communication system, symbols S3 and S4 that are different from symbols S1 and S2 are transmitted at time point i + 1, so that four symbols S1 to S4 can be transmitted in a period from time point i to time point i + 1. it can. That is, simply considering, when the space-time block coding technique is used, the data transmission efficiency is reduced to half that of general multi-antenna communication.

  The object of the present invention is to enable communication that could not be achieved by the prior art, such as that it is possible to obtain reception quality close to the maximum ratio combining without reducing the data transmission efficiency as compared with the transmission method using STBC. And providing a multi-antenna communication system capable of achieving this with a relatively small number of operations.

  The present invention also realizes retransmission suitable for a multi-antenna communication system.

One aspect of the wireless communication method of the present invention is a wireless transmission method for a wireless communication apparatus having a plurality of antennas, the step of transmitting a plurality of modulated signals having different transmission data to the communication partner using the plurality of antennas. And (i) a first retransmission method for retransmitting one modulated signal using one antenna, or (ii) using weights calculated from information shared by communication with the communication partner, A signal obtained by weighting and synthesizing a modulation signal for retransmission that is the same as or less than the number of a plurality of modulation signals is retransmitted to the same frequency band using the same or less number of antennas as the plurality of antennas. And a step of transmitting retransmission data to the communication partner using the selected retransmission method.

One aspect of the wireless communication apparatus of the present invention detects a retransmission request for transmission data transmitted to a communication partner using a plurality of antennas transmitting modulated signals and a plurality of modulation signals having different transmission data from the plurality of antennas. And (i) a first retransmission frame configuration for retransmitting one modulated signal using one antenna based on the retransmission request, or (ii) communication with the communication partner Using the weights calculated from the information shared by the number of the plurality of modulation signals, the number of the plurality of antennas in the same frequency band, a signal obtained by weighting and combining the modulation signals for retransmission with the same number or a smaller number than the number of the plurality of modulation signals. A frame configuration signal for selecting a second retransmission frame configuration to be retransmitted using the same or a smaller number of antennas and outputting a frame configuration signal indicating the selected retransmission frame configuration Based on the retransmission request, the wireless communication device retransmits the selected retransmission data based on the frame configuration signal, a data selection unit that selects retransmission data to the communication partner, based on the retransmission request And a transmission unit.

  According to the present invention, retransmission suitable for a multi-antenna communication system can be realized.

  To receive a plurality of modulated signals simultaneously transmitted from the multi-antenna transmitting apparatus and multiplexed on the propagation path by the multi-antenna receiving apparatus and perform signal point determination of each modulated signal to obtain data with good error rate characteristics, A huge amount of calculation is required. In particular, the greater the number of channels (the number of antennas) and the greater the number of modulation levels, the greater the number of calculations.

  One of the features of the present invention is that candidate signal points used when determining reception points of modulated signals to obtain received data are reduced by using determination values of other modulation signals other than the self-modulated signals. The determination (main determination) of the self-modulated signal is performed using the candidate signal points.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings.

(Embodiment 1)
FIG. 1 shows an overall configuration of a multi-antenna communication system described in this embodiment. In this embodiment, in order to simplify the description, a case where there are two transmission antennas and two reception antennas will be described, but M (M ≧ 2) transmission antennas and N (N ≧ 2) It can be applied to a multi-antenna system having a book receiving antenna.

  The multi-antenna transmission apparatus 110 of the multi-antenna communication system 100 obtains modulation signals Ta and Tb by performing predetermined modulation processing and conversion processing to radio frequencies on the transmission digital signals TA and TB in the transmission unit 111, This is transmitted from each antenna AN1, AN2. Multi-antenna receiving apparatus 120 inputs received signals R 1 and R 2 received by antennas AN 3 and AN 4 to receiving section 121. The receiving unit 121 performs reception processing on the reception signals R1 and R2 to obtain reception data RA and RB corresponding to the transmission digital signals TA and TB.

  Here, the modulation signal Ta transmitted from the antenna AN1 is received by the antennas AN3 and AN4 after receiving the channel fluctuations h11 (t) and h12 (t). The modulated signal Tb transmitted from the antenna AN2 is received by the antennas AN3 and AN4 after receiving the channel fluctuations h21 (t) and h22 (t).

Therefore, using the time parameter t, the signal transmitted from the antenna AN1 is Ta (t), the signal transmitted from the antenna AN2 is Tb (t), the signal received by the receiving antenna AN3 is R1 (t), and received. If the signal received by the antenna AN4 is R2 (t), the following relational expression is established.

  FIG. 2 shows the configuration of multi-antenna transmission apparatus 110. Multi-antenna transmission apparatus 110 inputs transmission digital signals TA and TB to encoding sections 201A and 201B. Encoding sections 201A and 201B perform encoded encoding processing S1A and S1B by performing convolutional encoding processing on transmission digital signals TA and TB according to frame configuration signal S10 from frame configuration signal generation section 210, and modulate this. The data is sent to the sections 202A and 202B.

  The modulation units 202A and 202B perform modulation processing such as QPSK and 16QAM on the encoded data S1A and S1B, and insert a channel estimation symbol at a timing according to the frame configuration signal S10. S2A and S2B are formed and sent to the diffusion units 203A and 203B. FIG. 3 shows a frame configuration example of each baseband signal.

  Spreading sections 203A and 203B obtain baseband signals S3A and S3B that are spread by multiplying the baseband signal by a spreading code, and send this to radio sections 204A and 204B. Note that different spreading codes are used in the spreading unit 203A and the spreading unit 203B. Radio sections 204A and 204B perform modulation processing Ta and Tb by performing radio processing such as up-conversion and amplification on the spread baseband signals S3A and S3B, and supply them to antennas AN1 and AN2.

  Thus, different modulation signals Ta and Tb convolutionally encoded in the time axis direction are transmitted simultaneously from the antennas AN1 and AN2.

  FIG. 4 shows the overall configuration of multi-antenna receiving apparatus 120. Multi-antenna receiving apparatus 120 supplies reception signals R1 and R2 received by antennas AN3 and AN4 to radio sections 401-1 and 401-2, respectively. Radio sections 401-1 and 401-2 perform baseband signals R <b> 1-1 and R <b> 2-1 by performing radio processing such as down-conversion and quadrature demodulation on the received signal, and obtain baseband signals R <b> 1-1 and R <b> 2-1. , 402-2.

  The despreading unit 402-1 performs a despreading process on the baseband signal R1-1 using the same spreading code as the spreading code used in the spreading unit 203A and the spreading unit 203B in FIG. A baseband signal R1-2 is obtained and transmitted to the channel fluctuation estimation unit 403-1A of the modulation signal A, the channel fluctuation estimation unit 403-1B of the modulation signal B, and the signal processing unit 404.

  Similarly, the despreading unit 402-2 performs despreading on the baseband signal R2-1 by performing a despreading process using the same spreading code as the spreading code used in the spreading unit 203A and the spreading unit 203B in FIG. The spread baseband signal R2-2 is obtained and transmitted to the channel fluctuation estimation unit 403-2A of the modulation signal A, the channel fluctuation estimation unit 403-2B of the modulation signal B, and the signal processing unit 404.

  Channel fluctuation estimation section 403-1A for modulated signal A obtains channel fluctuation estimated value h11 by estimating channel fluctuation of modulated signal A (modulated signal Ta transmitted from antenna AN1) based on the channel estimation symbol. Thereby, the channel fluctuation between the antenna AN1 and the antenna AN3 is estimated. Channel fluctuation estimation section 403-1B of modulated signal B estimates channel fluctuation of modulated signal B (modulated signal Tb transmitted from antenna AN2) based on the channel estimation symbol to obtain channel fluctuation estimated value h21. Thereby, the channel fluctuation between the antenna AN2 and the antenna AN3 is estimated.

  Similarly, channel fluctuation estimation section 403-2A of modulated signal A obtains channel fluctuation estimated value h12 by estimating the channel fluctuation of modulated signal A (modulated signal Ta transmitted from antenna AN1) based on the channel estimation symbol. . Thereby, the channel fluctuation between the antenna AN1 and the antenna AN4 is estimated. Channel fluctuation estimation section 403-2B of modulated signal B obtains channel fluctuation estimated value h22 by estimating the channel fluctuation of modulated signal B (modulated signal Tb transmitted from antenna AN2) based on the channel estimation symbol. Thereby, the channel fluctuation between the antenna AN2 and the antenna AN4 is estimated.

  The signal processing unit 404 receives channel fluctuation estimated values h11, h21, h12, and h22 in addition to the despread baseband signals R1-2 and R2-2, and the channel fluctuation estimated values h11, h21, h12, and h22. Is used to decode the baseband signals R1-2 and R2-2, detect them, etc., thereby obtaining the reception data RA and RB corresponding to the transmission digital signals TA and TB.

  FIG. 5 shows the configuration of the signal processing unit 404 of the present embodiment. The signal processing unit 404 inputs the baseband signals R1-2 and R2-2 and the channel fluctuation estimation values h11, h21, h12, and h22 to the separation unit 501.

  Separation section 501 applies baseband signals R1-2, R2-2 and channel fluctuation estimated values h11, h21, h12, h22 to equation (1) and performs inverse matrix operation of equation (1). Then, an estimated baseband signal 502 of the transmission digital signal TA and an estimated baseband signal 505 of the transmission digital signal TB are obtained. As described above, since the separation unit 501 performs signal separation by inverse matrix calculation instead of performing maximum likelihood determination (MLD), signal separation can be performed with a smaller circuit scale than when maximum likelihood determination is performed. it can. Separating section 501 sends estimated baseband signal 502 of transmission digital signal TA to soft decision section 503 and sends estimated baseband signal 505 of transmission digital signal TB to soft decision section 506.

  Soft decision sections 503 and 506 obtain soft decision values of estimated baseband signals 502 and 505, respectively, and then apply error correction processing to the soft decision values to obtain decision values 504 and 507 made of digital data. obtain. Determination value 504 obtained by soft decision section 503 is sent to signal point reduction sections 514 and 516. The determination value 507 obtained by the soft decision unit 506 is sent to the signal point reduction units 508 and 510.

  FIG. 6 shows the configuration of the soft decision units 503 and 506. Since the configurations of the soft decision unit 503 and the soft decision unit 506 are the same, only the configuration of the soft decision unit 503 will be described here. Soft decision section 503 inputs estimated baseband signal 502 to soft decision value calculation section 601. The soft decision value calculation unit 601 calculates the data series 602 of the estimated baseband signal 502 by obtaining the branch metric and path metric of the estimated baseband signal 502, and sends this data series 602 to the decision unit 603. The determination unit 603 performs error correction processing on the data series 602 and outputs the error-corrected data as a determination value 504.

  The processing in the soft decision units 503 and 506 will be specifically described with reference to FIG. FIG. 7 shows an example of signal point arrangement when the transmission digital signals TA and TB are QPSK modulated. In the figure, reference numeral 701 denotes a received signal point, which corresponds to the estimated baseband signals 502 and 505. The soft decision units 503 and 506 obtain, for example, the square of the Euclidean distance between the reception signal point 701 and the QPSK signal point in FIG. 7 and use this value as a branch metric, and obtain the path metric using this branch metric. When a convolutional code is used, decoding is performed according to, for example, the Viterbi algorithm, and a determination value 504 for the transmission digital signal TA and a determination value 507 for the transmission digital signal TB are obtained.

  In addition to this configuration, the signal processing unit 404 includes signal point reduction units 508 and 510 for the modulation signal A and signal point reduction units 514 and 516 for the modulation signal B.

  Signal point reduction sections 508 and 510 for modulated signal A receive the determination value for modulated signal B obtained by soft decision section 506. Further, the signal point reduction unit 508 receives the channel fluctuation value h12 of the modulation signal A and the channel fluctuation value h22 of the modulation signal B obtained based on the reception signal of one receiving antenna AN4, and the signal point reduction unit 510. Is input with the channel fluctuation value h11 of the modulated signal A and the channel fluctuation value h21 of the modulated signal B obtained based on the received signal of the other receiving antenna AN3.

  The signal point reduction unit 508 first estimates 16 candidate signal points 801 to 816 based on the channel fluctuation value h12 of the modulation signal A and the channel fluctuation value h22 of the modulation signal B as shown in FIG. Next, the signal point reduction unit 508 uses the determination value 507 of the modulation signal B obtained by the soft decision unit 506 to narrow down the number of candidate signal points to four as shown in FIG. FIG. 9 shows an example of candidate signal point reduction when the determination value 507 of the modulation signal B is (0, 0), that is, the two bits transmitted by the modulation signal B are determined to be (0, 0). It is. Then, the signal point reduction unit 508 sends information on the signal points 801, 806, 811, and 816 to the soft decision unit 512 as signal point information 509.

  Similarly, the signal point reduction unit 510 estimates 16 candidate signal points 801 to 816 based on the channel fluctuation value h11 of the modulation signal A and the channel fluctuation value h21 of the modulation signal B, and then obtains them by the soft decision unit 506. The number of candidate signal points is reduced to 4 using the determined determination value 507 of the modulated signal B, and information on the signal points of the 4 points is sent to the soft decision unit 512 as signal point information 511.

  The signal point reduction units 514 and 516 for the modulation signal B receive the determination value 504 for the modulation signal A obtained by the soft decision unit 503. Further, the signal point reduction unit 514 receives the channel fluctuation value h12 of the modulated signal A and the channel fluctuation value h22 of the modulation signal B obtained based on the received signal of one receiving antenna AN4, and the signal point reduction unit 516. Is input with the channel fluctuation value h11 of the modulated signal A and the channel fluctuation value h21 of the modulated signal B obtained based on the received signal of the other receiving antenna AN3.

  The signal point reduction unit 514 first estimates 16 candidate signal points 801 to 816 based on the channel fluctuation value h12 of the modulation signal A and the channel fluctuation value h22 of the modulation signal B as shown in FIG. Next, the signal point reduction unit 514 uses the determination value 504 of the modulation signal A obtained by the soft determination unit 503 to narrow down the number of candidate signal points to four as shown in FIG. FIG. 10 shows an example of candidate signal point reduction when the determination value 504 of the modulation signal A is (1, 0), that is, the two bits transmitted by the modulation signal A are determined as (1, 0). It is. Then, the signal point reduction unit 514 sends information on the signal points 805, 806, 807, and 808 as signal point information 515 to the soft decision unit 518.

  Similarly, the signal point reduction unit 516 estimates 16 candidate signal points 801 to 816 based on the channel fluctuation value h11 of the modulation signal A and the channel fluctuation value h21 of the modulation signal B, and then obtains them by the soft decision unit 503. The number of candidate signal points is reduced to 4 using the determination value 504 of the modulated signal A, and the information of the 4 signal points is sent to the soft decision unit 518 as signal point information 517.

  As described above, in multi-antenna receiving apparatus 120 according to the present embodiment, separation section 501 that separates modulated signals A and B by inverse matrix calculation of the channel variation matrix, and soft decision on separated modulated signals 502 and 505 are performed. In addition to the soft decision units 503 and 506, signal point reduction units 508, 510, 514, and 516 corresponding to the modulation signals A and B are provided, and the signal point reduction units 508, 510, 514, and 516 receive the self-modulation signals. The number of candidate signal points for the self-modulated signal is reduced using the soft decision values 507 and 504 of the other modulation signals.

  That is, each of the modulation signals A and B is provisionally determined by the separation unit 501 and the soft decision units 503 and 506, and the candidate signal points are reduced based on the provisional determination results 507 and 504 by the signal point reduction units 508, 510, 514, and 516. To do.

  The soft decision units 512 and 518 correspond to the transmission digital signals TA and TB by making soft decisions on the baseband signals R1-2 and R2-2 using candidate signal points for the reduced self-modulated signals. Received data RA and RB are obtained.

  This will be specifically described. Soft decision section 512 inputs information on candidate signal points 801, 806, 811, and 816 in FIG. 9 as signal point information 509 and 511, and also receives reception baseband signals R1-2 and R2-2. Soft decision section 512 makes a soft decision using candidate signal points 801, 806, 811, and 816 for both reception baseband signals R1-2 and R2-2. For example, if the reception point indicated by the reception baseband signal R1-2 is the signal point 800 in FIG. 9, the square of the Euclidean distance between the reception signal point 800 and the candidate signal points 801, 806, 811 and 816 is calculated. Thus, a branch metric (referred to as Bx) is obtained. Similarly, the reception point indicated by the reception baseband signal R2-2 is the signal point 800 in FIG. 9 (however, in fact, the reception point of the reception baseband signal R2-1 and the reception point of the reception baseband signal R2-2). Branch metric (referred to as By) is obtained by calculating the square of the Euclidean distance between the received signal point 800 and the candidate signal points 801, 806, 811 and 816.

  Then, the soft decision unit 512 obtains a path metric from the branch metric obtained by adding the branch metric Bx and the branch metric By. For example, when a convolutional code is used, the soft decision unit 512 performs decoding according to the Viterbi algorithm, thereby receiving the received data of the modulated signal A Get RA.

  Similarly, soft decision section 518 inputs candidate signal points 805, 806, 807, and 808 in FIG. 10 as signal point information 515 and 517, and also receives received baseband signals R1-2 and R2-2. . Soft decision section 518 makes a soft decision using candidate signal points 805, 806, 807, and 808 for both reception baseband signals R1-2 and R2-2. For example, if the reception point indicated by the reception baseband signal R1-2 is the signal point 800 in FIG. 10, the square of the Euclidean distance between the reception signal point 800 and the candidate signal points 805, 806, 807, and 808 is calculated. Thus, a branch metric (referred to as Bv) is obtained. Similarly, the reception point indicated by the reception baseband signal R2-2 is the signal point 800 of FIG. 10 (however, in fact, the reception point of the reception baseband signal R2-1 and the reception point of the reception baseband signal R2-2). Branch metric (referred to as Bw) is obtained by calculating the square of the Euclidean distance between the received signal point 800 and the candidate signal points 805, 806, 807, and 808.

  Then, the soft decision unit 518 obtains the path metric from the branch metric obtained by adding the branch metric Bv and the branch metric Bw. Obtain RB.

  Next, the operation of multi-antenna receiving apparatus 120 of this embodiment will be described. The multi-antenna receiving apparatus 120 receives two modulated signals A and B transmitted simultaneously from the two antennas AN1 and AN2 by the two antennas AN3 and AN4. The multi-antenna receiving apparatus 120 uses the channel fluctuation estimation units 403-1A, 403-1B, 403-2A, and 403-2B to transmit each of the transmission antennas AN1 and AN2 based on the known signals inserted into the modulated signals A and B. And channel fluctuations between the receiving antennas AN3 and AN4 are estimated.

  Here, when the modulated signal A and the modulated signal B are QPSK modulated, there are 4 × 4 = 16 signal points in the multiplexed received signal. That is, the number of candidate signal points formed based on the channel fluctuation estimated value is also 16.

  Here, in the conventional multi-antenna receiving apparatus, the signal point distance between the 16 candidate signal points and the receiving point is obtained, the candidate signal point having the smallest distance is detected, and the data indicated by this candidate signal point is obtained. Received data.

  On the other hand, in multi-antenna receiving apparatus 120 of the present embodiment, separation section 501 that separates modulated signals A and B by inverse matrix calculation of the channel variation matrix, and soft decision that softly determines the separated modulated signals. Units 503 and 506 are provided to temporarily obtain digital signals (determination values) of the modulation signals A and B, and narrow down candidate signal points of the modulation signals A and B using the digital signals. Then, an accurate determination is performed by the soft determination unit using only the narrowed candidate signal points. In other words, the separation unit 501 and the soft decision units 503 and 504 perform the temporary determination of the modulation signals A and B, narrow down the candidate signal points using the temporary determination values, and correct only the narrowed candidate signal points. It can be said that the digital judgment (main judgment) is performed.

  Thereby, compared with the case where a soft decision part 512,518 judges a receiving point using all candidate signal points, the amount of calculations can be reduced markedly. For example, in this embodiment, QPSK is used as the modulation method, but the effect is further increased as the number of multi-values increases. For example, if both the modulation signals A and B are modulated by 64QAM, if the number of signal points is not reduced, there are 64 × 64 = 4096 candidate signal points, and branch metrics are obtained for 4096 candidate signal points. Then, a very large circuit is required.

  In addition, the error rate characteristic can be improved as compared with the case where the reception data is obtained using only the inverse matrix operation, that is, compared with the case where the determination results of the soft decision units 503 and 506 are used as the reception data as they are. In particular, when reducing the number of signal points, if correct reduction is performed, a full diversity gain can be obtained, and the error rate characteristics can be further improved. A more preferable configuration for signal point reduction will be described in the following embodiments.

  Thus, according to the present embodiment, each modulation signal 502, 505 is provisionally determined based on each modulation signal 502, 505 separated using the inverse matrix calculation of the channel variation matrix, and a multiplexed modulation signal candidate signal is obtained. By reducing the number of points using the provisional determination results 504 and 507 and then performing accurate determination using the reduced candidate signal points so as to obtain the reception data RA and RB of each modulated signal, the amount of computation is reduced. Received data RA and RB with good error rate characteristics can be obtained. As a result, it is possible to realize a multi-antenna reception apparatus and a multi-antenna reception method that can simplify the apparatus configuration while maintaining the error rate characteristics.

  In the above-described embodiment, the case where the separation unit 501 performs the inverse matrix calculation of the channel variation matrix when separating the modulation signals for reducing the candidate signal points has been described. However, the separation method is reversed. For example, each modulation signal may be estimated and separated using a MMSE (Minimum Mean Square Error) algorithm.

  In the above-described embodiment, a case has been described in which the temporary determination of each modulation signal for reducing candidate signal points is performed by the separation unit 501 and the soft determination units 503 and 506. Not limited to. When the circuit scale is not a problem, for example, as shown in FIG. 11, the soft decision unit 1101 may perform the provisional determination without performing the inverse matrix calculation for separating the modulation signal.

  In FIG. 11, in which the same reference numerals are given to corresponding parts to FIG. 5, the soft decision unit 1101 of the signal processing unit 1100 includes baseband signals R1-2 and R2-2, channel fluctuation estimation values h11, h21, h12 and h22 are input. Soft decision section 1101 estimates 16 candidate signal points 801 to 816 based on channel fluctuation value h11 of modulated signal A and channel fluctuation value h21 of modulated signal B, as shown in FIG. Then, the received signal point 800 of FIG. 8 is estimated from the despread baseband signal R1-2, and for example, the square of each Euclidean distance between the received signal point 800 and the 16 candidate signal points 801 to 816 is obtained. Find branch metrics. Similarly, soft decision section 1101 obtains a branch metric from channel fluctuation signals h12 and h22 of modulated signal A and despread baseband signal R2-2. When using a convolutional code, soft decision section 1100 obtains a path metric from two branch metrics and outputs decision value 1102 for modulated signal A and decision value 1103 for modulated signal B.

(Embodiment 2)
In this embodiment, as compared with the first embodiment, the configuration of the part that performs provisional determination for reducing candidate signal points is made simpler, so that the error rate characteristics are better with a simpler configuration. A multi-antenna receiving apparatus capable of obtaining received data is proposed.

  FIG. 12, in which parts corresponding to those in FIG. 5 are assigned the same reference numerals, shows the configuration of signal processing section 1200 of the multi-antenna reception apparatus of this embodiment. Compared with the signal processing unit 404 in FIG. 5, the signal processing unit 1200 in FIG. 12 is a soft decision unit 506 (FIG. 5) for determining the estimated baseband signal 505 of the modulated signal B separated by the separation unit 501. Is omitted. The signal point reduction units 1201 and 1202 are made to receive the reception data RB of the modulated signal B obtained by the soft decision unit 518. Signal point reduction sections 1201 and 1202 are described in Embodiment 1 using received data RB obtained by soft decision section 518 instead of decision value 507 from soft decision section 506 (FIG. 5). The number of candidate signal points is reduced by the same method. As a result, the overall circuit configuration can be simplified by omitting the soft decision unit 506.

  Next, the operation of the signal processing unit 1200 of this embodiment will be described. The signal processing unit 1200 first decodes only the modulated signal A by the soft decision unit 503, uses the result to reduce the candidate signal points by the signal point reduction units 514 and 516, and decodes the modulation signal B by the soft decision unit 518. As a result, the reception data RB of the modulated signal B is obtained.

  Subsequently, the signal processing unit 1200 reduces the candidate signal points for the modulation signal A using the data RB of the modulation signal B by the signal point reduction units 1201 and 1202, and decodes the modulation signal A by the soft decision unit 512. Thus, the reception data RA of the modulated signal A is obtained. As described above, the signal processing unit 1200 according to the present embodiment does not decode the modulation signal A and the modulation signal B at the same time, but decodes the modulation signal A, decodes the modulation signal B, and decodes the modulation signal A. Decode alternately.

  Thus, according to the present embodiment, provisional determination is performed only for a certain modulation signal, instead of performing provisional determination for all modulation signals and reducing candidate signal points using the provisional determination result in all signal point reduction units. In addition to the effects of the first embodiment, the number of candidate signal points is reduced by using the final determination result (main determination result) for other modulation signals. An antenna receiver can be realized.

(Embodiment 3)
In this embodiment, in addition to obtaining reception data with good error rate characteristics with a small number of computations by performing main determination after reducing candidate signal points, it is possible to further improve A multi-antenna receiver capable of improving error rate characteristics is proposed.

  FIG. 13 in which parts corresponding to those in FIG. 5 are assigned the same reference numerals shows the configuration of signal processing section 1300 of the multi-antenna reception apparatus of this embodiment. That is, the signal processing unit 1300 is used in the multi-antenna receiving apparatus 120 in place of the signal processing unit 404 in FIG.

  The signal processing unit 404 in FIG. 5 described in Embodiment 1 differs from the signal processing unit 1300 in this embodiment in that the signal point reduction units 1301 and 1302 are added to the determination value 507 from the soft determination unit 506. The reception data RB from the soft decision unit 518 is input, and the signal point reduction units 1303 and 1304 receive the reception data RA from the soft decision unit 512 in addition to the decision value 504 from the soft decision unit 503. It is a point.

  As a result, the signal point reduction units 1301 to 1304 can increase the probability of correct signal point reduction compared to the signal point reduction units 508, 510, 514, and 516 of the first embodiment. As a result, the error rate characteristics of the reception data RA and RB finally obtained can be further improved.

  Next, the operation of the signal processing unit 1300 of this embodiment will be described with reference to FIG. As shown in FIG. 14, the signal processing unit 1300 performs soft decision and decoding of the modulation signals A and B in parallel. The signal point reduction of the modulation signal A is performed using the reception data RB of the modulation signal B obtained by the soft decision of the modulation signal B. Conversely, signal point reduction of the modulated signal B is performed using the received data RA of the modulated signal A obtained by the soft decision of the modulated signal A. Then, the reception data RA and RB of the modulation signals A and B are obtained by making a soft decision (main decision) for each of the modulation signals A and B. Furthermore, signal point reduction and soft decision (main decision) are performed repeatedly using the received data RA and RB of the obtained modulated signals A and B.

  This will be specifically described. The first soft decision and decoding operations are the same as the operations of the signal processing unit 404 in FIG. 5 described in the first embodiment. That is, signal point reduction is performed based on the provisional determination values (determination values 504 and 507) obtained by the soft determination units 503 and 504. On the other hand, the second and subsequent soft decisions and decoding are performed using the received data RA and RB obtained by the soft decision units 512 and 518.

  The signal processing unit 1300 transmits the modulated signal A using the modulated signal B using the soft decision value 507 of the modulated signal B in the first soft decision process shown in step ST1A. 8 is reduced, the 16 candidate signal points in FIG. 8 are reduced to the 4 signal points in FIG. 9, and the signal point information (4 signal points) 509 and 511 is sent to the soft decision unit 512. The soft decision unit 512 obtains reception data RA using the signal point information 509 and 511.

  Similarly, the signal processing unit 1300 uses the soft decision value 504 of the modulation signal A to modulate the modulated signal B in the first soft decision process shown in step ST1B. 8 is estimated, the 16 candidate signal points in FIG. 8 are reduced to the 4 signal points in FIG. 10, and the signal point information (4 signal points) 515 and 517 is soft-decided. The soft decision unit 518 uses the signal point information 515 and 517 to obtain the reception data RB.

  In the second soft decision processing shown in steps ST2A and ST3A, the signal processing unit 1300 uses the received data RB obtained in step ST1B to modulate signals in the second soft decision processing shown in steps ST2A and ST3A. 8 is estimated, the 16 candidate signal points in FIG. 8 are reduced to the 4 signal points in FIG. 9 (step ST2A), and the signal point information (4 signal points) 509, 511 is sent to soft decision section 512, and soft decision section 512 obtains reception data RA using signal point information 509 and 511 (step ST3A).

  Similarly, signal processing section 1300 uses received data RA obtained in step ST1A for signal point reduction sections 1303 and 1304 in the second soft decision processing shown in steps ST2B and ST3B for modulated signal B, respectively. 8 is estimated, the 16 candidate signal points in FIG. 8 are reduced to the 4 signal points in FIG. 10 (step ST2B), and the signal point information (4 signal points) ) 515 and 517 are sent to the soft decision section 518, and the soft decision section 518 uses the signal point information 515 and 517 to obtain received data RB (step ST3B).

  In the third soft decision processing shown in steps ST4A and ST5A, the signal processing unit 1300 uses the received data RB obtained in step ST3B to generate modulated signals in the third soft decision processing shown in steps ST4A and ST5A. 8 is estimated, the 16 candidate signal points in FIG. 8 are reduced to the 4 signal points in FIG. 9 (step ST4A), and the signal point information (4 signal points) 509, 511 is sent to soft decision section 512, and soft decision section 512 obtains reception data RA using signal point information 509 and 511 (step ST5A).

  Similarly, signal processing section 1300 uses reception data RA obtained in step ST3A for signal point reduction sections 1303 and 1304 in the third soft decision processing shown in steps ST4B and ST5B for modulated signal B, respectively. 8 is estimated, the 16 candidate signal points in FIG. 8 are reduced to the 4 signal points in FIG. 10 (step ST4B), and the signal point information (4 signal points) ) 515 and 517 are sent to the soft decision unit 518, and the soft decision unit 518 uses the signal point information 515 and 517 to obtain received data RB (step ST5B).

  As described above, in the signal processing unit 1300, signal point reduction after the second time is performed using the reception data RA and RB of the other modulation signal after the previous operation is completed.

  Then, the soft decision units 512 and 518 output the first received data RA and RB after the first soft decision and decoding, respectively. Next, when the second soft decision and decoding are performed, the second received data RA and RB are output instead of the first received data RA and RB. That is, when the nth soft decision and decoding are performed, the received data RA and RB, which are the nth soft decision decoding results, are output instead of the (n-1) th received data RA and RB.

  In this way, in reducing candidate signal points, iteration (iteration is performed) using data after error correction decoding of other modulation signals (assuming that error correction decoding processing is performed by soft decision sections 512 and 518). ), The probability that a correct candidate signal point can be left can be increased, and the error rate characteristics of the received data RA and RB can be further improved.

  FIG. 15 shows an image of the decoding processing procedure in the present embodiment. One frame of the modulation signal A and the modulation signal B is composed of a plurality of symbols. First, the first error correction for one frame is performed. Then, the number of states is reduced reflecting the first error correction result, and error correction for one frame is performed for the second time. As described above, after reducing the number of states reflecting the error correction result of (n−1) degrees, error correction for one frame of the nth time is performed.

  FIG. 16 shows a simulation result of reception characteristics (relationship between carrier power to noise power ratio (C / N) and bit error rate) when the signal processing unit 1300 of this embodiment is used. As is clear from this figure, both the modulation signals A and B improve the reception quality as the number of iterative decoding increases. However, it is not necessary to increase the number of times, and the effect of improving the reception quality is saturated after a certain number of times. Further, the reception quality of the modulated signals A and B is the same when the modulation scheme is the same.

  Thus, according to the present embodiment, in reducing candidate signal points, the data RA and RB of other modulation signals after error correction decoding (after main determination) are used, and an iteration process is performed to obtain a final result. By obtaining the reception data RA and RB, it is possible to obtain the reception data RA and RB with further improved error rate characteristics as compared with the first embodiment.

  In this embodiment, the case where the temporary determination of each modulation signal for reducing candidate signal points is performed by the separation unit 501 and the soft determination units 503 and 506 has been described. Without being limited to the circuit scale, for example, as shown in FIG. 17, for example, the soft decision unit 1705 may perform the tentative determination without performing the inverse matrix operation for separating the modulation signal. .

  In FIG. 17, in which parts corresponding to those in FIG. 13 are assigned the same reference numerals, the soft decision unit 1705 of the signal processing unit 1700 includes baseband signals R1-2 and R2-2, channel fluctuation estimation values h11, h21, h12 and h22 are input. Soft decision section 1705 estimates 16 candidate signal points 801 to 816 based on channel fluctuation value h11 of modulated signal A and channel fluctuation value h21 of modulated signal B, as shown in FIG. Then, the received signal point 800 of FIG. 8 is estimated from the despread baseband signal R1-2, and for example, the square of each Euclidean distance between the received signal point 800 and the 16 candidate signal points 801 to 816 is obtained. Find branch metrics. Similarly, soft decision section 1705 obtains a branch metric from channel fluctuation signals h12 and h22 of modulated signal A and despread baseband signal R2-2. When the convolutional code is used, the soft decision unit 1705 obtains a path metric from two branch metrics, sends the modulation signal A decision value 1706 to the signal point reduction units 1703 and 1704, and determines the modulation signal B. The value 1707 is sent to the signal point reduction units 1701 and 1702.

  Here, when the signal processing unit 1300 in FIG. 13 is compared with the signal processing unit 1700 in FIG. 17, the signal processing unit 1700 makes a determination on 16 candidate signal points in the soft decision unit 1705, so There is a drawback that the circuit scale of the metric and path metric is increased, and the overall circuit scale is larger than that of the signal processing unit 1300. In particular, in the case of QPSK, there are 16 points. However, since there are 4096 signal points at 64 QAM, the reality is lost as the modulation multi-level number increases.

  However, since the soft decision unit 1705 can obtain a determination value with higher accuracy than when the separation unit 501 and the soft decision units 503 and 506 are used, an error may occur even if the number of iterations is small. There is an advantage that reception data RA and RB with good rate characteristics can be obtained.

(Embodiment 4)
The feature of this embodiment is that, in the third embodiment, each modulation signal is soft-decision decoded in parallel, and the candidate signal points of the self-modulation signal are reduced using the soft-decision decoding results of other modulation signals. Thus, each modulation signal is alternately subjected to soft decision decoding, and the candidate signal points of the own modulation signal are reduced using the soft decision decoding results of other modulation signals. As a result, the number of computations when adopting the iteration technique for signal point deletion can be reduced, so that the circuit configuration can be further simplified.

  FIG. 18 in which parts corresponding to those in FIG. 13 are assigned the same reference numerals shows the configuration of the signal processing unit of the multi-antenna reception apparatus of this embodiment. Compared with the signal processing unit 1300 in FIG. 13 described in Embodiment 3, the signal processing unit 1800 has a configuration in which the soft decision unit 506 is omitted.

  Compared with the signal processing unit 1200 of FIG. 12 described in the second embodiment, the signal processing unit 1200 is added with an iteration process.

  In the signal processing unit 1800, the signal point reduction units 1803 and 1804 for the modulated signal B are the same as in the third embodiment, the decision value 504 obtained by the soft decision unit 503 and the error correction decoding obtained by the soft decision unit 512. The candidate signal points are reduced using both of the subsequent received data RA, but the signal point reduction units 1801 and 1802 for the modulated signal A only receive the received data RB after error correction decoding obtained by the soft decision unit 518. The candidate signal points are used to reduce the number of candidate signal points. As described above, the signal processing unit 1800 according to the present embodiment can simplify the overall circuit configuration compared to the signal processing unit 1300 according to the third embodiment by the amount that the soft decision unit 506 is omitted.

  Next, the operation of the signal processing unit 1800 of this embodiment will be described with reference to FIG. In the signal processing unit 1800, the signal processing unit 1300 of the third embodiment performs soft decision and decoding of the modulated signals A and B in parallel, whereas the first soft decision decoding performs only the modulated signal A, The soft decision decoding of the modulation signal A and the modulation signal B is alternately performed so that the soft decision decoding is performed only for the modulation signal B and the third soft decision decoding is performed only for the modulation signal A.

  This will be specifically described. First, signal processing section 1800 performs soft decision decoding only on modulated signal A by soft decision section 503 (step ST10A), and uses the result to reduce candidate signal points by signal point reduction sections 1803 and 1804 (step ST10B). Then, the soft decision unit 518 performs soft decision decoding on the modulated signal B (step ST11B), thereby obtaining the reception data RB of the modulated signal B. Next, signal processing section 1800 reduces candidate signal points using received data RB of modulated signal B by signal point reducing sections 1801 and 1802 (step ST11A), and soft-decision decoding of modulated signal A by soft-decision section 512. By performing again (step ST12A), the reception data RA of the modulated signal A is obtained. Similarly, the soft decision decoding of the modulation signal A and the soft decision decoding of the modulation signal B are alternately repeated while reducing candidate signal points using the other soft decision decoding result.

  FIG. 20 shows a simulation result of reception characteristics (relationship between carrier power to noise power ratio (C / N) and bit error rate) when the signal processing unit 1800 of this embodiment is used. As can be seen from this figure, even when soft-decision decoding is performed alternately on each modulation signal, received data with good error rate characteristics is the same as when soft-decision decoding is performed on each modulation signal in parallel (FIG. 16). Can be obtained. In addition, both the modulation signals A and B improve the reception quality as the number of iterative decoding increases. However, the number of times is not simply increased, and the effect of improving the reception quality is saturated at a certain number of times.

  Thus, according to the present embodiment, the process of reducing the candidate signal points of the self-modulation signal using the soft decision decoding result of the other modulation signal is performed alternately for each modulation signal. In addition to the effect of 3, since the number of times of decoding is halved, the circuit scale can be further reduced.

(Embodiment 5)
In this embodiment, in addition to Embodiments 1 to 4 described above, it is proposed to transmit modulated signals having different reception qualities from the respective antennas.

  An example is shown in FIG. FIG. 21 shows an example of signal point arrangement on the IQ plane when the modulation scheme of the modulation signal A is QPSK and the modulation scheme of the modulation signal B is 16QAM in consideration of the configurations of FIGS. . FIG. 22 shows the relationship between the carrier power-to-noise power ratio of QPSK and 16QAM and the bit error rate.

  Here, when the configuration of FIGS. 12 and 18 is adopted, when the modulation method of the modulation signal A is QPSK and the modulation method of the modulation signal B is 16 QAM, as shown in FIG. Since the modulation method of the modulation signal A is QPSK, the reception quality is good (compared with 16QAM), and the decision value 504 (digital signal) of the modulation signal A with good reception quality is obtained by the soft decision unit 503.

  Since the determination value of the digital signal of the modulation signal A thus obtained is accurate, the possibility of erroneous signal point reduction during signal point reduction is reduced, and the soft decision unit 518 performs soft decision decoding of the modulation signal B. The error rate characteristic of the reception data RB of the modulated signal B obtained when performing is improved. Here, considering the transmission rate, the modulation method of the modulation signal B may be 16QAM (or 64QAM may be sufficient), for example, having a higher modulation multi-value number than QPSK. Thereby, it is possible to achieve both improvement in reception quality and transmission speed.

  As described above, by reducing the modulation multi-level number of the modulation signal A to be smaller than the modulation multi-level number of the modulation signal B and ensuring the reception quality of the modulation signal A, it is possible to perform good signal point reduction. As a result, the reception quality of the modulated signal B can be ensured. As a result, it is possible to improve both the reception quality and the transmission speed.

  That is, if the reception quality of the modulation signal used for the initial provisional determination is improved, the signal point reduction can be accurately drawn, and a good determination result can be obtained in the subsequent main determination.

  Further, when iterative decoding (iteration) is performed, the number of iterations is reduced, and the circuit scale can be reduced.

  Further, the same effect can be obtained by setting the coding rates of the modulation signal A and the modulation signal B to be different. For example, the coding rate of the modulation signal A is 1/4 and the coding rate of the modulation signal B is 3/4. Then, since the reception quality of modulated signal A is good, there is a high possibility that signal point reduction will be performed correctly, and the reception quality of modulated signal B will also be improved.

  Further, the same effect can be obtained by making the spreading code lengths of the modulation signal A and the modulation signal B different. For example, the spreading code length of the modulation signal A may be longer than the spreading code length of the modulation signal B.

  Thus, according to the present embodiment, in addition to the configurations of the first to fourth embodiments, the modulation scheme, the coding rate, the spreading factor, etc. are changed between the modulation signals so that the reception quality of each modulation signal is different. As a result, in addition to the effects of the first to fourth embodiments, it is possible to improve both the error rate characteristics and the transmission speed.

(Embodiment 6)
In this embodiment, a multi-antenna transmission apparatus is proposed in which the interleave pattern of the modulation signal transmitted from each antenna is different between the modulation signals.

  FIG. 23, in which parts corresponding to those in FIG. Multi-antenna transmission apparatus 2300 except that interleaver 2301A is provided between encoding section 201A and modulation section 202A, and interleaver 2301B is provided between encoding section 201B and modulation section 202B. The configuration is the same as that of the multi-antenna transmission apparatus 110 of FIG. 2 described in the first embodiment.

  Interleaver 2301A receives encoded digital signal S1A, changes the order, and sends interleaved digital signal S10A to modulation section 202A. Similarly, interleaver 2301B receives encoded digital signal S1B as input, changes the order, and sends interleaved digital signal S10B to modulation section 202B.

  If interleaving processing is performed on the transmission device side in this way, it is necessary to perform deinterleaving processing on the reception side. A configuration example of the receiving apparatus in this case is shown in FIG. The configuration example in FIG. 24 corresponds to the signal processing unit 1300 described in the third embodiment. In FIG. 24, in which parts corresponding to those in FIG. 13 are assigned the same reference numerals, the signal processing unit 2400 includes deinterleavers 2401A, 2403A, and 2404A that return the signals rearranged by the interleaver 2301A on the transmission side. In addition, deinterleavers 2401B, 2403B, and 2404B are provided for returning the signals rearranged by the interleaver 2301B on the transmission side. The signal processing unit 2400 includes interleavers 2402A and 2405A that perform rearrangement similar to the interleaver 2301A, and includes interleavers 2402B and 2405B that perform rearrangement similar to the interleaver 2301B.

  With this configuration, the signal processing unit 2400 returns the estimated baseband signal for the transmission digital signal TA separated by the separation unit 501 to the original arrangement by the deinterleaver 2401A, and then sends the estimated baseband signal to the soft decision unit 503. The estimated baseband signal for the digital signal TB is returned to the original arrangement by the deinterleaver 2401B and then sent to the soft decision unit 506. The decision value obtained by soft decision section 503 is interleaved by interleaver 2402A and then sent to signal point reduction sections 1303 and 1304, and the decision value obtained by soft decision section 506 is interleaved by interleaver 2402B. And then sent to signal point reduction sections 1301 and 1302. Further, the decision values obtained by the soft decision unit 512 are input to the signal point reduction units 1303 and 1304 after being interleaved by the interleaver 2405A, and the signal point reduction units 1301 and 1302 are inputted by the soft decision unit 518. The obtained determination value is input after being interleaved by the interleaver 2405B.

  As a result, the signal point reduction sections 1301 and 1302 can obtain candidate signal points that are reduced for the modulated signal A by reducing the signal points of the interleaved modulated signal B from the interleaved received signals. However, since the reduced candidate signal points are interleaved signal points, they are input to the soft decision unit 512 after being deinterleaved by the deinterleavers 2403A and 2404A. Similarly, signal point reduction sections 1303 and 1304 can obtain reduced candidate signal points for modulated signal B by reducing the signal points of interleaved modulated signal A from the interleaved received signals. However, since this reduced candidate signal point is an interleaved signal point, it is input to the soft decision section 518 after deinterleaving by the deinterleavers 2403B and 2404B.

  Here, the configuration example for decoding the signal interleaved on the transmission side is described based on the signal processing unit 1300 described in the third embodiment. However, the first embodiment, the second embodiment, and the fourth embodiment are described. Also in the receiving apparatus described in the fifth embodiment, if a deinterleaver and an interleaver corresponding to the interleaver on the transmission side are provided as appropriate, signals of different interleave patterns are transmitted from the antennas as described above. , Each modulated signal can be decoded.

  Next, the interleave pattern (transmission signal replacement order) will be described in detail. The most important point in the present embodiment is that the interleave pattern for modulated signal A is different from the interleave pattern for modulated signal B. Thereby, the error rate characteristic on the receiving side can be improved. In particular, the reception quality can be greatly improved by selecting the interleave pattern so that the interleave pattern of modulated signal A and the interleave pattern for modulated signal B are close to uncorrelated. This point will be described in detail.

  FIG. 25 illustrates an example of a symbol state when the modulation signal A and the modulation signal B have the same interleave pattern. Assume that soft decision section 503 in FIG. 5 decodes modulated signal A, and as a result, the erroneously determined symbols as shown in FIG. Incidentally, when a convolutional code or the like is used, it is common that errors occur continuously. Then, when the signal point reduction units 514 and 516 reduce the signal points, as shown in FIG. 25B, an error occurs in the signal point selection by the signal point reduction for five consecutive symbols. As a result, when the modulated signal B is decoded by the soft decision unit 518, the reception quality is not effectively improved. This is because error correction codes have a low ability to correct continuous errors.

  Next, a case will be described in which the interleave pattern for modulated signal A and the interleave pattern for modulated signal B are different on the transmission side as in the present embodiment. In this case, when signal point reduction is performed, a symbol state as shown in FIG. 26 is obtained. It is assumed that the soft decision unit 503 in FIG. 24 decodes the modulated signal A, and as a result, the erroneously determined symbols as shown in FIG. Then, when the number of signal points is reduced in the signal point number reduction units 1303 and 1304, the interleave pattern of the modulation signal A and the interleave pattern of the modulation signal B are different from those in FIG. As in (B), the signal point selection error due to signal point reduction occurs discretely. That is, as shown in FIG. 25B, signal point selection errors due to signal point reduction do not occur continuously. As a result, when the soft decision unit 518 decodes the modulated signal B, the reception quality is effectively improved. This is because error correction codes have a high ability to correct discrete errors.

  This operation and effect are the same in the case of the configuration using the iteration technique.

  Even if the above operation is performed by replacing the modulation signal A with the modulation signal B and the modulation signal B with the modulation signal A, the same operation and effect can be obtained. As a result, the reception quality of decoding of the modulation signal A is also effective. Improve.

  Thus, according to the present embodiment, the interleaving pattern of the modulation signal transmitted from each antenna is different between the modulation signals, so that the influence of burst errors is reduced at the time of decoding on the receiving side, and the error rate is reduced. A multi-antenna transmission apparatus capable of obtaining received data with good characteristics can be realized.

  In particular, it is suitable for application to a multi-antenna transmission apparatus that transmits a modulated signal to a multi-antenna reception apparatus having a signal point reduction unit as in the first to fourth embodiments.

  FIG. 27 shows a simulation result of reception characteristics when using different interleave patterns between modulated signals as in the present embodiment and reception characteristics when using the same interleave pattern between modulated signals. In FIG. 27, the horizontal axis represents Eb / No (energy per bit-to-noise spectral density ratio), and the vertical axis represents BER (Bit Error Rate).

  Circles in the figure indicate signal processing using the configuration of this embodiment, that is, a signal transmitted from the multi-antenna transmission apparatus 2300 configured as shown in FIG. 23 and configured as shown in FIG. The characteristics in the case of receiving and demodulating by a multi-antenna receiving apparatus having unit 2400 are shown. On the other hand, triangles in the figure indicate reception characteristics when the same interleave pattern is used between the modulation signals. In the simulation, the characteristics when no iterative decoding is performed, when the iteration is performed only once, and when the iteration is performed five times were examined. In addition, this simulation is a result in the case where the propagation environment is a rice fading environment with a rice factor of 10 dB, the modulation method is QPSK, and a convolutional code with a coding rate of 1/2 is performed.

  As can be seen from the simulation results, when the interleave pattern is the same between the modulated signals, the reception quality is only slightly improved even if the number of iterations is increased as shown by the circles in the figure. On the other hand, when different interleave patterns are selected between the modulation signals, the reception quality can be effectively improved by increasing the number of repetitions, as indicated by a triangular mark in the figure.

  In this embodiment, when the interleave pattern of the modulation signal transmitted from each antenna is different, interleavers 2301A and 2301B are provided so that the order of symbols of each modulation signal is different between the modulation signals. However, the method of making the interleave pattern different between modulated signals is not limited to this.

  As a method for making interleaving different between the modulated signals, for example, the following methods can be considered.

(I) A method of making the arrangement of data constituting the symbols of each modulation signal different as in this embodiment

  A specific example of this method is shown in FIG. With respect to the modulation signal A, by interleaving the data aligned with the data 1, data 2,..., Data 200 before the interleaving, for example, the data is rearranged every 5th data, the data 1, the data 6,. Data 196, data 2, data 7, ... data 197, data 3, data 8, ... data 198, data 4, data 9, ... data 199, data 5, data 10, ... data Rearrange with 200. On the other hand, with respect to the modulation signal B, by interleaving the data that was aligned with data 1, data 2,..., Data 200 before the interleaving, for example, the data is rearranged every 8th data, Data 9, ... Data 193, Data 2, Data 10, ... Data 194, Data 3, Data 11, ... Data 195, Data 4, Data 12, ... Data 196, Data 5, Data 13 ,... 197, data 6, data 14,... 198, data 7, data 15,..., Data 199, data 8, data 16,. That is, the data arrangement itself is made different between the modulation signal A and the modulation signal B.

(Ii) The arrangement of symbols and data among the modulation signals is the same. However, when symbols and data are arranged in the frequency direction and time direction of subcarriers, for example, as will be described later with reference to FIG. How to make yourself different

  A specific example of this method is shown in FIG. As shown in FIG. 29 (A), by interleaving the data that was aligned with data 1, data 2,..., Data 200 before interleaving, for example, the data is rearranged every five data, Data 6, ... Data 196, Data 2, Data 7, ... Data 197, Data 3, Data 8, ... Data 198, Data 4, Data 9, ... Data 199, Data 5, Data 10 ,... Rearranged with the data 200. This is performed for each of the modulation signals A and B. That is, the interleave pattern between the modulated signals at this time is the same. Then, as shown in FIGS. 29B and 29C, the arrangement patterns of the modulated signals A and B on the subcarriers are made different. 29B and 29C show the case where the number of subcarriers of the OFDM signal is 200, and for the modulation signal A with respect to the frequency axis or the time axis, data 1, data 6,... 196, data 2, data 7, ... data 197, data 3, data 8, ... data 198, data 4, data 9, ... data 199, data 5, data 10, ... data 200 Line up. On the other hand, the modulation signal B is offset by 5 carriers with respect to the arrangement of the modulation signal A, and data 185, data 190, data 195, data 200, data 1, data 6,. Line up with 180. In this way, even when one modulation signal is offset from the other modulation signal by several carriers or by a certain time, the interleaving can be made different between the modulation signals.

(Iii) A method of using the above methods (i) and (ii) in combination

  That is, the different interleave patterns described in the present invention do not only indicate the case where the arrangement of symbols and data itself is different, but the arrangement of symbols and data in the frequency direction and / or in the time direction is different. To include. This is the same in any of the following embodiments for explaining the interleave pattern.

(Embodiment 7)
In this embodiment, a case where the features of the above-described embodiment are applied to multicarrier communication will be described. In particular, a case where an OFDM (Orthogonal Frequency Division Multiplexing) scheme is used will be described.

  FIG. 30, in which parts corresponding to those in FIG. 2 are assigned the same reference numerals, shows the configuration of the multi-antenna transmission apparatus of this embodiment. Compared with the multi-antenna transmission apparatus 110 of FIG. 2, the multi-antenna transmission apparatus 2700 performs serial-parallel conversion on the baseband signals S2A and S2B output from the modulation sections 202A and 202B instead of the spreading sections 203A and 203B. Similar to multi-antenna transmission apparatus 110 in FIG. 2 except that parallel conversion units (S / P) 2701A and 2701B and inverse Fourier transform units (idft) 2702A and 2702B for performing inverse Fourier transform on parallel signals S20A and S20B are included. It consists of.

  FIG. 31 shows a frame configuration on the time-frequency axis of an OFDM signal transmitted from multi-antenna transmission apparatus 2700. In this figure, as an example, an OFDM signal is composed of carrier 1 to carrier 5 and a symbol at the same time is transmitted simultaneously. The hatched portion in the figure is a pilot symbol (known signal), which is a symbol for estimating a propagation environment (channel fluctuation) in the receiving apparatus. Incidentally, although it is called a pilot symbol here, it may be called differently such as a preamble. The portion indicated by a blank is a data symbol.

  There are two types of data symbol encoding methods: a method of encoding in the frequency axis direction and a method of encoding in the time axis direction. When encoding is performed in the time axis direction, it is the same as that there are a plurality of carriers (five carriers in FIG. 31) having the frame configuration of FIG. One feature when using the OFDM method is that encoding can be performed in the frequency axis direction. It is also possible to encode in both the frequency axis and the time axis.

  FIG. 32 shows another configuration of the multi-antenna transmission apparatus of this embodiment. In this configuration, the multi-antenna transmission method using different interleave patterns in the sixth embodiment is applied to multi-carrier transmission. In FIG. 32, in which parts corresponding to those in FIG. 23 described in the sixth embodiment are denoted by the same reference numerals, multi-antenna transmission apparatus 2900 is output from modulation sections 202A and 202B instead of spreading sections 203A and 203B. It has serial-parallel conversion units (S / P) 2701A and 2701B for serial-parallel conversion of baseband signals S2A and S2B, and inverse Fourier transform units (idft) 2702A and 2702B for inverse Fourier transform of parallel signals S20A and S20B. Except for this, the configuration is the same as that of the multi-antenna transmission apparatus 2300 of FIG.

  As described above, as a method of selecting an interleave pattern when the characteristics of Embodiment 6 are applied to OFDM transmission, for example, the interleave pattern of interleaver 2301A is transferred from a low-frequency subcarrier to a high-frequency subcarrier. It is proposed that the interleave pattern of the interleaver 2301B is arranged by rearranging data from a subcarrier having a high frequency to a subcarrier having a low frequency.

  For example, when one frame is configured as shown in FIG. 31, interleaver 2301A arranges data for modulated signal A in the order of subcarrier 5, subcarrier 3, subcarrier 1, subcarrier 4, and subcarrier 2. Then, interleaver 2301B arranges data for modulated signal B in the order of subcarrier 1, subcarrier 3, subcarrier 5, subcarrier 2, and subcarrier 4. In this way, since the interleave pattern in the frequency direction can be made close to uncorrelated, the probability that both of the two OFDM-modulated signals are erroneous in a burst can be reduced.

  Similarly, as a method of selecting an interleave pattern when the features of the sixth embodiment are applied to OFDM transmission, for example, the interleave pattern of the interleaver 2301A is rearranged from the earlier time to the later time. It is proposed that the interleave pattern of the interleaver 2301B is arranged by rearranging data from the later time to the earlier time.

  For example, when one frame is configured as shown in FIG. 31, for example, in subcarrier 1, interleaver 2301 </ b> A converts data about modulated signal A to time 2, time 4, time 6, time 8, time 3, and time 5. , Time 7 and time 9 are arranged, and the interleaver 2301B arranges data about the modulated signal B in the order of time 9, time 7, time 5, time 3, time 8, time 6, time 4, time 2 Arrange. In this way, since the interleave pattern in the time direction can be made close to uncorrelated, the probability that both of the two OFDM modulation signals are erroneous in a burst manner can be reduced.

  Furthermore, each modulation signal may be randomly interleaved in both the frequency direction and the time direction. In this way, each modulated signal can be made more uncorrelated, so the probability that both of the two OFDM modulated signals are erroneous in a burst manner can be further reduced.

  FIG. 33, in which parts corresponding to those in FIG. The multi-antenna receiving apparatus 3000 is different from the multi-antenna receiving apparatus 120 of FIG. 4 described in Embodiment 1 in place of the despreading units 402-1 and 402-2, and a Fourier transform unit (dft) 3001-1. , 3001-2, except for having the same configuration as FIG. In addition, the signal processing unit 3002 may apply any configuration proposed in Embodiments 1 to 6.

  The Fourier transform unit 3001-1 performs a Fourier transform process on the baseband signal R1-1, and the Fourier transform signal R1-2 is converted into a channel variation estimation unit 403-1A of the modulation signal A and a channel variation of the modulation signal B. The data is sent to the estimation unit 403-1B and the signal processing unit 3002.

  Similarly, the Fourier transform unit 3001-2 performs a Fourier transform process on the baseband signal R2-1, and uses the signal R2-2 after the Fourier transform as the channel variation estimation unit 403-2A of the modulation signal A and the modulation signal B. To channel fluctuation estimation section 403-2B and signal processing section 3002.

  Each channel fluctuation estimation section 403-1A, 403-1B, 403-2A, 403-2B estimates the channel fluctuation for each subcarrier using pilot symbols arranged in each subcarrier as shown in FIG. To do. Thus, channel fluctuation estimation units 403-1A, 403-1B, 403-2A, and 403-2B obtain channel fluctuation estimation values for each channel and each subcarrier. That is, channel fluctuation estimated values h11, h21, h12, and h22 include channel fluctuation estimated values of subcarriers 1 to 5, respectively.

  Here, the signal processing unit 3002 receives the signals R1-2 and R2-2 after Fourier transform, the channel fluctuation signal groups h11 and h12 of the modulation signal A, and the channel fluctuation signal groups h21 and h22 of the modulation signal B as input, and changes the channel. By determining the signals R1-2 and R2-2 after Fourier transform using the signal groups h11, h12, h21, and h22, the reception data RA of the modulation signal A and the reception data RB of the modulation signal B are obtained.

  The signal processing flow of the signal processing unit 3002 is the same as that in the first to sixth embodiments. For example, a case where the signal processing unit 2400 described in Embodiment 6 is applied as the signal processing unit 3002 will be described as an example. Separation section 501 receives as input channel fluctuation estimation groups h11 and h12 of modulated signal A, channel fluctuation estimation groups h21 and h22 of modulated signal B, and signals R1-2 and R2-2 after Fourier transform, and performs inverse matrix calculation. Thus, the modulation signal A and the modulation signal B are separated. Each deinterleaver 2401A, 2401B, 2403A, 2404A, 2403B, 2404B performs deinterleave processing corresponding to the interleave pattern on the frequency-time axis, and each interleaver 2402A, 2402B, 2405A, 2405B performs frequency-time. Interleave processing corresponding to the interleave pattern in the axis is performed.

(Embodiment 8)
In this embodiment, it is proposed that the reliability in the provisional determination performed for signal point reduction is reflected in the main determination processing after signal point reduction. As a result, the error rate characteristics of the data obtained by the main determination process can be further improved. In the case of this embodiment, as a preferred example, a method of weighting the branch metric of each symbol in the main determination process using the path metric value of each symbol when soft determination is performed as provisional determination. suggest.

  In this embodiment, the signal processing unit 2400 having the configuration of FIG. 24 described in the sixth embodiment will be described as an example. That is, a case where modulated signal A and modulated signal B interleaved with different interleave patterns are received and separated and decoded will be described as an example.

  Here, as described in FIG. 26, assuming that the determination value of each symbol output from soft decision section 503 is in the state as shown in FIG. 34A, signal point reduction by signal point reduction sections 1303 and 1304 is performed. The later state is as shown in FIG. Here, as described in the sixth embodiment, the interleave pattern of the modulated signal A can be made discrete as the symbols from which erroneous candidate signal points are selected for the modulated signal B as shown in FIG. This is because the interleave pattern of the modulation signal B is different from that of the modulation signal B.

  In the present embodiment, the path metric obtained by the soft decision unit 503 is reflected in the soft decision process by the soft decision unit 518. Further, the path metric obtained by the soft decision unit 506 is reflected in the soft decision process in the soft decision unit 512. In practice, the path metric may be notified from the soft decision unit 503 of FIG. 24 to the soft decision unit 518, and the path metric may be notified from the soft decision unit 506 to the soft decision unit 512.

  Specifically, it is assumed that soft decision section 503 obtains a value as shown in FIG. 34 (B) as the minimum value of the path metric in path memory length n for each symbol of modulated signal A. The soft decision unit 518 uses the candidate signal point for the reduced modulation signal B to determine each symbol of the modulation signal B, and the minimum path metric of the modulation signal A symbol used for signal point reduction Judgment is made using the value.

  Here, there is a correlation between the minimum value of the path metric of each symbol for the modulation signal A and the error of the symbol. Specifically, the greater the minimum value of the path metric, the easier the error is for the symbol.

  In the present embodiment, as the minimum value of the path metric at the time of soft decision of another modulation signal (for example, modulation signal A) used for signal point reduction is larger, the reliability of the reduced signal point is also reduced. Based on the consideration that if the signal signal point is used to perform main determination of the self-modulated signal (for example, modulation signal B), the reliability of the determination is reduced, and other factors used for signal point reduction when performing main determination The minimum value of the path metric of the symbol of the modulation signal is used.

  In practice, the soft decision unit 518 obtains a path metric after obtaining the branch metric of the modulated signal B, and for each branch metric of each symbol, as shown in FIG. Multiplication is performed by the reciprocal of the minimum value of the path metric of the modulation signal A symbol used for the reduction of candidate signal points for the symbol. For example, for the symbol 3201 of the modulation signal B, the branch metric is multiplied by 1/20, and for the symbol 3202, the branch metric is multiplied by 1/52.

  As described above, when performing the main determination using the reduced signal points, the branch metric is multiplied by a value corresponding to the reliability of the signal point reduction, thereby improving the reliability of the path metric. It becomes like this. As a result, it is possible to improve the error rate characteristic of the data obtained by the main determination.

  Thus, according to the present embodiment, the reliability in the provisional determination (soft determination for other modulation signals) performed for signal point reduction is set as the main determination (soft determination for the self-modulation signal) after signal point reduction. ), The error rate characteristics of the data obtained by the main determination process can be further improved.

  In this embodiment, the case has been described in which the reliability of the temporary determination is reflected in the main determination by multiplying the reciprocal of the minimum value of the path metric at the time of the temporary determination by the branch metric at the time of the main determination. The method of reflecting the reliability at the time of tentative determination in the main determination is not limited to this, and the main determination may be performed using a coefficient related to the minimum value of the path metric.

  Further, as a method of reflecting the reliability at the time of temporary determination in the main determination, a difference between the minimum value of the path metric and the second smallest value of the path metric may be reflected in the main determination. Here, it can be said that the determination is more reliable as the difference between the minimum value of the path metric and the second smallest value of the path metric is larger. In consideration of this, the multiplication coefficient may be obtained using this difference instead of the reciprocal of the minimum value of the path metric described above.

  In this embodiment, the feature of this embodiment has been described with reference to FIG. 24, but the scope of application of this embodiment is not limited to this. The feature of this embodiment can be widely applied to the case where the candidate signal points of the own modulation signal are reduced using the determination result of other modulation signals and the own modulation signal is determined using the reduced candidate signal points. it can. For example, the present invention can be applied to all of Embodiments 1 to 7 described above.

(Embodiment 9)
A feature of this embodiment is that a specific symbol is transmitted at a predetermined timing in addition to the features of the first to eighth embodiments. In this embodiment, first, it is proposed to transmit a space-time code (in this embodiment, a space-time block code (STBC) is used) as a specific symbol. In this embodiment, secondly, it is proposed to transmit a special symbol as a specific symbol.

  As described above, by transmitting a specific symbol at a predetermined timing, in addition to the effects of the first to eighth embodiments, the error rate characteristics of received data can be further improved.

(I) When transmitting a space-time block code First, the principle of transmitting and receiving a space-time block code will be described. FIG. 35 shows a frame configuration example of modulated signal A and modulated signal B transmitted from each antenna of the transmission apparatus. As shown in the figure, the transmitting apparatus regularly transmits STBC symbols 3303 as modulation signals A in addition to channel estimation symbols 3301 and data symbols 3302, 3304, and 3306 from the first antenna AN1 (FIG. 1). It is supposed to be. In addition to the channel estimation symbol 3307 and the data symbols 3308, 3310, and 3312, the transmission apparatus regularly transmits the STBC symbol 3309 as the modulated signal B from the second antenna AN2 (FIG. 1). Yes.

  Note that the time axes of FIGS. 35A and 35B are the same. That is, channel estimation symbols 3301 and 3307, data symbols 3302 and 3308, STBC symbols 3303 and 3309, data symbols 3304 and 3310, STBC symbols 3305 and 3311, and data symbols 3306 and 3312 are transmitted at the same time. In the example of FIG. 35, two STBC symbols are inserted between four data symbols and transmitted.

The use of STBC for multi-antenna communication is a known technique, which will be briefly described with reference to FIG. In STBC, the modulated signal of the signal S1 is transmitted from the antenna 3401 and the modulated signal of the signal S2 is transmitted from the antenna 3402 at time t. At time t + 1, the modulated signal of the signal −S2 ^* is transmitted from the antenna 3401 and the modulated signal of the signal S1 ^* is transmitted from the antenna 3402. However, * shows conjugate complex.

At this time, if the received signal at time t of the antenna 3403 is R1 (t) and the received signal at time t + 1 is R1 (t + 1), the following relational expression is established.

  In the receiving unit, the transmission signals S1 and S2 are demodulated by solving the equation (2). As can be seen from the equation (2), a large diversity gain can be obtained. Can be obtained with good quality.

  Here, when STBC is inserted as shown in FIG. 35, convolutional coding, turbo, and data symbol 3302, 3304, 3306 and signal S1 in STBC symbol 3303, 3305 are formed when modulated signal A is formed. Encoding such as encoding and LDPC (Low Density Parity Check) encoding may be performed. When the modulation signal B is formed, the data symbol 3308, 3310, 3312 and the signal S2 in the STBC symbol 3309, 3311 are subjected to coding such as convolutional coding, turbo coding, LDPC coding. Good.

  Next, a configuration example of a multi-antenna transmission apparatus for transmitting a signal as shown in FIG. 35 and a configuration example of a multi-antenna reception apparatus for receiving and demodulating the signal will be described.

  For the multi-antenna transmission apparatus, the modulators 202A and 202B in FIGS. 2 and 30 may be configured as shown in FIG. 37, for example. Since the modulation unit 202A and the modulation unit 202B may have substantially the same configuration, only the modulation unit 202A will be described here.

  Modulation section 202A inputs encoded data S1A to data symbol signal generation section 3501 and STBC symbol signal generation section 3502. In addition, frame configuration signal S10 is input to data symbol signal generation section 3501, STBC symbol signal generation section 3502, channel estimation symbol signal generation section 3503, and signal selection section 3508.

  When the frame configuration signal S10 indicates a data symbol, the data symbol signal generation unit 3501 modulates the encoded data S1A and outputs a baseband signal 3504 of the data symbol. When the frame configuration signal S10 indicates an STBC symbol, the STBC symbol signal generation unit 3502 modulates the encoded data S1A and outputs a baseband signal 3506 of the STBC symbol. Channel estimation symbol signal generation section 3503 outputs baseband signal 3507 of the channel estimation symbol when frame configuration signal S10 indicates a channel estimation symbol.

  The signal selection unit 3508 selects the baseband signal indicated by the frame configuration signal S10 from the input baseband signals 3504, 3506, and 3507, and outputs it as the baseband signal S2A. As a result, a modulation signal having a frame configuration as shown in FIG. 35 can be transmitted.

  38 and 39 show configuration examples of the signal processing unit of the multi-antenna reception apparatus of this embodiment. FIG. 38 shows the configuration of the signal processing unit when iterative decoding is not used, and the same reference numerals are given to corresponding parts to FIG. FIG. 39 shows the configuration of the signal processing unit when iterative decoding is used.

  First, the configuration of the signal processing unit 3600 in FIG. 38 will be described. STBC symbol branch metric calculation section 4101 of signal processing section 3600 receives channel fluctuation estimation values h11, h21, h12, h22 and baseband signals R1-2, R2-2 as input, obtains a branch metric of STBC symbols, and generates STBC symbols. Branch metric signals 4102 and 4103 are output.

  At this time, two branch metric signals of STBC symbols are output. This is because branch metrics exist for S1 and S2 in the equation (2). Reference numeral 4102 denotes a branch metric signal of the STBC symbol transmitted as the modulation signal A, and reference numeral 4103 denotes a branch metric signal of the STBC transmitted as the modulation signal B.

  Separating section 501 performs signal separation according to equation (1) only for the data symbols in FIG. 35, and outputs estimated baseband signals 502 and 505.

  Data symbol branch metric calculation section 4104 receives estimated baseband signal 502 of modulated signal A, calculates a branch metric of the data symbol of modulated signal A, and outputs a branch metric signal 4105 of the data symbol. Similarly, data symbol branch metric calculation section 4106 receives estimated baseband signal 505 of modulated signal B, calculates a branch metric of the data symbol of modulated signal B, and outputs a data symbol branch metric signal 4107.

  Decoding section 4108 receives STBC symbol branch metric signal 4102 and data symbol branch metric signal 4105 as input, obtains and decodes path metric, and outputs determination value 504 for transmission digital signal TA. Similarly, decoding section 4109 receives STBC symbol branch metric signal 4103 and data symbol branch metric signal 4107 as input, obtains and decodes path metric, and outputs determination value 507 for transmission digital signal TB.

  Signal point reduction sections 508, 510, 514, and 516 perform signal point reduction on the data symbols as described in Embodiment 1, and output signal point information after signal point reduction. Data symbol branch metric calculation sections 4110 and 4112 have signal point information after signal point reduction and baseband signals R1-2 and R2-2 as inputs, and output data symbol branch metric signals 4111 and 4113. Decoding sections 4114 and 4115 receive the branch metric signal of the data symbol and the branch metric signal of the STBC symbol, obtain a path metric, and decode it.

  Next, the configuration of FIG. 39 will be described. As described above, FIG. 39 shows the configuration of the signal processing unit when iterative decoding is used, and is a combination of the configuration of FIG. 38 and the configuration of FIG. That is, the relationship between FIGS. 38 and 39 is the same as the relationship between FIGS. 5 and 13 already described. Therefore, the same reference numerals are given to the portions corresponding to FIG. 38 in FIG. 39, and the description thereof is omitted.

  Next, operations and effects at the time of reception when a space-time code is regularly transmitted as in the present embodiment will be described.

  FIG. 40 shows an example of a reception state when a signal having a frame configuration as shown in FIG. 35 is received. FIG. 40A shows the frame structure of modulated signal A. FIG. FIG. 40B shows a state example of the modulation signal A after the first determination. FIG. 40C shows the state of modulated signal B after signal point reduction. 40A and 40B are regarded as the modulation signal B, and FIG. 40C is regarded as the modulation signal A.

  Since the STBC symbol can obtain the diversity gain by encoding and the diversity gain at the receiving antenna, it is very reliable when the branch metric is obtained. The STBC symbol does not require signal point reduction as in the first to eighth embodiments. On the other hand, since the diversity gain is small in the data symbol, the reliability is low when the branch metric is obtained.

  Consider the state after the first soft decision of the modulation signal A in such characteristics. Since the reliability of the branch metric in the STBC symbol is very high, if a soft decision is made by obtaining the path metric of the STBC symbol, the possibility of obtaining a correct symbol becomes very high.

  Therefore, since the symbol determination of the modulation signal A can be performed correctly, if signal point reduction for the data symbol is performed using the determination result, the possibility of selecting an incorrect signal point is reduced. As a result, the reliability of the branch metric when the branch metric of the modulated signal B is obtained using the reduced signal points is increased.

  In addition, STBC symbols are also inserted into the modulated signal B, and the reliability of the branch metric obtained from the STBC symbols based on the diversity gain obtained by encoding the STBC symbols and the diversity gain at the receiving antenna is very high.

  With these two effects, it is possible to significantly improve the error rate characteristic of the modulated signal B when the path metric is obtained and soft decision decoding is performed.

  Considering the case of performing the iteration processing of Embodiments 3 and 4, the number of iterations for obtaining a good error rate characteristic by adopting a frame configuration in which STBC symbols are inserted as in this embodiment. As a result, the error rate characteristic is further improved. Further, if the interleave patterns of modulated signal A and modulated signal B are made different as in the sixth embodiment, the error rate characteristics are further improved. Since the configuration has been described in detail in Embodiment 6, the description thereof will be omitted here, but the point is that a plurality of interleavers having different interleave patterns are provided on the transmission side, and interleave with different interleave patterns from each antenna. In addition to transmitting the modulated signal, a deinterleaver and an interleaver corresponding to each interleaver may be provided on the receiving side.

(Ii) When transmitting special symbols Next, the principle of transmitting / receiving special symbols will be described. 41 and 42 show examples of special symbol frames.

  The frame configuration in FIG. 41 will be described in detail. In this frame configuration, a symbol 3703 composed of (0, 0) signals on the in-phase I-orthogonal Q plane is transmitted as the modulation signal B at the same time as the transmission of the data symbol 3701 as the modulation signal A. . That is, the modulated signal B is not transmitted. Also, at the same time as the time when the data symbol 3704 is transmitted as the modulation signal B, the modulation signal A is transmitted as a symbol 3702 composed of (0, 0) signals in the in-phase I-orthogonal Q plane. That is, the modulated signal A is not transmitted.

  In the example of FIG. 41, transmitting a data symbol from only one antenna and not transmitting any other antenna is referred to as a special symbol. That is, here, it is proposed to regularly transmit such special symbols instead of STBC symbols.

  Thereby, when the receiver receives the data symbol 3701 of the modulated signal A, since there is no signal in the modulated signal B, only the modulated symbol A is received by a plurality of antennas, so that diversity gain is obtained. A highly reliable branch metric can be obtained for the data symbol 3701. In addition, there is no need to perform signal point reduction. Similarly, when the receiver receives the data symbol 3704 of the modulated signal B, since there is no signal in the modulated signal A, only the modulated signal B is received by a plurality of antennas, so that diversity gain is obtained. A reliable branch metric can be obtained for the data symbol 3704. In addition, there is no need to perform signal point reduction.

  Incidentally, the data symbols 3701 and 3704 in the special symbols are encoded together with other data symbols that are temporally adjacent to this symbol. In this way, the special symbol is associated with other data symbols that precede and follow it.

  The frame configuration in FIG. 42 will be described in detail. In this frame configuration, modulated signal A is known data symbol 3801 and modulated signal B is known data symbol 3802, and these known data symbols 3801 and 3802 are transmitted at the same time. Here, the known data symbol is transmission of known data. That is, in the example of FIG. 42, transmitting known data symbols from a plurality of antennas is called a special symbol. That is, here, it is proposed to regularly transmit such special symbols instead of STBC symbols.

  Thus, when the receiver receives known data symbols 3801 and 3802 of modulated signal A and modulated signal B, each symbol can be reliably identified because these symbols are known. Therefore, sufficient diversity gain can be obtained for each modulation symbol by reception with a plurality of antennas, and a highly reliable branch metric can be obtained for each symbol. In addition, there is no need to perform signal point reduction.

  Incidentally, the known data symbols 3801 and 3802 in the special symbols are encoded together with other data symbols which are temporally and temporally related to this symbol. In this way, the special symbol is associated with other data symbols that precede and follow it.

  In FIG. 42, the example in which the known data symbol is composed of one symbol has been described. However, it may be composed of two symbols using the STBC method. In any case, it is important that the known data symbols are involved in the encoding.

  Next, a configuration example of a multi-antenna transmission apparatus for transmitting signals as shown in FIGS. 41 and 42 and a configuration example of a multi-antenna reception apparatus for receiving and demodulating the signals will be described.

  For the multi-antenna transmission apparatus, the modulators 202A and 202B in FIGS. 2 and 30 may be configured as shown in FIG. 43, for example. Since the modulation unit 202A and the modulation unit 202B may have substantially the same configuration, only the modulation unit 202A will be described here.

  Here, in the configuration of FIG. 43, the STBC symbol signal generation unit 3502 is replaced with a special symbol signal generation unit 4001 as compared with the configuration of FIG. A description thereof will be omitted. The special symbol signal generation unit 4001 receives the encoded data S1A and the frame configuration signal S10 as input, and when the frame configuration signal S10 indicates a special symbol, the special symbol signal generation unit 4001 outputs the baseband signal 4002 of the special symbol shown in FIG. 41 or FIG. Output.

  The configuration of the multi-antenna receiving apparatus that receives and demodulates a modulated signal with such a special symbol inserted may be obtained by replacing the STBC symbol branch metric calculation unit 4101 in FIGS. 38 and 39 with a special symbol branch metric calculation unit.

  FIG. 44 shows an example of a reception state when a special symbol is received. FIG. 44A shows a frame configuration of the modulation signal A. FIG. 44B shows an example of the state of the modulation signal A after the first determination. FIG. 44C shows the state of the modulation signal B after signal point reduction. 44 (A) and 44 (B) are the modulation signal B, and FIG. 44 (C) is the modulation signal A.

  If a special symbol is inserted in the same manner as when an STBC symbol is inserted, the reliability of the branch metric in the special symbol is very high. Therefore, if a path metric of the special symbol is obtained and a soft decision is performed, a correct symbol may be obtained. Becomes very high.

  Therefore, since the symbol determination of the modulation signal A can be performed correctly, if signal point reduction for the data symbol is performed using the determination result, the possibility of selecting an incorrect signal point is reduced. As a result, the reliability of the branch metric when the branch metric of the modulated signal B is obtained using the reduced signal points is increased.

  In addition, a special symbol is also inserted into the modulated signal B, and the reliability of the branch metric obtained with the special symbol is very high due to the diversity gain obtained by encoding the special symbol and the diversity gain at the receiving antenna.

  With these two effects, it is possible to significantly improve the error rate characteristic of the modulated signal B when the path metric is obtained and soft decision decoding is performed.

  Considering the case of performing the iteration processing of Embodiments 3 and 4, the number of iterations for obtaining a good error rate characteristic by adopting a frame configuration in which special symbols are inserted as in this embodiment. As a result, the error rate characteristic is further improved. Further, if the interleave patterns of modulated signal A and modulated signal B are made different as in the sixth embodiment, the error rate characteristics are further improved.

(Iii) Other Configuration Examples In the above-described embodiment, the STBC symbol is inserted at the position shown in FIG. 40 and the special symbols 3601 and 3602 are inserted at the positions shown in FIG. And the insertion position and interval of the special symbol are not limited to this. Symbols to be inserted between data symbols are not limited to STBC symbols and special symbols shown in FIGS. 41 and 42, but may be applied to symbols that have high branch metric reliability and do not require signal point reduction. With such a symbol, the same effect as described above can be obtained.

  Symbols with high branch metric reliability (STBC symbols in FIG. 40 and special symbols in FIG. 44) to be inserted can also be called pilot symbols for obtaining branch metrics with high reliability.

  In the above-described embodiment, the example applied to the spread spectrum communication system has been described. However, the present invention is not limited to this. For example, the present invention can also be applied to the OFDM system. In this case, the encoding can be performed in the direction along the time axis as shown in FIGS. 40 and 44. Also, the horizontal axis in FIGS. 40 and 44 is considered as the frequency axis, and the frequency axis is used. It is also possible to encode. In addition, encoding in both the time axis and the frequency axis is also possible. Of course, the present invention can also be applied to a single carrier system that is not a spread spectrum communication system.

  Further, the configuration of the receiving apparatus is not limited to the configurations of FIGS. 38 and 39, and for example, a configuration in which the modulation signal A and the modulation signal B are alternately demodulated as shown in FIGS. In this case, the circuit scale can be reduced as compared with the configurations of FIGS.

(Embodiment 10)
In this embodiment, it is proposed to always switch the antenna that transmits the modulation signal within the coding block. As a result, since the steady state due to the influence of the direct wave can be changed, it is possible to avoid a situation where the error rate characteristic is deteriorated over the entire coding block and to be brought into a state where the error rate characteristic is good. Become.

First, the principle of this embodiment will be described. Consider the outlook propagation environment. At this time, the channel matrix in the equation (1) includes channel elements h _{11, d} , h _{12, d} , h _{21, d} , h _{22, d of} direct wave components and channel elements h _{11, s} , h of scattered wave components. _{12, s 1} , h _{21, s 1} , h _{22, s} can be considered separately, and can be expressed as the following equation.

  It is known that when a channel element of a direct wave falls into a steady state, the reception field intensity is the same depending on the state, but the reception quality is completely different (for example, the MIMO system in the literature “Rice fading”). Analysis of “The Institute of Electronics, Information and Communication Engineers, IEICE Technical Report RCS 2003-90, pp. 1-6, July 2003”. In particular, in a line-of-sight environment in which direct waves are dominant, there is a possibility that a steady state is obtained in which the effect of having different interleave patterns between modulated signals as in the sixth embodiment does not sufficiently appear. In such a state, it is considered that good error rate characteristics cannot be obtained even if the received electric field strength is sufficient. This embodiment has been made based on such consideration.

  First, the description of the encoded symbol block is started. FIG. 45 shows an example of the configuration of the encoded symbol block and the transmission order in the present embodiment. FIG. 45A shows an example of the configuration of a coded symbol block. The encoded symbol has a finite length. An encoded symbol block means a block composed of a finite length (here, composed of 300 symbols). The numbers 1, 2,..., 299, 300 indicate the order of data encoding. When interleaving is performed, for example, data is transmitted in the order shown in FIG. 45B by dividing the data into units of 100 symbols and reading from the vertical direction of FIG.

  By the way, in the case of an environment where the direct wave is dominant, even when a conventional modulated signal that is not MIMO communication is transmitted in one system, the propagation environment is small and the interleaving effect is small, but the reception electric field strength is sufficient. Therefore, good reception quality (error rate characteristics) can be obtained.

  On the other hand, in the case of MIMO communication, since the fluctuation of the propagation environment is small in an environment where the direct wave is dominant, the effect of the interleaving is small as in the conventional case, but the difference is that the received electric field strength is not sufficient. Even so, depending on the state of the direct wave matrix in equation (3), the reception quality may deteriorate.

  Therefore, in the present embodiment, the antenna that transmits the modulated signal is always switched once in the coding block. A specific frame configuration example is shown in FIG. The modulated signal A is interleaved as shown in FIG. 45 (B), and FIG. 45 (B) is divided into three (hereinafter, each divided block is referred to as an XA block, a YA block, and a ZA block). Make sure that one of the blocks transmits from another antenna.

  For example, as shown in FIG. 46, in the modulated signal A, when the XA block corresponds to the data symbol 4402, the YA block corresponds to the data symbol 4404, and the ZA block corresponds to the data symbol 4406, the data symbols 4402 and 4404 (that is, the XA block). And YA block) are transmitted from the same antenna AN1, but the data symbol 4406 (that is, the ZA block) switches the transmitting antenna to another antenna AN2.

  Similarly, the modulation signal B is also interleaved as shown in FIG. 45B (however, as described in the sixth embodiment, the modulation signal B has an interleaving pattern different from that shown in FIG. 45B). 45B is divided into three (hereinafter, the divided blocks are referred to as XB block, YB block, and ZB block), and one of the divided blocks is always Try to transmit from another antenna.

  For example, as shown in FIG. 46, in the modulated signal B, when the XB block corresponds to the data symbol 4408, the YB block corresponds to the data symbol 4410, and the ZB block corresponds to the data symbol 4412, the data symbol 4408 (ie, XB block) is The data is transmitted from the antenna AN2, but the data symbols 4410 and 4412 (that is, the YB block and the ZB block) are transmitted from another antenna AN3.

  Here, when the modulated signals A and B are transmitted by the antennas AN1 and AN2, the state of the matrix that is stationary due to the influence of the direct wave is poor, and therefore the reliability of the branch metric is low even if the received electric field strength is sufficient. And Similarly, when the modulated signals A and B are transmitted by the antennas AN1 and AN3, the state of the matrix that is steady due to the influence of the direct wave is poor, and therefore the reliability of the branch metric is sufficient even if the received electric field strength is sufficient. Is low.

  On the other hand, when the modulated signals A and B are transmitted by the antennas AN2 and AN3, the state of the matrix that is stationary due to the influence of the direct wave is good, and therefore, the reliability of the branch metric is high.

  Thus, the state of the matrix when it falls into the steady state due to the direct wave can be changed by switching the antenna that transmits the modulated signal. As a result, the reliability of the branch metric can be changed by switching the antenna that transmits the modulated signal. Specifically, only a branch metric with low reliability can be obtained in the periods t1 and t2 in FIG. 46, but a branch metric with high reliability can be obtained in the period t3. Incidentally, when the antenna that transmits the modulated signal is switched, the received field strength does not change, but the state of the matrix changes. However, when the transmission antenna selection patterns are the same, the state is almost the same.

  In addition, since the antenna for transmitting the modulation signal is switched in the coding block, the reliable branch metric and the low branch metric can be rearranged at random in the coding block by deinterleaving. As a result, when a path metric is obtained and decoded, data with a certain degree of reliability can be obtained. When iterative decoding using signal point reduction and iteratively decoding data based on data with a certain degree of reliability, it becomes possible to obtain sufficiently reliable data.

  FIG. 47 shows a configuration example of the multi-antenna transmission apparatus of this embodiment. 47 corresponding to those in FIG. 2 are denoted by the same reference numerals. Antenna selection section 4501 of multi-antenna transmission apparatus 4500 receives modulated signals Ta and Tb and frame configuration signal S10 as input, and selects antennas AN1 to AN3 that transmit modulated signals Ta and Tb according to frame configuration signal S10. Thereby, the modulation signal having the frame configuration shown in FIG. 46 can be transmitted.

  Thus, according to the present embodiment, the steady state due to the influence of the direct wave can be changed by switching the antenna that transmits the modulation signal at least once in the coding block. It becomes possible to draw in a state where characteristics are improved. As a result, when combined with the features of Embodiments 1 to 9 described above, it is possible to obtain received data with better error rate characteristics. Incidentally, in order to draw a state with good error rate characteristics, it is effective to select different interleave patterns between modulated signals or to apply iterative decoding by signal point reduction.

(Embodiment 11)
In this embodiment, when transmitting modulated signals having different interleave patterns from the respective antennas, it is proposed to form modulated signals having different interleave patterns using bit interleaving. Furthermore, a bit interleaving method is proposed so that received data with good error rate characteristics can be obtained when considering signal point reduction on the receiving side.

  FIG. 48 shows an example of signal point arrangement on the IQ plane of modulated signal A and modulated signal B transmitted by the multi-antenna transmitting apparatus of the present embodiment.

  49, in which parts corresponding to those in FIG. 2 are assigned the same reference numerals, shows the configuration of the multi-antenna transmission apparatus of this embodiment. Here, when 16QAM is performed in modulation section 202A of multi-antenna transmission apparatus 4700, the signal point arrangement of modulated signal A (baseband signal 2A) is as shown in FIG. Specifically, any of the 16 points in FIG. 48A is assigned according to the four encoded bits Sa0, Sa1, Sa2, and Sa3 obtained by encoding the transmission digital signal TA.

  Similarly, when 16QAM is performed in modulation section 202B, the signal point arrangement of modulated signal B (baseband signal 2B) is as shown in FIG. Specifically, any of the 16 points in FIG. 48B is assigned according to the four encoded bits Sb0, Sb1, Sb2, and Sb3 obtained by encoding the transmission digital signal TB.

  Multi-antenna transmission apparatus 4700 inputs transmission digital signal TA to signal separation section 4701. The signal separation unit 4701 separates the transmission digital signal TA into a digital signal 4702 and a digital signal 4703, sends the digital signal 4702 to the (Sa0, Sa2) encoding unit 4704 and also uses the digital signal 4703 for (Sa1, Sa3). The data is sent to the encoding unit 4706. The (Sa0, Sa2) encoding unit 4704 encodes the digital signal 4702 to obtain an encoded bit string 4705 including encoded bits Sa0 and Sa2, and transmits this to the interleaver 4708. The (Sa1, Sa3) encoding unit 4706 encodes the digital signal 4703 to obtain an encoded bit string 4707 composed of encoded bits Sa1 and Sa3, and sends this to the interleaver 4710.

  Similarly, multi-antenna transmission apparatus 4700 inputs transmission digital signal TB to signal separation unit 4712. The signal separation unit 4712 separates the transmission digital signal TB into a digital signal 4713 and a digital signal 4714, sends the digital signal 4713 to the (Sb0, Sb2) encoding unit 4715, and uses the digital signal 4714 for (Sb1, Sb3). The data is sent to the encoding unit 4717. The (Sb0, Sb2) encoding unit 4715 encodes the digital signal 4713 to obtain an encoded bit string 4716 composed of encoded bits Sb0 and Sb2, and sends this to the interleaver 4719. The (Sb1, Sb3) encoding unit 4717 encodes the digital signal 4714 to obtain an encoded bit string 4718 composed of encoded bits Sb1, Sb3, and sends this to the interleaver 4721.

  Interleavers 4708 and 4710 obtain encoded bit sequences 4709 and 4711 by performing bit interleaving on the encoded bit sequences 4705 and 4707, respectively, and send these to modulation section 202A. Similarly, the interleavers 4719 and 4721 obtain encoded bit sequences 4720 and 4722 by bit interleaving the encoded bit sequences 4716 and 4718, respectively, and send them to the modulation unit 202B.

  In this embodiment, the interleave pattern of the interleaver 4708 and the interleaver 4719 is the same interleave pattern X, and the interleave pattern of the interleaver 4710 and the interleaver 4721 is the same interleave pattern Y.

  In this way, signal point reduction is performed on the receiving side by making the same set of bit interleave patterns between modulated signals, rather than making all bit interleave patterns for each modulated signal transmitted from each antenna different. Received data with good error rate characteristics can be obtained. The reason will be described later.

  An example of bit interleaving by the interleavers 4708, 4710, 4719, and 4721 is shown in FIG. FIG. 50 shows the order of data before and after interleaving.

  Data 1, data 2,..., Data 200 are added to the order before the interleaving of the coded bits Sa0 and Sa2 for the modulation signal A. If the interleaver 4708 performs bit interleaving for rearranging the order every five data, the data 1, data 6,. Next, data 2, data 7,..., Data 197 are arranged. .., Data 198, data 4, data 9,..., Data 199, data 5, data 10,. A similar rearrangement is performed by the interleaver 4719 for the data of the coded bits Sb0 and Sb2 for the modulation signal B as well.

  Further, the order of data 1, data 2,..., Data 200 is added to the order before the interleaving of the coded bits Sa1 and Sa3 for the modulation signal A. Here, assuming that the interleaver 4710 performs bit interleaving for rearranging the order every eight data, first, the data 1, the data 9,. Next, the data 2, data 10,..., Data 194 are arranged. Hereinafter, data 3, data 11, ..., data 195, then data 4, data 12, ..., data 196, then data 5, data 13, ..., data 197, then .., Data 198, then data 7, data 15,... Data 199, then data 8, data 16,. Similar rearrangement is performed by the interleaver 4721 for the data of the coded bits Sb1 and Sb3 for the modulation signal B as well.

  Next, the configuration and operation of the multi-antenna reception apparatus of this embodiment will be described. The overall configuration of the multi-antenna receiving apparatus is the same as in FIG. However, a signal processing unit 4900 configured as shown in FIG. 51 is provided as the signal processing unit 404 of FIG.

  In FIG. 51, in which parts corresponding to those in FIG. 24 are assigned the same reference numerals, the signal processing unit 4900 rearranges the order of the transmission digital signal A by the deinterleaver 2401A for the encoded bits Sa0 and Sa2. The band signal 502 is obtained, and then soft decision decoding is performed by the soft decision decoding unit 503 for the coded bits Sa0 and Sa2, thereby obtaining information of the coded bits Sa0 and Sa2. Next, the order is rearranged by the interleaver 2402A for the coded bits Sa0 and Sa2, and the coded bit string 504 of the coded bits Sa0 and Sa2 after the interleaving is output.

  The same applies to the modulated signal B, and the estimated baseband signal 505 is obtained by rearranging the order of the transmission digital signal B by the deinterleaver 2401B for the coded bits Sb0 and Sb2, and then for the coded bits Sb0 and Sb2. The soft decision decoding unit 506 performs soft decision decoding to obtain information on the encoded bits Sb0 and Sb2. Next, the order is rearranged by the interleaver 2402B for the encoded bits Sb0 and Sb2, and the encoded bit string 507 of the encoded bits Sb0 and Sb2 after the interleaving is output.

  Next, processing of the signal point reduction units 1301 and 1302 that operate will be described with reference to FIG.

  FIG. 52 (A) shows candidate signal points before signal point reduction (O: candidate signal points), and the candidate signal points transmit 256 bits in this embodiment, so 256 candidates. There will be signal points. Since 4 bits are determined from the interleaved encoded bits Sa0 and Sa2 information 504 and the interleaved encoded bits Sb0 and Sb2 information 507, the signal point reduction units 1301 and 1302 As in 52 (B), 256 candidate signal points are reduced to 16 candidate signal points.

  Likelihood determination section 4901 obtains the square of Euclidean distance between the 16 candidate signal points and the received baseband signal (■) as shown in FIG. Since the branch metric is obtained for each antenna, two systems are obtained. The likelihood determination unit 4901 obtains the sum of the branch metrics obtained for each antenna and encodes the coded bit Sa1 based on the branch metrics. , Sa3, Sb1, and Sb3 are output, and encoded bits Sa1 and Sa3 are output to deinterleaver 4902 and encoded bits Sb1 and Sb3 are output to deinterleaver 4905.

  The deinterleaver 4902 rearranges the order of the encoded bits Sa1 and Sa3, and sends the encoded bits Sa1 and Sa3 after deinterleaving to the decoding unit 4903. The decoding unit 4903 outputs information 4904 of the encoded bits Sa1 and Sa3 after error correction by performing, for example, hard decision decoding on the encoded bits Sa1 and Sa3 after deinterleaving.

  Similarly, the deinterleaver 4905 rearranges the order of the encoded bits Sb1 and Sb3, and sends the encoded bits Sb1 and Sb3 after deinterleaving to the decoding unit 4906. Decoding section 4906 outputs information 4907 of encoded bits Sb1 and Sb3 after error correction, for example, by performing hard decision decoding on the encoded bits after deinterleaving.

  Thus, the encoded bit Sa3 is obtained from the encoded bit Sa0, and the encoded bit Sb3 is obtained from the encoded bit Sb0. At this time, since signal points are reduced, it is possible to use only 16 operations to calculate 256 times of Euclidean distance for each antenna. Reduction can be achieved.

  Hereinafter, in order to further improve the reception quality, a method of applying iterative decoding and the reason why the interleave pattern is made different as described above will be described in detail.

  First, the application method of iterative decoding will be described in detail. Interleaver 4908 receives encoded bits Sa1 and Sa3 after error correction obtained as described above, performs interleaving for encoded bits Sa1 and Sa3, and outputs interleaved encoded bits Sa1 and Sa3 as signal points. The data is sent to the reduction units 1303 and 1304. Similarly, interleaver 4909 receives coded bits Sb1 and Sb3 after error correction obtained as described above, performs interleaving for coded bits Sb1 and Sb3, and coded bits Sb1 and Sb3 after interleaving. Is sent to signal point reduction sections 1303 and 1304.

  The signal point reduction units 1303 and 1304 have the interleaved encoded bits Sa1 and Sa3 information and the information of the interleaved encoded bits Sb1 and Sb3 as inputs, and 256 candidate signal points as shown in FIG. Using the 4 bits determined by the interleaved encoded bits Sa1 and Sa3 and the interleaved encoded bits Sb1 and Sb3, the number is reduced to 16 candidate signal points.

  Likelihood determination section 4910 obtains the square of the Euclidean distance between the 16 candidate signal points and the received baseband signal (■) as shown in FIG. Since the branch metric is obtained for each antenna, two systems are obtained. The likelihood determination unit 4910 obtains the sum of the branch metrics obtained for each antenna and encodes the coded bit Sa0 based on the branch metrics. , Sa2, Sb0, Sb2 are determined, and the encoded bits Sa0, Sa2 are output to the deinterleaver 4911 and the encoded bits Sb0, Sb2 are output to the deinterleaver 4914.

  The deinterleaver 4911 rearranges the order of the encoded bits Sa0 and Sa2, and sends the encoded bits Sa0 and Sa2 after deinterleaving to the decoding unit 4912. The decoding unit 4912 outputs information 4913 of the encoded bits Sa0 and Sa2 after error correction by performing, for example, hard decision decoding on the encoded bits Sa0 and Sa2 after deinterleaving.

  Similarly, the deinterleaver 4914 rearranges the order of the encoded bits Sb0 and Sb2, and sends the encoded bits Sb0 and Sb2 after deinterleaving to the decoding unit 4915. The decoding unit 4915 outputs information 4916 of the encoded bits Sb0 and Sb2 after error correction, for example, by performing hard decision decoding on the encoded bits Sb0 and Sb2 after deinterleaving.

  As described above, information 4913 and 4916 of the coded bits Sa0, Sa2, Sb0, and Sb2 with improved reception quality (error rate characteristics) are obtained.

  Further, interleaver 4917 receives information 4913 of coded bits Sa0 and Sa2 after error correction, and sends the information of coded bits Sa0 and Sa2 after interleaving to signal point reduction sections 1301 and 1302. Similarly, interleaver 4918 receives information 4916 of encoded bits Sb0 and Sb2 after error correction, and sends the information of encoded bits Sb0 and Sb2 after interleaving to signal point reduction sections 1301 and 1302.

  Then, the above-described operations are performed in the signal point reduction sections 1301 and 1302, the likelihood determination section 4901, the deinterleavers 4902 and 4905, and the decoding sections 4903 and 4906, so that the coded bits Sa1, Sa3 with improved reception quality are obtained. Information 4904, 4907 of Sb1, Sb3 is obtained.

  The reception quality can be improved by performing the above operations a plurality of times. A flowchart of these processes is shown in FIG.

  First, the coded bits Sa0 and Sa2 of the modulated signal A and the coded bits Sb0 and Sb2 of the modulated signal B are decoded (ST21A). Next, signal point reduction is performed based on the information of the obtained coded bits Sa0, Sa2, Sb0, Sb2 (ST21B), and the coded bits Sa1, Sa3 and the coded bits Sb1, Sb3 are decoded (ST22B). Next, signal point reduction is performed based on the information of the obtained coded bits Sa1, Sa3, Sb1, and Sb3 (ST22A), and the coded bits Sa0 and Sa2 and the coded bits Sb0 and Sb2 are decoded (ST23A). Thereafter, the same processing is repeated.

  In the present embodiment, the interleave pattern of the interleaver for the coded bits Sa0 and Sa2 and the interleaver for the coded bits Sb0 and Sb2 are the same, and the interleaver for the coded bits Sa1 and Sa3 and the coded bit Sb1 , Sb3 interleaver has the same interleave pattern. The effect of this is that the error rate at the time of signal point reduction can be made smaller than making all interleave patterns different.

  However, in essence, the interleaver for the coded bits Sa0 and Sa2 and the interleaver for the coded bits Sa1 and Sa3 are made different from each other in order to improve reception quality. The reason will be described in detail below.

  In FIG. 54, the interleave pattern of the interleaver for the coded bits Sa0 and Sa2 and the interleaver for the coded bits Sb0 and Sb2 is the same, and the interleaver for the coded bits Sa1 and Sa3 and the coded bits Sb1 and Sb3 are used. 2 shows an example of a reception state when the interleave patterns of the interleavers are the same and the interleave patterns for the coded bits Sa0 and Sa2 and the interleavers for the coded bits Sa1 and Sa3 are the same. That is, this is an example in which all interleavers have the same interleave pattern.

  As a result of decoding the coded bits Sa0 and Sa2 in the soft decision unit 503 in FIG. 51 under such an interleave pattern, symbols that are erroneously determined are continuously generated as shown in FIG. 54 (A). Suppose you did. Incidentally, when a convolutional code or the like is used, it is common that errors occur continuously. Then, when the signal point reduction units 1301 and 1302 reduce the number of signal points, as shown in FIG. 54B, an error occurs continuously in signal point selection by signal point reduction. As a result, even if the decoding units 4903 and 4906 decode the coded bits Sa1 and Sa3 and the coded bits Sb1 and Sb3, the reception quality (error rate characteristics) is not effectively improved. This is because error correction codes have a low ability to correct continuous errors.

  In FIG. 55, as in the present embodiment, the interleaver for the coded bits Sa0 and Sa2 and the interleaver for the coded bits Sb0 and Sb2 are the same, and the interleaver for the coded bits Sa1 and Sa3. And the interleaver pattern for the coded bits Sb1 and Sb3 are the same, and the interleaver pattern for the coded bits Sa0 and Sa2 and the interleaver for the coded bits Sa1 and Sa3 are different. An example of the reception state is shown.

  As a result of decoding the coded bits Sa0 and Sa2 in the soft decision unit 503 in FIG. 51 under such an interleave pattern, symbols that have been erroneously determined are continuously as shown in FIG. Assume that it has occurred. Then, when the number of signal points is reduced in the signal point reduction units 1301 and 1302, the interleave pattern of the coded bits Sa0 and Sa2 and the interleave pattern of the coded bits Sa1 and Sa3 are different from FIG. 55B. As a result of deinterleaving, signal point selection errors due to signal point reduction occur discretely as shown in FIG. 55 (B). That is, the signal point selection error due to signal point reduction does not occur continuously as shown in FIG. As a result, when the coded bits Sa1 and Sa3 and the coded bits Sb1 and Sb3 are decoded by the decoding units 4903 and 4906, the error rate characteristics are effectively improved. This is because error correction codes have a high ability to correct discrete errors.

  Further, since the interleave pattern for the coded bits Sa0 and Sa2 and the interleaver for the coded bits Sb0 and Sb2 are the same, the occurrence of errors in the coded bits Sa0 and Sa2 and the coded bits Sb0 and Sb2 is the same. Can be.

  For example, the error probability of the encoded bits Sa0 and Sa2 is 1/100, and the error probability of the encoded bits Sb0 and Sb2 is 1/100. At this time, if the occurrence of errors in the coded bits Sa0 and Sa2 and the coded bits Sb0 and Sb2 is the same, the probability of erroneous signal point reduction is 1/100. However, if the error occurrence patterns are different, the probability of signal point reduction error is 1/100 + 1/100 = 1/50. When the interleave pattern is different, the possibility that the error generation pattern is different increases.

  In this way, considering the signal point reduction, it is preferable that the interleave patterns of the interleaver for the coded bits Sa0 and Sa2 and the interleaver for the coded bits Sb0 and Sb2 are the same.

  However, even if all the interleave patterns are made different, signal point selection errors due to signal point reduction can be caused discretely as described above, so that the error rate characteristics can be improved in the same manner. That is, it is not an essential requirement to make all interleave patterns different, and if all are not the same pattern, the same is true in that signal point selection errors due to signal point reduction can be generated discretely. The effect can be demonstrated.

  Thus, according to the present embodiment, the bit interleave pattern of the modulated signal transmitted from each antenna is made different, so that the influence of the burst error is reduced at the time of decoding on the receiving side, and the error rate characteristic is good. A multi-antenna transmission apparatus capable of obtaining received data can be realized.

  Also, among the plurality of interleavers provided for each modulation signal, the same interleave pattern pair (interleaver 4708 and interleaver 4719, interleaver 4710 and interleaver 4721) is created by the interleaver between the modulation signals. As a result, the probability of an error occurring at the time of signal point reduction can be reduced, so that received data with better error rate characteristics can be obtained.

  In the above-described embodiment, the example applied to the spread spectrum communication system has been described. However, the present invention is not limited to this. For example, the present invention can also be applied to a single carrier system or OFDM system that is not a spread spectrum communication system. When applied to the OFDM system, the encoding method can be a method of encoding in the direction along the time axis as shown in FIG. 54, and the horizontal axis in FIG. 54 is considered as the frequency axis, and encoding is performed along the frequency axis. It is also possible to do. In addition, encoding in both the time axis and the frequency axis is also possible.

  At this time, as described in the seventh embodiment, the interleave pattern X is a pattern in which data is rearranged from a high-frequency subcarrier to a low subcarrier, and the interleave pattern Y is a low-frequency subcarrier to a high subcarrier. If the data is rearranged and arranged on the carrier, the error rate characteristics can be improved effectively, and the circuit configuration can be simplified.

  Furthermore, the interleaving method illustrated in this embodiment is merely an example, and the present invention is not limited to this. Ideally, random interleaving is appropriate.

  Moreover, although the case where the interleaving is performed by the interleaver has been described in the above-described embodiment, for example, when applied to the OFDM scheme, the interleaving may be performed by the mapping process to each subcarrier. In this way, the interleaver can be omitted.

(Embodiment 12)
In this embodiment, a bit interleaving method different from that in Embodiment 11 is proposed.

  FIG. 56, in which parts corresponding to those in FIG. 2 are assigned the same reference numerals, shows the configuration of the multi-antenna transmission apparatus of this embodiment. The multi-antenna transmission apparatus 5400 of this embodiment is different from the multi-antenna transmission apparatus 4700 of the eleventh embodiment in that the encoded bits Sa0 and Sa2 of the modulated signal A are encoded together with the encoded bits Sb0 and Sb2 of the modulated signal B. In addition, the encoded bits Sa1 and Sa3 of the modulated signal A are encoded together with the encoded bits Sb1 and Sb3 of the modulated signal B.

  Multi-antenna transmission apparatus 5400 inputs transmission digital signal 5401 to signal separation section 5402. The signal separation unit 5402 separates the transmission digital signal 5401 into two systems of a digital signal 5403 and a digital signal 5404, sends the digital signal 5403 to the encoding unit 5405, and sends the digital signal 5404 to the encoding unit 5412. The encoding unit 5405 encodes the digital signal 5403 (for example, convolutional encoding) to obtain an encoded bit string 5406, and transmits this to the interleaver 5407. The encoding unit 5412 encodes the digital signal 5404 (for example, convolutional encoding) to obtain an encoded bit string 5413, and sends this to the interleaver 5415.

  The interleaver 5407 interleaves the encoded bit string 5406 with the interleave pattern X, and sends the encoded bit string 5408 after the interleaving to the separating unit 5409. Separating section 5409 separates interleaved encoded bit string 5408 into encoded bit string 5410 including encoded bits Sa0 and Sa2 and encoded bit string 5411 including encoded bits Sb0 and Sb2, and converts encoded bit string 5410 to modulator The encoded bit string 5411 is sent to the modulation unit 202B to 202A.

  The interleaver 5415 interleaves the encoded bit string 5413 with the interleave pattern Y, and sends the encoded bit string 5416 after the interleaving to the separating unit 5417. The separation unit 5417 separates the interleaved encoded bit string 5416 into an encoded bit string 5418 including encoded bits Sa1 and Sa3 and an encoded bit string 5419 including encoded bits Sb1 and Sb3, and the encoded bit string 5418 is modulated. The encoded bit string 5419 is sent to the modulation unit 202B in 202A.

  Modulation section 202A assigns any of the 16 points in FIG. 48A according to coded bits Sa0 and Sa2 and coded bits Sa1 and Sa3, and outputs baseband signal S2A indicating the signal point. . Similarly, modulation section 202B assigns any one of 16 points in FIG. 48B according to coded bits Sb0 and Sb2 and coded bits Sb1 and Sb3, and provides baseband signal S2B indicating the signal point. Output.

  Here, the interleave pattern X of the interleaver 5407 and the interleave pattern Y of the interleaver 5415 are different. This makes it possible to obtain received data with good error rate characteristics when signal point reduction is performed on the receiving side.

  An example of bit interleaving by the interleavers 5407 and 5415 is shown in FIG. FIG. 57 shows the order of data before and after interleaving.

  FIG. 57 (A) shows an interleaving method of interleaving pattern X. Data 1, data 2,..., Data 200 are assigned to the order before interleaving. Here, the interleaver 5407 rearranges the order every five data (this processing is the same as the interleaving processing of the coded bits Sa0 and Sa2 in FIG. 50 described in the eleventh embodiment, and thus detailed description is omitted). To do).

  Then, the data arranged after interleaving is alternately allocated to the coded bits Sa0 and Sa2 and the coded bits Sb0 and Sb2. Therefore, the data of the coded bits Sa0, Sa2 are in the order of data 1, data 11,..., Data 185, data 195, and the data of the coded bits Sb0, Sb2 are data 6, data 16,. The order is data 190 and data 200.

  FIG. 57B shows an interleaving method for interleaving pattern Y. Data 1, data 2,..., Data 200 are assigned to the order before interleaving. Here, interleaver 5415 rearranges the order every eight data (this process is the same as the interleaving process of coded bits Sa1 and Sa3 in FIG. (Omitted).

  Then, the data arranged after the interleaving is alternately distributed to the coded bits Sa1, Sa3 and the coded bits Sb1, Sb3. Therefore, the data of the encoded bits Sa1, Sa3 are in the order of data 1, data 17,..., Data 176, data 192, and the data of the encoded bits Sb0, Sb2 are data 9, data 25,. The order is data 184 and data 200.

  Next, the configuration and operation of the multi-antenna reception apparatus of this embodiment will be described. The overall configuration of the multi-antenna receiving apparatus is the same as in FIG. However, a signal processing unit 5600 configured as shown in FIG. 58 is provided as the signal processing unit 404 of FIG.

  In FIG. 58 in which the same reference numerals are assigned to the corresponding parts in FIG. 13, the signal processing unit 5600 uses the deinterleaver 5601 for the pattern X to estimate the estimated baseband signal 504 of the modulated signal A and the estimated base of the modulated signal B. The band signals 505 are rearranged. The estimated baseband signal after deinterleaving is sent to soft decision decoding section 5602.

  Soft decision decoding section 5602 performs soft decision decoding processing on the deinterleaved estimated baseband signal to obtain information 5603 of coded bits Sa0 and Sa2 and coded bits Sb0 and Sb2, and this is obtained as pattern X To the interleaver 5604. The interleaver 5604 for the pattern X interleaves the pattern X with respect to the information 5603 of the encoded bits Sa0 and Sa2 and the encoded bits Sb0 and Sb2, and outputs the encoded bits Sa0 and Sa2 and the encoded bits Sb0, Sb0, Sb2 information 5605 is sent to signal point reduction sections 1301 and 1302.

  Next, processing of the signal point reduction units 1301 and 1302 that operate will be described with reference to FIG.

  FIG. 52 (A) shows candidate signal points before signal point reduction (O: candidate signal points), and the candidate signal points transmit 256 bits in this embodiment, so 256 candidates. There will be signal points. Then, since 4 bits are determined from the information 5605 of the encoded bits Sa0 and Sa2 and the encoded bits Sb0 and Sb2 after the interleaving, the signal point reduction units 1301 and 1302 are as shown in FIG. The 256 candidate signal points are reduced to 16 candidate signal points.

  Likelihood determination section 5606 obtains the square of Euclidean distance between the 16 candidate signal points and the received baseband signal (■) as shown in FIG. Since the branch metric is obtained for each antenna, two systems are obtained. The likelihood determination unit 5606 obtains the sum of the branch metrics obtained for each antenna and encodes the coded bit Sa1 based on the branch metrics. , Sa3 and encoded bits Sb1 and Sb3 are determined and output to the pattern Y deinterleaver 5607.

  The deinterleaver 5607 for pattern Y rearranges the order of information of the coded bits Sa1, Sa3 and coded bits Sb1, Sb3, and information of the coded bits Sa1, Sa3 and coded bits Sb1, Sb3 after deinterleaving Is sent to the decoding unit 5608. The decoding unit 5608 performs, for example, hard decision decoding on the information of the coded bits Sa1 and Sa3 and the coded bits Sb1 and Sb3 after deinterleaving, thereby performing coded bits Sa1 and Sa3 and coded bits Sb1, The information 5609 of Sb3 is output.

  As described above, the coded bit Sa3 is obtained from the coded bit Sa0, and the coded bit Sb3 is obtained from the coded bit Sb0. At this time, since signal points are reduced, it is possible to use only 16 operations to calculate 256 times of Euclidean distance for each antenna. Reduction can be achieved.

  Hereinafter, in order to further improve the reception quality, a method of applying iterative decoding and the reason why the interleave pattern is made different as described above will be described in detail.

  First, the application method of iterative decoding will be described in detail. The pattern Y interleaver 5610 performs pattern X interleaving on the information 5609 of the encoded bits Sa1 and Sa3 and the encoded bits Sb1 and Sb3 after error correction obtained as described above, and performs encoding after interleaving. Information 5611 of bits Sa1 and Sa3 and coded bits Sb1 and Sb3 is sent to signal point reduction sections 1303 and 1304.

  The signal point reduction sections 1303 and 1304 receive the interleaved encoded bits Sa1 and Sa3 and the interleaved encoded bits Sb1 and Sb3 information 5611 as input, and convert 256 candidate signal points after the interleave as shown in FIG. Using 4 bits determined by the coded bits Sa1 and Sa3 and the coded bits Sb1 and Sb3, the number is reduced to 16 candidate signal points.

  Likelihood determination section 5612 obtains the square of the Euclidean distance between the 16 candidate signal points and the received baseband signal (■) as shown in FIG. Since the branch metric is obtained for each antenna, two systems are obtained. The likelihood determination unit 5612 obtains the sum of the branch metrics obtained for each antenna and encodes the coded bit Sa0 based on the branch metrics. , Sa2, Sb0, Sb2 are determined, and the information of the coded bits Sa0, Sa2, Sb0, Sb2 is sent to the deinterleaver 5613 for the pattern X.

  The deinterleaver 5613 for the pattern X rearranges the order of information of the encoded bits Sa0 and Sa2 and the encoded bits Sb0 and Sb2, thereby allowing the encoded bits Sa0 and Sa2 and the encoded bits Sb0 and Sb2 after deinterleaving. Is sent to the decoding unit 5614. The decoding unit 5614 performs, for example, hard decision decoding on the encoded bits Sa0 and Sa2 and the encoded bits Sb0 and Sb2 information 5615 after deinterleaving, thereby performing error correction of the encoded bits Sa0 and Sa2 and Sb0 and Sb2. Information 5615 is output.

  As described above, information 5609 and 5615 of coded bits Sa0, Sa2, Sb0, and Sb2 with improved reception quality (error rate characteristics) are obtained.

  Further, the interleaver 5616 receives the information 5615 of the encoded bits Sa0 and Sa2 and the encoded bits Sb0 and Sb2 after error correction, and receives the information of the encoded bits Sa0 and Sa2 and the encoded bits Sb0 and Sb2 after the interleaving. It is sent to signal point reduction sections 1301 and 1302.

  Then, the above-described operations are performed by the signal point reduction sections 1301 and 1302, the likelihood determination section 5606, the deinterleaver 5607, and the decoding section 5608, so that the coded bits Sa1, Sa3, Sb1, and Sb3 with improved reception quality are obtained. Information 5609, 5615 is obtained.

  The reception quality can be improved by performing the above operations a plurality of times. A flowchart of these processes is shown in FIG.

  First, the coded bits Sa0 and Sa2 of the modulated signal A and the coded bits Sb0 and Sb2 of the modulated signal B are decoded (ST21A). Next, signal point reduction is performed based on the information of the obtained coded bits Sa0, Sa2, Sb0, Sb2 (ST21B), and the coded bits Sa1, Sa3 and the coded bits Sb1, Sb3 are decoded (ST22B). Next, signal point reduction is performed based on the information of the obtained coded bits Sa1, Sa3, Sb1, and Sb3 (ST22A), and the coded bits Sa0 and Sa2 and the coded bits Sb0 and Sb2 are decoded (ST23A). Thereafter, the same processing is repeated.

  In the present embodiment, it is important to improve the reception quality that the interleave pattern of the interleaver is different (pattern X and pattern Y are different). The reason will be described in detail below.

  FIG. 59 shows a reception state when pattern X (interleave pattern for coded bits Sa0, Sa2, Sb0, Sb2) and pattern Y (interleave pattern for coded bits Sa1, Sa3, Sb1, Sb3) are the same pattern. An example is shown.

  As a result of decoding the coded bits Sa0, Sa2, Sb0, and Sb2 in the soft decision unit 5602 of FIG. 58 based on such an interleave pattern, symbols that are erroneously determined are as shown in FIG. Assume that they occur continuously. Incidentally, when a convolutional code or the like is used, it is common that errors occur continuously. Then, when the signal point reduction units 1301 and 1302 reduce the number of signal points, as shown in FIG. 59B, an error occurs continuously in signal point selection by signal point reduction. As a result, even when the decoding unit 5608 decodes the encoded bits Sa1 and Sa3 and the encoded bits Sb1 and Sb3, the reception quality (error rate characteristic) is not effectively improved. This is because error correction codes have a low ability to correct continuous errors.

  In FIG. 60, pattern X (interleave pattern for coded bits Sa0, Sa2, Sb0, Sb2) and pattern Y (interleave pattern for coded bits Sa1, Sa3, Sb1, Sb3) are shown in FIG. An example of the reception state when different is shown.

  Based on such an interleave pattern, the soft decision unit 5602 in FIG. 58 decodes the encoded bits Sa0, Sa2, Sb0, and Sb2, and as a result, erroneously determined symbols are as shown in FIG. Are assumed to occur continuously. Then, when the number of signal points is reduced in the signal point number reduction units 1301 and 1302, since the interleave pattern X and the interleave pattern Y are different from FIG. 59B, deinterleaving results in FIG. 60B. In addition, signal point selection errors due to signal point reduction occur discretely. That is, the signal point selection error due to signal point reduction does not occur continuously as shown in FIG. Accordingly, when the coded bits Sa1 and Sa3 and the coded bits Sb1 and Sb3 are decoded by the decoding unit 5608, the error rate characteristic is effectively improved. This is because error correction codes have a high ability to correct discrete errors.

  Thus, according to the present embodiment, the bit interleave pattern of the modulated signal transmitted from each antenna is made different, so that the influence of the burst error is reduced at the time of decoding on the receiving side, and the error rate characteristic is good. A multi-antenna transmission apparatus capable of obtaining received data can be realized. Further, according to the present embodiment, the number of encoding units can be reduced as compared with the eleventh embodiment, which leads to a reduction in the amount of calculation and the circuit scale.

  In the above-described embodiment, the example applied to the spread spectrum communication system has been described. However, the present invention is not limited to this. For example, the present invention can be applied to a single carrier system or OFDM system that is not a spread spectrum communication system. When applied to the OFDM method, the encoding method can be a method of encoding in the direction along the time axis as shown in FIG. 60, and the horizontal axis in FIG. It is also possible to do. In addition, encoding in both the time axis and the frequency axis is also possible. In addition, encoding in both the time axis and the frequency axis is also possible.

  At this time, as described in the seventh embodiment, the interleave pattern X is a pattern in which data is rearranged from a high-frequency subcarrier to a low subcarrier, and the interleave pattern Y is a low-frequency subcarrier to a high subcarrier. If the data is rearranged and arranged on the carrier, the error rate characteristics can be improved effectively, and the circuit configuration can be simplified.

  Furthermore, the interleaving method illustrated in this embodiment is merely an example, and the present invention is not limited to this. Ideally, random interleaving is appropriate.

  Moreover, although the case where the interleaving is performed by the interleaver has been described in the above-described embodiment, for example, when applied to the OFDM scheme, the interleaving may be performed by the mapping process to each subcarrier. In this way, the interleaver can be omitted.

(Embodiment 13)
In the sixth embodiment, the multi-antenna transmission apparatus has been proposed in which the interleave pattern of the modulation signal transmitted from each antenna is different between the modulation signals. However, in this embodiment, when bit interleaving is applied as the interleaving, a specific example is given. A typical apparatus will be described. That is, the present embodiment is basically the same as the eleventh and twelfth embodiments in that bit interleaving is performed, but the bit interleaving is applied to the basic configuration of the sixth embodiment as it is. An example will be described.

  FIG. 48 shows an example of signal point arrangement on the IQ plane of modulated signal A and modulated signal B transmitted by the multi-antenna transmitting apparatus of the present embodiment.

  The basic configuration of the multi-antenna transmission apparatus according to the present embodiment is the same as that of FIG. 23 described in the sixth embodiment, and the operation thereof is the same as that of the sixth embodiment.

  A bit interleaving method using interleavers 2301A and 2301B in the case of adopting the configuration shown in FIG. 23 will be described in detail with reference to FIG. Incidentally, the interleaver 2301A performs an interleave process using the interleave pattern X of FIG. 61A, and the interleaver 2301B performs an interleave process using the interleave pattern Y of FIG. 61B.

  FIG. 61 shows an example of the order of data before interleaving, after interleaving, and after separation. FIG. 61A shows an interleaving method of the interleave pattern X. Data 1, data 2,..., Data 200 are assigned to the order before interleaving. The order is rearranged every five data by interleaving (this processing is the same as the interleaving processing of the coded bits Sa0 and Sa2 in FIG. 50 described in the eleventh embodiment, and thus detailed description is omitted). .

  Then, the data arranged after the interleaving is alternately distributed to the coded bits Sa0 and Sa2 and the coded bits Sa1 and Sa3. Therefore, the data of the coded bits Sa0, Sa2 are in the order of data 1, data 11,..., Data 185, data 195, and the data of the coded bits Sa1, Sa3 are data 6, data 16,. The order is data 190 and data 200.

  FIG. 61 (B) shows an interleaving method for interleaving pattern Y. Data 1, data 2,..., Data 200 are assigned to the order before interleaving. The order is rearranged every eight data by interleaving (this processing is the same as the interleaving processing of the coded bits Sa1 and Sa3 in FIG. 50 described in the eleventh embodiment, and detailed description thereof is omitted).

  Then, the data arranged after interleaving is alternately distributed to the coded bits Sb0 and Sb2 and the coded bits Sb1 and Sb3. Therefore, the data of the coded bits Sb0, Sb2 are in the order of data 1, data 17,..., Data 176, data 192, and the data of the coded bits Sb0, Sb2 are data 9, data 25,. The order is data 184 and data 200.

  Each of the modulation units 202A and 202B in FIG. 23 performs modulation by assigning any of the 16 points in FIGS. 48A and 48B in accordance with the encoded bits that have been bit-interleaved as described above. .

  Next, the configuration of a multi-antenna receiving apparatus that receives a plurality of modulated signals A and B that have been subjected to bit interleaving in this way will be described. As one of the configurations of the multi-antenna receiving apparatus (configuration of the signal processing unit), the configuration of FIG. 24 described in Embodiment 6 can be considered, and the operation is performed by bit deinterleaving processing and deinterleaving processing by each deinterleaver and interleaver. Except for performing the interleaving process, it is the same as in the sixth embodiment.

  In this embodiment, a configuration example as shown in FIG. 62 will be described as a configuration different from FIG. In FIG. 62, in which parts corresponding to those in FIG. 24 are assigned the same reference numerals, the maximum likelihood determination unit 6001 of the signal processing unit 6000 includes channel estimation values h11, h21, h12, h22, baseband signals R1-2, R2- 2 is input, and maximum likelihood determination is performed to obtain information 6004 of encoded bits Sa0, Sa1, Sa2, and Sa3, which is sent to the deinterleaver 2401A and the encoded bits Sb0, Sb1, Sb2, and Sb3 Information 6005 is obtained and sent to the deinterleaver 2401B.

  Information 6004 of coded bits Sa0, Sa1, Sa2, Sa3 and information 6005 of coded bits Sb0, Sb1, Sb2, and Sb3 deinterleaved by deinterleavers 2401A and 2401B are respectively obtained by hard decision decoding sections 6002 and 6003. Hard decision decoding is performed and output. Other parts operate in the same manner as in the sixth embodiment.

  In the signal processing unit 6000, the bit interleave pattern X of the modulation signal A and the bit interleave pattern Y of the modulation signal B are different from each other, and thus the same as described in the eleventh and twelfth embodiments. For the reason, it is possible to obtain received data RA and RB with good error rate characteristics.

  Thus, according to the present embodiment, when the interleave pattern of the modulated signal transmitted from each antenna is made different for each modulated signal, the bit interleave pattern for each modulated signal is made different so that decoding on the receiving side can be performed. In this case, it is possible to realize a multi-antenna transmission apparatus and a multi-antenna reception apparatus that can reduce the influence of a burst error and obtain reception data with good error rate characteristics.

  Note that the configuration of the signal processing unit 6000 described in the present embodiment can also be applied to embodiments such as the sixth embodiment. That is, maximum likelihood determination can be used for provisional determination.

  In the present embodiment, the configuration of the multi-antenna transmission apparatus in which the bit interleave pattern of each modulation signal is different has been described using the multi-antenna transmission apparatus 2300 in FIG. 23 and the bit interleave pattern in FIG. The overall configuration of the antenna device and the bit interleave pattern are not limited to those shown in FIGS. In particular, various bit interleave patterns can be applied. In short, the bit interleave pattern of each modulation signal may be different.

  As a configuration example different from the configuration of the multi-antenna transmission apparatus 2300 in FIG. 23, for example, a configuration as shown in FIG. In FIG. 63, in which parts corresponding to those in FIG. 23 are assigned the same reference numerals, the most characteristic point of multi-antenna transmission apparatus 6100 is that “the interleave pattern (interleaver 4708) and code for encoded bits Sa0 and Sa2” The interleave pattern (interleaver 6101) for the coded bits Sa1, Sa3 is the same, and the interleave pattern (interleaver 4719) for the coded bits Sb0, Sb2 and the interleave pattern for the coded bits Sb1, Sb3 (Interleaver 6102) is the same. " Thereby, the error correction capability is improved and the reception quality is improved in the receiving apparatus. Although all bit interleaving patterns may be different, reception quality is deteriorated as described in the eleventh embodiment.

  Furthermore, the interleaving method illustrated in this embodiment is merely an example, and the present invention is not limited to this. Ideally, random interleaving is appropriate.

(Embodiment 14)
This embodiment proposes a multi-antenna transmission apparatus capable of setting the above-described interleave pattern with a simple configuration when LDPC (Low Density Parity Check) is applied. In other words, a method for effectively utilizing LDPC in forming the above-described interleave pattern modulation signal is proposed.

  In the above-described sixth, seventh, eleventh, twelfth and thirteenth embodiments, basically, the interleaver 2301A and the interleaver 2301B have different interleave patterns as shown in FIG. The interleave pattern was made different. In this embodiment, when applying LDPC, it is proposed to replace interleavers 2301A and 2301B with LDPC encoders having different generation matrices G_i (check matrix H_i). As a result, the same effect as when different interleave patterns are provided by providing different interleavers can be obtained.

  FIG. 64, in which parts corresponding to those in FIG. 2 are assigned the same reference numerals, shows the configuration of the multi-antenna transmission apparatus of this embodiment. Compared to multi-antenna transmission apparatus 110 in FIG. 2, multi-antenna transmission apparatus 6200 is provided with LDPC encoders 6201A and 6201B in place of encoding sections 201A and 201B.

  The LDPC encoder 6201A for the modulation signal A is represented by a generator matrix Ga and a check matrix Ha, receives the transmission data TA, and outputs encoded data S1A. The LDPC encoder 6201B for the modulation signal B is represented by a generation matrix Gb and a check matrix Hb, receives the transmission data TB, and outputs encoded data S1B.

  The feature of this embodiment is that the generation matrix Ga of the LDPC encoder 6201A for the modulation signal A is different from the generation matrix G of the LDPC encoder 6201B for the modulation signal B. The reason will be described below.

  LDPC is a linear code defined by a very sparse check matrix, and one of its features is the flexibility to easily configure codes with various code lengths and coding rates. Similarly, it is easy to configure a plurality of types of codes having the same code length and coding rate. In many other error correction codes, configurable parameters are limited depending on the type of code.

  In decoding a turbo code or a convolutional code, all bits use information of adjacent bits in order to update likelihood information. Accordingly, when handling a channel with memory such as a fading environment, an interleaver is used to artificially whiten noise having a correlation between adjacent bits. For this reason, the present invention adopts a transmitter configuration that applies different interleave patterns for each stream.

  However, in LDPC decoding, when a certain bit n is updated by a parity check related to that bit, that is, check m, only information from some bits related to check m is used. Since the parity check matrix H of the LDPC code is generally constructed at random, the probability that information bits used for updating are adjacent is extremely small. Thus, even if bit n is in deep fade, the probability that other bits related to bit n are also in deep fade is low, and those bits are more reliable information for bit n. I will provide a.

  In other words, this means that the LDPC parity check matrix H has an intrinsic interleaving function. In principle, in the design of the parity check matrix H, the interleave gain is maximized for all non-zero elements of the matrix H. It is possible to arrange in

  This result is also reported in J. Hou, P. Siegel, and L. Milstein, “Performance Analysis and Code Optimization for Low Density Parity-Check Codes on Rayleigh Fading Channels” IEEE JSAC, Vol. 19, No. 5, May, 2001. It is shown.

Next, LDPC encoding will be described. Since LDPC is a kind of linear code, it can be obtained by multiplying the generator matrices Ga, Gb by information vectors (m1, m2,..., Mk), (n1, n2,..., Nk). That is, the generation matrices Ga and Gb corresponding to the check matrices Ha and Hb designed in advance are obtained (the generation matrices Ga and Gb satisfy G _a H _a ^T = 0 and G _b H _b ^T = 0), and c Codewords c and d can be obtained with = mG _a or d = nG _b .

  Next, LDPC decoding will be described. The overall configuration of the multi-antenna receiving apparatus in the present embodiment may be as shown in FIG. 4, for example. 4 may be configured as shown in FIG. 65, for example.

  In FIG. 65, in which the same reference numerals are assigned to the parts corresponding to those in FIG. 13, the signal processing unit 6300 replaces the soft decision units 503, 506, 512, and 518 of the signal processing unit 1300 in FIG. 13 with an LDPC decoding method. A certain probability region sum-product decoding unit 6301, 6302, 6303, 6304 is replaced.

  When LDPC is used, instead of the combination of a deinterleaver and a Viterbi decoding circuit (when convolutional code is applied), for example, a probability domain sum-product decoding method (probability), which is a typical decoding method when using LDPC, is used. domain sum-product decoding algorithm) or log domain sum-product decoding algorithm (log domain sum-product decoding algorithm) suitable for implementation in hardware and software can be used.

  Sum-product decoding units 6301 and 6303 of modulated signal A perform decoding corresponding to check matrix Ha used on the transmission side. Also, sum-product decoding units 6302 and 6304 of modulated signal B perform decoding corresponding to check matrix Hb used on the transmission side. The received digital signals of modulated signals A and B that have been subjected to error correction processing are re-encoded and modulated again using LDPCs of the same generation matrices Ga and Gb as at the time of transmission, and signal point reduction is performed.

  The detailed operation is the same as the operation described except for the LDPC, and only the operation related to the LDPC is different, and the other operations are the same as those of the above-described embodiment.

  Thus, according to the present embodiment, an LDPC having a different generation matrix G_i (check matrix H_i) is applied to each stream, and processing equivalent to interleaving processing is performed using the generation matrix. As shown in FIG. 23, it is possible to obtain the same effect as in the case of adopting a configuration using an interleaver having a different interleave pattern for each stream and an error correction encoder different from LDPC such as a turbo code or a convolutional code. . Furthermore, since the LDPC code includes the interleaver function itself, the circuit scale can be reduced.

  In this embodiment, the modulation signals A and B having different interleave patterns are formed only by the LDPC encoder. However, in addition to the LDPC encoder, an interleaver is used as in the sixth and seventh embodiments. It may be provided to form modulated signals having different interleave patterns, and even in this case, reception quality can be improved.

  Furthermore, as shown in FIG. 66, the same LDPC encoder may be used as the LDPC encoder 6201A and the LDPC encoder 6201B, and an interleaver 6601 may be provided in only one of them. Even in this case, the interleave pattern of the modulated signal A and the modulated signal B can be made different, so that the reception quality can be improved.

(Embodiment 15)
In this embodiment, a retransmission method suitable for the above-described receiving apparatus and method (that is, an apparatus and method for reducing candidate signal points and decoding) is proposed.

  First, FIG. 67 shows a frame configuration example of transmission signal A and transmission signal B transmitted on the transmission side (for example, base station). Channel estimation symbols 6801A and 6802B of modulated signals A and B are transmitted at the same time, and the receiver estimates channel fluctuations due to fading and the like using these channel estimation symbols 6801A and 6802B. The data symbols 6802A and 6802B of the modulation signals A and B are formed based on the transmission digital signals TA and TB, and data is transmitted by the data symbols 6802A and 6802B. In addition, CRC (Cyclic Redundancy Check) symbols 6803A and 6803B are added to the modulated signals A and B, respectively, and the receiver checks the CRC symbols 6803A and 6803B to detect errors in the data transmitted by the modulated signals A and B, respectively. Check if there was. The control information symbol 6804 is provided for frequency offset detection and AGC (Automatic Gain Control) at the receiver, and for identifying whether the signal is a retransmission signal.

  FIG. 68 shows a configuration example of a transmission system of a receiver (for example, a communication terminal). Error determination section 6902A receives as input digital signal 6901A of modulated signal A obtained by demodulating the signal transmitted by the base station, and uses CRC symbol 6803A of FIG. 67 to determine whether or not there is an error in modulated signal A. Error presence / absence information 6903A indicating the above is output. Similarly, error determination section 6902B receives as input digital signal 6901B of modulated signal B obtained by demodulating the signal transmitted from the base station, and uses CRC symbol 6803B of FIG. Error presence / absence information 6903B indicating whether or not there has been is output.

  Based on error presence / absence information 6903A and 6903B of modulated signals A and B, retransmission request section 6904 determines whether or not to request retransmission, and outputs retransmission request information 6905 (for example, ACK / NACK information). The retransmission request information 6905 is information indicating whether the modulated signal A is retransmitted or the modulated signal B is retransmitted.

  The data generation unit 6907 receives the retransmission request information 6905 and the transmission data 6906, generates a transmission digital signal 6908, and outputs it. The transmission unit 6909 performs a predetermined modulation process on the transmission digital signal 6908 to form a modulation signal 6910. Modulated signal 6910 is output as a radio wave from antenna 6911.

  FIG. 69 shows a frame configuration example of modulated signal 6910 transmitted by the communication terminal. Modulated signal 6910 includes channel estimation symbol 7001, data symbol 7002, and retransmission request information symbol 7003 for channel estimation on the receiving side.

  FIG. 70 shows the configuration of the multi-antenna transmission apparatus of this embodiment. Multi-antenna transmission apparatus 7000 is provided in a base station, for example.

  In FIG. 70, in which parts corresponding to those in FIG. 2 are assigned the same reference numerals, a multi-antenna transmission apparatus 7000 receives a signal transmitted from the communication terminal in FIG. Received signal 7102 (corresponding to the modulated signal in FIG. 69) is input to receiving section 7103. The receiving unit 7103 obtains a received digital signal 7104 by demodulating the received signal 7102 and outputs it. The retransmission request detection unit 7105 extracts retransmission request information 7106 from the received digital signal 7104 and outputs this. As described above, retransmission request information 7106 includes information on whether modulated signal A is retransmitted or modulated signal B is retransmitted.

  Data transmission units 7107A and 7107B and data selection units 7109A and 7109B are provided in the transmission system of multi-antenna transmission apparatus 7000, and retransmission request information 7106 is input to data selection units 7109A and 7109B. Data selection section 7109A selects and outputs retransmission data 7108A stored in data storage section 7107A when retransmission request information 7106 indicates that retransmission of modulated signal A is requested. Similarly, data selection section 7109B selects and outputs retransmission data 7108B stored in data storage section 7107B when retransmission request information 7106 indicates that retransmission of modulated signal B is requested. .

  More specifically, when the retransmission request information 7106 requests retransmission, and the data selection unit 7109A requests retransmission of the modulated signal A, the data selection unit 7109A uses the accumulated transmission digital signal 7108A of the modulated signal A. Select output. On the other hand, when the retransmission request information 7106 requests retransmission and requests retransmission of the modulated signal B, the data selection unit 7109A outputs nothing. When retransmission request information 7106 does not request retransmission, data selection unit 7109A selects and outputs transmission digital signal TA.

  Similarly, data selection section 7109B selects and outputs accumulated transmission digital signal 7108B of modulated signal B when retransmission request information 7106 requests retransmission and retransmission of modulated signal B is requested. . On the other hand, when the retransmission request information 7106 requests retransmission and requests retransmission of the modulated signal A, the data selection unit 7109B outputs nothing. When retransmission request information 7106 does not request retransmission, data selection unit 7109B selects and outputs transmission digital signal TB.

  Thus, in multi-antenna transmission apparatus 7000, when one modulation signal is retransmitted, the other modulation signal is not transmitted.

  Frame configuration signal generation section 210 determines the frame configuration based on retransmission request information 7106, and outputs frame configuration signal S10. An example of a frame configuration determination method will be described later with reference to FIG.

  FIG. 71 shows a flow of transmission signals of the base station and the communication terminal in the present embodiment. 71, the signal transmitted by the base station is actually a frame unit signal composed of control information, CRC symbols and the like in addition to data symbols.

First, the base station
Data 1A and data 1B are transmitted as in <1>.

Then, the terminal receives data 1A and data 1B, confirms that no error has occurred,
As in <2>, retransmission is not requested.

Next, the base station
Data 2A and data 2B are transmitted as in <3>.

Then, the terminal receives data 2A and data 2B and confirms that an error has occurred. At this time, the terminal compares the received electric field strength of modulated signal A with the received electric field strength of modulated signal B, and requests retransmission of a modulated signal having a lower received electric field strength. In the case of the figure, it has been detected that the received electric field strength of the modulation signal A is low.
As in <4>, a retransmission request for the modulated signal A is made. As described above, by retransmitting the modulated signal having the lower reception electric field strength, the effect of improving the error rate characteristic by the retransmission can be enhanced. This is because the modulation signal having a low reception electric field strength has a poor reception quality, and the reception quality of the modulation signal having a low reception electric field strength can be ensured by retransmission. In addition, by retransmitting the modulation signal with the lower received electric field strength, the accuracy in reducing candidate signal points for the other modulation signal using the modulation signal is improved, so that the other modulation signal is used. Signal error rate characteristics can also be improved.

  Here, the modulated signal having the lower received electric field strength is retransmitted. However, as another method, for example, when an error occurs in data 2B among data 2A and data 2B, retransmission of data 2B is performed. A simple method of requesting may be used.

When the base station receives the retransmission request signal for data 2A,
As in <5>, the data 2A is retransmitted.

Then, since the terminal did not cause an error in data 2A and data 2B,
As in <6>, no retransmission request is made.

Next, the base station
Data 3A and data 3B are transmitted as in <7>.

Then, the terminal receives data 3A and data 3B and confirms that an error has occurred. At this time, the terminal compares the received electric field strength of the modulated signal A with the received electric field strength of the modulated signal B, and detects that the received electric field strength of the modulated signal B is low.
As in <8>, a retransmission request for the modulated signal B is made.

Then the base station
As in <9>, the data 3B is retransmitted.

Still, if the terminal has an error in data 3A and data 3B,
Request retransmission again as in <10>. At this time, the terminal requests retransmission of a modulation signal different from the modulation signal requested for the first time. That is, the retransmission of the modulated signal A is requested. By doing in this way, the improvement effect of the error rate characteristic by resending can be heightened. In other words, the data 3B retransmitted for the first time has an excellent reception quality due to retransmission in <9>, whereas the reception quality of the data 3A has no effect of improving the reception quality due to retransmission at this point. Therefore, it is considered to be lower than the data 3B. Therefore, when the second retransmission is performed, it is preferable to retransmit the data of the modulation signal different from the first, such as data 3A.

When the base station receives the retransmission request signal for data 3B,
As in <11>, the data 3A is retransmitted.

  As described above, in this embodiment, the data of both modulation signal A and modulation signal B are not retransmitted, but only the data of one modulation signal is retransmitted. The reason for this will be described later.

  FIG. 72 shows the configuration of the multi-antenna reception apparatus of this embodiment. Multi-antenna reception apparatus 7200 is provided in a communication terminal, for example.

  In FIG. 72, in which parts corresponding to those in FIG. 4 are assigned the same reference numerals, multi-antenna reception apparatus 7200 receives and demodulates signals transmitted from multi-antenna transmission apparatus 7000 (FIG. 70).

  Control information detection section 7301 receives baseband signals R1-2 and R2-2 after despreading, and is indicated by control information symbol 6804 in the frame of FIG. 67 transmitted by multi-antenna transmission apparatus 7000 (base station). Detect control information. That is, based on the control information symbol 6804, the received signal is not the retransmitted signal but the modulated signal A and the modulated signal B are transmitted simultaneously, or if the received signal is the retransmitted signal, the retransmitted signal is the modulated signal A. Or control information indicating the modulation signal B is detected. The control information detection unit 7301 outputs the detected control information as transmission method information 7302.

  The signal processing unit 404 receives the channel fluctuation estimation values h11, h12, h21, h22, the despread baseband signals R1-2, R2-2, and the transmission method information 7302, and the transmission method information 7302 is the modulation signal A. , B indicates that the transmission method is a simultaneous transmission method, a demodulation operation is performed to obtain a reception digital signal RA of the modulation signal A and a reception digital signal RB of the modulation signal B. The detailed configuration of the signal processing unit 404 is, for example, as shown in FIGS. 5, 11, 12, 13, 13, 17, and the like, and the operation is as described above. The signal processing unit 404 operates, for example, at <1>, <3>, and <7> in FIG.

  Channel information / received signal storage section 7303 receives channel estimation values h11, h12, h21, h22 and despread baseband signals R2-1, R2-2, and stores these information. The channel information / received signal storage unit 7303 receives the transmission method information 7302 and indicates that the transmission method is transmitted when the transmission method information 7302 is retransmitted. The later baseband signal is output.

  Retransmission information detection section 7304 receives channel fluctuation estimated values h11, h12, h21, h22, despread baseband signals R1-2, R2-2, and transmission method information 7302, and transmission method information 7302 performs retransmission. In the case where it is shown that the retransmitted modulated signal is the modulated signal A, the modulated signal A is demodulated and the received digital signal RA of the modulated signal A is output. Also, retransmission information detection section 7304 demodulates modulated signal B, and receives modulated signal B when transmission method information 7302 indicates retransmission and the retransmitted modulated signal is modulated signal B. A digital signal RB is output. This operation is performed, for example, at <5>, <9>, and <11> in FIG.

  The signal processing unit 7305 includes accumulated channel estimation values h11, h12, h21, and h22 (not shown), accumulated debanded baseband signals R1-2 and R2-2 (not shown), and a transmission method. Information 7302 and received digital signal RA or RB of the retransmitted modulated signal (output of retransmission information detector 7304) are input.

  When transmission method information 7302 indicates retransmission and the retransmitted modulated signal is modulated signal A (corresponding to the situation <5> in FIG. 71), retransmission information detection section 7304 receives RA. (In the situation <5> in FIG. 71, data 2A is output), and the signal processing unit 7305 stores the accumulated channel estimation values h11, h12, h21, h22, Signal point reduction was used using despread baseband signals R1-2 and R2-2 and retransmitted digital signal RA of modulated signal A (data 2A in the situation <5> in FIG. 71). Demodulating operation is performed, and a digital signal RB of the modulated signal B is output (corresponding to data 2B in FIG. 71).

  On the other hand, when the transmission method information 7302 indicates retransmission and the retransmitted modulated signal is the modulated signal B (corresponding to the situation <9> in FIG. 71), the retransmission information detection unit 7304 RB is output (in the situation <9> in FIG. 71, data 3B is output), and the signal processing unit 7305 stores the accumulated channel estimation values h11, h12, h21, h22, Signal reduction using the despread baseband signals R1-2, R2-2 and the received digital signal RB of the retransmitted modulated signal (data 3B in the situation <9> in FIG. 71) Demodulation is performed, and a digital signal RA of the modulation signal A is output (corresponding to the data 3A in FIG. 71).

  FIG. 73, in which the same reference numerals are assigned to corresponding parts as in FIG. 5, shows a configuration example of the signal processing unit 7305. The signal processing unit 7305 inputs the transmission method information 7302 to the signal point reduction units 508, 510, 514, and 516. Also, received digital signal RA or RB output from retransmission information detection section 7304 is input to signal point reduction sections 508, 510, 514, 516, soft decision sections 512, 518 and data selection section 7401.

  Here, when transmission method information 7302 indicates retransmission and the retransmitted modulation signal is modulation signal A (corresponding to the situation <5> in FIG. 71), signal point reduction sections 508, 510, Received digital signal RA of modulated signal A is input to 514 and 516. At this time, the signal point reduction units 514 and 516 use the received digital signal RA of the modulation signal A that has been determined, and leave the signal point of only the modulation signal B as a candidate, as described in the above-described embodiment, The reduced signal point information 515 and 517 are output. Soft decision section 518 soft decision decodes modulated signal B and outputs received digital signal RB of modulated signal B. Data selection unit 7401 selects received digital signal RB of modulated signal B and outputs it as received digital signal 7402. At this time, the signal point reduction units 508 and 510 and the soft decision unit 512 do not operate.

  On the other hand, when transmission method information 7302 indicates retransmission and the retransmitted modulated signal is modulated signal B (corresponding to the situation <9> in FIG. 71), signal point reduction sections 508, 510, Received digital signal RB of modulated signal B is input to 514 and 516. At this time, the signal point reduction units 508 and 510 use the received digital signal RB of the determined modulation signal B, and leave the signal point of only the modulation signal A as a candidate, as described in the above embodiment. Reduced signal point information 509 and 511 are output. Soft decision section 512 performs soft decision decoding on modulated signal A and outputs received digital signal RA of modulated signal A. The data selection unit 7401 selects the received digital signal RA of the modulated signal A and outputs it as a received digital signal 7402. At this time, the signal point reduction units 514 and 516 and the soft decision unit 518 do not operate.

Next, data stored in channel information / received signal storage section 7303 of multi-antenna receiving apparatus 7200 will be described in detail with reference to FIG. For example, consider a case where data 2A and data 2B are transmitted as shown in <3> of FIG. Data 2A and data 2B are composed of 100 symbols, and each symbol is transmitted at time T2,0, T2,1,..., T2,99 as shown in FIG. At this time, assuming that time t = T2, 0, T2, 1,..., T2, 99, the channel fluctuations are h11 (t), h12 (t), h21 (t), h22 (t), and each antenna. Reception baseband signals received by AN3 and AN4 can be represented by R1-2 (t) and R2-2 (t). Further, the reception baseband signals R1-2 (t) and R2-2 (t) have channel fluctuations h11 (t), h12 (t), h21 (t), h22 (t), and the transmission signal Txa () of the modulation signal A. t) and the transmission signal Txb (t) of the modulation signal B can be expressed by the following equation.

  The channel information / received signal storage unit 7303 stores the channel fluctuations h11 (t), h12 (t), h21 (t), h22 (t) and the received baseband signals R1-2 (t), R2-2 (t ) Is accumulated.

  By the way, in the present embodiment, at the time of retransmission, only one of the modulation signals is retransmitted, so that the error rate characteristics of the received digital signals RA and RB can be improved. This will be described.

  When the data 2A is retransmitted as shown in <5> in FIG. 71, the modulated signal of the data 2A is received by the two receiving antennas AN3 and AN4, and is demodulated by the retransmission information detection unit 7304 by combining the maximum ratio. Therefore, retransmission information detector 7304 can obtain data 2A having very good reception quality (good error rate characteristics) as reception digital signal RA. Incidentally, as shown in <3> of FIG. 71, when the data 2A is demodulated from the signal in which the data 2A and the data 2B are mixed using the equation (4), the modulation signal of the data 2B becomes interference. It is difficult to obtain data 2A having better quality (good error rate characteristics) than when receiving alone. That is, in this embodiment, only one of the modulation signals is retransmitted, so that a good quality Txa (t) is estimated at times t = T2, 0, T2, 1,. A value (estimated value of the modulation signal A) can be obtained.

  Since all estimated values other than Txb (t) in the equation (4) are obtained at times t = T2, 0, T2, 1,..., T2, 99, the signal processing unit 7305 Then, channel fluctuations h11 (t), h12 (t), h21 (t), h22 (t) stored in the channel information / received signal storage unit 7303, received baseband signals R1-2 (t), R2- 2 (t) and the estimated value of Txa (t) with the highest ratio synthesized and demodulated (estimated value of the modulated signal A), Txb (corresponding to the maximum ratio synthesized with two receiving antennas ( t) can be demodulated. As a result, similarly to the data 2A, data 2B with excellent reception quality can be obtained. The above is a series of operations of the channel information / received signal storage unit 7303, the retransmission information detection unit 7304, and the signal processing unit 7305.

  Thus, according to the present embodiment, when retransmission is performed, the data of both the modulation signal A and the modulation signal B is not retransmitted, but only the data of one modulation signal is retransmitted. It is possible to obtain a merit that the possibility that the generated data can be reproduced becomes high.

  In particular, the reception quality is remarkably improved in a propagation environment in which direct waves exist predominantly as compared with the case where the data of both the modulation signal A and the modulation signal B are retransmitted. For example, when there is a direct wave, good reception quality may not be obtained even if the reception electric field strength is obtained as described in the tenth embodiment. In such a case, even if the data of both the modulation signal A and the modulation signal B are retransmitted again, a great improvement effect on the reception quality cannot be obtained.

  However, if a configuration is adopted in which only one modulation signal data is retransmitted as in this embodiment, as described above, the retransmitted modulation signal is demodulated with maximum ratio combining, that is, with a strong received electric field strength. As a result, excellent quality can be obtained. In addition, for the signal that has not been retransmitted, the maximum ratio combining is performed by canceling the retransmitted modulation signal from the mixed signal of both modulation signals (ie, the accumulated signal) and then demodulating it. Can be demodulated. As a result, even in an environment where the direct wave is dominant, both modulation signals can be demodulated in a maximum ratio combining situation, so that a significant improvement effect on the reception quality can be obtained.

  In this embodiment, the case where two modulated signals A and B are transmitted other than at the time of retransmission and only one of the modulated signals is transmitted at the time of retransmission has been described. If the number of signals is made smaller than the number of modulated signals to be transmitted other than at the time of retransmission, the same effect as in the above-described embodiment can be obtained.

  In this embodiment, the example used in the spread spectrum communication system has been described. However, the present invention is not limited to this example. For example, the present invention can be applied to a single carrier system or OFDM system that is not a spread spectrum communication system. When applied to the OFDM system, information can be transmitted not only in the time axis direction but also in the frequency axis direction. For example, when considering data 1A in FIG. 71, data 1A may be arranged in the time axis direction and the frequency axis direction. . The same applies to the sixteenth embodiment described below.

  In this embodiment, as shown in FIG. 71, the case where either one of modulated signal A or modulated signal B is transmitted at the time of retransmission has been described. However, the number of modulated signals to be transmitted is changed according to the retransmission. It may be.

  FIG. 75 shows an example of the signal flow between the base station and the communication terminal in this case. The base station transmits data 1A in modulated signal A and data 1B in modulated signal B as shown in <1> of FIG. Then, the terminal does not make a retransmission request unless an error occurs as in <2>.

  Next, the base station transmits data 2A in modulated signal A and data 2B in modulated signal B as shown in <3>. Then, when an error occurs as in <4>, the terminal makes a retransmission request.

  The base station retransmits data according to the retransmission request from the terminal. First, as in <5>, the base station retransmits data 2A in modulated signal A and data 2B in modulated signal B. When an error occurs again as in <6>, the terminal makes a retransmission request again.

  Then, the base station retransmits data according to the retransmission request from the terminal. At this time, retransmission is performed by a method different from the retransmission method of <5>. Here, only data 2A is retransmitted as shown in <7>. The terminal does not make a retransmission request unless an error occurs as in <8>. Then, the base station transmits the next data (data 3A and data 3B) as shown in <9>.

  In this way, the retransmission method is changed between the first retransmission and the second retransmission. In the example of FIG. 75, a retransmission method for retransmitting the same data from a plurality of antennas using a plurality of modulated signals at the first retransmission, and a retransmission method for retransmitting a modulated signal of only one channel at the second retransmission. It is carried out. If the number of modulated signals to be transmitted is changed in accordance with retransmission in this way, the number of retransmissions can be further reduced, leading to improved transmission efficiency.

  This is because a propagation environment suitable for improving reception quality differs between a method of retransmitting a modulated signal of only one channel and a method of retransmitting the same data from a plurality of antennas by a plurality of modulated signals. The method of retransmitting the modulation signal of only one channel is suitable for retransmission in an environment where the direct wave is dominant. On the other hand, the method of retransmitting the same data from a plurality of antennas using a plurality of modulated signals is suitable for an environment where scattered waves are dominant. Therefore, if the propagation environment is not estimated, retransmitting using a different retransmission method increases the probability that no data error will occur due to either retransmission, which can reduce the number of retransmissions. Data transmission efficiency is improved.

  Also, here, a case has been described where the retransmission method is different between the first retransmission and the second retransmission without estimating the propagation environment, but the propagation environment is estimated and the propagation environment estimation information is estimated between the base station and the terminal. Is shared, it is also effective to set a fixed one of the retransmission schemes based on this propagation environment estimation information.

  In this way, when using MIMO transmission, considering that a suitable retransmission method varies depending on the propagation environment, a plurality of retransmission methods (here, a method of retransmitting a modulated signal of only one channel, and a plurality of antennas) By using the method of retransmitting a plurality of modulated signals), it is possible to reduce the number of retransmissions and improve the data transmission efficiency.

  In this embodiment, the case where the number of transmission antennas is two has been described as an example. However, the present invention can be similarly implemented when the number of transmission antennas is three or more and the number of modulation signals is three or more.

(Embodiment 16)
In this embodiment, a retransmission method using a frame configuration different from the frame configuration of FIG. 71 described in Embodiment 15 is proposed as a retransmission method suitable for an apparatus and method for decoding and decoding candidate signal points. To do.

  The configuration of the base station, the configuration of the communication terminal, the configuration of one frame, the frame configuration of the modulation signal transmitted from the terminal to the base station, and the like are the same as those in the fifteenth embodiment. Here, the flow of signals of base stations and terminals different from those in FIG. 71 will be described with reference to FIG.

First, the base station
As in <1>, data 1A, data 1B, data 2A, data 2B, data 3A, data 3B, data 4A, and data 4B are transmitted.

Then, the terminal receives data 1A, data 1B, data 2A, data 2B, data 3A, data 3B, data 4A, and data 4B. When the terminal detects that an error has occurred in data 2A, data 2B, data 4A, and data 4B,
As in <2>, retransmission of these symbols is requested.

Next, the base station
As in <3>, data 2A and data 4A are retransmitted.

  Then, the terminal uses the accumulated channel estimation value, the baseband signal, and the retransmitted data 2A obtained in the case of <1>, and uses the accumulated baseband signal to convert the modulated signal of data 2A. Cancel and demodulate the data 2B from the signal after cancellation. Similarly, using the accumulated channel estimation value, the baseband signal, and the retransmitted data 4A, the modulated signal of the data 4A is canceled from the accumulated baseband signal, and the data 4B is obtained from the canceled signal. Demodulate.

If the terminal still detects that an error has occurred in data 2B,
As in <4>, retransmission of data 2B is requested.

Then the base station
Data 2B is transmitted as in <5>.

When the terminal receives the data 2B and confirms that no error has occurred,
As shown in <6>, the base station is notified that there is no request for retransmission.

And the base station is released from the retransmission operation,
As shown in <7>, new data, data 5A, data 5B, data 6A, data 6B, data 7A, data 7B, data 8A, and data 8B are transmitted.

  Such an operation is repeated.

  When retransmission is requested for each of a plurality of frames as in the present embodiment, the number of times the terminal transmits a retransmission request is reduced compared to the case where retransmission is requested for each frame as in the fifteenth embodiment. The transmission efficiency is improved.

(Embodiment 17)
In this embodiment, when the retransmission method as in the fifteenth embodiment and the sixteenth embodiment is adopted, a method for further improving the reception quality by retransmission is proposed by further devising the method of sending the retransmission data. Specifically, a space-time block code and cycled delay diversity are applied to the fifteenth and sixteenth embodiments.

  First, the process that led to the present embodiment will be described. When the retransmission is not performed, the multi-antenna transmission apparatus of the base station transmits the modulation signal A and the modulation signal B which are different modulation signals from the two antennas. Therefore, for data to be retransmitted, it is more stable as a system to use two antennas more effectively than to use only one antenna. In the present embodiment, paying attention to this point, retransmission data is transmitted by a transmission method capable of obtaining diversity gains such as space-time codes and cyclic delay diversity shown in FIG. As a result, retransmission data with good quality can be obtained on the receiving side, so that the error rate characteristics when the modulated signals A and B are demodulated can be further improved.

  Since the configuration of the space-time code has already been described, the cycled delay diversity will be described here with reference to FIGS. 92 and 77. FIG.

  FIG. 77 shows an example of a frame configuration when cycled delay diversity is performed using 12 symbols. The signal transmitted by the antenna AN1 in FIG. 92 is the transmission signal A in FIG. 77, and the signal transmitted by the antenna AN2 in FIG. 92 is the transmission signal B in FIG. For the transmission signal A, S1, S2,... S11, S12 are transmitted at times i + 1, i + 2,. The transmission signal B has a frame configuration shifted from the transmission signal A by a certain time. Here, S7, S8,..., S5, S6 are transmitted at times i + 1, i + 2,. With such a frame configuration, the receiving apparatus can obtain a diversity gain by equalizing the received signal, so that the reception quality of the signals S1 to S12 is improved and the error rate characteristic of the data is improved.

  FIG. 78 shows a configuration example of a multi-antenna transmission apparatus for realizing this. In FIG. 78, in which parts corresponding to those in FIG. 70 are assigned the same reference numerals, multi-antenna transmission apparatus 7700 receives encoded data S1A as well as modulator 202A and modulator 202B as well as encoded data S1B. Is the same as that of the multi-antenna transmission apparatus 7000 in FIG. 70 except that is input to the modulation unit 202A in addition to the modulation unit 202B.

  The retransmission operation according to the present embodiment will be described using FIG. 71 and FIG.

  Modulators 202A and 202B operate in the same manner as in the fifteenth embodiment, for example, when transmitting data that is not retransmission data such as data 1A and data 1B in FIG. On the other hand, for example, when data 2A is retransmitted as shown in <5> in FIG. 71, the encoded data S1A (that is, data 2A) is converted into a space-time code or cyclic delay diversity by the modulation unit 202A and the modulation unit 202B. Modulate according to the rules. Similarly, when data 3B is retransmitted as shown in <9> of FIG. 71, the encoded data S1B (that is, data 3B) is changed by the modulation unit 202A and the modulation unit 202B in accordance with the space-time code and cyclic delay diversity rules. Modulate.

  Incidentally, as a configuration of such a multi-antenna receiving apparatus for transmission, for example, a configuration as shown in FIG. 72 may be used. However, retransmission information detection section 7304 demodulates according to the rules of space-time code and cycled delay diversity. Other operations are the same as those described in the fifteenth and sixteenth embodiments.

  Thus, according to the present embodiment, in performing retransmission, in addition to retransmitting only data of one modulation signal rather than retransmitting data of both modulation signal A and modulation signal B, retransmission is performed using a space-time code. By using a transmission method that can obtain diversity gain such as Cyclic Delay Diversity, it is possible to further increase the possibility that data in which a frame error has occurred can be reproduced.

  In this embodiment, the example used in the spread spectrum communication system has been described. However, the present invention is not limited to this example. For example, the present invention can be applied to a single carrier system or OFDM system that is not a spread spectrum communication system. When applied to the OFDM system, the space-time code and cycled delay diversity can be realized by developing in the frequency axis direction in addition to the time axis direction.

(Embodiment 18)
In this embodiment, for example, by devising the MLD (Maximum Likelihood Detection) method performed in the soft decision units 1101 and 1705 of the signal processing unit shown in FIGS. A multi-antenna receiving apparatus and method capable of obtaining received digital data with improved rate characteristics are proposed.

  In detection by MLD, transmission is performed such that the square Euclidean distance between all candidate signal points created using the estimated channel fluctuations h11, h12, h21, and h22 and the received signals R1-2 and R2-2 is minimized. Judge as a signal.

  Detection by MLD can obtain the best reception quality (error rate characteristics) among detection methods such as ICD (Inverse Channel Detection) and MMSE (Minimum Mean Square Error) using inverse matrix operation. Since the distance is not uniform, soft decision decoding similar to the case where ICD is used cannot be performed.

Therefore, the BER is obtained by performing pseudo soft decision decoding by weighting the difference between the least square Euclidean distance U _min ² and the next smallest square Euclidean distance U _min2 ² with respect to the Hamming distance after the hard decision. The characteristics can be improved (in this embodiment, this detection / decoding method is called MLD-H (MLD-Hard Decision Decoding)).

The four Hamming distances when QPSK modulation is used for both channels A and B are defined as _{dH [0,0]} , dH _[0,1] , dH _[1,0] , and _{dH [1,1]} , respectively. Then, the branch metrics met Tx _{a [i, j]} and met Tx _{b [i, j]} of the channels A and B of the MLD-H decoding method are defined as follows.

  The MLD-H decoding method in the MIMO system is performed based on the equations (5) and (6), and the path that minimizes the sum of branch metrics is selected. Received digital data is obtained based on the selected path.

  Since the MLD-H decoding method performs hard decision decoding using the Hamming distance, there is a disadvantage that the coding gain becomes smaller than in the case of soft decision decoding using the Euclidean distance. In general, it is known that soft decision decoding has a larger coding gain than hard decision decoding.

  Considering this, in the present embodiment, as a soft decision decoding method in the case of using detection by MLD, candidate signal points are classified into two sets for each transmission bit, and the points of each set and received signal points An MLD-S (MLD-Soft Decision Decoding) decoding method that performs soft decision decoding using the least square Euclidean distance is proposed.

This will be specifically described. FIG. 79 shows a case where a QPSK-modulated transmission signal is decoded using the MLD-S decoding method. The least square Euclidean distance corresponding to a ₀ = 0 and a ₀ = 1 of the two bits a ₀ and a ₁ transmitted on channel A is obtained.

As shown in FIG. 79, 4 ² = 16 candidate signal points can be classified into a set of 8 points each corresponding to a ₀ = 0 and a ₀ = 1. In each set, the least square Euclidean distances U _{min (a0 = 0)} ² and U _{min (a0 = 1)} ² between the eight candidate signal points and the received signal points are calculated. Such classification and calculation are similarly performed on another bit a ₁ transmitted on channel A and 2 bits b ₀ and b ₁ transmitted on channel B, and softening is performed using these least square Euclidean distances. Perform decision decoding.

The branch metrics met Tx _{a [i, j]} and met Tx _{b [i, j]} for channels A and B are defined as follows:

Here, d _{S [i, j]} in the equation (7) is defined as the following equation.

Further, d _{S [i, j]} in the equation (8) is defined as the following equation.

  In the present embodiment, MLD-S decoding in the MIMO system is performed based on Equations (7) to (10), and a path that minimizes the sum of branch metrics is selected. Received digital data is obtained based on the selected path.

  Thus, according to the present embodiment, when using detection by MLD at the time of provisional determination, candidate signal points are classified into two (plurality) sets for each transmission bit, and the minimum 2 of the points of each set and the received signal points Since soft decision decoding is performed using the squared Euclidean distance, it is possible to perform MLD while suppressing a decrease in coding gain, so that it is possible to improve error rate characteristics at the time of provisional determination. become. As a result, received digital data with better error rate characteristics can be obtained.

  In this embodiment, the case where MLD-S decoding is performed using only the least square Euclidean distance in the set classified into two has been described, but there are a plurality of cases such as using the second smallest square Euclidean distance. MLD-S decoding may be performed using the squared Euclidean distance.

  Further, although this embodiment has been described using QPSK modulation, the modulation method can be similarly implemented with other modulation methods such as BPSK, 16QAM, and 64QAM.

  Further, the decoding method proposed in the present embodiment is not limited to an apparatus that performs iterative decoding, and the same effects as those of the above-described embodiment can be obtained even when the decoding method is performed alone.

(Embodiment 19)
In the present embodiment, when a specific symbol (STBC symbol or special symbol as shown in FIGS. 41 and 42) described in the ninth embodiment is inserted at regular timing, a specific symbol is determined in response to a retransmission request. It is proposed to change the interval of inserting.

  The schematic configuration of the terminal (retransmission requesting apparatus) of the present embodiment is as shown in FIG. 68, and the frame configuration of the modulation signal transmitted by the terminal is as shown in FIG. 69. The base station (retransmits data) The schematic configuration of the apparatus is as shown in FIG. 68, 69, and 70 have already been described in the fifteenth embodiment, and therefore, a duplicate description is omitted, and only the configuration peculiar to the present embodiment will be described here.

  In the present embodiment, encoders 201A and 201B and modulators 202A and 202B in FIG. 70 receive frame configuration signal S10, change the frame configuration based on the frame switching rule in FIG. Do.

  The frame configuration in FIG. 80 will be described in detail. Control information symbol 8901 is a symbol for transmitting frame configuration information and the like. The channel estimation symbol 8902 is a symbol for estimating each channel fluctuation due to fading or the like on the receiving side. Data symbol 8903 is formed based on each transmission digital signal TA, TB, and is a symbol for transmitting data. The data symbol of modulated signal A and the data symbol of modulated signal B at the same time are transmitted from different antennas.

  CRC symbol 8904 is a symbol for checking whether or not there is an error in the data symbols of modulated signals A and B on the receiving side. The specific symbol 8905 is a specific symbol (STBC symbol or a special symbol as shown in FIGS. 41 and 42) as described in the ninth embodiment.

  In this embodiment, as shown in FIGS. 80A, 80B, and 80C, transmission frames having different specific symbol insertion intervals are used. 80A shows a frame configuration in which no specific symbol is inserted, FIG. 80B shows a frame configuration in which specific symbols are inserted at intervals of 22 symbols, and FIG. 80C shows a specific symbol. A frame configuration to be inserted at intervals of 16 symbols is shown. In this embodiment, transmission is performed by selectively using any one of the frame configurations in FIGS. 80A, 80B, and 80C. For example, a specific symbol can be inserted by selecting one of FIGS. 80A, 80B, and 80C according to a request from the terminal (information such as received field strength and error rate). A method of changing the interval is conceivable. In the following, a method for changing the frame configuration at the time of retransmission will be described in detail.

  FIG. 81 shows the flow of transmission signals of the base station and terminal in the present embodiment. In FIG. 81, the diagram is simplified, but the signal transmitted from the base station is actually control information, CRC symbol, specific symbol (STBC symbol or as shown in FIGS. 41 and 42) in addition to the data symbol. The signal is a frame unit composed of a special symbol.

  In FIG. 81, as shown in <1>, the base station transmits data 1A with modulated signal A, data 1B with modulated signal B, and a frame configuration in which a specific symbol is not inserted as shown in FIG. . If the terminal receives this without error, the terminal does not request retransmission as in <2>.

  Next, as shown in <3>, the base station transmits data 2A with modulated signal A, data 2B with modulated signal B, and a frame configuration in which a specific symbol is not inserted as shown in FIG. If an error occurs when this is received, the terminal makes a retransmission request as shown in <4>.

  Then, the base station receives data 2A with modulated signal A and data 2B with modulated signal B as shown in <5>, as shown in FIG. 80 (B), in which the reception quality is improved as compared with the frame configuration of FIG. Is transmitted using a frame structure in which a specific symbol is inserted into the frame. If the terminal receives this without error, the terminal does not request retransmission as in <6>.

  Next, as shown in <7>, the base station transmits data 3A with modulated signal A, data 3B with modulated signal B, and a frame configuration in which a specific symbol is not inserted as shown in FIG. If an error occurs when this is received, the terminal makes a retransmission request as shown in <8>.

  Then, the base station receives data 3A with modulated signal A and data 3B with modulated signal B as shown in <9>, and the reception quality is improved as compared with the frame configuration of FIG. Is transmitted using a frame structure in which a specific symbol is inserted into the frame. If an error still occurs when the terminal receives this, it makes a retransmission request as shown in <10>.

  Then, the base station receives the data 3A with the modulation signal A and the data 3B with the modulation signal B as shown in <11>, and the reception quality is further improved than the frame configuration of FIGS. As shown in FIG. 80 (C), transmission is performed using a frame configuration in which the insertion interval of the specific symbol is shorter than the frame configuration of FIG. 80 (B).

  FIG. 82 shows a configuration example of a terminal reception apparatus in this embodiment. In FIG. 82, components that operate in the same manner as in FIG. 82 differs from the receiving apparatus in FIG. 72 in that there is no part that performs signal point reduction using a stored signal at the time of retransmission as shown in FIG. Control information detection section 7301 extracts information on the transmission method (frame configuration) from control information symbols 8901 included in the frame of FIG. 80, and outputs frame configuration signal 7302. Then, the signal processing unit 404 performs demodulation processing based on the frame configuration signal 7302 and outputs digital signals RA and RB.

  Thus, according to the present embodiment, as the number of retransmissions increases, the number of retransmissions can be reduced by shortening the insertion interval of specific symbols that contribute to the improvement of reception quality (in other words, increasing the number of insertions). Can do. Thereby, the data transmission efficiency can be further improved. This is because the reception quality improves as the number of insertions of a specific symbol increases (however, the transmission speed decreases). In other words, when there is a retransmission request and a modulated signal is transmitted, if a transmission method similar to the transmission method transmitted last time (the same number of insertions of a specific symbol) is taken, there is a possibility of an error again, and the possibility of error is reduced. This is because it is more suitable to increase the number of insertions of a specific symbol. Further, in the method of increasing the number of insertions of a specific symbol as the number of retransmissions increases, the data transmission rate decreases as the number of insertions of a specific symbol is increased. However, for example, when the modulation scheme is switched (the number of retransmissions increases). Compared with the case of using a modulation method that reduces the data transmitted in one symbol, the decrease in the data transmission rate is remarkably small.

  In this embodiment, the method and apparatus for changing the number of insertions of a specific symbol in accordance with the number of retransmissions have been described. However, changing the number of insertions of a specific symbol is not limited to being applied at the time of retransmission. For example, even if the number of insertions of a specific symbol is changed in response to a request from the terminal, the reception quality and the data transmission rate can be improved as in the embodiment.

Example 1
In this example, the multi-antenna transmission apparatus described in the previous embodiments, for example, the sixth embodiment and the seventh embodiment, which makes the interleave pattern of the modulation signal transmitted from each antenna different between the modulation signals. One embodiment related to the above will be described.

  FIG. 83 shows the configuration of the multi-antenna transmission apparatus in the case where FIG. 23 described in Embodiment 6 has one encoding unit and interleaver.

  Multi-antenna transmission apparatus 7900 has the same configuration as multi-antenna transmission apparatus 2300 of FIG. 23 described in Embodiment 6, except that encoding unit 201B and interleaver 2301B are removed from FIG. Therefore, description of portions that operate in the same manner as the operation described in Embodiment 6 is omitted.

  In this example, a method of performing reception quality improvement using different interleave patterns described in Embodiment 6 when there is one encoder and one interleaver will be described.

  Interleaver 2301A receives encoded digital signal S1A, changes the order, and sends interleaved digital signals S10A and S10B to modulators 202A and 202B.

  When the interleaving process is performed on the transmission device side as described above, it is necessary to perform the deinterleaving process on the reception side as in the case of FIG. A configuration example of the receiving apparatus in this case is shown in FIG. The configuration example in FIG. 84 corresponds to the signal processing unit 2400 in FIG. 24 described in the sixth embodiment. 24 is different from FIG. 24 in that the deinterleaver is only 2401A and 2403A, the interleaver is only 2402A and 2405A, and the soft decision units are only 503 and 512. Here, the interleavers 2401A and 2403A, the interleavers 2402A and 2405A, and the soft decision units 503 and 512 are parts that operate in the same manner as those shown separately in FIG. That's fine.

  The operation of this embodiment will be described in detail below. Further, description will be made using BPSK modulation as a modulation method.

  For example, if 12 bits are input, interleaver 2301A of multi-antenna transmitting apparatus 7900 arranges 6 bits in channels A and B after interleaving. The state at that time is shown in FIGS. 85-1 and 85-2. In these figures, # 1, # 2,..., # 12 indicate the order of data before interleaving. # 1 to # 6 are channel A symbols, and # 7 to # 12 are channel B symbols.

  Multi-antenna transmission apparatus 7900 interleaves input data as shown in FIG. 85-1 (A) with the first six symbols # 1 to # 6 and the second half with six symbols # 7 to # 12. Apply. The first six symbols are channel A symbols, and the last six symbols are channel B symbols.

  At this time, the multi-antenna transmission apparatus 7900 makes the change in the order of channel A data different from the change in the order of data in channel B. This is also evident from the fact that the arrangement of channel A symbols A1 to A6 and the arrangement of channel B symbols B1 to B6 shown in the interleaved data shown in FIG. 85-1 (B) are different. It is. However, the interleave pattern shown in FIG. 85-1 (B) is an example, and what is important here is that the interleave pattern is different for each channel.

  In this way, as shown in FIG. 85-1 (B), transmission frame configurations of channel A and channel B that are different from each other in the time axis direction are formed. As shown in FIG. 85-1 (C), for example, A5 and B1 symbols are transmitted from different antennas at the same time, and similarly, A3 and B5 symbols are transmitted from different antennas at the same time.

  Interleaver 2301A sends digital signals S10A and S10B after interleaving for channels A and B formed in this way to modulators 202A and 202B. Modulation section 202A generates a modulated signal based on S10A and transmits it from transmission antenna AN1. Similarly, modulation section 202B generates a modulated signal based on S10B and transmits it from transmission antenna AN2.

  In the signal processing unit 8000 on the reception side, the estimated baseband signal (both channels A and B) of the transmission digital signal separated by the separation unit 501 is returned to the original arrangement by the deinterleaver 2401A and then sent to the soft decision unit 503. Output.

  FIG. 85-1 (D) shows the correctness of each symbol after being decoded by the soft decision section 503. As described in the sixth embodiment, when a convolutional code or the like is used, errors generally occur continuously.

  The signal processing unit 8000 performs iterative decoding using the result shown in FIG. 85-1 (D).

  FIG. 85-1 (E) shows a replica signal (a signal estimated for signal point reduction) subjected to interleaving again based on the decoding result of FIG. 85-1 (D) in the signal processing unit 8000. This shows the situation and corresponds to the output of the interleaver 2402A of the signal processing unit 8000. In FIG. 85-1 (E), 6 symbols from the front are replicas of data transmitted on channel A, and 7th to 12th symbols from the front are replicas of data transmitted on channel B.

  FIG. 85-2 (F) shows a state where the candidate signal points of the self-modulated signal are reduced using the replica of the other channel signal, that is, a state when the signal is input to the deinterleaver 2403A of the signal processing unit 8000. FIG. 85-2 (G) shows the output of the deinterleaver 2403A, that is, the state when it is input to the soft decision unit 512.

  As can be seen from FIG. 85-2 (G), erroneous signal point selection occurs discretely. As a result, when the modulation signals A and B are decoded (for example, Viterbi decoding) by the soft decision unit 512, the modulation signals A and B described in the sixth embodiment are processed as shown in FIG. Similar to the case where the interleave pattern is made different, the reception quality is effectively improved.

  Note that the example described in this example can also be applied to multicarrier communication using OFDM as an example described in Embodiment 7. In this case, the encoding unit 201B and the interleaver 2301B may be excluded from the multi-antenna transmission apparatus 2900 described with reference to FIG. For example, what is necessary is just to comprise as shown in FIG. In FIG. 86, parts corresponding to those in FIG. The multi-antenna transmission apparatus 8200 in FIG. 86 is different from the multi-antenna transmission apparatus 2900 in FIG. 32 in that there is one encoding unit and one interleaver.

  Incidentally, when the configuration of FIG. 86 is adopted, encoding may be performed in the time axis direction or encoding in the frequency axis direction. In other words, the data order as shown in FIG. 85-1 may be rearranged so that the data of channel A and channel B are arranged in the time axis direction of a certain subcarrier (encoding in the time axis direction). The data rearrangement similar to that shown in FIG. 85-1 may be performed in the frequency axis direction (that is, over the subcarrier direction) (encoding in the frequency axis direction).

  By the way, in a configuration having one coding unit and one interleave, there is a possibility that a good reception quality may be given even if random interleaving is performed in the interleaving unit when considered in principle. However, there is a possibility that errors are biased to one modulation signal, for example, that many errors occur only in the modulation signal A. Therefore, it is technically important to use an interleave pattern in which an error is not biased to one modulation signal. An example is shown in FIG.

  As an application example of FIG. 85-1, a method of configuring one frame by repeating the frame of FIG. 85-1 a plurality of times as shown in FIG. In FIG. 87, A1, A2, and A3 correspond to the channel A symbol group (symbol composed of 6 symbols) in FIG. 85-1, and B1, B2, and B3 represent the channel B symbol group (FIG. 85-1). A1, A2, and A3 are transmitted from the antenna AN1 in FIG. 83, and B1, B2, and B3 are transmitted from the antenna AN2 in FIG. Also, an interleave pattern in which A1, A2, and A3 are different, and an interleave pattern in which B1, B2, and B3 are different may be used.

  What is important is that the symbols are interleaved in units of symbol groups composed of a plurality of symbols, and the symbols constituting the symbol groups may be interleaved.

  Therefore, the order shown in FIG. 88 may be used. At this time, A1, B2, and A3 are transmitted from the antenna AN1 in FIG. 83, and B3, A2, and B1 are transmitted from the antenna AN2 in FIG.

  Alternatively, interleaving may be performed across A1, A2, and A3, and interleaving may be performed across B1, B2, and B3. For example, if A1, A2, and A3 are each composed of 6 symbols, A1, A2, and A3 are a total of 18 symbols. The 18 symbols may be interleaved and divided into three symbol groups A1, A2, and A3. Furthermore, A1, A2, and A3 do not necessarily have the same number of symbols.

  The following is a summary of suitable interleaving methods. Here, a case will be described in which what has been called a symbol group so far is one symbol. First, one system of data is alternately distributed to channel A and channel B. This is shown in FIG. FIG. 89A shows the order of the original data, and names # 1 to # 24 are assigned according to the order of the data. It is assumed that this is alternately assigned to channel A and channel B and transmitted. Therefore, since “data # 1” is assigned to the first channel A, it is named “data # A1” as shown in FIG. 89B, and “data # 2” is first to channel B. Since it is assigned, it is named “data # B1” as shown in FIG. 89 (B). Similarly, since “data # 3” is assigned to the second channel A, it is named “data # A2”, and “data # 4” is assigned to the second channel B, so “data # 3” Named B2 ". Similarly, data sequentially assigned to channel A is named data # A3 to data # A12, and data sequentially assigned to channel B is named data # B3 to data # B12.

  Examples of data interleaving assigned to channels A and B as shown in FIG. 89 (B) are shown in FIGS. 90 (A) to (D). In FIG. 90, the horizontal axis represents frequency (subcarrier in OFDM), and the signals of channels A and B differ from 12 subcarriers of carriers 1 to 12 at the same time (for example, antennas AN1 and AN2 in FIG. 86). Shall be sent from.

  From the results, FIG. 90A is an example of interleaving with a small reception quality improvement effect, and FIGS. 90B, 90C, and 90D are examples of an interleaving with a large reception quality improvement effect. .

  First, the interleaving in FIG. 90A will be described. It is assumed that data is regularly arranged every three carriers in both channel A and channel B (however, arranging every three carriers regularly means the following processing: first, carrier 1, carrier 4, When the carrier 7 and the carrier 10 are arranged, the next operation returns to the carrier 2, followed by the carrier 5, the carrier 8, and the carrier 11. Next, the carrier 3 is returned, and then the carrier 6, the carrier 9, and the carrier 12 are arranged. Next, return to the carrier 1, and then arrange the carrier 4, the carrier 7, and the carrier 10). Consider a case where symbols are assigned to channel A and channel B according to this rule, channel A is deinterleaved on the receiving side, and the signal point reduction of channel B is performed using the result of decoding. If an error occurs in a burst manner in data # A6, data # A7, and data # A8 in the decoding result of channel A, in channel B, a signal point reduction error occurs in data # B9, data # B10, and data # B11. Will occur in bursts. As a result, the effect of improving the reception quality by interleaving is reduced.

  Next, interleaving in FIG. 90B will be described. The interleaving method shown in FIG. 90B is, in short, a method in which channel A and channel B have different symbol interleaving patterns themselves. As a result, it is possible to prevent an error in channel B burst signal point reduction caused by a burst error due to channel A decoding. As a result, the reception quality can be greatly improved for both channel A and channel B.

  Next, interleaving in FIG. 90C will be described. In FIG. 90 (C), channel A regularly arranges data every 3 carriers, and channel B regularly arranges data every 2 carriers (however, regularly arranging every 2 carriers) This means the following processing: First, when carrier 1, carrier 3, carrier 5, carrier 7, carrier 9, and carrier 11 are arranged, next returns to carrier 2, and then carrier 4, carrier 6, and carrier 8 , Carrier 10 and carrier 12. Then, return to carrier 1 and then follow the rules such as carrier 3, carrier 5, carrier 7, carrier 9 and carrier 11.

  Thus, channel A regularly arranges data every x carriers, and channel B regularly arranges every y (x ≠ y) carriers, resulting in burst errors due to decoding of channel A. An error of channel B burst signal point reduction can be prevented. As a result, the reception quality can be greatly improved for both channel A and channel B.

  In addition, the interleaving as shown in FIG. 90C, in other words, performs block interleaving for each channel (x = 2 in the figure) for each channel A and y for the channel B (y = 3 in the figure: x ≠ y). This corresponds to performing block interleaving for each symbol. As a result, errors in burst signal point reduction can be effectively reduced. Incidentally, in FIG. 28 already explained, it is said that channel A (modulated signal A) is subjected to block interleaving every 5 symbols and channel B (modulated signal B) is subjected to block interleaving every 8 symbols. be able to. In FIG. 45, it can be said that interleaving is performed every 100 symbols.

  Here, as a more preferable way of selecting x and y (x ≠ y), it is proposed that at least one of x and y is a prime number. As a result, interleaving closer to random interleaving can be realized between the channel A signal and the channel B signal, and burst errors can be further reduced.

  For example, x = 31 (prime number), y = 30, or x = 30, and y = 31 (prime number). At this time, the block size is 31 × 30 = 930. In channel A, interleaving is performed using an interleave pattern of block size 930 by a block interleaver for every 31 symbols. In channel B, interleaving is performed using an interleave pattern having a block size of 930 by a block interleaver every 30 symbols. Then, the period of the interleave pattern between channel A and channel B is 31 × 30. On the other hand, consider an interleave that is larger than 930 and has a block size of 1000. For example, x = 25, y = 40, or x = 40, y = 25. In channel A, interleaving is performed with an interleaving pattern having a block size of 1000 (= 25 × 40) by a block interleaver for every 25 symbols. Then, in channel B, interleaving is performed with an interleaving pattern having a block size of 1000 (= 40 × 25) by a block interleaver for every 40 symbols. Then, the period of the interleave pattern of channel A and channel B is 200, which is the least common multiple of x and y, and is smaller than 25 × 40. As a result, the randomness between the channel A signal and the channel B signal is reduced as compared with the case where either x or y is a prime number.

  This concept can also be applied when the number of channels is 3 or more. As an example, the case of transmission antennas with 3 and 3 channels will be described. Here, a case is considered in which block interleaving is applied to channel A every x symbols, channel B every y symbols, and channel C every z symbols. In this case, x ≠ y ≠ z and at least two values may be prime numbers. That is, when x and y are prime numbers, the block size is xyz which is their least common multiple. In channel A, interleaving is performed with a block size xyz interleave pattern by a block interleaver for each x symbol. In channel B, interleaving is performed with an interleave pattern of block size xyz by a block interleaver for each y symbol. In channel C, interleaving is performed with an interleave pattern of block size xyz by a block interleaver for each z symbol. Then, the period of the interleave pattern of channel A, channel B, and channel C is xyz. By doing so, since the cycle of the interleave pattern can be maximized, randomness can be ensured. Furthermore, the case where the number of antennas is increased and the number of transmission channels is increased can be considered similarly.

  In order to improve randomness, it is important that at least one of x and y is a prime number, and that the least common multiple of x and y is equal to the block size. For example, x = 16 and y = 27 may be set. At this time, the block size is 16 × 27 = 432. In channel A, interleaving is performed using an interleave pattern having a block size of 432 by a block interleaver every 16 symbols. In channel B, interleaving is performed using an interleave pattern of block size 432 by a block interleaver for every 27 symbols. Then, the period of the interleave pattern between channel A and channel B is 16 × 27.

  However, if at least one of x and y is a prime number, x × y is always the least common multiple of x and y, which is preferable because the design of the interleaver can be simplified.

  In addition, the interleaving process using the prime number proposed here is not limited to the application example in this embodiment. For example, “(i) Embodiment 6 of (6) the symbol of each modulation signal is used. Even when applied to an embodiment as described in the section `` Method for making the arrangement of data itself different '', as a method of performing interleaving processing with high randomness easily between channels using block interleaving It is valid. That is, it can be applied to all methods described in this specification.

  Further, in the interleaving of FIG. 90C, the channel B block interleaving is offset in the frequency direction with respect to the channel A block interleaving. As a result, burst errors can be further reduced.

  Incidentally, the probability that a burst error occurs can be reduced as the values of x and y are selected to be larger.

  Next, interleaving in FIG. 90D will be described. In FIG. 90D, data is regularly arranged every three carriers in both channel A and channel B, but channel A is from the higher frequency to the lower frequency, and channel B is from the lower frequency to the higher frequency. Arrange. Thereby, it is possible to prevent an error of channel B burst signal point reduction caused by a burst error due to decoding of channel A. As a result, the reception quality can be greatly improved for both channel A and channel B.

  90 (B), 90 (C), and 90 (D) may be realized by a multi-antenna transmission apparatus 7900 configured as shown in FIG. 83, for example. You may implement | achieve with the multi-antenna transmission apparatus 8400. 91, parts corresponding to those in FIG. 83 are denoted by the same reference numerals. The multi-antenna transmission apparatus 7900 in FIG. 83 and the multi-antenna transmission apparatus 8400 in FIG. 91 are different from each other in that the interleaving processing in FIGS. 90B, 90C, and 90D is performed by one interleaver 2301A. Or interleavers 8401A and 8401B provided for each channel. Specifically, in multi-antenna transmission apparatus 8400, the encoded data is serial-parallel converted and distributed to channel A interleaver 8401A and channel B interleaver 8401B. The channel A interleaver 8401A and the channel B interleaver 8401B perform the above-described interleaving.

  Note that the configuration of the transmission device is not limited to the configurations of FIGS. 86 and 91, and any configuration that can implement the interleaving processing shown in FIGS. 90B, 90C, and 90D. Such a configuration may be used. Further, as a configuration on the receiving side, any configuration may be used as long as it has a configuration including a portion for performing deinterleaving processing corresponding to interleaving and a portion for performing signal point reduction described above.

  90B, 90C, 90D, and the other suitable interleaving methods described in the above embodiments are examples of the frequency axis or the frequency axis in order to simplify the description. Although the case where interleaving is performed in any direction of the time axis has been described, even if the interleaving in the frequency axis direction is described as an example and applied in the time axis direction, it can be similarly performed, and similarly the time axis The present invention can be implemented in the same manner by applying the interleaving in the direction as an example to the frequency axis direction. Further, the above description is an example for explaining the features of the present invention, and the interleaving and deinterleaving methods are not limited to this, and the same effect can be obtained by using the feature portions of the above explanation. it can.

(Example 2)
In the first embodiment, an example in which the interleave pattern of the signal transmitted from each antenna is different depending on the interleaver has been described. However, in the present embodiment, the interleaver uses the same interleave pattern for the signal transmitted from each antenna. An example in which the same effect as in the first embodiment is obtained by performing interleaving processing and performing different symbol allocation among the antennas when allocating to subcarriers will be described.

  FIG. 92 shows the configuration of multi-antenna transmission apparatus 8300 in the present embodiment. In FIG. 92, components that operate in the same manner as in FIG. The difference between multi-antenna transmission apparatus 8300 and multi-antenna transmission apparatus 8200 is that signal allocation sections 8301A and 8301B for subcarriers are provided.

  Here, in the signal allocation units 8301A and 8301B, the output order of the input baseband signals S2A and S2B is different from each other, whereby the OFDM signal transmitted from the antenna AN1 and the OFDM signal transmitted from the antenna AN2 The order of the symbol arrangement on the subcarriers can be made different. As a result, the multi-antenna transmission apparatus 8300 has a function similar to that of forming a signal having a different interleave pattern between the respective channels (antennas) by the interleaver 2301A according to the first embodiment, and assigns signals to the subcarriers 8301A, It can be realized by 8301B.

  Next, the operation of multi-antenna transmission apparatus 8300 will be described with reference to FIG. As shown in FIG. 93 (A), the data before interleaving constitutes one data series with channel A and channel B. Here, it is assumed that one data series is composed of 60 pieces of data. The first half 30 are the channel A symbols (# 1 to # 30), and the second half 30 are the channel B symbols (# 31 to # 60). Since the data of # 1 is the first symbol of channel A, it is indicated by the number A1 in the figure. Similarly, the numbers up to A30 are sequentially given to the symbols of channel A. Also, since the data of # 31 is the first symbol of channel B, it is indicated by the number B1 in the figure. Similarly, the numbers up to B30 are given to the symbols of channel B.

  When the data as shown in FIG. 93 (A) is input, the interleaver 2301A forms blocks in units of 10 for the data from data A1 to data A30 as shown in FIG. 93 (B). The order of data is rearranged by reading. As a result, the interleaver 2301A, as the interleaved digital signal S10A, is data in the order of A1, A11, A21, A2, A12, A22,..., A10, A20, A30 as shown in FIG. Is output. Similarly, the interleaver 2301A forms blocks in units of 10 for the data from the data B1 to the data B30 as shown in FIG. 93C, and rearranges the order of the data by sequentially reading the blocks. . As a result, the interleaver 2301A, as the interleaved digital signal S10B, has data in the order of B1, B11, B21, B2, B12, B22,..., B10, B20, B30 as shown in FIG. Is output.

  The difference from the first embodiment described above is that, in the first embodiment, the interleaver 2301A changes the order of the data of the channel A and the order of the data of the channel B, but in this embodiment, Channel A and channel B use the same interleave pattern.

  When signal allocation section 8301A to subcarriers receives data of channel A after interleaving shown in FIG. 93 (D), each data A1 to A30 is arranged in subcarriers as shown in FIG. 93 (F). Output data 8202A is formed. On the other hand, when signal interleaved by subcarrier 8301B receives channel B data after interleaving shown in FIG. 93 (E), data B1 to B30 are arranged in subcarriers as shown in FIG. 93 (G). The output data 8202B is formed. In the example shown in FIGS. 93 (F) and 93 (G), the arrangement of channel B data on subcarriers is offset by three symbols with respect to the arrangement of subcarrier A data on subcarriers. In other words, this corresponds to interleaving channel A data (modulated signal A) and channel B data (modulated signal B) with different interleave patterns in the frequency direction.

  On the receiving side, as shown in FIG. 94, deinterleaving processing is performed on the signals transmitted in this way, and soft decision decoding processing is performed. Then, as described in the sixth embodiment, when a convolutional code or the like is used, errors continuously occur.

  However, when iterative decoding is performed by the signal processing unit 8000 in FIG. 84, the interleave pattern differs between channels as described in the sixth embodiment and the first embodiment. As in (G), erroneous signal point selection occurs discretely. Accordingly, when the data (modulated signals A and B) of each channel is decoded by the soft decision unit 512, the reception quality is effectively improved as shown in FIG. 85-2 (H).

  In the present embodiment, the case where the channel A (modulated signal A) and the channel B (modulated signal B) are set to different interleave patterns has been described with reference to FIG. 93, but the interleave pattern is not limited to this. . In short, the symbol allocation pattern on the subcarriers may be different between channels (antennas).

  Incidentally, the same processing as in the present embodiment can also be performed by the multi-antenna transmission apparatus 8200 having the configuration of FIG. In this case, the interleaver 2301A in FIG. 86 may be configured to include the functions of the subcarrier allocation units 8301A and 8301B in FIG.

  Further, in the first and second embodiments, the method of improving the reception quality by forming signals with different interleave patterns between channels when there is one encoder and one interleaver has been described. The data arrangement method is not limited to the methods of the first and second embodiments. In addition, when LDPC is applied as an error correction code, as described in the fourteenth embodiment, an interleaving method in which interleaving is not performed for channel A data and interleaving is performed for channel B data is The method performed by the interleaver can also be performed in the same manner as in the first and second embodiments.

  Further, in the first embodiment and the second embodiment, the case where the number of channels (antennas) is two has been described. However, the present invention is not limited to this, and even when the number of channels (antennas) is three or more, between each channel (antenna) By making the interleave pattern different, the same effect as described above can be obtained. The same applies to the other embodiments described above in which the interleave pattern is different between channels (antennas).

Example 3
In this example, the multi-antenna transmission in which the interleave pattern of the modulation signal transmitted from each antenna is different between the modulation signals described in the previous embodiments, for example, the sixth embodiment and the seventh embodiment. An example in which the same data is transmitted in each modulated signal in order to improve reception quality in the apparatus and the receiving apparatus that receives and demodulates the modulated signal will be described.

  FIG. 95 shows a configuration example of the multi-antenna transmission apparatus in the present embodiment. In FIG. 95, components that operate in the same way as in FIG. The multi-antenna transmission apparatus 9000 of FIG. 95 differs from the multi-antenna transmission apparatus 2300 of FIG. 23 in that the multi-antenna transmission apparatus 9000 transmits a modulated signal of channel A (modulated signal transmitted from the antenna AN1) and a modulated signal of channel B (antenna The modulation signal transmitted from AN2 is formed by applying different interleave patterns to the same data TA.

  FIG. 96 shows a frame configuration of a modulation signal transmitted by the multi-antenna transmission apparatus 9000 of the present embodiment. FIG. 96A shows the order of data # 1, # 2,..., # 11, # 12 before interleaving. FIG. 96 (B) shows the order of data # 1 to # 12 after being interleaved with different interleave patterns by interleavers 2301A and 2301B of multi-antenna transmission apparatus 9000. Interleaver 2301A performs interleaving so as to have a frame configuration like channel A in FIG. On the other hand, interleaver 2301B performs interleaving so as to have a frame configuration like channel B in FIG.

  The receiving apparatus in the present embodiment may be configured as shown in FIG. 4, for example. And what is necessary is just to comprise the signal processing part 404 of FIG. 4 as shown in FIG. In FIG. 97, components that operate in the same manner as in FIG. 97 is different from the signal processing unit 2400 in FIG. 24 in that the signal processing unit 9200 in FIG. 97 includes soft decision units 503 and 506 for temporary determination by the signal processing unit 2400 in FIG. Two soft decision units 512 and 518 for main determination are provided, whereas one soft decision unit 503 for temporary determination and one soft decision unit 512 for main determination are provided. It is a point that has.

  97 are deinterleavers for the interleaver 2301A in FIG. 95, and deinterleavers 2401B, 2403B, and 2404B are deinterleavers for the interleaver 2301B in FIG.

  Signals input to the soft decision unit 503 of the signal processing unit 9200 are estimated baseband signals 502 and 505 based on two channels, channel A and channel B, including information on the transmission digital signal TA. At this time, the soft decision unit 503 obtains the branch metric and path metric and performs decoding. However, since the estimated baseband signals 502 and 505 are interleaved with completely different interleave patterns, the estimated baseband signal The quality of the path metric obtained from 502 and the quality of the path metric obtained from the estimated baseband signal 505 are completely different. Therefore, by using the path metric obtained from both, the accuracy of soft decision is improved and the reception quality of data is improved. The same effect can be obtained in the soft decision unit 512.

  As a result, it is possible to obtain an effect that the reception quality of data is greatly improved.

  Note that the example described in this example can also be applied to multicarrier communication using OFDM as an example described in Embodiment 7. Incidentally, the encoding may be performed in the time axis direction or in the frequency axis direction.

  The method described in Embodiment 6 may be applied as a method for generating different interleave patterns. Further, the description has been given by taking the number of transmission antennas 2 and the number of reception antennas 2 as an example. However, the present invention is not limited to this. For example, when the number of transmission antennas is 3 Thus, it can be similarly implemented.

(Example 4)
By the way, as described in the first embodiment, when performing block interleaving for each x symbol in channel A and performing block interleaving for each y symbol symbol in channel B, the symbols are arranged in the frequency axis direction. Considering the correlation of propagation along the frequency axis and the correlation of propagation along the time axis when symbols in the time axis direction are arranged, it is better to increase x and y as much as possible. In the present embodiment, a block interleave design method for realizing this will be described.

  For example, in channel A and channel B, it is assumed that 48 block sizes are interleaved as data symbols. As an example of channel A and channel B interleaving in which the block size is the least common multiple in a block size of 48 symbols, channel A performs interleaving of block size 48 by an interleaver every 3 symbols, and channel B has 16 symbols. It is conceivable to perform interleaving with a block size of 48 by each interleaver. However, since the block interleaving for every 3 symbols applied to channel A is a small value such as every 3 symbols, the influence of the propagation correlation becomes high, and as a result, the reception quality tends to deteriorate.

  Therefore, a method is proposed in which, for example, 8 is added to the block size 48 to virtually obtain a block size of 48 + 8 = 56, and channel A and channel B signals are interleaved with this block size. Here, 8 is added to the block size 48 to obtain 56, because block interleaving for every 7 symbols and block interleaving for every 8 symbols are assigned to channel A and channel B, so that the condition of block size = least common multiple is , X and y are set so as to satisfy both conditions of making the values as large as possible.

  Hereinafter, the proposed method in this embodiment will be described in detail.

  In FIG. 98, for the sake of explanation, 56 data virtually constituting a block are numbered to indicate the order. FIG. 98A shows the order of channel A and channel B before interleaving. However, there are actually 8 items for which no data exists. As shown in FIG. 98 (A), in channel A, data # A2, # A5, # A9, # A13, # A44, # A48, # A52, and # A55 are assumed to have no data. In channel B, data # B2, # B5, data # B9, # B13, # B44, # B48, # B52, and # B55 are assumed to have no data.

  In channel A, regardless of the presence or absence of data, interleaving with a block size of 56 is performed by an interleaver every 7 symbols. In channel B, regardless of the presence or absence of data, interleaving with a block size of 56 is performed by an interleaver every 8 symbols. This is shown in FIG.

  Next, data # A2, # A5, # A9, # A13, # A44, # A48, # A52, # A55 for which no data exists in channel A are thinned out. Similarly, data # B2, # B5, # B9, # B13, # B44, # B48, # B52, and # B55 for which no data exists in channel B are thinned out. FIG. 98C shows the arrangement of channel A and channel B data after thinning.

  This FIG. 98C is the final result of interleaving. As a result, the correlation between the interleave patterns of channel A and channel B becomes low and the pattern period can be lengthened, so that the reception quality is greatly improved. Thus, if the method of virtually setting the block size = the least common multiple of x and y and increasing x and y is used, design flexibility can be increased.

  99 and 100 show an example of an actual interleaving process when such a method is performed. FIG. 99 shows a write state of transmission data # A1 to # A56 in the memory of the interleaver. Interleaving is realized by writing transmission data # A1 to # A56 sequentially with priority in the horizontal direction at the time of writing, and sequentially reading transmission data # A1 to # A56 with priority in the vertical direction at the time of reading. However, the order of writing and reading may be set as appropriate depending on the applied interleave pattern.

  FIG. 100 shows the relationship between the transmission data # A1 to # A56 and the memory address when the transmission data # A1 to # A56 is written as shown in FIG. In this embodiment, as shown in FIG. 100, data presence / absence information indicating whether or not data actually exists is provided in association with the address. As a result, when data is read from the memory, an address corresponding to the data presence / absence information (0) indicating that there is no data is skipped, so that an interleaving process with a virtually increased block size can be realized.

  Further, the effect of the interleaving method using the thinning described in this embodiment will be described. Although the interleaving method of the present embodiment is a block interleaving, it has a special effect of eliminating the correlation of propagation on the time axis or the frequency axis as in the case of random interleaving.

  For example, consider the case where the block size is 22. At this time, the interleaver of the block size 22 can be used by the interleaver every 2 symbols, or the interleave of the block size 22 can be used by the interleaver every 11 symbols. Therefore, it is difficult to eliminate the propagation correlation. However, using the above-described interleaving method, 3 is added to obtain a virtual block size of 25, and both channels A and B are interleaved with a block size of 25 by an interleaver every 5 symbols. By performing decimation in the same manner as in the procedure of FIG. 98 (C), interleaving with a block size of 22 interleaved by an interleaver for every 2 symbols or an interleaving in which propagation correlation has been eliminated by an interleaver for every 11 symbols rather than an interleave of a block size of 22 The special effect that can be given. Thus, the interleaving method of the present embodiment can obtain a special effect even with a single interleaving. In addition, the method described in this embodiment can be applied in combination with all the methods described in this specification.

(Other embodiments)
In the above-described embodiment, the case where the digital signal is obtained by performing the soft decision is mainly described. However, the present invention is not limited to this, and the present invention can be applied to the case where the digital signal is obtained by obtaining the hard decision. Even in this case, it is possible to obtain received data with good error rate characteristics with a small number of computations.

  In the above-described embodiment, the case where all the determination values provisionally determined by the separation unit 501 and the soft determination units 503, 506, and 1101 are used for the signal point reduction processing has been described. May be used as final received data. For example, data for which high reception quality is not required may be output as it is without performing main determination by the soft decision units 512 and 518.

  Further, in the above-described embodiment, the spread spectrum communication method has been mainly described as an example. However, the present invention is not limited to this. For example, a single carrier method without a spreading unit and an OFDM method can be similarly implemented. it can. In the case of a single carrier system, the configuration does not include a diffusion unit and a despreading unit. The same can be applied to the case where the multicarrier scheme and the spread spectrum communication scheme are used together (for example, OFDM-CDMA scheme).

  Further, a method of transmitting a modulation signal with different interleave patterns between the channels (antennas) described above is described in, for example, the document “Eigenbeam Space Division Multiplexing (E-SDM) System in MIMO Channel”, IEICE, IEICE. The same effect as described above can be obtained also when applied to a MIMO system that transmits a transmission signal in multi-beams as described in Academic Report RCS2002-53, May 2002. .

  FIG. 101 shows a schematic configuration of such a MIMO system. On the transmission side, a modulation unit 8601 receives a transmission data sequence and modulates it to form a plurality of transmission frames. Here, the modulation unit 8601 forms a modulation signal with different interleave patterns between channels (antennas). Channel analysis section 8602 calculates a plurality of transmission channel signature vectors for constituting a multiplexed channel, based on channel state information that is a propagation channel estimation result. Vector multiplexing section 8603 multiplies each transmission frame by a different channel signature vector and combines them, and sends the combined signal to transmission array antenna 8604. As a result, a multi-beamed signal is transmitted from the transmission array antenna 8604.

  On the reception side, the channel analysis unit 8611 calculates a plurality of reception channel signature vectors for separating the multiplexed transmission signals based on the channel state information that is the estimation result of the propagation channel. Multiplex signal separator 8613 receives the reception signals of reception array antenna 8612 and multiplies each reception signal by a different channel signature vector, thereby converting a signal in which a plurality of transmission frames are multiplexed into a plurality of reception signal frames. To separate. The signal processing unit 8614 obtains received data by demodulating and decoding the separated received signal frame. Here, the signal processing unit 8614 has the above-described deinterleave processing and signal point reduction processing functions. Thereby, it is possible to obtain received data with good error rate characteristics, as in the sixth embodiment and the first embodiment.

  Furthermore, when the retransmission method described in Embodiment 15 is applied to a MIMO system that performs beam forming as shown in FIG. 101, the signal quality at the time of retransmission in the MIMO system that performs beam forming can be improved. . That is, in performing retransmission, a beam is formed by reducing the number of modulated signals to be transmitted compared to the previously transmitted modulated signal. For example, when the modulated signals A and B are formed and transmitted for the first time, at the time of retransmission, the beams are not formed from the modulated signals A and B and retransmitted, but only from the data of one modulated signal. Form and retransmit. In this way, since the number of beams decreases at the time of retransmission, inter-beam interference can be reduced, and as a result, the quality of the retransmission signal can be improved.

  In this case, the beam position at the time of retransmission may be changed from that at the previous transmission. For example, if the modulation signal A is transmitted by the beam 1 at the previous transmission and the modulation signal B is transmitted by the beam 2 and the modulation signal B is requested to be retransmitted, the modulation signal B is transmitted by the beam 1 at the time of retransmission. This is preferable because the quality at the time of retransmission of the modulated signal B can be further improved. That is, the fact that there is no retransmission request for the modulated signal A is more likely to be a beam that can be transmitted with better quality than the beam 2. In this way, the quality of the retransmission signal can be further improved by sending the retransmission signal with a beam that can be transmitted with better quality at the time of retransmission.

  Further, in the embodiment described above, the case where the number of transmission antennas is 2 and the number of reception antennas is 2 has been described. However, the present invention is not limited to this, and the number of transmission antennas is 3 or more, or the number of reception antennas is 3 or more. In the case of, the same can be carried out.

  Furthermore, various methods can be applied as a special symbol insertion method when using an LDPC code. For example, unlike the case of using a convolutional code or a turbo code, when an LDPC code is used, the encoding unit also has an interleaving function, so that special symbols need not be regularly inserted. Therefore, special symbols may be inserted partially and continuously.

  The present invention is not limited to the embodiment described above, and can be implemented with various modifications.

  This specification includes Japanese Patent Application No. 2003-391860 filed on November 21, 2003, Japanese Patent Application No. 2004-38885 filed on January 9, 2004, Japanese Patent Application No. 2004-71780 filed on March 12, 2004, and May 5, 2004. Japanese Patent Application No. 2004-139241 filed on May 7, 2004, Japanese Patent Application No. 2004-146887 filed on May 17, 2004, Japanese Patent Application No. 2004-180277 filed on June 17, 2004, and Japanese Patent Application No. 2004 filed on November 1, 2004. Based on -318521. The contents are all included here.

  The present invention is suitable for application to a multi-antenna communication system in which high-speed data communication is performed using OFDM-MIMO (Multiple-Input Multiple-Output) technology or the like.

The figure which shows schematic structure of a multi-antenna communication system Block diagram showing the configuration of the multi-antenna transmission apparatus A diagram showing a frame configuration example of a baseband signal Block diagram showing the overall configuration of the multi-antenna receiving apparatus FIG. 3 is a block diagram showing a configuration of a signal processing unit of the multi-antenna reception apparatus according to Embodiment 1 The block diagram which shows the structure of the soft decision part 503 (506) The figure which uses for description of the process in the soft decision part 503 (506) The figure which shows the candidate signal point and receiving point of the multiplexed modulated signal A and the modulated signal B The figure which shows the reduced candidate signal point and receiving point about the modulation signal A The figure which shows the reduced candidate signal point and receiving point about the modulation signal B FIG. 3 is a block diagram showing another configuration example of the signal processing unit used in the multi-antenna reception apparatus of the first embodiment FIG. 9 is a block diagram showing a configuration of a signal processing unit of a multi-antenna reception apparatus according to Embodiment 2 FIG. 9 is a block diagram showing a configuration of a signal processing unit of a multi-antenna reception apparatus according to Embodiment 3 The figure for demonstrating the iteration operation | movement by Embodiment 3. FIG. The figure which shows the image of the decoding procedure in Embodiment 3. FIG. 6 is a characteristic curve diagram showing a simulation result of the multi-antenna receiving apparatus of Embodiment 3, where (A) is a characteristic curve diagram of modulated signal A, and (B) is a characteristic curve diagram of modulated signal B. FIG. 9 is a block diagram showing another configuration example of the signal processing unit used in the multi-antenna reception apparatus of the third embodiment FIG. 9 is a block diagram illustrating a configuration of a signal processing unit of a multi-antenna reception apparatus according to a fourth embodiment. The figure for demonstrating the iteration operation | movement by Embodiment 4. FIG. FIG. 6 is a characteristic curve diagram showing a simulation result of the multi-antenna receiving apparatus of Embodiment 4, where (A) is a characteristic curve diagram of modulated signal A, and (B) is a characteristic curve diagram of modulated signal B. FIG. 10 is a diagram illustrating a signal point arrangement example of each modulation signal in Embodiment 5 ((A) is a signal point arrangement of modulation signal A, and (B) is a signal point arrangement of modulation signal B) Characteristic curve diagram showing reception quality of QPSK and 16QAM Block diagram showing a configuration of a multi-antenna transmission apparatus according to a sixth embodiment Block diagram for explaining deinterleaver processing It is a figure which shows an example of the state of the symbol in case the interleave pattern of the modulation | alteration signal A and the modulation | alteration signal B is the same, (A) is the state of the modulation signal after the 1st determination, (B) is after reducing the number of signal points. Diagram showing the state of An example of the state of a symbol when the method of Embodiment 6 is applied to make the interleave pattern for modulated signal A different from the interleave pattern for modulated signal B is shown in FIG. (B) is a diagram showing a state after reducing the number of signal points The figure which shows the receiving characteristic when making the interleave pattern different between the modulation signals and when making it the same FIG. 17 is a diagram illustrating an example of an interleave pattern according to Embodiment 6, where (A) illustrates an interleave pattern X applied to modulated signal A, and (B) illustrates an interleave pattern Y applied to modulated signal B. FIG. 17 is a diagram illustrating an example of an interleaving pattern according to Embodiment 6, where (A) shows the arrangement of symbols before and after interleaving, (B) shows the arrangement of symbols of modulated signal A, and (C) shows the modulation signal B. Diagram showing symbol placement Block diagram showing a configuration of a multi-antenna transmission apparatus according to a seventh embodiment FIG. 18 is a diagram illustrating a frame configuration example of each modulation signal according to the seventh embodiment, where (A) is a frame configuration of modulation signal A, and (B) is a diagram illustrating a frame configuration of modulation signal B. FIG. 10 is a block diagram showing another configuration of the multi-antenna transmission apparatus according to the seventh embodiment. FIG. 9 is a block diagram showing a configuration of a multi-antenna reception apparatus according to a seventh embodiment. Diagram for explaining the principle of the eighth embodiment The figure which shows the example of a frame structure in the case of inserting an STBC symbol Diagram for explaining STBC transmission / reception Block diagram showing a configuration example for inserting STBC symbols FIG. 10 is a block diagram illustrating a configuration example of a signal processing unit of the multi-antenna reception apparatus according to the ninth embodiment. FIG. 10 is a block diagram illustrating a configuration example of a signal processing unit of the multi-antenna reception apparatus according to the ninth embodiment. The figure which shows the example of a receiving state when an STBC symbol is inserted Diagram showing an example frame configuration of special symbols Diagram showing an example frame configuration of special symbols Block diagram showing a configuration example for inserting STBC symbols The figure which shows the example of a reception state when a special symbol is inserted The figure which shows the encoding symbol block after the encoding symbol block and the interleaving FIG. 10 is a diagram for explaining the operation of the tenth embodiment. Block diagram showing a configuration of a multi-antenna transmission apparatus according to an embodiment 10 It is a figure where it uses for description of the signal point arrangement | positioning of each modulation signal, (A) shows the signal point arrangement | positioning of the modulation signal A, (B) shows the signal point arrangement | positioning of the modulation signal B Block diagram showing a configuration of a multi-antenna transmission apparatus according to an eleventh embodiment FIG. 10 is a diagram for explaining bit interleaving processing according to the eleventh embodiment. FIG. 11 is a block diagram illustrating a configuration example of a signal processing unit of the multi-antenna reception apparatus according to the eleventh embodiment. The figure used for explanation of signal point reduction processing by the signal point reduction unit The figure which shows the image of the decoding procedure in Embodiment 11 It is a figure which shows the mode of the signal point selection at the time of making the bit interleave pattern the same between modulation signals, (A) shows the state after the first determination, (B) shows the state after reducing the number of signal points. Illustration FIG. 20 is a diagram illustrating a state of signal point selection when the bit interleave pattern according to the eleventh embodiment is used, in which (A) shows a state after the first determination, and (B) shows a state after reducing the number of signal points. Illustration Block diagram showing a configuration of a multi-antenna transmission apparatus according to an embodiment 12 FIG. 19 is a diagram for explaining bit interleaving processing according to the twelfth embodiment, (A) shows an interleave pattern X, and (B) shows an interleave pattern Y. FIG. 13 is a block diagram illustrating a configuration example of a signal processing unit of the multi-antenna reception apparatus according to the twelfth embodiment. It is a figure which shows the mode of the signal point selection at the time of making the bit interleave pattern the same between modulation signals, (A) shows the state after the first determination, (B) shows the state after reducing the number of signal points. Illustration FIG. 19 is a diagram illustrating a state of signal point selection when the bit interleave pattern according to the twelfth embodiment is used, in which (A) shows a state after the first determination, and (B) shows a state after reducing the number of signal points. Illustration FIG. 19 is a diagram for describing bit interleaving processing according to the thirteenth embodiment, where (A) shows an interleave pattern X and (B) shows an interleave pattern Y. FIG. 15 is a block diagram illustrating a configuration example of a signal processing unit of the multi-antenna reception apparatus according to the thirteenth embodiment. FIG. 15 is a block diagram illustrating a configuration example of a multi-antenna transmission apparatus according to the thirteenth embodiment. Block diagram showing a configuration of a multi-antenna transmission apparatus of an embodiment 14 FIG. 14 is a block diagram illustrating a configuration example of a signal processing unit of a multi-antenna reception apparatus according to an embodiment 14; Block diagram showing another configuration example of the fourteenth embodiment The figure which shows the example of a transmission frame structure of Embodiment 15. FIG. FIG. 15 is a block diagram illustrating a configuration example of a transmission system of the multi-antenna reception apparatus according to the fifteenth embodiment. The figure which shows the flame | frame structure transmitted from the transmission system of a multi-antenna receiver. Block diagram showing a configuration of a multi-antenna transmission apparatus according to a fifteenth embodiment The figure for demonstrating the operation | movement of Embodiment 15. Block diagram showing a configuration of a reception system of a multi-antenna reception apparatus according to a fifteenth embodiment 72 is a block diagram showing the configuration of the signal processing unit of FIG. Diagram for explaining data stored in channel information / received signal storage The figure for demonstrating the operation | movement of Embodiment 15. The figure for demonstrating the operation | movement of Embodiment 16. Diagram for explaining Cycled Delay Diversity Block diagram showing a configuration of a multi-antenna transmission apparatus according to an embodiment 17 FIG. 19 is a diagram for explaining an MLD-S (MLD-Soft Decision Decoding) decoding method according to the eighteenth embodiment. The figure which shows the transmission frame structure of Embodiment 19. FIG. The figure showing an example of the retransmission operation of the nineteenth embodiment Block diagram showing a configuration of a multi-antenna reception apparatus according to an nineteenth embodiment 1 is a block diagram illustrating a configuration of a multi-antenna transmission apparatus according to a first embodiment. FIG. 3 is a block diagram illustrating a configuration example of a signal processing unit of the multi-antenna reception apparatus according to the first embodiment. It is a figure which shows the mode of the interleaving process and signal point selection in Example 1, (A) shows the arrangement | sequence of the data before interleaving, (B) shows the arrangement | sequence of the data after interleaving, (C) is after interleaving. (D) shows the data state after deinterleaving after the first decoding, and (E) shows the state of data after the first decoding after interleaving before signal point reduction. Figure showing It is a figure which shows the mode of the interleaving process and signal point selection in Example 1, (F) shows a state when a signal point is reduced using a replica, (G) reduces a signal point using a replica. The state after deinterleaving is shown, (H) is the state after Viterbi decoding 1 is a block diagram illustrating a configuration of a multi-antenna transmission apparatus according to a first embodiment. The figure which shows the transmission frame of Example 1. FIG. The figure which shows the transmission frame of Example 1. FIG. Diagram for explaining interleaving in the first embodiment Diagram for explaining interleaving in the first embodiment FIG. 3 is a block diagram showing another configuration of the multi-antenna transmission apparatus according to the first embodiment. Block diagram showing a configuration of a multi-antenna transmission apparatus according to a second embodiment. FIG. 10 is a diagram for explaining an interleaving process according to the second embodiment, where (A) shows the order of data before interleaving, (B) shows an interleaving method for channel A, and (C) shows an interleaving method for channel B. , (D) shows the order of channel A after interleaving, (E) shows the order of channel B after interleaving, (F) shows the assignment of channel A to subcarriers, and (G) shows channel B Of allocation to subcarriers FIG. 10 is a diagram for explaining reception processing according to the second embodiment. Block diagram showing a configuration of a multi-antenna transmission apparatus according to a third embodiment. The figure where it uses for description of the interleaving processing of execution example 3 FIG. 6 is a block diagram illustrating a configuration of a signal processing unit of a multi-antenna reception apparatus according to a third embodiment. FIG. 10 is a diagram for explaining an interleaving process according to the fourth embodiment, in which (A) shows the arrangement of the data of each channel before interleaving, (B) shows the arrangement of the data of each channel after interleaving, and (C) shows The figure which shows the arrangement of the data of each channel after thinning The figure where it uses for description of the interleaving processing of execution example 4 The figure where it uses for description of the interleaving processing of execution example 4 The block diagram which shows the structure of the MIMO system of other embodiment. The figure which shows schematic structure of a general multi-antenna communication system Block diagram showing the configuration of a conventional multi-antenna communication system Diagram for explaining space-time block codes

Claims (2)

A wireless communication method of a wireless communication device having a plurality of antennas,
Transmitting a plurality of modulated signals having different transmission data to the communication partner using the plurality of antennas;
(I) a first retransmission method for retransmitting one modulated signal using one antenna, or
(Ii) using a weight calculated from information shared by communication with the communication partner and weighting and combining signals for retransmission of the same number or a smaller number of the plurality of modulation signals, the same frequency band A second retransmission method for retransmitting using the same or less number of antennas as the plurality of antennas,
A step of selecting
Using the selected retransmission method, the wireless communication device transmitting retransmission data to a communication partner;
A wireless communication method including: A plurality of antennas for transmitting modulated signals;
A retransmission request detection unit for detecting a retransmission request of transmission data transmitted to a communication partner using a plurality of modulated signals having different transmission data from the plurality of antennas;
Based on the retransmission request,
(I) a first retransmission frame configuration for retransmitting one modulated signal using one antenna, or
(Ii) using a weight calculated from information shared by communication with the communication partner and weighting and combining signals for retransmission of the same number or a smaller number of the plurality of modulation signals, the same frequency band A second retransmission frame configuration for retransmitting using the same or less number of antennas as the plurality of antennas,
A frame configuration signal generator that outputs a frame configuration signal indicating the selected retransmission frame configuration;
Based on the retransmission request, a data selection unit for the wireless communication device to select retransmission data for the communication partner;
A transmitter that retransmits the selected retransmission data based on the frame configuration signal;
A wireless communication apparatus comprising:

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