JP4358890B2 - Local oscillator leakage control in direct conversion process - Google Patents

Description

[Field]
The present invention generally relates to wireless communications. In particular, the present invention relates to systems and methods for direct conversion transceivers.

[Background and related technologies]
The field of communications has experienced enormous growth, largely due to the improved capabilities of wireless devices. Wireless devices use radio waves to enable long-distance communications based on the line without the physical constraints of the system. Information such as voice, data, or paging information is transmitted by radio waves transmitted on a preset frequency band. The allocation of available frequency spectrum is regulated to ensure that a large number of users can communicate without undue interference.

  Most of the information to be transmitted from the source to the destination is not available in a format prepared for wireless transmission. Typically, a transmitter receives an input signal and formats it for transmission on a preset frequency band. An input signal, also referred to as a baseband signal, modulates the carrier in the required frequency band. For example, a wireless transmitter that receives an audio input signal modulates the carrier frequency with the input signal.

  A corresponding remote receiver tuned to the same carrier frequency as the transmitter must receive and demodulate the transmitted signal. This means that the remote receiver must recover the baseband signal from the modulated carrier. The baseband signal may be provided directly to the user or may be further processed prior to being provided to the user. Many consumer wireless devices such as radios, televisions, and pagers are simply receivers.

  A transceiver is a wireless device that integrates a transmitter and a receiver in a single package. The transceiver is capable of almost instantaneous bi-directional communication. Examples of transceivers include interactive radios, walkie-talkies, interactive pagers, and wireless telephones.

  In evaluating the effectiveness of receiver design, several figures-of-merit are important. Sensitivity determines the receiver's ability to detect weak signals. The receiver sensitivity must be such that the receiver can detect minute identifiable signals (MDS) from background noise. Noise gives random fluctuations in voltage and current. MDS is a measure of receiver-specific sensitivity that incorporates the bandwidth of a given system. On the other hand, receiver selectivity indicates protection from off-channel interference given to the receiver. The higher the selectivity, the better the receiver can eliminate unnecessary signals.

  Desense is a reduction in the overall sensitivity of the receiver due to artificial or natural radio frequency interference (RFI). Desensitization occurs when a very strong interference signal overloads the receiver, making it more difficult to detect weaker signals. The receiver's desensitization characteristics determine its ability to operate normally under strong interference such as jammers.

  Noise figure is another major measure of receiver characteristics. The noise figure degrades at each successive stage in the receive path. That is to increase. Amplification or attenuation techniques can be applied in the receiver to obtain an acceptable noise figure. Noise along with distortion determines the ratio expressed in decibels that characterizes the receiver in the presence of noise, signal-to-noise and distortion (SINAD).

  Distortion is the presence of unwanted signals at the output of the device in the receiver RF frequency path. The distortion may include harmonic distortion, intermodulation distortion, and intermodulation distortion. Harmonic distortion occurs when the required input signal is large enough to compress the receiver, and is typically a function of the frequency offset from the required signal at the baseband output. And as a function of the required signal power. Crossover distortion occurs when an amplitude-modulated component from a transmitter (eg, a CDMA radiotelephone) is moved to another carrier (jamming wave) at the device output (LNA output). The most common form of distortion is intermodulation distortion (IMD).

Intermodulation distortion is the result of two or more signals mixed together to produce additional unwanted distortion within the signal band. For two inputs, the intermodulation products occur at sums and differences of integer multiples of the original frequency. That is, for two input signals having frequencies f ₁ and f ₂ , the output frequency component can be denoted as mf ₁ ± nf ₂ . Here, m and n are integers equal to or greater than 1. The order of the intermodulation product is the sum of m and n. The “two-tone” third order components (2f ₁ -f ₂ and 2f ₂ -f ₁ ) can be generated at frequencies that are close to the required or interfering signals and are therefore easily filtered It is impossible. Higher order intermodulation products have smaller amplitudes, which alone makes them less problematic. Second order intermodulation jamming products can be generated at the baseband frequency if the tone spacing is within one-half of the signal bandwidth.

  FIG. 1 is a graph plotting the levels of the basic, second-order, and third-order IMD components against the input level. The theoretical points at which the secondary and tertiary levels block the base are known as the secondary cutoff point (IP2 or SOI) and the tertiary cutoff point (IP3 or TOI). The receiver IIP2 is the input level secondary cutoff point. IIP3 is an input level tertiary cutoff point.

  The third cutoff point and the receiver noise figure are directly related to the dynamic range of the receiver. The dynamic range defines the range of signals that the receiver can handle within the specified function of the receiver. That is the extent to which the receiver can produce an accurate output with an acceptable SINAD. In particular, for a baseband receiver such as an analog-to-digital converter, the dynamic range may be shown as a spurious free dynamic range (SFDR) that extends from the device noise floor to the maximum signal before clipping occurs.

  Local oscillator (LO) leakage occurs when the LO signal leaks to the receiver input. Such leakage may be transmitted as spurious radiation by the transceiver antenna, and it may interfere with other devices. Furthermore, LO leakage may be reflected back to the receiver itself and may desensitize the receiver if it is not removed prior to demodulation.

  Interference radio leakage occurs when an interfering radio wave signal leaks to the LO input or output of a device in the receiver. Such leakage may mix with the jamming signal to generate an unwanted signal, such as a direct current (DC) signal level proportional to the amplitude modulation (AM) component of the jamming signal. The AM interfering radio wave signal can be located at any frequency within the reception frequency band.

  Low frequency flicker (1 / f) noise is caused by defects in the emitter base junction of bipolar junction transistors. Although typically small, flicker noise and other such noise may need to be removed in the receiver in order to preserve signal integrity in baseband.

  Isolation is the ratio (dB) of the power level applied to one terminal pair of the device to the resulting power level with the same frequency appearing in the other terminal pair. Reverse isolation, which is the opposite of isolation, is a performance factor for the receiver components. Reverse isolation is a measure of how much energy injected into the output terminal pair returns to the input source. In order to obtain low LO and jamming leakage, high reverse isolation is required.

  The 1 dB compression point of the amplifier is a measure of the output power level when the amplifier gain is 1 dB lower than the small signal gain. The saturation point of the amplifier is a measure of the maximum output power capability of the amplifier. These performance factors are shown in FIG.

  The performance factors and signal phenomena described above should be considered when designing a wireless communication device. More generally, the wireless communications landscape has been dominated by code division multiple access (CDMA), spread spectrum formats, or broadband, communications in which wireless signals are spread over a very wide bandwidth. CDMA technology has become the basis for many modulation standards such as CDMA (IS-95 and CDMA200) and WCDMA (IMT2000). Each of these modulation or air interface standards is in many radio frequency bands including cellular (Japanese and US cellular), PCS (personal communications systems in the US and Korea bands), and IMT (International Telecommunication Union). It has an influence. Other modulation standards are FM (Frequency Modulation, IS-19), GSM (Global System for Mobile Communications), US-TDMA (IS-136), GPS (Global Positioning System), Wireless LAN (802. 11), and Bluetooth (Bluetooth®).

  Frequency bands have been assigned to various communication modes. For wireless transceivers, the US PCS receive (RX) frequency band is 1930-1990 MHz and the combined transmit (TX) frequency band is 1850-1910 MHz. The US cellular reception frequency band is 869-894 MHz and the combined transmission frequency band is 824-849 MHz. Similarly, reception and transmission frequency bands are allocated to Japanese cellular, IMT, and Korean PCS.

  The communication standard indicates the standard that the wireless communication device must satisfy. For example, spurious emissions, sensitivity, jamming (two tone intermodulation and single tone desensitization), and vestigial sideband standards must be met.

  Wireless communication has not yet been standardized on an international or even domestic basis. Current technology has recognized that transceivers capable of operating in more than one band or more than one mode have increased portability. In particular, a dual band handset operates on two frequency bands. For example, a two-band CDMA handset can operate in both the 800 MHz (US Cellular) and 1.9 GHz (US PCS) frequency bands. If a base station operating on these two frequencies uses the CDMA standard, then a mobile unit with a two-band CDMA handset can be served by either or both of these base stations. Is possible. In addition, a two-mode CDMA / FM handset can operate in both CDMA and FM modes. Nevertheless, given the current multiplicity of modulation standards and combined frequency bands, two-mode and two-band telephones offer at most limited compatibility for global communication systems. To provide.

  FIG. 2 is a high level block diagram for a conventional dual downconversion receiver. The receiver 101 adopts a superheterodyne architecture. In particular, the received RF signal 11 is transmitted along the RF signal path and preprocessed (stage 1). The preprocessed RF signal 13 is first converted or downconverted to a signal 15 having an intermediate frequency (IF) (stage 2). The IF signal 15 is then downconverted again to a baseband signal 17 containing “in-phase” (I) and “quadrature” (Q) phase components (stage 3). The I and Q baseband signal components are 90 degrees out of phase. The I and Q components are then sent to other parts of the receiver 101, such as a baseband processor, for further processing (stage 4). Similarly, in a dual upconversion transmitter, the analog I and Q baseband signals are first upconverted to an IF signal and the IF signal is upconverted to the RF signal transmitted there.

  FIG. 3 shows the receiver 101 in more detail. Receiver 101 has several inherent advantages. For example, the design provides superior sensitivity and selectivity, expanded signal dynamic range, flexible frequency planning, and lower dynamic range and current consumption for elements in receiver 101 after IF filter 70. provide. Furthermore, phase and amplitude matching between the I and Q channels 106, 107 can be more easily achieved because the IF signal is in a lower frequency band. Because of these advantages, the receiver 101 is suitable for multi-mode and multi-band applications. Here, the received RF signal—modulated in multiple modes and transmitted in multiple frequency bands—can be processed.

  In order to support multiple bands and modes of operation, receiver 101 must include some mode-specific components. For example, in a multi-band receiver, typically an individual RF signal path is required for each frequency band. In multi-mode receivers, individual baseband paths may be required for each mode according to jamming dynamic range requirements.

  In conventional receivers such as receiver 101, the IF signal path typically includes an amplifier, filtering, and automatic gain adjustment (AGC) circuitry. Thus, the receiver 101 can remove noise and jamming waves other than the signal band, and can compensate for the changing signal power and receiver gain change. In a multimode receiver, IF signal filtering is mode specific. As a result, the receiver 101 has one IF filter 70 for each mode. For example, the receiver of a two-mode telephone includes two IF SAWs (surface acoustic wave filters). For receivers supporting CDMA 1X, CDMA 3X, WCDMA, GSM, FM, Bluetooth, and GPS modes, 4-6 SAWs and 1 individual LC filter are required in the IF signal path May be.

  The need for an IF filter for each mode is a significant drawback of the receiver 101. Each IF filter increases the cost of the receiver, the number of critical components, and the board area of the receiver. Since each IF filter may have high losses, an IF preamplifier or AGC may also be required. An IF voltage controlled oscillator (VCO) and phase locked loop (PLL) 65 are also required to generate a local oscillator (LO) frequency that is input to the IF mixer 60. Additional disadvantages of receiver 101 include the need for a switch matrix or multiple IF amplifiers and AGC modules, and a low loss RF bandpass filter (BPF) to reduce unwanted sideband noise. And requiring an additional IF mixer. Thus, the IF stage of a dual downconversion receiver increases cost, design complexity, and circuit board area of such a receiver.

  FIG. 4 is a block diagram of a direct down-conversion, or zero IF receiver 200. In a direct downconversion receiver, the received RF signal 201 is directly downconverted to a baseband signal 225. Similarly, in direct upconversion, ie zero IF transmitters, the baseband signal is directly upconverted to the transmitted RF signal. In the receiver 200, the received RF signal is mixed with a local oscillator (LO) frequency to generate a baseband signal. In order not to incorporate the IF signal path, the receiver 200 includes cost, board area, and IF components including IF SAW, LC matching and individual filters, preamplifier, AGC, IF mixer, and IF VCO and PLL. Combined with reduce power consumption. In addition, there is less part-to-part and temperature variation.

  The receiver 200 design takes into account that further signal processing, such as channel selectivity filtering, occurs in the baseband analog or digital domain by the integrated circuit. Thus, the RF and analog portions of receiver 200 can be more general in nature. Since AGC is digital, it may require simplified calibration or even no calibration. For constant operating modes such as GPS, Bluetooth, and GSM, the receiver 200 can reduce the cross-modulation in CDMA cellular and PCS modes because the primary purpose of this filter is to reduce RF filtering. It is possible not to need. However, the GPS mode may require an RF filter if a GPS modulated signal is received simultaneously with other modulated signals.

  Despite the advantages described above, direct downconversion has not been widely incorporated into wireless telephones. The reason is that it is very difficult to achieve core receiver design goals while at the same time achieving the inherent dynamic range for the receiver. Design goals for a receiver such as receiver 200 include achieving high gain and low noise figure, high IIP3 and IIP2 values, and low power consumption. Multi-mode and multi-band receivers may require a very wide dynamic range. Therefore, achieving these design goals for these receivers is even more difficult.

  More specifically, local oscillator (LO) leakage and jammer leakage to the I and Q mixer LO terminal pairs cause significant problems for direct downconversion receivers. For cellular and PCS, spurious emission requirements are particularly stringent. Thus, higher reverse isolation is required. Furthermore, in direct downconversion receivers, LO leakage reflected back to the receiver itself, as well as jamming leakage to the LO terminal pair of the I and Q mixers, may be handled by the direct downconversion circuit. Absent. Thus, an unnecessary DC offset voltage, which may also contain baseband spectral components, may appear at the mixer output in addition to the required baseband signal. Therefore, the DC offset must be removed to ensure that the signal to noise ratio is high enough.

  In CDMA, sensitivity is tested with a signal set at a level such that a constant frame error rate (FER) is satisfied. IS-98 specifies that the device under test must be below 0.5% FER and satisfy a sensitivity level of -104 dBm (signal power). The intermodulation test is performed at a signal level set to -101 dBm (3 dB above the sensitivity test), with two tones of offset related to the RF signal (-43 dBm / tone at the offset producing the in-band distortion product, or typically Are ± 900 and ± 1700 kHz) and below 1% FER. Depending on the frequency band, the power level being tested and the frequency offset for jamming may vary. For a single tone desensitization test, the jammer level at the RF terminal pair of the I and Q mixers is 71 dB greater than the signal level at an offset equal to or greater than 900 kHz.

  The jamming power may leak into each mixer LO terminal pair and mix with the jamming RF level at the mixer RF terminal pair to produce a DC level proportional to the RF jamming amplitude. Typically, jamming is generated by the base station forward link in a competing wireless system. The power of the jammer may vary as a function of the modulation used or fading. The worst jammer may have an amplitude modulation equivalent to the required signal bandwidth. Thus, the AM component is at the top of any signal energy in the baseband after downconversion and cannot be removed by baseband filtering. This problem is exacerbated as the disturbing RF signal increases. For example, if the interfering RF signal increases by 10 dB, the baseband distortion increases by 20 dB. This baseband distortion is one-to-two if both the RF on the RF mixer LO isolation, which affects the self-mixing of jamming radio waves, and the RF mixer IIP2 representing the second order distortion effect are inadequate. It can actually be larger than the two-for-one) slope.

  Furthermore, in the direct down-conversion receiver, the jamming wave and LO leakage requirements for the mixer are very burdensome. Since such a receiver lacks IF filtering, the dynamic range of the receiver baseband element is increased by 30 dB or more due to baseband analog filtering and temperature variations of each part, frequency, and gain. You may need that. Residual sideband standards for various modulation standards must also be satisfied. Since such a receiver has less gain before its baseband stage, flicker noise in the baseband has a greater impact on the receiver's ability to process FM modulated signals. .

  As a result, what is needed is a direct conversion transceiver capable of modulating an RF signal in multiple bands and multiple modes.

[Overview]
The disclosed embodiments illustrate a new and improved system and method for generating a local oscillator (LO) frequency in a direct conversion wireless communication device. In one embodiment, the system incorporates a voltage controlled oscillator (VCO), a divider, and a mixer. The frequency divider has an input and an output generated by dividing the input signal. The divider input is operatively coupled to the VCO. The mixer has a first mixer input operatively coupled to the VCO, a second mixer input operatively coupled to the divider output, and an output. The mixer output provides the LO frequency in parallel to the phase shifter and the second divider.

  In other embodiments, the system incorporates a VCO, a first divider, a second divider, and a mixer. The first divider has an input and an output generated by dividing the input signal. The input of the first divider is operatively coupled to the VCO. The second divider has an input and an output generated by dividing the input signal. The input of the second divider is operatively coupled to the output of the first divider. The mixer includes a first mixer input operatively coupled to the output of the first divider, a second mixer input operatively coupled to the output of the second divider, and an output. have.

  In other embodiments, the system incorporates an LO oscillator, a frequency band selection mechanism, and an array selection mechanism. The LO oscillator has one or more arrangements and includes a mixer arranged to mix the VCO frequency with a divided-down version of the VCO frequency. Each array is combined with the frequency band of the RF signal and produces an output signal whose frequency is combined with the frequency band of the RF signal. The frequency band selection mechanism is arranged to select the frequency band of the RF signal. The array selection mechanism is arranged to select an array combined with a selected frequency band of the RF signal.

[Detailed description]
The features, objects, and advantages of the disclosed embodiments will become more apparent from the detailed description set forth below when taken in conjunction with the drawings. In the drawings, like reference numerals designate the same parts throughout.

  FIG. 4 is a high-level block diagram for a direct down-conversion receiver 200 according to one embodiment of the invention. Receiver 200 includes an RF signal path 210, a direct downconverter 220, and a baseband processor 230.

  The RF signal path 210 receives the RF signal 201. The RF signal 201 can include signals that are modulated into multiple modes and transmitted in multiple frequency bands. The RF signal path 210 can include a selection mechanism for selecting among different modes and different bands. In addition, the RF signal path 210 may include an amplifier or a filter to prepare the RF signal 201 for further processing. Such a prepared signal is shown as preprocessed RF signal 215 in FIG. Direct downconverter 220 receives preprocessed RF signals 215 from RF signal path 210 and downconverts these signals to baseband signals 225.

  Baseband processor 230 may include, for example, DC cancellation, matched and jammer filtering, sample decimation, automatic gain control, signal power measurement (received signal strength indicator, RSSI), despreading, deinterlacing Subsequent processing on the baseband signal 225 can be performed, such as leaving, error correction, and decoding into a digital data or audio stream. The processed information can then be routed to a suitable destination, such as a display, a loudspeaker, or an output mechanism in a wireless device that may include a data terminal pair. It should be noted that the baseband processor 230 can also be used by a transmitter that is complementary to the receiver 200.

  FIG. 5 shows the receiver 200 in more detail. Antenna 301 serves as an interface of receiver 200 for incoming RF signals. The antenna 301 can also broadcast an RF signal from a transmitter coupled to the antenna 301. Multiple antennas may be used to separate active bands or to isolate active modes from each other simultaneously. Interface 305 is capable of isolating the received RF signal from the transmitted RF signal so that both receiver 200 and transmitter can use antenna 301.

  The interface 305 can include one or more duplexers 312. The duplexer 312 filters the signal in the incoming reception band. Further, the duplexer 312 separates the incoming reception band signal from the outgoing transmission band signal. Multiple duplexers 312 can be used if multiple operating bands are required in individual receiver or transmitter applications. Assuming that the combined operating bands all fit within the duplexer 312 band, as shown in FIG. 5, one duplexer 312 is a signal modulated into CDMA, FM, and IMT modes. Can be processed.

  The interface 305 can also include one or more switches 314 and bandpass filters 316. Switch 314 selects between receive and transmit operations. For example, the switch 314 can support GSM or Bluetooth mode where signals are received and not transmitted simultaneously. Bandpass filter 316 filters GPS signals in the incoming reception band. Since GPS signals are received and not transmitted, the duplexer need not be used. Other band pass filters 316 may be included in receiver 200 for other similar receive only modes.

  A low noise amplifier (LNA) 320 is coupled to the interface 305 and amplifies the received RF signal. The LNA 320 can be selected to provide a minimum noise figure in the receive band, but high enough gain to minimize the noise figure contribution from the next stage in the receiver 200. . The gain of the LNA 320 can be controlled by the LNA gain control 324. The transmission power may leak from the interface 305 to the receiver 200. For example, the duplexer 312 may not be able to completely filter the transmit power. Thus, LNA 320 may require high compression and a third order breakpoint.

  LNA 320 is coupled to RX bandpass filter (BPF) 330. The BPF 330 further removes transmitter signals that are outside the receive band. It should be noted that BPF 330 may not be required in some embodiments of the present invention. For example, as pointed out above, a signal modulated in GSM mode may not be received and transmitted at the same time if the maximum data rate in GPRS (General Packet Radio Service) is not supported.

  FIG. 5 shows one RF signal path including one duplexer 312, one LNA 320, and one BPF 330. However, the receiver 200 can include multiple signal paths. Each signal path may correspond to one or more individual operating frequency bands of the receiver 200. For example, receiver 200 can include respective cellular, PCS, IMT, and GSM signal paths. Each RF path can include duplexers, switches, and / or bandpass filters, LNAs, BPFs, and I and Q mixers as needed. In addition, simultaneous GPS reception while operating in other modes requires separate LO oscillation, baseband amplification, analog low-pass filter, analog-to-digital converter, I / Q digital processing, and demodulation. Might do.

  The selection mechanism 310 switches between different RF signal paths according to the operating frequency band in operation at a given time. The selection mechanism 310 can include, for example, band selection devices connected to various duplexers and BPFs. The selection mechanism 310 can also be coupled to the I and Q channel mixers 340A, 340B. For example, for signals received in the US cellular band, selection mechanism 310 can switch to duplexer 312, LNA 320, and BPF 330 that appropriately filter and amplify the signals received together.

  The output of BPF 330 is coupled to the inputs of I and Q channel mixers 340A, 340B. In the exemplary embodiment, BPF 330 may have a differential output (not shown) for connection to a differential input (not shown) of mixers 340A, 340B. Thus, the positive and negative output terminals of BPF 330 can be coupled to the positive and negative input terminals of mixer 340A and to the positive and negative input terminals of mixer 340B. Such a differential signal path arrangement reduces LO and TX coupling to the RF signal path and increases common mode rejection of amplitude modulated jammers (higher secondary input cutoff level at the mixer input). . Thus, isolation and jamming rejection at the receiver 200 is improved.

  Alternatively, a transformer can be coupled to the single-ended output of BPF 330. The transformer can convert the single-ended signal into a differential signal that can be coupled to the differential inputs of the mixers 340A, 340B.

  As shown in FIG. 5, a local oscillator (LO) 350 is coupled to buffer amplifiers 351A, 351B. Buffer amplifiers 351A and 351B are coupled to second input 342A of mixer 340A and to second input 342B of mixer 340B, respectively. Buffer amplifiers 351A, 351B can have differential outputs if I and Q mixers 340A, 340B have differential inputs. In some embodiments, the buffer amplifier need not be included in the receiver 200 design.

  The LO 350 may include a frequency oscillator capable of generating output signals at various frequencies. For example, LO 350 may output a first signal and a second signal that is 90 degrees phase shifted from the first signal. The LO 350 may include a phase locked loop (PLL), a voltage controlled oscillator (VCO), a frequency mixing mechanism, and a phase shifting mechanism. The LO 350 may include a band selection 354 that controls the LO 350 according to the operating frequency of the received RF signal. In an exemplary embodiment, LO 350 uses a differential path to mitigate LO leakage and noise coupling into and out of the signal path at the I and Q mixer RF terminal pairs.

  Each mixer 340A, 340B mixes the RF signal received from the BPF 330 with the signal received from the LO 350 at the second input 342A, 342B of the mixer 340A, 340B. The mixing process multiplies the signals together. Thus, mixers 340A, 340B directly downconvert the received RF signal to I and Q baseband signals. In the exemplary embodiment, mixers 340A, 340B have associated gains that can be adjusted by mixer gain controls 341A, 341B.

  After downconversion, the I and Q signals are processed along their respective signal paths 365A, 365B. I signal path 365A is a sample of both signal paths and may include an amplifier 360A, an anti-aliasing filter 370A, and an I-channel analog to digital converter (ADC) 380A. Amplifier 360A is coupled to the output of mixer 340A. After processing and analog-to-digital conversion along each signal path, the digital I channel data 382 and Q channel data 385 may be further processed. In some embodiments, the I and Q signals can be processed along a path that is specific to the mode of operation. In other embodiments, the I and Q signal paths can be shared among the modes.

  The receiver 200 can include Bluetooth specific modules. As shown in FIG. 5, Bluetooth direct downconverter 390 and Bluetooth baseband processor 395 are functionally and structurally similar to the structures described above. However, since Bluetooth can operate simultaneously with other modes of operation, such as CDMA, Bluetooth direct downconverter 390 and baseband processor 395 may be implemented as Bluetooth-only modules. Similarly, GPS may operate simultaneously and may require separate baseband signal paths and LO oscillator circuitry.

  FIG. 6 illustrates a system 400 for generating a local oscillator frequency in accordance with an embodiment of the present invention. It should be noted that the system 400 can be incorporated into a wireless receiver, transmitter, or transceiver. For example, system 400 can be incorporated into receiver 200 as LO 350 in FIG. The system 400 includes a phase locked loop (PLL) 410, a loop filter 401, a mixer 450, a voltage controlled oscillator (VCO) 420, and a switch 440.

  The switch 440 may be arranged to have multiple positions. In FIG. 6, the switch 440 is a crosspoint switch having three positions. In the first position (1-2), referred to herein as “feed forward”, switch 440 couples VCO 420 to the input of divider 430. In the second position (2-3) “feedback”, switch 440 couples the output of mixer 450 to the input of divider 430. In the third position (1-3) “bypass”, the switch 440 couples the VCO 420 to the output of the mixer 450 and the output of the mixer 450 is disabled. Although system 400 is shown including a switch, in other embodiments, system 400 need not include a switch. For example, VCO 420 can be coupled directly to divider 430. The position of the switch 440 can be controlled by a control mechanism (not shown) such as band selection according to the frequency band of the received RF signal.

  VCO 420 may include a single-ended output VCO that is external to the chip that includes the combined receiver, transmitter, or transceiver. An external VCO can have better phase noise than a VCO integrated in an ASIC (application specific integrated circuit). However, an integrated VCO may be sufficient depending on the interference requirements specific to a given operating band. For an external VCO 420, the PLL 410 can be coupled directly to the VCO 420. Further, the PLL 410 can be coupled to the output of the mixer 450 if the PLL 410 is integrated into the system 400. The PLL 410 receives signals at a reference frequency 405 to create individual channel spacing within each active band.

  The embodiment shown in FIG. 6 includes a PLL input switch 445. Switch 445 can couple PLL 410 to VCO 420, to the output of mixer 450, or to the output of divider 430. As is well known in the art, PLL 410, loop filter 401, and VCO 420 cooperate to output a signal having a VCO frequency. The VCO frequency can be above or below the frequency of the received or transmitted signal. The frequency divider 430 may include a frequency divider that outputs a signal whose frequency is a divided form of the input signal. For example, divider 430 can divide by an integer N, where the value of N can be set by a control signal.

  VCO 420 is coupled to a first input of mixer 450. As stated above, based on the position of switch 440, the second input of mixer 450 is routed through divider 430 to VCO 420 (feed forward), mixer 450 output (feedback), or open. It can be coupled to a circuit (bypass). The mixer 450 can include a single sideband (SSB) mixer that outputs only one primary mixer product, or an image removal mixer. The SSB mixer minimizes unnecessary mixer products at the mixer output. In particular, the SSB mixer provides a frequency output that is either the sum of two input frequencies (upper sideband or USB) or the difference between the two input frequencies (lower sideband or LSB). The upper SSB mixer preserves the upper sideband and invalidates the lower SSB. Conversely, the lower SSB mixer preserves the lower sideband and invalidates the upper sideband. Mixer 450 can be arranged to operate between USB and LSB modes by a control signal coupled to mixer 450.

  System 400 can also include a second divider 470 to create a quadrature LO signal 490. The second divider 470 can divide the input frequency by an integer M and may include a flip-flop. When divider 470 includes two flip-flops, the first flip-flop records the rising edge of the input signal, while the second flip-flop records the falling edge. The individual outputs of the flip-flops can have a phase difference of 90 degrees. As described above, each flip-flop can differentially drive any one of the I and Q mixers 340A and 340B. In other embodiments, the buffer amplifiers 351A, 351B may be located between the second divider 470 and the I and Q mixers 340A, 340B. If M = 2, then the second divider 470 divides by 2, but the second divider 470 has an effect over a wide range of frequencies when used with the divider 430. It functions as a broadband phase shifter. The second divider 470 can generate I and Q mixer LO signals for the US and Japanese cellular bands.

  A phase shifter 460 can be included in the system 400 in parallel with the second divider 470. Instead, system 400 may include only phase shifter 460. A phase shifter 460, which may include an LCR network or active element, may be coupled to the output of the mixer 450. Phase shifter 460 may receive the input signal and generate a quadrature LO output signal 480. In the case of a receiver, each quadrature signal may be mixed with the received RF signal to downconvert the RF signal to I and Q baseband components. In an exemplary embodiment, phase shifter 460 is effective for higher operating bands of PCS (US or Korea) and IMT.

  In accordance with an embodiment of the invention, the value of N for divider 430, the position of switch 440, and the mode of mixer 450 can be varied to generate a wide range of LO frequencies. In addition, the value of M for the second divider 470 can be varied. Despite being able to generate a wide range of LO frequencies, the VCO 420 need only operate within a relatively narrow tuning range. Thus, system 400 can be implemented in wireless multi-band and multi-mode receivers, transmitters, or transceivers.

  In the exemplary embodiment, system 400 includes a differential signal path. For example, the output of the VCO 420 and the inputs and outputs of the mixer 450 and the divider 430 can be differential. The radiated I and Q LO energy and conducted coupling to the RF signal path in the wireless device incorporating the system 400 can be minimized.

  In a wireless device that includes the system 400, a micro processing unit (not shown) determines the frequency band applicable to the RF signal. Based on the selected band, an array selection mechanism, such as band selection 354 in FIG. 5, can select an array in system 400 that is combined with the selected frequency band. Thus, appropriate control signals for setting the value of N for divider 430, the position of switch 440, the mode of mixer 450, and the value of M for second divider 470 are within system 400. Can be generated.

Table 1 shows an exemplary arrangement for the system 400 when executed in a receiver environment. VCO 420 is controlled to operate at approximately 1600 to 1788 MHz. VCO 420 may be the first source of radiated and conducted noise in a wireless device. As Table 1 shows, the VCO frequency band is distinct from the combined RF receive frequency band. Thus, the following arrangement minimizes the effects of VCO noise in the wireless device.

Table 1 LO Control Arrangement for Multi-Band Direct Downconversion Receivers Consistent with the present invention, other arrangements can be used for different reception frequency bands and for different design techniques such as external VCO or integrated VCO implementation. In contrast, the tuning range and center frequency of the VCO 420 can be prepared to optimize. Additional frequency dividers may be included in system 400 to provide such an arrangement.

  Divider 430 and mixer 450 may generate unnecessary LO spurs that are outside the required receive bandwidth. However, the output of the mixer 450 will compress such a branch. Further, the terminal pairs 342A, 342B (see FIG. 5) of the I and Q mixers 340A, 340B may also include resonators that compress such branches. The RF signal path may also have multiple RF BPF responses that can remove jamming products at the same frequency as the LO branch.

  As discussed, second divider 470 creates quadrature LO signal 480. I and Q mixers 340A, 340B receive a quadrature LO signal 480 that can be passed by buffers 351A, 351B as inputs. Thus, phase variations within the load resistors and capacitors of I and Q mixers 340A, 340B may be a source of system error. However, the phase matching requirement can be satisfied by implementing the I and Q mixers 340A, 340B on the same chip. Thus, the vestigial sideband standard for the receiver can be satisfied.

  Amplitude matching between the I and Q channels may be required. A typical amplitude matching approach involves calibrating the I and Q channel gains through analog or digital gain compensation. In order to achieve analog gain compensation (not shown), an independent or switchable power detection mechanism is used to measure the received signal strength indicator (RSSI) of the channel and correspondingly gain. Can be coupled to each of the I and Q channels. The ASIC may store calibration values for the I and Q channels. Through a digital bus interface between the ASIC and the power detection mechanism, the calibration value can be examined and the gain can be compensated. To achieve digital gain compensation (not shown), the baseband path may include a digital multiplier after the ADC that multiplies both the I and Q signals. Therefore, the calibration values stored in the ASIC can be examined and the I and Q channel gains can be correspondingly compensated.

  In other embodiments (not shown), a GPS specific signal path can be included in the wireless receiver or transceiver. GPS modulated signals are received at only one frequency. Thus, the receiver need only tune to one GPS frequency. In particular, a GPS specific path can have a PLL and VCO exclusively for GPS signals. A VCO that may be on-chip or off-chip can operate at 3150.84 MHz, or twice the GPS frequency. The GPS VCO can then be coupled to and divided by a divider (divide by 2) to generate the LO frequency for direct downconversion of the GPS RF signal. Although a separate GPS RF signal path may be provided in the receiver, the GPS baseband path can nevertheless be separated or shared with signals modulated according to other modulation standards. . In the case of separation, the baseband processing of the GPS signal can occur simultaneously with the baseband processing of other modulated signals. When shared, savings in current and substrate area can be achieved.

  Since Bluetooth can operate simultaneously with other modes of operation such as CDMA, a separate VCO and LO oscillator can be used to help generate LO frequencies for direct downconversion of Bluetooth signals. May be included in the transceiver.

  FIG. 7 shows an alternative system 500 for generating a local oscillator frequency. System 500 includes PLL 570, loop filter 560, multi-band VCO 501, VCO divider 520, SSB mixer 540, SSB divider 530, and RX divider 550. Multi-band VCO 501, PLL 570, and loop filter 560 cooperate together to output the VCO frequency to different frequency ranges. Band selection 510 determines the appropriate frequency range for multi-band VCO 501.

  VCO divider 520 is coupled to multi-band VCO 501. The VCO divider 520 can divide the VCO frequency by an integer P such as 2. The divided output of VCO divider 520 is coupled to the input of SSB divider 530. The SSB divider 530 can divide the output frequency of the VCO divider 520 by an integer such as 2. The output of SSB divider 530 and the output of VCO divider 520 are coupled to respective inputs of SSB mixer 540. The SSB mixer 540 mixes the signals together. Depending on whether the SSB mixer 540 is operating as a USB mixer or LSB mixer, the sum or difference of the input signals is output by the mixer 540. As a result, VCO divider 520, SSB divider 530, and SSB mixer 540 cooperate to operate as a partial frequency multiplier. The output of mixer 540 is coupled to the input of RX divider 550. The RX divider 550 can divide the input signal by an integer such as 1 or 2.

Similar to the mode of the SSB mixer 540, changing the frequency band of the multi-band VCO 501 and the divider values of the VCO divider 520, SSB divider 530, and RX divider 550 allows a wide range of LO frequencies to be 500 can be generated. Table 2 shows an exemplary arrangement for the system 500 that creates the system 500 suitable for implementation in a multi-band wireless receiver.

Table 2 LO Control Arrangements for Multi-Band Direct Downconversion Receiver Other arrangements are possible in system 500. For example, system 500 can include a multiplier bypass switch 580 coupled to multi-band VCO 501 and RX divider 550. When the switch is closed, the multi-band VCO 501 can run at twice or four times the operating frequency of the received signal. RX divider 550 can then divide the VCO output frequency by 2 or 4, respectively, to generate the required LO frequency. In particular, to generate cellular I and Q mixer LO signals, the VCO 501 can be operated at four times the received frequency, and the RX divider 550 can be divided by four. is there. However, tuning may be more problematic due to the wide operating range of the multi-band VCO 501. System 500 may couple multi-band VCO 501 directly to RX divider 550, and multiplier bypass switch 580, SSB divider 530, SSB mixer 540, and VCO divider 520 are removed from system 500. What may be erased should be evaluated correctly.

  In addition, system 500 may include a switch (not shown) coupled to multi-band VCO 501 and SSB mixer 540 input 545. If the switch is closed, the SSB mixer 540 may mix the VCO frequency with the divided form of the VCO frequency. Thus, system 500 can generate I and Q mixer LO signals in a similar manner as used in system 400 above.

  FIG. 8 illustrates an embodiment of a direct upconversion, or zero IF transmitter 600. Transmitter 600 includes a system 602 that generates a local oscillator frequency. System 602 is similar to system 400 described above, but is specially configured and operates within a wireless direct upconversion transmitter. The system 602 includes a PLL 610, a loop filter 601, first and second SSB mixers 645, 650, a VCO 620, a PLL input switch 641, LO switches 640A and 640B, and a second divider 670.

  The phase noise requirement for the transmitter is less than that for the receiver that must satisfy the jamming requirement. As a result, the VCO 620 can be more easily integrated on the transmitter or transceiver ASIC. However, in other embodiments, the VCO 620 may be implemented off-chip. VCO 620, loop filter 601, PLL 610, and reference oscillator 605 work together to generate a VCO output frequency. The PLL input switch 641 can selectively couple the PLL 610 to the VCO 620, the output of the divider 630, or the output of the first SSB mixer 645. In this way, the input source for PLL 610 can be switched from VCO 620 to either the signal at the output of divider 630 or the signal at the output of first SSB mixer 645. Thus, if the required RF frequency is generated, locking to that frequency may occur.

  Each of the switches 640A and 640B has two positions. Additional locations are possible in other embodiments. In other embodiments, switches 640A and 640B need not be included. In the feed forward position of switch 640A, switch 640A couples VCO 620 to the input of first SSB mixer 645. In the feedback position, switch 640 A couples the output of SSB mixer 645 to the LO terminal pair of SSB mixer 645. In the feed forward position of switch 640B, switch 640B couples VCO 620 to the input of second SSB mixer 650. In the feedback position, switch 640B couples the output of SSB mixer 650 to the LO terminal pair of SSB mixer 650.

  VCO 620 is coupled to the input of divider 630. Divider 630 divides the VCO output frequency by an integer N. The frequency divider 630 generates first and second output signals. The first output of divider 630 is coupled to first SSB mixer 645. The second output of divider 630 is coupled to second SSB mixer 650. The signals at the first and second divider outputs are both divided in the form of the input frequency, but are 90 degrees out of phase.

  When switch 640A is in the feedforward position, the first SSB mixer 645 mixes the VCO output frequency with the divided form output by divider 630. Similarly, the second SSB mixer 650 mixes the VCO output frequency with the divided form output by the divider 630. The outputs of the first and second SSB mixers 645, 650 are identical in frequency and 90 degrees out of phase. The output of the first and second SSB mixers 645, 650 is the transmitter LO frequency of the system 602.

  The output of the second SSB mixer 650 is coupled to the second divider 670. The second frequency divider 670 can divide the input frequency by an integer M. The second frequency divider 670 generates first and second output signals. The first and second output signals are orthogonal. The output of second divider 670 is the transmitter LO frequency of system 602.

By changing the values of N and M, the mode of SSB mixers 645, 650, and the position of switches 640A, 640B, system 602 can generate a wide range of transmitter LO frequencies. Accordingly, the system 602 is suitable for implementation in a direct up-conversion transmitter such as the transmitter 600. Table 3 shows a typical arrangement combined with the transmitter operating band. Additional sequences may be prepared consistent with the teachings of the present invention. As described above, the required frequency band can be selected by a frequency band selection mechanism, and the combined array can be selected by an array selection mechanism.

Table 3 LO Control Arrangement for Multi-Band Direct Upconversion Transmitters In another embodiment (not shown), the system 602 mixes the received LO frequency for the receiver with a fixed offset LO frequency. It is possible to generate a transmit LO frequency. This approach allows the next modulation standard to have a fixed frequency offset between the TX and RX channels, as shown in Table 4.

Table 4 TX offsets related to RX channel frequency In particular, the LO generation circuitry of system 602 (PLL 610, loop filter 601, first and second SSB mixers 645, 650, VCO 620, and switches 640A, 640B) receive It is possible to generate the LO frequency. A second oscillator, which is a fixed offset LO, can be coupled to the respective inputs of the first and second SSB mixers 645,650. Thus, the first SSB mixer 645 and the second SSB mixer 650 can mix the received LO frequency with the offset LO to generate the transmitted LO frequency. However, it should be appreciated that the receive LO may generate spurious outputs. Thus, off-chip filtering in the transmitter or receiver may be required to meet the conducted spurious leakage standard in the receive band. Such filtering can eliminate spur products at the reception frequency.

  The transmitter 600 can use the LO frequency generated by the system 602 to transmit the RF signal. Baseband processor 608 may be external to transmitter 600 as shown in FIG. 8 or may be integrated within a transceiver that includes transmitter 600. Baseband processor 608 provides a pair of output signals. Each output signal can be implemented as a balanced or differential pair. The two outputs represent the I and Q baseband analog signals for each mode, and are provided as separate signal paths so that quadrature modulation of the signals can be performed at a later stage in transmitter 600. It is done.

  In the exemplary embodiment, transmitter 600 includes three RF outputs. Two of the outputs can correspond to a PCS or IMT signal band, and the other can correspond to a cellular band. For the PCS RF output, the first RF mixer 651 is coupled to the SSB mixer 645 and the first baseband output of the baseband processor 608. The first RF mixer 651 upconverts the baseband signal directly to the required RF frequency. The second RF mixer 653 is coupled to the second baseband output of the SSB mixer 650 and baseband processor 608. The second RF mixer 653 upconverts the baseband signal directly to the same RF frequency as at the output of the first RF mixer 651. The outputs of the first and second RF mixers 651, 653 are orthogonal due to the relative phase difference of the LO signal used to upconvert the baseband signal.

  The quadrature RF signal is then coupled to a signal adder 660 that combines the two quadrature signals into a single signal. The input of the signal adder 660 can be balanced corresponding to the balanced outputs from the first and second RF mixers 651, 653, respectively. The output of the signal adder 660 can also be a balanced signal to minimize signal interference from common mode noise sources.

  The output of the signal adder 660 can be coupled to two amplifier chains simultaneously. Both amplifier chains can be arranged to operate in the PCS transmission band. As shown in FIG. 8, the first amplifier chain may include AGC amplifiers 662 and 664. Second amplifier chain 670 may include AGC amplifiers 662 and 666.

  For cellular RF output, the third RF mixer 652 is coupled to the first output of the second divider 670 and the first baseband output of the baseband processor 608. The third RF mixer 652 directly upconverts the baseband signal to the required RF frequency. The fourth RF mixer 654 is coupled to the second output of the second divider 670 and the second baseband output of the baseband processor 608. The fourth RF mixer 654 directly upconverts the baseband signal to the same RF frequency as at the output of the third RF mixer 652. The outputs of the third and fourth RF mixers 652, 654 are orthogonal due to the relative phase difference of the LO signal used to upconvert the baseband signal.

  The quadrature RF signal is then coupled to a signal adder 670 that combines the two quadrature signals into a single signal. The input of the signal adder 670 can be balanced corresponding to the balanced outputs from the third and fourth RF mixers 652, 654, respectively. The output of the signal adder can also be a balanced signal to minimize signal interference from common mode noise sources.

  The output of signal adder 670 may be coupled to a third amplifier chain. The third amplifier chain may be arranged to operate in the cellular transmission band. As shown in FIG. 8, the third amplifier chain may include AGC amplifiers 672 and 674.

  The transmitter 600 can be arranged so that only one amplifier chain operates at any time. Thus, if the transmitter 600 is arranged to transmit in each frequency band, only the amplifier chain that supports that frequency band can operate. A dormant amplifier chain can be powered down by a control circuit (not shown) to conserve power. The three amplifier chains shown in FIG. 8 and other such amplifier chains may also include or be combined with transmit filters, isolators, or diplexers according to methods well known in the art. Is possible.

  The foregoing detailed description refers to the accompanying drawings that illustrate exemplary embodiments of the invention. Other embodiments are possible, and variations to the embodiments can be made without departing from the spirit and scope of the invention. For example, many of the devices described above can be indirectly coupled to each other, such as devices being separated by an intermediate device such as a filter or amplifier. Further, the teachings of the present invention can be applied to modulation standards and operating bands that will evolve in the future. As a result, this detailed description is not meant to limit the invention, but rather the scope of the invention is defined by the appended claims.

FIG. 6 is a graph plotting saturation and compression points, and second and third cutoff points. It is a high-level block diagram regarding the conventional dual conversion receiver. It is a block diagram regarding the conventional dual conversion receiver. FIG. 3 is a high level block diagram for a direct conversion receiver. It is a block diagram regarding a direct conversion receiver. 1 is a block diagram for a system for generating a local oscillator frequency in accordance with an embodiment of the present invention. FIG. 1 is a block diagram of a system for generating a local oscillator frequency in accordance with an embodiment of the present invention. Fig. 4 shows an embodiment of a zero IF transmitter.

Claims (46)

In multiband direct conversion wireless communication device, a method for generating a local oscillator (LO) frequency, said method comprises
Receiving a signal having a VCO frequency within a first region of frequency from a voltage controlled oscillator (VCO);
Selectively coupling the signal at the VCO frequency as an output signal at the LO frequency in a first array and selectively coupling the signal at the VCO frequency to a divider in a second array;
When the sequence of the second, in order to generate a signal having a divided frequency, divides the VCO frequency by a programmable number N, it noted the programmable number N partially Varied based on the control signal, and
When the sequence of the second, in order to produce the output signal having a second of said LO frequency within the range of frequencies that are determined by the first region and the programmable value N of the frequency Mixing the signal having the VCO frequency with the signal having the divided frequency . In multiband direct conversion wireless communication device, a method for generating a local oscillator (LO) frequency, said method comprises
Receiving a signal having a VCO frequency from a voltage controlled oscillator (VCO);
Selectively coupling the signal having the VCO frequency to an LO output;
The VCO frequency by numerical N to generate a signal having a divided frequency divides,
To produce a second signal having a further divided frequency divides the dividing frequency by numerical M, and,
To produce an output signal having the LO frequency in the LO output, the signal having a divided frequency, said has the further divided frequency second Mixed with the signal of the
A method involving that. In multiband direct conversion wireless communication device, a method for generating a local oscillator (LO) frequency, said method comprises
Determine the frequency band of the RF signal,
Based on the determined frequency band, the LO generator is controlled for one of the plurality of arrays , each array being associated with at least one frequency band of the RF signal , and the frequency of the RF signal generates an output signal associated with at least one frequency band, and a method of the VCO frequency to generate the LO frequency including <br/> that you mixed with the divided form of the VCO frequency . In multiband direct conversion wireless communication device, a system for generating a local oscillator (LO) frequency, the system comprising:
A voltage controlled oscillator (VCO) operating within a first region of frequency and selectively coupled to an oscillator output ;
The partially control signal program that is varied on the basis of the numerical N, input and an input signal divider frequency divider having an output generated by still the divider input is the VCO Selectively coupled to, and
First mixer input, a second mixer input, and a second region within the said frequency determined by a first region of said frequency engaged binding to the divider output that engaged binding to the VCO And a mixer having an output that provides an LO frequency to the oscillator output . In multiband direct conversion wireless communication device, a system for generating a local oscillator (LO) frequency, the system comprising:
A voltage controlled oscillator (VCO) operating within a first region of frequency ;
An input and the first frequency divider having an output generated by the first input signal dividing, still the input of the first divider is engaged sintered and said VCO ,
A second frequency divider having an output generated by the input and a second divider input signal dividing, still the input of the second divider the first division It is engaged the output and formation of vessels, and
First mixer input engaged the output and formation of the first divider, a second mixer input engaged the output and formation of the second frequency divider, and the first region of the frequency look including a mixer which has an output for providing a second said LO frequency within the region of the determined frequency, and
The VCO is selectively coupled to the output of the mixer . In multiband direct conversion wireless communication device, a system for generating a local oscillator (LO) frequency, the system comprising:
And one or LO generator having a more sequences, yet each array generates a LO frequency value for at least one frequency band of RF signals, and the said at least one frequency band of RF signals A mixer generating an LO output signal of an associated frequency and arranged to mix the VCO frequency with a divided form of the VCO frequency; and
An array arranged to select an array based on a determined frequency band of an RF signal by controlling at least the VCO frequency and a divider value to determine the divided form of the VCO frequency A system including a selection mechanism. The method of claim 1, further comprising dividing the LO frequency by a few M. The method of claim 1 including shifting the phase of the output signal. The method of claim 1, wherein the device comprises a receiver. Further comprising mixing the output signal with a signal having an offset frequency indicative of a frequency offset between transmission and reception of the wireless communication device to create an LO frequency for a transmitter in a multi-band direct conversion wireless communication device. Item 9. The method according to Item 9. The method of claim 1, wherein the device comprises a transmitter. 3. The method of claim 2, further comprising dividing the LO frequency by a number P. The method of claim 2, wherein the VCO is a multi-band VCO. The system of claim 4, wherein the VCO is external to a chip containing the device. The system of claim 14, wherein the VCO has a single-ended output. The system of claim 4, wherein the VCO is integrated in a chip containing the device. The system of claim 4, wherein the VCO operates below the frequency of the RF signal. The system of claim 4, wherein the VCO operates above the frequency of the RF signal. The system of claim 4, wherein the VCO operates at a frequency between 1600 and 1788 MHz. The VCO is coupled to a phase locked loop (PLL) and further includes a second PLL and a second VCO for signals received when in GPS mode, the second VCO comprising a received GPS signal The system of claim 4, operating at twice the frequency. 21. The system of claim 20, further comprising a third PLL and a third VCO for signals received when in Bluetooth mode. The system of claim 4, wherein the mixer comprises a single sideband (SSB) mixer. 24. The system of claim 22, wherein the SSB mixer is a low frequency side SSB mixer. 23. The system of claim 22, wherein the SSB mixer is a high frequency side SSB mixer. The system of claim 4, wherein the mixer output is coupled to a phase locked loop (PLL), the PLL being internal to a chip containing the device. The system of claim 4, wherein a switch selectively couples the divider input to the VCO. 27. The system of claim 26, wherein the switch is controlled by switch control based on a band of the received RF signal. The divider input of claim 4, wherein the divider input is selectively coupled to the mixer output using a switch controlled by switch control based on a band of an RF signal received by the multi-band wireless communication device. system. The system of claim 4, wherein the mixer output is selectively coupled to a VCO. 5. The system of claim 4, further comprising a phase shifter having an input coupled to the mixer output, the phase shifter having an output that generates a quadrature signal. 32. The system of claim 30, wherein the phase shifter comprises an active phase shifter. The system of claim 4 including a second divider having an input coupled to the mixer output and an output generated by dividing the input signal. 34. The system of claim 32, wherein the second divider divides by two. 33. The system of claim 32, wherein the second divider outputs a first signal and a second signal, the first signal being 90 degrees away from the phase of the second signal. In-phase (I) mixer, and
Including an orthogonal (Q) mixer,
The first signal drives one of the I and Q mixers in the device;
35. The system of claim 34. The device includes a receiver, where
The band of the received RF signal is a US personal communications system (PCS), and
The VCO operates between frequencies of 1716 MHz and 1769 MHz,
The divider divides by 8 and
The mixer is a high frequency side single sideband (SSB) mixer.
The system of claim 4. The device includes a receiver;
The frequency band of the received RF signal is the International Telecommunication Union (IMT), and
The VCO operates between 1688 MHz and 1736 MHz frequencies;
The divider divides by 4 and
The mixer is a high frequency side single sideband (SSB) mixer.
The system of claim 4. The system of claim 4, wherein the device is included in a wireless communication transceiver. The system of claim 4, wherein the device comprises a transmitter. The band of the transmitted RF signal is a US personal communications system (PCS), and
The VCO operates at a frequency between 1480 MHz and 1528 MHz;
The divider divides by 4 and
The mixer is a high frequency single sideband (SSB) mixer,
40. The system of claim 39. 40. The system of claim 39, further comprising a first amplifier chain arranged to operate within a first transmission frequency band, the amplifier chain being coupled to an upconverter. The device includes a receiver and a transmitter; and
And further comprising an offset LO having an offset frequency indicative of a frequency offset between a transmit and receive signal of a wireless communication device coupled to a third input of the mixer, wherein the mixer output provides the LO frequency for the transmitter The system of claim 4. The system of claim 4, wherein the first mixer input and the mixer output are differential. The system of claim 4, wherein the device includes a receiver incorporating a differential signal path. 6. The system of claim 5, further comprising a third divider coupled to the mixer output. The system of claim 5, wherein the VCO is a multi-band VCO.

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