JP3819184B2 - Control method of permanent magnet type synchronous motor - Google Patents
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Description
【0001】
【発明の属する技術分野】
本発明は、永久磁石形同期電動機の制御方法に係り、特に、回転子位置センサを用いることなく電動機のパラメータの推定を行い、電動機の制御を行う突極形永久磁石形同期電動機の制御方法に関するものである。
【0002】
【従来の技術】
従来、このような技術分野の先行技術文献としては、例えば、以下のようなものが挙げられる。
【0003】
(1)先行技術文献(1):竹下、市川、李、松井:「速度起電力推定に基づくセンサレス突極形ブラシレスDCモータ制御」電学論D,117,p.98(平9−1)
(2)先行技術文献(2):渡辺、竹下、松井:「センサレス突極形ブラシレスDCモータの零速制御」電気学会半導体電力変換研資,SPC−97−7,p37(平9−1)
(3)先行技術文献(3):竹下、市川、松井、山田、水谷:「センサレス突極形ブラシレスDCモータの初期位置角推定法」電学論D、116、p.736(平8−7)。
【0004】
永久磁石形同期電動機(PMSM)の小形、軽量、低価格、信頼性改善のためにPMSMの位置センサレス制御の各種提案がなされている。本発明者らも、既に、中高速度域において速度起電力を利用した方法を、停止時・低速度域において突極機のd−q軸インダクタンス差を利用した診断電圧印加による方法を提案し、全速度領域における位置センサレス駆動法を実現している。
【0005】
【発明が解決しようとする課題】
しかしながら、これらの位置センサレスアルゴリズムは、基本的にはd−q軸モデル上での電圧方程式から導出されているので、PMSMの電気的パラメータを事前に把握する必要がある。
【0006】
本発明は、上記状況に鑑みて、回転子位置センサを用いることなく電動機の駆動装置を用いて電動機のパラメータの推定を行うことにより、小形、軽量、低価格であり、信頼性の改善を図り得る永久磁石形同期電動機の制御方法を提供することを目的とする。
【0007】
【課題を解決するための手段】
本発明は、上記目的を達成するために、
〔1〕回転子位置センサを用いることなく電動機の駆動装置を用いて電動機のパラメータの推定を行い、電動機の制御を行う永久磁石形同期電動機の制御方法において、前記電動機のパラメータはインダクタンスであり、電動機停止時に初期電流零の状態で、インバータの特定のスイッチング素子に短時間Ts だけオン信号を与え、その時に検出される各相の電流Iu (Ts ),Iv (Ts ),Iw (Ts )に基づいてインダクタンスLd ,Lq を計測する永久磁石形同期電動機の制御方法であって、
前記インダクタンスL d ,L q は、以下の方法により得ることを特徴とする。
【0008】
回転子位置角θで停止しているとき、初期電流零で、スイッチング素子u + ,v - ,w - に時間T s だけオン信号を与えたときの検出電流I u1 (T s ),I v1 (T s ),I w1 (T s )から次式のIα u (T s ),Iβ u (T s )を計算し、
Iα u (T s )=√(2/3)・{I u1 (T s )−1/2・I v1 (T s )
−1/2・I w1 (T s )}
Iβ u (T s )=1/√2・{I v1 (T s )−I w1 (T s )}
一方、電動機の電圧方程式より、Iα u (T s ),Iβ u (T s )は次式で表わし、
Iα u (T s )=I 1 +I 2 cos2θ
Iβ u (T s )=I 2 sin2θ
ただし、
I 1 =√(2/3)・(L d +L q )V dc T s /(2L d L q ),
I 2 =√(2/3)・(L q −L d )V dc T s /(2L d L q )である。
【0009】
上式のI 1 ,I 2 が既知となれば、L d ,L q を次式で得る。
【0010】
L d =√(2/3)・V dc T s /(I 1 +I 2 ),
L q =√(2/3)・V dc T s /(I 1 −I 2 )
ただし、
I 1 ,I 2 を求めるために初期電流零でスイッチング素子u - ,v + ,w - に時間T s だけオン信号を与えたときの、検出電流I u2 (T s ),I v2 (T s ),I w2 (T s )及びスイッチング素子u - ,v - ,w + に時間T s だけオン信号を与えたときの検出電流I u3 (T s ),I v3 (T s ),I w3 (T s )から、次式のIα v (T s ),Iβ v (T s ),Iα w (T s ),Iβ w (T s )を計算し、 Iα v (T s )=√(2/3)・{I v2 (T s )−1/2・I w2 (T s )
−1/2・I u2 (T s )}
=I 1 +I 2 cos2(θ−2π/3)
Iβ v (T s )=1/√2{I w2 (T s )−I u2 (T s )}=I 2
sin2(θ−2π/3)
Iα w (T s )=√(2/3){I w3 (T s )−1/2・I u3 (T s )
−1/2・I v3 (T s )}
=I 1 +I 2 cos2(θ+2π/3)
Iβ w (T s )=1/2√{I u3 (T s )−I v3 (T s )}=I 2
sin2(θ+2π/3)
ただし、I 1 ,I 2 は次式で計算する。
【0011】
I 1 ={Iα u (T s )+Iα v (T s )+Iα w (T s )}/3
I 2 =√{2/3(Iβ u (T s ) 2 +Iβ v (T s ) 2
+Iβ w (T s ) 2 }または、
I 2 =√〔2/3{(Iα u (T s )−I 1 ) 2 +(Iα v (T s )
−I 1 ) 2 +(Iα w (T s )−I 1 ) 2 }〕
〔2〕回転子位置センサを用いることなく電動機の駆動装置を用いて電動機のパラメータの推定を行い、電動機の制御を行う永久磁石形同期電動機の制御方法において、前記電動機のパラメータは巻線抵抗であり、3種類のスイッチングパターンの電圧印加の電流応答により、更に2回の電圧印加により、初期位置角推定を行い、この初期位置角に基づき、前記電動機を回転させないようにd軸電流一定値に制御したときの指令電圧、検出電流により巻線抵抗を計測することを特徴とする。
【0012】
〔3〕回転子位置センサを用いることなく電動機の駆動装置を用いて電動機のパラメータの推定を行い、電動機の制御を行う永久磁石形同期電動機の制御方法において、前記電動機のパラメータは起電力係数であり、予めコントローラに記憶されている定格電圧と定格回転数から起電力係数の暫定値KEMを与え、既存のセンサレス制御法で電動機を回転させ、コントローラ内の推定起電力eM から得られる速度ωM が実際の速度ωに一致するように、コントローラ内の起電力係数KEMを調整することにより、起電力係数を計測することを特徴とする。
【0013】
〔4〕上記〔3〕記載の永久磁石形同期電動機の制御方法において、前記起電力係数は、以下の方法により得ることを特徴とする永久磁石形同期電動機の制御方法。
【0014】
コントローラ内で推定された起電力eM は実際の起電力eを正確に推定しているとし(e≒eM )、また、推定起電力eM に基づいた推定速度ωM (=eM /KEM)が得られる。
【0015】
一方、インバータ周波数よりモータの回転速度ωはコントローラ内でわかり、実際の起電力係数KE との間にω=e/KE の関係が得られる。
【0016】
以上より、速度誤差ω−ωM は、
ω−ωM =(1/KE −1/KEM)eM
と得られるので、速度差ω−ωM を用いてコントローラ内の起電力係数KEMを補正し、速度誤差が零に収束したときの起電力係数の暫定値KEMを起電力係数の計測値とする。
【0017】
【発明の実施の形態】
以下、本発明の実施の形態について詳細に説明する。
【0018】
まず、本発明の永久磁石形同期電動機の位置センサレス制御のためのパラメータ推定方法としてのインダクタンス計測について説明する。
【0019】
このインダクタンス計測では、原理的に電流を変化させる必要があり、電動機停止時に初期電流零の状態でインバータの特定のスイッチング素子に短時間Ts だけオン信号を与え、その時に検出される各相の電流Iu (Ts ),Iv (Ts ),Iw (Ts )からインダクタンスLd ,Lq を計測する。
【0020】
具体的方法は以下の通りである。
【0021】
まず、回転子位置角θで停止しているとき、初期電流零で、スイッチング素子u+ ,v- ,w- に時間Ts だけオン信号を与えたときの検出電流Iu1(Ts ),Iv1(Ts ),Iw1(Ts )から、次式のIαu (Ts ),Iβu (Ts )を計算する。
【0022】
Iαu (Ts )=√(2/3)・{Iu1(Ts )−1/2・Iv1(Ts )
−1/2・Iw1(Ts )}
Iβu (Ts )=1/√2{Iv1(Ts )−Iw1(Ts )}
一方、電動機の電圧方程式より、Iαu (Ts ),Iβu (Ts )は、次式で記せる。
【0023】
Iαu (Ts )=I1 +I2 cos2θ
Iβu (Ts )=I2 sin2θ
ただし、
I1 =√(2/3)・(Ld +Lq )VdcTs /(2Ld Lq ),
I2 =√(2/3)・(Lq −Ld )VdcTs /(2Ld Lq )である。
【0024】
上式のI1 ,I2 が既知となれば、Ld ,Lq は次式で得られる。
【0025】
Ld =√(2/3)・VdcTs /(I1 +I2 ),
Lq =√(2/3)・VdcTs /(I1 −I2 )
I1 ,I2 を求めるために、初期電流零でスイッチング素子u- ,v+ ,w- に時間Ts だけオン信号を与えたときの検出電流Iu2(Ts ),Iv2(Ts ),Iw2(Ts )及びスイッチング素子u- ,v- ,w+ に時間Ts だけオン信号を与えたときの検出電流Iu3(Ts ),Iv3(Ts ),Iw3(Ts )から、次式のIαv (Ts ),Iβv (Ts ),Iαw (Ts ),Iβw (Ts )を計算する。
【0026】
Iαv (Ts )=√(2/3)・{Iv2(Ts )−1/2・Iw2(Ts )
−1/2・Iu2(Ts )}
=I1 +I2 cos2(θ−2π/3)
Iβv (Ts )=1/√2・{Iw2(Ts )−Iu2(Ts )}=I2
sin2(θ−2π/3)
Iαw (Ts )=√(2/3)・{Iw3(Ts )−1/2・Iu3(Ts )
−1/2・Iv3(Ts )}
=I1 +I2 cos2(θ+2π/3)
Iβw (Ts )=1/2√{Iu3(Ts )−Iv3(Ts )}=I2
sin2(θ+2π/3)
I1 ,I2 は次式で計算できる。
【0027】
I1 ={Iαu (Ts )+Iαv (Ts )+Iαw (Ts )}/3
I2 =√{2/3(Iβu (Ts )2 +Iβv (Ts )2
+Iβw (Ts )2 }または、
I2 =√〔2/3{(Iαu (Ts )−I1 )2 +(Iαv (Ts )
−I1 )2 +(Iαw (Ts )−I1 )2 }〕
次いで、本発明の永久磁石形同期電動機の位置センサレス制御のための、パラメータ推定方法としての巻線抵抗の計測は、3種類のスイッチングパターンの電圧印加の電流応答により、更に2回の電圧印加により、初期位置角推定を行い、この初期位置角に基づき、前記電動機を回転させないようにd軸電流一定値に制御したときの指令電圧、検出電流により巻線抵抗を計測する。
【0028】
次いで、本発明の永久磁石形同期電動機の位置センサレス制御のための、パラメータ推定方法としての起電力係数の計測について説明する。
【0029】
この起電力係数の計測では、銘板の定格電圧と定格回転数から起電力係数の暫定値KEMを与え、既存のセンサレス制御法で電動機を回転させる。コントローラ内の推定起電力eM から得られる速度ωM が実際の速度ωに一致するように、コントローラ内の起電力係数KEMを調整し、この結果、起電力係数が計測できる。
【0030】
具体的には以下の通りである。
【0031】
コントローラ内で推定された起電力eM は実際の起電力eを正確に推定しているとする(e≒eM )。また、推定起電力eM に基づいた推定速度ωM (=eM /KEM)が得られる。
【0032】
一方、インバータ周波数よりモータの回転速度ωはコントローラ内でわかり、実際の起電力係数KE との間にω=e/KE の関係が得られる。以上より、速度誤差ω−ωM は、
ω−ωM =(1/KE −1/KEM)eM
と得られるので、速度差ω−ωM を用いてコントローラ内の起電力係数KEMを補正し、速度誤差が零に収束したときのKEMが起電力係数の計測値となる。
【0033】
以下、詳細に、本発明のパラメータ推定方法について説明する。
(1)インダクタンスの推定
図1は本発明に係るPMSMの解析モデル図である。
【0034】
この図において、1は逆突極形永久磁石形同期電動機、2はその電動機の逆突磁極 3は巻線、4はトランジスタインバータ、5はスイッチング素子(スイッチングトランジスタ)、6はダイオードである。
【0035】
図中に定義した静止座標αu −βu 軸上のモータ停止時の電圧方程式は次式で表される。
【0036】
【数1】
【0037】
【数2】
【0038】
ただし、vαu ,vβu :αu −βu 軸電機子電圧、iαu ,iβu :αu −βu 軸電機子電流、R:巻線抵抗、Ld ,Lq :d−q軸インダクタンス、p(=d/dt):微分演算子である。
【0039】
モータ停止時に初期電流零で時間Ts だけ電圧ベクトルv(100)〔+,−のスイッチング素子の導通に対してそれぞれ1.0を割り当てu+ ,v- ,w- 導通時のモータ印加電圧をv(100)と表現する〕を印加したときの電流iu (Ts )は、抵抗R≒0の近似のもとで次式で得られる。
【0040】
【数3】
【0041】
【数4】
【0042】
上記(4)式のI1 ,I2 がわかれば、Ld ,Lq は次式で計算できる。
【0043】
【数5】
【0044】
一方、αu −βu 軸に対して、2π/3,4π/3進みにそれぞれαv −βv ,αw −βw 軸を定義し、v(010),v(001)をTs だけ印加したときの電流は次の(6),(7)式で与えられる。
【0045】
【数6】
【0046】
【数7】
【0047】
上記(3),(6),(7)式のα軸からI1 は、
【0048】
【数8】
【0049】
と得られる。また、I2 はβ軸電流および上記(8)式のI1 とα軸電流から2通りで求められ、それぞれI21,I22として次の(9),(10)式で得られる。
【0050】
【数9】
【0051】
【数10】
【0052】
上記(9),(10)式のI21,I22を用いてそれぞれLd ,Lq を計算し、その平均値を推定値にする。また、iu (Ts ),iv (Ts ),iw (Ts )を用いて回転子位置θが推定できる(例えば、前記先行技術文献(3)参照)。
(2)巻線抵抗の推定
モータ停止時(θ=0)のd−q軸上の電圧方程式は次の(11)式で表される
【0053】
【数11】
【0054】
トルクを発生しないようにq軸電流を零とし、d軸に一定電流Id を流したときの指令電圧vd を用いて巻線抵抗Rを次の(12)式により推定できる。
【0055】
【数12】
【0056】
(3)速度起電力係数の推定
モータ銘版記載の定格電圧V0n,定格回転数ωn より速度起電力係数の初期値をKE0=V0n/ωn と与える。この初期値を用いて起電力推定に基づくセンサレス制御(例えば、前記先行技術文献(1)参照)により定格速度で無負荷運転する。このとき、1サンプル周期間Tのモータの位置変化量Δθw は速度起電力eと起電力係数KE を用いて、
【0057】
【数13】
【0058】
と得られる。一方、センサレス制御アルゴリズムよりΔθw は、
【0059】
【数14】
【0060】
と表される。ここで、eM (n)は速度起電力推定値、KEM(n−1)(=KE0)は速度起電力係数推定値、KθΔiγ(n)は位置の補正項で、Kθは位置推定ゲイン、Δiγ(n)は電流推定誤差である。
【0061】
一定速度状態ではe=eM (n)が成立するため、上記(13),(14)式より速度起電力係数誤差1/ΔKE (n−1)は、
【0062】
【数15】
【0063】
と記せる。したがって、図2に示すようにΔiγから1/KEM(n)を起電力係数推定ゲインKK を用いて、
【0064】
【数16】
【0065】
と推定できる。なお、図2において、11は起電力推定器である。
【0066】
【表1】
【0067】
表1の6極1.5kW,1500rpm供試機を用いて実験を行った。
【0068】
図3は、Ld ,Lq の推定結果である。1回につきπ/4づつ異なった8点における推定値をプロットしており、全部で256回の推定結果を示している。
【0069】
このとき、電圧ベクトル印加時間Ts は195μsとした。推定結果の平均値はLd =4.51mH、Lq =9.55mHで表1のモータパラメータとの誤差はそれぞれ4.9%、0.4%で推定できている。
【0070】
また、抵抗Rの推定については上記(12)式で、Id =4,7,10Aを与えたときの抵抗の平均値より求め、R=0.548Ωと推定できた。速度起電力係数KE の推定については、時定数0.44sとしてKK を設計した。
【0071】
図4は速度起電力係数の初期値KE0をKE の1.5倍、1.0倍、0.5倍として与え、定格回転時における1/KEMの推定結果である。
【0072】
図4から明らかなように、設計通りの時定数0.44sの収束特性が得られ、収束値はKEM=0.208V/rad/sとなり、表1のKE との誤差は5.0%で推定できている。
【0073】
本発明によれば、センサレス突極形PMSMのパラメータ計測方法(パラメータ:Ld ,Lq ,R,KE )について提案し、良好な結果が得られた。
【0074】
なお、本発明は上記実施例に限定されるものではなく、本発明の趣旨に基づいて種々の変形が可能であり、これらを本発明の範囲から排除するものではない。
【0075】
【発明の効果】
以上、詳細に説明したように、本発明によれば、回転子位置センサを用いることなく、電動機の駆動装置を用いて電動機のパラメータの推定を行い、電動機の制御を行うことができるので、小形、軽量、低価格であり、信頼性の改善を図ることができる。
【図面の簡単な説明】
【図1】 本発明に係るPMSMの解析モデル図である。
【図2】 速度起電力係数の推定の説明図である。
【図3】 インダクタンス推定結果を示す図である。
【図4】 速度起電力係数の収束特性を示す図である。
【符号の説明】
1 永久磁石形同期電動機
2 電動機の逆突磁極
3 巻線
4 トランジスタインバータ
5 スイッチング素子(スイッチングトランジスタ)
6 ダイオード
11 起電力推定器[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a control method for a permanent magnet type synchronous motor, and more particularly to a control method for a salient pole type permanent magnet type synchronous motor that estimates a motor parameter without using a rotor position sensor and controls the motor. Is.
[0002]
[Prior art]
Conventionally, as prior art documents in such a technical field, for example, the following can be cited.
[0003]
(1) Prior art document (1): Takeshita, Ichikawa, Lee, Matsui: “Sensorless salient pole type brushless DC motor control based on speed electromotive force estimation”, D. 117, p. 98 (flat 9-1)
(2) Prior art document (2): Watanabe, Takeshita, Matsui: “Zero speed control of sensorless salient pole type brushless DC motor” The Institute of Electrical Engineers of Japan, Semiconductor Power Conversion Research Institute, SPC-97-7, p37 (hira 9-1)
(3) Prior art document (3): Takeshita, Ichikawa, Matsui, Yamada, Mizutani: “Initial position angle estimation method of sensorless salient pole type brushless DC motor” Electronics D, 116, p. 736 (flat 8-7).
[0004]
Various proposals for position sensorless control of PMSM have been made in order to improve the small size, light weight, low cost, and reliability of a permanent magnet type synchronous motor (PMSM). The present inventors have already proposed a method using a speed electromotive force in a medium / high speed range, and a method by applying a diagnostic voltage using a dq axis inductance difference of a salient pole machine in a stop / low speed range, The position sensorless driving method in the entire speed range is realized.
[0005]
[Problems to be solved by the invention]
However, since these position sensorless algorithms are basically derived from voltage equations on the dq axis model, it is necessary to grasp the PMSM electrical parameters in advance.
[0006]
In view of the above situation, the present invention estimates the parameters of the electric motor using the electric motor drive device without using the rotor position sensor, and is small, light, and low in price, and improves reliability. It is an object of the present invention to provide a method for controlling a permanent magnet synchronous motor to be obtained.
[0007]
[Means for Solving the Problems]
In order to achieve the above object, the present invention provides
[1] In a control method of a permanent magnet type synchronous motor that estimates a motor parameter using a motor drive device without using a rotor position sensor and controls the motor, the parameter of the motor is an inductance; With the initial current zero when the motor is stopped, an on signal is given to a specific switching element of the inverter for a short time T s , and the currents I u (T s ), I v (T s ) of each phase detected at that time a inductance L d, a control method of L q a to that permanent magnet synchronous motor measured based on the I w (T s),
The inductances L d and L q are obtained by the following method.
[0008]
When the rotor is stopped at the rotor position angle θ, the detected currents I u1 (T s ) and I v1 when the ON current is given to the switching elements u + , v − and w − for the time T s with the initial current zero. The following formulas Iα u (T s ) and Iβ u (T s ) are calculated from (T s ) and I w1 (T s ) ,
Iα u (T s ) = √ (2/3) · {I u1 (T s ) −1 / 2 · I v1 (T s )
−1 / 2 · I w1 (T s )}
Iβ u (T s ) = 1 / √2 · {I v1 (T s ) −I w1 (T s )}
On the other hand, from the voltage equation of the motor, Iα u (T s ) and Iβ u (T s ) are expressed by the following equations:
Iα u (T s ) = I 1 + I 2 cos 2θ
Iβ u (T s ) = I 2 sin 2θ
However,
I 1 = √ (2/3) · (L d + L q ) V dc T s / (2L d L q ),
I 2 = √ (2/3) · (L q −L d ) V dc T s / (2L d L q ).
[0009]
If I 1 and I 2 in the above equations are known, L d and L q are obtained by the following equations.
[0010]
L d = √ (2/3) · V dc T s / (I 1 + I 2 ),
L q = √ (2/3) · V dc T s / (I 1 −I 2 )
However,
Detected currents I u2 (T s ), I v2 (T s ) when an ON signal is given to the switching elements u − , v + , w − for a time T s with zero initial current to obtain I 1 and I 2 ), I w2 (T s) and the switching element u -, v -, detection current I u3 (T s when received only on signal w + two hours T s), I v3 (T s), I w3 ( From T s , Iα v (T s ), Iβ v (T s ), Iα w (T s ), and Iβ w (T s ) are calculated as follows : Iα v (T s ) = √ (2 / 3) ・ {I v2 (T s ) −1/2 ・ I w2 (T s )
−1 / 2 · I u2 (T s )}
= I 1 + I 2 cos2 (θ-2π / 3)
Iβ v (T s ) = 1 / √2 {I w2 (T s ) −I u2 (T s )} = I 2
sin2 (θ-2π / 3)
Iα w (T s ) = √ (2/3) {I w3 (T s ) −1 / 2 · I u3 (T s )
−1 / 2 · I v3 (T s )}
= I 1 + I 2 cos2 (θ + 2π / 3)
Iβ w (T s) = 1 / 2√ {I u3 (T s) -I v3 (T s)} = I 2
sin2 (θ + 2π / 3)
However, I 1 and I 2 are calculated by the following equations.
[0011]
I 1 = {Iα u (T s ) + Iα v (T s ) + Iα w (T s )} / 3
I 2 = √ {2/3 (Iβ u (T s ) 2 + Iβ v (T s ) 2
+ Iβ w (T s ) 2 } or
I 2 = √ [2/3 {(Iα u (T s ) −I 1 ) 2 + (Iα v (T s )
−I 1 ) 2 + (Iα w (T s ) −I 1 ) 2 }]
[ 2 ] In a method for controlling a permanent magnet synchronous motor in which a motor parameter is estimated using a motor drive device without using a rotor position sensor and the motor is controlled, the motor parameter is a winding resistance. Yes, the initial position angle is estimated by applying the voltage response of the three types of switching patterns and by applying the voltage twice, and based on this initial position angle, the d-axis current is set to a constant value so as not to rotate the motor. The winding resistance is measured by a command voltage and a detection current when controlled.
[0012]
[ 3 ] In a control method of a permanent magnet type synchronous motor in which a motor parameter is estimated using a motor drive device without using a rotor position sensor and the motor is controlled, the parameter of the motor is an electromotive force coefficient. There, the speed of advance controller provides a provisional value K EM electromotive force coefficient from the rated voltage stored and the rated rotational speed, rotates the motor in the existing sensorless control method is obtained from the estimated electromotive force e M in the controller omega as M matches the actual speed omega, by adjusting an electromotive force coefficient K EM in the controller, characterized by measuring the electromotive force coefficient.
[0013]
[ 4 ] The method for controlling a permanent magnet synchronous motor according to [ 3 ], wherein the electromotive force coefficient is obtained by the following method.
[0014]
The electromotive force e M estimated in the controller and that accurately estimate the actual electromotive force e (e ≒ e M), also the estimated speed based on the estimated electromotive force e M ω M (= e M / K EM ) is obtained.
[0015]
On the other hand, the rotational speed ω of the motor is known in the controller from the inverter frequency, and the relationship of ω = e / K E is obtained with the actual electromotive force coefficient K E.
[0016]
From the above, the speed error ω-ω M is
ω−ω M = (1 / K E −1 / K EM ) e M
Since the resulting a, using the speed difference omega-omega M corrects the electromotive force coefficient K EM in the controller, the provisional value K EM of the electromotive force coefficient when the speed error has converged to zero measured value of the electromotive force coefficient And
[0017]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described in detail.
[0018]
First, inductance measurement as a parameter estimation method for position sensorless control of the permanent magnet type synchronous motor of the present invention will be described.
[0019]
In this inductance measurement, it is necessary to change the current in principle. When the motor is stopped, the initial current is zero and an ON signal is given to a specific switching element of the inverter for a short time T s , and each phase detected at that time is detected. The inductances L d and L q are measured from the currents I u (T s ), I v (T s ), and I w (T s ).
[0020]
The specific method is as follows.
[0021]
First, when the rotor is stopped at the rotor position angle θ, the detection current I u1 (T s ) when the ON current is given to the switching elements u + , v − , w − for the time T s with the initial current zero. From I v1 (T s ) and I w1 (T s ), Iα u (T s ) and Iβ u (T s ) of the following equations are calculated.
[0022]
Iα u (T s ) = √ (2/3) · {I u1 (T s ) −1 / 2 · I v1 (T s )
−1 / 2 · I w1 (T s )}
Iβ u (T s ) = 1 / √2 {I v1 (T s ) −I w1 (T s )}
On the other hand, from the voltage equation of the motor, Iα u (T s ) and Iβ u (T s ) can be expressed by the following equations.
[0023]
Iα u (T s ) = I 1 + I 2 cos 2θ
Iβ u (T s ) = I 2 sin 2θ
However,
I 1 = √ (2/3) · (L d + L q ) V dc T s / (2L d L q ),
I 2 = √ (2/3) · (L q −L d ) V dc T s / (2L d L q ).
[0024]
If I 1 and I 2 in the above equations are known, L d and L q can be obtained by the following equations.
[0025]
L d = √ (2/3) · V dc T s / (I 1 + I 2 ),
L q = √ (2/3) · V dc T s / (I 1 −I 2 )
In order to obtain I 1 and I 2 , the detected currents I u2 (T s ) and I v2 (T s ) when the ON current is given to the switching elements u − , v + , w − for the time T s with zero initial current. ), I w2 (T s) and the switching element u -, v -, detection current I u3 (T s when received only on signal w + two hours T s), I v3 (T s), I w3 ( From T s , Iα v (T s ), Iβ v (T s ), Iα w (T s ), and Iβ w (T s ) are calculated.
[0026]
Iα v (T s ) = √ (2/3) · {I v2 (T s ) −1 / 2 · I w2 (T s )
−1 / 2 · I u2 (T s )}
= I 1 + I 2 cos2 (θ-2π / 3)
Iβ v (T s ) = 1 / √2 · {I w2 (T s ) −I u2 (T s )} = I 2
sin2 (θ-2π / 3)
Iα w (T s ) = √ (2/3) · {I w3 (T s ) −1 / 2 · I u3 (T s )
−1 / 2 · I v3 (T s )}
= I 1 + I 2 cos2 (θ + 2π / 3)
Iβ w (T s) = 1 / 2√ {I u3 (T s) -I v3 (T s)} = I 2
sin2 (θ + 2π / 3)
I 1 and I 2 can be calculated by the following equations.
[0027]
I 1 = {Iα u (T s ) + Iα v (T s ) + Iα w (T s )} / 3
I 2 = √ {2/3 (Iβ u (T s ) 2 + Iβ v (T s ) 2
+ Iβ w (T s ) 2 } or
I 2 = √ [2/3 {(Iα u (T s ) −I 1 ) 2 + (Iα v (T s )
−I 1 ) 2 + (Iα w (T s ) −I 1 ) 2 }]
Next, the winding resistance measurement as a parameter estimation method for position sensorless control of the permanent magnet synchronous motor of the present invention is performed by applying a voltage response of three types of switching patterns and by applying two more voltage applications. Then, the initial position angle is estimated, and based on the initial position angle, the winding resistance is measured by the command voltage and the detected current when the d-axis current is controlled to be constant so as not to rotate the motor.
[0028]
Next, measurement of an electromotive force coefficient as a parameter estimation method for position sensorless control of the permanent magnet type synchronous motor of the present invention will be described.
[0029]
In the measurement of the electromotive force coefficient, a provisional value KEM of the electromotive force coefficient is given from the rated voltage and rated rotation speed of the nameplate, and the electric motor is rotated by an existing sensorless control method. The electromotive force coefficient K EM in the controller is adjusted so that the speed ω M obtained from the estimated electromotive force e M in the controller matches the actual speed ω, and as a result, the electromotive force coefficient can be measured.
[0030]
Specifically, it is as follows.
[0031]
It is assumed that the electromotive force e M estimated in the controller accurately estimates the actual electromotive force e (e≈e M ). Further, an estimated speed ω M (= e M / K EM ) based on the estimated electromotive force e M is obtained.
[0032]
On the other hand, the rotational speed ω of the motor is known in the controller from the inverter frequency, and the relationship of ω = e / K E is obtained with the actual electromotive force coefficient K E. From the above, the speed error ω-ω M is
ω−ω M = (1 / K E −1 / K EM ) e M
Since the resulting a, using the speed difference omega-omega M corrects the electromotive force coefficient K EM in the controller, K EM when the speed error has converged to zero is the measured value of the electromotive force coefficient.
[0033]
Hereinafter, the parameter estimation method of the present invention will be described in detail.
(1) Estimation of inductance FIG. 1 is an analysis model diagram of PMSM according to the present invention.
[0034]
In this figure, 1 is a reverse salient pole type permanent magnet synchronous motor, 2 is a reverse salient magnetic pole of the motor, 3 is a winding, 4 is a transistor inverter, 5 is a switching element (switching transistor), and 6 is a diode.
[0035]
The voltage equation when the motor is stopped on the stationary coordinate α u −β u axis defined in the figure is expressed by the following equation.
[0036]
[Expression 1]
[0037]
[Expression 2]
[0038]
However, vα u, vβ u: α u -β u -axis armature voltage, iα u, iβ u: α u -β u -axis armature current, R: winding resistance, L d, L q: d -q -axis Inductance, p (= d / dt): a differential operator.
[0039]
When the motor is stopped, the initial current is zero and the voltage vector v (100) [1.0 is assigned to the conduction of the switching elements of + and − for the time T s , and the motor applied voltage at the time of u + , v − , w − conduction is assigned. The current i u (T s ) when v (100) is applied is obtained by the following equation under the approximation of the resistance R≈0.
[0040]
[Equation 3]
[0041]
[Expression 4]
[0042]
If I 1 and I 2 in the above equation (4) are known, L d and L q can be calculated by the following equations.
[0043]
[Equation 5]
[0044]
On the other hand, α v -β v and α w -β w axes are respectively defined in advance by 2π / 3 and 4π / 3 with respect to the α u -β u axis, and v (010) and v (001) are expressed as T s. The current when only the voltage is applied is given by the following equations (6) and (7).
[0045]
[Formula 6]
[0046]
[Expression 7]
[0047]
From the α axis in the above equations (3), (6), (7), I 1 is
[0048]
[Equation 8]
[0049]
And obtained. I 2 is obtained in two ways from the β-axis current and the I 1 and α-axis currents of the above equation (8), and is obtained by the following equations (9) and (10) as I 21 and I 22 , respectively.
[0050]
[Equation 9]
[0051]
[Expression 10]
[0052]
L d and L q are calculated using I 21 and I 22 in the above expressions (9) and (10), respectively, and the average values are used as estimated values. Further, the rotor position θ can be estimated using i u (T s ), i v (T s ), and i w (T s ) (for example, see the prior art document (3)).
(2) Estimation of winding resistance The voltage equation on the dq axis when the motor is stopped (θ = 0) is expressed by the following equation (11).
[Expression 11]
[0054]
The winding resistance R can be estimated by the following equation (12) using the command voltage v d when the q-axis current is zero and a constant current I d is passed through the d-axis so as not to generate torque.
[0055]
[Expression 12]
[0056]
(3) Estimation of speed electromotive force coefficient The initial value of the speed electromotive force coefficient is given as K E0 = V 0n / ω n from the rated voltage V 0n and the rated speed ω n described in the motor nameplate . Using this initial value, no-load operation is performed at the rated speed by sensorless control based on electromotive force estimation (for example, see the prior art document (1)). In this case, first position change amount [Delta] [theta] w of the motor of the sample period between T by using the speed electromotive force e and the electromotive force coefficient K E,
[0057]
[Formula 13]
[0058]
And obtained. On the other hand, Δθ w is
[0059]
[Expression 14]
[0060]
It is expressed. Here, e M (n) is a speed electromotive force estimated value, K EM (n−1) (= K E0 ) is a speed electromotive force coefficient estimated value, KθΔiγ (n) is a position correction term, and Kθ is a position estimated. Gain, Δiγ (n) is a current estimation error.
[0061]
Since e = e M (n) is established in the constant speed state, the speed electromotive
[0062]
[Expression 15]
[0063]
It can be written. Thus, by using the electromotive force coefficient estimated gain K K a 1 / K EM (n) from Δiγ as shown in FIG. 2,
[0064]
[Expression 16]
[0065]
Can be estimated. In FIG. 2, 11 is an electromotive force estimator.
[0066]
[Table 1]
[0067]
Experiments were performed using the 6-pole 1.5 kW, 1500 rpm test machine shown in Table 1.
[0068]
FIG. 3 shows estimation results of L d and L q . The estimated values at 8 points different from each other by π / 4 are plotted, and a total of 256 estimation results are shown.
[0069]
At this time, the voltage vector application time T s was set to 195 μs. The average values of the estimation results are L d = 4.51 mH and L q = 9.55 mH, and the errors from the motor parameters in Table 1 can be estimated at 4.9% and 0.4%, respectively.
[0070]
The resistance R was estimated from the average value of the resistance when I d = 4, 7, 10 A was given by the above equation (12), and R = 0.548Ω was estimated. For estimation of speed electromotive force coefficient K E, it was designed K K as time constant 0.44S.
[0071]
FIG. 4 is an estimation result of 1 / K EM at the rated speed when the initial value K E0 of the speed electromotive force coefficient is given as 1.5 times, 1.0 times, and 0.5 times K E.
[0072]
As apparent from FIG. 4, obtained convergence characteristics of the time constant 0.44s as designed, the convergence value K EM = 0.208V / rad / s, and the the error between K E of Table 1 5.0 %.
[0073]
According to the present invention, a parameter measurement method (parameters: L d , L q , R, K E ) of the sensorless salient pole type PMSM was proposed, and good results were obtained.
[0074]
In addition, this invention is not limited to the said Example, A various deformation | transformation is possible based on the meaning of this invention, and these are not excluded from the scope of the present invention.
[0075]
【The invention's effect】
As described above in detail, according to the present invention, it is possible to estimate the parameters of the electric motor and control the electric motor using the electric motor drive device without using the rotor position sensor. Light weight, low price, and can improve reliability.
[Brief description of the drawings]
FIG. 1 is an analysis model diagram of PMSM according to the present invention.
FIG. 2 is an explanatory diagram of estimation of a speed electromotive force coefficient.
FIG. 3 is a diagram illustrating an inductance estimation result.
FIG. 4 is a diagram showing a convergence characteristic of a speed electromotive force coefficient.
[Explanation of symbols]
DESCRIPTION OF
6
Claims (4)
前記電動機のパラメータはインダクタンスであり、電動機停止時に初期電流零の状態で、インバータの特定のスイッチング素子に短時間Ts だけオン信号を与え、その時に検出される各相の電流Iu (Ts ),Iv (Ts ),Iw (Ts )に基づいてインダクタンスLd ,Lq を計測する永久磁石形同期電動機の制御方法であって、
前記インダクタンスL d ,L q は、以下の方法により得ることを特徴とする永久磁石形同期電動機の制御方法。
回転子位置角θで停止しているとき、初期電流零で、スイッチング素子u + ,v - ,w - に時間T s だけオン信号を与えたときの検出電流I u1 (T s ),I v1 (T s ),I w1 (T s )から次式のIα u (T s ),Iβ u (T s )を計算し、
Iα u (T s )=√(2/3)・{I u1 (T s )−1/2・I v1 (T s )
−1/2・I w1 (T s )}
Iβ u (T s )=1/√2・{I v1 (T s )−I w1 (T s )}
一方、電動機の電圧方程式より、Iα u (T s ),Iβ u (T s )は次式で表わし、
Iα u (T s )=I 1 +I 2 cos2θ
Iβ u (T s )=I 2 sin2θ
ただし、
I 1 =√(2/3)・(L d +L q )V dc T s /(2L d L q ),
I 2 =√(2/3)・(L q −L d )V dc T s /(2L d L q )である。
上式のI 1 ,I 2 が既知となれば、L d ,L q を次式で得る。
L d =√(2/3)・V dc T s /(I 1 +I 2 ),
L q =√(2/3)・V dc T s /(I 1 −I 2 )
ただし、
I 1 ,I 2 を求めるために初期電流零でスイッチング素子u - ,v + ,w - に時間T s だけオン信号を与えたときの、検出電流I u2 (T s ),I v2 (T s ),I w2 (T s )及びスイッチング素子u - ,v - ,w + に時間T s だけオン信号を与えたときの検出電流I u3 (T s ),I v3 (T s ),I w3 (T s )から、次式のIα v (T s ),Iβ v (T s ),Iα w (T s ),Iβ w (T s )を計算する。
Iα v (T s )=√(2/3)・{I v2 (T s )−1/2・I w2 (T s )
−1/2・I u2 (T s )}
=I 1 +I 2 cos2(θ−2π/3)
Iβ v (T s )=1/√2{I w2 (T s )−I u2 (T s )}=I 2
sin2(θ−2π/3)
Iα w (T s )=√(2/3){I w3 (T s )−1/2・I u3 (T s )
−1/2・I v3 (T s )}
=I 1 +I 2 cos2(θ+2π/3)
Iβ w (T s )=1/2√{I u3 (T s )−I v3 (T s )}=I 2
sin2(θ+2π/3)
ただし、I 1 ,I 2 は次式で計算する。
I 1 ={Iα u (T s )+Iα v (T s )+Iα w (T s )}/3
I 2 =√{2/3(Iβ u (T s ) 2 +Iβ v (T s ) 2
+Iβ w (T s ) 2 }または、
I 2 =√〔2/3{(Iα u (T s )−I 1 ) 2 +(Iα v (T s )
−I 1 ) 2 +(Iα w (T s )−I 1 ) 2 }〕 In the control method of the permanent magnet type synchronous motor that estimates the parameters of the motor using the motor drive device without using the rotor position sensor and controls the motor,
The parameter of the electric motor is inductance, and when the electric motor is stopped, the initial current is zero, and an ON signal is given to a specific switching element of the inverter for a short time T s , and each phase current I u (T s) detected at that time. ), I v (T s) , a inductance L d, a control method of L q a to that permanent magnet synchronous motor measured based on the I w (T s),
The inductances L d and L q are obtained by the following method.
When the rotor is stopped at the rotor position angle θ, the detected currents I u1 (T s ) and I v1 when the ON current is given to the switching elements u + , v − and w − for the time T s with the initial current zero. The following formulas Iα u (T s ) and Iβ u (T s ) are calculated from (T s ) and I w1 (T s ) ,
Iα u (T s ) = √ (2/3) · {I u1 (T s ) −1 / 2 · I v1 (T s )
−1 / 2 · I w1 (T s )}
Iβ u (T s ) = 1 / √2 · {I v1 (T s ) −I w1 (T s )}
On the other hand, from the voltage equation of the motor, Iα u (T s ) and Iβ u (T s ) are expressed by the following equations:
Iα u (T s ) = I 1 + I 2 cos 2θ
Iβ u (T s ) = I 2 sin 2θ
However,
I 1 = √ (2/3) · (L d + L q ) V dc T s / (2L d L q ),
I 2 = √ (2/3) · (L q −L d ) V dc T s / (2L d L q ).
If I 1 and I 2 in the above equations are known, L d and L q are obtained by the following equations.
L d = √ (2/3) · V dc T s / (I 1 + I 2 ),
L q = √ (2/3) · V dc T s / (I 1 −I 2 )
However,
Detected currents I u2 (T s ), I v2 (T s ) when an ON signal is given to the switching elements u − , v + , w − for a time T s with zero initial current to obtain I 1 and I 2 ), I w2 (T s) and the switching element u -, v -, detection current I u3 (T s when received only on signal w + two hours T s), I v3 (T s), I w3 ( From T s , Iα v (T s ), Iβ v (T s ), Iα w (T s ), and Iβ w (T s ) are calculated.
Iα v (T s ) = √ (2/3) · {I v2 (T s ) −1 / 2 · I w2 (T s )
−1 / 2 · I u2 (T s )}
= I 1 + I 2 cos2 (θ-2π / 3)
Iβ v (T s ) = 1 / √2 {I w2 (T s ) −I u2 (T s )} = I 2
sin2 (θ-2π / 3)
Iα w (T s ) = √ (2/3) {I w3 (T s ) −1 / 2 · I u3 (T s )
−1 / 2 · I v3 (T s )}
= I 1 + I 2 cos2 (θ + 2π / 3)
Iβ w (T s) = 1 / 2√ {I u3 (T s) -I v3 (T s)} = I 2
sin2 (θ + 2π / 3)
However, I 1 and I 2 are calculated by the following equations.
I 1 = {Iα u (T s ) + Iα v (T s ) + Iα w (T s )} / 3
I 2 = √ {2/3 (Iβ u (T s ) 2 + Iβ v (T s ) 2
+ Iβ w (T s ) 2 } or
I 2 = √ [2/3 {(Iα u (T s ) −I 1 ) 2 + (Iα v (T s )
−I 1 ) 2 + (Iα w (T s ) −I 1 ) 2 }]
前記電動機のパラメータは巻線抵抗であり、3種類のスイッチングパターンの電圧印加の電流応答により、更に2回の電圧印加により、初期位置角推定を行い、該初期位置角に基づき、前記電動機を回転させないようにd軸電流一定値に制御したときの指令電圧、検出電流により巻線抵抗を計測することを特徴とする永久磁石形同期電動機の制御方法。In the control method of the permanent magnet type synchronous motor that estimates the parameters of the motor using the motor drive device without using the rotor position sensor and controls the motor,
The parameter of the motor is winding resistance, and the initial position angle is estimated by applying the voltage response of three types of switching patterns and by applying the voltage twice, and the motor is rotated based on the initial position angle. A control method for a permanent magnet type synchronous motor, characterized in that winding resistance is measured by a command voltage and a detected current when the d-axis current is controlled to be a constant value so as not to cause a failure.
前記電動機のパラメータは起電力係数であり、予めコントローラに記憶されている定格電圧と定格回転数から起電力係数の暫定値KEMを与え、既存のセンサレス制御法で電動機を回転させ、コントローラ内の推定起電力eM から得られる速度ωMが実際の速度ωに一致するように、コントローラ内の起電力eM から得られる速度ωMが実際の速度ωに一致するようにコントローラ内の起電力係数KEMを調整することにより、起電力係数を計測することを特徴とする永久磁石形同期電動機の制御方法。In the control method of the permanent magnet type synchronous motor that estimates the parameters of the motor using the motor drive device without using the rotor position sensor and controls the motor,
The parameter of the electric motor is an electromotive force coefficient. A provisional value K EM of the electromotive force coefficient is given from the rated voltage and the rated rotational speed stored in advance in the controller, and the electric motor is rotated by an existing sensorless control method. The electromotive force coefficient K in the controller is set so that the speed ωM obtained from the electromotive force e M in the controller matches the actual speed ω, so that the speed ωM obtained from the estimated electromotive force e M matches the actual speed ω. A control method of a permanent magnet type synchronous motor, wherein an electromotive force coefficient is measured by adjusting EM .
コントローラ内で推定された起電力eM は実際の起電力eを正確に推定しているとし(e≒eM )、また、推定起電力eM に基づいた推定速度ωM (=eM /KEM)が得られる。
一方、インバータ周波数よりモータの回転速度ωはコントローラ内でわかり、実際の起電力係数KE との間にω=e/KE の関係が得られる。
以上より、速度誤差ω−ωM は、
ω−ωM =(1/KE −1/KEM)eM
と得られるので、速度差ω−ωM を用いてコントローラ内の起電力係数KEMを補正し、速度誤差が零に収束したときの起電力係数の暫定値KEMを起電力係数の計測値とする。4. The control method for a permanent magnet synchronous motor according to claim 3 , wherein the electromotive force coefficient is obtained by the following method.
The electromotive force e M estimated in the controller and that accurately estimate the actual electromotive force e (e ≒ e M), also the estimated speed based on the estimated electromotive force e M ω M (= e M / K EM ) is obtained.
On the other hand, the rotational speed ω of the motor is known in the controller from the inverter frequency, and the relationship of ω = e / K E is obtained with the actual electromotive force coefficient K E.
From the above, the speed error ω-ω M is
ω−ω M = (1 / K E −1 / K EM ) e M
Since the resulting a, using the speed difference omega-omega M corrects the electromotive force coefficient K EM in the controller, the provisional value K EM of the electromotive force coefficient when the speed error has converged to zero measured value of the electromotive force coefficient And
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JP4319377B2 (en) * | 2002-05-31 | 2009-08-26 | 三菱電機株式会社 | Permanent magnet motor drive device, hermetic compressor, refrigeration cycle device, and permanent magnet generator drive device |
AU2004237551B2 (en) | 2003-05-08 | 2007-07-26 | Mitsubishi Denki Kabushiki Kaisha | Method for providing energy saving service, method for deciding specification of electric motor, method for providing compressor version-up service, method for providing energy saving service using drive device for permanent magnet electric motor, compressor exchange method, and freezing/air conditioning device |
JP4692328B2 (en) * | 2006-02-27 | 2011-06-01 | 株式会社日立製作所 | Permanent magnet synchronous motor control device, control method, and module |
JP2009194974A (en) * | 2008-02-13 | 2009-08-27 | Panasonic Corp | Inverter apparatus |
FR2933550B1 (en) * | 2008-07-01 | 2012-10-12 | Schneider Toshiba Inverter Europe Sas | METHOD FOR DETERMINING THE INDUCTIONS OF A SYNCHRONOUS MACHINE WITH PERMANENT MAGNETS |
JP6261396B2 (en) * | 2014-03-13 | 2018-01-17 | 日本特殊陶業株式会社 | Permanent magnet type synchronous motor vector control device and magnet magnetic flux estimation device |
JP6343235B2 (en) * | 2014-12-17 | 2018-06-13 | 株式会社ミツバ | Brushless motor driving apparatus and driving method |
CN115864929B (en) * | 2022-12-27 | 2023-06-06 | 江西安百川电气有限公司 | Method and device for detecting rotor out-of-step of permanent magnet synchronous motor |
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