JP3780346B2 - Pulse compression method and apparatus for monostatic radar - Google Patents

Description

  The present invention relates to a pulse compression method and apparatus for monostatic radar that uses one common antenna by switching between transmission and reception.

  A pulse radar such as a wind profiler radar (WPR) is known. This is because radio waves are transmitted from the ground to the sky, weak radio waves scattered by atmospheric turbulence and precipitation particles are received, and the frequency difference (Doppler frequency shift) and received signal strength of the transmitted and received radio waves are measured. It is a radar that observes the altitude distribution of wind direction, wind speed, turbulence intensity, and rainfall intensity by measuring.

The average power of a pulse radar such as a wind profiler radar (WPR) is generally as small as several percent compared to the peak power (see Non-Patent Documents 1 and 2), and the maximum detection distance (R _max ) of the radar is increased. For this purpose, it is effective to increase the transmission time rate. If the transmission time rate can be increased, it not only contributes to the improvement of the S / N ratio of the radar, but the peak power of the transmission pulse can be reduced while maintaining the same S / N ratio (average power of the transmission wave). Cost can be reduced. A pulse compression method such as a coded pulse radar is an effective means for improving the S / N ratio of the radar.

  FIG. 10 is a diagram illustrating a relationship between a transmission pulse and a reception echo by a general pulse compression method. As illustrated, a transmission pulse train is transmitted as a transmission wave at the beginning of each cycle. In a monostatic radar that uses one common antenna by switching between transmission and reception, the antenna is switched to the reception side after receiving the transmission wave until it is switched to the transmission side in the next cycle, and receives a reception echo. . While switching to the reception side, the reception echo observation time is determined by the observation distance.

As a method for increasing the transmission time rate, a pulse compression method using a Barker code or the like is generally known. However, such a pulse compression method increases the side length of the autocorrelation function as the code length of the code sequence increases. Although the level is small, the minimum observable detection distance (R _min ) is large. In addition, it is well known that the side lobe level is likely to be easily affected by the Doppler shift due to the movement of the observation target (see Non-Patent Document 3), and the transmission bit rate is simply increased by increasing the number of bits of the code. There was a limit to raising it.
Masahisa Masuda, Toshio Ihara, Kenji Nakamura, Kenichi Okamoto, Tsutomu Onishi, "Development and initial observation data of radar for low-level atmospheric observation," Communications Research Laboratory quarterly report, vol.38, no.1, pp.15-33, Mar .1992. T Adachi, "Development of a 400MHz-band Wind Profiler Rader with RASS," Journal of the Communications Research Laboratory, vol.49, no.2, pp.211-216, June 2002. Tomonasa Kondo, Yoshimasa Ohashi, and Akira Akira Morimori, "Digital signal processing in measurement and sensors," Shigeo Sakurai (ed.), Shosodo Co., Ltd., 1993. RC Dixon, latest spread spectrum communication system, Satoshi Tateno, Shizuo Kataoka, Kiyoshi Iida (translation), JATEC Publishing, Tokyo, 1978. Shinya Sekizawa, Masaaki Heiwa, Yoshihide Kamio, Mitsuhiko Mizuno, "A system for measuring the spatio-temporal characteristics of multipath propagation paths using a virtual planar array," IEICE (B), vol.J83-B, no.9, pp .1303-1313, Sep. 2000. Akio Yasuda, “Prospects for GPS Technology,” Science Review (B), vol.J84-B, no.12, pp.2082-2091, Dec.2001.

  When the Doppler speed of the observation target is relatively slow like WPR, the time of one code period is sufficiently shorter than the time (correlation time) that the scattered wave from atmospheric turbulence can be regarded as coherent, and the Doppler shift in the time of one code period If the phase change of the scattered wave due to is sufficiently smaller than 2π, it is considered possible to use a code with a long period for the radar.

On the other hand, an M sequence (Maximal length sequence), which is one of PN (Pseudo Noise) codes and a finite code sequence is periodically repeated, has a sharp peak in its autocorrelation function (see Non-Patent Document 4). It is used in many fields such as measurement of impulse response of propagation path in mobile communication (see Non-Patent Document 5) and positioning by GPS (see Non-Patent Document 6). The advantage of applying this code to a radar is that the transmitted wave is a continuous wave, so that an amplifier with a large output power is not required, and the maximum detection distance (R _max ) is very high when a code sequence with a long period is used. The increase in size and the reduction of interference from other stations of the same frequency can be raised. However, the monostatic radar has been difficult to apply because it is difficult to sufficiently increase the degree of separation of transmitted and received signals in the receiver.

The present invention solves such a problem, and in the pulse compression technology of a monostatic radar, without increasing the minimum detectable distance (R _min ) of the radar and without decreasing the maximum detectable distance (R _max ). The purpose is to increase the transmission time rate and to reduce interference from other stations of the same frequency.

The pulse compression method for monostatic radar of the present invention uses one common antenna by switching between transmission and reception. A time length Tc of one cycle of the transmission pulse train is divided into a predetermined number N of divided time lengths T _IPP to generate a transmission pulse train on which bit information of the PN sequence code is placed. Each of the divided time lengths T _IPP is divided into a transmission time Ts and a remaining reception time, and a transmission pulse train of one period is divided into N parts of the transmission time Ts, modulated, and transmitted. Receive during reception time. By performing correlation processing between the time series data group demodulated from the received echo information from the observation target and the transmission pulse train, a complex time series of the received signal is calculated for each distance.

The pulse compression apparatus for monostatic radar according to the present invention uses one common antenna by switching between transmission and reception. The present invention relates to a generator for generating a PN sequence code, and a transmission in which the time length Tc of one transmission pulse train period is divided into a predetermined number N of divided time lengths T _IPP and the bit information of the PN sequence code is placed A transmission pulse train generator for generating a pulse train, and the divided time length T _IPP are divided into a transmission time Ts and a remaining reception time, respectively, and a transmission pulse train of one cycle is divided into N parts of the transmission time Ts. A transmission / reception switch that is modulated and transmitted and switched to receive during the remaining reception time, a time-series data group demodulated from reception echo information from an observation target, and the transmission pulse train By performing the correlation processing, the correlator is configured to calculate a complex time series of the received signal for each distance.

  According to the present invention, in the pulse compression technique of a monostatic radar, it is possible to increase the transmission time rate and increase the maximum detection distance without increasing the minimum detection distance of the radar.

  This not only contributes to an improvement in the S / N ratio of the radar, but can reduce the peak power of the transmission pulse while maintaining the same S / N ratio (average power of the transmission wave), thereby reducing the size and cost of the apparatus. .

  In addition, if incoherent integration is applied to the method of the present invention to improve the signal-to-noise ratio, it is possible to observe the atmosphere up to an altitude of 2 to 3 km even if the peak power of the transmitted pulse is 300 W and the power is relatively small. Obviously, if the method of the present invention is used, the radar apparatus can be reduced in size and cost, so that further WPR can be expected.

  In addition, since the own station and other stations having the same frequency can be separated using the difference in the types of M-sequences, it is possible to improve the frequency utilization efficiency.

  Hereinafter, the present invention will be described based on examples. FIG. 1 is a block diagram illustrating a schematic configuration of a pulse compression apparatus embodied according to the present invention. FIG. 2 is a diagram for explaining the operation of the pulse compression apparatus illustrated in FIG. 1 and shows the relationship between transmission pulses and reception echoes. FIG. 3 is a diagram for explaining the correlation processing in the correlator illustrated in FIG. In FIG. 2 and FIG. 3, the circled numbers added to the pulse train or signal indicate that they occur at the same circled numbers in FIG.

The PN sequence generator of FIG. 1 generates a PN sequence code, for example, an M sequence code having a predetermined code length L. The transmission pulse train generator generates a transmission pulse train {c ₁ , c ₂ ,..., C _N } on which bit information of the M-sequence code is placed.

The transmission pulse train (no information) shown at the top of FIG. 2 indicates that the time length Tc of one cycle of the transmission pulse train is divided into N (a predetermined number of integers) divided time lengths _TIPP . Further, each of the division time lengths T _IPP is divided into a transmission time Ts and a remaining reception time. Then, one cycle of the M-sequence code is divided into N parts of the N transmission times Ts, and the bit information is placed.

  The transmission wave in FIG. 2 shows a signal obtained by performing BPSK modulation on the above transmission pulse train using the carrier wave generated from the carrier oscillator by the BPSK modulator in FIG. This signal is amplified by the power amplifier and then transmitted from the transmitting / receiving antenna when switched to the transmission side by the transmission / reception switch.

Next, the information of the received echo (see FIG. 2) from the observation target is acquired as a time series data group {r ₁ , r ₂ ,..., R _N } in a period in which no transmission pulse is generated. That is, in FIG. 1, the received echo is input from the transmission / reception antenna to the low noise amplifier via the transmission / reception switcher, amplified here, and then input to the quadrature demodulator. The quadrature demodulator demodulates the time-series data group {r ₁ , r ₂ ,..., R _N } as a quadrature detection signal of the received echo using the same carrier as that used in the BPSK modulator.

The transmission pulse time interval T _IPP is set to be much shorter than the time length corresponding to the maximum detection distance R _max . The quadrature detection signal of the received echo to which the scattered waves having different distances are added is subjected to correlation processing with the transmission pulse train in the correlator of FIG. 1, so that a complex time series of the received signal can be calculated for each distance.

For example, the transmission pulse train illustrated in the lower side of FIG. 3 (six rows are shown) shows the transmission pulse train itself at the top, and the transmission pulse train is shifted to the next stage by a distance corresponding to the minimum detection distance R _min and correlated. The received signal strength as a result of processing is illustrated as a graph on the right side of the drawing. Hereinafter, similarly, a complex time series of received signals calculated by performing correlation processing while shifting the transmission pulse train to the maximum detection distance R _max is illustrated in the graph.

The time length T _c of one cycle of the transmission pulse train is N × T _IPP , and the theoretical maximum detection distance R _max is T _c · c / 2. Where c is the speed of light. However, as a use condition of this method, T _c needs to be determined in consideration of the correlation time and the allowable value of the range side lobe of the transmission pulse.

The main features of this method are as follows.
(1) Since the transmission time rate D _r (= pulse width T _s / T _IPP ) can be set high, the average transmission power P _av is large (the SN ratio is improved in proportion to P _av ).
(2) R _max is very large without increasing R _min in spite of using a relatively long cycle code.
(3) The range side lobe is very small under conditions where the Doppler shift does not affect the autocorrelation characteristics of the code.
(4) By using different types of M-sequences, interference from other stations with the same frequency as the own station can be suppressed.

On the other hand, (5) there is a drawback that the distance corresponding to the transmission period cannot be observed, and the application is limited. However, it is not a particular problem in applications where the observation results change slowly according to changes in the detection distance (altitude), as in the case of application to wind profiler radar (WPR). However, if reception is allowed during pulse transmission, it is possible in principle to solve (5) by setting _Dr to 100% in (1). However, in monostatic radar, it is desirable not to perform transmission and reception at the same time in order to increase the degree of separation between the transmission signal and the reception signal in the receiver. The present system having these features not only contributes to the improvement of the S / N ratio of the radar, but also can reduce the cost by reducing the peak power while maintaining the S / N ratio. In addition, it is possible to separate the local station and other stations of the same frequency using the difference in the type of M-sequence like the CDMA system of mobile communication, and the improvement of frequency utilization efficiency can be expected, so the same frequency was used. Various application examples are conceivable, such as a large number of radars can be arranged.

  FIG. 11 is a diagram showing an example in which the present invention is applied to an atmospheric high-density observation system. A wind profiler radar (WPR) equipped with the pulse compression device described above with reference to FIG. 1 is provided in each of a plurality of stations arranged at intervals of several kilometers. Each station assigns different M-sequence codes (C1, C2, C3...) As transmission pulses of the wind profiler radar (WPR), and each WPR performs atmospheric observation at the same time at the same frequency.

  Each WPR performs correlation processing using an M-sequence code assigned to its own station, thereby receiving and observing only its own signal. As a result, since interference signals from other stations with different types of M-sequence codes can be suppressed, a large number of WPRs can be arranged at a relatively close distance, so that high-density observation of the atmosphere can be performed.

  An arbitrary WPR can select and receive a signal of another station by performing a correlation process using the M-sequence code of the other station. As a result, since a plurality of WPRs can simultaneously observe a target at a certain position (Target) from different angles (bistatic observation), three-dimensional observation of atmospheric turbulence can be performed.

Next, it is shown that an increase in side lobe due to the influence of Doppler shift can be suppressed by using a code with an appropriate period for the radar. Moreover, it is confirmed by computer simulation that the peak power can be reduced while the 1.3 GHz band WPR to which the method of the present invention is applied keeps the average power of the transmission pulse low. Further, the effectiveness of the method of the present invention is confirmed by analyzing a simulation signal having a Doppler speed corresponding to the wind speed in the sky using a practical wind speed calculation method. Finally, it was confirmed that the wind at an altitude of 2 to 3 km can be observed even when the peak power of the transmission pulse is 300 W because the SN ratio is improved by the incoherent integration process in the method of the present invention.
1) Effect of Doppler shift on autocorrelation characteristics of M sequence In the method of the present invention, since the time length _{Tc of} one cycle of the transmission pulse train is relatively long, if the observation target is affected by Doppler shift caused by movement, the received signal Since the phase changes with time, the autocorrelation characteristics for a transmission pulse train carrying M-sequence bit information may be degraded. Therefore, using computer simulation, the autocorrelation characteristics between the received signal affected by the Doppler shift and the reference transmission pulse train were obtained, and the degree of degradation of the autocorrelation characteristics was investigated. However, the envelope of the transmission pulse is rectangular for simplicity. FIG. 4 shows a result of setting the transmission time rate _Dr to 1 and the code length N of the M sequence as a parameter. When the Doppler frequency f _d given to the transmission pulse train is constant, the amount of change Δφ in the phase of the received wave per T _c can be expressed by the following equation.

Δφ = 2π ・ T _c・ f _d [rad] (1)
However, f _d is 2v / λ, where v is the moving speed in the line-of-sight direction of the observation target and λ is the wavelength of the radio wave. Figure 4 (A) is the correlation peak for [Delta] [phi value E _p, FIG. 4 (B) shows the difference between G _p in dB value of the maximum value E _s of the correlation peak value E _p and the side lobes with respect to [Delta] [phi. 4 (A) and 4 (B), when Δφ = 0 (observation target is stationary), both E _p and G _p when the code length N is 2047 are both 66.2 dB, showing very good autocorrelation characteristics. You can see that However, as Δφ increases, E _p gradually decreases, but G _p decreases rapidly. This is due to the increase of Δφ which is proportional to the Doppler frequency, but decreases gradually correlation peak value E _p, sidelobe value E _s is shown to increase rapidly. Note that the results of FIG. 4 show substantially the same characteristics even when D _r <1.

When an M-sequence code having a relatively long period is used as in the present invention method, it is necessary to determine T _c so that the amount of degradation of autocorrelation characteristics due to the influence of f _d falls within an allowable range. In order to bring the deterioration of the autocorrelation characteristics of the code into an allowable range, the value of _Tc , which is the product of N and T _IPP , needs to be determined in consideration of the maximum expected Doppler frequency. Therefore, T _c of the transmission pulse train in the present invention system is, when Δφ is Δφ = BΔ [rad] when the condition of G _p = A _G is imposed, where f _dmax is the assumed maximum Doppler frequency, It is necessary to determine so as to satisfy the following condition obtained from equation (1).

T _c = N × T _IPP ≦ BΔ / (2π ・ f _dmax ) (2)
For example, when using the M-sequence of N = 2047 in the transmission pulse train, in order to secure the G _p 30 dB or more, Bideruta should be 110.4 deg follows from FIG. 4 (B). Accordingly, the value of T _c (= N × T _IPP ) is determined by substituting BΔ = 110.4 deg and f _dmax assumed in the equation (2).
2) Results of applying this method to WPR
2-1 Specifications of WPR to be examined The method of the present invention was applied to the 1.3 GHz band WPR shown in Table 1, and various studies were performed by computer simulation. From Table 1, the transmission frequency of WPR is 1.3575 GHz, and the peak power of the transmission signal is 300 W. The pulse width T _s is 0.8 μs and the altitude resolution is 120 m. The waveform shaping filter is Gaussian for simplicity, and its standard deviation σ is 0.1 T _s . However, the characteristic of the waveform shaping filter here is a summary of the characteristics of the transmission / reception filter. Transmission pulse train, Ts a 0.8 .mu.s, were generated carrying bit information of 11 stages M-sequence to a transmission pulse of 0.33 to D _r (code length N = 2047). The ratio between the transmission period and the reception period was 1: 1. Finally, the received signal for each altitude is obtained as time series data for each _Tc corresponding to the correlation processing period by the correlation processing. Here, _Tc is 4.91 ms, which is a sufficiently small value compared with the correlation time of 0.13 sec (see Non-Patent Document 1) measured with 1.3 GHz band WPR. This time corresponds to a coherent integration time performed in general WPR. The theoretical altitude R _m that can be observed under these conditions is a value of 360 m steps from the minimum observation altitude of 180 m to 736.7 km. The gain G _{a of the} transmission / reception antenna was 29 dBi, and its beam width θ _a was 5 deg (see Non-Patent Document 1). A simple comparison between this system shown in Table 1 and the monopulse radar that performs T _IPP = 50μs and coherent integration 100 times (coherent integration time 5 ms ≒ T _c ) under the same conditions as Table 1 with P _t and T _s In this method, the number of transmission pulses in the T _IPP period is about 20 times larger. For this reason, the average power of this method is about 20 times larger (13 dB), and the S / N ratio is superior by the same value.

2-2 Transmission Pulse and Autocorrelation Characteristics FIG. 5A shows a part of the waveform of the transmission pulse train after passing through the waveform shaping filter in the specifications shown in Table 1. FIG. 5B shows the autocorrelation characteristics between the transmission pulse train after shaping and the transmission pulse train before shaping. From this figure, it can be seen that the difference in dB value between the correlation peak and the side lobe of the autocorrelation characteristic is 66.2 dB, which is the same as the value of the autocorrelation characteristic of the M sequence when _Dr = 1 shown in FIG. It can be confirmed that there is.

2-3 Evaluation of the maximum observation altitude Next, the static characteristics of the method of the present invention in the specifications of Table 1 under the condition that the scattered wave due to the atmospheric turbulence in the sky is not affected by the Doppler shift due to the wind flow. Evaluate and examine the maximum observed altitude. First, the outline of the simulation performed this time will be described next. Radio waves were emitted from WPR to the sky, and scattered waves due to atmospheric turbulence at an observation altitude resolution of 120 m were received. The reflectivity of the atmosphere η at this time is _{^{0.38 · l -1/3 · C n 2}} [m -1], the structural parameters C _n ² with 10 ^{-0.000276R-13.862} [m ^-2/3] . R is the observed altitude. The phase of the scattered wave at each altitude was uniformly distributed.

6 (A) and 6 (B) show the correlation between the received power and phase deviation of the scattered wave obtained at every altitude by performing correlation processing on the time-series data group of received echoes obtained within the period of _Tc in this method. It is an example of a calculation result. Here, the system noise figure NF is used as a parameter. When the NF is 3 dB, the calculated value of the received power with an altitude of about 1.6 km or less is within ± 3 dB of the theoretical value and almost matches the theoretical value, and the deviation of the calculated value from the true value is also true for the phase. Is a small value within ± 30 deg. However, since the results of FIG. 6 vary in value due to the influence of noise, statistical evaluation was performed next. FIG. 7 shows the results of FIG. 6 obtained 1000 times, and the probability that the calculated values of the received power and the phase are ± 3 dB or less from the true value and ± 30 deg or less is obtained for each altitude. From the figure, it can be seen that the maximum observed altitude is 1.02 km and 1.72 km, respectively, in order that the probability of having the above accuracy is 0.9 and 0.5 when the NF is 3 dB. It can also be seen that the SN ratio when receiving scattered waves from the maximum observed altitudes of 1.02 km and 1.72 km is 3.2 dB and −3.3 dB from the theoretical curve of the altitude-to-SN ratio shown in FIG. Thus, in order to perform observation with a small error with a probability of 0.9, the maximum observation altitude must be 1.02 km, and the received power of scattered waves at this altitude exceeds 3 dB from the noise power. It is considered that the calculation result by this simulation is appropriate.

2-4 Analysis of Doppler spectrum Here, the effectiveness of the method of the present invention is confirmed by analyzing the wind speed in the sky using a simple Doppler velocity model. Normally, WPR measures wind direction and speed using multiple antenna beams with different zenith angles. In this study, we use one antenna beam to obtain the Doppler spectrum (frequency power spectrum) in the line-of-sight direction of the beam. Basic study will be conducted. The scattered waves used in this simulation were generated every 120 m, which is the observation altitude resolution, and the Doppler velocity of all the scattered waves was set to a constant value of 5 m / s (f _d = 45.2 Hz). However, the phase of each scattered wave was given by uniform distribution. This Doppler velocity value corresponds to a maximum horizontal wind velocity of 28.8 m / s when the zenith angle in the antenna viewing direction is 10 deg. In effect this Doppler velocity, [Delta] [phi is seen to 33.0 dB secured from (1) 79.9 deg next from the equation, G _p is 4 in N = 2047 (B).

Here, the frequency power spectrum of the scattered wave for each altitude was obtained as a single time-series data group with 256 received signals for each altitude obtained every T _c (= 4.91 ms). Under this condition, frequency power spectrum results can be obtained with a maximum Doppler velocity of ± 11.24 m / s (f _dmax of ± 101.7 Hz) and a Doppler velocity resolution of 87.8 mm / s (frequency resolution of 0.79 Hz). FIG. 9 shows the calculation result of the frequency power spectrum obtained by the simulation. At this time, NF was set to 3 dB. From this figure, the Doppler velocity of scattered waves can be observed accurately up to an altitude of 5 km, confirming the effectiveness of this method. However, since the scattered wave from the actual atmospheric turbulence has a spread around a certain Doppler velocity, its power is dispersed, so observation over 5 km is difficult as shown in the results of FIG. .

2-5 Analogue of maximum observed altitude by incoherent integration Generally, WPR performs FFT on time series data after coherent integration, and then incoherent integration on the time series data of the FFT result (see Non-Patent Document 1). )I do. When the noise component added to the signal is white noise, the S / N ratio improves in proportion to the number of integrations when coherent integration is performed. In incoherent integration also improves in proportion to the square root of the number of integrations (see Non-Patent Document 1). For example, if further incoherent integration is performed such that the frequency power spectrum obtained from 256 time-series data groups is integrated 80 times (about 100 seconds), it can be expected that the SN ratio is theoretically improved by 9.5 dB. In the above 2-3, it has already been described that it is necessary to secure the SN ratio of 3.2 dB and -3.3 dB in order to perform reasonable observations with the probabilities of 0.9 and 0.5. However, by performing incoherent integration, the SN ratio can be increased. If 9.5 dB can be further improved, it can be seen from FIGS. 7 and 8 that the maximum observation altitude increases to 2.14 km and 3.21 km. From this, when the method of the present invention is used, it can be expected that even if the peak power of the transmission pulse is 300 W, it can be observed up to an altitude of about 2-3 km.

It is a block diagram which illustrates schematic structure of the pulse compression device actualized based on this invention. FIG. 2 is a diagram for explaining the operation of the pulse compression apparatus illustrated in FIG. 1 and shows the relationship between transmission pulses and reception echoes. It is a figure explaining the correlation process in the correlator illustrated in FIG. It is a figure which shows the influence of the Doppler shift which gives to the autocorrelation characteristic of M series. It is a figure which shows the characteristic of the transmission pulse by this invention system. It is a figure which shows an example of the calculation result of the reception power and phase for every altitude. It is a figure which shows the probability that the calculated value of the received power and phase for every altitude will be ± 3 dB or less and ± 30 deg or less from the true value. It is a figure which shows the theoretical value of SN ratio. It is a figure which shows the altitude distribution of a frequency power spectrum. It is a figure which shows the relationship between the transmission pulse and reception echo by a general pulse compression method. It is a figure which shows the example which applied this invention to the high-density observation system of air | atmosphere.

Claims (4)

In a pulse compression method for monostatic radar using one common antenna switched between transmission and reception,
A time length Tc of one cycle of the transmission pulse train is divided into a predetermined number N of divided time lengths T _IPP to generate a transmission pulse train on which bit information of the PN sequence code is placed,
Each of the divided time lengths T _IPP is divided into a transmission time Ts and a remaining reception time, and a transmission pulse train of one period is divided into N parts of the transmission time Ts, modulated, and transmitted. Received during the reception time of
By performing correlation processing between the time-series data group demodulated from the received echo information from the observation target and the transmission pulse train, the complex time series of the received signal is calculated for each distance,
The value of the time length Tc is determined based on the assumed maximum Doppler frequency f _dmax so that the degradation amount of the autocorrelation characteristic due to the influence of the Doppler frequency f _d given to the transmission pulse train is within an allowable range ,
A plurality of stations each having the monostatic radar are arranged at intervals, and each station assigns a different code as a PN sequence code used for transmission pulse train generation, and each station performs atmospheric observation at the same frequency and at the same time. pulse compression method for monostatic radar consisting performed. Each station receives and observes only its own signal by performing correlation processing with the PN sequence code assigned to itself, and performs correlation processing with the PN sequence code of the other station. 2. A pulse compression method for monostatic radar according to claim 1 , comprising selecting and receiving a signal. In a pulse compression apparatus for monostatic radar using one common antenna for switching between transmission and reception,
A generator for generating a PN sequence code;
A transmission pulse train generator that divides a time length Tc of one cycle of the transmission pulse train into a predetermined number N of divided time lengths T _IPP and generates a transmission pulse train on which bit information of a PN sequence code is placed;
Each of the divided time lengths T _IPP is divided into a transmission time Ts and a remaining reception time, and a transmission pulse train of one period is divided into N parts of the transmission time Ts, modulated, and transmitted. A transmission / reception switcher that is switched to receive during the reception time of
And time-series data group demodulated from received echo information from the observation target, by performing a correlation processing between the transmission pulse train, comprising a correlator for calculating a complex time-series of the received signal to the distance-and,
The value of the time length Tc is determined based on the assumed maximum Doppler frequency f _dmax so that the degradation amount of the autocorrelation characteristic due to the influence of the Doppler frequency f _d given to the transmission pulse train is within an allowable range ,
A plurality of stations each having the monostatic radar are arranged at intervals, and each station assigns a different code as a PN sequence code used for transmission pulse train generation, and each station performs atmospheric observation at the same frequency and at the same time. Pulse compression device for monostatic radar consisting of performing . Each station receives and observes only its own signal by performing correlation processing with the PN sequence code assigned to itself, and performs correlation processing with the PN sequence code of the other station. 4. A pulse compression apparatus for monostatic radar as claimed in claim 3 , comprising selecting and receiving a signal.

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