JP3743417B2 - Receiver - Google Patents

Description

[0001]
BACKGROUND OF THE INVENTION
The present invention relates to a receiving apparatus in terrestrial digital broadcasting.
[0002]
[Prior art]
The configuration of a conventional receiving apparatus will be described with reference to FIG.
[0003]
The ISDB-T signal received by the antenna 1 is amplified by the RF amplifier 2 and input to the mixer 3. The PLL 4 generates an oscillation signal having a predetermined frequency, and this oscillation signal is supplied to the mixer 3 to be frequency-converted to the first intermediate frequency (fIF1 = 1400 MHz). The output of the mixer 3 is input to the BPF 5, and only the desired reception segment is selected and passed, the signals of other segments and adjacent channels are suppressed, input to the IF amplifier 6 and amplified, and then the mixer 7 and the mixer 8. And a quadrature mixer composed of the phase shifter 10. The output of the quadrature mixer is further suppressed by LPF 51 and LPF 52 to suppress signals of other segments and adjacent channels, only the desired reception segment is selected, amplified to a predetermined level of the next-stage ADC by BB amplifiers 30 and 31, and input to ADCs 32 and 33 Convert to digital signal. The signals from the ADCs 32 and 33 and the numerically controlled oscillator NCO 35 are frequency-converted by the complex multiplier 34. After the image frequency is suppressed by the LPFs 36 and 37, the frequency is input to the OFDM demodulator 38, demodulated in accordance with the modulation processing at the time of ISDB-T transmission, and TS is output to the TS output terminal.
[0004]
In addition, as prior art document information regarding the invention of this application, for example, Japanese Patent Application No. 2001-306121 of an unpublished in-house application is cited.
[0005]
[Problems to be solved by the invention]
In terrestrial digital broadcasting, analog broadcasting may exist in adjacent channels, and high suppression capability of adjacent channels is required.
[0006]
In particular, in terrestrial digital audio broadcasting performed using the VHF band, an analog broadcasting is always present in the upper channel and the lower channel, so that particularly high suppression capability is required.
[0007]
In terrestrial digital audio broadcasting, eight segments are connected and transmitted, and the spectrum shown in FIG.
[0008]
When the desired reception segment is the third from the bottom, as shown in FIG. 10 (b), both the lower adjacent audio carrier and the upper adjacent are sufficiently separated from the desired segment, and sufficient suppression is possible with the BPF 5, LPF 51, and LPF 52.
[0009]
When the first segment from the top is received, the frequency with the upper adjacent video carrier is close as shown in FIG.
[0010]
When the first segment from the bottom is received, the frequency with the upper side of the lower adjacent audio carrier is close as shown in FIG.
[0011]
Thus, the degree of suppression of BPF5, LPF51, and LPF52 is higher than when receiving the first segment from the top or the first segment from the bottom and when receiving the segment near the center of the channel, for example, the third segment from the bottom. More needed. In order to realize this characteristic, it is necessary to increase the order in order to make the characteristics of each filter steep. However, there is a problem that the shape of the filter increases as the order is increased. It is an object of the present invention to provide a receiving apparatus that can sufficiently obtain adjacent suppression capability regardless of the position of a receiving segment while maintaining the order of each filter.
[0012]
[Means for Solving the Problems]
The invention according to claim 1 of the present invention is a receiving device comprising a mixer circuit for frequency-converting a received signal and a filter for removing unnecessary signals from the frequency-converted received signal, and connecting a plurality of segments. Receiving apparatus having means for changing the cut-off frequency of a filter in accordance with the position of a segment to select and receive a segment of a signal modulated by OFDM, and capable of sufficiently obtaining adjacent suppression capability Can be obtained.
[0013]
The invention according to claim 2 of the present invention includes a first filter and a second filter for removing unnecessary signals from the frequency-converted received signal, and the second filter according to the characteristics of the first filter. The receiving device according to claim 1, further comprising means for changing a cutoff frequency, and obtaining a receiving device capable of sufficiently obtaining adjacent suppression capability even when the characteristics of the first filter vary in manufacturing. it can.
[0014]
According to a third aspect of the present invention, there is provided a frequency adjustment circuit connected to a reference clock and a DC voltage correction circuit connected to the frequency adjustment circuit, and the DC voltage correction circuit is a filter for different reference clock signal frequencies. 2. A receiving apparatus according to claim 1, further comprising means for making the cut-off frequency constant, wherein the receiving apparatus can perform frequency correction of a filter corresponding to a plurality of reference clocks and sufficiently obtain adjacent suppression capability. be able to.
[0015]
According to a fourth aspect of the present invention, there is provided a direct current voltage correction circuit between the second filter and the frequency adjustment circuit, and further comprising means for performing direct current voltage correction with respect to different reference clock signal frequencies. In this case, it is possible to obtain a receiving device that can correspond to a plurality of reference clocks, perform frequency correction of the filter in accordance with variations in manufacturing characteristics of the first filter, and can sufficiently obtain adjacent suppression capability.
[0016]
The invention according to claim 5 of the present invention is the receiving apparatus according to claim 1, comprising means for changing the cutoff frequency in accordance with the bit error rate, and adjacent suppression is performed even when there is a secular change of the filter. It is possible to obtain a receiving apparatus capable of obtaining sufficient capability.
[0017]
DETAILED DESCRIPTION OF THE INVENTION
Hereinafter, embodiments of the present invention will be described with reference to FIGS.
[0018]
FIG. 1 is a block diagram of a receiving apparatus according to the present invention.
[0019]
In FIG. 1, 1 is an antenna, 2 is an RF amplifier, 3 is a mixer, 4 is a PLL, 5 is a BPF, 6 is an IF amplifier, 7 and 8 are mixers, 9 is a PLL, 10 is a phase shifter, and 61 and 62 are variable Cut-off low-pass filter (LPF), 30 and 31 are baseband amplifiers (BB amplifiers), 32 and 33 are AD converters (ADC), 34 is a complex multiplier, 35 is a numerically controlled oscillator (NCO), 36, 37 is a digital LPF, 38 is an OFDM demodulator, 21 is a transport stream output terminal (TS output terminal), and 63 is a microprocessor. The cutoff frequency of the filter is changed according to the position of the segment received by the microprocessor 63.
[0020]
The configuration of this receiving apparatus is a receiving apparatus that performs partial reception of one segment. The ISDB-T signal received by the antenna 1 is amplified by the RF amplifier 2 and input to the mixer 3. The PLL 4 generates an oscillation signal having a predetermined frequency, and this oscillation signal is supplied to the mixer 3 to be frequency-converted to the first intermediate frequency (fIF1 = 1400 MHz). fIF1 is a fixed frequency.
[0021]
At this time, if the frequency of PLL2 is set higher than fIF1, the spectrum is inverted with respect to the antenna input in the first intermediate frequency band. Although it is possible to set the frequency of PLL2 on the lower side than fIF1, this embodiment will be described on the higher side.
[0022]
The first unnecessary signal is removed by the BPF 5 from the signal frequency-converted to fIF1. The fIF1 segment signal that has been band-limited by the BPF 5 is amplified by the IF amplifier 6 and multiplied by the PLL 9 in the mixer 8 and the signal obtained by shifting the PLL 9 signal by 90 ° by the phase shifter 10 in the mixer 7 to obtain a complex baseband signal. Is output.
[0023]
The output of the quadrature mixer is further suppressed by LPF 61 and LPF 62 to suppress signals of other segments and adjacent channels, only the desired reception segment is selected, amplified to the predetermined level of the next-stage ADC by BB amplifiers 30 and 31, and input to ADCs 32 and 33 Convert to digital signal. fOFS = 500 KHz is converted to baseband by complex multiplication of the NCO 35 signal and the ADCs 32 and 33 by the complex multiplier 34. After the frequency to be imaged is suppressed by the LPFs 36 and 37, it is input to the OFDM demodulator 38 and demodulated such as complex Fourier transform, frequency interleaving, time interleaving, and error correction in accordance with the modulation processing at the time of ISDB-T transmission. Processing is performed, and TS is output to the TS output terminal.
[0024]
Hereinafter, the suppression of the adjacent carrier by the BPF 5, LPF 61, and LPF 62 will be described with reference to FIGS. 6 and 7, assuming that the required value of the suppression level of the adjacent carrier according to the received segment position is 40 dB or more.
[0025]
FIG. 6A shows the spectrum of the first intermediate frequency band when the third segment from the bottom is received. The hatched portion is the reception segment, and the reception segment and the upper adjacent video carrier are separated by a certain distance or more, and for example, suppression of 40 dB or more is taken by the BPF 5, and suppression of 20 dB is taken for the lower adjacent audio carrier.
[0026]
FIG. 7A shows the characteristics of the baseband signal and the LPFs 61 and 62 when the third segment from the bottom is received. Since the cut-off of the LPFs 61 and 62 is lower than the frequency of the lower adjacent audio carrier wave, the level is sufficiently suppressed. The LPFs 61 and 62 can obtain 20 dB of suppression, and the required suppression amount (40 dB) can be obtained together with the suppression amount 20 dB of BPF5.
[0027]
FIG. 6B shows the spectrum of the first intermediate frequency band when the first segment from the top is received. The hatched portion is the received segment, and the received segment and the lower adjacent audio carrier are separated by a certain distance or more, and for example, suppression of 40 dB can be obtained by the BPF 5, but since the frequency is close to the upper adjacent video carrier, only 10 dB can be suppressed. Therefore, it is necessary to secure greater suppression with the LPFs 61 and 62.
[0028]
FIG. 7B shows the characteristics of the baseband signal and the LPFs 61 and 62 when the first segment from the top is received. As described above, since the suppression of the upper adjacent video carrier of BPF 5 is insufficient at 10 dB, it is necessary to supplement the characteristics with a low-pass filter, and the frequency of the reception segment and the carrier is closer than in FIG. The level of the upper adjacent video carrier is suppressed by controlling the cutoff of the LPFs 61 and 62 from the microprocessor. The setting point of the cutoff frequency in this case will be described. The lower the cut-off frequency, the more the carrier level can be suppressed, but the receiver segment undergoes modifications such as the spectrum of the received segment being deleted. The performance can be improved by setting the cut-off frequency so that the performance improvement due to the suppression of the level of the upper adjacent video carrier is greater than the performance deterioration due to the deformation of the spectrum.
[0029]
FIG. 7B shows a case where control is performed so that the LPFs 61 and 62 can suppress 30 dB or more, and the required value of 40 dB is obtained by combining the suppression amounts of the BPF 5 and LPFs 61 and 62, although the spectrum of some received segments is reduced. Therefore, the performance can be improved as compared with the case of using the characteristics of the LPFs 61 and 62 in FIG.
[0030]
FIG. 6C shows the spectrum of the first intermediate frequency band when the first segment from the bottom is received. The hatched portion is the reception segment, and the reception segment and the upper adjacent video carrier are separated by a certain distance or more, and for example, suppression of 40 dB can be obtained by the BPF 5, but since the frequency is close to the lower adjacent audio carrier, only suppression of 10 dB can be obtained. Therefore, it is necessary to secure greater suppression with the LPFs 61 and 62.
[0031]
FIG. 7B shows the characteristics of the baseband signal and the LPFs 61 and 62 when the first segment from the bottom is received. As described above, since the suppression of the lower adjacent audio carrier of BPF 5 is insufficient at 10 dB, it is necessary to supplement the characteristics with a low-pass filter, and the frequency of the reception segment and the carrier is closer than in FIG. Therefore, the level of the lower adjacent audio carrier is suppressed by controlling the cutoff frequency of the LPFs 61 and 62 from the microprocessor.
[0032]
The setting point of the cutoff frequency in this case will be described. The lower the cut-off frequency, the more the carrier level can be suppressed, but the receiver segment undergoes modifications such as the spectrum of the received segment being deleted. The performance can be improved by setting the cutoff frequency so that the performance improvement due to the suppression of the level of the lower adjacent audio carrier is greater than the performance deterioration due to the spectrum deformation.
[0033]
FIG. 7C shows a case where control is performed so that the LPFs 61 and 62 can suppress 30 dB or more. Although the spectrum of some received segments is cut, the required value 40 dB is obtained by adding the suppression amounts of the BPF 5 and LPFs 61 and 62 together. Therefore, the performance can be improved as compared with the case of using the characteristics of the LPFs 61 and 62 in FIG.
[0034]
FIG. 2 shows a configuration in which the performance of the BPF 5 is corrected using an OFDM demodulator 38 having general-purpose digital input terminals 65 and 66 in addition to the contents described in FIG. The BPF 5 uses a SAW (surface acoustic wave) filter or the like, but the characteristic variation is large at a high frequency of 1400 MHz. Therefore, in the configuration of FIG. 1, there are cases where sufficient suppression of adjacent carrier waves cannot be obtained or the signal is deleted.
[0035]
For example, when receiving the first segment from the top as shown in FIG. 6B, when the BPF 5 can obtain 10 dB suppression with respect to the upper adjacent video carrier, it is possible to ensure only 8 dB suppression due to variations. Are set to Low and High, respectively. The microprocessor 63 reads the logic of the terminals 65 and 66 and detects that the BPF 5 can ensure only suppression of 8 dB. The microprocessor 63 sets the cut-off frequency of the LPFs 61 and 62 to be lowered, and controls the LPFs 61 and 62 so as to obtain 32 dB of attenuation, so that the suppression of 40 dB can be secured by the BPF 5 and the LPFs 61 and 62.
[0036]
As shown in FIG. 6B, when the first segment from the top is received, the BPF 5 suppresses 10 dB with respect to the upper adjacent video carrier, and 15 dB is suppressed due to variations. In this case, the terminals 65 and 66 are set to High and High, respectively.
[0037]
The microprocessor 63 reads the logic of the terminals 65 and 66 and detects that the BPF 5 has secured 15 dB of excessive suppression.
[0038]
The microprocessor 63 is set so as to increase the cutoff frequency of the LPFs 61 and 62 and is controlled so as to obtain an attenuation of 25 dB by the LPFs 61 and 62, so that the suppression of 40 dB can be secured by the BPF5 and the LPFs 61 and 62.
[0039]
The desired suppression of 40 dB can be ensured without excessively reducing the cut-off frequency of the LPFs 61 and 62 without cutting the spectrum of the received segment more than necessary.
[0040]
As described above, the optimum cutoff frequency of the LPFs 61 and 62 can be set even when the BPF 5 has a manufacturing variation, so that the spectrum of the received segment is not cut more than necessary while obtaining desired carrier suppression.
[0041]
FIG. 3 shows a configuration in which the optimum cut-off frequency of the LPFs 61 and 62 according to the segment position can be obtained even when the LPFs 61 and 62 have manufacturing variations. The frequency of the LPFs 61 and 62 can be adjusted using the output signal of the frequency adjustment circuit 67.
[0042]
However, when this receiving apparatus is built in a small portable terminal, the frequency of the reference clock signal 68 varies from company to company. Therefore, it is considered that the DC voltage correction circuit 64 sets the correction voltage V1 obtained by correcting the control voltages of the LPFs 61 and 62 from the microprocessor 63 in accordance with the frequency of the reference clock signal 68.
[0043]
As described with reference to FIG. 1, since the optimum cutoff frequency of the LPFs 61 and 62 differs depending on the segment position, the optimum cutoff frequency is obtained by setting the conversion voltage V2 by changing the control voltage of the LPFs 61 and 62 according to the segment position. be able to.
[0044]
Since the frequency of the reference clock signal 68 and the reception segment are known, the microprocessor 63 controls the DC voltage correction circuit 64 and superimposes V1 + V2 on the output voltage of the frequency adjustment circuit 67, so that the frequency and segment of the reference clock signal 68 are obtained. Since control can be performed according to the position, an optimum cut-off frequency corresponding to the segment position can be obtained while correcting variations of the LPFs 61 and 62 for different reference clock signals 68.
[0045]
FIG. 4 shows a case where the microprocessor 63 controls the DC voltage correction circuit 64 by setting the logic of the general-purpose digital input terminals 65 and 66 according to the manufacturing variation of the BPF 5 in addition to the configuration of FIG. Even when there is a manufacturing variation of the LPF 61, LPF 62, and BPF 5 with respect to the signal 68, an optimal cutoff frequency can be obtained by controlling the cutoff frequencies of the LPF 61 and LPF 62 in accordance with the reception segment position.
[0046]
As described in the description of FIG. 3, the correction voltage V1 corresponding to the frequency of the reference clock signal 68 and the addition voltage (V1 + V2) of the correction voltage V2 corresponding to the segment position are superimposed on the control voltage from the frequency adjustment circuit 67. Thus, the cut-off frequency control of the LPF 61 and the LPF 62 is performed. In addition, when there is manufacturing variation in the BPF 5, the microprocessor 63 further superimposes the correction voltage V3 corresponding to the manufacturing variation by reading the logic of the general-purpose digital input terminals 65 and 66.
[0047]
By superimposing V1 + V2 + V3 on the control voltage from the frequency adjustment circuit 67 as described above, the cut-off frequency control of the LPF 61 and the LPF 62 is performed according to the manufacturing variation of the BPF 5, the LPF 61, 62, the reference clock signal 68, and the segment position. Thus, an optimum cut-off frequency can be obtained.
[0048]
In FIG. 5, a signal indicating the occurrence of bytes that could not be demodulated correctly by the OFDM demodulator 38 is connected to the error rate count circuit 69, and the error rate count circuit 69 is connected to the microprocessor 63.
[0049]
The BPF 5 is a filter that passes one segment in the 1400 MHz band, and is affected by temperature characteristics. In addition, since the strength of the adjacent carrier and the strength of the received segment differ depending on the reception location, the cut-off frequency required for the LPFs 61 and 62 varies depending on the temporal change of the BPF 5 and the reception location although there is a desired value for the worst condition.
[0050]
It is effective to count the error rate in order to perform the optimum cut-off adjustment of the LPFs 61 and 62 with respect to a change with time and a change in reception place.
[0051]
Initially, the microprocessor 63 controls the LPF 61 and LPF 62 so as to obtain a predetermined cutoff frequency according to the position of the segment. For example, when the frequency characteristic of the BPF 5 when receiving the top segment is deteriorated to 5 dB due to the time-dependent change of the frequency characteristic of the BPF 5 to 5 dB, the desired adjacent of the BPF 5 and the LPFs 61 and 62 is combined. If the video carrier suppression amount is 40 dB, the LPFs 61 and 62 can obtain only 30 dB attenuation as described with reference to FIG.
[0052]
The OFDM demodulator 38 performs demodulation processing such as complex Fourier transform, frequency interleaving, time interleaving, and error correction according to the modulation processing at the time of ISDB-T transmission, and outputs TS to the TS output terminal.
[0053]
The OFDM demodulator 38 performs RS decoding (Reed-Solomon decoding) after Viterbi decoding. During RS decoding, it is known that an RS byte that cannot be corrected for errors can be detected. When an RS byte is generated, the OFDM demodulator 38 sends a signal indicating that an error-correctable byte has been generated to an error rate counting circuit 69. Send to.
[0054]
The error rate counting circuit 69 calculates the error rate by counting the number within a predetermined time and informs the microprocessor 63 of the result. The microprocessor 63 does nothing if the calculated error rate is lower than the desired error rate, but if it is high, the microprocessor 63 performs control so that the cut-off frequencies of the LPFs 61 and 62 are sequentially changed from the initial values.
[0055]
The cut-off frequency is changed until the error rate calculated in the process of sequentially changing to steps becomes lower than the desired error rate. By performing such control, not only can a low bit error be obtained even when the BPF 5 changes over time, but also a low error rate can be obtained at the same time in response to the LPF 61 and LPF 62 over time, and even in locations where the adjacent carrier is strong. be able to.
[0056]
An example of a step-like change is shown using the cut-off frequency and error rate diagram of FIG.
[0057]
When there is no secular change, the cut-off frequency is at the point indicated by 105, and since it is lower than the desired error rate, no further control from the microprocessor 63 is required. When the initial value becomes 101 due to secular change, for example, if control is performed so as to change the step from 101 → 103 → 102 → 104, the error rate will be lower than the desired error rate when 104 is reached, so it is changed so far. As a result, the error rate can be made lower than the desired error rate.
[0058]
Analog broadcasting is abolished in the VHF band, and channel expansion such as another digital audio broadcasting is considered. Therefore, the required adjacent exclusion is expected to change.
[0059]
The receiving apparatus having the configuration shown in FIGS. 1 to 4 can easily cope with a change in the situation by changing the software of the microprocessor 63.
[0060]
In the case of FIG. 5, since the cut-off frequency is further changed depending on the error rate, it is not necessary to change the software.
[0061]
Furthermore, even when the level of the adjacent carrier wave is different from the standard depending on the region or channel, in the example of FIGS. 1 to 4, it can be flexibly changed by changing the software.
[0062]
【The invention's effect】
As described above, according to the present invention, in a demodulating apparatus that performs partial reception of an ISDB-T modulated signal, a receiving apparatus that can sufficiently obtain adjacent exclusion capability without increasing the order of the filter and regardless of the position of the received segment. The receiver can be miniaturized.
[Brief description of the drawings]
FIG. 1 is a block diagram of a receiving apparatus according to an embodiment of the present invention. FIG. 2 is a block diagram of a receiving apparatus according to another example of the present invention. FIG. 4 is a block diagram of a receiver according to the present invention. FIG. 5 is a block diagram of a receiver according to another example of the present invention. FIG. 6 is a spectrum diagram of the first intermediate frequency band according to the present invention. 7A to 7C are spectrum diagrams of the baseband frequency band of the present invention. FIG. 8 is a diagram of a cutoff frequency and an error rate of the present invention. FIG. 9 is a block diagram illustrating a configuration example of a conventional receiving apparatus. FIGS. 10A to 10D are spectrum diagrams for explaining a conventional receiving apparatus.
1 Antenna 2 RF amplifier 3, 7, 8 Mixer 4, 9 PLL
5 BPF
6 IF amplifier 10 Phase shifter 21 Transport stream output terminal (TS output terminal)
30, 31 Baseband amplifier (BB amplifier)
32, 33 AD converter (ADC)
34 Complex multiplier 35 Numerically controlled oscillator (NCO)
36,37 Digital LPF
38 OFDM demodulator 61, 62 Low pass filter (LPF)

Claims (5)

A receiving device including a mixer circuit that converts a frequency of a received signal and a filter that removes an unnecessary signal from the frequency-converted received signal. A receiving device comprising means for changing a cutoff frequency of a filter in accordance with a position of a segment to be selected and received. 2. A first filter and a second filter for removing unnecessary signals from the frequency-converted received signal, and means for changing a cutoff frequency of the second filter in accordance with the characteristics of the first filter. The receiving device described in 1. A frequency adjustment circuit connected to the reference clock and a DC voltage correction circuit connected to the frequency adjustment circuit, and the DC voltage correction circuit for the different reference clock signal frequencies are provided with means for making the cutoff frequency of the filter constant. The receiving device according to claim 1. 2. The receiving apparatus according to claim 1, further comprising a DC voltage correction circuit between the second filter and the frequency adjustment circuit, and means for correcting DC voltage with respect to different reference clock signal frequencies. The receiving apparatus according to claim 1, further comprising means for changing a cutoff frequency in accordance with a bit error rate.

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