JP3676944B2 - Control device for synchronous motor - Google Patents

Description

となる。すなわち出力トルクＴはトルク指令Ｔq*に比例する。次にＴq*＞Ｔrefの場合は数１、数２、数８、数９より、
【数１１】

となる。すなわち、この場合においても出力トルクＴはトルク指令Ｔq*に対して線形であるので、適切なＴref、Ｋd1を選定すれば正確な出力トルクを得ることができる。
【００１２】
次に本発明に基づく高精度なトルク制御方式において、定出力制御を行う方式について説明する。モータの回転速度をＮ、基底回転速度をＮbとするとき、Ｎ≧ＮbかつＴq*≦Ｔrefの場合は、ｄ軸界磁弱め電流定数をＫd0として、
【数１２】
ｉq*＝Ｋq1×Ｔq*×Ｎb÷Ｎ
【数１３】
ｉd*＝Ｋd0×（Ｎ−Ｎb）
なる演算によって電流指令を算出する。数１３は磁石による磁束を打ち消すための界磁弱め電流である。またＮ≧ＮbかつＴq*＞Ｔrefの場合は、
【数１４】
ｉq*＝Ｋq1×Ｔref×Ｎb÷Ｎ
【数１５】
ｉd*＝Ｋd0×（Ｎ−Ｎb）＋Ｋd1×（Ｔq*−Ｔref）
なる演算によって電流指令を算出する。
【００１３】
上述した数６、数７、数８、数９に基づくｄ軸電流指令ｉd*、ｑ軸電流指令ｉq*によって出力トルクＴを高精度に制御することができるが、その反面においてモータの端子電圧が比較的高くなり、限られたインバータ６の出力電圧範囲の中においては最大出力が確保できない場合がある。そこで図１の実施形態において、制御モード切換手段１３は外部より指定されたモード信号に従って、当該モード信号が高精度モードである場合には第１の電流指令算出部１２の算出したｉd*、ｉq*をｄｑ軸電圧指令算出部１０に対して出力し、また高精度モードでない場合には第２の電流指令算出部１４が算出したｉd*、ｉq*をｄｑ軸電圧指令算出部１０に対して出力している。ここで第２の電流指令算出部１４におけるｉd*、ｉq*の算出方法は従来の一般的な埋め込み磁石型同期電動機の制御装置と同様であり、ｑ軸電流定数をＫq2、ｄ軸電流定数をＫd2として次式によって算出する。これらＫq2、Ｋd2は一般的に最も大きなトルクを発生するように算出された係数である。
【数１６】
ｉq*＝Ｋq2×Ｔq*
【数１７】
ｉd*＝Ｋd2×Ｔq*
このように出力トルクの精度要求が比較的低い場合には、モード信号によってｄ軸電流指令ｉd*、ｑ軸電流指令ｉq*の演算方式を切り換えることによって、最も小さな電流値で効率よくトルクを出力できるため、モータの巻線抵抗によって発生する銅損を最小にすることができ、モータの発熱を低く抑えることができる。
【００１４】
図１におけるそのほかの構成要素の動作を説明する。ｄｑ軸電圧指令算出部１０では入力された電流指令ｉd*、ｉq*および電流検出値ｉd、ｉq、ロータ回転速度Ｎを元にｄ軸電圧指令値Ｅd*、ｑ軸電圧指令値Ｅq*を算出しており、その内部構成を図２に示す。
減算器１６はｄ軸電流指令ｉd*とｄ軸電流検出値ｉdとを減算してｄ軸電流誤差を算出している。Ｐｉ演算増幅器１７はｄ軸電流誤差を比例積分演算によってｄ軸電圧指令値を算出しているが、その出力には加算器１８および１９によって電流指令ｉd*およびｉq*から算出したｄ軸電圧の予測値をフィードフォワード補償している。同様に減算器２０ではｑ軸電流指令ｉq*とｑ軸電流検出値ｉqを減算してｑ軸電流誤差を求め、これをＰｉ演算増幅器２１によって増幅した結果に、加算器２２、２３および２４においてｑ軸電圧の予測値をフィードフォワード補償している。なお、ｑ軸電圧には磁石による速度誘起電圧成分も含まれているので乗算器２５において誘起電圧定数Ｋemを回転速度Ｎに乗算して速度誘起電圧を予測演算している。これらフィードフォワード補償値のｄ軸成分をＥdff、ｑ軸成分をＥqffとすると、これらの演算は巻線抵抗をＲ1、ｄ軸電流に対する巻線インダクタンスをＬd、ｑ軸電流に対するインダクタンスをＬqとして次式によって算出する。
【数１８】
Ｅdff＝Ｒ1・ｉd*＋Ｎ・Ｌq・ｉq*
【数１９】
Ｅqff＝Ｒ1・ｉq*−Ｎ・Ｌd・ｉd*＋Ｎ・Ｋem
【発明の効果】
本発明による同期電動機の制御装置においては、入力されたトルク指令値に対してモータの出力トルクを線形に制御できるため、常に高精度なトルク制御が可能となり、かつ複雑な演算処理を必要としないため、安価で信頼性の高い制御装置を実現することができる。この結果、位置制御を行う場合に位置誤差の発生を極小にできるので、工作機械の主軸における同期タップなどの用途において加工精度を向上できる効果がある。
【図面の簡単な説明】
【図１】 本発明による同期電動機の一実施形態を示すブロック図である。
【図２】 ｄｑ軸電圧指令算出部の動作を示すブロック図である。
【図３】 従来の誘導電動機の断面構造図である。
【図４】 本発明の同期電動機の制御装置によって駆動される埋め込み磁石型同期電動機の断面構造図である。
【符号の説明】
１ 二次導体、２ ロータコア、３ 磁石、４ エンコーダ、５ モータ、６インバータ、７ 電流検出器、８ 直流電源、９ ３相２相変換器、１０ ｄｑ軸電圧指令算出部、１１ ２相３相変換器、１２ 第１の電流指令算出部、１３ 制御モード切換手段、１４ 第２の電流指令算出部、１５ 微分器、１６ 減算器、１７ Ｐｉ演算増幅器、１８ 加算器、１９ 加算器、２０ 減算器、２１ Ｐｉ演算増幅器、２２ 加算器、２３ 加算器、２４ 加算器、２５ 乗算器、２６ 乗算器、２７ 乗算器、２８ 乗算器、２９ 乗算器、３０ 乗算器、３１ 乗算器。[0001]
BACKGROUND OF THE INVENTION
The present invention relates to an improvement of a control device for a synchronous motor used for driving a spindle of a machine tool, and in particular, has a structure in which a permanent magnet is embedded in a rotor core, and the torque generated by the magnetomotive force of the magnet and the rotor core The present invention relates to a control apparatus for a synchronous motor in which linearity of an output torque is improved in a control apparatus for an embedded magnet type synchronous motor using a reluctance torque generated by a change in magnetoresistance with respect to a rotation angle.
[0002]
[Prior art]
As an electric motor that drives a spindle of a machine tool, an induction motor that is robust in structure and capable of high-speed rotation is widely used. This induction motor can freely control torque and speed by vector control, and realizes various functions such as acceleration / deceleration operation necessary for machine tools.
[0003]
FIG. 3 shows a cross-sectional structure diagram of a general induction motor. As shown in the figure, the induction motor has a cage-type secondary conductor 1 in the rotor, and generates a rotational torque between the current flowing through the secondary conductor 1 and the magnetic flux interlinked with the rotor core 2. At this time, copper loss occurs in the rotor due to the current flowing through the secondary conductor 1, and the rotor cannot be sufficiently cooled as compared with the stator, so that the temperature rise of the rotor is increased. The stator can be sufficiently cooled by passing cooling air directly through the iron core or by passing a coolant. On the other hand, in order to suppress the temperature rise of the rotor to an appropriate value, it is necessary to increase the outer dimension of the rotor in order to secure a sufficient surface area. As a result, the rotor inertia increases and high-speed acceleration / deceleration cannot be realized. .
Furthermore, in recent machine tools, a built-in motor structure in which a motor rotor is directly mounted on the main spindle of the machine is becoming widespread. In such a structure, there is a problem that mechanical displacement is generated due to a rise in the temperature of the rotor, and it is expected to employ an electric motor instead of an induction motor.
[0004]
In order to solve these problems, recently, there is a conventional technique in which a synchronous motor having no secondary conductor, that is, a heat source, is used for driving a spindle of a machine tool. In particular, an embedded magnet type synchronous motor having a cross-sectional structure as shown in FIG. 4 is a system in which a permanent magnet 3 is embedded in the rotor core 2, and since the leakage inductance is large, the speed induced voltage by the magnet 3 is applied to the stator winding. It can be adjusted by the energized current, and field weakening control can be easily realized, so that it is easy to realize constant output characteristics required for the spindle of the machine tool. In addition, since the magnetic resistance of the rotor core 2 has a property that differs greatly in the same direction (d-axis direction) as the magnetic flux by the magnet and in the direction orthogonal to the magnetic flux (q-axis direction) by the magnet, Reluctance torque is generated. Since a torque obtained by adding and combining the reluctance torque and the torque generated by the magnet 3 can be output, a large output torque can be obtained with a small magnet volume. Due to these characteristics, embedded magnet type synchronous motors are beginning to be used in place of conventional induction motors for driving the spindle of machine tools.
[0005]
The embedded magnet type synchronous motor achieves a large output torque by using both the magnetomotive force of the magnet and the reluctance torque. Of these, the torque Tm due to the magnetomotive force of the magnet is expressed by the following equation, where P is the number of pole pairs, Kem is the induced voltage constant, and iq is the q-axis current component orthogonal to the magnetic flux.
Tm = P ・ Kem ・ iq (1)
That is, Tm is proportional to the q-axis current. The reluctance torque Tr is expressed by the following equation, where Ld is the inductance in the d-axis direction of the rotor, Lq is the inductance in the q-axis direction, and id is the d-axis current component.
Tr = P (Lq−Ld) id · iq (2)
In a conventional control device, id is generally controlled based on the following equation using a predetermined ratio α with respect to iq.
id = α · iq (3)
As a result, the reluctance torque Tr in the above equation (2) is expressed by the following equation, and the relationship with the amplitude of the q-axis current iq is nonlinear.
Tr = P · ^α (Lq−Ld) iq2 (4)
Since the output torque of the embedded magnet type synchronous motor is a combined torque of the Tr and the Tm, the reluctance torque Tr becomes non-linear with respect to the amplitude of the current in this way, so that the combined torque is also non-linear. For this reason, when the current amplitude is simply determined with respect to the input torque command, the relationship between the torque command value and the output torque becomes nonlinear.
[0006]
In order to solve the problem that the relationship between the torque command value and the output torque becomes nonlinear as described above, in Japanese Patent Laid-Open Nos. 9-149700 and 10-243679, the output torque is linearly set with respect to the torque command value. There has been proposed a control device that calculates a current command value to be performed. However, these conventional techniques require a square root calculation or a correction calculation using a memory in calculating the current command value, and the calculation process is complicated.
[0007]
[Problems to be solved by the invention]
As described above, in the embedded magnet type synchronous motor, since the reluctance torque is non-linear with respect to the current amplitude, a conventional control device that simply generates a current command value proportional to the input torque command value. Thus, it is not possible to obtain a linear output torque with respect to the torque command value. As a result, when performing highly accurate position control required for the machine tool spindle, such as a synchronous tap, there is a problem that the position accuracy during operation deteriorates and machining accuracy cannot be secured. In addition, if complicated calculation processing is performed to obtain a linear output torque with respect to the torque command value, there is a problem that the control device becomes expensive.
[0008]
[Means for Solving the Problems]
In order to solve the above-described problem, in the present invention, synchronous motor control using a combination of a torque generated by a magnetomotive force of a magnet embedded in the rotor core and a reluctance torque using a change in magnetoresistance with respect to the rotation angle of the rotor core. In the device
The three-phase alternating current detection value and commutation electrical angle command value of the three-phase alternating current flowing through the stator winding are input, and d is a component that generates a magnetic flux synchronized with the magnetic pole position in the three-phase alternating current detection value. A three-phase to two-phase converter that outputs an axial current detection value and a q-axis current detection value that is a component that generates a magnetic flux having a phase difference of 90 degrees from the magnetic pole position;
Current control means for feedback-controlling an alternating current to be supplied to the stator winding according to a deviation between the d-axis current command value and the detected d-axis current value and a deviation between the q-axis current command value and the q-axis current detected value When,
When the input torque command value Tq * is smaller than the predetermined value Tref,
iq * = Kq1 × Tq *
When the q-axis current command value iq * is calculated from the first coefficient Kq1 and the torque command value Tq * based on the following calculation, and when the torque command value Tq * is larger than the predetermined value Tref,
iq * = Kq1 × Tref
A q-axis current command calculation unit for calculating the q-axis current command value iq * based on the following calculation:
When the input torque command value Tq * is smaller than the predetermined value Tref,
id * = 0
And when the torque command value Tq * is larger than the predetermined value Tref,
id * = Kd1 × (Tq * −Tref)
And a d-axis current command calculation unit that calculates the d-axis current command value id * based on the second coefficient Kd1 and the torque command value Tq *.
In another embodiment of the present invention, as the q-axis current command calculation unit and the d-axis current command calculation unit, when the input torque command value Tq * is smaller than a predetermined value Tref and the rotational speed N of the rotor Is smaller than the base velocity Nb,
iq * = Kq1 × Tq *
The q-axis current command value iq * is calculated from the first coefficient Kq1 and the torque command value Tq * based on the following calculation, and when the torque command value Tq * is smaller than the predetermined value Tref and the rotation When the speed N is larger than the base speed Nb,
iq * = Kq1 × Tq * × Nb ÷ N
When the torque command value Tq * is larger than the predetermined value Tref and the rotational speed N is smaller than the base speed Nb,
iq * = Kq1 × Tref
When the torque command value Tq * is larger than the predetermined value Tref and the rotational speed N is larger than the base speed Nb,
iq * = Kq1 × Tref × Nb ÷ N
A q-axis current command calculation unit that calculates the q-axis current command value iq * based on the calculation, and when the torque command value Tq * is smaller than the predetermined value Tref and the rotational speed N is the base speed Nb. If smaller than
id * = 0
And when the torque command value Tq * is smaller than the predetermined value Tref and the rotational speed N is larger than the base speed Nb,
id * = Kd0 × (N−Nb)
The d-axis current command value id * is calculated from the second coefficient Kd0, the rotation speed N, and the base speed Nb based on the following calculation, and the torque command value Tq * is larger than the predetermined value Tref and When the rotational speed N is smaller than the base speed Nb,
id * = Kd1 × (Tq * −Tref)
The d-axis current command value id * is calculated from the third coefficient Kd1, the torque command value Tq *, and the predetermined value Tref, and the torque command value Tq * is larger than the predetermined value Tref. If the rotational speed N is greater than the base speed Nb,
id * = Kd0 × (N−Nb) + Kd1 × (Tq * −Tref)
And a d-axis current command calculation unit that calculates the d-axis current command value id * based on the following calculation.
Furthermore, in another embodiment of the present invention, there is provided control mode switching means for switching the output of the motor to the high accuracy mode by switching the calculation method of the d-axis current command value and the q-axis current command value. As the q-axis current command calculation unit and the d-axis current command calculation unit, when the input torque command value Tq * is smaller than a predetermined value Tref and the control mode switching means is in the high accuracy mode,
iq * = Kq1 × Tq *
The q-axis current command value iq * is calculated from the first coefficient Kq1 and the torque command value Tq * based on the calculation, and when the torque command value Tq * is larger than the predetermined value Tref and the control mode When the switching means is in the high accuracy mode,
iq * = Kq1 × Tref
The q-axis current command value iq * is calculated based on the following calculation, and when the control mode switching means is not in the high accuracy mode,
iq * = Kq2 × Tq *
A q-axis current command calculating unit that calculates the q-axis current command value iq * based on the second coefficient Kq2 and the torque command value Tq * based on
When the input torque command value Tq * is smaller than the predetermined value Tref and the control mode switching means is in the high accuracy mode,
id * = 0
When the torque command value Tq * is larger than the predetermined value Tref and the control mode switching means is in the high accuracy mode,
id * = Kd1 × (Tq * −Tref)
When the d-axis current command value id * is calculated from the third coefficient Kd1 and the torque command value Tq * based on the following calculation, and when the control mode switching means is not in the high accuracy mode,
id * = Kd2 × Tq *
And a d-axis current command calculation unit that calculates the d-axis current command value id * based on the fourth coefficient Kd2 and the torque command value Tq *.
[0009]
DETAILED DESCRIPTION OF THE INVENTION
FIG. 1 is a block diagram showing an embodiment of a control apparatus for a synchronous motor according to the present invention. The motor 5 in the figure is the above-described embedded magnet type synchronous motor, and is driven by applying a variable voltage and variable frequency three-phase AC voltage by an inverter 6. The inverter 6 controls the voltage applied to the motor 5 in accordance with the three-phase voltage commands eu *, ev * and ew * with the DC power supply 8 as an input.
The three-phase current flowing through the motor 5 is detected by the current detector 7, and the detected values iu, iv, and iw are input, and the three-phase to two-phase converter 9 performs the following calculation to obtain the two-phase signals id and iq. Output.
[Expression 1]
id = iu · sin θ + iv · sin (θ−120 °) + iw · sin (θ + 120 °)
[Expression 2]
iq = iu · cos θ + iv · cos (θ−120 °) + iw · cos (θ + 120 °)
Here, θ is an electrical angle obtained by multiplying the rotation angle of the rotor by the number of pole pairs (1/2 of the number of poles), and the position of the magnetic flux generated by the magnet is 0 degree. The id obtained in this way is a direct current amount representing a component that generates a magnetic flux synchronized with the magnetic pole position in the alternating current flowing through the stator winding, and is referred to as a d-axis current detection value. Iq is a direct current amount representing a component that generates a magnetic flux having a phase difference of 90 degrees from the magnetic pole position, and is referred to as a q-axis current detection value.
These id and iq are input to the dq-axis voltage command calculation unit 10, and the d-axis voltage commands ed * and eq * are calculated according to the deviation from the respective command values id * and iq *. Further, ed * and eq * are converted into three-phase AC voltage commands eu *, ev *, and ew * by the following calculation in the two-phase / three-phase converter 11.
[Equation 3]
eu * = ed * · sinθ + eq * · cosθ
[Expression 4]
ev * = ed * · sin (θ−120 °) + eq * · cos (θ−120 °)
[Equation 5]
ew * = ed * · sin (θ + 120 °) + eq * · cos (θ + 120 °)
By distributing the three-phase alternating current that is passed through the stator in this way to the d-axis current and the q-axis current and performing feedback control, the torque generated by the motor is determined by the d-axis current command value id * and the q-axis current command value. It becomes possible to arbitrarily control with iq *.
[0010]
A torque command Tq * is input as an external input command to the synchronous motor control device of FIG. In response to this torque command Tq *, the first current command calculation unit 12 compares a predetermined torque command threshold value Tref with Tq * and, based on the comparison result, the d-axis current command value id as follows. *, Q-axis current command iq * is calculated. That is, when the torque command Tq * is smaller than Tref, the q-axis current constant is set to Kq1,
[Formula 6]
iq * = Kq1 × Tq *
[Expression 7]
id * = 0
And calculate. When the torque command Tq * is larger than Tref, the d-axis current constant is Kd1,
[Equation 8]
iq * = Kq1 × Tref
[Equation 9]
id * = Kd1 × (Tq * −Tref)
And calculate.
[0011]
The output torque T when the d-axis current command id * and the q-axis current command iq * are calculated as described above is expressed as follows. First, in the case of Tq * ≦ Tref, from Equation 1, Equation 2, Equation 6, and Equation 7,
[Expression 10]

It becomes. That is, the output torque T is proportional to the torque command Tq *. Next, when Tq *> Tref, from Equation 1, Equation 2, Equation 8, and Equation 9,
[Expression 11]

It becomes. That is, even in this case, since the output torque T is linear with respect to the torque command Tq *, an accurate output torque can be obtained by selecting appropriate Tref and Kd1.
[0012]
Next, a method for performing constant output control in the highly accurate torque control method based on the present invention will be described. When the rotational speed of the motor is N and the base rotational speed is Nb, if N ≧ Nb and Tq * ≦ Tref, the d-axis field weakening current constant is Kd0,
[Expression 12]
iq * = Kq1 × Tq * × Nb ÷ N
[Formula 13]
id * = Kd0 × (N−Nb)
The current command is calculated by the following calculation. Equation 13 is a field weakening current for canceling the magnetic flux generated by the magnet. If N ≧ Nb and Tq *> Tref,
[Expression 14]
iq * = Kq1 × Tref × Nb ÷ N
[Expression 15]
id * = Kd0 × (N−Nb) + Kd1 × (Tq * −Tref)
The current command is calculated by the following calculation.
[0013]
Although the output torque T can be controlled with high accuracy by the d-axis current command id * and the q-axis current command iq * based on the above-described equations 6, 7, 8, and 9, the terminal voltage of the motor is on the other hand. In some cases, the maximum output cannot be secured within the limited output voltage range of the inverter 6. Therefore, in the embodiment of FIG. 1, the control mode switching means 13 follows the mode signal designated from the outside, and when the mode signal is in the high-accuracy mode, the id *, iq calculated by the first current command calculation unit 12 * Is output to the dq-axis voltage command calculation unit 10, and if not in the high-accuracy mode, id * and iq * calculated by the second current command calculation unit 14 are output to the dq-axis voltage command calculation unit 10. Output. Here, the calculation method of id * and iq * in the second current command calculation unit 14 is the same as that of a conventional general embedded magnet type synchronous motor control device, and the q-axis current constant is Kq2 and the d-axis current constant is Kd2 is calculated by the following equation. These Kq2 and Kd2 are generally coefficients calculated so as to generate the largest torque.
[Expression 16]
iq * = Kq2 × Tq *
[Expression 17]
id * = Kd2 × Tq *
When the output torque accuracy requirement is relatively low, the torque is efficiently output with the smallest current value by switching the calculation method of the d-axis current command id * and the q-axis current command iq * according to the mode signal. Therefore, the copper loss caused by the winding resistance of the motor can be minimized, and the heat generation of the motor can be kept low.
[0014]
The operation of the other components in FIG. 1 will be described. The dq-axis voltage command calculation unit 10 calculates the d-axis voltage command value Ed * and the q-axis voltage command value Eq * based on the input current commands id * and iq *, the current detection values id and iq, and the rotor rotational speed N. The internal structure is shown in FIG.
The subtractor 16 calculates a d-axis current error by subtracting the d-axis current command id * and the detected d-axis current value id. The Pi operational amplifier 17 calculates the d-axis voltage command value by proportional-integral calculation of the d-axis current error, but the output of the d-axis voltage calculated from the current commands id * and iq * by the adders 18 and 19 The predicted value is feedforward compensated. Similarly, the subtracter 20 subtracts the q-axis current command iq * and the q-axis current detection value iq to obtain a q-axis current error, which is amplified by the Pi operational amplifier 21, and is added to the adders 22, 23 and 24. The predicted value of the q-axis voltage is feedforward compensated. Since the q-axis voltage includes a speed-induced voltage component due to the magnet, the multiplier 25 multiplies the rotational speed N by the induced voltage constant Kem to predict the speed-induced voltage. Assuming that the d-axis component of these feedforward compensation values is Edff and the q-axis component is Eqff, these operations are expressed by the following equations, with the winding resistance being R1, the winding inductance for the d-axis current being Ld, and the inductance for the q-axis current being Lq. Calculated by
[Expression 18]
Edff = R1 ・ id * + N ・ Lq ・ iq *
[Equation 19]
Eqff = R1 ・ iq * −N ・ Ld ・ id * + N ・ Kem
【The invention's effect】
In the synchronous motor control apparatus according to the present invention, since the motor output torque can be controlled linearly with respect to the input torque command value, high-accuracy torque control is always possible, and complicated calculation processing is not required. Therefore, an inexpensive and highly reliable control device can be realized. As a result, the occurrence of a position error can be minimized when performing position control, and there is an effect that machining accuracy can be improved in applications such as a synchronous tap on the spindle of a machine tool.
[Brief description of the drawings]
FIG. 1 is a block diagram showing an embodiment of a synchronous motor according to the present invention.
FIG. 2 is a block diagram showing an operation of a dq-axis voltage command calculation unit.
FIG. 3 is a sectional structural view of a conventional induction motor.
FIG. 4 is a sectional structural view of an embedded magnet type synchronous motor driven by the synchronous motor control device of the present invention.
[Explanation of symbols]
DESCRIPTION OF SYMBOLS 1 Secondary conductor, 2 Rotor core, 3 Magnet, 4 Encoder, 5 Motor, 6 inverter, 7 Current detector, 8 DC power supply, 9 3 phase 2 phase converter, 10 dq axis voltage command calculation part, 11 2 phase 3 phase Converter, 12 First current command calculation unit, 13 Control mode switching means, 14 Second current command calculation unit, 15 Differentiator, 16 Subtractor, 17 Pi operational amplifier, 18 Adder, 19 Adder, 20 Subtraction 21 Pi operational amplifier, 22 adder, 23 adder, 24 adder, 25 multiplier, 26 multiplier, 27 multiplier, 28 multiplier, 29 multiplier, 30 multiplier, 31 multiplier.

Claims (3)

In the control device for a synchronous motor using both the torque generated by the magnetomotive force of the magnet embedded in the rotor core and the reluctance torque using the change in magnetoresistance with respect to the rotation angle of the rotor core,
The three-phase alternating current detection value and commutation electrical angle command value of the three-phase alternating current to be passed through the stator winding are input, and d is a component that generates a magnetic flux synchronized with the magnetic pole position in the three-phase alternating current detection value. A three-phase to two-phase converter that outputs an axial current detection value and a q-axis current detection value that is a component that generates a magnetic flux having a phase difference of 90 degrees from the magnetic pole position;
Current control means for feedback-controlling an alternating current to be supplied to the stator winding according to a deviation between the d-axis current command value and the detected d-axis current value and a deviation between the q-axis current command value and the q-axis current detected value When,
If the input torque command value Tq * is smaller than the predetermined torque command threshold value Tref is
iq * = Kq1 × Tq *
Made to calculate the q-axis current command value iq * first coefficient Kq1 based on calculation and by the torque command value Tq *, also when the torque command value Tq * greater than the torque command threshold value Tref is ,
iq * = Kq1 × Tref
A q-axis current command calculation unit for calculating the q-axis current command value iq * based on the following calculation:
If the inputted torque command value Tq * is smaller than the predetermined torque command threshold value Tref is
id * = 0
Based on it becomes operational, and when the torque command value Tq * greater than the torque command threshold value Tref is
id * = Kd1 × (Tq * −Tref)
A d-axis current command calculation unit that calculates the d-axis current command value id * from the second coefficient Kd1 and the torque command value Tq * based on the following calculation :
Equipped with a,
Thereby, the control apparatus of the synchronous motor which improved the linearity of the output torque . In the control device for a synchronous motor using both the torque generated by the magnetomotive force of the magnet embedded in the rotor core and the reluctance torque using the change in magnetoresistance with respect to the rotation angle of the rotor core,
The three-phase alternating current detection value and commutation electrical angle command value of the three-phase alternating current to be passed through the stator winding are input, and d is a component that generates a magnetic flux synchronized with the magnetic pole position in the three-phase alternating current detection value. A three-phase to two-phase converter that outputs an axial current detection value and a q-axis current detection value that is a component that generates a magnetic flux having a phase difference of 90 degrees from the magnetic pole position;
Current control means for feedback-controlling an alternating current to be supplied to the stator winding according to a deviation between the d-axis current command value and the detected d-axis current value and a deviation between the q-axis current command value and the q-axis current detected value When,
When the input torque command value Tq * is smaller than the predetermined value Tref and the rotational speed N of the rotor is smaller than the base speed Nb,
iq * = Kq1 × Tq *
The q-axis current command value iq * is calculated from the first coefficient Kq1 and the torque command value Tq * based on the following calculation, and when the torque command value Tq * is smaller than the predetermined value Tref and the rotation When the speed N is larger than the base speed Nb,
iq * = Kq1 × Tq * × Nb ÷ N
When the torque command value Tq * is larger than the predetermined value Tref and the rotational speed N is smaller than the base speed Nb,
iq * = Kq1 × Tref
When the torque command value Tq * is larger than the predetermined value Tref and the rotational speed N is larger than the base speed Nb,
iq * = Kq1 × Tref × Nb ÷ N
A q-axis current command calculation unit for calculating the q-axis current command value iq * based on the following calculation:
When the torque command value Tq * is smaller than the predetermined value Tref and the rotational speed N is smaller than the base speed Nb,
id * = 0
And when the torque command value Tq * is smaller than the predetermined value Tref and the rotational speed N is larger than the base speed Nb,
id * = Kd0 × (N−Nb)
The d-axis current command value id * is calculated from the second coefficient Kd0, the rotation speed N, and the base speed Nb based on the following calculation, and the torque command value Tq * is larger than the predetermined value Tref and When the rotational speed N is smaller than the base speed Nb,
id * = Kd1 × (Tq * −Tref)
The d-axis current command value id * is calculated from the third coefficient Kd1, the torque command value Tq *, and the predetermined value Tref, and the torque command value Tq * is larger than the predetermined value Tref. If the rotational speed N is greater than the base speed Nb,
id * = Kd0 × (N−Nb) + Kd1 × (Tq * −Tref)
And a d-axis current command calculation unit that calculates the d-axis current command value id * based on the following calculation. In the control device for a synchronous motor using both the torque generated by the magnetomotive force of the magnet embedded in the rotor core and the reluctance torque using the change in magnetoresistance with respect to the rotation angle of the rotor core,
The three-phase alternating current detection value and commutation electrical angle command value of the three-phase alternating current to be passed through the stator winding are input, and d is a component that generates a magnetic flux synchronized with the magnetic pole position in the three-phase alternating current detection value. A three-phase to two-phase converter that outputs an axial current detection value and a q-axis current detection value that is a component that generates a magnetic flux having a phase difference of 90 degrees from the magnetic pole position;
Current control means for feedback-controlling an alternating current to be supplied to the stator winding according to a deviation between the d-axis current command value and the detected d-axis current value and a deviation between the q-axis current command value and the q-axis current detected value When,
Control mode switching means for switching the output of the electric motor to a high-accuracy mode by switching the calculation method of the d-axis current command value and the q-axis current command value;
When the input torque command value Tq * is smaller than the predetermined value Tref and the control mode switching means is in the high accuracy mode,
iq * = Kq1 × Tq *
The q-axis current command value iq * is calculated from the first coefficient Kq1 and the torque command value Tq * based on the calculation, and when the torque command value Tq * is larger than the predetermined value Tref and the control mode When the switching means is in the high accuracy mode,
iq * = Kq1 × Tref
The q-axis current command value iq * is calculated based on the following calculation, and when the control mode switching means is not in the high accuracy mode,
iq * = Kq2 × Tq *
A q-axis current command calculating unit that calculates the q-axis current command value iq * based on the second coefficient Kq2 and the torque command value Tq * based on
When the input torque command value Tq * is smaller than the predetermined value Tref and the control mode switching means is in the high accuracy mode,
id * = 0
When the torque command value Tq * is larger than the predetermined value Tref and the control mode switching means is in the high accuracy mode,
id * = Kd1 × (Tq * −Tref)
When the d-axis current command value id * is calculated from the third coefficient Kd1 and the torque command value Tq * based on the following calculation, and when the control mode switching means is not in the high accuracy mode,
id * = Kd2 × Tq *
And a d-axis current command calculation unit that calculates the d-axis current command value id * from the fourth coefficient Kd2 and the torque command value Tq * based on the calculation.

Images