JP3404880B2 - Inverter device - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an inverter device for converting DC power obtained by rectifying and smoothing an AC power supply into high frequency power and supplying the power to a load.

[0002]

2. Description of the Related Art Conventionally, a chopper circuit has been widely used as a circuit system for reducing a harmonic component of an input current from a commercial AC power supply. However, when a fluorescent lamp or the like is turned on, it is necessary to convert the DC voltage obtained by the chopper circuit into high frequency power again, which causes a problem that the circuit becomes complicated. Therefore, recently, an inverter device (for example, Japanese Patent Application No. 63-235982) in which a switching element of a chopper circuit and an inverter circuit is shared is proposed. FIG. 21 shows the circuit.

The circuit configuration of FIG. 21 will be described below.
It Transistor Q₁, Q₂Is a bipolar transistor
It consists of a star. Transistor Q₁The emitter of Transi
Star Q ₂Connected to the collector. Transistor Q
₁, Q₂The collector and emitter of the diode
D₁, D₂The cathode and anode of
ing. Transistor Q₁Between the base and emitter of
The first rectangular wave signal is input, and the transistor Q₂
The first rectangular wave signal has a high level between the base and emitter of
Low level, the first square wave signal is at a low level.
The second rectangular wave signal, which is high level when
ing. As a result, the transistor Q₁, Q₂Alternate
It is turned on and off. Transistor Q₁The collector of
Iodo D₃Connected to the cathode of diode D₃of
Anode is diode D_FourConnected to the cathode of the die
Aether D_FourIs the transistor Q₂To the emitter of
It is connected. Transistor Q₁The collector of
Indexer C₂Is connected to one end of the capacitor C₂The other end of
Is the capacitor C₃Connected to one end of the capacitor C₃of
The other end is a transistor Q₂Connected to the emitter.

Transistor Q₁, Q₂Connection points and
Sensor C₂, C₃The load circuit R is connected between the connection points of
Has been. Here, in order to simplify the explanation, the load circuit
A resistance element is used as R, but inductive reactance
Or capacitive reactance may be included. Transis
Q₁, Q₂Is connected to one end of the AC power supply Vs.
ing. The other end of the AC power supply Vs has an inductor L₁, L₂
Through the diode D₃, D_FourConnected to the connection point of
There is. Inductor L₁, L₂Connection point and AC power supply Vs
Between the one end, the capacitor C₁Are connected. I
Inductor L₁And capacitor C₁Constitutes the AC filter 3
is doing. Also, the transistor Q₁, Q ₂And diode
D₁, D₂And capacitor C₂, C₃Is the diode D
₃, D_FourAnd inductor L₂With chopper circuit 2
And the inverter circuit 1 together with the load circuit R
is doing.

22 to 25 are circuit diagrams for explaining the operation of the above circuit. First, when the transistor Q ₁ is turned on when the AC power supply Vs has a positive half cycle, as shown in FIG. 22, the inductor L ₂ , the diode D ₃ , and the transistor Q ₁ are used to connect the AC power supply Vs to the inductor L.
_A current flows in ₂ , and the current value increases with a slope proportional to the instantaneous value of the input AC voltage Vin. At this time, the transistor Q ₁ also functions as a switching element for the inverter, and causes a current to flow from the capacitor C ₂ to the load circuit R via the transistor Q ₁ .

Next, when the transistor Q ₁ is turned off, as shown in FIG. 23, the inductor L ₂ , the diode D ₃ ,
Capacitors C ₂ , C ₃ , diode D ₂ , AC power supply Vs
In the path through, the energy in inductor L ₂ is released, charging capacitors C ₂ and C ₃ . At this time, the transistor Q ₂ is turned on, and a current flows through the load circuit R in the direction opposite to the direction shown in FIG. 22 on the path from the capacitor C ₃ to the load circuit R and the transistor Q ₂ .

As described above, in the positive half cycle of the AC power supply Vs, the transistor Q ₁ serves both as the switching element for the chopper and the switching element for the inverter, and the transistor Q ₂ functions only as the switching element for the inverter.

Next, when the transistor Q ₂ is turned on while the AC power supply Vs is in the negative half cycle, as shown in FIG. 24, the AC power supply Vs, the transistor Q ₂ , and the diode D.
₄ , a current flows through the inductor L _{2 along a} path passing through the inductor L ₂ , and the current value increases with a slope proportional to the instantaneous value of the input AC voltage Vin. At this time, the transistor Q ₂ also functions as a switching element for the inverter, from the capacitor C ₃ to the load circuit R and the transistor Q ₂
A current is passed through the load circuit R along a path passing through.

[0009] Next, when the transistor Q ₂ is turned off, as shown in FIG. 25, the AC power source Vs, the diode 1, a capacitor C _2, C _3, diode D _4, a path through the inductor L _2, the inductor L ₂ Energy is released,
Charge capacitors C ₂ and C ₃ . At this time, the transistor Q ₁ is on, and a current flows from the capacitor C ₂ through the transistor Q ₁ to the load circuit R in the direction opposite to the direction shown in FIG.

As described above, in the half cycle in which the AC power source Vs is negative, the transistor Q ₂ functions as a switching element for the chopper and a switching element for the inverter, and the transistor Q ₁ functions only as a switching element for the inverter. To do. The diodes D ₁ and D ₂ in the above circuit charge the capacitors C ₂ and C ₃ to supply a stable smoothed output to the load circuit R. That is, when the diodes D ₁ and D ₂ in FIG. 21 are removed, there is a path for charging the capacitors C ₂ and C ₃ , but there is a path through the load circuit R, so that a stable smoothed output is supplied. However, there is a possibility that the control of the transistors Q ₁ and Q ₂ may be devised and that the impedance value of the load circuit R may be restricted.

In the above circuit, the switching element for the inverter also serves as the switching element for the chopper and is constituted by a small number of elements, which has the advantages of low power loss and a simple circuit configuration. Also,
Each transistor Q ₁ , Q for every half cycle of the AC power supply Vs
_{Since 2} works alternately as a switching element for the chopper, there is an advantage that the stress per switching element is reduced, and the power loss of the switching elements (transistors Q ₁ , Q ₂ ) is balanced. Therefore, for example, the heat dissipation structure may be the same. Furthermore, since the switching elements (transistors Q ₁ and Q ₂ ) also operate as switching elements for the inverter, it is not necessary to separately provide a chopper drive circuit, and the configuration of the drive circuit is simplified.

[0012]

In the circuit of FIG. 21, the drive signals for the transistors Q ₁ and Q ₂ are:
It is common to give mutually inverted rectangular wave signals having the same on-time, but in that case, the conduction period of the chopper current (in the inductance component of the chopper) is divided into a high period and a low period of the input voltage Vin from the AC power supply Vs. The period during which the stored energy is released) differs greatly. That is, near the peak of the input voltage Vin, there is almost no pause period in the chopper current, whereas near the zero crossing of the input voltage Vin, energy is released immediately after the ON period of the transistor that functions as the chopper, and the chopper current pauses. The period becomes longer and, as a result, the input current Ii
There is a problem that n has a distorted waveform as shown in FIG.

When the load circuit shown in FIG. 21 is a fluorescent lamp load, a sinusoidal high-frequency current flows in the load circuit R because it is generally composed of an LC resonance circuit. Therefore, when the transistors Q ₁ and Q ₂ are PWM-controlled only during the period also used for the chopper operation, the transistors Q ₁ and Q ₂ are turned off at the same time, and the operation of the inverter circuit becomes unstable. Therefore, it is necessary to control _{one of} the transistors Q ₁ and Q _{2 so} that one of them is always on.

The present invention has been made in view of the above points, and an object thereof is to perform stable pulse width control while using a switching element for a chopper and a switching element for an inverter as well. The purpose is to make the input current waveform close to a sine wave and not to adversely affect the output of the inverter device.

[0015]

According to the present invention, in order to solve the above problems, as shown in FIG. 1, an AC power supply Vs is switched to obtain a DC voltage in a smoothing capacitor C ₂ . In an inverter device in which at least one switching element is shared by the chopper circuit and the inverter circuit which switches the DC voltage of the smoothing capacitor C ₂ and supplies the high frequency power to the load circuit Z, an AC power source is used.
Means for detecting the instantaneous value of the input voltage from Vs, and
It is characterized in that a control means is provided for changing the ON section of the switching element shared by the chopper circuit and the inverter circuit within a range which does not become zero in accordance with the instantaneous value of the input voltage .

The present invention is not limited to the circuit shown in FIG. 1 in which the two switching elements are shared by the chopper circuit and the inverter circuit, and the switching elements on the shared side are alternately switched depending on the polarity of the AC power supply. As shown in FIG. 8, the present invention can be applied to a circuit in which only one switching element among the two switching elements constituting the inverter circuit is shared with the chopper circuit. Further, the drive system of the inverter device may be a self-excited system or a separately excited system.

[0017]

According to the present invention, the input from the AC power source is thus
A means for detecting the instantaneous value of the input voltage is provided, and a control means for changing the ON section of the switching element shared by the chopper circuit and the inverter circuit according to the detected instantaneous value of the input voltage from the AC power supply is provided. Since it is provided, for example, when the instantaneous voltage value of the AC power supply is high, the ON section of the switching element on the common side does not become zero.
By in short, unstable operation of the inverter circuit
And the output voltage of the chopper circuit,
That is, the input voltage of the inverter circuit can be prevented from rising cyclically, a stable DC voltage can be obtained even with a small-capacity smoothing capacitor, and the output of the inverter circuit can be stabilized. Further, when the instantaneous voltage value of the AC power supply is low, the on-section of the switching element on the shared side can be lengthened to eliminate the pause section of the chopper current, which allows the input current to approach a sine wave. It is possible to reduce the harmonic component of the input current, reduce the input current distortion, and increase the input power factor.

[0018]

1 shows the configuration of a main circuit according to a first embodiment of the present invention. The circuit configuration will be described below. The transistors Q ₁ and Q ₂ are power MOSFETs.
The source of the transistor Q ₁ is connected to the drain of the transistor Q ₂ . The cathodes and anodes of the diodes D ₁ and D ₂ are connected to the drains and sources of the transistors Q ₁ and Q ₂ , respectively. Transistor Q
_{The first} drive signal is input from the terminals a and b of the drive circuit 6 of the control circuit 5 between the gate and source of ₁ and the drive circuit of the control circuit 5 between the gate and source of the transistor Q _2. The second drive signal is input from the terminals c and d of 6. As a result, the transistors Q ₁ and Q ₂ are alternately turned on / off. The drain of the transistor Q ₁ is connected the cathode of a diode D _3, the anode of the diode D ₃ is connected to the cathode of the diode D _4, the anode of the diode D ₄ is connected to the source of the transistor Q _2. The drain of the transistor Q ₁ is connected to one end of the capacitor C _2, the other end of the capacitor C ₂ is connected to the source of the transistor Q _2.

A load circuit Z formed by connecting an inductor L ₃ in series with a parallel circuit of a load R and a capacitor C ₄ is connected to both ends of the transistor Q ₁ via a capacitor C ₃ . The capacitor C ₄ and the inductor L _{3 form} an LC series resonance circuit. Further, the capacitor C ₃ is a coupling capacitor for cutting a DC component, has a sufficiently large capacity as compared with the capacitor C ₄ for resonance, and does not contribute to resonance. The connection point of the transistors Q ₁ and Q ₂ is connected to one end of the AC power supply Vs. The other end of the AC power supply Vs is connected to the connection point of the diodes D ₃ and D ₄ via the inductors L ₁ and L ₂ . A capacitor C ₁ is connected between the connection point of the inductors L ₁ and L ₂ and one end of the AC power supply Vs. The inductor L ₁ and the capacitor C _{1 form} an AC filter. Also, the transistors Q ₁ and Q ₂
The diodes D ₁ and D ₂ and the capacitor C ₂ form a chopper circuit with the diodes D ₃ and D ₄ and the inductor L ₂ , and an inverter circuit with the load circuit Z.

The control circuit 5 and the main circuit are connected via terminals a to g. The voltage of the capacitor C ₂ which is the input voltage of the inverter circuit is divided by the resistors R ₁ and R ₂ and input to the control circuit 5 via the terminal g. The instantaneous voltage of the AC power supply Vs is detected by the voltage detection circuit 4 and input to the oscillation circuit 7 of the control circuit 5 via the terminals e and f. Also, the transistor Q on the high potential side
_A capacitor C ₁₈ is connected to both ends of ₁ via a resistor R ₃ . The voltage of the capacitor C ₁₈ is converted to a constant voltage by the Zener diode ZD and supplied to the drive circuit 6 as the power supply voltage E ₂ . The gates and sources of the transistors Q ₁ and Q ₂ are connected to the drive circuit 6 of the control circuit 5 via terminals a to d.

Next, the configuration of the control circuit 5 is shown in FIGS.
did. The circuit configuration of each unit will be described below. First,
The circuit of FIG. 2 will be described. Current mirror circuit CM
Is a pair of transistors, one of which is
A current proportional to the current flowing in the other transistor
DC power supply E is a circuit configured to flow₁From
Resistance r₁The constant current flowing in the capacitor C_FiveAlso flow
It As a result, the capacitor C_FiveIs linear with constant current
Be charged. TM is a timer circuit, a general-purpose timer
-IC (eg NEC's μPD5555),
Capacitor C_FiveIs the threshold of the timer circuit TM
When the voltage reaches the potential, the capacitor C_FiveStarts discharging.
The charge / discharge waveform at this time is a sawtooth waveform. Conde
Sensor C_FiveThe cycle of charging and discharging is₁And capacitor C_Five
It is determined by the time constant of. Sawtooth wave by the above circuit
The generation circuit 8 is configured. Capacitor C_FiveVoltage
Is the comparator CP₁Is input to the positive input terminal of
It Comparator CP₁AC power source V is connected to the negative input terminal of
full-wave rectify the input voltage from s₂, R₃
The reference voltage divided by is input. Comparator C
P₁Output terminal of the resistor r _FourPulled up by
Inversion circuit N₁Is inverted by and output to the terminal h.
It The ON section determination circuit 9 is configured by the above circuits.
There is.

Next, the circuit of FIG. 3 will be described. on
The output of the section determination circuit 9 is an inverting circuit via a terminal h.
N₂, N₃, N₇Has been entered in. Inversion circuit N₂Out of
Power is capacitor C₇Inversion circuit N via_FourEntered in
There is. Inversion circuit N₃Output is capacitor C₈Anti through
Transfer circuit N_FiveHas been entered in. Inversion circuit N_FourIs often entered
Anti r_FiveDC power supply via₁Is pulled up to the level of
ing. In addition, the inverting circuit N _FiveInput is resistance r₆Through
It is pulled down to ground level. Inversion circuit N_Five
Output of AND circuit A₁Connected to one input
It Inversion circuit N_FourOutput is inverting circuit N₆AND through
Circuit A₁Connected to the other input of. AND circuit A
₁Is output from the terminal j, and the inverting circuit N₇The output of
It is output from the terminal i. Dead by the above circuit
The off-time setting circuit 10 is configured. Dead off
Time is resistance r_Five, Capacitor C₇Time constant and resistance
r₆, Capacitor C₈It is set by the time constant of. Deck
The output signal for which the dead time is set is the AND circuit
A₂, A₃, A_Four, A_Five, AN_Five, AN₆And capacitor
C₉, C_TenThe signal distribution circuit 11 consisting of
Through each transistor Q₁, Q₂Input to the drive circuit 6
Has been done.

Next, the circuit of FIG. 4 will be described. First, the drive circuit of the transistor Q ₁ will be described. When the terminal m becomes High level, a current flows through the light emitting element of the photocoupler PC ₁ through the resistor r _11, and the light receiving element of the photocoupler PC ₁ becomes conductive. As a result, the current flowing from the resistor r ₁₂ to the transistor Q ₁₁ is bypassed, the transistor Q ₁₁ is turned off, the current flows between the base and the emitter of the transistor Q ₁₂ via the resistor r ₁₃ , and the transistor Q ₁₂ is turned on. And the resistances r ₁₄ , r ₁₅ ,
An electric current flows through r ₁₆ . This current causes the resistance r ₁₆
Of the MOS transistor Q ₁ is input between the gate and source of the MOS transistor Q ₁ via the terminals a and b. Next, when the terminal m goes low, the transistor Q ₁₁ turns on, the transistor Q ₁₂ turns off, and the transistor Q ₁₃ turns on. As a result, the charge accumulated between the gate and the source of the MOS transistor Q ₁ is discharged through the diode D ₆ and the transistor Q ₁₃ ,
The MOS transistor Q ₁ is turned off immediately. MOS
The drive circuit of the transistor Q ₂ has the same structure, and when the terminal n becomes High level, the terminals c,
An ON signal is input between the gate and source of the MOS transistor Q ₂ via d. Further, when the terminal n becomes low level, the MOS transistor Q ₂ is quickly turned off.

Next, the circuit of FIG. 5 will be described. Real
In the embodiment, the capacitor C₂Voltage Vc₂Resistance R₁, R
₂The voltage is divided by₂
It is input to the positive input terminal of. Comparator CP₂of
DC power supply E is connected to the negative input terminal. ₁The resistance r₉, R_TenPartial pressure
The constant voltage is input. Comparator CP₂Out of
Force is input to the set input terminal S of the flip-flop FF
Has been done. The flip-flop FF is a general-purpose IC (for example,
For example, it consists of NEC μPD4013) and its reset
The input terminal R has a resistor r₇, R₈And capacitor C₁₁Yo
The power-on reset circuit is connected. this
As a result, the flip-flop FF is reset immediately after the power is turned on.
Is set. With the above circuit, the abnormal voltage protection circuit 12
Is configured. When there is no load or the discharge lamp comes off
When the load is light, the capacitor C is used as the output of the chopper.₂
The voltage of the capacitor C is abnormally boosted, but in this embodiment, the capacitor C
₂When the voltage of exceeds the reference voltage, the comparator CP₂of
The output becomes High level, and the flip-flop FF
Is set and its inverting output terminal goes from High level to L
Change to ow level. As a result, the terminal k
AND circuit AN_Five, AN₆Low level signal is input to
Transistor Q₁, Q₂Drive signals to
It becomes Low level and the oscillation of the inverter circuit is stopped.
The circuit can be protected.

Next, the circuit of FIG. 6 is a circuit diagram of the voltage detection circuit 4 used in the present embodiment, in which the diode bridge DB
₂ , full-wave rectifies the input voltage from the AC power source Vs, and outputs the pulsating voltage to the terminal e. Although the terminal f is not used in this embodiment, the terminal f may be used in the embodiments described later. In the present embodiment, the reference voltage applied to the negative input terminal of the comparator CP ₁ in the ON interval determination circuit 9 for determining the ON interval of the switching element composed of the transistors Q ₁ and Q ₂ is input from the AC power supply Vs. Full-wave rectify the voltage and apply it to resistor r ₂ ,
The voltage is divided by r ₃ . Therefore, the ON section of the shared switching element changes according to the pulsating current voltage obtained by full-wave rectifying the AC power supply Vs. That is, in a period in which the pulsating current is high, the reference potential of the comparator CP ₁ is also high, so that the ON interval of the shared switching element is narrowed and the ON interval of the shared switching element is widened as the pulsating current voltage approaches zero.

Transistors Q ₁ and Q ₂ in this embodiment
FIG. 7 shows the drive signal for the signal. In the figure, A and C are drive signals for the shared switching elements, and B and D are drive signals for the non-shared switching elements. The drive signals A and B are drive signals in the period when the pulsating voltage is high, and the drive signals C and D are drive signals in the period when the pulsating voltage is low.
In the present embodiment, since the sawtooth wave generation cycle of the sawtooth wave generation circuit 8 that determines the oscillation frequency is constant, the switching frequency of the inverter is also constant.

By controlling the switching element as described above, the chopper current rest period is reduced when the input voltage from the AC power source Vs is near zero cross, and it is possible to prevent the chopper current from rising in the vicinity of the peak, and the input current of the chopper current can be prevented. It is possible to reduce harmonic components. Moreover, since the frequency is constant, the output of the discharge lamp, which is the inverter output, is hardly adversely affected, and a stable optical output can be supplied.

Further, the above-mentioned control circuit is the second circuit shown in FIG.
As in the embodiment, it can be applied to a power supply device in which a single switching element is shared by the inverter operation and the chopper operation. In this embodiment, the AC power supply Vs is connected to the AC input terminal of the diode bridge DB ₁ via an AC filter including the inductor L ₁ and the capacitor C ₁ , and the inductor L ₂ is connected to the DC output terminal of the diode bridge DB _1. A series circuit of the transistor Q ₂ is connected. The diode D ₁ is connected across the transistor Q _2.
A capacitor C ₂ is connected via. When the transistor Q ₂ is switched at high frequency, the transistor Q ₂
Energy is stored in inductor L ₂ when ₂ is on, and inductor L ₂ is off when transistor Q ₂ is off.
The energy of ₂ is superimposed on the output of the diode bridge DB ₁ and charged in the capacitor C ₂ via the diode D ₁ . This constitutes a boost chopper circuit. In addition, the transistor Q is connected across the capacitor C _2.
A series circuit of ₁ and Q ₂ is connected, and transistor Q
_A load circuit Z formed by connecting an inductor L ₃ in series with a parallel circuit of a capacitor C ₄ and a load R is connected to both ends of ₁ via a capacitor C ₃ . Each transistor Q ₁ ,
Diodes D ₁ and D ₂ are connected in antiparallel to Q ₂ , respectively. By alternately turning on / off the transistors Q ₁ and Q ₂ , a current flows in the load circuit Z alternately in the opposite direction. This constitutes an inverter circuit. In this circuit, the transistor Q ₂ on the low potential side is shared by the chopper circuit and the inverter circuit, and the transistor Q ₁ on the high potential side is used only in the inverter circuit. The present invention can also be applied to such a power supply device of the one-stone switching element sharing type.

Next, FIG. 9 shows drive signal waveforms of the transistors Q ₁ and Q ₂ in the third embodiment of the present invention. In the figure,
A and C are drive signals for shared switching elements, and B and D are drive signals for non-shared switching elements. The drive signals A and B are drive signals in the period when the pulsating voltage is high, and the drive signals C and D are drive signals in the period when the pulsating voltage is low. In this embodiment, for example, in the main circuit of FIG. 1, the ON section of the shared switching element is short in the section where the pulsating voltage is high and long in the section where the pulsating voltage is low, and the ON duty (the ON section in one cycle is The ratio) is controlled to be approximately 50%, and the ON period of the switching element on the non-shared side is controlled so that the switching frequency becomes constant.
That is, in the section where the pulsating voltage is low, the ON sections of the transistors Q ₁ and Q ₂ are controlled to be equal.

FIG. 10 shows the voltage detection circuit 4 used in this embodiment.
Shows the configuration of. In this circuit, the input voltage from the AC power source Vs is full-wave rectified by the diode bridge DB ₂ , and the full-wave rectification is performed by the capacitors C ₁₂ , C ₁₃ , diodes D ₉ , D ₁₀ , D.
_It is converted into a ripple voltage having a smoothing ratio of about 50% by a partial smoothing circuit consisting of _11, and is divided by resistors r ₂ and r ₃ via a terminal e to be input to the negative input terminal of the comparator CP ₁ of the ON section determining circuit 9. You are typing. Therefore, the ON section of the shared switching element changes following the partially smoothed reference voltage. That is, near the peak of the reference voltage, the ON section of the shared switching element becomes short, and it becomes almost constant in the smooth portion of the valley portion of the reference voltage. Here, the ratio of the voltage dividing resistors r ₂ and r ₃ is adjusted so that the common switching element (the ratio of the ON section in one cycle) becomes 50% in the smooth portion of the valley where the reference voltage is. Set. Further, the oscillation frequency at this time, that is, the switching frequency of the inverter is constant. By controlling the switching elements like this, the chopper current rest period is reduced near the power supply voltage near zero crossing, and it is possible to prevent the chopper current from rising near the peak and reduce the harmonic components of the input current. it can. Moreover, since the switching frequency is substantially constant, the inverter output is hardly adversely affected. In this embodiment, as shown in the main circuit of FIG. 1, since the abnormal voltage protection circuit is provided, it goes without saying that the circuit can be protected from the abnormal boost of the chopper output voltage. The operation thereof is the same as that of the first embodiment, so the explanation is omitted.
As shown in FIG. 8, the control circuit used in the present embodiment can be applied to a power supply device of the one-stone switching element sharing type.

Next, FIG. 11 is a circuit diagram of a fourth embodiment of the present invention, which is a discharge lamp lighting device using a self-excited inverter circuit. The secondary winding n ₂ wound around the resonance inductor L ₃ of the inverter circuit is connected between the base and the emitter of the transistor Q ₁ , and the transistor Q ₁ is also connected.
Between _two base-emitter, the secondary winding n ₃ of the resonance inductor L ₃ is connected. Both windings n ₂ and n ₃ are wound in opposite directions, and output voltages of opposite polarities are applied between the base and emitter of both transistors Q ₁ and Q ₂ to drive one of them ON. Sometimes the other is turned off. A series circuit of a resonance capacitor C ₄ and a resonance inductor L ₃ is connected between the collector and the emitter of the transistor Q ₁ via a DC component cutting capacitor C ₃ . Resonance capacitor C
A primary winding of an output transformer T, which is an insulating transformer, is connected in parallel as a load R to both ends of ₄ , and a discharge lamp DL made of a fluorescent lamp is connected in parallel to a secondary winding of the output transformer T. The filaments f ₁ and f ₂ of the discharge lamp DL are connected to the two preheating windings of the output transformer T, respectively. The starting means 14 is not particularly limited as long as it is for driving the transistor Q ₂ when the power is turned on. Other configurations are similar to those of the embodiment shown in FIG.

Next, the operation of the circuit shown in FIG. 11 will be described. Well
After the start-up operation, the commercial AC power supply Vs has a negative half
If it is an icle, commercial AC power supply Vs, Transis
Q ₂, Diode D_Four, Inductor L₂, L₁, Commercial
A current flows through the path of the AC power supply Vs. Current value at this time
Increases with a slope proportional to the instantaneous value of the power supply voltage. Well
Transistor Q₂Is the switching of the inverter circuit
Since it also functions as an element, the capacitor C₂, Conden
SA C₃, C_Four, Resonance inductor L₃Primary winding n₁,
Transistor Q₂, Capacitor C₂By the route of Conden
SA C₂The charge of is discharged. At this time, the output tiger
The electric power supplied to the discharge lamp DL via the
Inductor L due to high frequency switching of the data circuit₃
And capacitor C_FourResonance induced by series resonance of
Obtained by pressure. Here, the transistor Q₂Noi
Converter current and chopper current flow in the same direction,
Is in use. Transistor Q₁, Q₂On
・ OFF control is based on inductor L₃Winding n wound around
₂, N₃The feedback voltage induced by
The feedback voltage of the transistor Q₁, Q₂Are turned on at the same time
The oscillating voltage is fed back alternately so that the
It is wound so that the motion continues. Output tiger
Inductor T, discharge lamp DL, inductor forming a series resonance circuit
L₃And capacitor C_FourTo the oscillating current of the oscillating circuit
Than transistor Q₂Off signal, transistor Q₁
ON signal to winding n₂, N₃Given by.

Transistor Q₂Turned off,
Inductor L₂, L₁The energy stored in
Ducta L₂, L₁, Commercial AC power supply Vs, diode
D₁, Capacitor C₂, Diode D_Four, Inductor L
₂, L₁Is discharged in the path of₁Charge
It On the other hand, the resonance inductor L₃The residual energy of
Inductor L₃, Diode D₁, Capacitor C₃,
Indexer C_FourAnd load R, inductor L₃Released by
To be done. Diode D₁Is the inverter current and chopper
Current flows in the same direction and is in a shared state.
Eventually, inductor L ₃Residual energy is released
And next, DC cut capacitor C₃Charge is
Langista Q₁The operation is released via. This
When the inverter current is₃Charge charge
Is used as a power source and a capacitor C₃, Transistor Q₁, Inn
Ducta L₃, Resonance capacitor C_FourAnd load R, conde
Sensor C₃Flow through the path of the capacitor C₃Charge of
Is discharged. On the other hand, the chopper current is the inductor
L₂, L₁, Commercial AC power supply Vs, diode D₁, Con
Densa C₂, Diode D_Four, Inductor L₂, L₁of
It flows in the path and is in the opposite direction to the direction of the inverter current.
Diode D₃And D_FourFrom the connection point of inductor L₃,Both
Swing capacitor C_FourAnd load R, capacitor C₃The route
A shunt of the sneak current is generated. Therefore, this time
The leakage current overlaps with the inverter current and causes resonance noise.
Inductor L₃Produces a characteristic effect on the voltage generated in
You The effect is the peak of the voltage waveform of the commercial AC power supply Vs.
It becomes noticeable near (around the peak value) and the valley of the commercial AC power supply Vs
The department has a very small impact.

That is, in the valley portion of the commercial AC power source Vs, since the energy stored in the resonance inductor L ₃ is small,
The value of the sneak current to the inverter circuit side is small, and the ON width of the transistor Q ₁ that performs only the inverter operation is substantially equal to the ON width of the transistor Q ₂ that is shared by the inverter operation and the chopper operation. At this time,
FIG. 12 shows the waveforms of the voltages V ₁ and V ₂ across the transistors Q ₁ and Q ₂ and their forward currents I ₁ and I ₂ . Further, in the mountain portion of the commercial AC power supply Vs, the energy accumulated in the resonance inductor L ₃ is high, so that the current flowing into the inverter is large and the voltage generated in the resonance inductor L ₃ is high, so that the inverter operation is performed. since the voltage induced in the transistor Q turns n ₂ for oN signal of _one only it is high, so that greatly extend the on width transistors Q _1. FIG. 13 shows the waveforms of the voltages V ₁ and V ₂ across the transistors Q ₁ and Q ₂ and the forward currents I ₁ and I ₂ at this time.

Now, the resonance inductor L₃Winding n₁To
The oscillating current that flows is winding n₂, N₃Transistor Q
₁, Q₂Is fed back to the transistor Q₁Off signal,
Langista Q₂ON signal is given to. By this
Next, the resonance inductor L ₃The stored energy of
Move to the source operation. Transistor Q₂ON signal
Is given, the transistor Q₂Is turned on, the quotient
AC power supply Vs, transistor Q₂, Diode D_Four,
Inductor L₂, L₁, Current in the path of commercial AC power supply Vs
Flows, inductor L₂Magnetic energy is stored in
It On the other hand, the resonance inductor L₃As a power source and
Inductor L₃, Capacitor C_FourAnd load R, capacitor
C₃, Capacitor C₂, Diode D₂, For resonance
Kuta L₃The current flows through the path. In this route, Inver
Current is flywheel diode D₂Flowing through
The chopper current is diode D₂Connected in parallel with
Transistor Q₂Will flow through. others
Therefore, the flywheel current is transistor Q₂Flow to each
It is canceled by the chopper current. Inverter current
Is almost constant, while the chopper current is commercial AC
Because it depends on the instantaneous voltage value of the power supply Vs, a commercial AC power supply
At the valley of Vs, the diode D_FourThe current flowing through is a shadow
It is not affected, but as it approaches the mountainous area of the commercial AC power supply Vs
The diode D _FourThe current flowing through the
Aether D_FourON time is shortened and ON duty is
It will be shortened.

As an effect of the above operation, a commercial AC power source V
In the valley of s, the transistor Q ₁
The ON width of the transistor Q ₂ shared with the chopper operation is substantially equal to keep the frequency constant, and the ON width of the transistor Q ₁ only for the inverter operation becomes long in the mountain portion of the commercial AC power supply Vs. The ON width of the transistor Q ₂ which is shared with the chopper operation becomes short. Therefore, in the end, the operation is performed in the direction in which the frequency becomes constant, the pause section of the input current Iin is eliminated, and an envelope waveform closer to a sine wave is obtained. FIG. 14 shows the waveforms of the input voltage Vin and the input current Iin from the commercial AC power supply Vs in this embodiment.

Next, FIG. 15 is a circuit diagram of a fifth embodiment of the present invention. The circuit configuration will be described below. The AC power supply Vs is full-wave rectified by the diode bridge DB ₁ , and its rectified output is an inductor L ₂ and a transistor Q _2.
Applied to the series circuit of. A smoothing capacitor C ₂ is connected to both ends of the transistor Q ₂ via a diode D ₁ . A series circuit of transistors Q ₁ and Q ₂ is connected to both ends of the capacitor C ₂ . Each of the transistors Q ₁ and Q ₂ has a diode D ₁ and D ₂ respectively.
Are connected in anti-parallel. At both ends of the transistor Q _1, a primary winding n ₁ of an inductor L ₄ for current feedback, an inductor L ₃ for resonance, a fluorescent lamp as a load R, and a capacitor C ₃ for cutting a DC component are connected in series. The circuit is connected. Between the non-power supply side terminals of the filament of the fluorescent lamp,
A resonance capacitor C ₄ is connected in parallel. In this embodiment, the transistor Q ₂ on the low potential side and the diode D ₁ on the high potential side are shared by the chopper circuit and the inverter circuit.

The transformer of the inverter circuit in this embodiment is
Dista Q₁, Q₂Is inductor L_FourInduced by
Voltage applied to the secondary winding n₂, N₃By returning from
Self-excited drive. First, the transistor Q₁Is on
The capacitor C of the time constant circuit at the same time₁₆Is usually
Anti-R₁₆Is charged via the capacitor C₁₆Potential of
Is a Zener diode ZD₁Zener voltage and MOS transistor
Langista Q₇Higher than the sum of the threshold voltages of
Then, the resistance R₁₇Through the MOS transistor Q ₇The game
To the MOS transistor Q₇Is
Turned on, feedback winding n₂To transistor Q₁The ba
The current flowing into the transistor is bypassed and the transistor Q₁
Turn off. Thus, the transistor Q₁Is a time constant
It is turned on only for a certain time set on the road. That is,
Transistor Q₁The ON section of is fixed regardless of the pulsating voltage.
It is fixed. Next, inductor L_FourOscillating current reverses
And secondary winding n₃Through transistor Q₂ON signal
Is sent to the transistor Q ₂Turns on. Also, Tran
Dista Q₂ON signal to the transistor Q ₆What
DC power supply E_FourDriven by
Transistor Q_FiveTurns off. Therefore, the resistance
R₁₂, Capacitor C₁₅Of the time constant circuit composed of
Power is started. Capacitor C₁₅The voltage of the comparator
CP₃It is input to the positive input terminal of. In addition, direct current
Source E_FourResistance R_Ten, R₁₁Reference voltage obtained by dividing by
Is the comparator CP₃Is input to the negative input terminal of
It Comparator CP₃Is the capacitor C₁₅Is based on
When the quasi voltage is exceeded, the transistor Q _FourTurn on
Langista Q₃Turn on. Therefore, the transistor
Q₂ON signal to transistor Q₃Pulled out by
Transistor Q₂Turns off. That is, the transition
Star Q₂Resistance of the ON section of₁₂And capacitor C₁₅Timed
Determined by the number.

In this embodiment, the AC power source Vs
Input voltage is diode bridge DB ₁Pulse full-wave rectified by
The flowing voltage is the resistance R of the voltage detection circuit 4._Five, R₆Partial pressure with
Iodo D₁₂Through the capacitor C of the time constant circuit₁₅Enter
Since it works well, it is shared by the chopper and the inverter.
Transistor Q₂Depending on the magnitude of the pulsating voltage, the ON section of
It is possible to change it. That is, on the non-shared side
Transistor Q₁Works in a certain ON interval and
One shared transistor Q₂Full-wave power supply voltage
The ON section is short near the peak of the rectified pulsating current voltage.
Therefore, the ON section becomes longer near zero.

By controlling the transistors Q ₁ and Q ₂ as described above, the idle period of the chopper current is reduced when the power supply voltage of the AC power supply Vs is near zero cross, and the rise of the chopper current is prevented even near the peak. And the harmonic component of the input current can be reduced. Also,
Although the output on the inverter side changes to some extent according to the pulsating current voltage, stable operation can be maintained.
Further, the present embodiment can be configured with a relatively simple circuit, although the switching operation is similar to that of the fourth embodiment.

Next, FIG. 16 shows drive signal waveforms of the transistors Q ₁ and Q ₂ in this embodiment. In the figure, A and C are drive signals to the shared switching element, and B and
D is a drive signal to the switching element on the non-shared side.
The drive signals A and B are drive signals in the period when the pulsating voltage is high, and the drive signals C and D are drive signals in the period when the pulsating voltage is low. In this embodiment, the ON section of the shared switching element is shortened in the section where the pulsating voltage is high and long in the section where the pulsating voltage is low, and the ON period of the non-shared side switching element is controlled to be constant. It was done.
That is, in the section where the pulsating current voltage is low, the ON sections of the switching elements Q ₁ and Q ₂ are controlled to be equal. This control method can also be applied to the circuit of FIG. 1, FIG. 8 or FIG.

Next, FIG. 17 shows drive signal waveforms of the transistors Q ₁ and Q ₂ in the sixth embodiment of the present invention. In the figure, A and C are drive signals for the shared switching elements, and B and D are drive signals for the non-shared switching elements. The drive signals A and B are drive signals in the period when the pulsating voltage is high, and the drive signals C and D are drive signals in the period when the pulsating voltage is low. In this embodiment, the ON section of the shared switching element is controlled such that it is short in a section where the pulsating voltage is high and long in a section where the pulsating voltage is low. It is controlled so that it is short in the high voltage section and long in the low voltage section. However, in this case, the condition is that the ON section of the shared switching element that changes according to the pulsating voltage is larger than the ON section of the non-shared switching element.

In this embodiment, a circuit as shown in FIG. 18 is used as the voltage detection circuit 4 in the main circuit shown in FIG. 1, and the pulsating current voltage obtained by the diode bridge DB ₂ is passed through the terminal f. Is input to the capacitor C ₅ of the sawtooth wave generation circuit 8, and the other terminal e is connected to the negative input terminal of the comparator CP ₁ of the ON section determination circuit 9 and serves as a reference voltage. Therefore, according to the magnitude of the detected pulsating current voltage, the frequency is controlled to be high when the pulsating current voltage is high, and to be low when the pulsating current voltage is low.

Also in this embodiment, the ON section of the shared switching element changes according to the magnitude of the pulsating current voltage. The ON section is short at the high pulsating voltage and the ON section is low at the low pulsating voltage. Since it is controlled to be long, the harmonic component of the input current is reduced. However, since the switching element on the non-shared side is also controlled in the same manner, the duty change in the ON section of the shared switching element is smaller than that in each of the embodiments. Further, the output of the inverter also has a large frequency change in accordance with the pulsating voltage as compared with the above-described respective embodiments, and the output change also becomes large. It is needless to say that in this embodiment as well, since the abnormal voltage protection circuit is provided, the circuit can be protected from abnormal boosting of the chopper output. Furthermore, the control circuit used in this embodiment is
The present invention can also be applied to the one-stone switching element shared type as shown in FIG.

FIG. 19 is a circuit diagram of the seventh embodiment of the present invention. In the present embodiment, in the self-excited power supply device of the one-stone switching element sharing type, the ON section of the shared switching element is short in the section where the pulsating voltage is high,
The control is performed so that the switching element on the non-shared side is turned on so as to be long in a low section, and is controlled to be long in a section where the pulsating current voltage is high and short in a section where the pulsating current voltage is low. At this time, the drive signals to the transistors Q ₁ and Q ₂ are as shown in FIG. In the figure, A and C are drive signals for the shared switching elements, and B and D are drive signals for the non-shared switching elements. The drive signals A and B are drive signals in the period when the pulsating voltage is high, and the drive signals C and D are drive signals in the period when the pulsating voltage is low.

In this embodiment, the transistor on the non-shared side is
Q₁Is the inductor L₃Feedback winding n₂Controlled only by
To be done. Therefore, the transistor Q₁The ON section of is
Iodo Bridge DB₁The large pulsating current voltage rectified by
Inductor L of the chopper circuit according to the size₂Accumulated in
Energy that is different, transistor Q₁On ward
The interval changes depending on the magnitude of the pulsating voltage. Large pulsating voltage
Transistor Q₁The ON section of becomes longer,
When the pulsating voltage is low, the transistor Q₁On section of
It gets shorter. Also, the transistor Q₂For the operation of
This is the same as the control circuit of the fifth embodiment shown in FIG.
When the flow voltage is large, the transistor Q ₂ON section of
When it becomes short and the pulsating current voltage is small, the transistor Q₂
The ON section of becomes longer.

By controlling the transistors Q ₁ and Q ₂ as described above, the idle section of the chopper current decreases when the input voltage from the AC power supply Vs is near zero cross, and it is possible to prevent the chopper current from rising near the peak. , The harmonic components of the input current can be reduced. In addition, the present embodiment has an advantage that it can be configured with a relatively simple circuit as compared with the separately excited power supply device.

It is to be noted that, by using the drive signal as shown in FIG. 20, the transistors Q ₁ , in the main circuit of FIG.
By separately driving Q ₂ , the ON section of the shared switching element is controlled to be short in the section where the pulsating voltage is high and long in the section where the pulsating voltage is low, and the ON section of the switching element on the non-shared side is controlled. May be controlled to be long in a high pulsating voltage section and short in an ON section in a low pulsating voltage section. By controlling the transistors Q ₁ and Q _{2 in} this way,
When the power supply voltage of the AC power supply Vs is near zero cross, the chopper current rest period is reduced, and near the peak, it is possible to prevent the chopper current from rising and the harmonic component of the input current can be reduced. Since the main circuit of FIG. 1 or 8 is provided with the abnormal voltage protection circuit as shown in FIG. 5, the circuit can be protected from the abnormal boost of the chopper output.

In the above embodiments, the structure of the load circuit is not particularly limited, but a circuit using a transformer as shown in FIG. 11 or a circuit not using a transformer as shown in FIG. 19 may be used. In the description of the embodiment,
As the circuit method, the method in which two switching elements are shared by the chopper operation and the inverter operation and the method in which one switching element is shared are illustrated, and the driving method is a method in which each switching element is driven separately and a self-excited method. The driving method is illustrated, and the control method is shown in FIGS.
6, the method shown in FIGS. 17 and 20 has been illustrated, but it goes without saying that these circuit methods, drive methods, and control methods can be used in any combination.

[0050]

According to the invention described in claim 1 or 2,
Provided with means for detecting the instantaneous value of the input voltage from the AC power supply
In addition, the power supply voltage detected in the ON section of the switching element that shares the chopper operation and the inverter operation.
The operation of the inverter circuit becomes unstable because it is changed within the range that does not become zero according to the instantaneous value of
In addition, there is an advantage that a high input power factor and a low input current distortion factor can be realized since the idle section of the chopper current can be eliminated regardless of the magnitude of the power supply voltage. In particular, according to the invention described in claim 1, since the switching elements of two stones are shared by the chopper operation and the inverter operation, the switching elements on the shared side are alternately switched according to the polarity of the AC power supply. According to the invention of claim 2, since the stress of each switching element can be dispersed, only one switching element is shared by the chopper operation and the inverter operation, so that the control circuit is simplified. The effect is that you can do it.

Further, according to the invention of claim 3, since the switching element for performing the inverter operation is separately driven, there is an effect that a highly accurate control can be performed. According to the invention of claim 4, For example, by self-exciting the switching element that performs the inverter operation, there is an effect that the circuit configuration can be simplified. Further, in any of the inventions, the drive circuit of the switching element also used for the chopper operation can be used also as the drive circuit of the switching element for the inverter operation, so that there is an advantage that the cost can be reduced.

Next, according to the invention of claims 5 to 9, exchange
Provision of means for detecting the instantaneous value of the input voltage from the current source
At the same time, the ON section of the switching element that shares the chopper operation and the inverter operation is narrowed within the range where the detected instantaneous value of the power supply voltage is not zero in the high period, and widened during the low instantaneous value of the power supply voltage. Since it is controlled, the operation of the inverter circuit does not become unstable.
In addition, regardless of the magnitude of the instantaneous value of the power supply voltage, the pause period of the chopper current can be eliminated, the input current can have a sinusoidal waveform, the input current distortion can be reduced, and the input power factor can be reduced. There is an effect that can be raised. Further, in particular, according to the invention of claim 5 or 6,
The constant switching frequency has the advantage of facilitating the design of a filter circuit for removing noise.

Further, when the load circuit is an LC resonance circuit including a discharge lamp as in claim 10, there is an advantage that a highly efficient lighting device can be realized, and further, in claim 11, With an abnormal voltage protection circuit, no load,
Since the oscillation is stopped at the time of a light load such as disconnection of the lamp, there is an effect that the circuit can be protected from an excessive boost of the chopper circuit.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a first embodiment of the present invention.

FIG. 2 is a circuit diagram of an oscillation circuit section of a control circuit used in the first embodiment of the present invention.

FIG. 3 is a circuit diagram of a logic circuit section of a control circuit used in the first embodiment of the present invention.

FIG. 4 is a circuit diagram of a drive circuit used in the first embodiment of the present invention.

FIG. 5 is a circuit diagram of an abnormal voltage protection circuit used in the first embodiment of the present invention.

FIG. 6 is a circuit diagram of a voltage detection circuit used in the first embodiment of the present invention.

FIG. 7 is an operation waveform diagram of the first embodiment of the present invention.

FIG. 8 is a circuit diagram of a second embodiment of the present invention.

FIG. 9 is an operation waveform diagram of the third embodiment of the present invention.

FIG. 10 is a circuit diagram of a voltage detection circuit used in a third embodiment of the present invention.

FIG. 11 is a circuit diagram of a fourth embodiment of the present invention.

FIG. 12 is an operation waveform diagram in the valley portion of the pulsating current voltage according to the fourth embodiment of the present invention.

FIG. 13 is an operation waveform diagram in the peak portion of the pulsating current voltage according to the fourth embodiment of the present invention.

FIG. 14 is a waveform chart of an input voltage and an input current in the fourth embodiment of the present invention.

FIG. 15 is a circuit diagram of a fifth embodiment of the present invention.

FIG. 16 is an operation waveform diagram of the fifth embodiment of the present invention.

FIG. 17 is an operation waveform diagram of the sixth embodiment of the present invention.

FIG. 18 is a circuit diagram of a voltage detection circuit used in a sixth embodiment of the present invention.

FIG. 19 is a circuit diagram of a seventh embodiment of the present invention.

FIG. 20 is an operation waveform diagram of the seventh embodiment of the present invention.

FIG. 21 is a circuit diagram of a conventional example.

FIG. 22 is a first circuit diagram for explaining the operation of the conventional example.

FIG. 23 is a second circuit diagram for explaining the operation of the conventional example.

FIG. 24 is a third circuit diagram for explaining the operation of the conventional example.

FIG. 25 is a fourth circuit diagram for explaining the operation of the conventional example.

FIG. 26 is a waveform diagram of an input voltage and an input current in a conventional example.

[Explanation of symbols]

Z load circuit Q ₁ transistor Q ₂ transistor C ₂ capacitor L ₂ inductor 4 voltage detection circuit 5 control circuit 6 drive circuit 7 oscillator circuit

─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Shinkazu Miki 1048 Kadoma, Kadoma City, Osaka Prefecture Matsushita Electric Works, Ltd. (56) References JP-A-2-211065 (JP, A) JP-A-4-8174 (JP, A) JP 5-64460 (JP, A) JP 3-198670 (JP, A) JP 4-137499 (JP, A) (58) Fields investigated (Int. Cl. ⁷⁾ , DB name) H02M 7/48 H05B 41/282

Claims (12)

(57) [Claims] 1. A circuit in which a first switching element and a second switching element, which are alternately turned on and off in the forward direction and which do not block a reverse current, are connected in series so that the forward direction is matched, and a first and a second circuit. A circuit in which two diodes are connected in series so that their forward directions match with each other is connected in parallel so that the forward direction of each diode matches the reverse direction of each switching element, and the first and second switching elements are connected. An AC power source is connected via an inductor between the connection point and the connection point of the first and second diodes, and a first capacitor is connected in parallel to both ends of the series circuit of the first and second switching elements. In an inverter device in which a load circuit and a series circuit of a second capacitor are connected in parallel with the switching element of, the instantaneous value of the input voltage from the AC power supply is detected.
And a control means for changing the ON section of the switching element in which the input current from the AC power supply and the output current to the load circuit simultaneously flow in accordance with the detected instantaneous value of the input voltage within a range that does not become zero. Characteristic inverter device. 2. A series circuit of first and second switching elements is connected in parallel to a first capacitor, and first and second switching elements are connected in parallel.
The first and second diodes are respectively connected in anti-parallel to the switching elements of, the series circuit of the load circuit and the second capacitor is connected in parallel with one of the switching elements, and the DC output of the rectifier for full-wave rectifying the AC power source In an inverter device having means for connecting a second switching element and an inductor series circuit to a terminal and alternately turning on and off the first and second switching elements, an input from an AC power source is used.
A means for detecting the instantaneous value of the input voltage and the detected input voltage
Depending on the instantaneous value of, the ON section of the second switching element
An inverter device comprising a control means for changing the value within a range that does not become zero . 3. The inverter device according to claim 1, wherein the control means includes a self-excited drive circuit. 4. The inverter device according to claim 1, wherein the control means includes a separately-excited drive circuit. 5. A switching device comprising at least one chopper circuit for switching an AC power supply to obtain a DC voltage on a smoothing capacitor and an inverter circuit for switching a DC voltage on the smoothing capacitor to supply high frequency power to a load circuit. Means for detecting the instantaneous value of the input voltage from the AC power supply in the inverter device sharing the same
According to the detected instantaneous value of the input voltage, while keeping the frequency of the switching element shared by the chopper circuit and the inverter circuit substantially constant,
An inverter device comprising means for controlling such that when the instantaneous voltage value of the AC power supply is high in a range where it does not become zero, it is short, and when it is low, it is long. 6. The ON section of the switching element shared by the chopper circuit and the inverter circuit is set to have a length corresponding to about half the switching cycle when the instantaneous voltage value of the AC power supply is near zero. The inverter device according to claim 5, which is characterized in that. 7. A chopper circuit for switching an AC power supply to obtain a DC voltage in a smoothing capacitor, and an inverter circuit for switching the DC voltage of the smoothing capacitor to supply a high frequency power to a load circuit. Means for detecting the instantaneous value of the input voltage from the AC power supply in the inverter device sharing the same
According to the detected instantaneous value of the input voltage, while keeping the OFF section of the switching element shared by the chopper circuit and the inverter circuit substantially constant, the ON section of the AC power supply is maintained within a range that does not become zero . An inverter device comprising means for controlling such that when the instantaneous voltage value is high, it is short, and when it is low, it is long. 8. A at least one switching element comprising a chopper circuit for switching an AC power supply to obtain a DC voltage to a smoothing capacitor and an inverter circuit for switching the DC voltage of the smoothing capacitor to supply a high frequency power to a load circuit. Means for detecting the instantaneous value of the input voltage from the AC power supply in the inverter device sharing the same
According to the detected instantaneous value of the input voltage, short and low when the instantaneous voltage value of the AC power supply is high within a range in which the ON section and the OFF section of the switching element shared by the chopper circuit and the inverter circuit are not zero. An inverter device characterized in that it is provided with means for controlling it so that it sometimes becomes longer. 9. A switching device comprising at least a chopper circuit for switching an AC power supply to obtain a DC voltage in a smoothing capacitor, and an inverter circuit for switching the DC voltage in the smoothing capacitor to supply high frequency power to a load circuit. Means for detecting the instantaneous value of the input voltage from the AC power supply in the inverter device sharing the same
According to the detected instantaneous value of the input voltage, the ON period of the switching element shared by the chopper circuit and the inverter circuit is short when the instantaneous voltage value of the AC power supply is high and long when the instantaneous voltage value of the AC power supply is low, within a range that does not become zero. Inverter device characterized in that it is provided with a means for controlling so as to be long when the instantaneous voltage value of the AC power supply is high and short when the instantaneous voltage value of the AC power supply is low within a range where the off period does not become zero . 10. The inverter device according to claim 1, wherein the load circuit is an LC resonance circuit including a discharge lamp. 11. The abnormal voltage protection circuit for forcibly stopping the switching element of the inverter circuit when the DC voltage obtained in the smoothing capacitor is a predetermined value or more, according to any one of claims 5 to 10. The inverter device according to claim 1. 12. A chopper circuit for switching an AC power supply to obtain a DC voltage to a smoothing capacitor, and an inverter circuit for switching a DC voltage of the smoothing capacitor to supply high frequency power to a load circuit.
Means to detect the instantaneous value of the input voltage from the AC power supply in an inverter device that shares two switching elements
And an inverter device provided with control means for changing the ON section of the switching element shared by the chopper circuit and the inverter circuit within a range not becoming zero according to the detected instantaneous value of the input voltage .