JP3402322B2 - Brushless motor - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a brushless motor which supplies a sinusoidal drive current in synchronization with the rotation of the motor.

[0002]

2. Description of the Related Art In recent years, brushless motors have been widely used which detect the rotational position of a rotor and switch the current to the drive winding to drive the rotation in a predetermined direction.

FIG. 23 shows the structure of a conventional brushless motor. The position detector 502, which is an optical rotary encoder provided coaxially with the rotor magnet 501, has an optical slit 505 provided between the light emitting diodes 503a, 503b, 503c and three sets of photocouplers consisting of the phototransistors 504a, 504b, 504c. Then, the position of the optical slit 505 changes as the rotor magnet 501 rotates, and the phototransistors 504a, 504b,
The output of 504c is changed. Phototransistor 50
4a, 504b, 504c output current is the resistance 507a,
507b and 507c convert into three-phase detection voltage. The comparators 510a, 510b, 510c compare the respective detected voltages with the reference voltage of the reference voltage source 508 to generate three-phase digital signals. Comparator 5
The output digital signals of 10a, 510b, 510c are
Power is amplified by the amplifiers 511a, 511b, and 511c, respectively, and three-phase drive windings 520a, 520b, and 5 are provided.
20c.

The position of the optical slit 505 changes as the rotor magnet 501 rotates, and the comparators 510a and 510a
The output digital signals of 10b and 510c change. As a result, the drive voltage applied to the drive windings 520a, 520b, 520c is switched, and the torque in the predetermined direction is continuously generated.

[0005]

However, the above-mentioned conventional configuration has the following problems.

First, in the conventional structure, since the comparatively simple position detector for detecting the rotational position of the rotor is used, the electric power is supplied to the drive winding by the rectangular wave voltage. As a result, current distortion occurs due to the winding inductance, and the drive torque varies greatly. In addition, current distortion due to digital voltage switching causes motor vibration and noise, which is a serious problem.

Secondly, since the optical rotary encoder used in the position detector uses three sets of light emitting diodes / phototransistors and optical slits, the number of parts and the number of wirings are large, and the manufacturing is extremely complicated. Met. Further, since the position detecting parts arranged near the rotor magnet and the drive winding are used in a bad environment where the temperature is high and there is much dust, it is preferable that the number of parts is as small as possible.

The main object of the present invention is to provide the above-mentioned first prior art.
An object of the present invention is to provide a brushless motor that supplies a sinusoidal drive current using a simple rotation detector. Further, the subordinate object of the present invention is to
The first and second problems described above are solved, and an object thereof is to provide a brushless motor that supplies a sinusoidal drive current using a very simplified rotation detector.

[0009]

In order to achieve this object, a brushless motor having the structure of the present invention is attached to a rotor and uses a magnetic flux generated by a permanent magnet to generate a P pole (here,
P is an even number of 2 or more) and is attached to the stator and a magnetic field portion forming a magnetic field pole 3 which interlinks with the magnetic flux of the magnetic field portion 3
Phase drive winding, three MOS type upper drive transistors, three MOS type lower drive transistors, three upper diodes connected in parallel to each upper drive transistor, and each lower side A power supply unit that includes three lower diodes connected in parallel to the drive transistor and supplies power to the drive windings of three phases, and rotation of the rotor is detected to generate a sinusoidal drive command signal. Drive command means for driving the upper drive transistor and the lower drive transistor of the power supply means in response to the drive command signals of the three phases to generate a sinusoidal three-phase drive current for the three phases. A brushless motor comprising: a drive unit that supplies the drive winding; wherein the drive command unit is a rotation detection unit that obtains a pulse signal that changes in synchronization with the rotation of the rotor; A time measuring means for measuring a timing interval synchronized with the rotation of the rotor by the pulse signal of the detecting means, and an estimated electric power corresponding to the rotational position of the rotor for each time interval shorter than the timing interval of the time measuring means. And a deviation amount of the estimated electrical angle of the electrical angle estimating means with respect to a predetermined electrical angle corresponding to the rotational position of the rotor at the timing of generating the pulse signal of the electrical angle estimating means and the rotation detecting means. An estimated displacement detection / correction unit that performs a required correction according to the displacement amount each time the electrical angle estimation unit estimates a new estimated electrical angle, and the three-phase drive command corresponding to the estimated electrical angle. Command generating means for generating a signal, and the driving means responds to the three-phase driving command signals generated corresponding to the estimated electrical angle. , The upper drive transistor and MOS of the MOS type of the power supply unit
Shaped lower drive transistor is subjected to pulse width modulation operation at a predetermined frequency, and has a sine wave shape 3 smoothed by the upper drive transistor, the lower drive transistor, the upper diode, the lower diode and the drive winding. It is configured to include means for supplying a three-phase drive winding with a three-phase drive current.

In order to achieve this object, a brushless motor according to another aspect of the present invention is attached to a rotor and uses a magnetic flux generated by a permanent magnet to generate a P pole (here,
P is an even number of 2 or more) and is attached to the stator and a magnetic field portion forming a magnetic field pole 3 which interlinks with the magnetic flux of the magnetic field portion 3
Phase drive winding, three MOS type upper drive transistors, three MOS type lower drive transistors, three upper diodes connected in parallel to each upper drive transistor, and each lower side A power supply means including three lower diodes connected in parallel to the drive transistor for supplying power to the drive windings of three phases; a current command means for creating a two-phase current command signal; The current detecting means for obtaining a two-phase current feedback signal corresponding to the current supplied to the winding, the conversion comparing means for inputting the current command signal and the current feedback signal and performing an error detecting operation, and the conversion comparing means. Drive means for driving and controlling the upper drive transistor and the lower drive transistor of the power supply means in response to a three-phase output signal, and supplying a sinusoidal three-phase drive current to the three-phase drive winding. And A brushless motor that Bei,
However, the conversion comparing means is a rotation detecting means for generating a pulse signal that changes in synchronization with the rotation of the rotor, and a time for measuring a timing interval synchronized with the rotation of the rotor by the pulse signal of the rotation detecting means. Measuring means,
At each time interval shorter than the timing interval of the time measuring means, an electrical angle estimating means for estimating an estimated electrical angle corresponding to the rotational position of the rotor, and a generation timing of the pulse signal of the rotation detecting means, The amount of deviation of the estimated electrical angle of the electrical angle estimating means with respect to a predetermined electrical angle corresponding to the rotational position of the rotor is detected, and the electrical angle estimating means responds to the amount of deviation every time it estimates the new estimated electrical angle. Estimated deviation detection and correction means for performing the required correction,
Conversion feedback means for obtaining a two-phase conversion feedback signal by coordinate-converting the two-phase current feedback signals using the estimated electrical angle;
Control creating means for obtaining a two-phase control signal in response to a comparison result of the two-phase current feedback signal and the two-phase converted feedback signal;
Conversion control creating means for obtaining conversion control signals of three phases by coordinate conversion of the control signals of two phases using the estimated electrical angle;
3 of the conversion comparing means responsive to the conversion control signal of the phase
Output producing means for obtaining the output signal of a phase, the drive means responding to the output signals of the three phases of the output producing means, and the MOS type upper drive transistor of the power supply means. A pulse width modulation operation of the MOS-type lower drive transistor at a predetermined frequency, and the upper drive transistor, the lower drive transistor,
A three-phase sinusoidal drive current smoothed by the upper diode, the lower diode, and the drive winding is set to 3
It is configured to include means for supplying the drive winding of the phase.

With this configuration, the estimated electrical angle corresponding to the rotational position of the rotor is sequentially estimated using only the pulse signal of the rotation detection means, and the drive command signal is created or the coordinate conversion is performed using the estimated electrical angle. Then, a sinusoidal drive current corresponding to the estimated electrical angle is supplied. Further, the deviation amount of the estimated electrical angle is detected at the timing of generation of the pulse signal of the rotation detecting means, and a necessary correction according to the deviation amount is performed every time the electrical angle estimating means estimates the new estimated electrical angle. As a result, even if a large amount of deviation of the estimated electrical angle occurs, the estimated electrical angle does not change at once due to the correction of the amount of displacement. That is, even if there is a large amount of deviation in the estimated electrical angle at the generation timing of the pulse signal, it is gradually corrected each time a new estimated electrical angle is estimated,
The amount of deviation at the timing of generation of the next pulse signal is reduced. Therefore, regardless of whether the rotor rotation speed is high or low, the electrical angle can be accurately estimated, and the sinusoidal drive current synchronized with the rotor rotation position can be supplied. In particular, since the estimated electrical angle is prevented from being greatly corrected / changed in the operation of correcting the deviation amount of the estimated electrical angle, the disturbance of the drive current is reduced, and the noise / vibration can be significantly reduced. Also, MOS
3 upper drive transistors, 3 lower drive transistors of MOS type, 3 upper diodes connected in parallel to each upper drive transistor, 3 connected in parallel to each lower drive transistor The MOS type upper drive transistor and the MOS type lower drive transistor are pulsed in response to the three-phase drive command signal created corresponding to the estimated electrical angle, using the power supply means including the lower diode. The width modulation control is performed so that a sinusoidal three-phase drive current is supplied to the three-phase drive winding. This makes noise
Vibration can be significantly reduced. As described above, since the sinusoidal drive current is supplied to the drive winding, the drive current changes smoothly and the current distortion due to the winding inductance is significantly reduced. As a result, a uniform drive torque with little torque fluctuation is obtained, the motor is driven to rotate smoothly, and motor vibration and noise are significantly reduced.

Further, the pulse signals of the rotation detecting means are not necessarily required for three phases, and the present invention can be constructed even when one detecting element outputs one pulse signal. As a result, the number of parts provided near the drive winding is significantly reduced, and a simple rotation detecting means can be adopted. The electrical angle described in the present invention is 2 of the field part.
The pole corresponds to 360 degrees.

[0013]

BEST MODE FOR CARRYING OUT THE INVENTION (Embodiment 1) An embodiment of the present invention will be described below with reference to the drawings.

1 to 7 show a brushless motor according to a first embodiment of the present invention. FIG. 2 shows the motor structure of the first embodiment. The rotor permanent magnet 12 that generates the magnetic flux of the four-pole field magnetic pole is fixed to the rotor rotation shaft 10 together with the inner yoke 11 of the ferromagnetic material that forms the inner magnetic path. The permanent magnet 12 has four magnetic poles (N, S, N, S) at equal angular intervals (90 degrees) or approximately equal angular intervals, and an outer yoke 13 of a ferromagnetic material is fixed to the outer peripheral surface side. . The outer yoke 13 forms a magnetic flux path at a position that covers the magnetic pole surface of the permanent magnet 12, and the yoke blocks 13a, 13b, 13c, 13 are formed.
The mechanical connection portion of the four yoke blocks has a very small radial thickness and is magnetically separated due to magnetic saturation. That is, the magnetic flux directly passing through the connecting portion of the yoke block is very small and can be ignored. Rotating shaft 10, inner yoke 11, permanent magnet 12, outer yoke 13
Are integrated to form a rotor, and the permanent magnet 12 of the rotor
A magnetic field having four magnetic poles is formed by using the magnetic flux generated by.

The stator iron core 14 has twelve salient poles arranged at equal angular intervals (30 degrees) or substantially equal angular intervals.
Each drive winding A1, A2, A to wind one salient pole
3, A4, B1, B2, B3, B4, C1, C2, C
The windings C3 and C4 are out of phase with each other. Drive winding A
1, A2, A3, A4 are connected in series so that the current directions are sequentially reversed, and form the first-phase drive winding 20A. Similarly, the drive windings B1, B2, B3, B4 are connected in series so that the current directions are sequentially reversed to form a second-phase drive winding 20B. Furthermore, drive windings C1, C
2, C3, C4 are connected in series so that the current directions are sequentially reversed, and form a third-phase drive winding 20C.

The magnetic flux generated by the permanent magnet 12 of the field magnet portion flows into each salient pole of the stator core 14 through the yoke blocks 13a, 13b, 13c and 13d, and interlinks with each drive winding. Regarding the interlinkage magnetic flux due to the permanent magnets 12, the drive angles of the first phase, the second phase, and the third phase are each 120 electrical degrees.
There is a phase difference of degrees. In this embodiment, the mechanical angle 18
0 degree (the mechanical angle of two poles) corresponds to the electrical angle of 360 degrees.

A detection element 17 is provided on a part of the stator core 14.
Detects the magnetic flux generated by the permanent magnet 12 attached to the rotor and generates an electric signal corresponding to the magnetic flux density. For the detection element 17, for example, a Hall element, a magnetically biased magnetoresistive element, a supersaturated reactor, or the like is used.

FIG. 1 shows the circuit configuration of the brushless motor of the first embodiment. In FIG. 1, 20A, 20B, 20
C is the above-mentioned three-phase drive winding, 21 is a drive command unit, 22 is a drive controller, 23 is a power supply unit, and 24a, 24b, 24.
c is a current detector. The power supply unit 23 includes the upper drive transistors 31a, 31b, 31c and the upper diode 3
2a, 32b, 32c and the lower drive transistor 33a,
33b, 33c and lower diodes 34a, 34b, 34
It is configured to include c. Here, MOS transistors are used as the drive transistors 31a, 31b, 31c, 33a, 33b, and 33c. The drive command unit 21 is configured to include a rotation detector 41, a time measuring unit 42, a drive command creating unit 43 (electrical angle estimator 44 and command creating unit 45), and a controller 46 as necessary. .

The rotation detector 41 of the drive command unit 21 uses the output signal of the detection element 17 to generate a pulse signal g having a frequency proportional to the rotation speed of the rotor. FIG. 3 shows a configuration example of the rotation detector 41. The output signal e of the detection element 17 is
The amplitude is amplified by a low-pass type or band-pass type amplifying circuit 51, and the waveform is shaped by a shaping circuit 52 into a pulse signal g. Since the detection element 17 detects the magnetic flux of the permanent magnet 12 attached to the rotor, the pulse signal g changes in synchronization with the rotation of the rotor, and one pulse is generated by the rotation of two poles. That is, electrical angle 3
One pulse is generated by the rotation of 60 degrees. Also,
The time when the pulse changes corresponds to the time when the detecting element 17 faces the switching position of the magnetic pole of the permanent magnet 12.

The time measuring device 42 receives the output pulse signal g of the rotation detector 41 and measures the generation timing interval of the pulse signal g which changes in inverse proportion to the rotation speed of the rotor. FIG. 4 shows a configuration example of the time measuring device 42. The first differentiating circuit 61 generates a first differentiating pulse having a predetermined time width "H" (high potential state) by using the falling edge of the pulse signal g as a trigger, and the second differentiating circuit 62 produces a falling edge of the first differentiating pulse. A second differential pulse y having a predetermined time width "H" is generated by using the edge as a trigger. The first counter circuit 64 is reset by the generation of the second differential pulse y, and thereafter counts up using the clock pulse ck1 of the first clock circuit 63 as a clock. The count value of the first counter circuit 64 is latched by the first latch circuit 65 at the time of generation of the first differential pulse of the first differentiating circuit 61, and this latched value is output as the output signal f of the time measuring device 42.
As a result, the output signal f of the first latch circuit 65 is a time measurement result corresponding to the generation timing interval of the pulse signal g.

The drive command creating section 43 is composed of an electrical angle estimator 44 and a command creating unit 45, which inputs the measurement result signal f of the time measuring device 42 to estimate the electrical angle corresponding to the rotor rotational position to estimate the estimated electrical power. Three-phase sinusoidal drive command signals ja, jb, jc using angles are output. The amplitudes of the drive command signals ja, jb, and jc are the current command signal j of the controller 46.
It changes in response to q and jd. FIG. 5 shows a configuration example of the electrical angle estimator 44. The multiplication circuit 71 of FIG.
Is multiplied by the correction signal n of the correction coefficient circuit 78 described later (which is almost equal to 1). The second latch circuit 72 uses the multiplication signal of the multiplication circuit 71 as the second differential pulse y.
Latch by the occurrence of. The second counter circuit 74 is
The clock pulse ck2 of the second clock circuit 73 is used as a clock to count down, and when the count value becomes zero, an internal timing signal z (zero detection pulse) having a predetermined time width is output, and when the next clock pulse CK2 arrives. At, the latch value of the second latch circuit 72 is loaded into the second counter circuit 74, and then the countdown operation is continued. The second counter circuit 74 repeats the above operation every time the count value becomes zero, and the second latch circuit 72
Internal timing signal z at a time interval according to the latch value of
(Zero detection pulse) is output. The clock pulse ck2 of the second clock circuit 73 changes a required times faster than the clock pulse ck1 of the first clock circuit 63. In this embodiment,
For the sake of explanation, (clock frequency of ck2) / (clock frequency of ck1) = 12 (the larger the ratio, the better, and 36 or more is preferable). As a result, the second count circuit 74 outputs the internal timing signal z at time intervals of about 1/12 of the detection timing interval of the time measuring device 42 (the generation timing interval of the pulse signal g). The third counter circuit 75 counts up using the internal timing signal z as a clock. When the count value v of the third counter circuit 75 reaches the second set value v2, the second set value detection circuit 77 operates to output the first set value to the third counter circuit 75 at the time of generation of the next internal timing signal z. Circuit 7
The first set value v1 of 6 is loaded. After that, the internal timing signal z is generated to sequentially count up. As a result, the third counter circuit 75 causes the first set value v1
From the second set value v2 to a count value v corresponding to the estimated electrical angle. In this embodiment, for the sake of explanation, v1 = -180 degrees and v2 = 180 degrees- (for one step) = 150 degrees in terms of electrical angle.

The correction coefficient circuit 78 is the third counter circuit 7
The correction signal n is obtained from the count value v of 5. Since the timing of using the correction signal n is the time of generation of the second differential pulse y for operating the second latch circuit 72, the correction signal n at this time will be described. Correction coefficient circuit 78
Sets the correction signal n to 1 when the count value v (electrical angle conversion) of the third counter circuit 75 is equal to zero, and sets the correction value to the negative value k that responds to the ratio to the electrical angle of 360 degrees when the count value v is negative. The correction signal n is set to (1 + k) as
When the count value v is positive, the correction signal n is (1 + k) with the positive value k corresponding to the ratio to the electrical angle of 360 degrees as the correction value.
I have to. As a result, the deviation amount v3 of the count value v (estimated electrical angle) from the predetermined value (zero) at the time of generation of the second differential pulse y is detected, the correction signal n corresponding to the deviation amount v3 is obtained, and the time is measured. The measurement output signal f of the instrument 42 is multiplied and corrected, and stored in the second latch circuit 72 and saved. Since the latch value of the second latch circuit 72 becomes the data for determining the cycle time interval of the second counter circuit 74 (the time interval of generation of the internal timing signal z), the correction signal n of the correction coefficient circuit 78 generates the internal timing signal z. Correcting the time interval.

FIG. 8 shows signal waveforms (represented by analog waveforms) for explaining the operation relations of the main parts of the rotation detector 41, the time measuring device 42, and the electrical angle estimator 44. A magnetic flux detection signal of the rotor permanent magnet 12 by the detection element 17 (see FIG. 8).
The waveform of (a)) is shaped by the rotation detector 41 and output as a pulse signal g (FIG. 8B). Time measuring instrument 4
The first counter circuit 64 of No. 2 digitally measures the timing of occurrence of the falling edge of the pulse signal g (FIG. 8C), and the measurement result is the first latch circuit 65.
Output signal f of The second counter circuit 74 of the electrical angle estimator 44 periodically counts down using the latch value of the second latch circuit 72 in response to the measurement result signal f of the time measuring device 42 as a load value (FIG. 8 (d)). , Each time the count value becomes zero, an internal timing signal z (zero detection pulse) is created. Each time the internal timing signal z is generated, the count value of the third counting circuit 75 corresponding to the estimated electrical angle (see FIG. 8).
(e)) is changed and output as the electrical angle signal v. A deviation amount v3 between the estimated electrical angle and the predetermined value (zero) at the time of the falling edge of the pulse signal g is detected, and the measurement result signal f of the time measuring device 42 is multiplied by the correction signal n corresponding to the deviation amount v3. Then, the multiplication result is latched by the second latch circuit 72. As a result, the generation time interval of the internal timing signal z by the second counter circuit 74 is corrected according to the deviation amount v3. That is, when the estimated electrical angle is delayed (v3 <0), the generation time interval of the internal timing signal z is corrected to be short, and when the estimated electrical angle is advanced (v3> 0), the internal timing is reduced. The generation time interval of the signal z is corrected to be long. As a result, the amount of deviation at the time when the next falling edge of the pulse signal g occurs is zero or small. As a result, the third counter circuit 75 synchronized with the pulse signal g indicating the rotational position of the rotor
A count signal v (corresponding to the estimated electrical angle) of is obtained.

The command generator 45 receives the count signal v, the internal timing signal z, and the current command signals jq, jd, and outputs three-phase sinusoidal drive command signals ja, jb, jc. The command generator 45 is composed of a microcomputer and performs the arithmetic processing of the flowchart shown in FIG.

(1) Interrupt start processing 80 The following interrupt processing is performed by the generation of the internal timing signal z.

(2) Input processing 81 The count signal v and the current command signals jq, jd (two-phase current command signals) are input.

(3) Two-phase rotation / stationary conversion process 82 The conversion electrical angle w in which the phase is adjusted from the count signal v
To calculate.

W = k0 (v + v0) where k0 is a proportional coefficient and v0 is a phase shift value. Next, the coordinate conversion between the rotating coordinate system and the stationary coordinate system is performed according to the following (Equation 1), and the converted current command signal is converted in proportion to the two-phase current command signals jq, jd and the electrical angle w. hq,
Find hd. The converted current command signals hq and hd are two-phase signals having a phase difference of 90 degrees in electrical angle.

[0029]

[Equation 1]

(4) Two-phase / three-phase conversion processing 83 Three-phase drive command signals ja, jb, jc are obtained from the two-phase converted current command signals hq, hd by the following equation.

[0031]

[Equation 2]

Here, Jo is a proportional constant. The drive command signal j obtained by this two-phase / three-phase conversion process
a, jb, and jc are three-phase signals having a phase difference of 120 degrees in electrical angle.

(5) Output processing 84 The drive command signals ja, jb, and jc are DA converted and output.

(6) End processing 85 Terminate interrupt processing.

The controller 46 controls the two-phase current command signals jq,
jd is given to the command generator 45. In the present embodiment, the speed command signal r and the measurement result signal f of the time measuring device 42 are compared, and a predetermined speed control calculation is performed so as to make the difference zero, thereby obtaining the current command signals jq, jd. . As shown in the flowchart of FIG. 6 described above, the amplitudes of the drive command signals ja, jb, jc of the command generator 45 are the current command signals jq, jd.
Changes in proportion to.

The drive controller 22 drives the drive command signals ja, j.
b, jc and the current feedback signals da, db, dc are respectively compared, and drive transistor groups are provided so as to supply the drive currents Ia, Ib, Ic corresponding to the drive command signals to the drive windings 20A, 20B, 20C, respectively. PWM control (pulse width modulation control) is performed. FIG. 7 shows the configuration of the drive controller 22 and the connection with the power supply unit 23 and the drive windings 20A, 20B, 20C. The drive controller 22 includes differential amplifier circuits 91a and 91a.
b, 91c, comparators 92a, 92b, 92c, and a triangular wave generating circuit 93. The differential amplifier circuit 91a amplifies and outputs a difference signal between the drive command signal ja and the current feedback signal da of the current detector 24a. The comparator 92a compares the output of the differential amplifier circuit 91a with the triangular wave signal of the predetermined frequency (about 20 kHz) of the triangular wave generating circuit 93 to generate a PWM signal (pulse width modulation signal). The PWM signal of the comparator 92a turns on and off the upper drive transistor 31a and the lower drive transistor 33a, and the upper diode 32a and the lower diode 34 are driven.
The drive current Ia smoothed by a and the drive winding is supplied to the drive winding 20A. Therefore, the current detector 24a, the differential amplifier circuit 91a, the comparator 92a, the drive transistors 31a and 33a, the diodes 32a and 34a, and the drive winding form a feedback loop, and the drive current Ia is proportional or substantially proportional to the drive command signal ja. It becomes a sinusoidal current. Similarly, the current detector 24b, the differential amplifier circuit 91b, the comparator 92b, the drive transistors 31b and 33b, the diodes 32b and 34b, and the drive winding form a feedback loop, and the drive current Ib is proportional to or substantially equal to the drive command signal jb. It becomes a proportional sinusoidal current. Further, the current detector 24c, the differential amplifier circuit 91c, the comparator 92c, and the drive transistor 3
A feedback loop is formed by 1c and 33c, diodes 32c and 34c, and a drive winding, and a drive current Ic
Is a sinusoidal current proportional or substantially proportional to the drive command signal jc.

As shown in this embodiment, the rotation detector, the time measuring device, the electrical angle estimator, and the command generator obtain an estimated electrical angle, and a three-phase sinusoidal drive command corresponding to the estimated electrical angle is obtained. Generate signals ja, jb, jc, and drive command signals ja, j
Three-phase sinusoidal drive currents Ia, I proportional to b, jc
If b and Ic are supplied to the drive windings 20A, 20B and 20C, it is possible to obtain a uniform torque with little torque fluctuation. This will be described. The generated torque is generated by the interaction between the drive currents Ia, Ib, Ic of the respective phases and the magnetic flux of the permanent magnet 12, and normally, the generated magnetic flux density of the permanent magnet 12 also changes sinusoidally. As a result, the generated torque is Tor = Kr * {sinw * sin (w + w1) + sin (w-120) * sin (w + w1-120) + sin (w-240) * sin (w + w1-240)} = (3/2)・ Kr · cosw1 and a uniform drive torque Tor can be obtained (where w1 corresponds to the phase shift between the current and the magnetic flux density). Therefore, motor vibration and noise are extremely small.

Further, as shown in the present embodiment, the generation timing interval of the pulse signal that changes in synchronization with the rotation of the rotor is measured, and the estimated electrical angle is changed at each time interval in response to the measurement result, If a sinusoidal drive command signal corresponding to the estimated electrical angle is generated, a sinusoidal drive that smoothly changes in synchronization with the rotation of the rotor is used while using an extremely simple detection element (one detection element). A command signal can be obtained.

Further, as shown in this embodiment, the deviation amount between the estimated electrical angle and the predetermined value at the time when the edge of the pulse signal is generated is detected, and the generation time interval of the internal timing signal is corrected based on this deviation amount. If so, the estimated electrical angle can be gradually corrected to match the value synchronized with the rotation of the rotor. As a result, the drive command signal has a sinusoidal waveform that is very well synchronized with the rotation of the rotor, and it is possible to prevent out-of-synchronization and step-out of the motor. Further, the change in the estimated electrical angle becomes continuous within the range of the predetermined step angle, so that the sinusoidal drive command signal is not discontinuous. Therefore,
As described above, a smooth drive current can be supplied, a uniform drive torque can be obtained, and motor vibration and noise are reduced.

Further, as shown in this embodiment, the magnetic fluxes generated by the magnetic poles of the permanent magnet 12 embedded inside the outer yoke 13 of the rotor field part are generated by the ferromagnetic yoke blocks 13a, 13b, 13c. , 13d, and a motor structure that allows the stator winding to flow into and out of the salient poles provided with the drive winding, and the drive winding has a three-phase sinusoidal drive current I
When a, Ib, and Ic are supplied, the drive command signals ja, jb, and jc are phase-advanced appropriately by changing the ratio of the current command signals jq and jd to correspond to the drive command signals. By setting the current phases of the drive currents Ia, Ib, Ic to the lead phases, it is possible to increase the torque generated during high speed rotation and increase the maximum rotation speed. This will be described. The counter electromotive voltage (velocity electromotive force) due to the permanent magnet magnetic flux generated at the time of high-speed rotation blocks the flow of the drive current, and as a result, the drive current at the time of high-speed rotation becomes small and the generated torque becomes small. In addition, the maximum rotation speed is also limited by the back electromotive force and is kept low. On the other hand, if the phase-advancing drive current is passed, an inductance voltage due to the drive current is generated in the direction of subtracting the back electromotive force, and the margin voltage for passing the drive current is increased. As a result, the drive current during high-speed rotation can be increased, and the generated torque and the maximum rotation speed can be increased. In particular, in the motor structure of this embodiment, the inductance seen from the drive winding becomes considerably large due to the presence of the outer yoke block, the subtraction effect due to the inductance voltage becomes large, and the effect of increasing the torque and increasing the maximum rotation speed can be increased. Further, by advancing the phase of the drive current, the reluctance torque by the outer yoke block can also be used, and the generated torque is further increased. These effects are one of the advantages obtained by supplying a sinusoidal drive current. Further, the drive command signal and the amount of phase advance of the drive current can be changed as necessary. The phase advance is zero or small at low speed rotation, and the phase advance is increased at high speed rotation. Further, such a phase advance of the drive command signal can be realized by advancing the estimated electrical angle itself with respect to the electrical angle corresponding to the actual rotational position, and is included in the present invention.

When the brushless motor of this embodiment is started, a three-phase drive command signal that changes in a predetermined cycle is applied to the drive controller 22 by a start-up circuit (not shown), and the drive windings 20A, 20B, By forcibly switching the drive current to 20C, rotation in a predetermined direction is performed. A pulse signal g of the rotation detector 41 is generated along with the rotation of the rotor, and the time measuring device 42 operates to generate steady sinusoidal drive command signals ja, jb, and jc, and the above-described steady drive is performed. The operation is started (at this time, the starting circuit is stopped).

(Second Embodiment) A second embodiment of the present invention will be described below with reference to the drawings.

FIGS. 9 to 12 show a second embodiment of the brushless motor of the present invention. FIG. 9 shows the circuit configuration of the second embodiment (the same components as those in the above-mentioned first embodiment are designated by the same reference numerals). In the present embodiment, the drive command generation unit 43 of the drive command unit 21 is configured by the microcomputer device 101 and the timer device 102 to enhance the function as described later. Further, the number of current detectors was reduced to two, and the current detection of the other one phase was performed by the adder 103 and the inverting amplifier 104. The configuration and operation of the other parts are the same as those of the above-described embodiment, and the description thereof will be omitted. The motor structure of this embodiment is similar to that shown in FIG.

First, the current detection will be described. Since the combined value of the three-phase drive currents Ia, Ib, Ic is zero, the combined value of the current feedback signals da, db, dc is also zero. Therefore, the signal for one phase can be obtained from the signals for the other two phases by the following equation.

Dc =-(da + db) Therefore, the adder 10
If the current feedback signals da and db for two phases are added by 3 and the sign of the addition result is inverted by the inverting amplifier 103,
The current feedback signals dc of the remaining phases can be obtained.

Next, the operations of the microcomputer 101 and the timer 102 of the drive command generator 43 will be described. The microcomputer device 101 inputs the measurement result signal f of the time measuring device 42, the second differential pulse y, and the current command signals jq, jd, performs predetermined arithmetic processing, and outputs drive command signals ja, jb, jc. . 10 and 11 show flowcharts of the microcomputer 101.

First, the interrupt processing by the second differential pulse y shown in FIG. 10 will be described. This process is executed every time the time measuring device 42 obtains a new measurement result.

(1) Interrupt processing 110 The following interrupt processing is performed by the generation of the second differential pulse y.

(2) Input processing 111 The measurement result signal f of the time measuring instrument 42 is input and its value is F
And

(3) Basic timer value setting process 112 The measurement result value F is set to the timer value Tt for the timer device 102. That is, Tt = F (4) Deviation amount detection processing 113 Count value V of an internal counter described later at this point
Put (corresponding to the estimated electrical angle) in V3. That is, V3 = V is set and the amount of deviation of the internal count value from the predetermined value (zero) is obtained. When the phase of the internal count value is delayed, V3 is negative, and when the phase is advanced, V3 is positive.

(5) Deviation correction process 114 The timer value Tt is corrected according to the detected deviation amount V3. That is, Tt = Tt + k1V3 where k1 is a proportional coefficient. When the phase of the internal count value is delayed, V3 <0 and Tt are corrected to be small. When the phase of the internal count value is advanced, V3> 0 and Tt are corrected to be increased. To be done.

(6) Acceleration / deceleration detection processing 115 The acceleration / deceleration value Ac representing the acceleration / deceleration state is calculated from the new measurement result value F and the one-time old measurement result value F1, and thereafter, it is set to F1 for the next processing. Substitute F.

Ac = F-F1 F1 = F The acceleration / deceleration value Ac has a negative value because F <F1 during acceleration, and has a positive value because F> F1 during deceleration.

(7) Acceleration / deceleration correction processing 116 The timer value Tt is corrected according to the acceleration / deceleration value Ac. That is, Tt = Tt + k2 · Ac, where k2 is a proportional coefficient. During acceleration, Ac <0 and Tt are corrected to be small, and during deceleration Ac> 0.
And Tt is corrected to be large.

(8) Output processing 117 to the timer device The timer value Tt is output to the timer device 102 and stored therein. The timer device 102 down-counts this Tt as the load value from the next timer counting, and the internal timing signal z is output to the microcomputer device 101 every time it becomes zero.
Then, Tt is loaded and the down count is continued. Therefore, the internal timing signal z can be obtained at each internal timing time interval according to the timer value Tt.

(9) Interrupt termination processing 118 Terminate interrupt processing.

Next, the interrupt processing by the internal timing signal z shown in FIG. 11 will be described. This process
It is executed every time the timer device 102 generates the internal timing signal z.

(1) Interrupt processing 120 The following interrupt processing is performed by the generation of the internal timing signal z.

(2) Input processing 121 Input the current command signals jq and jd.

(3) Internal count processing 122 The internal count value V corresponding to the estimated electrical angle is incremented. That is, V = V + 1. When V becomes equal to (or becomes larger than) the second set value V2, V is reset to the first set value V1. Here, the first set value V1 is -180 in terms of electrical angle.
The second set value V2 is 180 degrees-
It is a positive value corresponding to (1 count). Therefore, the internal count value V counts up each time the internal timing signal z arrives and is repeatedly counted between the first set value V1 and the second set value V2.

(4) Two-phase rotation / stationary conversion process 123 The conversion electrical angle w is calculated from the internal count value V.

W = k0 (V + V0) where k0 is a proportional coefficient and V0 is a phase shift value. Next, coordinate conversion between the rotating coordinate system and the stationary coordinate system is performed to obtain the converted coordinate converted current command signals hq and hd (the specific equation is the above-mentioned (Equation 1)).

(5) Two-phase / three-phase conversion processing 124 The three-phase drive command signals ja, jb, and jc are calculated from the two-phase converted current command signals hq and hd (the concrete formula is the above-mentioned 2)).

(6) Output processing 125 The drive command signals ja, jb, and jc are DA converted and output.

(7) End processing 127 Terminate interrupt processing.

The structure and operation of the motor structure, the rotation detector 41, the time measuring device 42, the controller 43, the drive controller 22, and the power supply unit 23 of this embodiment are the same as those of the first embodiment. , Description is omitted. Drive currents Ia, I of this embodiment
b and Ic also become three-phase sinusoidal currents proportional to the drive command signals ja, jb, and jc.

In this embodiment, the deviation amount of the estimated electrical angle at the time of edge generation of the pulse signal is detected, and the generation time interval of the internal timing signal is corrected based on this deviation amount,
The estimated electrical angle is gradually made to coincide with the value synchronized with the rotation of the rotor. At the same time, the generation time interval of the internal timing signal is corrected based on the acceleration / deceleration value Ac that detects the acceleration / deceleration state of the rotor, and the time interval is shortened during acceleration and lengthened during deceleration. As a result, the estimated electrical angle in the acceleration / deceleration state also matches the rotational position of the rotor very well, and uniform torque with little fluctuation can be obtained.

It should be noted that these corrections can be eliminated in a timely manner. For example, if the deviation correction process 114 in FIG. 10 is eliminated, only acceleration / deceleration correction will be performed. Also,
If the acceleration / deceleration correction process 116 is eliminated, only the deviation correction process is performed. Further, if both the deviation correction process 114 and the acceleration / deceleration correction process 116 are eliminated, the correction process is not performed at all.

Further, it is possible to simplify the deviation correction process and directly correct the estimated electrical angle to a predetermined value at the timing of generation of the pulse signal g. FIG. 12 shows a specific flowchart thereof (replaced with the flowchart of FIG. 10).

(1) Interrupt processing 130 The following interrupt processing is performed by the generation of the second differential pulse y.

(2) Input processing 131 The measurement result signal f of the time measuring device 42 is input, and its value is F
And

(3) Timer value setting processing 132 The measurement result value F is set to the timer value Tt. That is, Tt = F (4) deviation correction processing 133 The internal count value V is set to a predetermined value Vr (for example, zero). That is, V = Vr.

(5) Output Processing to Timer Unit 134 The timer value Tt is output to and stored in the timer unit 102, and the internal timing signal z is output at each internal timing time interval according to the timer value Tt.

(6) Interrupt end processing 135 Terminate interrupt processing.

This method has a drawback that a comparatively large discontinuity in the estimated electrical angle is likely to occur and the drive command signal is not smooth, but the estimated electrical angle can be instantaneously synchronized with the rotor rotation. There are advantages. Therefore, this method is effective when the amount of deviation is small, such as in a steady constant speed rotation state.

When the brushless motor of this embodiment is started up, it changes at a predetermined cycle by a start-up processing program (not shown) of the microcomputer 101.
By applying a phase drive command signal to the drive controller 22 and forcibly switching the drive current to the drive windings 20A, 20B, 20C, rotation in a predetermined direction is performed. A pulse signal g of the rotation detector 41 is generated in accordance with the rotation of the rotor, and the time measuring device 42 operates to generate steady sinusoidal drive command signals ja, jb, and jc, and the above-described steady drive is performed. The operation is started (at this time, the startup processing program is stopped).

Further, in the present embodiment, the rotation detector 41 is constructed by using the detection element for detecting the field magnetic flux of the permanent magnet, but the present invention is not limited to such a case, and the one-phase drive winding is used. A rotation detector that detects a signal in response to a back electromotive force (speed electromotive force) generated in
Included in the present invention. If the pulse signal is obtained from the counter electromotive voltage generated in the drive winding, a special detecting element is not required and the motor structure is simplified.

Further, in the present embodiment, the drive transistor is drive-controlled by the result of comparing the three-phase drive command signals ja, jb, jc and the three-phase current feedback signals da, db, dc, respectively. Not limited to such a case, for example, the drive command signals ja and jb for two phases and the current feedback signals da and db for two phases are compared with each other to generate a comparison error signal for two phases. Minute comparison error signals are added and the sign is inverted to generate the remaining one phase comparison error signals, and the drive transistors may be drive-controlled by the three phase comparison error signals thus obtained. Included in the present invention.

In the present embodiment, the three-phase drive command signals ja, jb, jc and the three-phase current feedback signals da, db, dc are compared as analog signals, respectively. For example, the current feedback signals da, db,
dc may be AD-converted and input as a digital signal into a microcomputer to digitally compare the drive command signal and the current feedback signal, which is included in the present invention. The sinusoidal drive command signal and drive current described in the present invention have a sinusoidal drive command signal and drive current with respect to changes in the estimated electrical angle when the current command signals jq and jd are constant. It means changing.

(Embodiment 3) A third embodiment of the present invention will be described below with reference to the drawings.

13 to 17 show a third embodiment of the brushless motor of the present invention. FIG. 13 shows the circuit configuration of the third embodiment, and FIG. 14 shows the motor structure. In this embodiment, the number of detection elements is increased, the processing of the rotation detector 201 and the microcomputer 101 is improved, and the starting operation using the position detection signal can be performed. Regarding other configurations and operations, the same numbers are assigned to the same parts as those in the above-described embodiment.

First, the motor structure shown in FIG. 14 will be described. The inner yoke 11 and the outer yoke 13 made of a ferromagnetic material attached to the rotor rotation shaft 10 have four axially symmetric (9
It has a thin connecting portion in 0 degree symmetry). As a result, the mechanical connection is maintained, but magnetic saturation occurs and they are magnetically separated. Permanent magnets 12a, 12b, 12c and 12d magnetized in the radial direction are embedded in the four gaps between the inner yoke 11 and the outer yoke 13 while alternately changing their polarities. Further, the yoke block 1 of the outer yoke 13 is provided on the outer peripheral side of each magnetic pole.
3a, 13b, 13c, 13d are arranged. As a result, the field portion including the permanent magnets 12a, 12b, 12c, and 12d and the outer yoke 13 has N, S, and
Four magnetic field poles of N and S are formed at equal angular intervals (90 degrees) or substantially equal angular intervals.

The stator core 14 has three-phase drive windings (A1, A2, A3, A4), (B1, B2, B3, B).
4) and (C1, C2, C3, C4) are wound with a predetermined phase difference, (A1, A2, A3, A4) forms the first-phase drive winding 20A, and (B1, B2) , B3, B4)
Forms the second phase drive winding 20B, and (C1, C2, C
3, C4) form the third phase drive winding 20C.
Further, a detection element 211a for detecting the magnetic flux generated in the field magnet portion,
211b, 211c are three-phase drive windings 20A, 20
It is arranged corresponding to B and 20C.

The rotation detector 201 has three detection elements 21.
Three-phase position signals ga, gb, gc corresponding to the rotational position of the rotor obtained from the detection outputs of 1a, 211b, 211c.
And a pulse signal g that is a combination of these position signals is output. FIG. 15 shows a specific configuration of the rotation detector 201.
The outputs of the detection elements 211a and 211b are amplification circuits 222.
After being amplified a required number of times by a and 222b, waveform shaping is performed by shaping circuits 223a and 223b to obtain position signals ga and gb. Further, the output of the detection element 211c is inverted and amplified to the opposite sign by the inverting amplifier circuit 221, amplified by a required number of times by the amplifier circuit 222c, and further shaped by the shaping circuit 223c to obtain the position signal gc. As a result, the position signals ga, gb, gc become three-phase digital signals having a phase difference of 120 degrees electrically. The position signals ga, gb, gc are supplied to the AND circuit 22.
4, 225, 226 and the OR circuit 227 are logically combined to output a pulse signal g. The pulse signal g becomes “H” when the two position signals are “H” (high potential state), and becomes “H” when the two position signals are “L” (low potential state).
L ″. Position signals ga and g are shown in FIGS.
The waveform relationship between b and gc and the pulse signal g is shown.

The pulse signal g of the rotation detector 201 is input to the time measuring device 42, the timing of occurrence of the falling edge of the pulse signal g is measured, and the measurement result signal f and the second
Obtain the differential pulse y. The time measuring device 42 has the same configuration as that shown in FIG.

The microcomputer 101 inputs the measurement result signal f, the second differential pulse y, the current command signals jq, jd, and the position signals ga, gb, gc, performs predetermined processing, and drives command signals ja, jb. , Jc are output.

In a steady rotation state, the microcomputer 101 executes the processing shown in FIG. 16 and the above-mentioned flowchart of FIG. The interrupt processing by the second differential pulse y shown in FIG. 16 will be described.
This process is executed every time the time measuring device 42 obtains a new measurement result.

(1) Interrupt processing 230 The following interrupt processing is performed by the generation of the second differential pulse y.

(2) Input processing 231 The measurement result signal f of the time measuring instrument 42 is input and its value is F
And

(3) Basic timer value setting process 232 The measurement result value F is set to the timer value Tt. That is, Tt = F (4) Input process 233 The position signals ga, gb, gc are input.

(5) Deviation amount detection process 234 Internal count value V at this point (corresponding to the estimated electrical angle, internal count process 122 of the flowchart of FIG. 11)
The expected value Vp of the count value) obtained in (1) is selected according to the states of the position signals ga, gb, gc. The generation timing of the falling edge of the pulse signal g is the position signal g
Since there are three states of a, gb, and gc, the corresponding expected value is selected from the three expected values that are 120 degrees apart from each other as the estimated electrical angle. Next, the deviation amount V3 between the internal count value V and the selected expected value Vp at this time is detected (V3 = V-Vp). Therefore, V3 is negative when the phase of the internal count value is delayed, and V3 is positive when the phase is advanced.

(6) Deviation correction process 235 The timer value Tt is corrected according to the detected deviation amount V3. That is, Tt = Tt + k1V3 where k1 is a proportional coefficient. When the phase of the internal count value is delayed, V3 <0 and Tt are corrected to be small. When the phase of the internal count value is advanced, V3> 0 and Tt are corrected to be increased. To be done.

(7) Output processing 236 to timer unit 236 The timer value Tt is output to the timer unit 102 and stored therein. The timer device 102 down-counts this Tt as the load value from the next timer counting, and the internal timing signal z is output to the microcomputer device 101 every time it becomes zero.
Then, Tt is loaded and the down count is continued. Therefore, the internal timing signal z can be obtained at each internal timing time interval according to the timer value Tt.

(8) Interrupt termination processing 237 Terminate interrupt processing.

In this way, the timer value is set in the timer unit 102 which generates the internal timing signal z.

Internal timing signal z of the timer unit 102
The interrupt processing by means of FIG. 11 is shown in the flow chart of FIG. 11, and the internal timing signal z is generated at each time interval in response to the timer value Tt, and the count value V of the internal counter is changed by the generation of the internal timing signal z. By updating to obtain a new estimated electrical angle, the current command signals jq, jd are subjected to rotational / stationary coordinate conversion using this estimated electrical angle, and further two-phase / three-phase conversion is performed to generate a three-phase sinusoidal wave. The drive command signals ja, jb, and jc are output (similar to the description of the flowchart of FIG. 11, and detailed description thereof is omitted). Although the falling edge of the pulse signal g arrives three times while the internal count value V changes by an electrical angle of 360 degrees, the deviation amount is detected at each arrival time point, and the deviation amount is detected by the next arrival time point. The interval at which the internal timing signal z is generated (timer value Tt) is corrected so that

The configuration and operation of the drive controller 22, the power supply unit 23, etc. are the same as those of the above-mentioned embodiment, and the sinusoidal drive currents Ia, I proportional to the drive command signals ja, jb, jc.
b and Ic are supplied to the drive windings 20A, 20B and 20C.
As a result, a uniform torque with little fluctuation is obtained, and the motor continues to rotate smoothly.

Next, the operation at startup will be described. FIG. 17 shows a microcomputer device 1 including processing at the time of startup.
A flowchart of 01 is shown.

(1) Start processing 240 This process is started when the power is turned on.

(2) Judgment process 241 It is judged whether the routine process is being performed. When the motor is rotating and the drive command signal is generated using the measurement result signal f of the time measuring device 42, the process branches to step 245. If it is in the activated state, the process branches to step 242, and a drive command signal for activation is created. Note that the determination of the steady rotation state is made by, for example, the second differential pulse y of the time measuring device 42.
Is repeatedly determined within a predetermined time interval. If it is determined that the internal timing signal z is activated, the interrupt processing of the internal timing signal z is not performed.

(3) Input processing 242 The position signals ga, gb, gc are input.

(4) Drive command generation processing 243 Drive command signals ja, jb, jc from position signals ga, gb, gc
Is created by the following formula.

Ja = Jm.multidot. (Ga-gb) jb = Jm.multidot. (Gb-gc) jc = Jm.multidot. (Gc-ga) where ga, gb, and gc are "H", "1" and "L". Sometimes set to 0. Therefore, ja, jb, jc are Jm, 0, -Jm
It changes in 3 states. 18 (e), (f), and (g) show the waveforms of the drive command signals ja, jb, and jc at the time of startup.

(5) Output processing 244 The drive command signals ja, jb, and jc for starting are DA-converted and output.

(6) Main Routine Process 245 The required main process is performed, and the process branches to the determination process 241.
It should be noted that the processing amount of the main processing is reduced so as not to affect the creation of the drive command signal at the time of startup (the main processing may be omitted if necessary).

In this embodiment, three-phase position signals ga, gb,
Position detection of gc (position detection mechanism), position signals ga, g
A rectangular drive command signal ja, jb, jc that changes stepwise using b and gc is created as a start drive command signal (start drive command creation mechanism), and reliable rotation drive is performed by the start drive command signal. Drive command signals ja, jb, j in a sinusoidal shape using the estimated electrical angle during rotation.
Smooth rotation is performed by c. as a result,
A stable and reliable motor starting operation is performed, a uniform drive torque with little fluctuation is obtained during rotation, and motor vibration and noise are significantly reduced.

In this embodiment, three-phase position signals ga, gb,
Since the generation timing interval of the pulse signal g obtained by synthesizing gc is time-measured, the number of times of measurement per 360 electrical angle increases three times, and more accurate electrical angle estimation is performed.
In addition, the amount of deviation detection / correction is tripled, and the amount of deviation is significantly reduced. Further, the acceleration / deceleration detection processing 115 and the acceleration / deceleration correction processing 116 shown in FIG. 10 are inserted between the processing 235 and the processing 236 of the flowchart of this embodiment so that the acceleration / deceleration detection / correction operation is performed. However, it is included in the present invention.

Further, in this embodiment, the rotation detector 201 is constructed by using three detecting elements for detecting the field magnetic flux of the permanent magnet.
However, the present invention is not limited to such a case, but a pulse signal is applied to a signal in response to a back electromotive force generated in a three-phase drive winding during steady rotation, thereby obtaining a position signal and a pulse signal. A detector may be used (at this time, the starting operation of the motor is as described in the first embodiment above,
The drive current may be switched at an appropriate cycle). If the counter electromotive voltage generated in the drive winding is detected, no special detection element is required and the motor structure is simplified.

(Embodiment 4) A fourth embodiment of the present invention will be described below with reference to the drawings.

19 to 22 show the fourth embodiment of the brushless motor of the present invention. FIG. 19 shows the circuit configuration of the fourth embodiment (the same components as those in the above-mentioned second embodiment are designated by the same reference numerals). In the present embodiment, the computing unit 311 of the conversion comparing unit 301 is composed of a microcomputer unit 312 and a timer unit 313, and as will be described later, current command signals jq, jd of the current command unit 302 and current detectors 24a, 2a.
The current feedback signals da and db of 4b are input, a predetermined conversion / comparison operation is performed using the estimated electrical angle, and an error detection operation and a control operation operation are performed. Further, the PWM device 303 creates a PWM signal (pulse width modulation signal) from the output signals ma, mb, mc of the conversion comparison unit 301, and drives the drive transistor 3
The on / off control of 1a, 31b, 31c, 33a, 33b, 33c is performed. Other configurations and operations are the same as those in the above-described embodiment, and detailed description thereof will be omitted. The motor structure of this embodiment is similar to that shown in FIG.

The rotation detector 41 of the conversion comparison unit 301 uses the output signal of the detection element 17 to generate a pulse signal g having a frequency proportional to the rotation speed of the rotor. Rotation detector 4
1 uses the configuration example shown in FIG. Time measuring instrument 4
2 measures the generation timing interval of the output pulse signal g of the rotation detector 41, and measures the measurement result signal f and the second differential pulse y.
Is output. The time measuring device 42 uses the configuration example shown in FIG.

The calculation unit 311 of the conversion comparison unit 301 is composed of a microcomputer unit 312 and a timer unit 313, and the measurement result signal f of the time measuring unit 42, the second differential pulse y, and the current command signal jq of the current command unit 302. , Jd and the current feedback signals da, db of the current detectors 24a, 24b are input, and predetermined arithmetic processing is performed to output signals ma, mb, mc.
Is output. 20 and 21 show flowcharts of the microcomputer 312.

First, the interrupt processing by the second differential pulse y shown in FIG. 20 will be described (the same as the contents shown in the flowchart of FIG. 10). This process is executed every time the time measuring device 42 obtains a new measurement result.

(1) Interrupt processing 320 The following interrupt processing is performed by the generation of the second differential pulse y.

(2) Input process 321 The measurement result signal f of the time measuring instrument 42 is input, and its value is F
And

(3) Basic timer value setting process 322 The measurement result value F is set to the timer value Tt for the timer 313. That is, Tt = F (4) Deviation amount detection processing 323 Count value V of an internal counter described later at this point
Put (corresponding to the estimated electrical angle) in V3. That is, V3 = V is set and the amount of deviation of the internal count value from the predetermined value (zero) is obtained. When the phase of the internal count value is delayed, V3 is negative, and when the phase is advanced, V3 is positive.

(5) Deviation correction processing 324 The timer value Tt is corrected according to the detected deviation amount V3. That is, Tt = Tt + k1V3 where k1 is a proportional coefficient. When the phase of the internal count value is delayed, V3 <0 and Tt are corrected to be small. When the phase of the internal count value is advanced, V3> 0 and Tt are corrected to be increased. To be done.

(6) Acceleration / deceleration detection processing 325 The acceleration / deceleration value Ac representing the acceleration / deceleration state is calculated from the new measurement result value F and the one-time old measurement result value F1, and thereafter, it is set to F1 for the next processing. Substitute F.

Ac = F-F1 F1 = F The acceleration / deceleration value Ac has a negative value because F <F1 during acceleration, and has a positive value because F> F1 during deceleration.

(7) Acceleration / deceleration correction processing 326 The timer value Tt is corrected according to the acceleration / deceleration value Ac. That is, Tt = Tt + k2 · Ac, where k2 is a proportional coefficient. During acceleration, Ac <0 and Tt are corrected to be small, and during deceleration Ac> 0.
And Tt is corrected to be large.

(8) Output process 327 to timer device The timer value Tt is output to and saved in the timer device 313. The timer device 313 counts down from the next timer count using this Tt as a load value, and the internal timing signal z is output to the microcomputer device 312 every time it becomes zero.
Then, Tt is loaded and the down count is continued. Therefore, the internal timing signal z can be obtained at each internal timing time interval according to the timer value Tt.

(9) Interrupt end processing 328 Terminate interrupt processing.

Next, the interrupt processing by the internal timing signal z shown in FIG. 21 will be described. This process
It is executed every time the internal timing signal z of the timer 313 is generated.

(1) Interrupt processing 340 When the internal timing signal z is generated, the following interrupt processing is performed.

(2) Internal Count (Electrical Angle Estimation) Process 341 The internal count value V corresponding to the estimated electrical angle is incremented. That is, V = V + 1. When V becomes equal to (or becomes larger than) the second set value V2, V is reset to the first set value V1. Here, the first set value V1 is -180 in terms of electrical angle.
The second set value V2 is 180 degrees-
It is a positive value corresponding to (1 count). Therefore, the internal count value V counts up each time the internal timing signal z arrives and is repeatedly counted between the first set value V1 and the second set value V2. From the internal count value V, the conversion electrical angle w (estimated electrical angle) for which phase matching has been performed is calculated.

W = k0. (V + V0) Here, k0 is a proportional coefficient and V0 is a phase shift value.

(3) Input processing 342 The current feedback signals da and db are AD-converted and digitally input.

(4) Conversion feedback signal creation processing 343 Coordinate conversion between the stationary coordinate system and the rotating coordinate system is performed on the current feedback signals da and db using the estimated electrical angle w, and the coordinate-converted conversion feedback is performed. The signals gd and gq are obtained by the following equation.

[0129]

[Equation 3]

In the above equation, the converted feedback signals gd and gq are directly obtained from the two-phase current feedback signals da and db. This is 2
The current feedback signals dc for the remaining one phase are obtained from the phase current feedback signals da and db, and da, db, and dc are converted into three-phase and two-phase,
Further, it is equivalent to the coordinate conversion based on the estimated electrical angle w.

(5) Input processing 344 Input the current command signals jq, jd.

(6) Control signal generation processing 345 The current command signals jd and jq and the conversion feedback signal g are calculated by the following equation.
The error signals ed and eq are obtained by comparing d and gq.

[0133]

[Equation 4]

Here, Jo and Go are predetermined constants.

The following control calculation is performed on the error signals ed and eq to obtain control signals nd and nq.

[0136]

[Equation 5]

Here, No is a predetermined constant. (In the control calculation of (Equation 5), proportional control is performed, but proportional / integral control calculation or proportional / integral / derivative control calculation may be performed.) (7) Conversion control signal creation process 346 Estimate Using the electrical angle w, coordinate conversion between the rotating coordinate system and the stationary coordinate system is performed on the control signals nd and nq, and the coordinate-converted conversion control signals pa, pb, pc are obtained by the following equation.

[0138]

[Equation 6]

In the above equation, the control signals nd and nq are estimated to be the electrical angle w
It corresponds to the two-phase / three-phase conversion after the coordinate conversion by.

(8) Output signal generation processing 347 The output signals ma, mb and mc corresponding to the conversion control signals pa, pb and pc are obtained by the following equation.

[0141]

[Equation 7]

Here, Mo is a predetermined constant.

(9) Output process 348 The output signals ma, mb and mc are DA converted and output.

(7) End processing 349 Terminate interrupt processing.

The current command unit 302 is composed of the controller 46, and converts the two-phase current command signals jq and jd into the conversion and comparison unit 30.
1 to the arithmetic unit 311. In the present embodiment, the speed command signal r and the measurement result signal f of the time measuring device 42 are compared, and a predetermined speed control calculation is performed so as to make the difference zero, thereby obtaining the current command signals jq, jd. .

Output signals ma, mb, m of the conversion comparator 301
c is input to the PWM device 303 and drives and controls the drive transistor. FIG. 22 shows the configuration of the PWM device 303 and the connection with the power supply unit 23 and the drive windings 20A, 20B, 20C. The PWM device 303 includes a comparator 360a,
It is composed of 360b and 360c and a triangular wave generation circuit 361. The comparator 360a includes the conversion comparison unit 3
The output signal ma of 01 and the triangular wave signal of the predetermined frequency (about 20 kHz) of the triangular wave generation circuit 361 are compared, and PWM
Create a signal (pulse width modulated signal). Comparator 3
The PWM signal of 60a turns on / off the upper drive transistor 31a and the lower drive transistor 33a, and supplies the drive current Ia smoothed by the upper diode 32a, the lower diode 34a and the drive winding to the drive winding 20A. . Similarly, the comparator 360b compares the output signal mb of the conversion comparison unit 301 with the triangular wave signal of the triangular wave generation circuit 361 to generate a PWM signal, and turns on the upper drive transistor 31b and the lower drive transistor 33b.
Turned off, upper diode 32b, lower diode 3
The drive current Ib smoothed by 4b and the drive winding is supplied to the drive winding 20B. Similarly, the comparator 36
0c compares the output signal mc of the conversion comparison unit 301 and the triangular wave signal of the triangular wave generation circuit 361 to generate a PWM signal, drives the upper drive transistor 31c and the lower drive transistor 33c on / off, and drives the upper diode 32.
c, the lower diode 34c and the drive current Ic smoothed by the drive winding are supplied to the drive winding 20C.

Therefore, the calculation unit 311 of the conversion comparison unit 301
The microcomputer 312, the PWM device 303, the power supply unit 23, the drive windings 20A, 20B, 20C and the current detectors 24a, 24b constitute a feedback loop, and have a sine amplitude corresponding to the current command signals jq, jd. Wavy three-phase drive currents Ia, Ib, Ic are supplied to the respective drive windings 20A, 20B, 20C (here, the sine wave drive current of the present invention means that the current command signals jq, jd are constant. It means that the drive current changes sinusoidally when the estimated electrical angle changes.

In the present embodiment, the deviation amount of the estimated electrical angle at the time when the edge of the pulse signal occurs is detected, and the generation time interval of the internal timing signal is corrected based on this deviation amount,
The estimated electrical angle is gradually made to coincide with the value synchronized with the rotation of the rotor. At the same time, the generation time interval of the internal timing signal is corrected based on the acceleration / deceleration value Ac that detects the acceleration / deceleration state of the rotor, and the time interval is shortened during acceleration and lengthened during deceleration. As a result, the estimated electrical angle in the acceleration / deceleration state also matches the rotational position of the rotor very well, and uniform torque with little fluctuation can be obtained.

Further, these corrections can be eliminated in a timely manner. For example, if the deviation correction process 324 of FIG. 20 is eliminated, only acceleration / deceleration correction will be performed. Also,
If the acceleration / deceleration correction process 326 is eliminated, only the deviation correction process is performed. Furthermore, if both the deviation correction process 324 and the acceleration / deceleration correction process 326 are eliminated, the correction process will not be performed at all. Furthermore, it is possible to simplify the deviation correction process and directly correct the estimated electrical angle to a predetermined value at the timing of generation of the pulse signal g (replace the flow chart of FIG. 12 with the flow chart of FIG. 20).

When the brushless motor of this embodiment is activated, the microcomputer unit 3 of the conversion / comparison section 301 is also used.
PW of three-phase output signals ma, mb, and mc that change in a predetermined cycle is generated by a start-up processing program 12 (not shown).
In addition to M unit 303, drive windings 20A, 20B, 20C
By forcibly switching the drive current to, the rotation in the predetermined direction is performed. A pulse signal g of the rotation detector 41 is generated in accordance with the rotation of the rotor, and the time measuring device 42 operates to shift to the operation of supplying a steady sinusoidal drive current (at this time, the startup processing program is Stop). Also in the present embodiment, the above-mentioned third
As described in the above embodiment, the number of position detecting elements is increased and the rotational position signals ga, gb, corresponding to the rotational position of the rotor,
gc is obtained, and in response to the rotational position signals ga, gb, gc at the time of startup, three-phase output signals ma, mb, mc of the conversion comparator 301
(For example, start-up processing is performed by replacing ja, jb, jc with ma, mb, mc in the flowchart of FIG. 17) to forcibly switch the drive current to the drive windings 20A, 20B, 20C. However, the present invention is also included in the present invention (note that, during steady rotation, a sinusoidal drive current is supplied by the above operation).

The arithmetic processing shown in the flow chart of FIG. 21 of the present embodiment can be modified in various ways. For example, the current control arithmetic operation of the control signal generation processing 345 is proportional to
The present invention is also included in the present invention, and may be an integral type or may include a compensation signal that cancels the influence of the back electromotive force (speed electromotive force) in the control signal generation process 345 or the output signal generation process 347.

Further, in the present embodiment, the rotation detector 41 is constructed by using the detecting element for detecting the field magnetic flux of the permanent magnet, but the present invention is not limited to such a case, and the one-phase drive winding is used. A rotation detector that obtains a pulse signal by detecting a signal in response to a back electromotive force (velocity electromotive force) generated in the above may be used and is included in the present invention.

Although the brushless motor having the three-phase drive windings has been described in the above embodiment, the present invention is not limited to such a case, and in general, the K-phase (K
A brushless motor having a drive winding of 2 or more) can be configured. For example, in a motor structure having two-phase drive windings, the above-mentioned hq and hd may be used as drive command signals to supply a drive current proportional to these hq and hd. Further, the time measuring device may measure not only the falling edge of the pulse signal but also the generation timing intervals of both the falling edge and the rising edge. In addition, simple averaging of measurement results by the time measuring device
The set value of the timer may be set after the weighted average processing or the filter processing. In addition, the power supply unit is PWM
Instead of driving, the driving voltage may be changed in an analog manner. Further, the drive transistor may be a bipolar transistor, an IGBT or the like instead of the MOS type transistor. Further, the motor structure is not limited to the structure described above in which the permanent magnet is embedded in the ferromagnetic yoke, and the permanent magnet may be exposed on the surface so as to face the stator core. In addition, the drive winding of the stator is 1
One salient pole may be wound with one winding, and the salient poles are not limited to the above-described embodiments. Further, the detection element may be arranged not as a magnetic flux of a permanent magnet for generating torque but as a component for rotation detection as another structure. At the same time, the detection element is not limited to the magnetoelectric conversion element, and detection elements of other principles may be used. Also, if a rotation detector that detects a signal that responds to the one-phase or three-phase back electromotive force (speed electromotive force) generated in the drive winding to obtain a pulse signal is used, eliminate the special detection element. You can Further, the controller is not limited to speed control, and for example, torque control or position control may be performed to output a current command signal (if there is a current command signal, the controller is not always necessary. Absent). Also, the method of giving the current command signal does not have to be two signals,
For example, it may be a single signal. Further, the arithmetic expression for obtaining the drive command signal and the drive current is not limited to the above-mentioned configuration. Or a negative slope part has an electrical angle of about 60 degrees) or a triangular wave (for example, one side of the positive slope part or the negative slope part has an electrical angle of about 90 degrees).
Etc. are also included in the range of the sine wave drive command signal and drive current described in the present invention. Further, the number of steps of the estimated electrical angle is not limited to the above-described configuration (360 ° / 10 ° = 36 because a resolution of 10 ° or less in electrical angle is preferable.
Steps or more are preferred). Further, the acceleration / deceleration detection may use, for example, a change in the speed command signal r or directly generate an acceleration command instead of using the measurement result of the time measuring device. Further, in the above-described embodiment, the drive transistor is drive-controlled based on the result of comparison between the three-phase drive command signal and the three-phase current feedback signal, but the present invention is not limited to such a case and, for example, 2
The drive command signals for the phases are compared with the current feedback signals for the two phases to obtain the error signals for the two phases, and the control signals for the two phases are obtained by performing a predetermined current control operation on the error signals. The control signals for the remaining one phase are created by adding the control signals for the two phases and inverting the sign, and the drive transistors are driven and controlled in response to the control signals for the three phases thus obtained. May be. Further, in order to improve the accuracy of the current control, the current control calculation may be of a proportional / integral type, or a compensating voltage for canceling the influence of the counter electromotive force (speed electromotive force) may be added. Moreover, the calculation using the estimated electrical angle is not limited to the above-described embodiment, and various modifications can be made. Besides, it goes without saying that various modifications can be made without changing the gist of the present invention and are included in the present invention.

[0154]

As described above, according to the brushless motor of the present invention, the timing time interval of the pulse signal synchronized with the rotation of the rotor is measured, and the estimated electrical angle is changed at each time interval corresponding to the measurement result, and the estimation is performed. Since a sinusoidal drive current corresponding to the electrical angle is supplied to the drive winding, the effect of current distortion due to the winding inductance is significantly reduced, a uniform drive torque with less fluctuation is obtained, and the motor rotates smoothly. Driven, motor vibration and noise are significantly reduced. Further, it is possible to reduce the pulse signal for rotation detection, and it is possible to employ a simple rotation detector having a small number of parts.

[Brief description of drawings]

FIG. 1 is a circuit configuration diagram of a first embodiment of the present invention.

FIG. 2 is a motor structure diagram in the first embodiment.

FIG. 3 is a circuit configuration diagram of a rotation detector 41 in the first embodiment.

FIG. 4 is a circuit configuration diagram of a time measuring device 42 according to the first embodiment.

FIG. 5 is a circuit configuration diagram of an electrical angle estimator 44 in the first embodiment.

FIG. 6 is a flow chart of a command generator 45 in the first embodiment.

FIG. 7 is a circuit configuration diagram of a drive controller 22 and a power supply unit 23 in the first embodiment.

FIG. 8 is a waveform diagram for explaining the operation in the first embodiment.

FIG. 9 is a circuit configuration diagram of a second embodiment of the present invention.

FIG. 10 is a first flowchart of the microcomputer 101 according to the second embodiment.

FIG. 11 is a second flowchart of the microcomputer 101 according to the second embodiment.

FIG. 12 is a third flowchart used in place of the first flowchart (FIG. 10) of the microcomputer 101 in the second embodiment.

FIG. 13 is a circuit configuration diagram of a third embodiment of the present invention.

FIG. 14 is a motor structure diagram according to a third embodiment.

FIG. 15 is a circuit configuration diagram of a rotation detector 201 according to a third embodiment.

FIG. 16 is a first flowchart of the microcomputer 101 according to the third embodiment.

FIG. 17 is a second flowchart of the microcomputer 101 according to the third embodiment.

FIG. 18 is a waveform chart for explaining the operation in the third embodiment.

FIG. 19 is a circuit configuration diagram of a fourth embodiment of the present invention.

FIG. 20 is a first flowchart of the microcomputer 312 in the fourth embodiment.

FIG. 21 is a second flowchart of the microcomputer 312 in the fourth embodiment.

FIG. 22 is a circuit configuration diagram of the PWM device 303 and the power supply unit 23 in the fourth embodiment.

FIG. 23 is a configuration diagram of a conventional brushless motor.

[Explanation of symbols]

10 Rotor Rotating Shafts 12, 12a, 12b, 12c, 12d Permanent Magnet 13 Outer Yoke 14 Stator Iron Cores A1 to A4, B1 to B4, C1 to C4, 20A to 20C
Drive windings 17, 211a to 211c Detection element 21 Drive command unit 22 Drive controller 23 Power supply units 24a, 24b, 24c Current detectors 31a, 31b, 31c Upper drive transistors 33a, 33b, 33c Lower drive transistors 41, 201 Rotation detector 42 Time measuring device 43 Drive command preparation unit 44 Electric angle estimator 45 Command preparation device 101, 312 Microcomputer device 102, 313 Timer device 301 Conversion comparison unit 302 Current command unit 303 PWM device 311 Calculation unit

─────────────────────────────────────────────────── --Continued from the front page (56) References JP-A-4-236190 (JP, A) JP-A-1-126191 (JP, A) JP-A-3-198687 (JP, A) JP-A-6- 253582 (JP, A) (58) Fields investigated (Int.Cl. ⁷ , DB name) H02P 6/16 H02P 6/06

Claims (6)

(57) [Claims] 1. A field magnetic section attached to a rotor and forming a field pole of a P pole (where P is an even number of 2 or more) using a magnetic flux generated by a permanent magnet; Three-phase drive windings interlinking with the magnetic flux of the magnetic part, three MOS-type upper drive transistors, three MOS-type lower drive transistors, and three connected in parallel to each upper drive transistor. A plurality of upper diodes and three lower diodes connected in parallel to the respective lower drive transistors, and a power supply unit that supplies power to the three-phase drive windings, and detects rotation of the rotor. Drive command means for generating a sinusoidal drive command signal, and drive control of the upper drive transistor and the lower drive transistor of the power supply means in response to the three-phase drive command signals to generate a sinusoidal wave Phase drive voltage And a drive means for supplying three-phase drive windings to the drive winding, wherein the drive command means is rotation detection means for obtaining a pulse signal that changes in synchronization with rotation of the rotor. A time measuring unit that measures a timing interval synchronized with the rotation of the rotor by the pulse signal of the rotation detecting unit; and a rotation position of the rotor for each time interval shorter than the timing interval of the time measuring unit. Electrical angle estimating means for estimating the estimated electrical angle, and the estimated electrical angle of the electrical angle estimating means with respect to a predetermined electrical angle corresponding to the rotational position of the rotor at the generation timing of the pulse signal of the rotation detecting means. Estimated displacement detection that detects a displacement amount and performs necessary correction according to the displacement amount each time the electrical angle estimation means estimates a new estimated electrical angle. And a command creating unit that creates the drive command signals of three phases corresponding to the estimated electrical angle. The driving unit includes three phases created corresponding to the estimated electrical angle. In response to the drive command signal, the MOS type upper drive transistor and the MOS type lower drive transistor of the power supply means are subjected to pulse width modulation operation at a predetermined frequency to drive the upper drive transistor and the lower drive. A brushless motor comprising a transistor, the upper diode, the lower diode, and means for supplying a three-phase sinusoidal drive current smoothed by the drive winding to the three-phase drive winding. 2. The drive command means further obtains an acceleration / deceleration value representing an acceleration / deceleration state of the rotor, and sets the acceleration / deceleration value to the acceleration / deceleration value each time the electrical angle estimation means estimates a new estimated electrical angle. The brushless motor according to claim 1, wherein the brushless motor is configured to include an acceleration / deceleration detection correction unit that performs a required correction according to the correction. 3. A field magnetic section attached to a rotor and having a field pole of a P pole (where P is an even number of 2 or more) formed by using a magnetic flux generated by a permanent magnet; and a field attached to a stator. Three-phase drive windings interlinking with the magnetic flux of the magnetic part, three MOS-type upper drive transistors, three MOS-type lower drive transistors, and three connected in parallel to each upper drive transistor. A plurality of upper diodes and three lower diodes connected in parallel to each lower drive transistor, and a power supply means for supplying power to the three-phase drive windings, and a two-phase current command signal. A current command means to be created, a current detection means for obtaining a two-phase current feedback signal corresponding to the current supplied to the drive winding, and a conversion for inputting the current command signal and the current feedback signal to perform an error detection operation. Comparing means and the conversion comparing hand The upper drive transistor and the lower drive transistor of the power supply means are driven and controlled in response to the output signals of the three phases of the stage to supply a sinusoidal three-phase drive current to the three-phase drive winding. A brushless motor comprising: a drive unit, wherein the conversion comparison unit generates a pulse signal that changes in synchronization with rotation of the rotor; and a pulse signal of the rotation detection unit. Time measuring means for measuring a timing interval synchronized with the rotation of the rotor, and electrical angle estimation for estimating an estimated electrical angle corresponding to the rotational position of the rotor for each time interval shorter than the timing interval of the time measuring means. And a means for estimating the electrical angle with respect to a predetermined electrical angle corresponding to the rotational position of the rotor at the timing of generation of the pulse signal of the rotation detecting means. An estimated deviation detection / correction unit that detects a deviation amount of the estimated electrical angle of a step, and performs a necessary correction according to the deviation amount each time the electrical angle estimation unit estimates a new estimated electrical angle; A conversion feedback means for obtaining a two-phase conversion feedback signal obtained by coordinate-converting the two-phase current feedback signals by using an angle; Control creating means for obtaining a phase control signal, conversion control creating means for obtaining a three-phase conversion control signal by coordinate conversion of the two-phase control signals using the estimated electrical angle, and three-phase conversion control signals Output producing means for obtaining the three-phase output signals of the conversion comparing means in response to the three-phase output signals of the output producing means, and the power supply means. MOS type upper drive transistor And a sine smoothed by the upper drive transistor, the lower drive transistor, the upper diode, the lower diode, and the drive winding. A brushless motor including means for supplying a wavy three-phase drive current to the three-phase drive windings. 4. The conversion comparing means further obtains an acceleration / deceleration value indicating an acceleration / deceleration state of the rotor, and the acceleration / deceleration value is set to the acceleration / deceleration value each time the electrical angle estimation means estimates a new estimated electrical angle. The brushless motor according to claim 3, wherein the brushless motor is configured to include acceleration / deceleration detection correction means for performing required correction according to the correction. 5. The electrical angle estimating means is configured to include means for estimating a new estimated electrical angle by changing the previous estimated electrical angle by a required value at each time interval. Item 5. The brushless motor according to any one of Items 4. 6. The field magnet means comprises field magnet pole means having a plurality of permanent magnets, and ferromagnetic yoke means for embedding and holding the permanent magnets. The brushless motor according to claim 5.