JP3088613B2 - Corner reflector antenna - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention is applied to, for example, a base station antenna of a mobile communication system and uses two or more frequencies in common and increases a beam width in a horizontal plane (hereinafter abbreviated as "beam width"). The present invention relates to a corner reflector antenna.

[0002]

2. Description of the Related Art FIG. 4A is a plan view of a conventional corner reflector antenna. One side edge of the same rectangular reflectors 11 and 12 is connected to each other at an angle α (α <180 °) to form a corner reflector 13. At an equidistant position, a half-wavelength dipole excitation element 14 parallel to both the reflection plates 11 and 12 is arranged. In the figure, the excitation element 1
4 is perpendicular to the paper. A power supply line 15 that is passed through a small hole at the corner position of the corner reflector 13 is connected to the excitation element 14. The distance between the corner of the corner reflector 13 and the excitation element 14 (vertical angle distance of the excitation element) is L, the opening angle of the corner reflector 13 is α, and each of the reflection plates 11, 1
The width of 2 is W.

FIG. 4B shows the calculation of the beam width (width of the main beam of directivity in a plane perpendicular to the reflectors 11, 12 and the excitation element 14) for a frequency of 0.6f to 2f in the conventional corner reflector antenna. This is shown. here,
f is 1 GHz. The antenna is designed to have an equivalent diameter of 100 mmφ in consideration of making the antenna dimension as small as possible. At this time, the opening angle α is 120 °, the width W of the reflecting plate is fixed at 50 mm, and the apex angle distance L is changed. As is clear from the figure, as the frequency increases, the beam width decreases. Here, at the frequency f (= 1 GHz), a beam width of about 125 ° is obtained regardless of the apex angle distance L.
The beam width is 2 ° to 113 ° and does not match. When the apex angle distance L is converted to a wavelength, when L = 60 mm, f is equal to 0.
At 2 wavelengths and 2f, the wavelength becomes 0.4 wavelength, and the apex angle distance in wavelength conversion differs. Therefore, the beam width varies depending on the frequency used. By resonating two different frequencies with such a corner reflector antenna, the beam width in a plane perpendicular to each of the reflectors 11, 12 and the excitation element 14, usually in a horizontal plane, is matched with each other for each frequency. For this purpose, conventionally, as shown in FIG. 4C, two excitation elements 14 and 16 are used, and a 2f excitation element 16 having a high working frequency is set closer to the reflection plates 11 and 12 and its apex angle is set. The beam width is adjusted so that the distance L is equal to 0.2 wavelength, that is, the wavelength-converted apex angle distances of the two excitation elements 14 and 16 are matched.

[0004]

As described above, conventionally, there is a disadvantage that two excitation elements must be arranged in order to operate in a wide band or at two frequencies. An object of the present invention is to provide a corner reflector antenna that uses one excitation element and has a beam width in a horizontal plane that hardly changes over a wider frequency band than before.

[0005]

According to the present invention, a sub-reflector substantially along the radiation direction of the antenna main beam is connected to both ends of the corner reflector. Further, in the invention of claim 2, the opening angle of the reflector is approximately 140 to 150 degrees,
The width of each reflector is approximately 5 / 7L to 2 /
3L, and the width of each sub-reflector is approximately 1 / 7L to 1 / 3L
It is.

[0006]

FIG. 1A shows an embodiment of the present invention, and portions corresponding to those in FIG. 4A are denoted by the same reference numerals. In the present invention, both ends of the corner reflector 13, that is, the reflection plate 1
The sub-reflectors 21 and 22 are connected to the side edges opposite to the connection side edges of the corner reflectors 1 and 12, respectively, and the sub-reflectors 21 and 22 are substantially parallel to the extension direction of the antenna main beam. It is substantially parallel to a straight line connecting the corner and the excitation element 14. In order to make the entire occupied space as small as possible, the sub-reflectors 21 and 22 are located inside the circle inscribed at the three points of the corner and both ends of the corner reflector 13.
Are desirably located, and are not limited to the case where they are parallel to each other as shown in the figure, but may be slightly inwardly facing each other. Let T be the width of each of the sub reflectors 21 and 22. Reflector 1
The effects of the width W of the sub-reflectors 1 and 12, the width T of the sub-reflectors 21 and 22, the opening angle α and the apex angle distance L on the beam width will be described below.

FIG. 2A shows a reflector 1 having the antenna structure of FIG. 1A.
This is a calculation of frequency characteristics with respect to the beam width when the width W of the first and the second 12 is changed.
As in the tendency shown in B, the beam width becomes narrower as the frequency becomes higher. However, W = 8 / 13-10 / 13
At L, the beam width tends to increase again as the frequency increases. From the above, W = 8/13 to 10 /
13L is required. Here, T = 2 / 13L, opening angle α = 140 °, and L is the apex angle distance.

FIG. 2B shows the relationship between the width T of the sub-reflectors 21 and 22 and the beam width. When there is no sub-reflectors 21 and 22 (T = 0), the beam width becomes narrower as the frequency increases. ing. However, from T = 2 / 13L to 4 / 13L, the beam width tends to increase again as the frequency increases, indicating that a directivity of 120 ° beam width can be obtained. From the above, the width T of the sub-reflectors 21 and 22 is 2 / 13L.
~ 4 / 13L is required. However, the reflection plates 11 and 12
, W = 10 / 13L, opening angle α = 140 °, and L is the apex distance.

FIG. 3A shows the frequency characteristics of the beam width with respect to the opening angle α, where the opening angle α is 140 ° to 150 °.
It turns out that it is optimal. However, the width W of the reflection plate is 1
0 / 13L, the width T of the sub-reflector T = 2 / 13L, L is the apex angle distance. FIG. 3B shows the relationship between the apex angle distance L and the beam width. When the apex angle distance L is 80 mm, the beam width increases as the frequency increases. However, when the frequency increases from 60 to 70 mm, the beam width increases again. Becomes wider or the beam width hardly changes. When the apex angle distance L is reduced to 50 mm, the beam width tends to be extremely wide as the frequency increases with respect to the frequency. From the above, the apex angle distance is 60 to 70 mm.
It will be optimal. However, the width W of the reflection plate is 50 mm, the width T of the sub-reflection plate is 10 mm, and the opening angle α is 140 °.

From the above points, preferred numerical examples are as follows. Reflector width W = 50 mm, auxiliary reflector width T = 10 mm, opening angle α = 140 °, vertex angle distance L = 60
mm at this time, the beam width at f is 122 °, and at 2f is 120 °, it is understood that there is a dimension where the beam width hardly changes. At this time, the excitation element 14
Can be realized by using a dual frequency antenna equipped with a parasitic element. As such a dual-frequency antenna, for example, as shown in FIG. 1B, a ground layer 25 is formed on one surface of one end of a rectangular dielectric plate 24, and a strip extending from the edge of the dielectric plate 24 is formed at the center of the other surface. A line (not shown on the reverse side in the figure) is formed, and two parallel lines 26 each having one end connected to the strip line and the ground layer 25 are formed to face each other via the dielectric plate 24, and the two parallel lines are formed. At one end of 26, one element 27 of a dipole is formed on one surface of the dielectric plate 24 at right angles thereto, and two parallel lines 26 are formed on the other side and the other element 2 of the dipole is formed.
8 is formed on the other surface of the dielectric plate 24, and a parasitic element 29 is formed on the dielectric plate 24 in close proximity to the dipole elements 27 and 28. At the corner of the corner reflector 13, the earth layer 25 side is passed, and the dipole elements 27 and 28 are located almost at the location of the excitation element 14. The light is radiated at the resonance frequency determined by the dipole elements 27 and 28, and is radiated at the resonance frequency determined by the parasitic element 29.

The dual-frequency antenna is described in, for example, Karimitsu and Ebisu, "Characteristics of Printed Dipole Antenna with Parasitic Element", IEICE Tech. As the excitation element 14, a broadband antenna may be used in addition to the dual frequency antenna.

[0012]

As described above, according to the present invention, by providing the sub-reflector, it is possible to obtain an antenna with a wide frequency characteristic with respect to the beam width and a reduced size using a single excitation element. it can.

[Brief description of the drawings]

FIG. 1A is a plan view schematically showing a corner reflector antenna of the present invention, and FIG. 1B is a perspective view showing an example of a dual-frequency printed dipole antenna with a parasitic element.

2A is a frequency characteristic diagram of a beam width with respect to a reflector width of the corner reflector antenna of the present invention, and FIG. 2B is a frequency characteristic diagram of a beam width with respect to a sub-reflector width of the corner reflector antenna.

3A is a frequency characteristic diagram of a beam width with respect to an opening angle of the corner reflector antenna of the present invention, and FIG. 3B is a frequency characteristic diagram of a beam width with respect to a vertex angle distance of the corner reflector antenna.

4A is a plan view schematically showing a conventional corner reflector antenna, FIG. 4B is a frequency characteristic diagram of a beam width with respect to an apex angle distance, and FIG. 4C is a plan view showing a conventional dual-frequency corner reflector antenna. .

──────────────────────────────────────────────────続 き Continuation of the front page (72) Kunio Takema, Inventor 1-126 Matsumotocho, Higashimatsuyama-shi, Saitama (72) Inventor, Tamaki Ebata 2-33-8, Higashi-Oizumi, Nerima-ku, Tokyo Fuller 8101 (56) References JP-A-62-29205 (JP, A) JP-A-60-214606 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H01Q 19/10 H01Q 15 / 18

Claims (2)

(57) [Claims] An opening angle of each one side edge is substantially 1
It is connected with an opening angle of 40 to 150 degrees, and the width is almost
The first and second reflectors, which are 5 / 7L to 2 / 3L, are connected to the other side edges of the first and second reflectors, respectively.
Corner reflector antenna from the connected position.
Approximately parallel to the main radiation beam direction of the
The first and second sub-reflectors extended by 1 / 3L are substantially equidistant from the first and second reflectors, and
1. A distance L away from the position where the second reflector is connected
A corner reflector antenna , comprising: an excitation element arranged at a position . 2. A method according to claim 1 Symbol mounting corner reflector antenna, wherein said driven element is a Dual Band Printed Dipole Antenna with Parasitic Elements.