JP3058820B2 - Demodulation method for spread spectrum communication and demodulation device using the method - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a code division multiple access (CDMA) used for mobile communication such as a personal communication system (PCS) and a digital cellular system.
In particular, the present invention relates to a demodulation device used for CDMA communication.

[0002]

2. Description of the Related Art Conventionally, a radio terminal (hereinafter referred to as a mobile station) used for CDMA communication and a receiving section of a base station have been operated as follows. On the transmitting side, a signal transmitted after performing either phase modulation or frequency modulation and spread modulation is received by an antenna for both transmission and reception.
Phase demodulation or frequency demodulation is performed by a signal output from the carrier frequency oscillator of the receiving unit and having the same frequency as that used for phase modulation or frequency modulation.
Thereafter, despreading is performed using a correlation detector.

[0003] The carrier frequency oscillator of the receiving section needs to output a demodulation signal with high frequency accuracy for the following reasons.

In CDMA communication, adjacent cell areas use the same frequency band. Therefore, in a forward link of CDMA communication (transmission from a base station to a mobile station), one mobile station (hereinafter, referred to as a tracking station) is adjacent to a cell area covered by a base station that must establish synchronization. Thus, there is a cell area covered by a base station using the same frequency band (hereinafter, referred to as an interference station).

[0005] Here, the base station with which synchronization must be established is usually located closest to the following station,
The base station with the highest received power from that base station,
Hereinafter, it is called a desired station.

[0006] Also, a large number of mobile stations exist in one cell area, and one base station covering this cell area has as many receivers (specifically, correlation detection) as the line capacity allows. Container). Therefore, in the reverse link (transmission from the mobile station to the base station), when one correlation detector is considered as one station (a tracking station in the reverse link), synchronization is established for the tracking station and communication is performed. There is a mobile station (desired station) that needs to perform the operation, and a mobile station (interfering station) that does not need to establish synchronization.

In the above environment, since the desired station and the interfering station use the same frequency band as a carrier, in order for the following station to distinguish between the desired station and the interfering station,
The use of a high-precision carrier frequency oscillator and the good characteristics of the PN code used for spreading and despreading;
Two conditions are required.

The reason for using a high-precision carrier frequency oscillator is as follows. When the reception signal is phase-modulated, for example, when the reception unit performs phase demodulation on the reception signal, the signal output from the carrier frequency oscillator is multiplied by the reception signal. If the accuracy of the frequency of the signal output from the carrier frequency oscillator is poor, the difference between the phase of this signal and the phase of the received signal increases, and the magnitude of the output obtained by multiplication decreases. The demodulated signal is input to the correlation detector, and the desired station and the interfering station are identified based on the correlation value output from the correlation detector. If the demodulation output is reduced, the correlation value is also reduced.
Therefore, the difference between the correlation value for the signal from the desired station and the correlation value for the signal from the interfering station, which is obtained by using the correlation detector, also becomes small, and it becomes difficult to discriminate the desired station from the interfering station.

Conventionally, as a countermeasure against this, a carrier frequency oscillator capable of outputting a signal of a predetermined frequency with high accuracy has been used to make the frequency coincide with the receiving side. In the case of phase demodulation, it is necessary to match the phases. However, as a method of matching the phases, a phase shift is detected from the intensity of the signal after the phase demodulation, and the phase timing of the carrier frequency oscillator is adjusted. Was being done.

The reason why the characteristics of the PN code used for spreading and despreading are required to be good is as follows. The desired station and the interfering station are distinguished by different PN codes, but the good characteristics of the PN codes used for spreading and despreading mean that the cross-correlation value between different PN codes is as close to zero as possible. The more the cross-correlation value is closer to zero, the more erroneous recognition between the desired station and the interference station can be prevented. Poor characteristics mean that the cross-correlation value is large. When the cross-correlation value is large, the difference between the correlation value for the signal from the desired station and the correlation value for the signal from the interfering station, which is obtained using the correlation detector of the receiving unit, becomes small, and the interference with the desired station is reduced. Station identification becomes difficult.

As described above, when either of the above two conditions is not satisfied, the correlation value for the signal from the desired station and the correlation value for the signal from the interfering station, which are obtained by using the correlation detector of the receiving unit, are obtained. The difference from the correlation value becomes small, and it becomes difficult to distinguish between the two.

When there are a plurality of interfering stations, when the correlation value of each interfering station is compared with the correlation value of the desired station, the correlation value of each interfering station becomes equal to the correlation value of the desired station. If the correlation value for the desired station is small even if it is clearly smaller than the value (even if the correlation value between the desired station and one interfering station is discernible), the And the difference between the correlation value of the desired station and the correlation value of the desired station are reduced, and as a result, the desired station cannot be distinguished.

[0013]

As described above, the demodulator according to the prior art requires a high-precision carrier frequency oscillator, and the configuration of the high-precision carrier frequency oscillator is complicated. For this reason, a high-precision carrier frequency oscillator is expensive, and there is a problem that the cost of the terminal and the base station increases.

An object of the present invention is to solve the above-mentioned drawbacks of the prior art and to provide a terminal and a base station which do not require a high-precision carrier frequency oscillator.

[0015]

SUMMARY OF THE INVENTION In order to solve the above-mentioned problems, the present invention relates to a spread spectrum communication demodulator for receiving a spread modulated received signal and demodulating the received signal. Phase-locked loop means for receiving the signal, and receiving the signal output from the phase-locked loop means, performing correlation detection, and outputting a detection output. The loop gain of the loop means depends on the detection output of the correlation detector.

[0016]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment in which a demodulator for spread spectrum communication according to the present invention is applied to a mobile station (mobile phone) for CDMA communication will be described in detail with reference to the accompanying drawings.

An embodiment of the demodulator 10 for spread spectrum communication according to the present invention will be described in detail with reference to FIGS.
This device 10 receives a transmission signal from base station 12, performs phase demodulation and despreading, and finally obtains a voice signal and the like.
When the power of the device 10 is turned on, the device 10 first
It is necessary to synchronize with a transmission signal from the base station 12 located closest to the device 10. The transmission signal also includes a pilot signal for establishing synchronization. In this embodiment, a pilot signal is used to synchronize with a transmission signal from a base station. After the synchronization is established, the processing for obtaining the audio signal is performed.

FIG. 1 is a block diagram showing a portion related to the present invention, that is, a portion related to phase demodulation and synchronization establishment, of a demodulator 10 for spread spectrum communication of a mobile station. In the figure, other parts of the demodulation device and the transmission device of the mobile station that are not necessary for understanding the present invention are omitted.

First, an overview of the demodulator for spread spectrum communication shown in FIG. 1 will be described. The present apparatus 10 receives a spread-modulated and phase-modulated signal 20 from a transmitting side (base station) 12 via a propagation path 14 and an antenna (not shown). In the synchronization establishment stage, the frequency and phase of the carrier frequency oscillator 16 match the frequency and phase of the received signal 20 in the synchronization establishment stage, and after the synchronization, the phase locked loop functioning as a phase demodulator (also called a phase discriminator) (PL
L) having 18. Since the PLL 18 has such a function, the PLL 18 can be configured using a carrier frequency oscillator (voltage controlled oscillator (VCO)) 16 having low frequency stability. The received signal 20 is input to the PLL 18. Carrier frequency oscillator
After the 16 frequencies and phases are accurately synchronized with the frequency and phase of the received signal 20, a signal 22 that has been accurately phase-demodulated is obtained as an output of the PLL 18.

The output 22 of the PLL 18 is input to a traffic signal correlation detector 38 and a pilot signal correlation detector) 24. The traffic signal correlation detector 38 performs despreading of the phase-demodulated traffic signal, and outputs a correlation value 28 as a result of the despreading. It should be noted that the traffic signal including the voice data is transmitted from the base station 12 after the synchronization is established, and that the correlation value 28 is significant after the synchronization is established. The correlation value (digital data) 38 is multiplied by the output 36 of the pilot signal correlation detector 24 by the multiplier 36,
Thereafter, processing such as error correction is performed. As a result of these processes, digital audio signals or digital data (such as facsimile data) are obtained.

The multiplier 36 outputs the digital data after the phase demodulation.
This is for correcting the phase shift included in 38, and corrects the phase shift by multiplying the data 38 by the output 26 of the pilot signal correlation detector 24. The principle by which the phase shift is corrected by this is not directly related to the present invention, and a description thereof will be omitted. Traffic signal correlation detector
Since 38 and the multiplier 36 are not directly related to the present invention, further detailed description will be omitted.

The pilot signal correlation detector 24 in FIG. 1 despreads the pilot signal and outputs a correlation value 26 which is the result of the despreading. The output (correlation value) 26 of the pilot signal correlation detector 24 is input to the PLL 18 via the compensator 30.

When the received signal 22 demodulated with high accuracy is input to the correlation detector 24, the received signal from the base station (desired station) located at a short distance from the tracking station is more received than the received signal from the interfering station. Because the intensity is high, the compensator
The correlation value 26 output to the PLL 18 via 30 is also larger for the tracking station. The correlation value 26 is calculated by the multiplier 62
Since the signal is taken into the PLL 18, the loop gain of the PLL 18 changes depending on the correlation value 26. That is, the loop gain of the PLL 18 relating to the signal from the desired station is larger than the loop gain of the PLL 18 relating to the signal received from the interfering station.

In general, the larger the loop gain of the PLL, the shorter the time required to establish synchronization with the input signal, and the position of the input signal and the output signal of the VCO after the synchronization is established. It is known that the phase difference (stationary phase error) is reduced.

Therefore, when a received signal with a large loop gain and a received signal with a small loop gain are input, the PLL 18 quickly follows the received signal with a large loop gain. Utilizing this, the determination circuit 32 to which the phase discrimination output 28 of the PLL 18 is input, at a point in time when a certain time has elapsed, if the phase discrimination output 28 of the PLL 18 is smaller than a predetermined value, synchronization is established. Judge.

The instruction circuit 40 instructs the controller 42 in the determination circuit 32 to initialize the inside of the controller 42 when the power of the mobile station is turned on.
Start 16

Next, details of each section of the demodulation device 10 will be described. Before going into the detailed description of each part, using FIG.
In the case of the present embodiment, how modulation is performed on the transmitting side 12 will be described in a range necessary for understanding the present embodiment. FIG. 2 is a block diagram showing a part of the base station 12 related to the transmitting device. In the figure, other parts of the transmission device that are not necessary for understanding the present embodiment and the demodulation device of the base station are omitted.

Input data to be transmitted at the base station 12
64 is spread-modulated by the traffic signal spreading device 44 using a PN code. Next, the PN code used in this embodiment will be described.

In CDMA communication, TDMA (Time Division
multiple Access) communication and FDMA (Frequency Division mu)
Unlike ltiple Access) communication, the frequency used between adjacent cells is the same. Cell identification, that is, base station identification, is performed using the fact that the PN code used for spreading modulation on the forward link differs for each base station. In this embodiment, different PN codes are one M sequence
(Maximum Length Code). Specifically, M sequences are shifted at different times for each base station. In this embodiment, the property that the correlation value between the M sequences shifted in time is zero is used. In this method, a plurality of base stations synchronize with the PN
The code needs to be transmitted and this scheme is called synchronous CDMA.

The PN code is also used in the pilot signal spreading device 46 to spread-modulate the pilot signal. The pilot signal is a signal used by the demodulator 10 of FIG. 1 to establish synchronization, and is always “1”. The outputs of the two spreading devices 44 and 46 are added by an adder 48. Thereafter, the output of the adder 48 and the carrier signal output from the carrier oscillator 52 are multiplied by the multiplier 50 to perform phase modulation. After the phase modulation, the signal is transmitted from an antenna (not shown) and transmitted to the demodulation device 10 via the propagation path 14.

As a phase modulation method, PSK (Phase Shift
Keying) and QPSK (Quadrature Phase Shift Keying). The present invention can be applied to any of the phase modulation schemes, and can also be applied to the frequency modulation scheme. The phase modulation of this embodiment employs the QPSK scheme. The QPSK method generates in-phase component data and quadrature component data from digital data to be transmitted, and performs phase modulation on each of the in-phase component data and the quadrature component data using two carrier signals having phases different from each other by 90 degrees. Things.

In FIGS. 1 and 2, of the transmitting device and the demodulating device used for the forward link of the CDMA communication,
Only the part according to the present invention, that is, the part related to the pilot signal and the in-phase component data is shown. Therefore, the input data 64 shown in FIG. 2 is in-phase component data. As shown in FIG. 2, the in-phase component data 64 is phase-modulated after being added to a pilot signal used for establishing synchronization.

As a phase demodulation method of quadrature component data not shown in FIG. 1, for example, the output 54 of the VCO 16 in FIG. There is a method of generating a signal whose phase is advanced by 90 degrees and multiplying the received signal 20 by the generated signal.

Returning to FIG. 1, each part of the demodulator 10 will be described in detail. First, the PLL 18 will be described. PLL 18
Accepts the received signal 20. The received signal 20 input to the PLL 18 is multiplied by an output 54 of the VCO 16 by a multiplier 74. The multiplier 74 has a function of obtaining a phase difference between the received signal 20 and the output 54 by performing multiplication. Therefore, after the synchronization is established, the output (baseband signal) 56 of the multiplier 74 becomes a signal obtained by phase demodulating the received signal 20.

The output of the multiplier 74 is sent to the low-pass filter 60. By the multiplication by the multiplier 74, a high-frequency component is simultaneously generated in addition to the phase difference component.
Is cut by Only a signal whose phase is the phase difference between the phase of the received signal 20 and the phase of the output 54 is a low-pass filter.
Output from 60.

The signal 22 after the high frequency component has been cut is
The signal is sent to the loop filter 66 and also sent to the traffic signal correlation detector 34 and the pilot signal correlation detector 24.

In the pilot signal correlation detector 24, despreading, ie, multiplication by the same PN code as the PN code used for the spread modulation shown in FIG. 2, is performed, and a correlation value 26 is obtained.
The correlation value 26 is input to the multiplier 62 via the compensator 30 for compensating for the frequency dependence of the correlation detector 24. Details of the correlation detector 24 and the compensator 30 will be described later.

The multiplier 62 multiplies the output of the compensator 30 and the output of the loop filter 66 by changing the magnitude of the loop gain of the PLL 18 according to the magnitude of the correlation value 26 and
This is for the purpose of synchronizing with the 26 large stations, that is, the desired station earlier than the interfering station. The relationship between the magnitude of the loop gain and the speed of synchronization will be described later.

After the output of the compensator 30 and the output of the loop filter 66 are multiplied, they are DC-amplified by the amplifier 68. amplifier
The output 28 of 68 is sent to the decision circuit 32 and the multiplier 72. The determination circuit 32 is a circuit that determines whether synchronization has been established based on the output 28. When the determination circuit 32 determines that synchronization has not been established within a predetermined time, it changes the frequency band,
Output an impulse signal to establish synchronization
Output as 70. Details of the determination circuit 32 will be described later.

The output 70 is the output of the amplifier 68 by the adder 72.
28 is added. The signal 58 after the addition is input to the VCO 16. The VCO 16 is a circuit that outputs a signal having a frequency proportional to the input voltage signal 58. Since the transfer function is an integration circuit represented by an integration element (1 / s), when an impulse signal is input from the determination circuit 32, the frequency of the output signal jumps. As a result, a process for establishing synchronization is performed in the new frequency band after the jump.

Next, the pilot signal correlation detector 24 will be described. FIG. 3 shows an example of the PN code used in the correlation detector. The PN code takes a value of “+1” or “−1” and is multiplied by transmission data to perform spread modulation. In FIG. 3, T is the length of one chip of the PN code, and is called a spread chip length. mT is the length of one bit of transmission data, and is called a bit section. As shown in the figure, the length of one bit of transmission data usually corresponds to a plurality of chips. Further, the length (number of chips) of the PN code when calculating the correlation value in the correlation detector is called the correlation length of the correlator, and is represented by n hereinafter. In the present embodiment, the correlation detector 24
It is assumed that the length of the PN code stored in the ROM in is the same length n as the correlation length. Note that the length of the PN code may be longer than the correlation length.

The correlation detector 24 includes, for example, a matched filter
According to the ing method, the input signal 22 is multiplied by a chip unit with the PN code stored in the ROM in the correlation detector 24 to obtain a sum (correlation value) 26 over the correlation length of the multiplied value.
FIG. 4 shows an embodiment of the correlation detector 24 using the matched filtering method. The correlation detector 24 stores the signals 22 sequentially input to the correlation detector 24 by a correlation length in a shift register 120, and obtains a correlation value 122 for each time unit equal to or less than the chip length T.

The correlation value 122 is obtained as follows. The output of the shift register 120 is extracted by taps 124 and 126.
At that time, the tap 124 to be added to the adder 128 and the adder
A tap 126 to be subtracted from 128 is stored in the ROM in the correlation detector 24.
Is determined according to the PN code stored in. In FIG. 4, the correlation length (= the number of stages of the shift register) n is 10, and the PN code stored in the ROM in the correlation detector 24 is "1,-".
1, -1, 1, 1, -1, 1, 1, -1, -1, 1 ". At this time, adder 1
Taps 124 to be added to 28 are 1, 4, 5,
Taps 126 to be subtracted from adder 128
Are the second, third, sixth, eighth, and ninth from the right side of FIG.

When this calculation is performed for each time unit shorter than the chip length T, a large correlation value 122 is obtained when the timing of the PN code of the input signal 22 and the PN code of the correlation detector 24 match. The PN code of the input signal 22 is a correlation detector
Since the correlation value 122 is calculated while sequentially shifting the position of the PN code included in the PN code by the shift register 120, once during one cycle (time corresponding to the length n of the PN code) Is because the timings match. In addition,
In the synchronous CDMA system as in the present embodiment, the coincident timing differs for each base station.

The peak of the correlation value 122 is determined by the peak determination circuit 13
0 is detected and output as the correlation value 26. The peak determination circuit 130 performs a peak determination on the correlation value 122 during one cycle, and outputs the peak value obtained during one cycle as the correlation value 26. As a result, the correlation value 122
Is that the magnitude fluctuates during one cycle, but the correlation value is 26
Is a constant value over one cycle.

The magnitude of the correlation value 26 increases as the distance from the base station 12 decreases. Further, it depends on whether the reception signal 22 and the output signal of the VCO 16 are synchronized. The shift register 120 of the correlation detector 24 is provided with a CCD (Charge Coupled).
Device or SAW (Surface Acoustic Wave) element can be configured as the main component.

Next, a method for canceling the frequency dependency of the correlation detector 24 will be described. For this purpose, the frequency dependence of the correlation detector 24 is determined. Since the frequency dependency can be represented by a transfer function, a transfer function representing the processing performed by the correlation detector 24 is obtained.

In the process of obtaining the correlation value performed by the correlation detector 24, the product of the signal 22 and the PN code built in the correlation detector 24 is calculated in a time interval [t-nT, t] (t is an arbitrary time). ) As an integration interval.

When the transfer function is obtained according to the definition of the transfer function using the signal 22 as the input of the transfer function, the transfer function of the correlation detector 24 is (1-EXP (-nTs)) / s. This equation includes s, which means that the correlation detector 24 has frequency dependence. If the transfer function has frequency dependency, a part of the frequency spectrum of the input signal will be cut off and output, and the correlation detector
24 output decreases. Further, since having the frequency dependency deteriorates the characteristic of wideband spreading which is a feature of the CDMA communication, it is desirable that the output of the correlation detector 24 does not have the frequency dependency.

Therefore, in this embodiment, in order to eliminate the frequency dependence of the correlation detector 24, a compensator 30 is provided. The compensator 30 may be any as long as its transfer function cancels a term including s included in the transfer function of the correlation detector 24. Regarding the connection method between the compensator 30 and the correlation detector 24,
The compensator 30 can be provided in parallel with the correlation detector 24, or can be provided in series. In the present embodiment, the compensator 30 is provided in series because the circuit of the compensator 30 is simpler than the parallel circuit. The transfer function of the correlation detector 24 is (1-EXP (-
nTs)) / s.
May be the reciprocal of (1-EXP (-nTs)) / s.
One embodiment of the compensator 30 having such a transfer function is shown in FIG.
Shown in

In FIG. 5, 301, 302, and 303 are adders, 304 and 306 are integrators (the transfer functions are 1 / s), and 306 is a delay circuit that delays by time nT (the transfer function is EXP ( -n
T)).

Next, FIG. 6 is a block diagram showing the demodulation device 10 shown in FIG. 1 expressed by the above transfer function. This device 10
FIG. 6 shows the phase of each signal in order to perform phase demodulation. The phase comparator 76 of FIG. 6 is a multiplier of FIG.
This corresponds to a circuit portion composed of 74 and the low-pass filter 60. The phase comparator 76 including the multiplier 74 and the low-pass filter 60 is provided with an adder 78 that determines the phase difference between the two input signals when focusing only on the phases of the signals input to and output from the phase comparator 76. And the circuit 80 for calculating the sin of the obtained phase difference.

This will be described. Receive signal 20 is V _PM (t),
Let output 54 be V _C (t),

[0054]

It is assumed that V _PM (t) = sin (θ (in (t))) V _C (t) = sin (θ (out (t))). Here, θ (in (t)) is the phase of V _PM (t), and θ (o
ut (t)) is the phase of V _C (t). When the received signal 20 and the VCO 16 are multiplied by the multiplier 74, the phase sum θin (t) + θout
(t) and the phase difference φ (t) = θin (t) −θout (t)
Is generated. The signal having the phase sum θin (t) + θout (t), which is the high-frequency component, is cut by the low-pass filter 60. Thus, the phase comparator 76 outputs the signal 22 (sin (φ (t))) having the phase difference φ (t) 84 as the phase. The signal 22 is sent to the correlation detector 24 and the loop filter 66.

In the transfer function of the correlation detector 24 shown in FIG.
P is a spreading gain by a spreading code. The denominator nT is the integration time. Usually, the transfer function is divided by the integration time in order to normalize the transfer function. C (s) is a transfer function of the compensator 30, one example of which is as shown in FIG.

The loop filter 66 may be a low-pass filter. In this embodiment, a lag-lead type loop filter often used for a PLL is shown. The transfer function of this filter is (s + a) / (s + b).

The amplifier 68 performs DC amplification, and its gain is K. The output of amplifier 68 is input to VCO 16.
By the way, since the VCO 16 is an integrator when viewed as a transfer element, the input of the VCO 16 is equal to the output of the VCO 16 differentiated. For this reason, in FIG. 6, the input of the VCO 16, that is, the output of the amplifier 68 is shown as a derivative of the output θout (t) of the VCO 16.

The correlation detector 24, the compensator 30, and the multiplier 62 can be regarded as an amplifier together. The gain P 'of the deemed amplifier is determined by the transmission element of the correlation detector 24 and the compensator 30.
By multiplying by the transfer element of Therefore,
P '= P / (nT). Regarding this gain, since the spreading gain P for the desired station is larger than the spreading gain P for the interfering station, the gain P ′ for the desired station is obtained.
Is larger than the gain P 'for the interfering station.

The gain P ′ and the gain K of the amplifier 68 are combined.
The loop gain of the entire PLL 18 is K · P ′. Similarly to the gain P ′, the loop gain K · P ′ of the desired station is larger than that of the interfering station.
It is larger than K ・ P '.

FIG. 7 shows an equivalent circuit of only the portion related to the PLL circuit in FIG. 6 using the loop gain K · P ′. As can be seen from the figure, the PLL circuit of this embodiment is
This is equivalent to a PLL circuit 82 including an amplifier 82 having a loop gain K · P ′. The PLL circuit of this embodiment has different loop gains K · P ′ between the desired station and the interfering station. As a result, the equivalent PLL circuit of this figure follows the desired station faster than the interfering station.

Hereinafter, this will be described. Time required from the start of synchronization establishment to the end of synchronization establishment (synchronization establishment time)
As for Ts, a PLL having a lag-lead type loop filter as shown in FIG. 5 is analyzed in "How to Use PLL-IC-Electronics Selection", 1991, Masayasu Hata, Keisuke Furukawa, and Akiba Publishing. According to FIG. 7,
First, a differential equation relating to the phase difference φ (t) 84 between the received signal 20 from the tracked station (desired station or interference station) and the output signal 54 of the carrier frequency oscillator 16 of the tracked station is determined. It looks like this:

[0062]

Φ ″ = −e ・ φ′-e ・ Δω -K ・ P ′ ・ φ'cosφ -K ・ P '・ a ・ sinφ where φ''is φ (t) twice in time The derivative,
φ ′ is obtained by differentiating φ (t) once with respect to time.
Note that φ ′ is equal to the frequency difference between the received signal 20 and the output signal 54. Δω is an initial frequency difference at the start of synchronization establishment.
e and a are constants included in the transfer function of the loop filter 66 described above. Equation 2 is described as equation (5.18) on page 62 of the reference. According to this document, the synchronization establishment time Ts is obtained as follows based on Equation 2.

[0063]

(Equation 3) From Equation 3, the synchronization establishment time Ts becomes shorter when the loop gain K · P ′ is larger. Therefore, the synchronization establishment time is shorter when tracking the desired station is started than when tracking the interfering station is started. When synchronization is established, φ
Since the value of (t) is small, it can be determined whether or not the phase discrimination output 28 is small.

The determination circuit 32 in FIG. 1 uses this to determine whether synchronization has been established. That is, the timer
The time is measured by 86, and the magnitude of the phase discrimination output 28 after a certain period of time is checked to determine whether or not synchronization with the desired station has been made. Next, the determination circuit 32 will be described.

The determination circuit 32 receives the phase discrimination output 28 for determining the establishment of synchronization.
Detects that the operation for establishing synchronization has started,
Used to determine whether synchronization has been completed. Therefore, the phase discrimination output 28 input to the determination circuit 32 is input to the trigger circuit 88 and the controller 42.

The trigger circuit 88 detects that the operation for establishing synchronization has started, and notifies the timer 86 of the start of the operation for establishing synchronization. The trigger circuit 88 differentiates the input phase discrimination output 28 to detect the start of the operation. The phase discrimination output 28 is output when the demodulator 10 is turned on and the VCO
When the frequency band of the output of the VCO 16 is largely changed by the controller 42 in order to change the frequency band for which synchronization is to be established, the frequency greatly changes. Therefore, the trigger circuit 88, which is a differentiating circuit, can detect the start of the operation at these times, and outputs a signal 90 indicating the start to the timer 86.

The timer 90 starts time measurement in response to the signal 90. When a predetermined time has elapsed and the synchronization can be determined, the timer 90 outputs a timer end signal 92 indicating that. This is a circuit for outputting to the controller 42. This predetermined time is equal to or longer than the time for establishing synchronization with the desired station,
This time is shorter than the time for establishing synchronization with the interference station. This value is stored in a memory (not shown) inside the timer 90.

The time stored in the memory is usually fixed, but the mobile station may reset itself in consideration of the communication state. Alternatively, the time may be reset based on transmission data that specifies the time and is transmitted from the base station.

The controller 42 receiving the signal 92 from the timer 90 determines whether or not the synchronization has been established. If the synchronization has not been established, the controller 42 jumps the frequency of the output signal 54 of the VCO 16 and jumps the frequency. When synchronization is not established even after performing the predetermined number of times, a predetermined display is performed.

Upon receiving the signal 92, the controller 42 compares the phase discrimination output 28 at that time with a threshold value incorporated in the controller 42 to determine whether or not synchronization has been established. When the discrimination output 28 is smaller than the threshold value, it is considered that synchronization has been established.

On the contrary, when the discrimination output 28 is larger than the threshold value, the value of a counter (not shown) built in the controller 42 which records the number of times the frequency is jumped is checked. If the value of the counter is equal to or more than the allowable number of jumps (maximum number of times) stored in a memory (not shown) in the controller 42, it is determined that synchronization cannot be established even if the frequency band is changed a predetermined number of times. I do.

At this time, the display unit (not shown) of the mobile station displays that the desired station could not be found. Specifically, a display such as "out of communication service available area" is displayed on a display or the like of the mobile station.

Here, the maximum number of times stored in the memory is, for example, a value of about 10. When the value of the counter is less than the maximum number, the controller 42 increases the counter by one. After the increase, the controller 42 outputs the impulse signal as the output signal 70.

The phase discrimination output 28 and the output signal 70 which is an impulse signal are added and added to the VCO 16. As a result, the frequency of the output 54 of the VCO 16 greatly changes, and the phase discrimination output 28 sharply changes. As a result, the trigger circuit 88
Output the signal 90 again.

Next, the instruction circuit 40 will be described. When the power of the mobile station is turned on, the instruction circuit 40 detects the power-on, sends a start signal 96 to the controller 42, and instructs the controller 42 to perform an initial setting. Upon receiving this instruction, the controller 42 resets a counter inside the controller 42. The instruction circuit 40 outputs an instruction for initial setting,
After the time required for the initialization by the controller 42 has elapsed, the power supply voltage is applied to the VCO 16 to activate the VCO 16.

Next, the operation of the apparatus 10 will be described with reference to the flowchart of FIG. When the power of the mobile station is turned on, the instruction circuit 40 detects this. Indication circuit 40 detected
Sends a start signal 94 to the controller 42 (step 101).
Upon receiving the start signal 94, the controller 42 initializes the counter. The instruction circuit 40 then applies the power supply voltage to the VCO 16. The VCO 16 starts outputting (step 101). VC
Since O16 has been activated, the phase discrimination output 28 changes abruptly (step 102). The trigger circuit 88 operates in response to this change, and outputs a signal 90 which is a differential value of the phase discrimination output 28 to the timer 86.
(Step 103). The timer 86 starts measuring time (step 104). When the predetermined time has elapsed, the timer 86 sends a timer end signal 92 to the controller 42 (step 105).

Upon receiving the timer end signal 92, the controller 42 compares the phase discrimination output 28 with a threshold value.
(Step 106). When the discrimination output 28 is smaller than the threshold value, it is determined that synchronization has been established (step 107). Discrimination output 28
Is larger than the threshold, the value of the counter is compared with the maximum number of times (step 108). If the value of the counter is equal to or greater than the maximum number, it is regarded that synchronization establishment has failed, and a display indicating that the desired station has not been found is displayed on the display unit of the mobile station (step 109).

If the value of the counter is less than the maximum number, the value of the counter is increased by 1 (step 110). Next, the controller 42 outputs an impulse signal as the signal 70. This impulse signal is phase-discriminated by the adder 72.
28 is added (step 111). When the discrimination output to which the impulse signal is added is input to the VCO 16, the VCO 16
Greatly changes the frequency of the output signal 54 (step 11).
2).

As a result, the phase discrimination output 28 changes greatly (step 102). In response to this change, the trigger circuit 88 as a differentiating circuit operates again (step 103).

As described above, according to the demodulation device according to the present embodiment, since a carrier frequency oscillator with low accuracy is incorporated in the PLL, it is possible to accurately synchronize with the received signal.

The PLL in the circuit related to the synchronization of the demodulation device
Is made a function of the output of the correlation detector, so that it can be correctly synchronized with the desired station. If phase demodulation is performed only with a PLL without using a correlation detector, it is possible that the desired station and the interfering station cannot be distinguished from each other, so that they may synchronize with the interfering station. By incorporating the correlation detector in the PLL, it is possible to synchronize only with the desired station.

Further, by using a correlation detector in combination with a PLL, the gain of the correlation detector is effectively increased, and even when a PN code having poor characteristics is used,
The desired station and the interfering station can be distinguished.

In this embodiment, as a method for determining whether or not synchronization has been established, a method for evaluating the magnitude of the phase discrimination output after a predetermined time has elapsed is adopted. However, the present invention is not limited to this. Therefore, the determination can be made without waiting for a certain time to elapse. For example, the determination circuit may repeat the integration of the phase discrimination output for a certain time period, and determine that synchronization has been established when the integrated value becomes smaller than the certain value.

The following advantages are obtained by performing integration over a fixed time interval. That is, depending on the circuit parameters of the PLL, the phase discrimination output oscillates, and the vibration may be so large that it is erroneous to judge the establishment of synchronization if it is determined only by the instantaneous value of the phase discrimination output. Synchronization establishment can be correctly determined.

Further, in the above embodiment, the case where the M-sequence is used as the PN code has been described. However, the present invention is not limited to this.
Any device that is used for communication can be used.

Further, the code string used for the spread modulation is
The present invention is not limited to the PN code, and any code having an orthogonal relationship can be used in the present invention. Such a code is, for example, a Walsh code.

Further, in the above embodiment, the frequency and the phase of the received signal are made to match the frequency and the phase of the carrier frequency oscillator in the demodulator by the PLL, but this device uses the pilot signal. Therefore, the degree of phase matching may be poor.

Here, a method of correcting a phase shift using a pilot signal will be described. FIG. 9 shows how a pilot signal, a traffic signal and a control signal are modulated on the transmission side. Pilot signal 132
(Always its value is "1") is spread modulated with the PN code PN. The in-phase component Xi of the traffic signal is spread-modulated with the PN code PN and further modulated with the Walsh code W1,
It is added to the spread modulated pilot signal. The signal after the addition is defined as I. Quadrature component Xq of traffic signal
Is spread-modulated with the PN code PN and further modulated with the Walsh code W1. The control signal Xs is spread-modulated with the PN code PN and further modulated with the Walsh code W2. After these modulations, the orthogonal components of the traffic signal and the control signal are added. The signal after the addition is defined as Q.

The signal I and the signal Q are phase-modulated by the modulator 134.
After being subjected to (QPSK), it is sent to the receiving side 10 via the propagation path 14. The received signal 20 received usually has a phase shift due to the state of the propagation path, and also has a phase shift in the output signal of the carrier frequency oscillator on the receiving side 10. The received signal is multiplied by the signal output from the carrier frequency oscillator on the receiving side, and the in-phase component signal Ri and the quadrature component signal Rq
Is obtained.

The in-phase component signal Ri and the quadrature component signal Rq usually include a phase shift due to the propagation path, and also have a shift in the phase of the output signal of the carrier frequency oscillator on the receiving side 10. Considering the spread modulation on the transmitting side and the phase shift, the in-phase component signal Ri and the quadrature component signal Rq are as follows.

[0091]

[Equation 4] Ri = PN ・ (1 + Xi ・ W1) ・ β ・ cosθ-PN ・ (Xq ・ W1 + Xs ・ W2) ・ β ・ sinθ Rq = PN ・ (1 + Xi ・ W1) ・ β ・ sinθ + PN · (Xq · W1 + Xs · W2) · β · cosθ where PN is a PN code (± 1), W1 and W2 are Walsh codes (± 1), and Xi is an in-phase component of the traffic signal ( ±
1), Xq is the quadrature component (± 1) of the traffic signal, Xs
Indicates a control signal (± 1), β indicates an amplitude after being changed by a transmission line or the like, and θ indicates a phase shift.

From the signals Ri and Rq, the pilot signal is demodulated by despreading as shown in FIG. The signal Ri and the signal Rq are multiplied by a PN code to obtain a signal Ri 'and a signal Rq'. The signal Ri ′ and the signal Rq ′ are represented as follows.

[0093]

Equation 5 Ri ′ = (1 + XiW1) · β · cos θ− (Xq · W1 + Xs · W2) · β · sinθ Rq ′ = (1 + Xi · W1) · β · sinθ + (Xq · W1 + XsW2) · β · cosθ Next, as shown in FIG. 10, when the sum of these signals is obtained for N symbols, the sum of Walsh codes has a property of being zero. Disappears, and the signal CHi,
CHq is obtained.

[0094]

CHi = β · cos θ CHq = β · sin θ Using this signal, demodulation of the traffic signal and correction of the phase shift are performed as shown in FIG. That is, the signal
Ri and the signal Rq are multiplied by the product of the PN code and the Walsh function, and the signals Ri1 and Rq1 are obtained as follows.

[0095]

## EQU7 ## Ri1 = (W1 + Xi) ・ β ・ cos θ− (Xq + Xs ・ W1 / W2) ・ β ・ sinθ Rq1 = (W1 + Xi) ・ β ・ sinθ + (Xq + Xs ・ W1 / W2) • β · cos θ Next, as shown in FIG. 11, when the sum of these signals is calculated for one symbol, the signal Ri2 and the signal Rq2 become the following because of the property that the sum of W1 and W2 and the sum of W1 become zero. It is obtained as follows.

[0096]

## EQU00008 ## Ri2 = Xi.beta.cos.theta.-Xq.beta.sin.theta. Rq2 = Xi.beta.sin.theta. + Xq.beta.cos.theta. When the spread signals CHi and CHq are multiplied as shown in FIG. 11, an in-phase component Yi and a quadrature component Yq of the traffic signal that do not include a phase shift are finally obtained as follows.

[0097]

(Equation 9) Yi = Xi ・ β ²・ cos ² θ-Xq ・ β ²・ cos θ ・ sin θ + Xi ・ β
²・ sin ² θ + Xq ・ β ²・ cos θ ・ sin θ = Xi ・ β ² Yq = Xi ・ β ²・ cos θ ・ sin θ + Xq ・ β ²・ cos ² θ-Xi ・ β ²・ cos θ
· Sin θ + Xq · β ² · sin ² θ = Xq · β ² By the way, it is assumed that the demodulation device of the present embodiment is configured by an analog circuit, but the circuit forming the demodulation device according to the present invention shown in FIG. Can be configured entirely by digital circuits. In this case, the received signal 20 shown in FIG. 1 may be converted into a digital signal using an A / D converter, and the converted digital signal may be processed by a digital circuit.

Although the present invention has been described with respect to a demodulation device used for CDMA communication, the present invention is not limited to CDMA communication, and is applicable to any communication system using ordinary demodulation such as phase demodulation and a correlator. It can be used for any communication system.

[0099]

As described above, according to the present invention, it is possible to provide a terminal and a base station which do not require a high-precision carrier frequency oscillator.

[Brief description of the drawings]

FIG. 1 is a block diagram showing an embodiment of a demodulator for spread spectrum communication according to the present invention.

FIG. 2 is a block diagram showing an example of a transmitter of a CDMA communication base station corresponding to the demodulator for spread spectrum communication shown in FIG.

FIG. 3 is a diagram showing a time waveform of a PN code used in the embodiment shown in FIG. 1;

FIG. 4 is a block diagram showing one embodiment of a correlation detector used in the embodiment shown in FIG. 1;

FIG. 5 is a block diagram showing one embodiment of a compensator used in the demodulation device shown in FIG.

FIG. 6 is a block diagram illustrating the demodulation device illustrated in FIG. 1 using a transfer function.

FIG. 7 is an equivalent PLL of a circuit included in the demodulation device shown in FIG.
It is a block diagram showing a circuit.

FIG. 8 is a flowchart of a synchronization establishing operation performed by the demodulation device shown in FIG. 1;

FIG. 9 is an explanatory diagram of a method of correcting a phase shift using a pilot signal.

FIG. 10 is an explanatory diagram of a method for correcting a phase shift using a pilot signal.

FIG. 11 is an explanatory diagram of a method for correcting a phase shift using a pilot signal.

[Explanation of symbols]

 10 Demodulator 16 Voltage controlled oscillator 18 PLL 24 Pilot signal correlation detector 30 Compensator 32 Judgment circuit 42 Controller 86 Timer 88 Trigger circuit

────────────────────────────────────────────────── ─── Continuation of the front page (56) References JP-A-7-193557 (JP, A) JP-A 5-227123 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) H04B 1/69-1/713 H04J 13/00-13/06 H04L 27/00-27/38

Claims (6)

(57) [Claims] 1. A spread spectrum communication demodulator for receiving a spread-modulated received signal and demodulating the received signal, wherein the device receives the received signal and synchronizes with the received signal. Phase-locked loop means; and correlation detection means for receiving a signal output from the phase-locked loop means, performing correlation detection, and outputting a detection output, wherein the loop gain of the phase-locked loop means is A demodulator for spread spectrum communication which depends on a detection output of a detector. 2. The demodulator for spread spectrum communication according to claim 1, wherein said phase locked loop means is configured to control the phase locked loop means based on an output of the phase locked loop means when a predetermined time has elapsed. A demodulator for spread spectrum communication, comprising: a determination unit for determining whether or not the signal is synchronized with a received signal. 3. The demodulation device for spread spectrum communication according to claim 2, wherein the device is configured to perform the phase synchronization when the determination unit determines that the phase locked loop unit is not synchronized with the received signal. A demodulator for spread spectrum communication, comprising a changing unit for changing a frequency of a signal output from a synchronous loop unit. 4. The demodulator for spread spectrum communication according to claim 1, wherein said demodulator has a compensation circuit for preventing the output of said correlation detection means from being frequency dependent. Spread spectrum communication demodulator. 5. A spread spectrum communication demodulation method for receiving a spread modulated received signal and demodulating the received signal, the method comprising: synchronizing with the received signal by a phase locked loop; A demodulation method for spread spectrum communication, wherein a correlation detection is performed on a signal output by a synchronous loop, a detection output is output, and a loop gain of the phase locked loop depends on a detection output of the correlation detection. 6. The demodulation method for spread spectrum communication according to claim 5, wherein when a predetermined time has elapsed, the phase locked loop is synchronized with the reception signal based on an output of the phase locked loop. A demodulation method for spread spectrum communication.