JP3048258B2 - Pulse radar and its components - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

FIELD OF THE INVENTION The present invention relates to radar target tracking devices for use in guided missiles, and more particularly to frequencies at which optical techniques can be used to reduce the size of such devices, reduce costs, and improve performance. An active target tracking device that operates.

[0002]

2. Description of the Related Art A terminally guided small bomb (submun)
Anti-armored weapons systems using pits have been developed to automatically follow, identify and attack targets in high ground clutter backgrounds. In order to provide all weather, such small bombs generally use millimeter wave radar target trackers, and to achieve the required target discrimination, millimeter wave target trackers can be used with relatively complex radar systems, e.g., synthetic radar systems. An aperture radar system or a polarization radar system must be used. However, the structure of each of these types of radars is relatively complicated. The complexity of such radar systems is, for example, 94 GHz
At operating frequencies, conventional waveguide dimensions are on the order of about 0.050 to 0.100 inch, and in many rigorous assemblies about 0.025 mm.
(0.001 inch) or more is required. While it is possible to produce such millimeter-wave hardware at a somewhat lower cost using modern robotics, the costs associated with tuning and testing such tight tolerance hardware are too high. Becomes

[0003] Problems associated with incorporating and adjusting an active millimeter-wave target tracker into a conventional small bomb include a polarization or dual-polarization monopulse target tracker using a waveguide element and having no monopulse tracking capability. It will be appreciated that may require more than 20 different waveguide elements to control the routing and duplexing of the various signals exiting the transmitter and returning to the receiver. If monopulse tracking capability is required, all of the above waveguide elements would be required to track each other in both amplitude and phase. At an operating frequency of 94 GHz, one inch (about 25.
One thousandth of each 4 mm) is equivalent to about 2 ° of phase. Therefore, it must be understood that the required phase and amplitude tracking between the various channels is very difficult.

Another problem inherent in active millimeter wave radar target trackers using waveguide devices is providing sufficient isolation between the transmitter and the receiver. The problem is that waveguide switches and circulators that provide a high degree of isolation require 94 GH
The fact that it is not generally available at operating frequencies of z makes it even more severe. As a result, it is generally required to shut down the transmitter during the inter-pulse period of the radar to achieve the required separation. However, such attempts have involved the use of complex phase lock control loops as described in pending U.S. Patent Application No. 356,696, filed March 3, 1982, assigned to the same assignee as the present application. It is necessary to ensure that the phase of the transmitter is properly controlled during the pulse transmission period.

Another problem inherent in millimeter-wave radar systems using waveguide elements is the relatively low operating bandwidth, primarily due to tight waveguide tolerances. The relatively narrow operating bandwidth increases the sensitivity of millimeter wave radar to electronic interference.

[0006]

SUMMARY OF THE INVENTION Accordingly, it is a primary object of the present invention to provide a low cost and small volume active millimeter wave radar target tracking device with the foregoing background in mind.

Another object of the present invention is to provide a millimeter wave radar system having improved isolation between transmit and receive channels.

Yet another object of the present invention is to provide a broadband active millimeter wave target tracking device.

Yet another object of the present invention is to provide an active millimeter-wave target tracking device having monopulse tracking capability and being more compact than known active millimeter-wave type target tracking devices.

Another object of the present invention is to provide an active millimeter wave radar target tracking device capable of transmitting and receiving a circularly polarized or linearly polarized signal.

For the above and other objects of the present invention,
In general, an optical diplexer is used to provide the necessary separation between the transmitter and receiver, while simultaneously transmitting a circularly polarized signal and simultaneously receiving both left and right circularly reflected signals. This is achieved by providing an active millimeter wave radar target tracker that provides polarization twist capability. The diplexer is optically coupled to an antenna system that includes a 45 ° hyperbolic reflector, a parabolic reflector, and an elliptical plate scanning reflector. The left-handed and right-handed circularly polarized reflected signals are converted by a diplexer into corresponding planar (linear) polarization signals, which are applied to corresponding output ports of the diplexer. The output signal from the diplexer is optically coupled to a radio frequency mixer array in a radio frequency (RF) receiver and heterodyned with a local oscillator signal that is optically coupled from the transmitter output channel to the RF mixer array. Down-converted to a first intermediate frequency (IF) of 500 MHz. This IF output signal from the mixer array is combined in a conventional monopulse comparator, and the resulting sum and difference signals corresponding to each of the received polarizations are quadrature detected.
It is digitized and provided as an input signal to a conventional digital signal processor to derive the required guidance command signal.

Other objects and many advantages of the present invention will be readily and better understood by reference to the following detailed description when taken in conjunction with the accompanying drawings.

[0013]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Referring first to FIG. 1, an active millimeter wave radar target tracking apparatus 10 according to the present invention
3 is shown to include a transmitter 11 connected to a diplexer 15. The diplexer, which will be described in detail below, converts the signal from the transmitter 11 to the signal shown in FIGS.
It serves to convert to a circularly polarized output signal for the antenna system described in detail with respect to B. Here, the antenna system consists of a hyperbolic mirror 17, tilted at an angle of 45 °, a parabolic mirror 19, and an elliptical flat plate scanning mirror 2 fixed to a conventional gimbal system (not shown).
Let's just mention that it contains 1. In the transmission mode, the light incident on the fixed 45 ° hyperbolic mirror 17 is 9
The 4 GHz radiation is directed to the elliptical flat plate scanning mirror 21 via the parabolic mirror 19. Those skilled in the art will appreciate that the elliptical flat plate scanning mirror 21 operates to scan incident energy over twice the gimbal angle. In the receiving mode, the radar reflection signal incident on the elliptical flat plate scanning mirror 21 is transmitted through the spherical mirror 19 and the fixed 45 ° hyperbolic mirror 17 to the diplexer 1.
5, the reciprocal relationship is maintained. In the receive mode, diplexer 15 converts both left-handed and right-handed circularly polarized reflected signals (corresponding to reflections from either "single reflection" or "double reflection" objects) into planar signals. It decomposes into (linear) polarization signals and serves to optically couple such planar polarization signals to a receiver 25 described in detail below with respect to FIGS. Transmitter 11
An internally generated local oscillation signal is also provided to receiver 25 via horn antenna 27. Receiver 25
(A) downconverts the radar reflected signal to a first IF signal, eg, 500 MHz, (b) forms a monopulse sum and difference signal at this IF frequency, and (c) converts the first IF signal to 1 GHz. (D) quadrature detection of the monopulse sum and difference signal. An output signal from the receiver 25 is output to an analog / digital (A / D) converter 27.
And is provided to a fast Fourier transform (FFT) signal processor 29 as an input signal. This processor serves to perform the target detection function in a well-known manner. The output signal from the FFT processor 29 is sent to a digital computer 31, which now has an Intel Model 808 from Santa Clara, 95051, California, USA.
Six 16-bit microprocessors, which effectively implement target tracking and radar timing, among other things. The guidance control signals generated inside the digital computer 31 are sent to a conventional autopilot 33, where the necessary control signals for a small bomb control surface (not shown) are generated.

Referring now to FIGS. 1A and 1B, the scanning mirror antenna system of the present invention is fixed at 45 °, tilted at an angle of 45 ° with respect to the small bomb's longitudinal axis (unnumbered). It is shown to include a hyperbolic mirror 17. The hyperbolic mirror 17 fixed at 45 ° is supported by a support post 14 attached to the body (no number) of the small bomb, and the center of the hyperbolic mirror 17 fixed at 45 ° is a diplexer 15 (see FIG. The focus coincides with the focus of the output lens of 1). In the transmit mode, the 45 ° fixed hyperbolic mirror 17 directs incoming 94 GHz radiation to a parabolic mirror 19 which extends along half of the inner circumference of the small bomb (unnumbered). Work like. The parabolic mirror 19 has a shape that directs the substantially collimated beam toward the elliptical flat mirror 21. An elliptical flat mirror 21 is mounted on a conventional gimbal (not shown) so that the substantially collimated beam can scan over an entire scan angle of at least ± 30 ° azimuth and a minimum elevation of 25 °. The elliptical flat mirror 21 has a clearance hole (no number) provided at the center thereof for the hyperbolic mirror 17 fixed at 45 °. The elliptical shape of the flat scan mirror is necessary to prevent beam obscuration during vertical and horizontal scanning. The rear face (unnumbered) of the hyperbolic mirror 17 fixed at 45 ° is made of RF absorbing material (not shown) to minimize the problem of secondary reflection of the reflected signal 25 of the radar directly incident on it. Covered.

Next, referring to FIG. 2, the transmitter 11 will be described in detail. However, before the description, it should be noted that for purposes of signal processing, the example millimeter wave radar target tracker 10 (FIG. 1) uses a chirp waveform. Such a chirp waveform is between 15.66 and 15.75 GH
It is generated by applying a voltage ramp waveform to a Ku band voltage controlled oscillator (VCO) 43 as a control signal, tunable over a frequency range of z. The voltage ramp waveform is generated in a normal voltage ramp generator 41 in response to a control signal provided by a timing generator 49. This timing generator further includes a digital computer 31 (FIG. 1).
As controlled by the control signal given by The chirp waveform from the VCO 43 is amplified in a field effect transistor (FET) amplifier 45, and the output signal from this amplifier is provided to an absorption PIN diode modulator 47, which is a timing generator 49. To produce a pulse waveform in response to a control signal provided by The chirp pulse waveform from the absorption PIN diode modulator 47 is a varactor
The frequency is multiplied by the diode multiplier 51 to 9
This produces an output signal between 4.0 and 94.5 GHz. Such an output signal is sent to the IMPATT diode oscillator 55 via the circulator 53 as an injection lock signal. This oscillator, here a single diode device, is modulated by an IMPATT diode modulator 57 which is further triggered by a control signal from a timing generator 49. The amplified output signal from the IMPATT diode oscillator 55, which in this example is at a level of about 150 milliwatts, is sent as an input signal to the diplexer 15 (FIG. 1) via the circulator 53 and the horn antenna 13 (FIG. 1).

As mentioned briefly above, the LO signal for receiver 25 (FIG. 1) is also generated within transmitter 11. The LO signal is a 93.90 GHz stable local oscillator (STAL) which in this example is a Gunn diode oscillator.
O) 59. Since this signal is not a chirp waveform, a LO signal having a second chirp waveform must be provided to the receiver 25 (FIG. 1) to remove the chirp waveform of the radar reflected signal. Such a second LO signal is generated in the directional coupler 61 by coupling a part of the output signal from the STALO 59 to the harmonic mixer 63. A second input signal to such a mixer is obtained by combining a portion of the chirped output signal from FET amplifier 45 via combiner 65. Resulting IF from harmonic mixer 63
The output signal is a 1.0 GHz crystal controlled oscillator (X
The output signal from the (CO) 71 is amplified in an amplifier 67 before being up-converted into an L-band signal by heterodyning in a mixer 69. The output signal from mixer 69 is filtered in a high pass filter 73 that serves to pass only the higher sideband signal from mixer 69. The resulting 1.10 to 1.60 GHz output signal from filter 73 is the second LO
Before being provided as a signal to receiver 25 (FIG. 1), it is amplified in amplifier 75.

Referring now to FIG. 3, the input port 81 of the diplexer 15 includes a lens 83 which serves to collimate the 94 GHz signal received from the transmitter 11 (FIG. 2) via the horn antenna 13 (FIG. 1). Is shown in In this example, Irv of 92714, California, USA
ine, Kettering Street 1770
Model Number 01LQB of Melles Griot
The lens 83 which is 028 is a fused quartz lens having two convex surfaces. The lens 83 has a diameter of about 20.3 mm
(0.800 inch) and has a focal length of about 1.0 inch (25.4 mm). The lens 83 is covered with a non-reflective layer (not shown) of polyethylene having a thickness of a quarter wavelength of a dielectric medium (polyethylene having a relative dielectric constant of 2.0). The 94 GHz signal from the lens 83 is incident on a conventional polarization grating 85 inclined at an angle of 45 ° with respect to the vertical axis, and in this example, a wire parallel to one side is used.
Includes a half-wave thickness quartz substrate covered with a non-reflective layer on which a strip pattern (not shown) is deposited. As in the present example, when the electric field (E) of the incident signal is at right angles to a wire strip (not shown), the strip presents a capacitive effect, allowing the polarized signal to pass unhindered by the grating 85. Will do. The 94 GHz signal output from the polarization grating 85 passes through a second lens 87, which is the same as the lens 83, and strikes a Faraday rotator 89.

Referring now briefly to FIG. 4, Faraday rotator 89 includes an annular cobalt permanent magnet 91 having a ferrite disk 93 disposed centrally. The ferrite disk 93 has a diameter of about 13.97 mm in this example.
(0.550 inch), about 2.54mm (0.10mm) thick
0 inch) magnesium ferrite material (in this example, a model TT1-3000 material from Trans-Tech, Gaithersburg, Maryland, USA). On either side of the ferrite disk are positioned fused quartz matching disks 95, 97 having a thickness of about 0.016 inches. The annular samarium-cobalt permanent magnet 91 has a diameter of about 3
8.1 mm (1.50 inches), about 12.7 mm thick
(0.500 inch) produces an axial electric field component exceeding 3000 Gauss. Those skilled in the art will be familiar with the Faraday rotator 8.
It will be appreciated that 9 acts to rotate the direction of polarization for a linearly polarized input signal. Therefore, when the horizontal E field component of the signal incident on the Faraday rotator 89 is parallel to the vertical axis, the horizontal E field component for the data signal from the rotator 89 is ( Rotated 45 ° (counterclockwise).

Referring again to FIG. 3, the data signal from the Faraday rotator 89 travels through another lens 99 (same as lenses 83 and 87) to the second polarization grating 101. Parallel strips of such a grid (not shown)
At an angle of 45 ° to the vertical axis. As a result, the E-field of the incident signal is parallel to the metal strip, so that the signal is reflected to the same lens 103 as lenses 83,87,89. The signal from the lens 103 is
It is incident on a quarter-wave plate 105 of sapphire. This quarter-wave plate 105 is, in this example,
Approximately 17.78 mm (0.70 inch) in diameter and about 2.388 mm (0.094 inch) in thickness, and quarter-wave thickness on both sides covered with MYLAR (trademark, not shown) Having a reflective non-reflective layer. Quarter wavelength plate 105
Works to convert a linear (plane) polarized signal incident thereon into a circularly polarized output signal. For the optical path through diplexer 15 described above, the resulting signal at the output of quarter-wave plate 105 is circularly polarized clockwise.

In the receive mode, a counterclockwise circularly polarized signal incident on the quarter-wave plate 105 (corresponding to reflection from a so-called "single reflection" scatterer such as clutter) is described above. Contrary to that, the E field has a direction of 45 ° with respect to the vertical axis, but the polarization grating 101
The upper metal strip (not shown) is converted to linear (plane) polarized radiation having a perpendicular direction. As a result, such a linearly (plane) -polarized signal traverses the lens 103 and then passes through the polarization grating 101 and lens 1.
07 (same as the lenses 83, 87, 99, 103) and exits from the so-called left circular receiving port 109.

The clockwise circularly polarized reflected signal (corresponding to the reflected signal from a so-called "double reflection" scatterer such as an artifact) has a quarter-wave plate 105 with the E field on the vertical axis. The polarization grating 101 having a direction of 45 °
It will be converted to a linear (plane) polarization signal with a direction parallel to the upper metal strip (not shown).
After traversing the lens 103, such a signal is reflected by the polarization grating 101 again through the lens 99 to the Faraday rotator 89. In the receiving mode, the rotor
Due to its non-reciprocating characteristic, the incident signal is rotated by 45 ° in the same direction (direction) as the transmitted signal, and the signal output from the Faraday rotator 89 is parallel to the metal strip on the polarization grating 85 and on the vertical axis. On the other hand, it means rotating the 90 ° E field. Therefore, this signal passes through the lens 87, is reflected by the polarization grating 85, and is reflected by the lens 111 (the lenses 83, 87, 99, 103, and 107).
) To the so-called right circular receiving port 113.

The housing 121 of the diplexer 15
In this example, it is composed of two halves 122a and 122b. These halves are molded plastic to reduce both weight and cost. Each of the halves 122a, 122b has a polarization grating 85, 101, a lens 87, 99,
Further, a concave portion (no number) for holding the Faraday rotator 89 is provided. Two halves 122a, 122b
Are tightened together by a convenient method such as a screw (not shown) inserted into the clearance hole 123 so as to screw with a screw hole 124 provided in the lower half 122b.
A total of five screw holes (not numbered) are provided in the housing 121 for various input / output lenses. Lens 8
Each of 3, 107, 111 is held in place on housing 121 by a collar 125 with plastic screws. The two holes (unnumbered) provided in the upper half 122a are slightly larger in diameter than the corresponding holes (unnumbered) provided in the lower half 122b. This is necessary to accommodate the lens 103 and the plastic element 126 which is threaded for the sapphire quarter-wave plate 105. These two elements are housed in a housing 12 by means of a threaded plastic collar 127.
1, and a spacer 128 is provided to separate the lens 103 from the quarter wave plate 105. Finally, the opening (unnumbered) provided in the upper half 122a
Is closed by the cap 129.

Next, in FIG. 5, the diplexer 15
The manner in which the output signal from FIG. 3 is optically coupled with the receiver 25 is described in detail. However, here, in the receiver 25 of the present invention, the 94 GHz input signal from the diplexer 15 (FIG. 3) is coupled to the 93.9 GHz LO from the transmitter 11 which is also optically coupled to the receiver 25.
By heterodyne with a signal, for example, 100M
Recall that it is down-converted to the first IF signal at Hz. Also, the oppositely polarized input signal from the diplexer 15 (FIG. 3) is switched so that a sum and difference signal of monopulses can be formed at the first IF frequency for each input signal polarized in the opposite direction. It should also be noted that down-conversion is performed using an array of two balanced mixers. Finally, for the sake of illustration, one of the dual polarization receivers 25
Only one polarization channel will be described in detail.

The radiation of 94 GHz from the circular receiving port 109 (FIG. 3) on the left side of the diplexer 15 (FIG. 3) is about 20.32 mm (0.80 mm) provided on the receiver body 130.
0 inch) diameter and then 1/32 inch D about 0.794 mm (1/32 inch) supporting an IF circuit (not shown)
Ground plane 131 and dielectric 133 of a microstrip board (unnumbered) made of uroid ™
, And is shown to be incident on the receiver 25. At the entrance of the circular hole is arranged in this example a polarization grating 135 identical to the polarization grating 101 (FIG. 3), the metal strip of which is perpendicular to the E-field of the incident signal so that said signal passes. Be consistent. This signal also traverses a quartz substrate 137 that supports the mixer array (FIG. 6), which is described in detail herein below with respect to FIG. Crystal substrate 13
7, and thus the mixer array is located at a point corresponding to the focal point of lens 107 (FIG. 3). The 94 GHz signal coming out of the quartz substrate 137 is incident on the second polarization grating 141 and its metal strip (not shown) is aligned parallel to the E-field of these signals. As a result, the signal is reflected back to the crystal substrate 137 so as to be connected to the mixer diode (shown in detail in FIG. 6). The polarization grating 141 controls the amount of 94 GHz signal connected to the mixer diode to match the complex impedance of the mixer diode, so that the axis is aligned with the quartz substrates 137 and 139 in a housing (not shown). Can move in any direction.

9 from the horn antenna 27 (FIG. 1)
The 3.9 GHz LO signal is coupled to the polarization grating 135 via the polarization grating 141 and the quartz substrate 137, and is reflected back from here to the quartz substrate 137. The manner in which the 93.9 GHz LO signal is coupled to the mixer diode is described in detail below with respect to FIG. Here, since the polarization grating 135 also controls the coupling of the 93.9 GHz LO signal to the mixer diode to match the coupling impedance, the quartz substrate 1 in the housing 139 is also used.
Suffice it to say that it can move axially relative to 37.

The quartz substrate 137 is only about 0.178 mm
(0.007 inches) thick, and thus the substrate 1
37 is a dielectric foam material (Emerson-C, Canton, Mass., USA) to provide structural rigidity.
It should be noted that it is embedded in Ummings EccofoamFPH material.

Next, referring to FIG. 6, a mixer array 150 of the present invention has four square holes (also not numbered) formed by etching four metal holes 152 provided on the top surface of a quartz substrate 137. Shown to include a mixer (unnumbered). Each mixer (unnumbered) is, in this example, an M from Dodington Road, Lincoln, UK.
arconi Electronic Devices
A pair of beams, a model DC1346 device from
It is shown to include read diodes 151a, 151b. In these devices, the anode of the diode 151a and the cathode of the diode 151b are connected by a connection 153.
Are connected in series so that the cathode of the diode 151a and the anode of the diode 151b are connected to the metal plate 152, that is, the ground. These diodes 151a and 151b are square holes (no number).
94 are heat-compressed to both ends of the
The GHz signal produces a current that is "in phase" with the E-field of the 94 GHz radiation. That is, the diode 151a,
151b is in parallel with the incident E-field, thus
The H field of the incident radiation will induce a current in the latter. The 93.9 GHz LO signal is output from the quartz substrate 137.
Optically coupled to the mixer (unnumbered) via both sides of the However, the E field of the LO signal is 94.0 GHz.
, A quarter-wave monopole 155 arranged at right angles to each pair of diodes 151a, 151b connects the LO signal to each pair of diodes 151a, 151b. As provided.

The polarization grating 141 (FIG. 5) not only serves to prevent the 94 GHz signal radiation from being coupled to the LO channel (unnumbered), but also to move it at the frequency of the 94 GHz signal. Also works to match the complex impedance of the mixer (unnumbered). Similarly, the polarization grating 135 (FIG. 5) has an LO of 93.9 GHz.
Blocking signals from entering diplexer 15 (FIG. 3),
It also serves to match the complex impedance of the mixer (unnumbered) with the monopole 155 LO signal frequency of 93.9 GHz. Since the polarization gratings 135, 141 (FIG. 5) can be adjusted independently, matching of the mixer (unnumbered) at both the signal and local oscillator frequencies can be achieved over a relatively large strip bandwidth.

The low-pass matching filter 157 formed by the linear medium in the same plane is used to output the I-wave from the mixer (unnumbered).
Provided to connect the F output signal to receiver 25 (FIG. 1). The design of such a low-pass matching filter is
It is well known to those skilled in the art and therefore will not be repeated here.

Next, the operation of the receiver 25 will be described in detail with reference to FIG. However, it should be noted that before that description, only the half of the receiver 25 corresponding to the so-called "left circular polarization channel" is shown. It will be appreciated that the right circular polarization channel (not shown) is the same as described herein.

The first IF output signal from the mixer array 150 is synthesized by a well-known monopulse comparator 161 which generates a sum, pitch and yaw signal of the monopulses. Such a monopulse signal is also chirped and therefore de-chirped (correlated) and controlled in mixer 163 by a control signal provided by digital computer 31 (FIG. 1). Delay line 16 with well-known taps shown as
It will be appreciated that the transmitted signal obtained by passing the second LO signal from the transmitter 11 (FIG. 2) through 5 is upconverted to the L-band frequency by heterodyning with a delayed replica of the transmitted signal. Mixer 16
3 is filtered in strip bandpass filter 165 to remove unwanted spurious signals generated in mixer 163. The filtered output signal from strip bandpass filter 165 is in-phase (I) from L-band crystal controlled oscillator 171 and L-band crystal controlled oscillator 171 at mixers 169I, 169Q before being quadrature detected and downconverted to a baseband video signal. 4
It is amplified in the amplifier 167 by being heterodyned by the 1 / phase (Q) output signal. Mixer 1
A quarter phase reference signal for 69Q is generated by passing a portion of the output signal from crystal controlled oscillator 171 through 90 ° phase shifter 173. Baseband I and Q output signals from receiver 25 are provided to A / D converter 27 as input signals.

While the preferred embodiment of the present invention has been described, it will be apparent to those skilled in the art that other embodiments incorporating the inventive concept of the present invention may be used. Accordingly, the invention is not limited to the embodiments disclosed herein, but only by the spirit and scope of the appended claims.

[Brief description of the drawings]

FIG. 1 is a schematic block diagram showing an active millimeter wave radar target tracking device according to the present invention. 1A and 1B are a cross-sectional view and a plan view, respectively, of the antenna system of FIG.

FIG. 2 is a schematic block diagram showing the transmitter of FIG.

FIG. 3 is a sectional view showing the diplexer of FIG. 1 according to the present invention;

FIG. 4 is a sectional view of a Faraday rotator used in the diplexer of FIG.

FIG. 5 is a schematic diagram showing how the diplexer of FIG. 3 is optically coupled with the receiver of FIG. 1;

FIG. 6 is a plan view showing one of the mixer arrays of the receiver of FIG. 5;

FIG. 7 is a schematic block diagram showing the receiver of FIG. 1;

[Explanation of symbols]

 REFERENCE SIGNS LIST 10 Active millimeter wave radar target tracking device 11 Transmitter 13 Horn antenna 15 Diplexer 17 45 ° fixed (hyperbolic) mirror 19 Parabolic mirror 21 Elliptic flat plate scanning mirror 25 Receiver 27 Analog / digital (A / D) converter 29 FET processor 31 Digital computer 33 Autopilot

────────────────────────────────────────────────── ─── Continuation of the front page (56) References JP-A-5-107336 (JP, A) JP-A-5-164838 (JP, A) JP-A-5-315833 (JP, A) 231182 (JP, A) JP-A-60-257376 (JP, A) (58) Fields investigated (Int. Cl. ⁷ , DB name) G01S ⁷ /00-7/42 G01S 13/00-13/95

Claims (8)

(57) [Claims] 1. A first detector in a monopulse receiver operating in the band above the 94 GHz band, comprising: (a) first, second, third and third groups gathered around a center point on a common substrate. 4. Balanced detectors of claim 4, wherein each of said detectors comprises a pair of similarly polarized beam lead diodes; and (b) detecting via the beam lead of each said diode. (C) means for optically coupling the first local oscillator signal to each pair of beam lead diodes; and (d) means for optically coupling the first local oscillation signal to each pair of beam lead diodes. An output line electrically coupled to the connection point of each pair. 2. The common substrate comprising: (a) a dielectric sheet substantially transparent to detected radio frequency energy; and (b) a metallic coating on one surface of the dielectric sheet. The coating is formed through and collected around a center point on the common substrate;
There are second, third and fourth openings, each of which is
Claims: A substantially square portion and a substantially rectangular portion coupled to a first side of the substantially square portion at a center thereof and extending to a periphery of the dielectric material sheet. Item 1. The first detector according to Item 1. (A) each pair of said beam lead diodes is connected in series between a center of a corresponding second and third side of said square portion; and (b) a center of an electrical transmission line. 3. The first detector of claim 2, wherein a conductor is formed on the periphery of the dielectric sheet at the center of the substantially rectangular portion from a connection point of each pair of the beam lead diodes. 4. The first detector according to claim 3, wherein the center conductor is shaped inside a rectangular portion of the opening to form a low-pass filter. 5. The first detector according to claim 4, wherein the first local oscillation signal is coupled via a portion of a center conductor inside a square portion of the opening. 6. A detected radio frequency signal is transmitted in a first beam from a first side of the common substrate, and the first local oscillation signal is transmitted in a second beam from a second side of the common substrate. Transmitted, the directions of polarization of the two signals are at right angles, and the longitudinal axes of the two beams are at right angles to the common substrate,
The first detector according to claim 5. 7. A third polarization screen and a fourth polarization screen disposed parallel to and opposite to the common substrate,
7. The first detector according to claim 6, wherein said third polarization screen reflects a detected radio frequency signal and said fourth polarization screen reflects said first local oscillation signal. 8. The third and the fourth from the common substrate.
The first detector according to claim 7, wherein the distance of each of the polarization screens is adjustable.