JP2986225B2 - DC-DC converter and switching power supply - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a DC-DC converter and a switching power supply, and more particularly to a DC-DC converter having a higher frequency.

[0002]

2. Description of the Related Art Conventionally, as this type of switching power supply, for example, there is one shown in FIG.

FIG. 4 shows a basic circuit of a full bridge converter system which is a typical circuit of a conventional switching power supply. In this method, power FETs (Q1 to Q4), which are four bridge-connected switching elements, are connected to each other by two pairs (Q1 and Q4, Q2 and Q3).
This is a method in which a rectangular wave voltage is applied to the primary side of the transformer T1 by N-OFF, and the output voltage is set by changing the conduction time.

FIG. 2 shows the ON-OF of each of the FETs Q1 to Q4.
The F signal, the primary voltage V1 of the transformer T1 and its current I1
3 is a timing chart of FIG.

[0005]

However, in such a conventional switching power supply, as shown in FIG. 3, when the switching element is switched from OFF to ON, when the switching element is switched from OFF to ON, and during the ON time. Loss occurs.

Hereinafter, the loss amount will be described.

(A) The loss during the time τ1 for switching from OFF to ON is given by the following equation.

[0008]

(Equation 1)

(B) The loss during the ON time τ2 is given by the following equation.

[0010]

(Equation 2)

(C) The loss during .tau.3 at which ON-OFF switching is performed is given by the following equation.

[0012]

(Equation 3)

In the above equation, τ1, τ3 and RON depend on the characteristics of the switching element.

VDS and ID depend on the input voltage EIN and the output power.

Further, P1 and P3 are switching frequencies f
Is proportional to

P2 is almost constant regardless of the frequency because τ2 becomes shorter as f becomes higher.

The fact that P2 is substantially constant will be described below.

FIG. 15 shows VD, ID, and loss when the frequency is f = 1 / T. FIG. 16 shows VD, ID, and loss when the frequency is doubled and f = 2 / T.

Since RON is a fixed value determined by the characteristics of the FET element, P2 is as follows.

[0020]

(Equation 4)

If the frequency is doubled, τ2 becomes about τ2 / 2 from FIGS. 15 and 16, and P2 becomes almost constant irrespective of the frequency.

Therefore, when the switching frequency is increased to reduce the size, the loss of P1 and P3 lowers the efficiency, so that there is a limit to the higher frequency and there is a disadvantage that the size cannot be reduced.

As another method of increasing the frequency, there are a current resonance method and a voltage resonance method.
There are the following.

Control is difficult because the control frequency must be variable. Further, the resonance frequency depends on the temperature characteristics of the resonance capacitor. In addition, the stress of the element due to the resonance current and the resonance voltage is large.

An object of the present invention is to provide a DC-DC converter capable of increasing the frequency without using a current resonance or voltage resonance system.

[0026]

SUMMARY OF THE INVENTION The present invention has been made in view of such conventional problems.

It has two or more switching means, a transformer, and switching control means. The switching means is alternately turned on (conducting) -off (non-conducting) to change the direction of the current flowing through the transformer. Alternatively, in a DC-DC converter for performing DC-DC conversion, a coil connected in parallel to the transformer and a diode connected in series to the coil, wherein the coil and the diode are: When connected in series and there is no output from the transformer,
The current in the coil is maintained.

[0028]

When the switching means is turned on by the switching control means, the coil is energized through the switching means to store energy.

In the above coil, the switching means is OFF.
From this time until the next ON, the battery is discharged (discharges energy) via the diode to maintain the voltage between both ends of the switching means in a zero voltage state. At this time, the switching control means Zero voltage switching is performed by turning on the switching means.

[0030]

DESCRIPTION OF THE PREFERRED EMBODIMENTS The present invention will be described below with reference to the drawings.

FIG. 1 is a block diagram showing one embodiment of the switching power supply according to the present invention.

In the present embodiment, in order to reduce the size of the DC-DC converter, the efficiency and the frequency are increased, and the control method is based on a conventional PWM control method.

The principle of this method is that a switching power supply in which two or more switching elements are alternately turned on and off alternately is a PW which is a conventional control method.
According to the M method, the voltage between both ends of the switching element is set to zero voltage by the energy accumulated in the choke coil at a certain timing before the time when the switching element is turned on, and the element is turned on during the zero voltage. . According to this method, the loss during switching is only the loss due to the conduction resistance of this element.

This switching power supply includes a rectifier RECT for rectifying an AC input, a smoothing circuit SMC for smoothing an output of the rectifier RECT, and a DC-DC converter DDC.
, And supplies the DC-DC converted DC voltage to the load RD.

The DC-DC converter DDC includes a feedback circuit FB for monitoring the output and a PWM control circuit I which is a switching control means for determining a pulse width based on a DC signal output from the feedback circuit FB.
C1, IC2, power switching elements (FETs) Q1 to Q4 as switching means, a choke coil L1 as a coil, and a diode inserted in series with the choke coil L1 (this diode uses a diode inside the FET. CR1 to CR4, secondary side rectifier diodes CR5 to CR6, capacitors (including output capacitors of switch elements) C1 to C4, secondary side choke coil L0, and secondary side smoothing. Capacitor C6
And transformers (1, 2 are primary side coils, 3, 4 and 4, 5
Has a secondary side coil) T1.

Ad is a power switch element (FET)
Are the input DC voltage.

Next, the operation of the above embodiment will be described.

FIG. 5 shows a timing chart of this embodiment.

The output (DC signal) of the feedback circuit FB is a signal proportional to the secondary output voltage. If the output voltage is higher than the rated value, the DC signal will also be higher and PWM
The pulse width of the outputs a and b of the control circuit IC1 becomes narrow, and the operation is performed so that the secondary voltage becomes a constant value. Note that the pulse widths of the outputs c and d of the PWM control IC 2 have a constant value τ2 regardless of the secondary side output voltage. To

(1) Operation from time t1 to t2 (power is being supplied to the secondary side)
And FET Q4 are ON, FET Q2 and FET Q5 are OFF
State. Therefore, the equivalent circuit between t1 and t2 is shown in FIG.

The currents iT1 and iLO flowing through the transformer T1 during the period from t1 to t2 are the same as the currents in the conventional switching method. The current flowing through the choke coil L1 is iLI = (EIN.t) / L1.

(2) Operation from time t2 to t3 (state in which electric power is not supplied to the secondary side)
Q4 is ON, and the other FETs Q1 to Q3 are OFF.
FIG. 9 shows an equivalent circuit.

At time t2, the FET Q1 turns OFF and EX
Although the voltage decreases, the capacitances of the capacitors C1 and C2 are set so that there is no problem with the loss in the FET Q1 until the time (tf) when the FET Q1 is completely turned off.

The loss in the FET Q1 in the circuit according to the present invention at t2 to tf and the loss in the FET Q1 in the conventional circuit shown in FIG. 4 are compared below.

FIG. 11 shows an equivalent circuit from t1 to t2 of the circuit according to the present invention, including the capacitors C1 and C2. FIG. 12 shows an equivalent circuit at time t2. From t2 to t2 + tf (Fall Time), iL1 + iT1 is considered to be a constant current value.

Then, during the period from t2 to t2 + tf, iQ1 is given by iQ1 = (iL1 + iT1) ・ (1- (t-t2) / tf).
Is as follows: ic = (iL1 + iT1). (T-t2) / tf.

Therefore, EX = EIN−ic · (t−t2)
/ (C1 + C2).

The drain-source voltage of the FET Q1 is ΔV
Then, ΔV = EIN−EX = ic · (t−t2) / (C1 + C2)
) = (IL1 + iT1) ・ (t-t2) 2 / (C1 + C2)
Tf 損失 Loss energy Pof while FET Q1 is completely turned off
f becomes like the following formula.

[0049]

(Equation 5)

From this equation, a specific numerical value will be obtained. As an example, C1 + C2 = 2000PF = 2 × 1
If 0-9F, tf = 50 ns=5.times.10@-8 sec, and iL1 + iT1 = 2 A, then Poff = {22 / (2.times.10@-9)}. Multidot. (5.times.10@-8) @ 2
・ (1/12) = 0.417 × 10−6 When the frequency is 1 MHz and the loss when the FET Q1 is turned off is P1, P1 = f × Poff = 10 6 × 0.417 × 10−6 = 0.
417W.

Next, the loss is calculated for the conventional circuit shown in FIG. FIG. 13 shows the current of the FET Q1 at this time.

IQ1 is defined as iQ1 = iT1 · tf during tf.
(1-t / tf).

FIG. 14 shows the drain-source voltage of the FET Q1 at this time.

EQ1 is equal to EQ1 = EIN (1
-T / 2tf) From this, Poff is as follows.

[0055]

(Equation 6)

IT1 = 2A, 100V at EIN, tf = 5
Assuming × 10−8 sec, Poff = (5/12) · 2 · 100 × 5 × 10−8 = (5
/12).10@-5 =4.26.times.10@-6 When the frequency is 1 MHz, the loss P1 when the FET Q1 is turned off is P1=f.times.Poff=106.times.4.26.times.10@-6 = 4.2.
6W.

This concludes the description of the effects of the capacitors C1 and C2. The same applies to the effects of the capacitors C3 and C4.

When Ex becomes 0 V, the current flowing through the choke coil L1 flows through the diode CR2 as shown in FIG.

Here, since the forward voltage of the diode CR2 is low (generally about 0.6 V), the change in the current of iL1 is very small. It will continue to flow.

The fact that this current is constant will be described in more detail.

FIG. 17 shows an equivalent circuit on the primary side in FIG.

Each current equation when the SW is ON (t1 to t2) is as follows. Here, E0 is the voltage across the coil L, and IOFF is the current flowing through L immediately before the SW is turned on.

[0063]

(Equation 7)

[0064]

(Equation 8)

Now, assuming that I0 = 0, the following equation is obtained at t2 (immediately before OFF).

IR = E0 / R: constant value iL = E0 × t1 / L The energy PL stored in the coil L at this time is PL = L × (E0 × t1 / L) ^{2
/ 2 = E0 2 × t1 2} / become (2L).

Each current equation when the SW is OFF (t2 to t3) is the release of energy stored in the coil.
The equivalent circuit shown in FIG. In FIG. 18, R = 0
Assuming that

[0068]

(Equation 9)

Therefore, if R = 0, E = 0.

That is, iL = ION = E0 × t1 / L.

This means that a constant current continues to flow because there is nothing to consume the energy of the coil (superconducting state).
In reality, since the coil itself has resistance, energy is consumed there. That is, when R ≠ 0, the following equation is obtained.

E = R × iL

[0073]

(Equation 10)

DiL / dt = R × iL / L Therefore, iL is given by the following equation.

[0075]

[Equation 11]

FIG. 19 shows a summary of the above results.

The description of the fact that the current flowing through the choke coil L1 is constant has been completed. Since Ex ≒ 0 V, the current ir1 = 0 of the transformer T1.

On the secondary side, the smoothing choke coil L0 keeps flowing current to the load side through the diodes CR5 and CR6 by the stored energy.

Of course, the third side of the secondary side of the transformer T1
Since the number of turns of 4, 4-5 is the same and the direction of the current is as shown in FIG. 9, no voltage is generated between both ends of the winding.

(3) Operation from time t3 to t4 (L1
During this time, all the FETs are OFF.

FIG. 10 shows an equivalent circuit.

At time t3, FET Q4 turns off, but F
As in the case of the ETQ1, the OFF loss is reduced by the capacitors C3 and C4.

Ex is in the state of 0V, and the voltage of EY increases.

The current flowing through the choke coil L1 becomes
Diode CR2 → choke coil L1 → diode C
The current iL1a flows through R3 and the current iL1b flows through transformer T1.

Current iL1 flowing through choke coil L1 = i
The inductance of the choke coil L1 is determined so that L1a + iL1b is ≥0 (A) even at time t4. This is because, in principle, iL1 can output 0A at time t4, but in order to surely switch the FETs Q2 and Q3 to zero voltage, the voltage between both ends of the diodes CR2 and CR3 is low and the variation of the elements is reduced. Considering this, iL1 ≧ 0.

At time t4, Ex ≒ 0 V, EY ≒ E1N, and when FET Q2 and FET Q3 are turned on at time t4, the voltage between both ends of FET Q2 and FET Q3 becomes voltage ≒ 0.
V and zero-voltage switching becomes possible.

(4) Operation at time T4 ≒ t7 (FE
(TQ2 and FET Q3 are ON and FET Q1 and FET Q4 are OFF.) IT1 is opposite to the current direction of item (1), and iL1 is after the energy accumulated in the choke coil L1 is completely released (ie1 = 0A). Thereafter, the current direction is opposite to the current direction of the item (1), and the operation is performed on the same principle as the items (1), (2), and (3).

Therefore, from time t6 to time t7, Ex ≒ E1N,
EY ≒ 0 V, and the FET Q1 and the FET Q4 can also realize zero voltage switching.

The above (1) to (4) are summarized in FIG.
This will be described with reference to FIG.

FIG. 24 is a circuit diagram showing an equivalent circuit (from the secondary circuit and the transformer T in FIGS. 8, 9, and 10) at times t1 to t4.
1 is omitted), the voltage applied to both ends of the choke coil L1, and the current (i
L1) are collectively shown.

FIG. 25 is for explaining the operation of the coil shown in FIG. 24 and corresponds to a principle diagram in the present invention, and is applied (or may be generated) to both ends of the coil. The relationship between the voltage V and the current i flowing through the coil is shown.

As shown in FIG. 25, the voltage Vp is applied between t1 and t2, energy is accumulated in the coil, and between t2 and t3, the magnitude of the current does not change since both ends of the coil are 0 voltage. . From t3 to t4, the coil releases a reverse voltage of Vm while releasing energy.
Causes EX to be 0V even if EY increases.
Is held in.

During the period from time t1 to time t2, EX is applied to both ends of the choke coil L1, so that iL1 increases with time. (Vp in FIG. 24 corresponds to EX.)
From time t2 to time t3, 0 voltage is applied to both ends of the choke coil L1, and the forward voltage of the diode CR2 is low, so that a constant current is maintained.

From time t3 to time t4, from t3 to t3 ', iL1 flows through capacitors C3 and C4, and choke coil L1 releases energy to generate back electromotive force, thereby charging capacitor C4. At the same time, the capacitor C3 is discharged to raise EY.

At time t3 ', the choke coil L
1 means that EY ≒ EI
After time N (t3 'to t4), back electromotive force is similarly generated, iL1 flows through diodes CR2 and CR3, and Ey is held at EIN by diode CR3, and EX ≒ EIN by diode CR2. Hold.

Thus, at time t4, the FET Q3
Q4 allows for zero voltage switching.

[0097] Incidentally, in T4～t ₇ is the direction of the current of the choke coil L1 is reversed, the principle of operation, t1
Same as ~ t4.

Next, the switching control means will be described.

Τ1 which is the output of the switching control means
(T1 to t2, t4 to t5) are controlled by a conventional PWM, and τ2 (t1 to t3, t4 to t6) is a fixed time.

Τ2 is determined so that t3 to t4 and t6 to t7 become the minimum required switch switching timing times.

The time from t1 to t9 is determined by the control unit using two element resistors and a capacitor.

More specifically, the control is performed using a control IC. As an example, μpc494 (N
EC) using the PWM control circuits IC1 and IC
2 is shown in FIG.

Μpc494PC1 and PC2 are directly
Since it is not an IC that can drive the ET, the FETs Q1 to Q4 are connected via the FET drive circuits FED1 and FED2.
Output pulse. In order to synchronize the signals a and b with the signals c and d, the μpc
Connect pin 5 of 494PC2 and send a synchronization signal.
The period t1 to t9 is determined by the capacitor CT connected to the 5th pin of the μpc494PC1 and the resistor RT of the 6th pin. These ICs output the signals a, b, c, and d in FIG. The pulse width τ1 of these signals changes depending on the output signal of the feedback circuit FB in FIG.

In FIGS. 1C and 1D, a fixed time τ2 from the rise of a and b can be formed by a monostable circuit.
Alternatively, by synchronizing the control ICs here, one can generate the signals a and b and the other can generate the signals c and d.

The choke coil L1 stores energy from t1 to t2, holds energy from t2 to t3, and stores t3 to t3.
At t4, the secondary side and a part release energy to the power supply side.

From t4 to t7, energy is stored and released in the opposite direction to t1 and t4.

In the circuits of FIGS. 8 to 10, a series saturable inductance is inserted with the primary side or the secondary side of the transformer T1, and the current inb of the transformer in time t3 to t4 in the item (3) of the operation description. Is brought to a state close to 0, iL1
Current can be reduced.

The principle is the same even if the signal timing is changed between a and d or b and c receiving the ON-OFF signal from the PWM.

That is, the basic ON-OFF operation is as follows.
When Q1 and Q4 are ON, Q2 and Q3 are OFF and conversely Q1
When signals Q4 and Q4 are OFF, signals a and d may be exchanged because Q2 and Q3 are operating with ON. Further, even if b and c are exchanged, L1 is maintained while all the switching elements Q1 to Q2 are OFF.
However, the basic operation is the same except that the current loop flowing through is different (the high voltage side of the primary voltage is on the low voltage (GND) side).

The capacitors C1 and C2 are replaced by one capacitor having the same capacity as C1 + C2, and the capacitors C3 and C4 are replaced by C3 +
The operation is the same even with one capacitor having the same capacity as C4. This is shown in FIGS. Further, if an FET is used as the switching element, the diodes CR1 to CR4 can be used because the diodes CR1 to CR4 exist inside the FET. C1 to C4 are the output capacitances C of the FETs.
An OFF capacitance value may be used.

According to this method, since zero voltage switching is performed, the efficiency can be made the same at a higher frequency than in the current resonance method. Therefore, the size can be reduced by increasing the efficiency.

The control is performed at a constant frequency PW.
The control is simple because of the M method. In the current and voltage resonance, the control frequency is variable, and the control circuit is complicated.

Further, since zero voltage switching is performed, voltage stress and current stress are small, and reliability can be improved.

The switching elements are FETs
However, the invention is not limited thereto, and a bipolar transistor (BT) or an insulated gate bipolar transistor (IGBT) may be used.

FIG. 6 shows another embodiment.

FIG. 6 shows an embodiment of the half bridge circuit.

Q1 to Q4 are switch elements (here, FEs).
T), C1 and C2 are capacitors, CR1 to CR4 are diodes, T1 is a transformer, L1 and L2 are coils, C3 and C2
C4 is a capacitor on the input side.

In FIG. 6, the secondary side, the control section, and the feedback circuit are omitted. FIG. 7 is a timing chart of FIG.

T1 to t2 are the same as before, and t2 to t3 are F
Since ETQ2 is OFF, Ex = 1 / 2Ex.

Since the FET Q3 is ON, iL1 has EIN
As a result, the current continues to flow while increasing.

At time t3, FET Q3 is turned off and iL1 flows through capacitors C1 and C2 and starts to flow in T1 in the opposite direction to iT1. On the other hand, L2 also has a diode CR3 → CR
4 flows to the coil L1. Ex is 1 / 2E1N → 0
And when ExＣＲ0V, CR2 conducts and keeps Ex = 0V. During the time when Ex is 0V, the FET at time t4
By turning on Q2 and FET Q4, FET Q2 and FE
TQ4 can perform zero voltage switching. FET at t5
Ex ≒ E1N when Q2 is OFF iL2 is FET Q
Since 4 is also ON, the current (voltage 1 / 2E1N) continues to flow while increasing from the capacitors C3 and C4.

At time t6, FET Q4 turns off, and iC2 flows through capacitors C1 and C2 and begins to supply power to T1 in the direction of iT1. On the other hand, diode CR3 → CR also flows to L1.
Flow through 4 Ex changes from 1 / 2E1N to E1N. When Ex = EIN, the diode CR1 conducts, and Ex = E1N.
Keep at 1N. Then, the FETs Q1 and Q3 are turned on. Therefore, the FETs Q1 and Q3 also perform zero voltage switching.

As described above, according to the present invention, the following effects can be obtained.

ZERO voltage switching is possible.
That is, the switching loss of the switching element is reduced by switching when the voltage across the switching element is 0 V. As a result, the efficiency can be improved and the frequency can be increased. This means that the power supply device can be downsized.

Further, a pulse width modulation method can be used. That is, the control is simple because it is a conventional pulse width modulation (PWM) method. Since the control is based on the conventional PWM method, this part may be a conventional technique. In addition, most commercially available ICs use the PWM method as compared with the frequency variable method, and the degree of freedom in selecting the control IC is high. And the protection circuit is also an extension of the conventional technology.

Further, since it is a non-resonant type, there is no need to consider the deviation of the resonance frequency unlike the resonance type, and the current and voltage stresses are smaller than in the resonance type. further,
Since it is not a resonance system, L and C for resonance are unnecessary. Therefore, it is not necessary to consider changes in the temperature and the like of the resonance component.

[0127]

As described above, according to the present invention, it is possible to provide a DC-DC converter capable of increasing the frequency without using the current resonance or voltage resonance method.

[Brief description of the drawings]

FIG. 1 is a circuit diagram of a full bridge system according to the present invention.

FIG. 2 is an operation timing chart of the circuit of FIG. 4;

FIG. 3 is an explanatory diagram of a loss of a switch element in the circuit of FIG. 4;

FIG. 4 is a circuit diagram of a full-bridge type switching power supply according to the related art.

FIG. 5 is an operation timing chart of the circuit of FIG. 1;

FIG. 6 is a circuit diagram of a half bridge system according to the present invention.

FIG. 7 is an operation timing chart of FIG. 6;

FIG. 8 is an explanatory diagram of the operation of the circuit of FIG. 1;

FIG. 9 is an explanatory diagram of the operation of the circuit of FIG. 1;

FIG. 10 is an explanatory diagram of the operation of the circuit in FIG. 1;

FIG. 11 is an equivalent circuit diagram between times t1 and t2.

FIG. 12 is an equivalent circuit diagram at time t2.

FIG. 13 is a graph showing the voltage and current of the FET Q1 of the circuit according to the present invention.

FIG. 14 is a graph showing the voltage and the current of the FET Q1 of the circuit according to the prior art.

FIG. 15 is a graph showing a loss of a circuit according to the related art.

FIG. 16 is a graph showing a loss of a circuit according to the related art.

FIG. 17 is an equivalent circuit diagram of FIG. 1;

FIG. 18 is an equivalent circuit diagram of FIG.

FIG. 19 is a diagram illustrating the operation of a coil.

FIG. 20 is a block diagram of μpc494.

FIG. 21 is a circuit diagram showing another embodiment of the DC-DC converter.

FIG. 22 is a circuit diagram showing another embodiment of the DC-DC converter.

FIG. 23 is a block diagram showing an embodiment of a PWM control circuit.

FIG. 24 is a diagram illustrating the operation of a coil.

FIG. 25 is a diagram illustrating the operation of a coil.

[Explanation of symbols]

L0: smoothing choke coil, L1, L11, L21: choke coil, Q1, Q2, Q3, Q4, Q11, Q21,
Q31, Q41 ... FET, T1, T11 ... transformer.

──────────────────────────────────────────────────続 き Continued on the front page (72) Inventor Toshihiko Uemura 2-16-46 Minami Kamata, Ota-ku, Tokyo Inside Tokimec Co., Ltd. (72) Inventor Yasuo Ishikawa 2-16-46 Minami Kamata, Ota-ku, Tokyo (56) References JP-A-2-155467 (JP, A) (58) Fields investigated (Int. Cl. ⁶ , DB name) H02M 3/28 H02M 3/335

Claims (5)

(57) [Claims] The present invention has two or more switching means, a transformer, and a switching control means. The switching means is alternately turned ON (conducting) -OFF (non-conducting) to reduce the current flowing through the transformer. A DC-DC converter for performing DC-DC conversion by changing the direction, comprising a coil connected in parallel to the transformer and a diode connected in series to the coil, wherein the coil and the diode Are connected in series and when there is no output from the transformer,
DC- characterized by maintaining the current in the coil
DC converter. And a switching control means for switching the switching means between ON (conducting) and OFF (non-conducting) alternately so as to reduce a current flowing through the transformer. In a DC-DC converter that performs DC-DC conversion by changing the direction, a coil connected in parallel to the transformer is connected in parallel with the switching means and in a direction opposite to the operation direction of the switching means. A diode, wherein the coil and the diode are connected in series and maintain current in the coil when there is no output from the transformer.
C-DC converter. 3. A switch comprising four switching means connected in series and parallel, a transformer, and a switching control means, wherein the switching means are alternately turned on (conducting) and off (non-conducting).
Then, by changing the direction of the current flowing through the transformer, D
In a DC-DC converter that performs C-DC conversion, the DC-DC converter includes a diode connected in parallel to the switching means, and a coil connected between intermediate points of the switching means connected in series, The coil and the diode maintain a current in the coil when there is no output from a transformer.
converter. 4. The DC-DC converter according to claim 1, wherein a capacitor is provided in parallel with said switching means, and when the switching means is turned off, a change in a voltage across the switching means is reduced. DC converter. 5. The DC- according to claim 1, 2, 3, or 4.
A switching power supply having a DC converter.