JP2916391B2 - Array antenna control method and control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a method and an apparatus for controlling an array antenna.

[0002]

2. Description of the Related Art Conventionally, a phased array antenna for satellite communication (hereinafter referred to as a first conventional example) mounted on a vehicle or the like and automatically tracking the direction of a geostationary satellite has been prototyped by the Communications Research Laboratory of the Ministry of Posts and Telecommunications. ing. The phased array antenna of the first conventional example is composed of 19 microstrip antenna elements, and one element is provided for each element except one element.
Eight microwave phase shifters are provided to electrically scan the direction of the transmission beam without mechanical drive. Here, as a sensor for controlling the directivity of the antenna and tracking the direction of the incoming beam, a magnetic sensor for detecting the direction of geomagnetism and calculating the direction of the geostationary satellite seen from the vehicle that is known in advance,
Further, an optical fiber gyro is provided for detecting the rotational angular velocity of the vehicle and accurately keeping the beam direction constant. By combining these two sensors, regardless of the presence or absence of an incoming beam, the antenna directs the transmission directivity in a certain direction, and keeps the same directivity in the same direction even when the vehicle moves. Have been. In addition, by providing the microwave phase shifter with frequency characteristics, directivity is formed in the same direction in both transmission and reception even when the transmission and reception frequencies are different.

[0003]

However, the phased array antenna of the first prior art can direct a transmission beam in the direction of a signal source if the direction of the signal source is known regardless of the presence or absence of an incoming signal. However, unless the direction of the signal source is unknown or the signal source itself moves such as a low orbit satellite, unless the movement is all predictable,
Tracking is impossible. In addition, even when the direction of the signal source is known, the magnetic sensor can know the absolute azimuth, but is easily affected by the magnetic field due to the surrounding metal, and the optical fiber gyro is affected by the magnetic field due to the surrounding metal. No,
Since the angular velocities are detected and integrated to determine the absolute azimuth, errors are easily accumulated. The method of performing accurate tracking while complementing each other by combining these features has a problem that the configuration is complicated and the performance is limited as described above.

In order to solve the above-mentioned problems, as a method of forming a transmission beam in the direction of an incoming signal, a reception phase difference is used as it is, or the transmission phase difference is converted according to transmission and reception frequencies. (Hereinafter referred to as a second conventional example) is disclosed by the present applicant in Japanese Patent Application No. 6-20 / 1990.
No. 3258 is proposed in the patent application. In the second conventional example, a transmission beam can be automatically formed in the arrival direction of a received wave without using a special sensor for knowing the direction of the arrival wave.

[0005] However, in the case of the transmission beam forming method of the second conventional example, when only one wave is received, even if the direction of the signal source is unknown, without using an azimuth sensor or the like, A transmit beam can be formed in the direction of the received wave, but if multiple multiplex waves are received from different directions due to reflection on a building, the ground, etc., the transmit beam should also be formed in the direction of arrival of those multiplex waves. Becomes This property is because when the transmission and reception frequencies are equal, the transmission waves transmitted through multiple paths of the incoming multiplex wave are received in the same phase at the partner station due to the reversibility of transmission and reception. No problem. However, when the transmission and reception frequencies are different, the reversibility of transmission and reception does not hold, and the transmission waves received by the partner station through multiple paths of the incoming multiplex wave are not necessarily in phase with each other at the partner station. It is not received and communication quality is degraded. Also, if the inclination angle of the equal phase plane of the received signal with respect to the antenna surface becomes large, phase uncertainty occurs due to the fact that the reception phase difference between the antenna elements exceeds the range of −π or more and + π or less. However, there is a problem that a transmission beam is not formed correctly.

SUMMARY OF THE INVENTION The object of the present invention is to solve the above problems and to solve the problem even when the direction of a partner station serving as a signal source is unknown.
Without using an azimuth sensor or the like, a transmission beam is formed in the direction of the arriving wave based on the reception signal for each antenna element obtained from the arriving wave transmitted from the signal source, and a plurality of multiplex waves arrive. It is an object of the present invention to provide an array antenna control method and a control apparatus capable of forming a single transmission main beam only in the direction of the maximum received wave even when the phase is uncertain in the environment or the reception phase difference.

[0007]

According to a first aspect of the present invention, there is provided an array antenna control method for controlling an array antenna including a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape. In the antenna control method, a plurality of reception signals received by the respective antenna elements of the array antenna are respectively used by using a common local oscillation signal, and an in-phase component having the same phase as the local oscillation signal, and the local oscillation signal The signal is converted into a quadrature signal composed of quadrature components orthogonal to each other, and a product of quadrature signals and quadrature components of quadrature signals in each of two first and second antenna elements which are mutually different pairs among the plurality of antenna elements. , The first data calculated by the sum of the products of: the in-phase component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. A predetermined transfer function is used to calculate the product of the cross component and the second data calculated by the difference between the product of the quadrature component of the quadrature signal of the first antenna element and the product of the quadrature signal of the quadrature signal of the second antenna element. And then filtered through a noise suppression filter having
Divides the filtered second data by the following data and calculates the arc tangent of the result of the division to obtain the difference between the received signal of the first antenna element and the received signal of the second antenna element. Is calculated, and the above calculated 2 is calculated.
A matrix comprising reception phase differences corresponding to all candidates of phase uncertainty caused by the fact that the reception phase difference between the two first and second antenna elements is limited to the range from -π to + π; By solving the Wiener-Hopf equation using the matrix of the position coordinates of a plurality of antenna elements constituting the array antenna and using the least squares method, all the candidates for the phase uncertainty are supported. An equation representing the first-order regression plane having the same phase is derived, and a plurality of first-order regression planes having the same phase corresponding to all the candidates for the phase uncertainty are calculated. The phase uncertainty is removed using the sum of squares of the residual between the first-order regression plane and the gradient coefficient of the first-order regression plane having the same phase, and the equal phase corresponding to the maximum received wave is removed. Features only one linear regression plane By correcting the received phase difference, the sign of the corrected received phase difference is inverted into a transmission phase difference by inverting the sign of the corrected reception phase difference. By transmitting a transmission signal from each of the antenna elements with a transmission phase difference and an equal amplitude, a transmission main beam is formed only in the direction of the maximum reception wave.

[0008]

[0009]

According to a second aspect of the present invention, there is provided a method of controlling an array antenna according to the first aspect of the present invention, wherein the phase uncertainty is eliminated from the first-order regression plane having the same phase. The phase difference is multiplied by the ratio of the transmission frequency to the reception frequency to obtain a transmission phase difference, which is converted into a transmission phase difference.

According to a third aspect of the present invention, there is provided an array antenna control device for controlling an array antenna including a plurality of antenna elements closely arranged in a predetermined arrangement shape. Using a common local oscillation signal for each of the plurality of reception signals received by each antenna element of the array antenna, an in-phase component that is in phase with the local oscillation signal and a quadrature component that is orthogonal to the local oscillation signal First conversion means for converting the signals into quadrature signals respectively, and the product of the in-phase components of the quadrature signals and the quadrature components among the two first and second antenna elements which are mutually different pairs of the plurality of antenna elements. , The in-phase component of the quadrature signal of the first antenna element and the quadrature signal of the second antenna element. A predetermined transfer function is used to calculate the product of the quadrature components and the second data calculated by the difference between the product of the quadrature component of the quadrature signal of the first antenna element and the product of the in-phase component of the quadrature signal of the second antenna element. And then filtered through a noise suppression filter having
Divides the filtered second data by the following data and calculates the arc tangent of the result of the division to obtain the difference between the received signal of the first antenna element and the received signal of the second antenna element. First calculating means for calculating the reception phase difference of
All candidates for phase uncertainty caused by the fact that the reception phase difference between each of the two first and second antenna elements calculated by the first calculation means is limited to the range from -π to + π. Solving a Wiener-Hopf equation using a least squares method, using a matrix consisting of a reception phase difference corresponding to the above and a matrix consisting of position coordinates of a plurality of antenna elements constituting the array antenna. Derives an equation representing the first-order regression plane of the same phase corresponding to all the candidates of the phase uncertainty, and obtains a plurality of 1s of the same phase corresponding to all the candidates of the phase uncertainty.
A second calculating means for calculating a next regression plane; a sum of squares of residuals between the reception phase difference and the first phase regression plane having the same phase; a gradient coefficient of the first phase regression plane having the same phase; Using,
Correcting means for removing the phase uncertainty and specifying only one first-order regression plane having the same phase corresponding to the maximum received wave to correct the received phase difference; and a receiving position corrected by the correcting means. Second conversion means for converting the sign of the phase difference into a transmission phase difference by inverting the sign of the phase difference, wherein the transmission phase difference between each of the two first and second antenna elements converted by the second conversion means is provided. In addition, a transmission main beam is formed only in the direction of the maximum reception wave by transmitting a transmission signal from each of the antenna elements with equal amplitude.

[0012]

[0013]

According to a fourth aspect of the present invention, there is provided the array antenna control device according to the third aspect, wherein the second conversion means includes the one-phase equal-phase signal having the phase uncertainty removed. The transmission phase difference is obtained by multiplying the reception phase difference calculated from the next regression plane by the ratio of the transmission frequency to the reception frequency to obtain a transmission phase difference.

[0015]

In the method of controlling an array antenna according to the first aspect of the present invention, a plurality of reception signals received by each antenna element of the array antenna are used in common with the local oscillation signal by using a common local oscillation signal. A quadrature signal composed of a certain in-phase component and a quadrature component orthogonal to the local oscillation signal, respectively, and a quadrature signal in each of two first and second antenna elements that are mutually different pairs among the plurality of antenna elements And the first data calculated by the sum of the product of the in-phase components and the product of the quadrature components, and the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element. Product and second data calculated by the difference between the product of the quadrature component of the quadrature signal of the first antenna element and the in-phase component of the quadrature signal of the second antenna element, After filtered is passed through a noise suppression filter having a constant of the transfer function, the is the filtered 1
Divides the filtered second data by the following data and calculates the arc tangent of the result of the division to obtain the difference between the received signal of the first antenna element and the received signal of the second antenna element. Is calculated, and the above calculated 2 is calculated.
A matrix comprising reception phase differences corresponding to all candidates of phase uncertainty caused by the fact that the reception phase difference between the two first and second antenna elements is limited to the range from -π to + π; By solving the Wiener-Hopf equation using the matrix of the position coordinates of a plurality of antenna elements constituting the array antenna and using the least squares method, all the candidates for the phase uncertainty are supported. An equation representing the first-order regression plane having the same phase is derived, and a plurality of first-order regression planes having the same phase corresponding to all the candidates for the phase uncertainty are calculated. The phase uncertainty is removed using the sum of squares of the residual between the first-order regression plane and the gradient coefficient of the first-order regression plane having the same phase, and the equal phase corresponding to the maximum received wave is removed. Features only one linear regression plane By correcting the received phase difference, the inverted received phase difference is converted into a transmission phase difference by inverting the sign of the corrected reception phase difference, and the converted first and second antenna elements are converted between the two first and second antenna elements. By transmitting a transmission signal from each of the antenna elements with a transmission phase difference and an equal amplitude, a transmission main beam is formed only in the direction of the maximum reception wave.

[0016]

[0017]

Further, in the array antenna control method according to the second aspect, in the array antenna control method according to the first aspect, a calculation is performed from the first-order regression plane having the same phase from which the phase uncertainty has been removed. The received phase difference
By multiplying the transmission phase ratio by multiplying the ratio of the transmission frequency to the reception frequency, the signal is converted into a transmission phase difference.

According to a third aspect of the present invention, in the array antenna control device, the first converting means converts a plurality of reception signals received by each antenna element of the array antenna into a common local oscillation signal. The signal is converted into a quadrature signal composed of an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal. Next, the first calculating means calculates the sum of the product of the in-phase components of the quadrature signals and the sum of the products of the quadrature components of each of the two first and second antenna elements, which are mutually different pairs, of the plurality of antenna elements. The first calculated by
And the product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element, the quadrature component of the quadrature signal of the first antenna element, and the second antenna The second data calculated by the difference between the products of the in-phase components of the quadrature signals of the elements are respectively passed through a noise suppression filter having a predetermined transfer function to be filtered, and then the filtered first The second filtered above with data
, And by calculating the arc tangent value of the division result, the reception phase difference between the reception signal of the first antenna element and the reception signal of the second antenna element is calculated. The second operation means is configured such that the reception phase difference between each of the two first and second antenna elements calculated by the first operation means is limited to a range from -π to + π. Using a matrix consisting of reception phase differences corresponding to all candidates of phase uncertainty occurring in the above and a matrix consisting of position coordinates of a plurality of antenna elements constituting the array antenna, using a method of least squares, Hopf (W
By solving the equation (ener-Hopf), an equation representing the first-order regression plane of the equiphase corresponding to all the candidates for the phase uncertainty is derived, and the equiphase corresponding to all the candidates for the phase uncertainty is obtained. Are calculated. Further, the correction means uses the sum of squares of the residual between the received phase difference and the first-order regression plane having the same phase and the gradient coefficient of the first-order regression plane having the same phase to obtain the phase. The uncertainty is removed, the reception phase difference is corrected by specifying only one primary regression plane having the same phase corresponding to the maximum reception wave, and the second conversion means is corrected by the correction means. The signal is converted into a transmission phase difference by inverting the sign of the received phase difference. Then, by transmitting a transmission signal from each of the antenna elements with a transmission phase difference and equal amplitude between each of the two first and second antenna elements converted by the second conversion means, the maximum reception wave The transmission main beam is formed only in the direction.

[0020]

[0021]

According to a fourth aspect of the present invention, there is provided the array antenna control device according to the third aspect, wherein the second converting means includes the second phase conversion means for eliminating the phase uncertainty. The transmission phase difference is converted into a transmission phase difference by multiplying the reception phase difference calculated from the first-order regression plane by the ratio of the transmission frequency to the reception frequency.

[0023]

Embodiments of the present invention will be described below with reference to the drawings. <First Embodiment> FIG. 1 is a block diagram of a receiving section of an automatic beam capture and tracking device for a communication array antenna according to a first embodiment of the present invention. The automatic beam capture and tracking device of the communication array antenna according to the present embodiment has a predetermined length of, for example, 受 信 of the reception frequency, の of the transmission frequency, or の of the average of the reception frequency and the transmission frequency. The directivity of the array antenna 1 including a plurality of N antenna elements A1, A2,..., Ai,. Aim at high speed in the direction of the incoming beam of a radio signal wave such as a wave, and track it. Here, in particular, the acquisition and tracking device of the present embodiment includes the DBF unit 4 and the transmission weight operation circuit 3
0 is provided. Then, even when the azimuth of the partner station serving as the signal source is unknown, the arriving signal based on the baseband signal for each antenna element obtained from the arriving wave transmitted from the signal source without using an azimuth sensor or the like. Forming a transmission beam in the direction of the wave, and also in an environment where multiple multiplexes arrive, or even when a phase uncertainty occurs in the reception phase difference, remove the effects and phase uncertainty of those multiplexes, A single transmit main beam is formed only in the direction of the maximum received wave.

As shown in FIG. 1, the array antenna 1
It includes N antenna elements A1 to AN and circulators CI-1 to CI-N as transmission / reception separators. The receiving modules RM-1 to RM-N are respectively
Using the low-noise amplifier 2 and the common first local oscillation signal output from the first local oscillator 11, down-converts a received radio signal having a radio frequency into an intermediate frequency signal having a predetermined intermediate frequency. And a converter (D / C) 3.

The receiving unit of the acquisition and tracking device further uses N A / D converters AD-1 to AD-N and a common second local oscillation signal output from the second local oscillator 12 to The intermediate frequency signal after the A / D conversion is quasi-synchronously detected and two baseband signals orthogonal to each other (hereinafter, these 2
One baseband signal is called a quadrature baseband signal. N) quasi-synchronous detection circuits QD-1 through QD
D-N and said converted reception weight W ₁ ^RX for each quadrature baseband signal as maximum ratio combining based on the quadrature baseband signal, W ₂ ^RX, ..., and calculates the W _N ^RX, each The reception weights W ₁ ^RX , W ₂ ^RX ,..., W _N ^RX calculated on the quadrature baseband signal are multiplied, in-phase synthesized, and output to the demodulator 5. Receive weights W ₁ ^RX , W ₂ ^RX , ..., W _N
Based on ^RX , the transmission weight W is calculated by the method according to the present invention.
₁ ^TX , W ₂ ^TX ,..., W _N ^TX
And a demodulation for performing synchronous detection or delay detection by a predetermined baseband demodulation process from a baseband signal output from the DBF unit 4 and extracting desired digital data and outputting the digital data as reception data. And a vessel 5.

In the receiving section, each of the antenna elements A1 to AN in the array antenna 1 to the DBF section 4
The cascade connection is made for each antenna element system. Since the signal processing for each system of each antenna element in the receiving unit is similarly executed, the radio signal received by the antenna element Ai (Ai is represented as one of the antenna elements A1 to AN). The processing for waves will be described.

The radio signal wave received by the antenna element Ai is transmitted to the circulator CI-i and the receiving module RM-
The signal is input to the down converter 3 via the low noise amplifier 2 in i. The down converter 3 in the receiving module RM-i is connected to the common first
Using the local oscillation signal, the input radio signal is frequency-converted into an intermediate frequency signal having a predetermined intermediate frequency.
Output to the quasi-synchronous detection circuit QD-i via the D converter AD-i. The quasi-synchronous detection circuit QD-i is connected to the second local oscillator 1
2 using the common second local oscillation signal output from the A / D converter, quasi-synchronous detection of the input A / D-converted intermediate frequency signal and conversion into two orthogonal baseband signals I _i , Q _i , and a DBF
Output to section 4.

The DBF unit 4 receives reception weights W ₁ ^RX , W ₂ ^RX ,..., For each of the orthogonal baseband signals for performing maximum ratio combining based on the converted orthogonal baseband signals.
W _N by calculating the ^{R X,} reception weight W ₁ ^RX computed with respect to the respective quadrature baseband ^{_{signal, W 2 RX, ..., W}} N RX of the by-phase synthesis was multiplied outputs to the demodulator 5. Further, based on the reception weights W ₁ ^RX , W ₂ ^RX ,..., W _N ^RX calculated by the DBF unit 4, the transmission weight calculation circuit 30 converts the transmission beam in the direction of the direct wave by the method according to the present invention. Also, even in an environment where a plurality of multiplexed waves arrive, or even when a phase uncertainty occurs in the reception phase difference, the influence and phase uncertainty of those multiplexed waves are removed, and only in the direction of the maximum received wave. The transmission weights W ₁ ^TX , W ₂ ^TX ,..., W _N ^TX are calculated and output to the transmission local oscillator 10 so as to form a single transmission main beam. On the other hand, the demodulator 5 performs synchronous detection or delay detection from the baseband signal output from the DBF unit 4 by a predetermined baseband demodulation process, extracts desired digital data, and outputs it as received data. The circuit processing of the DBF unit 4 and the transmission weight calculation circuit 30 will be described later in detail.

FIG. 2 is a block diagram of a transmission unit of the acquisition and tracking device. The transmission unit includes N transmission modules TM-1 to TM-N, N quadrature modulation circuits QM-1 to QM-N, and an in-phase distributor 9 as shown in FIG. Here, each of the quadrature modulation circuits QM1 to QM-N includes a quadrature modulator 6 and a transmission local oscillator 10, and each of the transmission modules TM-1 to TM-N converts the input intermediate frequency signal into a predetermined signal. Up-converter (U) that converts the frequency of the
/ C) 7 and a transmission power amplifier 8. Here, each of the transmission local oscillators 10 in the quadrature modulation circuits QM-1 to QM-N is, for example, an oscillator using a DDS (Direct Digital Synthesizer) driven by the same clock, and a transmission weight operation circuit. Based on the transmission weights W ₁ ^TX , W ₂ ^TX ,..., W _N ^TX input from _N , _N transmission local oscillation signals having respective phases corresponding to them are generated.

The transmission baseband signal S which is transmission data
^{After the TX} is input to the in-phase distributor 9, it is distributed in-phase and input to each of the quadrature modulators 6 of the quadrature modulation circuits QM-1 to QM-N. For example, the quadrature modulator 6 in the quadrature modulation circuit QM-1 converts the transmission local oscillation signal generated by the transmission local oscillator 10 into quadrature modulation such as QPSK according to the transmission baseband signal ^STX input from the in-phase distributor 9. After doing
The intermediate frequency signal after the quadrature modulation is transmitted to the transmission module TM-1.
The signal is input to the circulator CI-1 in the array antenna 1 as a transmission radio signal via the up-converter 7 and the transmission power amplifier 8 inside. Here, the quadrature modulator 6
Converts the input transmission baseband signal S ^TX into a transmission orthogonal baseband signal by performing serial / parallel conversion, and then transmits each other in accordance with the transmission orthogonal baseband signal.
The intermediate frequency signal is obtained by orthogonally modulating and combining the transmission local oscillation signals having a phase difference of °. Then, the transmission radio signal is transmitted and radiated from the antenna element A1. Further, similar signal processing is performed in each system of the transmission unit connected to the antenna elements A2 to AN. Therefore, the transmission signals weighted by the transmission weights W ₁ ^TX , W ₂ ^TX ,..., W _N ^TX are radiated from the antenna elements A1 to AN. In this embodiment, the transmission signal transmitted from each antenna element Ai is weighted by transmission weights W ₁ ^{T X} , W ₂ ^TX ,..., W _N ^TX as described in detail later. The weights are transmitted at equal amplitudes only by changing the phase.

In this embodiment, for example, N = 16 antenna elements A1 to A16 are juxtaposed at a predetermined interval in a lattice shape. As described above, the interval is a half wavelength of the transmission frequency, a half wavelength of the reception frequency, or a half wavelength of an average thereof. The antenna elements A1 to AN are, for example, circular patch microstrip antennas. In a modification of the one-dimensional linear array antenna, four antenna elements A1 to A4 are arranged in a straight line at a distance from each other.

FIG. 4 is a block diagram showing signal processing in the DBF unit 4. In the signal processing in the DBF unit 4 of the present embodiment, A is set for each of the antenna elements A1 to AN.
This is performed on a quadrature baseband signal composed of an I component and a Q component that has been / D converted and quasi-coherently detected. Here, assuming that the number of antenna elements of the array antenna 1 is N, an antenna element Ar serving as a phase reference and an arbitrary antenna element Ai including the antenna element Ar (1 ≦ r ≦ N, 1 ≦ i ≦ N)
The baseband signal S _r, S _i, respectively as follows when expressed by a complex number in. Here, referred to as the reference base band signal of the base band signal S _r, the base band signal S _i that the processing baseband signal. The antenna element serving as a phase reference (hereinafter referred to as antenna element Ar) is a predetermined one of the N antenna elements. The antenna element receiving the processed baseband signal S _i of processing antenna element Ai.

[0033]

[Number 1] _{S r = I r + jQ r} = a r exp {j (φ m + θ r)}

S _i = I _i + jQ _i = a _i exp {j (φ _m + θ _i )} = a _i exp {j (φ _m + θ _r + Δθ _{r, i} )}

Here, a _r is the amplitude component of the reference baseband signal, a _i is the amplitude component of the processed baseband signal, and φ _m is the modulation phase. Θ _r is the phase difference between the reference baseband signal S _r and the local oscillation signal generated by the second local signal oscillator 12, and θ _i is the processed baseband signal S _i and the second local signal oscillator 12 Δθ _{r, i} is the phase difference from the local oscillation signal generated by
Is the reference baseband signal S _r and the processed baseband signal S _i
Is the phase difference between At this time, the processing antenna element A
The received signal power | S _i | ^{2 at} _i is represented by the following equation (3).

[0035]

| S _i | ² = I _i ² + Q _i ² = a _i ²

In this embodiment, it is desirable that the antenna element having the maximum received signal power in each processing antenna element Ai be used as a phase reference for in-phase in terms of the in-phase accuracy, but in practice, the reference is used. If the antenna element is switched during communication, a phase jump will occur. In this case, this is predetermined and fixed. Next, φ _m and θ _r in Equations 1 and 2 can be eliminated by using a complex conjugate product arithmetic expression represented by the following Equation 4.

[0037]

[Number 4] ^{_{S r * · S i = a}} r a i · exp (jΔθ r, i)

Here, * represents a complex conjugate. The complex conjugate product operation unit 21 in FIG. The real part and the imaginary part in Equation 4 are expressed by the following Equations 5 and 6, respectively.

[0039]

[Number 5] ^{_{Re [S r * · S i}} ] = a r a i · cosΔθ r, i = I r I i + Q r Q i

[6] ^{_{Im [S r * · S i}} ] = a r a i sinΔθ r, i = I r Q i -I i Q r

[0040] Thus, a complex conjugate of the number ^{_{4 (S r * · S i}} ) (S r * · S i) * a by multiplying the baseband signal S _i of the antenna element Ai, baseband signal processing S _i is in-phase with the reference baseband signal S _r, and the processed base band signal S _i ′ after in-phase is expressed by the following equation (7).

[0041]

(7) S _i ′ = (1 / │S _r │) ・ (S _r ^*・ S _i ) ^*・ S _i = (1 / │S _r │) ・ (S _r・ S _i ^* ) ・ S _i = A _i ² exp {j (φ _m + θ _r )} where

[Equation 8] _{│S r │ = a r = √} (I r 2 + Q r 2)

Where | S _r | is the reference antenna element Ar
A magnitude of the reference baseband signal S _r in, the reciprocal, as shown in Equation 7, by multiplying the common to the antenna elements Ai, the baseband signal processing S _i
Are normalized by the total received power received by the array antenna 1. When the above equation 7 is expressed by a vector, the following equation 9 is obtained.

[0043]

(Equation 9)

By performing this vector rotation operation on all the antenna elements Ai, all the processed baseband signals _Si are relatively in-phase. Since the method of this embodiment according to the present invention does not perform the operation of tan ^-1 and directly uses the results of Equations 5 and 6 as rotation matrix elements, as can be seen from Equation 9, the processing baseband signal S _i The amplitude | a _i | is automatically multiplied as a coefficient. Therefore, combining this with all antenna elements Ai is nothing but performing maximum ratio combining (MRC). However, in actual communication, receiver noise, modulation components, band limitation, and the like cause errors in the in-phase and amplitude fluctuations, and accordingly, the weight of the maximum ratio combining also increases the error. In order to suppress these effects, the above equation 9 is replaced as follows by using low-pass filters 22 and 23 which are digital filters having a filter function F (•).

[0045]

(Equation 10)

Here, the cutoff frequency of the low-pass filters 22 and 23 will be described. The low-pass filters 22 and 23 in FIG. 4 are constituted by digital filters such as an FIR filter or an IIR filter. The higher the cutoff frequency, the more easily the reception noise is affected.
If the received power per element is low, the accuracy of acquisition and tracking tends to deteriorate. Conversely, the lower the cut-off frequency, the more the influence of the reception noise is removed, so that even if the reception power per antenna element is low, the acquisition and tracking can be performed. However, the time constant of the bandpass filter becomes longer as the band becomes narrower.
The ability to follow a sudden change in the direction of arrival of the received wave becomes dull. The change in the direction of direct arrival of the received wave in ordinary mobile communication or the like is sufficiently slow as compared with the calculation time for performing beamforming, so that the reception noise becomes dominant. For this reason, the cutoff frequency of the low-pass filters 22 and 23 is determined by the received signal power versus the received noise power, and when the received power is small as in satellite communication, the cutoff frequency may be set as low as the hardware allows. preferable. The low-pass filter 2
Specifically, the cutoff frequencies 2 and 23 are set to about 1/100 to 1/1000 of the sampling frequency.

In the DBF 4 shown in FIG. 4, taking into account the delay caused by the low-pass filters 22 and 23, the timing is adjusted so that the phases of the two signals input to the multipliers 26 and 27 are the same. The delay buffer circuits 24 and 25 for adjusting the delay time are inserted.

The configuration and operation of the above-described DBF unit 4 will be described with reference to FIG. The reference baseband signal S _r is calculated by an absolute value calculating unit 20 and a complex conjugate product calculating unit 2.
1 and to a multiplier 26 via a delay buffer circuit 24. On the other hand, the processing baseband signal S _i is input to the complex conjugate product operation unit 21 and also to the multiplier 27 via the delay buffer circuit 25. The absolute value calculator 20 calculates the reference baseband signal S _r
The absolute value | S _r | is calculated based on
A signal indicating S _r | is output to dividers 28 a and 28 b via a low-pass filter (LPF) 22. On the other hand, the complex conjugate product calculating section 21, the reference baseband signal based on the S _r processed baseband signal _{S i, (S r · S} i *) low frequency signals indicative of the result of the operation performed operations Pass filter 2
3 to the multiplier 27 and the divider 28b. The multiplier 26 multiplies the two input signals and processes the signal indicating the result of the multiplication by processing the reference baseband signal S _r ′.
Output as On the other hand, the multiplier 27 multiplies the two input signals by a signal that indicates a result of the multiplication.
a, and a divider 28a divides the signal input from the multiplier 27 by the signal input from the low-pass filter 22, and processes the signal indicating the result of the division to obtain an in-phase processed baseband. The signal is output to the in-phase synthesizer 29 as a signal S _i ′. The divider 28b divides the signal input from the low-pass filter 23 by the signal input from the low-pass filter 22, and outputs a signal indicating the result of the division by the reception weight W
Output to the transmission weight calculation circuit 30 as _i ^RX . Then, the in-phase combiner 29 outputs all N processed baseband signals S _i ′ (i = 1, 2,...,
N) are combined in phase and output to the demodulator 5. Accordingly, as can be seen from FIG. 4 and the above description, weighting for maximum ratio combining is automatically performed in the process of in-phase, and the configuration of the DBF unit 4 is extremely simple.

On the other hand, as shown in FIG. 1, the quasi-synchronous detection is used for the detection of the baseband signal.
Is not synchronized with the second local oscillation signal for reception. Therefore, it is necessary to connect the baseband processing type demodulator 5 to the subsequent stage of the DBF unit 4 to synchronize the carrier phase.
Further, when the symbol delay of the multipath wave signal is so large that it cannot be ignored, it is necessary to add a suitable adaptive equalizer (EQL), not shown. As a result of these processes, the apparatus simultaneously forms a plurality of main beams in the direction of a direct wave and a multipath delayed wave (hereinafter, referred to as a multipath wave), and combines them into a carrier power to noise power ratio (received CNR). From the viewpoint, the images are optimally synthesized and tracked. Since a feedback loop is not used for beamforming, a low reception CNR is obtained as in the second conventional example.
Can operate stably at high speed.

Next, the transmission weight calculation circuit 30 shown in FIG.
Will be described. First, consider a case where the spacing between the antenna elements of the transmitting array antenna and the spacing between the antenna elements of the receiving array antenna are equal in terms of wavelength. In this case, in order to form a transmission beam in the same direction as the received incoming wave is typically complex conjugate (W _i ^RX) of the received weights W _i ^RX used in recipient ^* a may be used as the transmission weight W _i ^TX . That is, it is expressed as follows by inverting the sign of the phase of the reception weight and using it as the phase of the transmission weight.

[0051]

[Expression 11] S _i ^TX = W _i ^TX · S ^TX = (W _i ^RX ) ^* · S ^TX

[Number 12] ^{_{W i RX = {1 / F}} (│S r │)} · F (S r · S i *)

Here, S ^TX is a transmission baseband signal input to the device, S _i ^TX is a transmission base band signal supplied to the antenna element Ai, and W _i ^TX is a transmission weight in the antenna element Ai. is there. As a result, a transmission beam having the same shape as the reception beam is formed.However, when a large multipath delayed wave exists, the beam is formed not only in the direction of the direct wave but also in the direction of the delayed wave. Will be done. In the case where transmission and reception can be regarded as having the same frequency and both paths are almost equal, such as TDD (abbreviation of Time Division Duplex) that performs transmission alternately using the same frequency, this alone is sufficient, and diversity can be easily achieved. A transmission system can be configured. However, if the transmission and reception frequencies are different, the phase difference between the paths becomes unequal, so that a diversity transmission system cannot be constructed, and transmission in the direction of the delayed wave must be suppressed as much as possible. Therefore, here, assuming that the level of the direct wave is the highest among a plurality of multipath waves, a method of forming a single main beam only in the direction of the direct wave by removing the effect of the delayed wave is described. This will be described below.

From the above equations (5) and (6), the reception phase difference Δθ between the reference antenna element Ar and an arbitrary antenna element Ai is obtained.
_{r and i} are expressed by the following equation (13).

[0054]

Equation 13] ^{_{Δθ r, i = tan -1 {}} F (I r · Q i -I i · Q r) / F (I r · I i + Q r · Q i)}

However, since Δθ _{r, i} obtained here is in the range from −π to + π, the phase difference becomes many rotations as the antenna element interval increases (that is, 2π).
Becomes an integral multiple of. ) Causes phase uncertainty. The method of removing the phase uncertainty will be described later in detail, but here it is assumed that this has already been removed. Assuming that there is no delay wave and no noise, the phase difference Δθ _{r, i} should be on a certain first-order phase plane, but if there is a delay wave or noise, it will be around the plane. It will be dispersed. It is considered that a single transmission main beam is formed only in the arrival direction of the direct wave having the highest level by setting the value obtained by regressing these on the phase plane as the excitation phase and exciting at the same amplitude. As a method for returning to the first-order phase plane, a minimum 2
Regression analysis (LSR) using the multiplication method can be used. First, the first-order phase regression plane is set as follows.

[0056]

## EQU14 ## Δθ _{r, i} ^LSR = ax + by + c

Here, it is assumed that the array antenna 1 is located on the xy plane of the xyz coordinate system as shown in FIG. Coefficients a, b, and c are given by the following Wiener-Hopf (Wi)
ener-Hopf) equation.

[0058]

X ^T · X · A = X ^T · Θ where,

[0059]

(Equation 16)

[Equation 17]

(Equation 18)

Here, the coordinates of the antenna element Ai of the array antenna 1 are (x _i , y _i ) (i = 1, 2,..., N), and X is a matrix determined by the arrangement of the antenna element Ai. A is a coefficient a, b, indicating the first-order phase regression plane.
is a matrix composed of c, and Θ is a matrix composed of the phase difference Δθ _{r, i} in each antenna element Ai. The matrix A in the above equation 15 is obtained by rewriting the equation 15 to obtain the following equation 19
It is represented as

[0061]

A = (X ^T · X) ⁻¹ · X ^T · Θ

In the above Expression 19, (X ^T · X) ^-1 X ^T is a 3 × N matrix determined by the element arrangement of the array antenna 1 and can be calculated in advance. The parameter A of the regression plane can be obtained from the obtained phase matrix で by N times of product-sum operations. On the other hand, as described above, the phase difference Δθ _{r, i} obtained from Expression 13 has a phase uncertainty. Is no longer required. Therefore, the following three types of phase uncertainty and phase correction in that case are executed.

[0063]

## EQU20 ## Correction (I) Δθ ′ _{i-1, i} = Δθ _{i-1, i} (no correction)

Correction (II) If Δθ _{i-1, i} <−k, Δθ ′ _{i−1, i} = Δθ _{i−1, i} + 2π Otherwise, Δθ ′ _{i−1, i} = Δθ _{i-1, i} (no correction)

## EQU22 ## Correction (III) If k ≦ Δθ _{i−1, i} Δθ ′ _{i−1, i} = Δθ _{i−1, i} −2π Otherwise, Δθ ′ _{i−1, i} = Δθ _{i-1, i} (no correction)

Here, the phase difference Δθ _{i−1, i} is the phase difference between the nearest neighboring antenna elements, and is expressed by the following equation (23).

[0065]

## EQU23 ## Δθ _{i-1, i} = Δθ _{r, i} −Δθ _{r, i-1}

K is 0 <k <π and is a phase threshold value indicating the degree of disturbance of the reception phase difference due to the multipath wave, and is set according to the assumed strength of the multipath wave. Here, the setting of the phase threshold value k in the examination of the reception phase uncertainty will be described below.

In this embodiment, three types of phase uncertainties and phase corrections as shown in Expressions 20 to 22 are performed, and the positive phase threshold value k (> 0) set there is determined by the phase It is a parameter that determines the sensitivity of correction. That is, the smaller k is, the higher the correction sensitivity is, and the sensitivity becomes maximum when k = 0. Conversely, the correction sensitivity decreases as k increases,
Above π, the phase is hardly corrected. Therefore, if the received wave is only one direct arriving wave and the reception intensity of the multiple arriving wave is sufficiently smaller than the direct arrival wave, k ≒ 0 is desirable. And, when the arrival direction of the direct wave is near the front of the antenna,
As shown in FIG. 13, an erroneous correction may occur because the reception phase plane is not flat. This is because the sensitivity of the correction is too high. By setting k to a certain value of k> 0, if the correction sensitivity is slightly reduced, a correct correction phase can be obtained. If the phase threshold value k is set to about π / 6, correct phase correction can be performed even when a multiple arriving wave having substantially the same level as the direct arriving wave is received. Therefore, in this embodiment, the phase threshold value k is preferably set to π / 6.

Here, when the array antenna 1 is arranged in the xy coordinate system as shown in FIG. 5, the phase plane is expressed by the following equation (24).

[0069]

[Equation 24] Δθ _{r, i} ^LSR = ax + by + c

Here, while there are three correction methods (I) to (III) in the x-axis direction, there are three correction methods (I) to (III) in the y-axis direction. Therefore, a total of nine types of phase regression planes are obtained. Hereinafter, for example, the correction (I-II) indicates a phase regression plane when the correction (I) is performed in the x-axis direction (actually not corrected) and the correction (II) is performed in the y-axis direction. 9 corresponding to three types of phase uncertainty for each axis and represented by the following equation 25 in total
As a result, three phase regression planes are obtained.

[0071]

(A) At the time of correction (II), Δθ _{r, i
^{LSR (II) = a I x}} + b when _I y + c (b) Correction of _{(I-II), Δθ r} , i LSR (I-II) = a I
x + b _II y + c (c) For correction (I-III), Δθ _{r, i} ^{LSR (I-III)} = a
When _{I x + b III y + c} (d) Correction _{(III), Δθ r, i} LSR (III) = a II
x + b _I y + c (e) For correction (II-II), Δθ _{r, i} ^{LSR (II-II)} = a
_II x + b _II y + c (f) For correction (II-III), Δθ _{r, i} ^{LSR (II-III)} =
a _II x + b _III y + c (g) At the time of correction (III-I), Δθ _{r, i} ^{LSR (III-I)} = a
_III x + b _I y + c (h) For correction (III-II), Δθ _{r, i} ^{LSR (III-II)} =
a _III x + b _II y + c (i) For correction (III-III), Δθ _{r, i} ^{LSR (III-III)}
= A _III x + b _III y + c

Here, the residual sum of squares for each correction is defined as in the following equation (26).

[0073]

(A) Correction (II) (B) Correction (I-II) (C) Correction (I-III) (D) Correction (II-I) (E) For correction (II-II) (F) Correction (II-III) (G) For correction (III-I) (H) For correction (III-II) (I) For correction (III-III)

From this, the phase uncertainty is determined by the residual square sum S
Using S = Σ (Δθ _{r, i} −Δθ _{r, i} ^LSR ) ² and the phase gradient | a |, | b | A phase regression plane is selected.

A phase regression plane selection process in the case of a two-dimensional array will be described with reference to FIGS. As shown in FIG. 8, in step S11, the correction (I-
I), (I-II), (II-I) and (II-II) residual sum of squares SS _(II) , SS _{(I- II)} , SS _(II-I) , SS
_(II-II) , and in step S12, the residual square sum SS
_{If (II )} is the minimum, the correction (I−
The phase regression plane in I) is selected, and the process ends. If the residual sum of squares SS _(I-II) is the smallest in step S13, the corrections (I-II) and (I-II) are made in step S22.
−III) gradient | b | _(I-II) , | b |
Compare _(I-III) . Then, in step S23, | b |
_{If (I-II)} <| b | _(I-III) , step S24
To select the phase regression plane in the correction (I-II) and terminate the processing. On the other hand, in step S23, | b | _(I-II)
If ≧ | b | _(I-III) , the phase regression plane in the correction (I-III) is selected in step S25, and the process ends.

If NO in step S13, and if the residual sum of squares SS _(II-I) is the minimum in step S14 in FIG. 9, correction is made in steps S26 (II-I) and (III-I).
| A | _(II-I) , | a | _(III-I)
Compare. Then, in step S27, | a | _(II-I) <
If | a | _(III-I) , the correction (I
The phase regression plane in II) is selected, and the process ends. On the other hand, in step S27, | a | _(II-I) ≧ | a |
_{If (III-I)} , the correction is made in step S29 _(III-I).
The phase regression plane in I) is selected, and the process ends.

If NO in step S14, the corrections (II-II) and (III-II) are made in step S30 in FIG.
| A | _(II-II) , | a |
Compare _(III-II) . Then, in step S31, | a |
_{If (II-II)} <| a | _(III-II) , step S4
When 0, the gradients | b | _(II-II) and | b | _(II-III) of the regression plane in the corrections (II-II) and _(II-III) are compared. Then, in step S41, | b | _{(II- II)} <| b | _(II-III)
If, the phase regression plane in the correction (II-II) is selected in step S42, and the process ends. on the other hand,
Step S41 In _{| b | (I III) ≧} | b | (II-III) is when a selects a phase regression plane in the correction in the step S43 (II-III) and finishes the process.

Further, in step S31, | a | _(II-II)
If | ≧ | a | _(III-II) , the gradient of the regression plane in the corrections (III-II) and (III-III) in step S32 |
b | _(III-II) and | b | _(III-III) . And
If | b | _(III-II) <| b | _(III-III) in step S33, the phase regression plane in the correction (III-II) is selected in step S44, and the process ends. on the other hand,
If | b | _(III-II) ≥ | b | _(III-III) in step S33, the phase regression plane in the correction (III-III) is selected in step S45, and the process ends.

Next, for simplicity, a method of removing phase uncertainty in the case of a one-dimensional linear array antenna (modification) will be described. That is, when the N antenna elements Ai are juxtaposed on a straight line, the phase plane is expressed by the following Expression 27.

[0080]

(27) Δθ _{r, i} ^LSR = ax + c

Here, for each of the cases of Equations 20 to 22,
By applying the above Expression 27, the following three types of phase regression planes can be obtained.

[0082]

Equation 28] [Delta] [theta] _r When (a) Correction ^{_{(I), i LSR (I}} ) = a I x + c Δθ r When _I (b) Correction ^{_{(II), i LSR (II}} ) = a II x + c II (c ) Correction (III) Δθ _{r, i} ^{LSR (III)} = a _III x + c _III

Here, the residual sum of squares for each correction is defined by the following equation (29).

[0084]

(A) Correction (I) (B) Correction (II) (C) Correction (III)

From the above, the phase uncertainty is calculated by using the sum of squares of the residual SS = Σ (Δθ _{r, i} −Δθ _{r, i} ^LSR ) ² and the phase gradient | a | of the regression plane as shown in FIG. It is removed by the regression plane selection processing, and one phase regression plane is selected.

The phase regression plane selection processing in the case of a one-dimensional array will be described with reference to FIG. As shown in FIG. 7, in step S1, the residual sums of squares SS _(I) and SS _(II) in the corrections (I) and _(II) are compared, and in step S2, SS _(I) <SS _(II). In step S3, the phase regression plane in the correction (I) is selected in step S3, and the process ends. On the other hand, when SS _(I) ≧ SS _(II) in step S2, the gradient | a | in corrections (II) and (III)
_(II) and | a | _(III) are compared, and | a |
_{If (II)} <| a | _(III) , the phase regression plane in the correction (II) is selected in step S6, and the process ends. On the other hand, in step S5, | a | _(II) ≧ | a | _(III)
If, the phase regression plane in the correction (III) is selected in step S7, and the process ends.

FIG. 11 is a diagram showing a 1-phase signal obtained by the least square method of the reception phase.
FIG. 12 is an explanatory diagram showing regression processing to the next plane, and FIG.
FIG. 9 is an explanatory diagram showing inspection and removal of phase uncertainty in that case. As shown in FIG. 11, when only the direct wave is received, the reception phase difference Δθ _{r, i} between the antenna elements Ai is located on a straight line depending on the position of the antenna element Ai. If received, it will deviate from that straight line. FIG. 12 shows a case where the process reaches step S6 and selects the phase regression plane in the correction (II).

In the above process of selecting a phase regression plane,
The phase plane corresponding to the direction of the strongest direct wave can be estimated and detected. In the other phase planes, the residual sum of squares increases and the phase gradient becomes steep. From the reception phase difference Δθ _{r, i} ^LSR thus determined, the transmission weight W _i ^TX
Can be calculated by the following 30.

[0089]

## EQU30 ## W _i ^TX = exp (jθ _i ^TX ) = exp (−jΔθ _{r, i} ^LSR )

Here, the amplitude component of the transmission weight is set to 1 commonly for all antenna elements Ai, and the wave source distribution is made uniform. Furthermore, when the transmitting and receiving array antenna 1 is used and the transmitting and receiving frequencies are different, the transmission main beam can be correctly formed directly in the direction of the incoming wave by multiplying the excitation phase by the frequency ratio. That is,
It is expressed by the following equation 31. Here, f ^TX and f ^RX are a transmission frequency and a reception frequency, respectively.

[0091]

W _i ^TX = exp (jθ _i ^TX ) = exp ｛−j (f ^TX / f ^RX ) Δθ _{r, i} ^LSR 【

FIG. 6 is a block diagram showing a transmission weight calculation circuit 30 for executing the above processing. In FIG. 6, the phase difference calculator 31-i (i = 1, 2,..., N)
Based on the reception weight W _i ^RX input from DBF unit 4, and calculates the phase difference [Delta] [theta] _r, the _i by performing tan ^-1 calculation of the reception weight W _i ^RX, least squares processor 32
−j (j = 1, 2,..., 9). The least square processing units 32-j (j = 1, 2,..., 9) are provided with nine processing units corresponding to the nine types of phase regression planes represented by Expression 25. Each least-squares processing unit 32-j calculates the coefficients a, b, and c of the equation of the phase plane set by solving the Wiener-Hopf equation represented by Expression 15, and calculates the calculated coefficients. By substituting a, b, and c into Equation 25, the received phase difference Δθ _{r, i} ^LSR (i =
1, 2,..., N) and outputs the result to the selector 34.
On the other hand, the phase regression plane selection unit 33
Based on the phase regression plane calculated by 2-j, the phase regression plane selection processing shown in FIGS. 8 to 10 is executed to determine a phase regression plane to be selected, and the phase regression plane determined to be selected is selected. The information is output to the selector 34. The selector 34 selects only the N reception phase differences Δθ _{r, i} ^LSR input from the least square processing unit 32-k corresponding to the phase regression plane determined to be selected, and selects the transmission weight calculation unit 3
5 is output. In response, the transmission weight calculator 3
5 calculates the transmission weight W _i ^TX (i = 1, 2,..., N) by executing the operation of ^Expression 31 based on the N received phase differences Δθ _{r, i} ^LSR input. Output.

Further, the results of a simulation performed by the inventor on the apparatus configured as described above will be described below. In order to evaluate the apparatus of this embodiment, a numerical simulation was performed under the conditions shown in Table 1. As a modified example, a basic four-element half-wavelength-spaced linear array antenna is used as the array antenna 1, and the modulation method is differentially coded synchronous detection QPSK (transmission speed: 16 kb).
ps). Also, as the low-pass filters 22 and 23 for in-phase reception signals, a second-order narrow band IIR (In
finite Impulse Response) filter was used.

[0094]

[Table 1] Specifications of the simulation ─────────────────────────────────── Modulation method 16 kbps differential coding Synchronous detection QPSK Modulation frequency 32 kHz (used as intermediate frequency) Sampling frequency 128 kHz (16 samples / symbol) A / D resolution 8 bits Additional noise Gaussian noise Antenna 4-element linear array of point radiation sources Antenna element spacing Carrier half-wavelength roll-off filter 10-tap FIR filter, roll-off rate 50%, cut-off frequency 8 kHz, transmission band-pass filter, bandwidth bit length product BT = 2, reception band-pass filter, bandwidth bit length product BTm = 1, carrier regeneration method feed-forward phase estimation clock regeneration method, decision directory Ted method ───────────────────── ──────────────

FIG. 14 shows that a direct wave arrives from a direction of -45 °, and a multipath wave is at a level of -3 dB and a phase difference of π / 2 (at the center of the array antenna 1) at + 15 ° compared to the direct wave. , The directivity pattern by the maximum ratio combining (MRC) reception when arriving from the direction of Equal Gain Combining (EGC) in which the received signals received by each antenna element Ai are combined with equal gain, and there is no multipath wave. It is a comparison with the case. Note that the ratio of received carrier power to noise power of a direct wave (hereinafter referred to as received CNR) is 4 dB.
And In equal gain combining, the effect on the directivity pattern by the multipath wave is slight, but in the maximum ratio combining, a beam is also formed in the arrival direction of the multipath wave, and as a result, both the direct wave and the multipath wave are captured It can be seen that directional diversity for re-synthesis is realized.

FIGS. 15 and 16 show directivity patterns when the phase of the multipath wave with respect to the direct wave changes. Here, the phase delay value is 0, π / 2 or (3π) /
2 and π. A phase delay value = 0 indicates that the phases of the two waves are the same at the center of the antenna. In order to clarify the characteristics of the directivity pattern, the reception CNR of the direct wave is set to 30 dB. In the case of FIG. 15 where the directions of the direct wave and the multipath wave are relatively close (when the arrival direction of the multipath wave is −15 °), when the phase delay value = 0, two waves are captured by the same beam, whereas the phase delay Value = π (negative phase)
It can be seen that the beam is formed so as to be captured by the next adjacent beam. On the other hand, in the case of FIG. 16 in which the incident directions of the two waves are apart (when the arrival direction of the multipath is 30 °), the beams to be captured are shifted by one in the in-phase incidence and the opposite-phase incidence, but they are also limited. It can be seen that the beam is formed in a direction that can be effectively captured within the range of the degree of freedom of the antenna. In other words, directional diversity has been realized that combines direct waves and multipath waves with directivity corresponding to their power.

FIG. 17 is a simulation result of the bit error rate (BER) by the maximum ratio combining reception under the same conditions as in FIG. However, the symbol delay for the direct multipath wave was assumed to be negligible.
Bit error rate when one multipath wave arrives (BE
R) is about 1.5 dB compared to the case where only direct waves arrive
It can be seen that the value is improved and approaches the value theoretically expected by the maximum ratio combining (improvement of about 1.8 dB).

Next, simulation results of transmission beam formation will be described. FIGS. 18 and 19 show an example in which a transmission beam is formed when two waves of a direct wave and a multipath wave arrive using the apparatus of the present embodiment. Where 2
FIG. 18 shows the case where the arrival directions of the direct wave and the multipath wave are −45 ° and + 15 °, respectively, and FIG. 19 shows the case where the arrival directions of the direct wave and the multipath wave are changed. The case where the arrival directions of the path waves are −15 ° and + 30 ° respectively is shown. The array antenna 1 is shared for transmission and reception, and the transmission frequency is 1.
066 times. In each case, the transmitting main beam is formed only in the direction of the direct wave, is not affected by the multipath wave, and the radiation in the direction of the multipath wave is suppressed to about the side lobe level at most. You can see that.

As described above, this embodiment according to the present invention has the following unique effects. (1) Unlike the first conventional example, a special direction sensor and position information of the partner station are not required, so that there is no influence from surrounding magnetic disturbance or accumulation of a bearing detection error, and the partner station moves. In this case, the transmission beam can be automatically formed in the direction of the incoming wave transmitted from the partner station, and the size and cost can be reduced. (2) Instead of converting the reception phase difference at the reception antenna to the transmission phase difference as it is as in the second conventional example,
Based on the least-squares method, the phase uncertainty is removed and the influence of multiple waves other than the maximum received wave is removed.
Regardless of the direction in which the maximum received wave arrives in a multiplexed wave environment, a transmission beam can be formed in the direction in which the maximum received wave arrives reliably. Can be reduced. (3) As seen from the apparatus of the embodiment, in forming a transmission beam, it is possible to adopt a configuration in which there is no mechanically driven portion of the antenna and no feedback loop, so that the reception baseband signal is If it is obtained, the transmission weight can be immediately determined, and a transmission beam can be formed at high speed in real time. (4) Further, as seen in the apparatus of the embodiment, since the transmission weight can be determined by digital signal processing, if transmission beam formation is also performed by digital processing, baseband processing including modulation is performed. All of them can be integrated into a digital signal processor, and if a device with a high degree of integration is used, the entire system can be reduced in size and cost.

<Second Embodiment> FIG. 3 is a block diagram showing a transmission section of an automatic beam acquisition and tracking device for a communication array antenna according to a second embodiment. For other parts,
The configuration is the same as that of the first embodiment. Hereinafter, differences from the first embodiment of FIG. 2 will be described in detail.

The transmission local oscillator 10a is, for example, a DDS (Direct Digital Synthesize) driven by the same clock.
An oscillator using r), which generates a transmission local oscillation signal of a predetermined frequency. On the other hand, the transmission baseband signal S ^TX which is transmission data is input to the in-phase distributor 9 and distributed in-phase to N transmission baseband signals S ^TX , respectively.
The signals are input to the phase correction units 13-1 to 13-N. Phase correcting unit 13-i (i = 1,2, ..., N) is transmitted to the transmission baseband signal S ^TX input weight W ₁ ^{TX,
_{W 2 TX, ..., W N}} TX of the multiplication to the multiplication result transmission baseband signal ^{_{S i TX (i = 1,2,}} ..., N) a quadrature modulator 6a
Output to -i. The quadrature modulators 6a-i perform serial / parallel conversion of the input transmission baseband signal to convert it to a transmission quadrature baseband signal, and then, according to the transmission quadrature baseband signal, have a transmission phase difference of 90 ° with respect to each other. The intermediate frequency signal is obtained by orthogonally modulating and combining the signals. Then, the intermediate frequency signal after the quadrature modulation is input to the circulator CI-i in the array antenna 1 as a transmission radio signal via the up-converter 7 and the transmission power amplifier 8 in the transmission module TM-i. Then, the transmission radio signal is transmitted to the antenna element A
Transmitted and radiated from i. Therefore, the transmission signals weighted by the transmission weights W ₁ ^TX , W ₂ ^TX ,..., W _N ^TX are radiated from the antenna elements A1 to AN. Therefore, the transmitting section in the second embodiment operates in the same manner as that of the first embodiment and has the same effect.

[0102]

As described above in detail, according to the method for controlling an array antenna according to the first aspect of the present invention, an array antenna composed of a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape is controlled. In a method of controlling an array antenna for performing, using a common local oscillation signal each of a plurality of reception signals received by each antenna element of the array antenna, an in-phase component that is in phase with the local oscillation signal, A quadrature signal composed of a local oscillation signal and a quadrature component orthogonal to each other, is converted into a quadrature signal. And the first data calculated by the sum of the products of the quadrature components and the in-phase component of the quadrature signal of the first antenna element and the second data
Product of the orthogonal components of the orthogonal signals of the antenna elements
Pass the quadrature component of the quadrature signal of the second antenna element and the second data calculated by the difference between the product of the in-phase component of the quadrature signal of the second antenna element to a noise suppression filter having a predetermined transfer function. After filtering, the filtered second data is divided by the filtered second data, and the arc tangent of the result of the division is calculated, thereby receiving the first antenna element. A reception phase difference between the signal and the reception signal of the second antenna element is calculated, and the calculated reception phase difference between each of the two first and second antenna elements is in a range from −π to + π. Using a matrix consisting of the reception phase differences corresponding to all the candidates for the phase uncertainty that occurs due to being limited to the matrix and a matrix consisting of the position coordinates of a plurality of antenna elements constituting the array antenna, a minimum of 2
Wiener-Hop using multiplication
f) solving an equation to derive an equation representing a first-order regression plane of the equiphase corresponding to all the candidates of the phase uncertainty, and obtaining a plurality of equiphases corresponding to all the candidates of the phase uncertainty; Is calculated using the sum of squares of the residuals between the reception phase difference and the equiphase primary regression plane, and the gradient coefficient of the equiphase primary regression plane.
The phase uncertainty is removed, the reception phase difference is corrected by specifying only one primary regression plane having the same phase corresponding to the maximum reception wave, and the sign of the corrected reception phase difference is inverted. By transmitting the transmission signal from each of the antenna elements with the transmission phase difference between the two first and second antenna elements and with the same amplitude, the maximum reception wave is obtained. The transmission main beam is formed only in the direction. Therefore, the present invention has the following specific effects. (1) Unlike the first conventional example, a special direction sensor and position information of the partner station are not required, so that there is no influence from surrounding magnetic disturbance or accumulation of a bearing detection error, and the partner station moves. In this case, the transmission beam can be automatically formed in the direction of the incoming wave transmitted from the partner station, and the size and cost can be reduced. (2) Instead of converting the reception phase difference at the reception antenna to the transmission phase difference as it is as in the second conventional example,
Based on the least-squares method, the phase uncertainty is removed and the influence of multiple waves other than the maximum received wave is removed.
Regardless of the direction in which the maximum received wave arrives in a multiplexed wave environment, a transmission beam can be formed in the direction in which the maximum received wave arrives reliably. Can be reduced. (3) In forming a transmission beam, it is possible to adopt a configuration in which there is no mechanically driven part of the antenna and no feedback loop, so that if a reception baseband signal is obtained, the transmission weight is immediately determined. The transmission beam can be formed at high speed in real time. (4) Since the transmission weight is determined by digital signal processing, if transmission beam formation is also performed digitally, all baseband processing including modulation can be integrated in a digital signal processor, and the degree of integration can be reduced. The use of expensive devices can reduce the size and cost of the entire system.

[0103]

[0104]

Further, according to the array antenna control method of the second aspect, in the array antenna control method of the first aspect, a calculation is performed from the first-order linear regression plane of the same phase from which the phase uncertainty has been removed. The transmission phase difference is converted to a transmission phase difference by multiplying the reception phase difference by the ratio of the transmission frequency to the reception frequency. Thereby, the reception phase difference can be converted into the transmission phase difference by a simpler method.

According to the third aspect of the present invention, there is provided an array antenna control device for controlling an array antenna including a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape. A plurality of reception signals respectively received by the antenna elements of the array antenna, using a common local oscillation signal, an in-phase component having the same phase as the local oscillation signal, and a quadrature component orthogonal to the local oscillation signal. And first and second conversion means for respectively converting the quadrature signals into quadrature signals, and the product and quadrature of the in-phase components of the quadrature signals in each of the two first and second antenna elements which are different pairs among the plurality of antenna elements. First data calculated by the sum of products of the components, the in-phase component of the quadrature signal of the first antenna element, and the second data
Product of the orthogonal components of the orthogonal signals of the antenna elements
Pass the quadrature component of the quadrature signal of the second antenna element and the second data calculated by the difference between the product of the in-phase component of the quadrature signal of the second antenna element to a noise suppression filter having a predetermined transfer function. After filtering, the filtered second data is divided by the filtered second data, and the arc tangent of the result of the division is calculated, thereby receiving the first antenna element. First calculating means for calculating a reception phase difference between a signal and a received signal of the second antenna element, and a first calculating means for calculating a reception phase difference between the two first and second antenna elements calculated by the first calculating means; And a matrix consisting of reception phase differences corresponding to all candidates for phase uncertainty that occurs because the reception phase difference is limited to the range from -π to + π, and a plurality of antenna elements constituting the array antenna. Consists of position coordinates By solving the Wiener-Hopf equation using the least squares method with the matrix, the equation representing the equiphase first-order regression plane corresponding to all the candidates for the phase uncertainty is derived. A second calculating means for calculating a plurality of equal-phase primary regression planes corresponding to all candidates for the phase uncertainty; The phase uncertainty is removed using the sum of squares of the residual and the gradient coefficient of the first-order regression plane of the same phase, and only one first-order regression plane of the same phase corresponding to the maximum received wave is obtained. A correcting means for correcting the received phase difference by specifying the signal; and a second converting means for converting the received phase difference corrected by the correcting means into a transmission phase difference by inverting the sign of the received phase difference. Each converted by the conversion means One first and by a transmission signal in and equal amplitude transmission phase difference between the second antenna element for transmitting from each antenna element to form a transmission main beam only in the direction of maximum received wave. Therefore,
The present invention has the following specific effects. (1) Unlike the first conventional example, a special direction sensor and position information of the partner station are not required, so that there is no influence from surrounding magnetic disturbance or accumulation of a bearing detection error, and the partner station moves. In this case, the transmission beam can be automatically formed in the direction of the incoming wave transmitted from the partner station, and the size and cost can be reduced. (2) Instead of converting the reception phase difference at the reception antenna to the transmission phase difference as it is as in the second conventional example,
Based on the least-squares method, the phase uncertainty is removed and the influence of multiple waves other than the maximum received wave is removed.
Regardless of the direction in which the maximum received wave arrives in a multiplexed wave environment, a transmission beam can be formed in the direction in which the maximum received wave arrives reliably. Can be reduced. (3) In forming a transmission beam, it is possible to adopt a configuration in which there is no mechanically driven part of the antenna and no feedback loop, so that if a reception baseband signal is obtained, the transmission weight is immediately determined. The transmission beam can be formed at high speed in real time. (4) Since the transmission weight is determined by digital signal processing, if transmission beam formation is also performed digitally, all baseband processing including modulation can be integrated in a digital signal processor, and the degree of integration can be reduced. The use of expensive devices can reduce the size and cost of the entire system.

[0107]

[0108]

According to the array antenna control device of the fourth aspect, in the array antenna control device of the third aspect, the second conversion means may be configured so that the second phase conversion means eliminates the phase uncertainty. By multiplying the reception phase difference calculated from the first-order regression plane by the ratio of the transmission frequency to the reception frequency, the transmission phase difference is converted to the transmission phase difference. Thereby, the reception phase difference can be converted into the transmission phase difference by a simpler method.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating a receiving unit of an automatic beam acquisition and tracking device for a communication array antenna according to a first embodiment of the present invention.

FIG. 2 is a block diagram illustrating a transmitting unit of the automatic beam acquisition and tracking device of the communication array antenna according to the first embodiment.

FIG. 3 is a block diagram illustrating a transmitting unit of an automatic beam acquisition and tracking device for a communication array antenna according to a second embodiment.

FIG. 4 is a block diagram showing a DBF unit 4 of FIG. 1;

FIG. 5 is a plan view showing an arrangement of antenna elements in the example.

FIG. 6 is a block diagram of a transmission weight calculation circuit 30 of FIG. 1;

FIG. 7 is a flowchart showing a phase regression plane selection process executed by the phase regression plane selection unit 33 in FIG. 6 when the antenna elements are in a one-dimensional array (a modification).

8 is a flowchart showing a first part of a phase regression plane selection process performed by the phase regression plane selection unit 33 in FIG. 6 when the antenna elements are two-dimensionally arranged (embodiment).

FIG. 9 is a flowchart illustrating a second part of the phase regression plane selection process performed by the phase regression plane selection unit 33 in FIG. 6 when the antenna elements are in a two-dimensional array (embodiment).

FIG. 10 is a flowchart showing a third part of the phase regression plane selection processing performed by the phase regression plane selection unit 33 in FIG. 6 when the antenna elements are two-dimensionally arranged (embodiment).

FIG. 11 is an explanatory diagram showing a regression process to a primary plane by the least squares method of a reception phase in the transmission weight calculation circuit 30 of FIG. 6;

FIG. 12 is an explanatory diagram showing inspection and removal of phase uncertainty in the transmission weight operation circuit 30 of FIG. 6;

FIG. 13 is an explanatory diagram showing setting of a phase threshold value k in a check of uncertainty of a reception phase in the transmission weight calculation circuit 30 of FIG. 6;

FIG. 14 is a simulation result of the automatic beam capture and tracking device of the communication array antenna of FIGS. 1 and 2;
9 is a graph showing a directivity pattern indicating beam forming by maximum ratio combining reception.

FIG. 15 is a simulation result of the automatic beam acquisition and tracking device of the communication array antenna of FIGS. 1 and 2;
It is a graph which shows the directivity pattern when the angle of the arrival direction of a multipath wave is 15 degrees.

FIG. 16 is a simulation result of the automatic beam acquisition and tracking device of the communication array antenna of FIGS. 1 and 2;
It is a graph which shows the directivity pattern when the angle of the arrival direction of a multipath wave is 30 degrees.

FIG. 17 is a simulation result of the automatic beam capture and tracking device of the communication array antenna of FIGS. 1 and 2;
5 is a graph showing bit error rate characteristics in maximum ratio combining reception.

FIG. 18 is a simulation result of the automatic beam acquisition and tracking device of the communication array antenna of FIGS. 1 and 2;
The angles of arrival of the direct wave and the multipath wave are each -4.
It is a graph of the directivity pattern which shows formation of a transmission / reception beam in the case of 5 degrees and +15 degrees.

FIG. 19 is a simulation result of the automatic beam acquisition and tracking device of the communication array antenna of FIGS. 1 and 2;
The angles of arrival of the direct wave and the multipath wave are each -1.
It is a graph of the directivity pattern which shows formation of a transmission / reception beam in the case of 5 degrees and +30 degrees.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 ... Array antenna, 2 ... Low noise amplifier, 3 ... Down converter, 4 ... DBF part, 5 ... Demodulator, 6, 6a-1 to 6a-N ... Quadrature modulator, 7 ... Up converter, 8 ... Transmission power amplifier, 9: In-phase distributor, 10, 10a: transmitting local oscillator, 11: first local oscillator, 12: second local oscillator, 13-1 to 13-N: phase correcting unit, 20: absolute value calculating unit, 21: complex Conjugate product operation unit, 22, 23 low-pass filter, 24, 25 delay buffer circuit, 26, 27 multiplier, 28a, 28b divider, 29 in-phase synthesizer, 30 transmission weight operation circuit, 31 -1 to 31-N: phase difference calculation unit, 32-1 to 32-9: least square regression processing unit, 33: phase regression plane selection unit, 34: selector, 35: transmission weight calculation unit, A1 to AN: Ante Elements, CI-1 to CI-N: circulator, RM-1 to RM-N: receiving module, AD-1 to AD-N: A / D converter, QD-1 to QD-N: quasi-synchronous detection circuit, QM-1 to QM-N: quadrature modulation circuits; TM-1 to TM-N: transmission modules.

Continuation of the front page (72) Inventor Yoshio Karasawa Kyoto, Soraku-gun, Seika-cho, 5F, Inani, 5F, Sanpira, ATR Optical Co., Ltd. Inside the Radio Communication Research Laboratory (56) References JP-A-4-273601 (JP) JP-A-3-85927 (JP, A) JP-A-1-161180 (JP, A) JP-A-3-220804 (JP, A) JP-A-61-237502 (JP, A) 4-252525 (JP, A) JP-A-6-90193 (JP, A) JP-A-7-50627 (JP, A) IEICE technical report (Technical Report of IEICE, Vol. 93 No. 290), RCS93 −66, “Feedforward In-phase Synthetic Digital Beamforming Method for Mobile Satellite Communications for High-Speed Beam Acquisition and Tracking”, Ryu Miura et al., Pp31-37, pp31-37, Oct. 22, 1993 IEICE Technical Research Report (IEICE Technical Report, Vol. 94, No. 311), RCS94-85, "Mobile Devices Using ASICs" “Digital Beamforming Antenna for Communication”, Toyohisa Tanaka et al., Pp. 69-74, published October 27, 1994 IEICE Technical Report (IEICE Technical Report Vol. 94 No. 501), RCS 94-141, “ “Beam Space CMA Adaptive Array Antenna Using ASIC”, Toyohisa Tanaka et al., Pp31-36, published February 27, 1995. (58) Fields investigated (Int. Cl. ⁶ , DB name) H01Q 3 / 00-3/46 H01Q 21/00-21/30 H01Q 23/00 H01Q 25/00-25/04 JICST file (JOIS) WPI / L (QUESTEL)

Claims (4)

(57) [Claims] An array antenna control method for controlling an array antenna including a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape, comprising: a plurality of antenna elements each of which is received by each antenna element of the array antenna. Each of the received signals is converted into an in-phase component having the same phase as the local oscillation signal and a quadrature signal including an orthogonal component orthogonal to the local oscillation signal, using a common local oscillation signal. First data calculated by the sum of the product of the in-phase components of the quadrature signals and the product of the quadrature components of the two orthogonal signals in each of the two first and second antenna elements that are different from each other; And the quadrature component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the first antenna element and the second component of the quadrature signal of the second antenna element. The second data calculated by the difference between the products of the in-phase components of the quadrature signals of the antenna elements are respectively passed through a noise suppression filter having a predetermined transfer function to be filtered. Divides the filtered second data by the following data and calculates the arc tangent of the result of the division to obtain the difference between the received signal of the first antenna element and the received signal of the second antenna element. Is calculated, and the phase uncertainty that occurs because the calculated reception phase difference between each of the two first and second antenna elements is limited to the range from -π to + π And a matrix consisting of position coordinates of a plurality of antenna elements constituting the array antenna, using a matrix consisting of reception phase differences corresponding to the candidates of
By solving the (r-Hopf) equation, an equation representing the first-order regression plane of the same phase corresponding to all the candidates of the phase uncertainty is derived. Are calculated using the sum of squares of the residuals between the reception phase difference and the equiphase primary regression plane, and the gradient coefficient of the equiphase primary regression plane. Then, the phase uncertainty is removed, the reception phase difference is corrected by specifying only one primary regression plane having the same phase corresponding to the maximum reception wave, and the sign of the corrected reception phase difference is changed. By inverting the signal, the signal is converted into a transmission phase difference, and the transmission signal is transmitted from each of the antenna elements at the same transmission phase difference and equal amplitude between the two first and second antenna elements. Form the transmit main beam only in the direction of the received wave Control method for an array antenna, characterized in that. 2. A transmission phase difference obtained by multiplying a reception phase difference calculated from the first-order regression plane of the same phase from which the phase uncertainty is removed by a ratio of a transmission frequency to a reception frequency. 2. The method according to claim 1, wherein the conversion is performed to a transmission phase difference. 3. An array antenna control device for controlling an array antenna composed of a plurality of antenna elements arranged in close proximity in a predetermined arrangement shape, comprising: a plurality of antenna elements respectively received by each of the antenna elements of the array antenna. A first conversion unit that converts the received signal into a quadrature signal composed of an in-phase component having the same phase as the local oscillation signal and a quadrature component orthogonal to the local oscillation signal, using a common local oscillation signal, First data calculated by the sum of the product of the in-phase components of the quadrature signals and the product of the quadrature components of each of the two first and second antenna elements that are different pairs from each other among the plurality of antenna elements; The product of the in-phase component of the quadrature signal of the first antenna element and the quadrature component of the quadrature signal of the second antenna element and the quadrature signal of the first antenna element After passing the cross data and the second data calculated by the difference between the products of the in-phase components of the quadrature signals of the second antenna elements, the data are passed through a noise suppression filter having a predetermined transfer function and filtered. By dividing the filtered second data by the filtered first data and calculating the arctangent of the result of the division, the reception signal of the first antenna element and the second antenna A first calculating means for calculating a reception phase difference between the reception signal of the element and a reception phase difference between each of the two first and second antenna elements calculated by the first calculation means; A matrix consisting of the reception phase differences corresponding to all the candidates for the phase uncertainty generated due to being limited to the range from to + π, and a matrix consisting of the position coordinates of a plurality of antenna elements constituting the array antenna. Using the least squares method Is used to solve the Wiener-Hopf equation to derive an equation representing the equiphase first-order regression plane corresponding to all the candidates for the phase uncertainty. Multiple equiphase 1s corresponding to the candidate
Second calculating means for calculating a next regression plane; a sum of squares of residuals between the reception phase difference and the same-phase first-order regression plane; a gradient coefficient of the same-phase first-order regression plane; A correcting means for correcting the received phase difference by removing the phase uncertainty and specifying only one first-order regression plane having the same phase corresponding to the maximum received wave; and correcting by the correcting means. A second converting means for inverting the sign of the received reception phase difference to convert the reception phase difference into a transmission phase difference, wherein each of the two first and second antenna elements converted by the second conversion means is provided. A control device for an array antenna, wherein a transmission main beam is formed only in a direction of a maximum received wave by transmitting a transmission signal from each of the antenna elements with a transmission phase difference and an equal amplitude. 4. The second conversion means multiplies a reception phase difference calculated from the equiphase primary regression plane from which the phase uncertainty has been removed by a ratio of a transmission frequency to a reception frequency. 4. The array antenna control device according to claim 3, wherein the transmission phase difference is converted into a transmission phase difference.