JP2700746B2 - Frequency hopping communication system - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION The present invention is used for digital radio communication. The present invention is used for spread spectrum communication. The present invention is suitable for use in mobile radio communication. According to the present invention, one symbol of a symbol sequence of an input modulated signal is modulated by K (K is an integer of 2 or more) time-varying carrier frequencies and transmitted as a K-chip modulated signal. It relates to improvement of a hopping communication system. A "chip" is one of the signals divided into K and modulated at one frequency. In particular, the present invention relates to a frequency hopping communication system that performs coherent detection.

[0002]

2. Description of the Related Art In mobile radio communication, there is a fading fluctuation of a transmission line, and it is necessary to take measures against interference of an interfering wave.
Spread spectrum communication is considered to be a very effective system for this purpose. Spread-spectrum communication has a large frequency occupied bandwidth, but spread-spectrum communication can improve communication quality. You can use it.

[0003] Spread spectrum communication systems can be broadly classified into direct sequence (DS) systems and frequency hopping (FH) systems. Among them, the frequency hopping method further includes information 1
A high-speed frequency hopping (FFH, Fast FH) system that performs frequency hopping at least once with a symbol, and a low-speed frequency hopping (SFH, Sl) that performs frequency hopping for each burst signal formed based on a signal of several symbols or more.
ow FH). In the frequency hopping method, a frequency diversity effect is obtained every time the frequency is changed, so that a method with a low bit error rate can be realized. In particular, in the high-speed frequency hopping method, a very high-quality transmission path can be realized by the frequency diversity effect for each symbol.

FIG. 11 is a block diagram of a conventional transmitting apparatus. A terminal I is supplied with an input modulation signal of a symbol sequence. The signal is modulated by the mixer MIX with the carrier frequency generated from the frequency synthesizer LT controlled by the frequency control circuit FCONT. This synthesizer LT generates K different frequencies in a predetermined order during one symbol period of the input modulation signal. The output of the mixer MIX passes through the bandpass filter BPF and is transmitted from the antenna. The transmitted modulated signal is a signal in which one symbol is composed of K chips. Assuming that K = 4, the carrier frequency changes to f ₁ , f ₂ , f ₃ , and f ₄ in one symbol period.

FIG. 12 is a block diagram of a conventional receiver. This example shows a receiving apparatus when K = 4. The signal of the transmitting apparatus received by the antenna is divided and supplied to _four mixers MIX _{1 to} MIX ₄ . These four mixers MIX _{1 to} MIX ₄ have local frequencies of f ₁ + f _IF , f ₂ + f _IF , f ₃ + f _IF , f ₄ + f, respectively.
_IF supplied. Where f _IF is the intermediate frequency.
The output of the mixer MIX passes through a band-pass filter BPF that passes an intermediate frequency, and is square-detected by a correlation detection circuit COR and a squaring circuit to extract a signal level corresponding to each chip. Take the level sum and use it as the detection output.

In such a conventional apparatus, the K different carrier frequencies generated by the synthesizer LT of the transmitting apparatus are not controlled so that their phases become continuous at the time of hopping. Even if the carrier frequency is controlled so as to be continuous, the local frequency for demodulation in the receiving apparatus is frequency-synchronized with the received carrier frequency but not in phase. That is, the above-described conventional device is non-coherent. This is because, in an actual mobile radio communication system, since the fading fluctuation period of the transmission path is generally short, it is difficult to synchronize the phase of the received carrier frequency.Conventionally, the local frequency of the receiving device is set to the carrier frequency of the transmitting device. Used without phase synchronization.

[0007]

In a synthesizer LT of a transmission apparatus for generating a transmission carrier frequency, it is easy to realize control of K different carrier frequencies so that the phase is continuous at the time of hopping. it can. And, although it is conceivable to perform coherent detection in which the local frequency of the receiving apparatus is phase-synchronized with the carrier frequency of the transmitting side obtained from the received signal and performs detection, it is quite difficult, and it is realized. A specific circuit for performing coherent detection has not been studied. In general, it is theoretically clear that the coherent detection has better transmission characteristics than the non-coherent detection. For example, in BPSK (Binary Phase Shift Keying), non-coherent detection achieves the same error rate as coherent detection because the carrier has the same error rate. Noise to noise ratio (CNR,
It is known that Carrier to Noise Ratio theoretically degrades by about 6 dB.

The present invention has been made in such a background, and an object of the present invention is to provide a frequency hopping method capable of performing coherent detection in a feasible method. An object of the present invention is to improve the quality of a wireless transmission path and to effectively use frequency resources by effectively utilizing the advantages of the spread spectrum communication system and the frequency hopping system.

[0009]

SUMMARY OF THE INVENTION According to the present invention, there is provided a transmitting apparatus in which a carrier frequency phase of a signal composed of K chips to be transmitted is continuous at the time of a frequency change of each chip, and a receiving apparatus includes K chips. , A complex envelope corresponding to each chip demodulated from the received signal composed of the symbols is synthesized, and symbol judgment is performed from the synthesized output. By synthesizing the complex envelope, even if the carrier frequency received at the receiving device is not phase-locked, this is to perform coherent detection, and the local frequency of the receiving device is phase-locked to the carrier frequency of the transmitting device. It is equivalent to that.

That is, according to the present invention, K symbols (K is an integer of 2 or more) in a symbol sequence of an input modulated signal are used.
A transmitting device provided with a modulation circuit for modulating the carrier frequency with the time-varying carrier frequency, receiving a transmission signal from the transmitting device, and synchronizing with the K time-varying carrier frequencies to produce a time-series And a receiving device having a demodulation circuit for demodulating with a local frequency that changes to the carrier frequency. The carrier frequency and the local frequency have a continuous phase at the time when the frequency changes, and the demodulation circuit has a K corresponding to one symbol. Synthesis circuit (C
SUM) and a determination circuit (DEC) for determining a symbol from the output of the synthesis circuit.

It is preferable that the synthesizing circuit (CSUM) includes means for calculating a synthetic output by a least square method.

The K complex envelopes are added to the synthesis circuit (C
Multiplication circuits (CM _{1 to} CM _K ) for multiplying complex coefficients (w _{1 to} w _K ) before input of the SUM), and a difference circuit (CSUB) for calculating a difference between an input and an output of the determination circuit (DEC) And an arithmetic and control circuit (CCONT) that generates the complex coefficient corresponding to the K complex envelopes so as to reduce the difference.

[0013] The arithmetic control circuit (CCONT) may be configured to include means for setting an initial value of the complex coefficient when a training signal of a preset signal pattern is transmitted from the transmitting device.

[0014]

According to the frequency hopping communication system of the present invention, the carrier frequency of the transmitting device and the local frequency of the receiving device are configured such that their phases become continuous at the time of the frequency hopping change. Then, the demodulation circuit of the receiving device detects the complex envelope of each chip and performs symbol determination from the combined output obtained by combining the complex envelopes, so that symbol determination taking into account the phase is performed. That is, coherent detection is performed.

It is convenient for this synthesizing circuit to synthesize the complex envelope of each chip into one complex envelope using the least squares method. It is preferable that each complex envelope is multiplied by a complex coefficient (corresponding to a weight coefficient) before the synthesizing circuit, and the complex coefficient is adaptively changed according to the received signal.
By adaptively changing the complex coefficient according to the received signal, coherent detection can be continued even if fading or interference occurs. When adaptively changing this complex coefficient, its initial value is to transmit a training signal of a preset signal pattern from the transmitting device,
It is convenient to determine from the determination error when receiving the training signal.

[0016]

FIG. 1 is a block diagram of a transmitting apparatus according to a first embodiment of the present invention. FIG. 2 is a block diagram of the receiving apparatus according to the first embodiment of the present invention. Referring to FIG. 1, terminal I is provided with a symbol sequence of an input modulation signal. The frequency synthesizer LT generates K time-varying carrier frequencies in one symbol period of the input modulation signal under the control of the frequency control circuit FCONT-T. The input signal modulated by the mixer MIX is modulated by the carrier frequency, and a transmission signal of K chips is generated in one symbol period. This signal passes through the band pass filter BPFT and is transmitted from the antenna.

This signal received by the antenna in FIG.
Is converted to an intermediate frequency f _IF is mixed with a local frequency by the mixer MIXR. The local frequency is controlled by the frequency control circuit FCONT-R, and generates K time-varying frequencies in one symbol period in synchronization with the carrier frequency of the transmitting device. When the K time-varying carrier frequencies generated by the frequency synthesizer of the transmitting device are f ₁ , f ₂ ,..., F _K , the K time sequences generated by the frequency synthesizer of the receiving device The local frequencies which fluctuate are f ₁ + f _IF , f ₂ + f _IF ,..., F _K + f _IF . Hereinafter, in this example, K = 4. Intermediate frequency f _IF by the band-pass filter BPFR from the output of the mixer MIXR is selected, is detected by the detection circuit DET.

Here, the frequency synthesizer L of the transmitting device
The feature of the present invention resides in that the phase of each frequency hopping is preserved or continuous in T and the frequency synthesizer LR of the receiving apparatus. This can be easily realized by known frequency synthesizer technology.
For example, a ROM (Read Only Memory) is provided in the frequency synthesizer, and the ROM is read by a clock signal,
A circuit for converting the read output into an analog signal by a digital / analog conversion circuit is used.
This can be realized by storing data in the OM in advance so that the phase of the output frequency is continuous at the time of the phase change.

Further, a feature of the present invention resides in a detection circuit DET. FIG. 3 is a block diagram of the detection circuit DET of the first embodiment of the present invention. FIG. 3 is a specific block diagram of the detection circuit DET of FIG. The terminal I _D 2
The output signal I _D of band-pass filter BPFR of inputs.
This signal _ID is quasi-synchronously detected by a detector IQD to extract a complex envelope. This complex envelope is the switch SW
Is divided into K chips by K (in this example, K =
4) Distributed and stored in the memory CMEM. This switch SW is controlled in synchronization with the change of the local frequency by a switch control signal SWC generated from the frequency control circuit FCONT-R of FIG.

The complex envelope r stored in the memory CMEM
₁ , r ₂ , r ₃ , and r ₄ are complex coefficients w ₁ , w ₂ , w ₃ , w
₄ is multiplied by the complex multipliers CM1 to CM4, and the product signal is synthesized by the synthesizing circuit CSUM. Complex coefficients w ₁ , w
₂ , w ₃ and w ₄ correspond to weight coefficients. This combining circuit C
A function for calculating the least squares method is set in the SUM, and the composite output is calculated using the least squares method. That is, one complex envelope is calculated from the four complex envelopes by the least square method. Obtaining a demodulated output d _i a synthesized output y _i at the output of the combining circuit CSUM determination to the determination circuit DEC.

Here, the complex coefficients w ₁ , w ₂ , w ₃ , w
₄ , the value is adaptively controlled by the received signal. That is, the difference e _i between the input and the output of the decision circuit DEC is taken by the subtractor CSUB, and the operation control circuit CCON is set so that the difference e _i is minimized, ideally zero.
It is calculated by T. The fact that the difference e _i is zero means that there is no decision error of the decision circuit DEC. By automatically controlling the complex coefficients w ₁ , w ₂ , w ₃ , w ₄ so that there is no decision error, the transmission line The best reception state can be maintained adaptively with respect to the state change.

The complex coefficients w ₁ , w ₂ , w ₃ , w ₄ are values which fluctuate adaptively with respect to the received signal during the operation of the receiving apparatus, but at the time of starting or resetting the receiving apparatus. The initial value of the complex coefficient is set by the training signal. The training signal may be any pattern signal as long as the signal has a preset pattern. The simplest is a repetition signal of "1" and "0", but generally a more complicated PN signal is used so that synchronization can be clearly confirmed. The pattern of the training signal is set in advance in each of the transmitting device and the receiving device. Then, the training signal is transmitted from the transmitting device, and the receiving device receives the complex coefficient w so that the receiving of the training signal can be correctly determined.
_{1, w 2, w 3,} w 4 it may set its initial value by controlling the.

[0023] Figure 4 the horizontal axis represents frequency, the received high frequency f _R in the receiving apparatus, the reception intermediate frequency f _IF, the transmission signal bandwidth W _R of this scheme, a diagram showing the intermediate frequency bandwidth W _IF relationship is there. The bandwidth of the K chips spread signal To and the signal bandwidth W _S when no diffusion, K × W
Becomes _S. Here, a 4W _S because it is K = 4. Here signals are hopping are evenly arranged in the transmission signal bandwidth W _R. When converting this to an intermediate frequency, the local frequency, as this signal are all at an intermediate frequency f _IF, a frequency synthesizer LR is controlled.

This operation will be described in more detail using mathematical expressions.

Symbol sequence b input from input terminal I
(t) is a sine wave c output from the frequency synthesizer LT
Modulated by mixer MIXT using _t (t). Frequency of the sine wave c _t (t) is switched in response to a control signal output from the frequency control circuit FCONT-T. Modulated wave S
(t) is as follows.

S (t) = b (t) _ct (t) (1) where b (t) = b _i , iT ≦ t <(i + 1) T (2) Further, assuming that the hopping frequency for the k-th (1 ≦ k ≦ K) chip is ω _k and the initial phase is φ _k , c _t (t) is as follows.

[0027]

(Equation 1) The value of the hopping frequency ω _k changes for each chip in a predetermined order. The diversity effect is obtained by distributing the value over the entire band w _R of the system and preventing the same frequency from appearing within one symbol. The phase of each hopping frequency is stored, and this phase control is also performed by the frequency control circuit FCONT-T.

FIG. 5 is a diagram for explaining the above modulation method in the case where the number of chips K = 4. 1 in every cycle T input symbol sequence b _i was changed into 1,1.
These symbols are divided into signals of 4 chips per symbol. Next, modulation is performed by allocating to each chip a carrier frequency of hopping frequency and phase of f ₁ and φ ₁ , f ₂ and φ ₂ , f ₃ and φ ₃ , and f ₄ and φ _{4 respectively} . By repeating this operation for each symbol, a frequency-hopping transmission wave is generated.

Next, the specific operation of the demodulation circuit DET will be described using equations. The input signal r (t) can be expressed by the following equation.

[0030] r (t) = s (t ) A k c r * (t) = A k b (t) eXp [-jθ (t)] (4) where

[0031]

(Equation 2) It is. c ^* _r (t) represents the complex conjugate of _cr (t). Where A
_k represents a complex envelope fluctuation due to fading of the transmission path, and is divided into an amplitude and a phase component as in the following equation.

A _k = | A _k | exp [jArg (A _k )] (6) The input signal r (t) is, as described with reference to FIG.
Are weighted synthesized with complex coefficients w _i-1 obtained at the time of the previous (i-1) using a M1~CM4 and combining circuit CSUM. The composite signal y _i is as follows.

[0033]

(Equation 3) However

[0034]

(Equation 4) It is. Determining the combined signal y _i, and outputs the determination result d _i. Using these, the difference e _i is calculated by the following equation.

E _i = d _i −y _i (9) The difference e _i is input to the control circuit CCONT, and the complex constant w _i is calculated based on the least squares method. Ideally, each component w _k of the complex constant w _i is equal to its complex contributor A _k ^* for the complex envelope component A _k . Using the complex constant w _i , y _{i + 1} is obtained from the composite signal. Symbols are demodulated by repeating the above operation.

FIG. 5 is a diagram for explaining the state of frequency hopping with time on the horizontal axis. That is, the symbol sequence b
This shows how the transmission frequency changes in order from f ₁ to f ₄ when i changes between +1 and −1 at one symbol period T.

Here, when the carrier frequency is instantaneously switched as shown in FIG. 5, the phase and frequency changes are abrupt, so that the spectrum of the transmission signal contains many side lobes as shown in FIG. This means that a wide bandwidth is required as a transmission frequency band. An apparatus for improving this as shown in FIG. 7 is a second embodiment of the present invention described below.

FIG. 8 is a block diagram of the transmitting apparatus of the apparatus according to the second embodiment of the present invention. In the device of the second embodiment, the receiving device is equivalent to the device of the first embodiment described above. A signal equivalent to the input modulation signal I of the first embodiment described with reference to FIG. 1 is input to the input modulation signal I of FIG. This input modulation signal I
Are branched into K (K = 4) branches (k = 1 to 4) by the switch circuit SWT. The signals are shaped by passing through the low-pass filters LPFT1 to LPFT4, respectively, and converted into respective chip frequencies by the mixing circuits MIX1 to MIX4. The outputs of the mixing circuits MIX1 to MIX4 are respectively bandpass filters BPFT of chip frequencies.
1 to BPFT 4 to extract necessary high frequency components. This is synthesized by the synthesis circuit COM.

In the transmitting apparatus of FIG. 8, the frequency synthesizers LT1 to LT4 for generating carrier frequencies each generate a carrier frequency corresponding to a chip, which is controlled by one control circuit (not shown). ,
Control is performed so that the phase at the frequency transition point of the chip that has passed through the synthesizing circuit COM is continuous.

The transmission apparatus is constructed as described above, and the low-pass filters LPFT1 to LPFT4 and the band-pass filters BP
By properly setting FT1 to BPFT4, the spurious frequency can be removed, and the waveform of the transmission signal can be made an ideal waveform as shown in FIG.

FIG. 9 is a block diagram of a transmitting apparatus of the third embodiment of the present invention. The feature of the third embodiment is that the modulation signal I (corresponding to the input modulation signal I of the first embodiment described with reference to FIG. 1) is already a phase-multiplexed signal. That is, the four input signals b _{i1 to} b _i4 are respectively orthogonal to four kinds of orthogonal functions h ₁ , h ₂ ,
h ₃ , h ₄ (In general, h _m (t), where m =
,..., 4) are multiplied and added by an adder circuit MUX to be multiplexed and input to a terminal I. And
The signal at the terminal I is modulated by frequency hopping equivalent to that of the first embodiment described with reference to FIG. 1 and transmitted.

FIG. 10 is a block diagram of a main part of a receiving apparatus of the third embodiment of the present invention. This pin I _D from the terminal I / Q demodulation circuit of the first embodiment apparatus described in FIG. 3
FIG. 2 is a partial block configuration diagram corresponding to FIG. That is, in the receiving apparatus of the third embodiment, the terminals from the terminal _ID to the terminal I / Q of the demodulation circuit of the first embodiment shown in FIG.
It is realized by substituting as follows. The orthogonal functions for separating phase multiplexing are h ₁ ′, h ₂ ′, h ₃ ′, and h ₄ ′ are orthogonal functions h ₁ , h ₂ , h ₃ , h _{4 of the} transmitter shown in FIG.
Is set corresponding to. When this apparatus is used for the mobile communication system, the transmitting apparatus of the base station apparatus performs phase multiplexing on a plurality of channels (here, four channels) as shown in FIG. It is sufficient to demodulate only the channel, and FIG. 10 shows only one channel h ₁ ′.
Is displayed. The configuration after the terminal I / Q is the same as that of the first embodiment described with reference to FIG.

In the third embodiment, the frequency utilization efficiency is improved by the amount of phase multiplexing. The operation of the third embodiment will be described using mathematical expressions. That is, in the third embodiment, multiplexing is performed by multiplying four symbol sequences by four types of orthogonal functions h _m (t), where m = 1,..., 4. However, h _m (t) is the time of the chip _{k h m (t) = h} m, k (10), the orthonormal satisfy the condition of following equation.

[0044]

(Equation 5) Examples of the orthogonal signal h _m (t) include a Walsh function and a frequency offset signal. Also, when receiving such a multiplexed signal, it is necessary to multiply the same function as the signal multiplied at the time of transmission by the receiver.

FIG. 10 shows an example of a configuration for receiving the first signal multiplexed on the transmission side, where h ₁ '(t) is multiplied after the demodulation circuit IQD. At this time, when the influence of fading is small and the level of each chip is substantially the same, the value of the complex envelope variation A _k becomes substantially constant. Therefore, the complex coefficient w _k is calculated by the least squares method with its complex conjugate value A _k ^* . Become equal. Therefore, each chip is synthesized with a constant weight, and a signal of m = 1 is extracted by the equation (11-a).
Further, the signal of m ≠ 1 becomes 0 according to the equation (11-b). When the effect of fading is large and there is a level difference between chips, a larger weight is given to a chip having a higher level by the least squares method. In this case, since the levels of the respective chips to be combined are not equal, the orthogonal condition is not satisfied, and the signal of m ≠ 1 cannot be completely canceled. However, an operation of weighting a chip with a large level with a larger weight is almost equivalent to the maximum ratio combining in diversity reception, so that a diversity effect can be expected.

As described above, multiplexing is possible in coherent combining, and it is possible to avoid a decrease in frequency use efficiency due to signal band expansion accompanying frequency hopping.
This property is due to coherent detection and is a feature not found in non-coherent detection.

[0047]

As described above, according to the present invention, a frequency hopping system having excellent transmission characteristics can be obtained by coherent detection. According to the method of the present invention, the carrier to noise ratio (CNR, Carrier to Noise) for obtaining the same error rate is different from that of the conventional non-coherent detection.
It is apparent that the input modulation signal can be phase-multiplexed, and the frequency utilization efficiency can be improved.

[Brief description of the drawings]

FIG. 1 is a block diagram of a transmission apparatus of a first embodiment of the present invention.

FIG. 2 is a block diagram of a receiving apparatus of the first embodiment of the present invention.

FIG. 3 is a block diagram of a demodulation circuit (DET) provided in the receiver of the first embodiment of the present invention.

FIG. 4 is a view for explaining a signal spectrum of the device according to the embodiment of the present invention.

FIG. 5 is a view for explaining signal waveforms of the device according to the embodiment of the present invention.

FIG. 6 is a view for explaining a signal spectrum of the apparatus according to the embodiment of the present invention.

FIG. 7 is a view for explaining a signal spectrum of the apparatus according to the embodiment of the present invention.

FIG. 8 is a block diagram of a transmission device of the device according to the second embodiment of the present invention.

FIG. 9 is a block diagram of a transmission device of the device according to the third embodiment of the present invention.

FIG. 10 is a block diagram of a main part of a receiving apparatus of a third embodiment of the present invention.

FIG. 11 is a block diagram of a conventional apparatus transmission apparatus.

FIG. 12 is a block diagram of a conventional apparatus receiving apparatus.

Claims (5)

(57) [Claims] 1. A symbol sequence of an input modulated signal,
A transmitting device having a modulation circuit for modulating a symbol with K (K is an integer of 2 or more) time-varying carrier frequency; and a K-time-sequential signal receiving a transmission signal from the transmitting device. And a receiving device having a demodulation circuit that demodulates with a local frequency that changes in a time series in synchronization with a carrier frequency that changes in the frequency hopping communication system, wherein the carrier frequency and the local frequency are changed at the time when the frequency changes. The demodulation circuit includes a synthesis circuit (CSUM) that receives K complex envelopes corresponding to one symbol as inputs, and a determination circuit (DEC) that determines a symbol from an output of the synthesis circuit. A frequency hopping communication method comprising: 2. The frequency hopping communication system according to claim 1, wherein said combining circuit (CSUM) includes means for calculating an output by a least square method. 3. The complex coefficients (w ₁ to w ₁ ) are input to the K complex envelopes before input to the synthesis circuit (CSUM).
a multiplier circuit for multiplying the _{w K) (CM 1 ~CM K} ), wherein a difference circuit for calculating an input and output difference of the decision circuit (DEC) (CSUB), the complex coefficient as the difference becomes smaller 3. The frequency hopping communication system according to claim 1, further comprising: an operation control circuit (CCONT) generated corresponding to the K complex envelopes. 4. The arithmetic control circuit (CCONT) according to claim 3, further comprising means for setting an initial value of said complex coefficient when a training signal of a preset signal pattern is transmitted from said transmitting device. Frequency hopping communication method. 5. The frequency hopping communication system according to claim 1, wherein said input modulated signal is a signal obtained by phase-multiplexing a plurality of modulated signals.