JP2024060638A - Power Supplies - Google Patents

Abstract

[Problem] To provide a power supply device that can realize zero-volt switching without reducing the maximum output power, even when the current flowing through the primary winding is small. [Solution] The power supply device includes an inverter circuit 2 having switching elements TR1 to TR4, and a control circuit 5. Switching elements TR1 and TR3 are connected in series, switching elements TR2 and TR4 are connected in series, and switching elements TR1 and TR2 are connected to the same polarity side. The control circuit 5 generates drive signals P1 and P4, a drive signal P3 that is shifted by a half period from the drive signal P1, and a drive signal P2 that is shifted by a half period from the drive signal P4, and performs output current control by adjusting the phase difference between the drive signals P1 and P4, and increases the oscillation frequency of the drive signals P1 and P4 as the current through the inverter circuit 2 decreases. The drive signals P1 to P4 are input to the switching elements TR1 to TR4, respectively. [Selected Figure] FIG.

Description

The present invention relates to a power supply device.

There is a known power supply device in which an inverter circuit converts a DC voltage input from a DC power supply into a high-frequency voltage, a transformer transforms the high-frequency voltage, and a rectifier circuit rectifies and outputs the voltage. Phase shift control is also known as a control method for a single-phase full-bridge inverter circuit. Patent Document 1 discloses a power supply device controlled by phase shift control. In phase shift control, a phase difference is provided during the on-period between a high-side switching element of one arm of the inverter circuit and a low-side switching element of the other arm, and the output is adjusted according to the phase difference.

Figure 2 is a diagram for explaining phase shift control. In period A shown in Figure 2, the switching elements TR1 and TR4 are on, and the current supplied by the DC power supply 1 flows to the primary winding 31 of the transformer 3 through the switching elements TR1 and TR4. In period B, the switching element TR1 is off, but the current continues to flow to the primary winding 31 through the freewheel diode of the switching element TR3 and the switching element TR4 due to the energy stored in the leakage inductance of the primary winding 31 of the transformer 3. In period C, the switching element TR3 is on. When the switching element TR3 is switched on, a current flows through the freewheel diode, so the voltage between the drain and source of the switching element TR3 is "0". Therefore, zero-volt switching is performed. In period D, the switching element TR4 is off, but the current continues to flow to the primary winding 31 through the freewheel diode of the switching element TR2, the smoothing capacitor of the DC power supply 1, and the freewheel diode of the switching element TR3. In period E, switching element TR2 is on, and a current flows in the opposite direction to that in period A through the primary winding 31 via switching elements TR2 and TR3. When switching element TR2 is switched on, if a current is flowing through the freewheeling diode, the drain-source voltage of switching element TR2 is "0". This results in zero-volt switching. From then on, the high and low sides of periods B, C, and D are reversed, and the period returns to period A. In this way, with phase shift control, zero-volt switching can be achieved when each switching element TR1 to TR4 is switched on.

JP 2004-322189 A

However, when the leakage inductance of the primary winding 31 of the transformer 3 is small and the current flowing through the primary winding 31 is small, the energy stored in the leakage inductance of the primary winding 31 is small. If the energy stored in the leakage inductance is consumed between the time when the switching element TR1 is turned off and the time when the switching element TR3 is turned on, no current flows through the freewheel diode of the switching element TR3, and zero-volt switching of the switching element TR3 cannot be realized. If a coil is connected in series to the primary winding 31 of the transformer 3 to increase the leakage inductance, zero-volt switching can be realized. However, when the leakage inductance increases, the energy stored in the leakage inductance increases, but the energy transmitted to the secondary side decreases. In other words, increasing the leakage inductance reduces the maximum output power of the power supply device.

The present invention was conceived in light of the above circumstances, and its purpose is to provide a power supply device that can achieve zero volt switching without reducing the maximum output power, even when the current flowing through the primary winding is small.

To solve the above problems, the present invention provides the following technical solutions:

In order to solve the above-mentioned problems, the present invention provides:
an inverter circuit having a first arm and a second arm, each of which has two switching elements connected in series;
a control circuit that generates a first drive signal input to a first switching element of the first arm, a second drive signal input to a second switching element connected in the second arm to the same polarity side as the first switching element, a third drive signal input to a third switching element of the first arm, and a fourth drive signal input to a fourth switching element connected in the second arm to the same polarity side as the third switching element;
Equipped with
The control circuit includes:
generating a signal obtained by shifting the first drive signal by half a period as the third drive signal;
generating a signal obtained by shifting the fourth drive signal by half a period as the second drive signal;
performing output current control by adjusting a phase difference between a rising timing of a pulse of the first drive signal and a rising timing of a pulse of the fourth drive signal;
The oscillation frequency of the first drive signal and the fourth drive signal is set to be higher as the current setting value of the inverter circuit is smaller.
It is a power supply.

The invention of claim 2 is as follows:
a current sensor for detecting a current in the inverter circuit, the control circuit being configured to set the oscillation frequency to a higher value as the detection value of the current sensor becomes smaller;
2. A power supply device according to claim 1.

The invention of claim 3 is as follows:
The current sensor is provided in the inverter circuit unit.
The power supply device according to claim 2.

According to the present invention, the dead time td is shortened by increasing the oscillation frequency F as the output current of the power supply device is smaller. Therefore, when the output current of the power supply device is small and the energy stored in the leakage inductance of the primary winding of the transformer 3 is small, the dead time td (see period B in FIG. 2) can be shortened. As a result, the power supply device according to the present invention prevents the energy stored in the leakage inductance of the transformer 3 from being consumed during the dead time td, thereby realizing zero-volt switching. In addition, since the leakage inductance of the primary winding 31 of the transformer 3 is not increased, the power supply device according to the present invention does not reduce the maximum output power.

1 is a diagram showing an overall configuration of a welding power supply according to a first embodiment; 5A to 5C are waveform diagrams showing an example of each drive signal input to each switching element. FIG. 13 is a diagram showing the overall configuration of a welding power supply according to a second embodiment.

The following describes in detail an embodiment of the present invention with reference to the drawings, taking as an example a case in which the power supply device according to the present invention is used as a welding power supply device.

First Embodiment
FIG. 1 is a diagram for explaining a welding power supply according to a first embodiment, showing the overall configuration of the welding power supply.

The welding power supply supplies power to the workpiece 7 to perform welding. As shown in FIG. 1, the welding power supply includes a DC power supply 1, an inverter circuit 2, a transformer 3, a rectifier circuit 4, a control circuit 5, a drive circuit 6, and a current sensor 8.

The DC power supply 1 outputs a DC voltage, and includes, for example, a rectifier circuit that rectifies the AC voltage input from the power grid, and a smoothing capacitor that smoothes the AC voltage. The configuration of the DC power supply 1 is not limited, and it may be any configuration that outputs a DC voltage to the inverter circuit 2.

The inverter circuit 2 converts the DC voltage input from the DC power supply 1 into a high-frequency voltage and outputs it to the transformer 3. The inverter circuit 2 is a single-phase full-bridge inverter, and includes four switching elements TR1 to TR4. In this embodiment, MOSFETs (Metal Oxide Semiconductor Field Effect Transistors) are used as the switching elements TR1 to TR4. Note that the switching elements TR1 to TR4 are not limited to MOSFETs, and may be bipolar transistors, IGBTs (Insulated Gate Bipolar Transistors), etc.

The switching element TR1 and the switching element TR3 are connected in series with the source terminal of the switching element TR1 and the drain terminal of the switching element TR3 connected. The drain terminal of the switching element TR1 is connected to the positive side of the DC power supply 1, and the source terminal of the switching element TR3 is connected to the negative side of the DC power supply 1 to form a bridge structure. The switching element TR2 and the switching element TR4 are connected in series with the source terminal of the switching element TR2 and the drain terminal of the switching element TR4 connected. The drain terminal of the switching element TR2 is connected to the positive side of the DC power supply 1, and the source terminal of the switching element TR4 is connected to the negative side of the DC power supply 1 to form a bridge structure. The bridge structure formed by the switching element TR1 and the switching element TR3 is the first arm 21, and the bridge structure formed by the switching element TR2 and the switching element TR4 is the second arm 22. An output line C1 is connected to the connection point between switching element TR1 and switching element TR3 of the first arm 21, and an output line C2 is connected to the connection point between switching element TR2 and switching element TR4 of the second arm 22.

A free wheel diode is connected in anti-parallel to each of the switching elements TR1 to TR4. A snubber capacitor is connected between the drain terminal and the source terminal of each of the switching elements TR1 to TR4. Drive signals P1 to P4 (described later) output from the drive circuit 6 are input to the gate terminal of each of the switching elements TR1 to TR4. Each of the switching elements TR1 to TR4 is switched on and off based on the drive signals P1 to P4, respectively. This converts the DC voltage into a high-frequency voltage. The configuration of the inverter circuit 2 is not limited.

The transformer 3 transforms the high-frequency voltage output by the inverter circuit 2 and outputs it to the rectifier circuit 4. The transformer 3 has a primary winding 31 and a secondary winding 32. One input terminal of the primary winding 31 is connected to the output line C1, and the other input terminal is connected to the output line C2. One output terminal of the secondary winding 32 is connected to one input terminal of the rectifier circuit 4, and the other output terminal is connected to the other input terminal of the rectifier circuit 4. The secondary winding 32 is provided with a center tap in addition to the two output terminals. The primary winding 31 and the secondary winding 32 are each wound around a core (not shown) and can be magnetically coupled to each other. The configuration of the transformer 3 is not limited.

The rectifier circuit 4 is a full-wave rectifier circuit using the center tap of the transformer 3, and rectifies the high-frequency current output by the transformer 3 and outputs it as a direct current. The rectifier circuit 4 includes two rectifier diodes 41 and 42 and a direct current reactor 43. The rectifier diodes 41 and 42 have their anode terminals connected to the output terminals of the secondary winding 32 of the transformer 3, respectively, and their cathode terminals connected to each other. The direct current reactor 43 is connected in series between the connection point on the cathode terminal side of the rectifier diodes 41 and 42 and the output terminal a of the welding power supply device, and stabilizes the output current. The center tap of the transformer 3 is connected to the output terminal b of the welding power supply device. The direct current output by the rectifier circuit 4 flows to the welding workpiece 7 as a welding current. The configuration of the rectifier circuit 4 is not limited.

The current sensor 8 is disposed on the connection line between the center tap of the transformer 3 and the output terminal b, and detects the output current of the welding power supply device and outputs the result as a welding current signal Io to the control circuit 5. The location of the current sensor 8 is not limited as long as it can detect the output current of the welding power supply device.

The control circuit 5 is configured to control the inverter circuit 2, and is realized by, for example, a microcomputer. The control circuit 5 generates drive signals P1 to P4 for controlling the inverter circuit 2 and outputs them to the drive circuit 6. The control circuit 5 performs output current control, and feedback controls the output current of the welding power supply based on the welding current signal Io input from the current sensor 8. The control circuit 5 also performs control similar to phase shift control. Specifically, the control circuit 5 controls the output current by adjusting the phase difference of the on-period of the switching elements TR1 and TR4 (switching elements TR3 and TR2) of the inverter circuit 2, similar to the phase shift control. The control circuit 5 increases the phase difference to shorten the period during which the switching elements TR1 and TR4 (switching elements TR3 and TR2) are simultaneously on, thereby reducing the output current of the welding power supply compared to when there is no phase difference (when the phase difference is "0"). In this embodiment, the control circuit 5 changes the oscillation frequency F of the drive signals P1 to P4 according to the current set value Iset, which is the target value of the output current. The control circuit 5 has, as its functional configuration, a current setting unit 51, a subtraction unit 52, a phase difference setting unit 53, a period setting unit 55, and a drive signal generation unit 54.

The current setting unit 51 sets the current setting value Iset, which is the target value of the output current of the welding power supply. The current setting unit 51 outputs the set current setting value Iset to the subtraction unit 52 and the period setting unit 55. The current setting unit 51 sets the current setting value Iset based on the welding conditions input by the operator or pre-programmed. The period setting unit 55 increases the oscillation frequency F based on the current setting value Iset, the greater the current setting value Iset. For example, when the current setting value Iset is the minimum current value, the frequency is 120 kHz, and when the current setting value Iset is the maximum current value, the frequency is 80 kHz. The oscillation frequency F may be changed continuously or stepwise according to the current setting value Iset.

The subtraction unit 52 calculates the deviation ΔI (=Iset-Io) between the welding current signal Io input from the current sensor 8 and the current set value Iset input from the current setting unit 51, and outputs it to the phase difference setting unit 53. Based on the input deviation ΔI, the phase difference setting unit 53 calculates the phase difference tθ for making the deviation ΔI "0", and outputs it to the drive signal generation unit 54. Note that the method of calculating the phase difference tθ is not limited. The phase difference tθ becomes smaller as the deviation ΔI becomes larger, and becomes larger as the deviation ΔI becomes smaller.

The drive signal generating unit 54 generates drive signals P1 to P4 before one cycle begins based on the oscillation frequency F input from the cycle setting unit 55 and the phase difference tθ input from the phase difference setting unit 53, and outputs them to the drive circuit 6. The drive signals P1 to P4 are amplified by the drive circuit 6 and input to the switching elements TR1 to TR4, respectively.

Figure 2 is a waveform diagram showing an example of each drive signal P1 to P4. The drive signals P1 to P4 are pulse signals having a predetermined pulse width Ton at a predetermined period T. The period T is determined by the oscillation frequency F of the period setting unit 55 at T = 1/F. The pulse width Ton is fixed at a time ratio of about 40% of the period T and is determined by Ton = 1/F x 40%. The phases of the drive signals P1 and P3 are shifted by half a period (T/2). In addition, the dead time td provided between the pulse of the drive signal P1 and the pulse of the drive signal P3 is also fixed at a time ratio of about 10% of the period T and is determined by td = 1/F x 10%. In Figure 2, the leftmost pulse of each drive signal P1 to P4 shows the case where the phase difference tθ = 45°. The second pulse from the left of each drive signal P1 to P4 shows the case where the phase difference tθ = 90°. The third pulse from the left of each drive signal P1 to P4 shows the case where the phase difference tθ = 135°.

On the other hand, the drive signal P2 is a waveform shifted by a phase difference tθ from the drive signal P1, and the drive signal P4 is a waveform shifted by a phase difference tθ from the drive signal P3. When the inverter does not output, the drive signal P2 and the drive signal P4 are waveforms shifted by a phase difference tθ = 180° from the drive signal P1 and the drive signal P3, and the Tons of the drive signal P1 and the drive signal P4 do not overlap, and the Tons of the drive signal P3 and the drive signal P2 do not overlap, so the inverter is in a state of not outputting. When the inverter outputs, it changes according to the phase difference tθ input from the phase difference setting unit 53. When the output of the inverter is to be increased, the phase difference tθ is adjusted so that the phase difference becomes smaller, such as 180° → 135° → 90° → 45°, and the overlap time of the Tons of the drive signal P1 and the drive signal P4 and the drive signal P3 and the drive signal P2 can be increased, thereby increasing the output. Conversely, to reduce the output of the inverter, the phase difference tθ can be adjusted to increase from 45° to 90° to 135° to 180°, thereby reducing the overlap time of Ton between drive signals P1 and P4, and between drive signals P3 and P2, and thereby reducing the output.

Here, the cycle setting unit 55 is configured to increase the oscillation frequency F as the current setting value Iset decreases based on the current setting value Iset. Therefore, it can be seen that the lower the current setting value Iset, the shorter the dead time td becomes. For example, assume that the dead time td is set to 10% of the oscillation frequency F and the pulse width Ton is set to 40% of the oscillation frequency F. If the oscillation frequency F = 120 kHz when the current setting value Iset is the minimum current value, the dead time td = at the minimum current value is (1/120 kHz) x 0.1 = 0.83 μS. On the other hand, if the oscillation frequency F = 80 kHz when the current setting value Iset is the maximum current value, the dead time td at the maximum current value is (1/80 kHz) x 0.1 = 1.25 μS. As described above, the smaller the current setting value Iset, the higher the oscillation frequency F, the shorter the dead time td can be, and the switching element can be turned on with current flowing through the freewheel diode, achieving zero-volt switching. Note that the oscillation frequency F is set so that the dead time td is within the time during which no energy is consumed in response to the current setting value Iset.

In this embodiment, before one cycle begins, the drive signal generating unit 54 generates drive signals P1 to P4 for each cycle based on the oscillation frequency F and phase difference tθ. The overlapping times of drive signals P1 and P4 and drive signals P3 and P2 during one cycle are the same, and the current-carrying time of the positive side and the current-carrying time of the negative side of the primary winding 31 of the transformer 3 are equal, which also has the effect of suppressing biased magnetism of the transformer 3.

Second Embodiment
3 is a block diagram for explaining a welding power supply according to a second embodiment of the present disclosure. The welding power supply according to this embodiment differs from the welding power supply according to the first embodiment in that a drive signal generating unit 54 adjusts an oscillation frequency F based on an inverter current signal Ip input from a current sensor 9. The configuration and operation of other parts of this embodiment are similar to those of the first embodiment.

In this embodiment, the period setting unit 55 sets the oscillation frequency F based on the inverter current signal Ip flowing through the primary winding 31 of the transformer 3, input from the current sensor 9, instead of the current set value Iset. The current sensor 9 is provided in the inverter circuit unit 2 to detect the current flowing through the primary winding 31 of the transformer 3. In this embodiment, the period setting unit 55 operates so that the smaller the inverter current signal Ip, the higher the oscillation frequency F. The oscillation frequency F may be changed continuously or in stages according to the value of the inverter current signal Ip.

In this embodiment, the current flowing through the primary winding 31 of the transformer 3 is detected by the current sensor 9 provided in the inverter circuit unit 2, and the inverter current signal Ip detected by the current sensor 9 also includes the excitation current flowing through the primary winding 31 of the transformer 3. The inverter current signal Ip and the time during which the energy stored in the leakage inductance of the transformer 3 is consumed are approximately proportional to the square of the inverter current signal Ip. Therefore, it is more preferable to adjust the oscillation frequency F using the inverter current signal Ip than to adjust the oscillation frequency F using the current set value Iset in the first embodiment. In addition, the oscillation frequency F may be adjusted using the welding current signal Io of the current sensor 8, but the welding current signal Io does not include the excitation current flowing through the primary winding 31 of the transformer 3, and when the transformer 3 is in a biased magnetization state, it is difficult to estimate the time during which the energy stored in the leakage inductance is consumed, so it is preferable to detect the inverter current signal Ip using the current sensor 9 provided in the inverter circuit unit 2 and determine the oscillation frequency F.

In the above first and second embodiments, the present invention has been described as being applied to a welding power supply, but this is not limited to this. The present invention is applicable to all power supplies that convert DC voltage to high-frequency voltage using an inverter circuit.

The power supply device according to the present invention is not limited to the above-described embodiment. The specific configuration of each part of the power supply device according to the present invention can be freely designed in various ways.

1: DC power supply 2: Inverter circuit 21: First arm 22: Second arm 3: Transformer 31: Primary winding 32: Secondary winding 4: Rectifier circuit 5: Control circuit 6: Drive circuit 7: Welding workpiece 8: Current sensor 9: Current sensors 41, 42: Rectifier diode 43: DC reactor 51: Current setting section 52: Subtraction section 53: Phase difference setting section 54: Drive signal generation section 55: Period setting section a, b: Output terminals C1, C2: Output line F: Oscillation frequency Iset: Current set value Io: Welding current signal Ip: Inverter current signals P1 to P4: Drive signals TR1 to TR4: Switching element tθ: Phase difference td: Dead time T: Period Ton: Pulse width ΔI: Deviation

Claims (3)

an inverter circuit having a first arm and a second arm, each of which has two switching elements connected in series;
a control circuit that generates a first drive signal input to a first switching element of the first arm, a second drive signal input to a second switching element connected in the second arm to the same polarity side as the first switching element, a third drive signal input to a third switching element of the first arm, and a fourth drive signal input to a fourth switching element connected in the second arm to the same polarity side as the third switching element;
Equipped with
The control circuit includes:
generating a signal obtained by shifting the first drive signal by half a period as the third drive signal;
generating a signal obtained by shifting the fourth drive signal by half a period as the second drive signal;
performing output current control by adjusting a phase difference between a rising timing of a pulse of the first drive signal and a rising timing of a pulse of the fourth drive signal;
The oscillation frequency of the first drive signal and the fourth drive signal is set to be higher as the current setting value of the inverter circuit is smaller.
Power supply. a current sensor for detecting a current in the inverter circuit, the control circuit being configured to set the oscillation frequency to a higher value as the detection value of the current sensor becomes smaller;
2. The power supply device of claim 1. The current sensor is provided in the inverter circuit.
3. The power supply device of claim 2.

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