JP2023115267A - Radar apparatus - Google Patents

Abstract

To inhibit unwanted a grating lobe from generating and achieve a desired directivity pattern even for a sub-array antenna configuration.SOLUTION: A radar apparatus comprises a transmission array antenna including a plurality of transmission antennas, and a receiving array antenna including a plurality of receiving antennas. The plurality of transmission antennas are arranged in different positions in a first direction and the plurality of receiving antennas are arranged in different positions in the first direction. A gap between two adjacent transmission antennas of the plurality of transmission antennas is one wavelength or more in the first direction, and a gap between two adjacent receiving antennas of the plurality of receiving antennas is one wavelength or more in the first direction. An absolute value of a difference between the gap of the two adjacent transmission antennas and the gap of the two adjacent receiving antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction. At least one of the plurality of transmission antennas and the plurality of receiving antennas includes a plurality of antenna elements arranged in the first direction.SELECTED DRAWING: Figure 6

Description

The present disclosure relates to radar equipment.

2. Description of the Related Art In recent years, studies have been made on radar devices using short-wavelength radar transmission signals including microwaves or millimeter waves that can provide high resolution. Further, in order to improve outdoor safety, there is a demand for the development of a radar device that detects objects (targets) including pedestrians in a wide-angle range in addition to vehicles.

For example, as a radar device, a pulse radar device that repeatedly transmits pulse waves is known. The signal received by the wide-angle pulse radar that detects vehicles/pedestrians in a wide-angle range is a mixture of multiple reflected waves from a target (e.g. vehicle) at a short distance and a target (e.g. pedestrian) at a long distance. signal. Therefore, (1) the radar transmission unit is required to have a configuration for transmitting a pulse wave or a pulse-modulated wave having autocorrelation characteristics (hereinafter referred to as low-range sidelobe characteristics) with low range sidelobes, and (2) A radar receiver is required to have a configuration with a wide reception dynamic range.

As the configuration of the wide-angle radar device, there are the following two configurations.

The first is to transmit a radar wave by mechanically or electronically scanning a pulsed or modulated wave with a narrow-angle (beam width of several degrees) directional beam, and generate a narrow-angle directional beam. is used to receive the reflected wave. In this configuration, since many scans are required to obtain high resolution, the followability to a target moving at high speed is degraded.

The second method is to receive reflected waves by an array antenna consisting of multiple antennas (antenna elements) and estimate the arrival angle of the reflected waves using a signal processing algorithm based on the reception phase difference with respect to the antenna spacing (Direction of Arrival). (DOA) estimation). In this configuration, even if the scanning interval of the transmission beam in the radar transmission section is thinned out, the arrival angle can be estimated in the radar reception section. improves. For example, direction-of-arrival estimation methods include Fourier transform based on matrix operation, Capon method and LP (Linear Prediction) method based on inverse matrix operation, or MUSIC (Multiple Signal Classification) and ESPRIT (Estimation of Signal Parameters) based on eigenvalue operation. via Rotational Invariance Techniques).

Further, as a radar device, in addition to the radar receiving unit, the radar transmitting unit also has a plurality of antennas (array antennas), and has a configuration in which beam scanning is performed by signal processing using the transmitting/receiving array antenna (sometimes called MIMO radar). has been proposed (see, for example, Non-Patent Document 2).

Japanese Patent Publication No. 2011-526370

Budisin, S.Z., "New complementary pairs of sequences," Electron. Lett., 1990, 26, (13), pp.881-883 Jian Li, Stoica, Petre, "MIMO Radar with Colocated Antennas," Signal Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007

By the way, in order to increase the directivity gain of an array antenna, a sub-array antenna may be used in which each of the antenna elements (hereinafter referred to as array elements) constituting the array antenna is further composed of a plurality of antenna elements.

It is difficult to arrange the elements of the array antenna at intervals narrower than the size of the array elements. However, when using a subarray antenna configuration, the size of the array elements increases, so it is necessary to widen the intervals between the subarray antennas, and grating lobes may occur on the directivity pattern of the array antenna.

One aspect of the present disclosure provides a radar apparatus capable of suppressing generation of unnecessary grating lobes and realizing a desired directivity pattern even in the case of a subarray antenna configuration.

A radar apparatus according to an aspect of the present disclosure includes a transmission array antenna, a reception array antenna, a radar transmission unit that transmits a radar signal using the transmission array antenna, and a reflected wave signal obtained by reflecting the radar signal from a target. using the receiving array antenna, the transmitting array antenna including a plurality of transmitting antennas, the receiving array antenna including a plurality of receiving antennas, and the plurality of transmitting antennas are arranged at different positions in the first direction, the plurality of receiving antennas are arranged at different positions in the first direction, and the distance between two adjacent transmitting antennas among the plurality of transmitting antennas is the 1 wavelength or more in a first direction, the distance between two adjacent receiving antennas among the plurality of receiving antennas is 1 wavelength or more in the first direction, and the two adjacent transmission antennas The absolute value of the difference between the antenna spacing and the spacing between the two adjacent receiving antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction, and the plurality of transmitting antennas and the plurality of includes a plurality of antenna elements arranged in the first direction.
A radar device according to another aspect of the present disclosure transmits a first array antenna, a second array antenna, and a radar signal using one of the first array antenna and the second array antenna. a radar transmission unit; and a radar reception unit that receives a reflected wave signal of the radar signal reflected by a target using the other of the first array antenna and the second array antenna, comprises a plurality of first antennas, each of said plurality of first antennas being arranged at different positions in a first direction, said second array antenna comprising a plurality of second antennas wherein each of the plurality of second antennas is arranged at a different position in the first direction, and the distance between two adjacent first antennas among the plurality of first antennas is the first the distance between two adjacent second antennas among the plurality of second antennas is one wavelength or more in the first direction, and the distance between the two adjacent second antennas is The absolute value of the difference between the spacing between the first antenna and the spacing between the two adjacent second antennas includes 0.5 wavelength or more and 0.75 wavelength or less in the first direction, At least one of the plurality of first antennas and the plurality of second antennas includes a plurality of antenna elements arranged in the first direction.

These generic or specific aspects may be realized by systems, methods, integrated circuits, computer programs, or recording media, and any of the systems, devices, methods, integrated circuits, computer programs and recording media may be implemented in any combination.

According to one aspect of the present disclosure, it is possible to suppress generation of unnecessary grating lobes and realize a desired directivity pattern even in the case of a subarray antenna configuration.

Further advantages and advantages of one aspect of the present disclosure are apparent from the specification and drawings. Such advantages and/or advantages are provided by the several embodiments and features described in the specification and drawings, respectively, not necessarily all provided to obtain one or more of the same features. no.

Diagram showing a configuration example of a subarray element A diagram showing a configuration example of an array antenna composed of subarray elements 1 is a block diagram showing the configuration of a radar device according to an embodiment of the present disclosure; FIG. A diagram showing an example of a radar transmission signal according to an embodiment of the present disclosure Block diagram showing another configuration of the radar transmission signal generator according to an embodiment of the present disclosure A diagram showing an example of transmission timing and measurement range of a radar transmission signal according to an embodiment of the present disclosure FIG. 4 is a diagram showing antenna arrangements of a transmission array, a reception array, and a virtual reception array according to an embodiment of the present disclosure; A diagram showing a directivity pattern according to an embodiment of the present disclosure FIG. 10 is a diagram showing antenna arrangements of a transmission array, a reception array, and a virtual reception array according to variation 1 of an embodiment of the present disclosure; FIG. 10 is a diagram showing a horizontal directivity pattern according to variation 1 of an embodiment of the present disclosure; A diagram showing a vertical directivity pattern according to Variation 1 of an embodiment of the present disclosure FIG. 10 is a diagram showing antenna arrangements of a transmission array, a reception array, and a virtual reception array according to Variation 2 of an embodiment of the present disclosure; FIG. 11 shows a horizontal directivity pattern according to Variation 2 of an embodiment of the present disclosure; A diagram showing a vertical directivity pattern according to Variation 2 of an embodiment of the present disclosure FIG. 10 is a diagram showing antenna arrangements of a transmission array, a reception array, and a virtual reception array according to Variation 3 of an embodiment of the present disclosure; FIG. 10 is a diagram showing a horizontal directivity pattern according to Variation 3 of an embodiment of the present disclosure; FIG. 11 is a diagram showing an example of a diagram showing a vertical directivity pattern according to Variation 3 of an embodiment of the present disclosure;

[Circumstances leading to one aspect of the present disclosure]
FIG. 1A shows an example of an antenna element having a subarray configuration (hereinafter sometimes referred to as a subarray element). The sub-array element shown in FIG. 1A is composed of 2×2=4 antenna elements. Also, in the example shown in FIG. 1A, the size of the subarray elements is 0.8 wavelengths in both the horizontal and vertical directions.

FIG. 1B shows an example of an array antenna configured by arranging four sub-array elements shown in FIG. 1A in series. As shown in FIG. 1B, since the size of each subarray element is 0.8 wavelength (see FIG. 1A), the interval between subarray elements must be approximately one wavelength or longer.

For example, the array element spacing (desired element spacing) for not generating a grating lobe within ±90° of the main lobe is 0.5 wavelength. In the array antenna shown in FIG. 1B, since the element spacing of the subarray elements is about one wavelength or more, it is difficult to set the desired element spacing, and grating lobes occur within the range of ±90° of the main lobe. become.

Thus, when the size of the subarray elements is 0.5 wavelength or more, it may be difficult to set the element interval of the array antenna to 0.5 wavelength. Therefore, an unnecessary grating lobe is generated within ±90° of the main lobe, and a virtual image is generated during angle measurement, which causes erroneous detection.

Here, Patent Document 1 discloses an array antenna configuration using sub-array elements having a width d of approximately one wavelength. In Patent Document 1, the element spacing of the transmitting antennas Tx0 and Tx1 is six wavelengths, and the element spacing of the receiving antennas RX0, RX1, RX2, and RX3 is 1.5 wavelengths ±(λ/8) (λ represents one wavelength. ). Further, in Patent Document 1, a radar transmission signal is transmitted by switching transmission antennas Tx0 and Tx1 in a time division manner, and the reception antennas RX0, RX1, RX2, It has a configuration that acquires the received signal with RX3.

With such a configuration, a phase change due to a change in the position of the transmitting antenna is superimposed on the received signal acquired by the receiving array antenna, so the effect of virtually increasing the aperture length of the receiving antenna can be obtained. Hereinafter, a virtual receiving array antenna whose effective aperture length is increased by arranging antenna elements in the transmitting/receiving array antenna will be referred to as a "virtual receiving array".

However, in Patent Document 1, since the element spacing of the receiving array antenna is 1.5 wavelengths ±λ/8, a grating lobe is generated in a direction deviated from the main beam direction by about 40°.

One aspect of the present disclosure suppresses generation of unnecessary grating lobes and realizes a desired directivity pattern even when array elements having a sub-array configuration are used.

Hereinafter, embodiments according to one aspect of the present disclosure will be described in detail with reference to the drawings. In addition, in the embodiments, the same constituent elements are denoted by the same reference numerals, and redundant description thereof will be omitted.

[Configuration of radar device]
FIG. 2 is a block diagram showing the configuration of the radar device 10 according to this embodiment.

The radar device 10 has a radar transmitter 100 , a radar receiver 200 and a reference signal generator 300 .

The radar transmitter 100 generates a high-frequency radar signal (radar transmission signal) based on the reference signal received from the reference signal generator 300 . Radar transmission section 100 transmits a radar transmission signal at a predetermined transmission cycle using a transmission array antenna composed of a plurality of transmission antennas 106-1 to 106-Nt.

The radar receiver 200 receives a reflected wave signal, which is a radar transmission signal reflected by a target (not shown), using a reception array antenna composed of a plurality of reception antennas 202-1 to 202-Na. The radar receiver 200 uses the reference signal received from the reference signal generator 300 to perform signal processing on the reflected wave signal received by each antenna 202 to detect the presence or absence of a target, estimate its direction, and the like. A target is an object to be detected by the radar device 10, and includes, for example, a vehicle or a person.

The reference signal generator 300 is connected to each of the radar transmitter 100 and the radar receiver 200 . The reference signal generation unit 300 commonly supplies a reference signal as a reference signal to the radar transmission unit 100 and the radar reception unit 200 to synchronize the processing of the radar transmission unit 100 and the radar reception unit 200 .

[Configuration of radar transmission unit 100]
Radar transmission section 100 has radar transmission signal generation sections 101-1 to 101-Nt, transmission radio sections 105-1 to 105-Nt, and transmission antennas 106-1 to 106-Nt. That is, the radar transmission section 100 has Nt transmission antennas 106, and each transmission antenna 106 is connected to an individual radar transmission signal generation section 101 and transmission radio section 105, respectively.

The radar transmission signal generator 101 generates a timing clock by multiplying the reference signal received from the reference signal generator 300 by a predetermined number, and generates a radar transmission signal based on the generated timing clock. Then, the radar transmission signal generator 101 repeatedly outputs the radar transmission signal at a predetermined radar transmission cycle (Tr). A radar transmission signal is represented by _rz (k,M)= _Iz (k,M)+ _jQz (k,M). Here, z represents a number corresponding to each transmitting antenna 106, and z=1, . . . , Nt. Also, j represents an imaginary number unit, k represents a discrete time, and M represents an ordinal number of a radar transmission cycle.

Each radar transmission signal generation section 101 is composed of a code generation section 102 , a modulation section 103 and an LPF (Low Pass Filter) 104 . Each component in the radar transmission signal generator 101-z corresponding to the z-th (z=1, . . . , Nt) transmission antenna 106 will be described below.

Specifically, the code generator 102 generates a code a(z) _n (n=1, . The codes a(z) _n (z=1, . Examples of code sequences include Walsh-Hadamard codes, M-sequence codes, and Gold codes.

The modulation unit 103 performs pulse modulation (amplitude modulation, ASK (Amplitude Shift Keying), pulse shift keying) or phase modulation (Phase Shift Keying) on the code a(z) _n received from the code generation unit 102 to generate a modulated signal is output to the LPF 104 .

LPF 104 outputs a signal component below a predetermined band limit in the modulated signal received from modulation section 103 to transmission radio section 105 as a baseband radar transmission signal.

The z-th (z=1, . A radio frequency (RF) band radar transmission signal is generated, amplified to a predetermined transmission power P [dB] by a transmission amplifier, and output to the z-th transmission antenna 106 .

The z-th (z=1, . . . , Nt) transmitting antenna 106 radiates the radar transmission signal output from the z-th transmitting radio section 105 into space.

FIG. 3 shows radar transmission signals transmitted from Nt transmission antennas 106 of radar transmission section 100 . A pulse code sequence of code length L is included in the code transmission interval Tw. In each radar transmission period Tr, a pulse code sequence is transmitted during the code transmission interval Tw, and the remaining interval (Tr-Tw) is a no-signal interval. By performing pulse modulation using No samples per pulse code (a(z) _n ), a signal of Nr (=No×L) samples is generated in each code transmission interval Tw. is included. That is, the sampling rate in modulation section 103 is (No×L)/Tw. It is also assumed that the no-signal interval (Tr-Tw) includes Nu samples.

The radar transmission section 100 may include a radar transmission signal generation section 101a shown in FIG. 4 instead of the radar transmission signal generation section 101. FIG. Radar transmission signal generation section 101a does not have code generation section 102, modulation section 103 and LPF 104 shown in FIG. The code storage unit 111 preliminarily stores code sequences generated by the code generation unit 102 (FIG. 2), and cyclically reads out the stored code sequences. The DA conversion section 112 converts the code series (digital signal) output from the code storage section 111 into an analog signal.

[Configuration of radar receiver 200]
In FIG. 2, the radar receiver 200 includes Na receiving antennas 202 to form an array antenna. The radar receiver 200 also has Na antenna system processors 201 - 1 to 201 -Na and a direction estimator 214 .

Each receiving antenna 202 receives a reflected wave signal, which is a radar transmission signal reflected by a target (object), and outputs the received reflected wave signal to the corresponding antenna system processing section 201 as a received signal.

Each antenna system processing section 201 has a receiving radio section 203 and a signal processing section 207 .

Reception radio section 203 has amplifier section 204 , frequency converter 205 , and quadrature detector 206 . The reception radio section 203 generates a timing clock by multiplying the reference signal received from the reference signal generation section 300 by a predetermined number, and operates based on the generated timing clock. Specifically, the amplifier 204 amplifies the received signal received from the receiving antenna 202 to a predetermined level, the frequency converter 205 frequency-converts the received signal in the high frequency band to the baseband band, and the quadrature detector 206 converts the frequency of the received signal to the baseband. The band received signal is converted into a baseband received signal including the I signal and the Q signal.

The signal processing unit 207 has AD conversion units 208 and 209 and separation units 210-1 to 210-Nt.

The I signal is input from the quadrature detector 206 to the AD conversion section 208 , and the Q signal is input from the quadrature detector 206 to the AD conversion section 209 . The AD conversion unit 208 converts the I signal into digital data by sampling the baseband signal including the I signal at discrete times. The AD conversion unit 209 converts the Q signal into digital data by sampling the baseband signal including the Q signal at discrete times.

Here, in the sampling of the AD converters 208 and 209, Ns discrete samples are performed per time Tp (=Tw/L) of one subpulse in the radar transmission signal. That is, the number of oversamples per subpulse is Ns.

In the following description, using the I signal Ir(k, M) and the Q signal Qr(k, M), discrete time Denote the baseband received signal at k as the complex signal x(k,M)=Ir(k,M)+jQr(k,M). In the following description, the discrete time k is based on the start timing of the radar transmission cycle (Tr) (k=1), and the signal processing unit 207 uses k Operates periodically until = (Nr + Nu) Ns/No. That is, k=1, . . . , (Nr+Nu)Ns/No. where j is the imaginary unit.

Signal processing section 207 includes Nt separation sections 210 equal to the number of systems corresponding to the number of transmission antennas 106 . Each separator 210 has a correlation calculator 211 , an adder 212 and a Doppler frequency analyzer 213 . The configuration of the z-th (z=1, . . . , Nt) separation unit 210 will be described below.

The correlation calculator 211 receives the discrete sample values Ir(k, M) and Qr(k, M) from the AD converters 208 and 209 for each radar transmission period Tr, and the discrete sample values x(k, M), A correlation calculation is performed with the pulse code a(z) _n (where z=1, . . . , Nt, n=1, . For example, the correlation calculator 211 performs a sliding correlation calculation between the discrete sample value x(k, M) and the pulse code a(z) _n . For example, the correlation calculation value AC _(z) (k, M) of the sliding correlation calculation at the discrete time k in the M-th radar transmission cycle Tr[M] is calculated based on the following equation.

In the above equation, the asterisk (*) represents the complex conjugate operator.

Correlation calculation section 211 performs correlation calculation over a period of k=1, .

Note that the correlation calculation unit 211 is not limited to performing the correlation calculation for k=1, . . . , (Nr+Nu)Ns/No. The range (ie range of k) may be limited. As a result, in the radar device 10, it is possible to reduce the computation processing amount of the correlation computing section 211. FIG. For example, the correlation calculator 211 may limit the measurement range to k=Ns(L+1), . . . (Nr+Nu)Ns/No-NsL. In this case, as shown in FIG. 5, the radar device 10 does not perform measurement in the time interval corresponding to the code transmission interval Tw.

As a result, in the radar apparatus 10, even when the radar transmission signal directly enters the radar receiving section 200, the correlation calculation section 211 does not perform processing during the period during which the radar transmission signal enters (at least the period of less than τ1). Therefore, it is possible to measure without the influence of wraparound. Further, when the measurement range (range of k) is limited, the measurement range (range of k) is similarly limited to the processing of the addition unit 212, the Doppler frequency analysis unit 213, and the direction estimation unit 214 described below. It is necessary to apply the processing described above. Thereby, the amount of processing in each component can be reduced, and the power consumption in the radar receiver 200 can be reduced.

The addition unit 212 uses the correlation calculation value AC _(z) (k, M) received from the correlation calculation unit 211 at each discrete time k of the M-th radar transmission cycle Tr to perform radar transmission a predetermined number of times (Np times). The correlation calculation value AC _(z) (k, M) is added (coherent integration) over a period of period Tr (Tr×Np). The addition (coherent integration) processing of the addition number Np over the period (Tr×Np) is represented by the following equation.

Here, CI _(z) (k, m) represents the added value of the correlation calculation value (hereinafter also referred to as the correlation added value), Np is an integer value of 1 or more, and m is It is an integer of 1 or more indicating the ordinal number of the number of additions when the number of additions Np is one unit. Also, z=1,...,Nt.

The adder 212 performs addition Np times with the output of the correlation calculator 211 obtained in units of the radar transmission period Tr as one unit. That is, the addition unit 212 aligns the timing of the discrete time k with the correlation calculation values AC _(z) (k, Np(m−1)+1) to AC _(z) (k, Np×m) as one unit. The correlation value CI _(z) (k, m) added by the k is calculated at each discrete time k. As a result, the adder 212 can improve the SNR of the reflected wave signal in a range in which the reflected wave signal from the target has high correlation due to the effect of adding the correlation calculation values Np times. Therefore, it is possible to improve the measurement performance for estimating the arrival distance of the target.

In order to obtain an ideal addition gain, it is necessary to satisfy the condition that the phase components of the correlation calculation values are aligned within a certain range in the addition interval of the number of additions Np of the correlation calculation values. In other words, the number of additions Np is preferably set based on the assumed maximum moving speed of the target to be measured. This is because the larger the assumed maximum velocity of the target, the larger the amount of variation in the Doppler frequency contained in the reflected wave from the target, and the shorter the time period with high correlation. In this case, since the number of additions Np is a small value, the effect of increasing the gain by the addition in the addition section 212 becomes small.

The Doppler frequency analysis unit 213 calculates the Nc outputs of the addition unit 212 obtained at each discrete time k, CI _(z) (k, Nc(w-1)+1) to CI _(z) (k, Nc ×w) as one unit, coherent integration is performed with discrete times k aligned. For example, the Doppler frequency analysis unit 213 performs coherent integration after correcting the phase variation Φ(fs)=2πfs(Tr×Np)ΔΦ corresponding to 2Nf different Doppler frequencies fsΔΦ as shown in the following equation. .

Here, FT_CI _(z) ^Nant (k, fs, w) is the w-th output of the Doppler frequency analysis unit 213, and the Doppler frequency fsΔΦ at discrete time k in the Nant-th antenna system processing unit 201. Coherent integration results are shown. However, Nant = 1 to Na, fs = -Nf + 1, ..., 0, ..., Nf, k = 1, ..., (Nr + Nu) Ns/No, w is an integer of 1 or more , ΔΦ is the phase rotation unit.

^, _{FT_CI ( z)} ^Nant (k, Nf−1, w) is obtained every Np×Nc periods (Tr×Np×Nc) of the radar transmission period Tr. Note that j is an imaginary unit and z=1, . . . , Nt.

When ΔΦ=1/Nc, the above-described processing of the Doppler frequency analysis unit 213 performs a discrete Fourier transform (DFT) on the output of the addition unit 212 at a sampling interval Tm=(Tr×Np) and a sampling frequency fm=1/Tm. It is equivalent to processing.

Also, by setting Nf to a number that is a power of 2, the Doppler frequency analysis unit 213 can apply fast Fourier transform (FFT) processing, and can greatly reduce the amount of arithmetic processing. At this time, when Nf>Nc, by performing zero-filling processing with CI _(z) (k, Nc(w-1)+q)=0 in the region where q>Nc, FFT processing can be applied and the amount of computational processing can be greatly reduced.

Further, in the Doppler frequency analysis unit 213, the product-sum operation shown in the above equation (3) may be sequentially performed without performing the FFT processing. That is, the Doppler frequency analysis unit ₂₁₃ calculates fs Coefficients exp[-j2πf _s T _r N _p qΔφ] corresponding to =−Nf+1, . . . , 0, . where q=0 to Nc-1.

In the following description, the w-th output FT_CI _(z) ¹ (k, fs, w), FT_CI _(z) ² obtained by performing the same processing in each of the Na antenna system processing units 201 (k, fs, w),..., FT_CI _(z) ^Na (k, fs, w) is expressed as a virtual receive array correlation vector h(k, fs, w) as shown in the following equation. The virtual receive array correlation vector h(k, fs, w) includes Nt×Na elements, which is the product of the number of transmit antennas Nt and the number of receive antennas Na. The virtual receive array correlation vector h(k, fs, w) is used for explanation of the process of estimating the direction of the reflected wave signal from the target based on the phase difference between the receive antennas 202, which will be described later. where z=1,...,Nt and b=1,...,Na.

The processing in each component of the signal processing unit 207 has been described above.

Direction estimating section 214 applies array correction value h_cal _[y] is used to calculate a virtual reception array correlation vector _{h_after_cal} (k, fs, w) in which the phase deviation and amplitude deviation between the antenna system processing units 201 are corrected. A virtual receive array correlation vector _{h_after_cal} (k, fs, w) is expressed by the following equation. Note that y = 1,..., (Nt x Na).

Then, the direction estimation unit 214 uses the virtual reception array correlation vector h _{_after_cal} (k, fs, w), based on the phase difference of the reflected wave signal between the reception antennas 202, direction estimation processing in the horizontal direction and the vertical direction. I do. The azimuth estimation unit 214 calculates a spatial profile by varying the azimuth direction θ and the elevation angle direction Φ in the direction estimation evaluation function value P(θ, φ, k, fs, w) within a predetermined angle range, and calculates the spatial profile A predetermined number of maximal peaks are extracted in descending order, and the azimuth and elevation directions of the maximal peaks are used as direction-of-arrival estimation values.

There are various evaluation function values P(θ, φ, k, fs, w) depending on the direction-of-arrival estimation algorithm. For example, an estimation method using an array antenna disclosed in Reference Non-Patent Document 1 may be used.

(Reference non-patent document 1) Direction-of-arrival estimation using signal subspace modeling Cadzow, J.A.; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28, Issue: 1 Publication Year: 1992, Page(s): 64 - 79

For example, the beamformer method can be expressed as follows. Other techniques such as Capon and MUSIC are also applicable.

where the superscript H is the Hermitian transpose operator. Also, a(θ _u , φ _v ) indicates the direction vector of the virtual receiving array for the incoming wave in the azimuth direction θ _u and elevation direction φ _v .

As described above, the direction estimation unit 214 outputs the calculated w-th direction-of-arrival estimated value, the discrete time k, the Doppler frequency fsΔΦ, and the angle _θu as the radar positioning result.

Here, the directional vector a(θ _u , φ _v ) is the element of the complex response of the virtual receiving array when the reflected wave for the radar transmission signal arrives from the azimuth θ _u direction and the elevation angle direction φ _v (Nt×Na ) is the next column vector. The complex response a(θ _u , φ _v ) of the virtual receiving array represents the phase difference calculated geometrically based on the element spacing between antennas.

Also, θ _u is obtained by changing the azimuth range for estimating the direction of arrival at a predetermined azimuth interval β ₁ . For example, θ _u is set as follows.
_θu = θmin + _uβ1 , u=0,…,NU
NU=floor[(θmax−θmin)/ _β1 ]+1
where floor(x) is a function that returns the largest integer value that does not exceed the real number x.

Also, φ _v is obtained by changing the elevation angle range for estimating the direction of arrival at a predetermined elevation angle interval β ₂ . For example, φ _v is set as follows.
_φv = φmin + vβ ₂ , v=0,…, NV
NV=floor[(φmax-φmin)/ _β2 ]+1

In this embodiment, it is assumed that direction vectors of virtual reception arrays are calculated in advance based on virtual reception array arrangements VA#1, . . . , VA#(Nt×Na) described later. Elements of the directional vector of the virtual receiving array represent phase differences geometrically optically calculated at element intervals between antennas in VA#1, .

Also, the time information k described above may be converted into distance information and output. The following equation can be used to convert time information k into distance information R(k). Here, Tw represents the code transmission period, L represents the pulse code length, and _C0 represents the speed of light.

Also, the Doppler frequency information (fsΔΦ) may be converted into a relative velocity component and output. When converting the Doppler frequency fsΔΦ into the relative velocity component vd(fs), the following equation can be used. Here, λ is the wavelength of the carrier frequency of the RF signal output from radio transmission section 107 .

[Antenna Arrangement in Radar Device 10]
The arrangement of the Nt transmitting antennas 106 and the Na receiving antennas 202 in the radar apparatus 10 having the above configuration will be described.

FIG. 6 shows the antenna arrangement of a transmitting array composed of Nt=2 transmitting antennas 106 (Tx#1, Tx#2), Na=3 receiving antennas 202 (Rx#1, Rx#2, Rx# 3) and the antenna arrangement of a virtual reception array (number of elements: Nt×Na=6) constructed based on these transmission and reception array antennas.

Each of the transmit antenna 106 and the receive antenna 202 is constructed using a sub-array element containing two antenna elements.

Also, let D _subarry be the size (width) of the subarray element, and De be the desired antenna element interval at which no grating lobe occurs in the radar detection angle range. In FIG. 6, the subarray element size D _subarry is larger than the desired antenna element spacing De (D _subarry >De). As the desired antenna element spacing De, a value of 0.5 wavelength or more and 0.75 wavelength or less is used.

Let Dt be the sub-array element spacing of the transmitting array antenna, and Dr be the sub-array element spacing of the receiving array antenna. For example, in FIG. 6, the subarray element spacing Dt of the transmitting array antenna is 1.5λ (1.5 wavelengths), and the subarray element spacing Dr of the receiving antenna is 1λ (1 wavelength). That is, the subarray element intervals Dt and Dr are approximately one wavelength (λ) or longer.

In this embodiment, the subarray element size D _subarry is wider than the desired antenna element interval De at which no grating lobe occurs in the radar detection angle range (D _subarry >De). In this case, the transmission array and the reception array are arranged so that the relationship shown in the following equation is satisfied between the subarray element spacing Dt of the transmission array antenna and the subarray element spacing Dr of the reception array antenna.
|Dt - Dr | = De (10)

That is, the absolute value of the difference between the subarray element spacing Dt of the transmitting array antenna and the subarray element spacing Dr of the receiving array antenna is the same as the desired antenna element spacing De.

As an example, FIG. 6 shows a case where De=λ/2, the subarray element spacing of the transmitting array antenna is Dt=1.5λ, and the subarray element spacing of the receiving array antenna is Dr=λ.

In this case, as shown in FIG. 6, the element spacing near the center of the virtual receiving array (other than the ends) is the desired antenna element spacing De (=|Dt-Dr|=λ/2). That is, in the virtual receiving array, an array arrangement is obtained in which no grating lobe occurs in the radar detection angle range.

FIG. 7 shows a directivity pattern (Fourier beam pattern, main beam: 0° direction) in the transmitting/receiving array antenna arrangement (De=0.5λ, Dt=1.5λ, Dr=λ) shown in FIG. As shown in FIG. 7, it can be seen that no grating lobe is generated in the angle range of ±90° from the main beam direction.

Thus, in the present embodiment, the difference (absolute value) between the element spacing of the transmitting array antenna composed of transmitting antenna 106 and the element spacing of the receiving array antenna composed of receiving antenna 202 is the grating lobe. Transmit antenna 106 and receive antenna 202 are positioned to equal the desired element spacing at which no .

By doing so, it is possible to set the element spacing of the virtual receiving array configured according to the arrangement relationship between the transmitting antenna 106 and the receiving antenna 202 to a desired element spacing that does not generate grating lobes. This makes it possible to eliminate erroneous detection due to grating lobes when performing direction estimation processing in the direction estimation unit 214 .

Therefore, according to the present embodiment, it is possible to suppress generation of unnecessary grating lobes and realize a desired directivity pattern even when array elements having a sub-array configuration are used.

Note that FIG. 6 shows, as an example, a configuration in which array antennas are arranged in a straight line in the horizontal direction in order to estimate the direction of arrival in the horizontal direction. However, in the present embodiment, in order to estimate the direction of arrival in the vertical direction, even when array antennas are arranged in a straight line in the vertical direction, the desired element spacing that does not generate grating lobes in the vertical direction is the same. virtual receive arrays can be deployed.

(Variation 1)
In Variation 1, a case will be described in which direction-of-arrival estimation is performed in both the horizontal direction and the vertical direction.

Transmit array elements or receive array elements are arranged in two dimensions, vertical and horizontal.

FIG. 8 shows the antenna arrangement of a transmitting array composed of Nt=6 transmitting antennas 106 (Tx#1 to Tx#6), Na=3 receiving antennas 202 (Rx#1, Rx#2, Rx# 3) and the antenna arrangement of a virtual reception array (number of elements: Nt×Na=18) constructed based on these transmission/reception array antennas.

In FIG. 8, the transmission array has sub-array elements arranged two-dimensionally, two in the horizontal direction and three in the vertical direction.

In FIG. 8, the size of the subarray element in the horizontal direction is _Dsubarry , and the size of the subarray element in the vertical direction is De or less. That is, the size of the antenna elements is larger than the desired antenna element spacing De in the horizontal direction and equal to or less than the desired antenna element spacing De in the vertical direction.

In FIG. 8, as an example, the desired antenna element spacing is De=λ/2, the horizontal sub-array element spacing Dt of the transmission array antenna is 1.5λ, and the vertical transmission array antenna element spacing is De. In addition, it is assumed that the sub-array element spacing in the horizontal direction of the receiving antenna is Dr=λ.

In this case, as shown in FIG. 8, in the horizontal direction, the absolute value of the difference between the subarray element spacing Dt of the transmitting array antenna and the subarray element spacing Dr of the receiving array antenna is the same as the desired antenna element spacing De. . Also, as shown in FIG. 8, in the vertical direction, the element spacing of the transmission array antenna is the same as the desired antenna element spacing De.

As a result, as shown in FIG. 8, in the horizontal direction, the element spacing near the center (other than the ends) of the virtual receiving array becomes the desired antenna element spacing De (=|Dt−Dr|=λ/2). .

Also, as shown in FIG. 8, in the vertical direction, the element spacing of the virtual reception array is the desired antenna element spacing De, like the vertical element spacing of the transmission array.

That is, in the virtual receiving array, an array arrangement is obtained in which no grating lobe occurs in the radar detection angle range in both the horizontal and vertical directions.

When direction _- of-arrival estimation in the horizontal and vertical directions is performed in the direction estimator 214, the direction estimation evaluation function value P( _{θ u} , φ _v , k, fs, w) are calculated, and the azimuth direction and elevation direction in which the maximum value is obtained are taken as the direction-of-arrival estimated value DOA(k, fs, w).

where u=1,...,NU. Note that arg max P(x) is an operator whose output value is the value of the domain that maximizes the function value P(x).

Note that there are various evaluation function values P (θ _u , φ _v , k, fs, w) depending on the direction-of-arrival estimation algorithm. For example, the estimation method using an array antenna disclosed in Reference Non-Patent Document 1 mentioned above may be used. For example, the beamformer method can be expressed as follows. Other techniques such as Capon and MUSIC are also applicable.

Here the superscript H is the Hermitian transpose operator. Also, a(θ _u , φ _v ) indicates a direction vector for incoming waves in the azimuth direction θ _u and elevation angle direction φ _v .

9A and 9B show directivity patterns in the horizontal and vertical directions (Fourier beam pattern. Main beam: 0° direction).

As shown in FIG. 9A, in the horizontal direction, it can be seen that no grating lobes are generated in the angle range of ±90° from the main beam direction. Also, as shown in FIG. 9B, it can be seen that a beam pattern is formed in which no grating lobe is generated even in the vertical direction.

By using such a transmission/reception array antenna arrangement, it is possible to eliminate erroneous detection due to grating lobes in both the horizontal direction and the vertical direction when performing direction estimation processing in the direction estimation unit 214 .

Therefore, according to Variation 1, it is possible to suppress the generation of unnecessary grating lobes and realize a desired directivity pattern even when using two-dimensionally arranged array elements having a sub-array configuration.

In FIG. 8, _the case in which the horizontal size of the subarray element is D _subarry (>De) has been described. can be applied as well. In this case, in the vertical arrangement of the transmission array, the difference (absolute value) between the element spacing of the transmission array antenna and the element spacing of the reception array antenna is equal to the desired element spacing that does not generate grating lobes. All you have to do is place the transmit array.

(Variation 2)
Variation 2 describes another example in which direction-of-arrival estimation is performed in both the horizontal direction and the vertical direction.

Specifically, in the transmission array antenna, when the element spacing in the horizontal direction is Dt (>De) and the element spacing in the vertical direction is the desired antenna element spacing De, the transmission array antennas are adjacent in the vertical direction, Two subarray element arrays arranged in a straight line in the horizontal direction are arranged with a gap equal to the desired antenna element gap De in the horizontal direction.

FIG. 10 shows the antenna arrangement of a transmitting array composed of Nt=6 transmitting antennas 106 (Tx#1 to Tx#6), Na=3 receiving antennas 202 (Rx#1, Rx#2, Rx# 3) and the antenna arrangement of a virtual reception array (number of elements: Nt×Na=18) constructed based on these transmission/reception array antennas.

In FIG. 10, the transmission array has sub-array elements arranged two-dimensionally, two in the horizontal direction and three in the vertical direction.

In FIG. 10, the size of the subarray element in the horizontal direction is _Dsubarry , and the size of the subarray element in the vertical direction is De or less. That is, the size of the antenna elements is larger than the desired antenna element spacing De in the horizontal direction and equal to or less than the desired antenna element spacing De in the vertical direction.

In FIG. 10, similarly to FIG. 8, the desired antenna element spacing De is λ/2, the horizontal sub-array element spacing Dt of the transmission array antenna is 1.5λ, and the vertical transmission array antenna element spacing is De. . In addition, it is assumed that the sub-array element spacing in the horizontal direction of the receiving array antenna is Dr=λ.

As in variation 1 (FIG. 8), as shown in FIG. 10, in the horizontal direction, the absolute value of the difference between the subarray element spacing Dt of the transmitting array antenna and the subarray element spacing Dr of the receiving array antenna is the desired antenna element Identical to interval De. Also, as shown in FIG. 10, in the vertical direction, the element spacing of the transmission array antenna is the same as the desired antenna element spacing De.

Furthermore, in FIG. 10, the transmitting antennas 106 separated by the antenna element spacing De in the vertical direction of the transmitting array antenna (transmitting antennas 106 adjacent in the vertical direction) are arranged at the same spacing as the antenna element spacing De in the horizontal direction. be. In other words, in the transmission array antenna, two subarray element arrays that are adjacent in the vertical direction and linearly arranged in the horizontal direction are arranged with the same spacing as the desired element spacing in the horizontal direction.

For example, the arrangement of transmitting antennas Tx#1 and Tx#2 shown in FIG. 10 (that is, subarray element arrangement; the same shall apply hereinafter) and the arrangement of transmitting antennas T#3 and Tx#4 adjacent to the arrangement in the vertical direction are , are shifted at the same interval as the antenna element interval De. Similarly, the array of transmitting antennas Tx#3 and Tx#4 and the array of transmitting antennas T#5 and T#6 vertically adjacent to the array are shifted in the horizontal direction by the same interval as the antenna element interval De. are placed.

In FIG. 10, in the horizontal direction, the element spacing near the center of the virtual receiving array (other than the ends) is the desired antenna element spacing De (=|Dt-Dr|=λ/2). Also, as shown in FIG. 10, in the vertical direction, the element spacing of the virtual reception array is the desired antenna element spacing De, similar to the vertical element spacing of the transmission array. That is, in the virtual receiving array, an array arrangement is obtained in which no grating lobe occurs in the radar detection angle range.

Furthermore, as shown in FIG. 10, in the vertical direction of the virtual receiving array, the arrangement of the central (second stage) array element is compared with the arrangements of the other array elements (first and third stages). are arranged with a horizontal shift of De. As a result, in FIG. 10, the antenna elements are more closely spaced in the two-dimensional plane in which the virtual receiving array is arranged compared to Variation 1 (FIG. 8). This allows the virtual receive array to reduce the side lobe level.

11A and 11B show directivity patterns (Fourier beam patterns) in the horizontal and vertical directions of the transmission/reception array antenna arrangement (when De=0.5λ, Dt=1.5λ, and Dr=λ) shown in FIG. 0° direction).

As shown in FIG. 11A, it can be seen that in the horizontal direction, no grating lobes are generated in the angular range of ±90° from the main beam direction. In addition, as shown in FIG. 11B, it can be seen that a beam pattern in which no grating lobe occurs also in the vertical direction is formed.

Furthermore, compared with Variation 1 (FIG. 9A), it can be seen that the sidelobe level is reduced in the horizontal directivity pattern, as shown in FIG. 11A.

By using such an arrangement of the transmitting/receiving array antennas, it is possible to eliminate erroneous detection due to grating lobes and side lobes in both the horizontal direction and the vertical direction when performing direction estimation processing in the direction estimation unit 214. can.

Therefore, according to Variation 2, even when using two-dimensionally arranged array elements having a sub-array configuration, generation of unnecessary grating lobes and side lobe levels can be suppressed, and a desired directivity pattern can be realized. can.

(Variation 3)
Variation 3 describes another example in which direction-of-arrival estimation is performed in both the horizontal direction and the vertical direction.

Specifically, in the transmission array antenna, the interval between the subarray element arrays adjacent in the vertical direction and linearly arranged in the horizontal direction is the interval obtained by multiplying the desired antenna element interval De by a constant α. Two sub-array element arrays adjacent to each other in the direction and linearly aligned in the horizontal direction are displaced in the horizontal direction by an interval obtained by multiplying a desired antenna element interval De by a constant β.

FIG. 12 shows the antenna arrangement of a transmitting array composed of Nt=6 transmitting antennas 106 (Tx#1 to Tx#6), Na=3 receiving antennas 202 (Rx#1, Rx#2, Rx# 3) and the antenna arrangement of a virtual reception array (number of elements: Nt×Na=18) constructed based on these transmission/reception array antennas.

In FIG. 12, the transmission array has sub-array elements arranged two-dimensionally, two in the horizontal direction and three in the vertical direction.

In FIG. 12, the size of the subarray element in the horizontal direction is _Dsubarry , and the size of the subarray element in the vertical direction is De or less. That is, the size of the antenna elements is larger than the desired antenna element spacing De in the horizontal direction and equal to or less than the desired antenna element spacing De in the vertical direction.

In FIG. 12, as in FIG. 8, the desired antenna element spacing De=λ/2, the horizontal sub-array element spacing of the transmitting array antenna Dt=1.5λ, and the horizontal sub-array element spacing Dr=λ of the receiving array antenna. and In addition, it is assumed that the sub-array element spacing in the horizontal direction of the receiving array antenna is Dr=λ.

As in variations 1 and 2 (FIGS. 8 and 10), as shown in FIG. 12, in the horizontal direction, the absolute value of the difference between the subarray element spacing Dt of the transmitting array antenna and the subarray element spacing Dr of the receiving array antenna is , is the same as the desired antenna element spacing De.

On the other hand, as shown in FIG. 12, in the vertical direction, the element spacing of the transmission array antenna is the spacing αDe obtained by multiplying the desired antenna element spacing De by a constant α.

Also, in FIG. 12, the transmitting antennas 106 separated by the element spacing αDe in the vertical direction of the transmitting array antenna (transmitting antennas 106 adjacent in the vertical direction) are obtained by multiplying the desired antenna element spacing De in the horizontal direction by a constant β. They are arranged with an interval βDe. In other words, in the transmission array antenna, two sub-array element arrays that are adjacent in the vertical direction and linearly arranged in the horizontal direction are arranged with an interval of β times the desired element interval in the horizontal direction.

For example, the array of transmitting antennas Tx#1 and Tx#2 shown in FIG. 12 and the array of transmitting antennas T#3 and Tx#4 adjacent to the array in the vertical direction are arranged with a gap of βDe. Similarly, the array of transmitting antennas Tx#3 and Tx#4 and the array of transmitting antennas T#5 and T#6 vertically adjacent to the array are arranged with an interval βDe in the horizontal direction.

For example, α=(3) ^0.5 /2≈0.866 and β=0.5.

In FIG. 12, in the horizontal direction, the element spacing near the center of the virtual receiving array (other than the ends) is the desired antenna element spacing De (=|Dt-Dr|=λ/2).

Also, as shown in FIG. 12, in the vertical direction, the element spacing of the virtual reception array is αDe (=(3) ^0.5 De), similar to the vertical element spacing of the transmission array.

That is, in the virtual receiving array, an array arrangement is obtained in which no grating lobe occurs in the radar detection angle range.

Furthermore, as shown in FIG. 12, in the vertical direction of the virtual receiving array, the arrangement of the central (second stage) array element is compared with the arrangements of the other array elements (first and third stages). are arranged with a horizontal shift of βDe (=0.5De).

As a result, in FIG. 12, similar to Variation 2 (FIG. 10), compared to Variation 1 (FIG. 8), the spacing between the antenna elements in the two-dimensional plane on which the virtual receiving array is arranged becomes closer. This allows the virtual receive array to reduce the side lobe level.

Here, as shown in FIG. 12, near the center of the virtual receiving array, the distance between three adjacent antenna elements on the two-dimensional plane on which the virtual receiving array is arranged is the desired antenna element distance De. In other words, a straight line connecting three adjacent array elements on the two-dimensional plane on which the virtual receiving array is arranged forms an equilateral triangle whose one side is the antenna element interval De. The equilateral triangular lattice arrangement has higher grating lobe suppression performance than the rectangular lattice arrangement with the same aperture length.

That is, the constants α and β are set so that the element spacing between three adjacent array elements in two dimensions in the vertical and horizontal directions is the desired antenna element spacing De (an equilateral triangle with one side De). should be set to

13A and 13B show the horizontal and vertical directions of the transmission/reception array antenna arrangement (De=0.5λ, Dt=1.5λ, Dr=1λ, α=(3) ^0.5 /2, β=0.5) shown in FIG. A directivity pattern (Fourier beam pattern, main beam: 0° direction) is shown.

As shown in FIG. 13A, in the horizontal direction, no grating lobes are generated in the angle range of ±90° from the main beam direction. In addition, as shown in FIG. 13B, it can be seen that a beam pattern in which no grating lobe occurs also in the vertical direction is formed.

Furthermore, when compared with Variation 1 (FIG. 9A), it can be seen that the side lobe level is reduced in the horizontal directivity pattern as shown in FIG. 13A.

Also, when compared with Variation 2 (FIG. 11A), as shown in FIG. It turns out that it is reduced.

By using such an arrangement of the transmitting/receiving array antennas, it is possible to eliminate erroneous detection due to grating lobes and side lobes in both the horizontal direction and the vertical direction when performing direction estimation processing in the direction estimation unit 214. can.

Therefore, according to Variation 3, even when using two-dimensionally arranged array elements having a sub-array configuration, generation of unnecessary grating lobes and side lobe levels can be suppressed, and a desired directivity pattern can be realized. can.

The embodiment according to one aspect of the present disclosure has been described above.

Note that the operations according to the above-described embodiment and each modification may be appropriately combined and implemented.

Further, in the above embodiment, the case where the number of transmitting antennas 106 Nt=2 or 3 and the number of receiving antennas 202 Na=3 has been exemplified. However, the number Nt of transmitting antennas 106 and the number Na of receiving antennas 202 are not limited to these numbers.

Further, in the above embodiment, a case has been described in which the transmitting antenna 106 and the receiving antenna 202 are sub-array elements composed of two antenna elements. It may be composed of more than one element.

Also, in variations 1 to 3 of the above embodiment, the case where the transmission array antennas are arranged two-dimensionally in the horizontal direction and the vertical direction, and the reception array antennas are arranged in the one-dimensional direction in the horizontal direction has been described. However, the present disclosure may have the receive array antennas arranged in two dimensions and the transmit array antennas arranged in one dimension. In this case, the arrangement of the subarray elements in the transmission array antenna described above may be applied to the arrangement of the subarray elements in the reception array antenna.

Further, in the above embodiment, the case where the size of the antenna elements is larger than the desired antenna element spacing De in the horizontal direction and equal to or less than the desired antenna element spacing De in the vertical direction has been described. It may be larger than the desired antenna element spacing De in the vertical direction and less than or equal to the desired antenna element spacing De in the horizontal direction. In this case, the arrangement of the subarray elements in the transmission/reception array antenna described above may be switched between the horizontal direction and the vertical direction.

Also, in the above embodiment, the case of using a coded pulse radar has been described, but the present disclosure is also applicable to a radar system using a frequency-modulated pulse wave, such as a chirp pulse radar.

In addition, in the radar device 10 shown in FIG. 2, the radar transmission section 100 and the radar reception section 200 may be individually arranged at physically separated locations.

In addition, in the radar device, the radar transmission unit transmits different transmission signals that are code division multiplexed from a plurality of transmission antennas, and the radar reception unit separates each transmission signal and performs reception processing. The configuration of the radar device is not limited to this, and the radar transmission unit transmits different transmission signals that are frequency division multiplexed from a plurality of transmission antennas, and the radar reception unit separates each transmission signal and performs reception processing. may be configured. Similarly, the configuration of the radar apparatus may be a configuration in which the radar transmission section transmits time-division multiplexed transmission signals from a plurality of transmission antennas, and the radar reception section performs reception processing, as in the above embodiment. effect is obtained.

Although not shown, the radar device 10 has, for example, a CPU (Central Processing Unit), a storage medium such as a ROM (Read Only Memory) storing a control program, and a working memory such as a RAM (Random Access Memory). . In this case, the functions of the respective units described above are realized by the CPU executing the control program. However, the hardware configuration of the radar device 10 is not limited to this example. For example, each functional unit of the radar device 10 may be realized as an IC (Integrated Circuit), which is an integrated circuit. Each functional unit may be individually integrated into one chip, or may be integrated into one chip so as to include part or all of it.

<Summary of this disclosure>
A radar apparatus according to the present disclosure includes a radar transmitter that transmits a radar signal at a predetermined transmission cycle using a transmission array antenna, and a reception array antenna that receives a reflected wave signal of the radar signal reflected by a target. and a radar receiver, wherein the transmission array antenna and the reception array antenna each include a plurality of subarray elements, and the plurality of subarray elements in the transmission array antenna and the reception array antenna have a first arranged on a straight line in a direction, each of said sub-array elements comprising a plurality of antenna elements, the size of said sub-array elements being larger than a desired antenna element spacing in said first direction, and sub-array elements of said transmitting array antenna The absolute value of the difference between the spacing and the sub-array element spacing of the receiving array antenna is the same as the desired antenna element spacing.

Further, in the radar device of the present disclosure, the desired antenna element spacing is 0.5 wavelength or more and 0.75 wavelength or less.

Further, in the radar apparatus of the present disclosure, in either one of the transmission array antenna and the reception array antenna, the plurality of subarray elements are arranged in a second direction orthogonal to the first direction, and If the subarray element size is larger than the desired antenna element spacing in the first direction and is equal to or smaller than the desired antenna element spacing in the second direction, An absolute value of a difference between a subarray element spacing and a subarray element spacing of the receiving array antenna is the same as the desired antenna element spacing, and in the second direction, the subarray element spacing is equal to the desired antenna element spacing. Identical to interval.

Also, in the radar apparatus of the present disclosure, the plurality of subarray elements arranged in the second direction are arranged with the same spacing as the desired antenna element spacing shifted in the first direction.

Further, in the radar apparatus of the present disclosure, in either one of the transmission array antenna and the reception array antenna, the plurality of subarray elements are arranged in a second direction orthogonal to the first direction, and When the size of the subarray elements is larger than the desired antenna element spacing in the first direction and equal to or less than the desired antenna element spacing in the second direction, The absolute value of the difference between the subarray element spacing and the subarray element spacing of the receiving array antenna is the same as the desired element spacing, and in the second direction, the subarray element spacing is equal to the desired antenna element spacing. ((√3)/2) times the length of the plurality of sub-array elements arranged in the second direction is (1/2) the desired antenna element spacing in the first direction. Double spacing is shifted and arranged.

Various embodiments (variations) have been described above with reference to the drawings, but it goes without saying that the present disclosure is not limited to such examples. It is obvious that a person skilled in the art can conceive of various modifications or modifications within the scope described in the claims, and these also belong to the technical scope of the present disclosure. Understood. Also, the constituent elements in the above-described embodiments (variations) may be combined arbitrarily within the scope of the disclosure.

In each of the above-described embodiments, the present disclosure has been described as an example configured using hardware, but the present disclosure can also be realized by software in cooperation with hardware.

Also, each functional block used in the description of each of the above embodiments is typically implemented as an LSI, which is an integrated circuit. The integrated circuit may control each functional block used in the description of the above embodiments and may have inputs and outputs. These may be made into one chip individually, or may be made into one chip so as to include part or all of them. Although LSI is used here, it may also be called IC, system LSI, super LSI, or ultra LSI depending on the degree of integration.

Also, the method of circuit integration is not limited to LSI, and may be implemented using a dedicated circuit or a general-purpose processor. An FPGA (Field Programmable Gate Array) that can be programmed after LSI manufacturing, and a reconfigurable processor (Reconfigurable Processor) that can reconfigure connections or settings of circuit cells inside the LSI may be used.

Furthermore, if an integration technology that replaces the LSI appears due to advances in semiconductor technology or another derived technology, the technology may naturally be used to integrate the functional blocks. Application of biotechnology, etc. is possible.

The present disclosure is suitable as a radar device that detects a wide-angle range.

REFERENCE SIGNS LIST 10 radar device 100 radar transmitter 200 radar receiver 300 reference signal generator 400 controller 101, 101a radar transmission signal generator 102 code generator 103 modulator 104 LPF
105 Transmission radio section 106 Transmission antenna 111 Code storage section 112 DA conversion section 201 Antenna system processing section 202 Reception antenna 203 Reception radio section 204 Amplifier 205 Frequency converter 206 Quadrature detector 207 Signal processing section 208, 209 AD conversion section 210 Separation section 211 Correlation calculator 212 Adder 213 Doppler frequency analyzer 214 Direction estimator

Claims (10)

a transmitting array antenna;
a receiving array antenna;
a radar transmitter that transmits a radar signal using the transmission array antenna;
a radar receiver that receives a reflected wave signal of the radar signal reflected by a target using the reception array antenna;
and
the transmit array antenna includes a plurality of transmit antennas;
the receiving array antenna includes a plurality of receiving antennas;
The plurality of transmitting antennas are arranged at different positions in a first direction,
The plurality of receiving antennas are arranged at different positions in the first direction,
An interval between two adjacent transmitting antennas among the plurality of transmitting antennas is one wavelength or more in the first direction,
An interval between two adjacent receiving antennas among the plurality of receiving antennas is one wavelength or more in the first direction,
the absolute value of the difference between the interval between the two adjacent transmitting antennas and the interval between the two adjacent receiving antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction;
at least one of the plurality of transmit antennas and the plurality of receive antennas includes a plurality of antenna elements arranged in the first direction;
radar equipment. At least one of the plurality of transmitting antennas and the plurality of receiving antennas, including the plurality of antenna elements arranged in the first direction, has a size in the first direction equal to the spacing between the two adjacent transmitting antennas. greater than the absolute value of the difference between the distances between the two adjacent receiving antennas;
The radar system according to claim 1. At least one of the plurality of transmitting antennas and the plurality of receiving antennas, including the plurality of antenna elements arranged in the first direction, has a size in the first direction equal to the spacing between the two adjacent transmitting antennas. smaller than the narrowest interval between the two adjacent receiving antennas;
3. The radar device according to claim 1 or 2. A virtual receiving array antenna formed by the transmitting array antenna and the receiving array antenna has a plurality of virtual receiving antennas,
At least one of two adjacent virtual reception antenna intervals among the plurality of virtual reception antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction.
Radar device according to any one of claims 1 to 3. the wavelength is determined by the frequency of the radar signal;
Radar device according to any one of claims 1 to 4. a first array antenna;
a second array antenna;
a radar transmitter that transmits a radar signal using one of the first array antenna and the second array antenna;
a radar receiver that receives a reflected wave signal of the radar signal reflected by a target using the other of the first array antenna and the second array antenna;
and
The first array antenna includes a plurality of first antennas,
Each of the plurality of first antennas is arranged at a different position in a first direction,
the second array antenna includes a plurality of second antennas,
Each of the plurality of second antennas is arranged at a different position in the first direction,
The distance between two adjacent first antennas among the plurality of first antennas is one wavelength or more in the first direction,
an interval between two adjacent second antennas among the plurality of second antennas is one wavelength or more in the first direction;
The absolute value of the difference between the interval between the two adjacent first antennas and the interval between the two adjacent second antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction. including something
at least one of the plurality of first antennas and the plurality of second antennas includes a plurality of antenna elements arranged in the first direction;
radar equipment. At least one of the plurality of first antennas and the plurality of second antennas, which includes a plurality of antenna elements arranged in the first direction, has a size in the first direction equal to that of the two adjacent second antennas. larger than the absolute value of the difference between the spacing of one antenna and the spacing of the two adjacent second antennas;
The radar device according to claim 6. At least one of the plurality of first antennas and the plurality of second antennas, which includes a plurality of antenna elements arranged in the first direction, has a size in the first direction equal to that of the two adjacent second antennas. smaller than the narrowest interval between the interval between one antenna and the interval between the two adjacent second antennas;
The radar device according to claim 6 or 7. a virtual reception array antenna composed of the first array antenna and the second array antenna has a plurality of virtual reception antennas;
At least one of two adjacent virtual reception antenna intervals of the plurality of virtual reception antennas is 0.5 wavelength or more and 0.75 wavelength or less in the first direction.
The radar device according to any one of claims 6 to 8. the wavelength is determined by the frequency of the radar signal;
Radar installation according to any one of claims 6 to 9.