JP2021061699A - Motor control device, motor control method, and electric power steering device - Google Patents

Abstract

To provide a motor control device, a motor control method, and an electric power steering device capable of easily and appropriately realizing a maximum torque control when performing a drive control of a permanent magnet synchronous motor.SOLUTION: A motor control device has: a deflection angle calculation unit calculating a deflection angle between a dq coordinate system which is a rotation coordinate system and a ft coordinate system which is a maximum torque control coordinate system, based on dq-axis current of a permanent magnet synchronous motor and a parameter which may be variable due to magnet saturation; a t-axis current command generation unit generating a t-axis current command value in the ft coordinate system based on a torque command value of the permanent magnet synchronous motor; a f-axis current command generation unit generating a f-axis current command value as zero in the ft coordinate system; and a dq-axis current command generation unit generating a dq-axis current command value by a rotation coordinate conversion based on the deflection angle calculated by the deflection angle calculation unit, the t-axis current command value generated by the t-axis current command value generation unit, and the f-axis command value generated by the f-axis current command generation unit.SELECTED DRAWING: Figure 3

Description

The present invention relates to a motor control device, a motor control method, and an electric power steering device.

Conventionally, a control method called vector control has been widely used as a control method for motor drive control. In this control method, the dq coordinates are considered with the main magnetic flux direction of the rotor as the d-axis (direct) and the direction electrically perpendicular to it as the q-axis (quadrature), and the three-phase current and voltage are converted into dq coordinates. , This is a method of controlling as a vector on the dq coordinate. When this dq vector control is used, the maximum torque control that maximizes the torque for the same current amplitude can be performed more easily than the control with three phases, but in order to realize the maximum torque control. The formula for calculating the d-axis current is complicated.
Therefore, in order to easily realize the maximum torque control, for example, there is a technique described in Patent Document 1. This technology measures the axis error, which is the phase difference between the rotating coordinate system (dmqm coordinate system) different from the dq coordinate system and the control coordinate system (γδ coordinate system) different from the dq coordinate system, and the output torque and torque of the motor. This is a technology that controls according to the current for weakening magnetic flux so that the torque error between the command values is reduced. Patent Document 1 discloses that maximum torque control is realized when the dm-axis current idm = 0.

Japanese Unexamined Patent Publication No. 2012-130183

In the technique described in Patent Document 1, the armature interlinkage magnetic flux of the permanent magnet, the d-axis inductance, and the q-axis inductance are used as control parameters, but in consideration of the fact that these fluctuate due to magnetic saturation. Not. Therefore, there is a possibility that the maximum torque control cannot be appropriately realized.
Therefore, an object of the present invention is to provide a motor control device, a motor control method, and an electric power steering device that can easily and appropriately realize maximum torque control in driving control of a permanent magnet synchronous motor.

In order to solve the above problems, the motor control device of one aspect of the present invention is a motor control device that drives and controls a permanent magnet synchronous motor, and fluctuates due to the dq axis current of the permanent magnet synchronous motor and magnetic saturation. Based on the possible parameters, the deviation angle calculation unit that calculates the deviation angle between the dq coordinate system that is the rotation coordinate system and the ft coordinate system that is the maximum torque control coordinate system, and the torque command value of the permanent magnet synchronous motor Based on this, the t-axis current command generation unit that generates the t-axis current command value in the ft coordinate system, the f-axis current command generation unit that generates the f-axis current command value in the ft coordinate system as zero, and the deviation angle. Based on the deviation angle calculated by the calculation unit, the t-axis current command value generated by the t-axis current command generation unit, and the f-axis current command value generated by the f-axis current command generation unit. It includes a dq-axis current command generation unit that generates a dq-axis current command value by rotational coordinate conversion.
In this way, since the ft coordinate system, which is the maximum torque control coordinate system, is used and the current of the f-axis component is controlled to be 0, the maximum torque control can be easily realized. Further, since the parameter fluctuation due to magnetic saturation is taken into consideration when calculating the declination between the dq coordinate system and the ft coordinate system, the maximum torque control can be appropriately realized.

Further, in the above motor control device, the parameters may be a permanent magnet interlinkage magnetic flux, a d-axis inductance, and a q-axis inductance. In this case, the maximum torque control can be appropriately realized in consideration of the fluctuation of the motor parameters due to magnetic saturation.

Further, in the above motor control device, the deviation angle calculation unit makes the difference between the d-axis inductance and the q-axis inductance constant with respect to the dq-axis current, and sets the permanent magnet interlinkage magnetic flux to 2 of the q-axis current. The deflection angle may be calculated as a next function.
As described above, it is assumed that the fluctuations of the inductances of the d-axis and q-axis due to magnetic saturation show substantially similar behavior, and the dq-axis inductance difference may be constant with respect to the dq-axis current. In this case, the declination can be calculated by a simple calculation. Further, by making the permanent magnet interlinkage magnetic flux a quadratic function of the q-axis current, the fluctuation of the permanent magnet interlinkage magnetic flux can be appropriately considered, and the deviation angle can be calculated with high accuracy.

Further, in the above motor control device, the deviation angle calculation unit sets the difference between the d-axis inductance and the q-axis inductance to be constant with respect to the dq-axis current, and sets the permanent magnet interlinkage magnetic flux to 1 of the q-axis current. The deflection angle may be calculated as a next function.
As described above, it is assumed that the fluctuations of the inductances of the d-axis and q-axis due to magnetic saturation show substantially similar behavior, and the dq-axis inductance difference may be constant with respect to the dq-axis current. In this case, the declination can be calculated by a simple calculation. Further, by making the permanent magnet interlinkage magnetic flux a linear function of the q-axis current, the declination can be calculated by a simpler calculation.

Furthermore, in the motor control device, the declination calculation unit uses the declination from π / 2 according to the formula obtained by differentiating the d-axis current with the q-axis current, which is derived from the torque formula of the permanent magnet synchronous motor. The angle may be calculated by subtracting the above angle, and the argument may be calculated from the angle.
In this case, the declination can be calculated by a simple calculation. For example, when the permanent magnet interlinkage magnetic flux is considered as a parameter that can fluctuate due to magnetic saturation, the d-axis current is the q-axis current even when the permanent magnet interlinkage magnetic flux is considered as a quadratic function of the q-axis current. The formula differentiated by is a simple formula. Therefore, the declination can be calculated easily and appropriately.

Further, in the above motor control device, the declination calculation unit may calculate the declination according to the equation obtained by differentiating the q-axis current with the d-axis current derived from the torque equation of the permanent magnet synchronous motor. Good.
In this case, the declination can be calculated directly. Further, when the permanent magnet interlinkage magnetic flux is considered as a parameter that can fluctuate due to magnetic saturation, and when the permanent magnet interlinkage magnetic flux is considered as a linear function of the q-axis current, the q-axis current is differentiated by the d-axis current. The formula can be a relatively simple formula. Therefore, the declination can be calculated easily and appropriately.

Further, the motor control device has a torque ripple estimation unit that estimates the torque ripple of the permanent magnet synchronous motor, and a torque command that corrects the torque command value based on the torque ripple estimated by the torque ripple estimation unit. The t-axis current command generation unit may further include a correction unit, and the t-axis current command generation unit may generate the t-axis current command value based on the torque command value corrected by the torque command correction unit.
In this case, torque ripple can be appropriately suppressed. Further, since the torque ripple estimated value can be used as it is as a correction value and superimposed on the torque command value, the torque ripple can be easily suppressed.

Further, the motor control method according to one aspect of the present invention is a motor control method for driving and controlling a permanent magnet synchronous motor, which is based on the dq-axis current of the permanent magnet synchronous motor and parameters that can fluctuate due to magnetic saturation. Then, based on the step of calculating the deviation angle between the dq coordinate system which is the rotation coordinate system and the ft coordinate system which is the maximum torque control coordinate system and the torque command value of the permanent magnet synchronous motor, t in the ft coordinate system. The step of generating the shaft current command value, the step of generating the f-axis current command value in the ft coordinate system as zero, the deviation angle, the t-axis current command value, and the f-axis current command value are Based on this, it includes a step of generating a dq-axis current command value by rotational coordinate conversion.
In this way, since the ft coordinate system, which is the maximum torque control coordinate system, is used and the current of the f-axis component is controlled to be 0, the maximum torque control can be easily realized. Further, since the parameter fluctuation due to magnetic saturation is taken into consideration when calculating the declination between the dq coordinate system and the ft coordinate system, appropriate maximum torque control can be realized.

Further, the electric power steering device according to one aspect of the present invention includes any of the above motor control devices.
As a result, the maximum torque control of the in-vehicle permanent magnet synchronous motor for performing steering assist control (steering assist) can be easily and appropriately realized. Therefore, it is possible to obtain an electric power steering device in which high efficiency is realized.

According to one aspect of the present invention, the maximum torque control can be easily and appropriately realized in the drive control of the permanent magnet synchronous motor.

FIG. 1 is a configuration diagram of a motor system. FIG. 2 is a functional block diagram of the motor control device. FIG. 3 is a functional block diagram showing details of the current command generation unit. FIG. 4 is a diagram illustrating a method of calculating the declination δ. FIG. 5 is an explanatory diagram of maximum torque control. FIG. 6 is a block diagram of the command generation unit in the second embodiment. FIG. 7 is a functional block diagram of the current command generation unit in the second embodiment. FIG. 8 is a diagram of a vehicle equipped with an electric power steering device. FIG. 9 is a schematic view of the electric power steering device.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
The scope of the present invention is not limited to the following embodiments, and can be arbitrarily changed within the scope of the technical idea of the present invention.

(First Embodiment)
FIG. 1 is a configuration diagram of a motor system 300 according to the present embodiment.
The motor system 300 includes a motor 310, a motor control device 320, and an inverter 330. The motor control device 320 includes a command generation unit 321 and a current controller 322, and drives and controls the motor 310 via an inverter 330.
Here, the motor 310 is a permanent magnet synchronous motor. For example, the motor 310 can be various permanent magnet synchronous motors such as a surface magnet type SPMSM (Surface Permanent Magnet Synchronous Motor), an embedded magnet type IPMSM (Interior Permanent Magnet Synchronous Motor), and a claw pole type motor.

The command generation unit 321 generates a current command value i ^{* from the torque command value τ *} of the motor 310.
Current controller 322, based on the current command value i ^*, and outputs a voltage command value v ^* by feedback control of the motor current.
The inverter 330 turns on / off the switching elements constituting the inverter 330 based on the PWM signal generated based on the voltage command value v ^{*, and supplies the voltage v to the motor 310.} As a result, torque τ is generated in the motor 310.

Hereinafter, the functions of the motor control device 320 will be specifically described.
FIG. 2 is a functional block diagram of the motor control device 320.
As shown in FIG. 2, the motor control device 320 includes an angle detection unit 301, an angular velocity calculation unit 302, a three-phase / two-phase conversion unit 303, a current command generation unit 304, a current control unit 305, and two. A phase / three-phase conversion unit 306 is provided.

The angle detection unit 301 converts the motor angle (mechanical angle) θ _M detected by the angle sensor 310a into the motor angle (electrical angle) θ _E.
The angular velocity calculation unit 302 calculates the motor angular velocity ω by differentiating the _{motor angle θ E output from the angle detection unit 301.}
The three-phase / two-phase conversion unit 303 converts the three-phase motor currents iu, iv and iwa into three-phase / two-phase and outputs the dq-axis currents id and iq.

The current command generator 304 ^{inputs the torque command value τ *} , the motor angle θ _E , the motor acceleration ω, the dq axis current id, and iq, and uses the dq axis current command value id ^{* as the current command value i * in FIG.} Generate iq ^*. Here, the current command generation unit 304 generates dq-axis current command values id ^* and iq ^* that realize maximum torque control.
In the present embodiment, the maximum torque / current (MTPA: Maximum Torque Per Ampere) control is targeted as the maximum torque control. Current command generating unit 304, dq axis current command value id ^* from the torque command tau ^*, when generating iq ^*, utilizing ft coordinate system is the maximum torque control coordinate system, and controls the f-axis current to 0 This realizes maximum torque control. The specific functional configuration of the current command generation unit 304 will be described later.

The current control unit 305 calculates the current deviations between the dq-axis current command values id ^* and iq ^* and the dq-axis current id and iq, respectively. Then, the current control unit 305 performs proportional integral control with respect to the calculated current deviation, and calculates the dq-axis voltage command values vd and vq as ^{the voltage command values v * in FIG. 1.} In this way, the current control unit 305 performs vector control based on the ^{dq-axis current command values id *} and iq ^*.
The two-phase / three-phase conversion unit 306 converts the two-phase voltage command values vd and vq into the three-phase voltage command values vu, vv and vw. Drive power is supplied to the motor 310 based on the voltage command values vu, vv and vw.

FIG. 3 is a functional block diagram showing a specific configuration of the current command generation unit 304 shown in FIG.
As shown in FIG. 3, the current command generation unit 304 includes a declination calculation unit 304a, a t-axis current command generation unit 304b, an f-axis current command generation unit 304c, and a dq-axis current command generation unit 304d. Be prepared.
The declination calculation unit 304a controls the maximum torque with respect to the rotating coordinate system (dq coordinate system) shown in FIG. 4 based on the detected dq-axis currents id and iq of the motor 310 and parameters that can fluctuate due to magnetic saturation. The declination δ of the coordinate system (ft coordinate system) is calculated. The method of calculating the declination δ in the declination calculation unit 304a will be described later.

The t-axis current command generation unit 304b converts the torque command value τ ^* into a current on the t-axis of the maximum torque control coordinate system, and outputs ^{this as the t-axis current command value it *.} For example, the t-axis current command generation unit 304b can generate the t-axis current command value it ^{* from the torque command value τ *} by using the conversion table between the torque and the t-axis current. The conversion table shall be created in advance by actual measurement or analysis and stored in a memory or the like.
Incidentally, a technique for generating a t-axis current command value it ^* from the torque command value tau ^* is not limited to the method using the above conversion table. For example, may be generated t-axis current command value it ^* by using a function such as to derive a t-axis current command value it ^* from the torque command value tau ^*.

The f-axis current command generation unit 304c ^{outputs the f-axis current command value if *} = 0.
The dq axis current command generation unit 304d performs coordinate conversion (rotational coordinate conversion) from the ft axis to the dq axis based on ^{the ft current command values it *} , if ^{* and the declination δ, and the dq axis current command value id. *} , iq ^* is generated. The dq-axis current command values id ^* and iq ^* thus obtained are the current command values that realize the maximum torque control.
As described above, the motor control method in the present embodiment is based on the dq axis current of the permanent magnet synchronous motor and the parameters that can fluctuate due to magnetic saturation, and is a rotating coordinate system dq coordinate system and a maximum torque control coordinate system. The step of calculating the deviation angle from the ft coordinate system, the step of generating the t-axis current command value in the ft coordinate system based on the torque command value of the permanent magnet synchronous motor, and the f-axis current command in the ft coordinate system. It includes a step of generating a value as zero and a step of generating a dq-axis current command value by rotating coordinate conversion based on the above-mentioned deviation angle, t-axis current command value and f-axis current command value.

Hereinafter, the method of calculating the declination δ in the declination calculation unit 304a will be specifically described.
In the present embodiment, as described above, the declination calculation unit 304a calculates the declination δ of the maximum torque control coordinate system based on the dq-axis currents id and iq and the parameters that can fluctuate due to magnetic saturation. Here, the parameters that can fluctuate due to magnetic saturation are the permanent magnet interlinkage magnetic flux Ψa, the d-axis inductance Ld, and the q-axis inductance Lq. The following assumptions are made for these parameters.

Assumption 1: The dq-axis inductance difference Lq-Ld is constant with respect to the dq-axis current.
Assumption 2: Permanent magnet interlinkage flux Ψa can be approximated as a quadratic function of the q-axis current iq.
Ψa ＝ a _Ψ iq ² ＋ b _Ψ iq ＋ c _Ψ (iq> 0) ……… (1)
In the present embodiment, when considering the parameter fluctuation due to magnetic saturation, it is assumed that the salient pole ratio, which is the ratio of the dq-axis inductance, is relatively small, and that the d-axis inductance and the q-axis inductance also fluctuate due to magnetic saturation. As in Assumption 1 above, it is assumed that the dq-axis inductance difference Lq-Ld is constant with respect to the dq-axis current.

The torque T of the synchronous motor can be expressed by the following torque formula.
T = P {Ψa + (Ld-Lq) id} iq
= P {a _Ψ iq ² + b _Ψ iq + c _Ψ + (Ld-Lq) id} iq ……… (2)
Here, P is the number of pole pairs, Ψa is the permanent magnet interlinkage magnetic flux (magnetic flux interlinking the coil of the motor from the permanent magnet), Ld is the d-axis inductance, Lq is the q-axis inductance, id is the d-axis current, iq. Is the q-axis current.
In the above torque equation, assuming that the torque is constant (thinking on the constant torque curve), solving the above equation (2) with respect to the d-axis current id gives the following equation (3).

When the d-axis current id represented by the above equation (3) is differentiated by the q-axis current iq, the following equation (4) is obtained.

Then, by substituting the torque equation represented by the above equation (2) into the torque T of the above equation (4), the following equation (5) is obtained.

The did / diq represented by the above equation (5) is a value determined by the dq-axis current id and iq, and is from the q-axis of the tangent line C of the constant torque curve A at the point B on the constant torque curve A shown in FIG. It is the inclination that I saw. Since this tangent line C is a straight line in the f-axis direction, the angle δ'from the q-axis of the f-axis is a value (tan ^-1 (did / diq)) obtained by taking the inverse tangent of did / diq.
Therefore, the declination δ from the dq axis of the ft axis can be calculated by subtracting the angle δ'from π / 2.

As described above, the declination calculation unit 304a uses the above equation (5) obtained by differentiating the d-axis current id by the q-axis current iq, which is derived from the torque equation of the motor 310 represented by the above equation (2). , Π / 2 minus the declination δ to calculate the angle δ'. Then, the declination calculation unit 304a calculates the declination δ by subtracting the calculated angle δ'from π / 2, as shown in the above equation (6).
By using the declination δ calculated by the above equation (6) and performing the rotation coordinate conversion by the declination δ, it is possible to convert from the dq coordinate to the maximum torque control coordinate (ft coordinate) or vice versa. Become.

However, the above declination δ derivation formula is a formula that holds only when the q-axis current iq is in the positive region. Therefore, in order to extend the q-axis current iq to the case where it is 0 or in the negative region, the declination δ derivation formula is rewritten as follows.
(1) In order to avoid division by q-axis current iq = 0, the q-axis current iq in the denominator portion of the above equation (5) is replaced with iq + ε. Here, ε is a sufficiently small positive number (eg, 1 × 10 ^-6 ).
(2) Even in the region where the q-axis current iq is negative, since only the magnitude of the q-axis current is referred to for magnetic saturation, the absolute value of the q-axis current iq is taken and the sign of the declination δ is the sign of the q-axis current iq. Same as the code. That is, the above equation (6) is rewritten as the following equation.

Here, abs () is a function that takes the absolute value of the value in parentheses, and sgn () is a function that returns one of -1, 0, and 1 depending on the sign of the value in parentheses.
As described above, the maximum torque control coordinate system can be used for the q-axis current iq of the entire real number. However, in practice, there is a limit on the maximum system current.

As described above, the motor control device 320 in the present embodiment easily controls the maximum torque of the motor 310 by controlling the f-axis current to 0 by using the maximum torque control coordinate system (ft coordinate system). It can be realized. As shown in FIG. 5, the operating point at the time of the maximum torque control is the point E where the constant torque curve A and the constant current circle D meet. In FIG. 5, the constant current circle D is a circle drawn around the origin, and the operating point E is the point where the distance from the origin is the shortest with respect to the constant torque curve A.
Further, when the motor control device 320 uses the maximum torque control coordinate system, the permanent magnet interlinkage magnetic flux Ψa and the dq axis inductances Ld and Lq are used as motor parameters, but it is considered that these parameters fluctuate due to magnetic saturation. Then, the deviation angle δ of the maximum torque control coordinate system is calculated. Therefore, it is possible to realize an appropriate maximum torque control in consideration of magnetic saturation.

Specifically, in the motor control device 320, the dq-axis inductance difference Lq-Ld is constant with respect to the dq-axis current, and the permanent magnet interlinkage magnetic flux Ψa can be approximated as a quadratic function of the q-axis current iq. Calculate the angle δ.
Generally, it is said that the d-axis inductance Ld does not fluctuate much with respect to the dq-axis current due to magnetic saturation, and the q-axis inductance Lq decreases due to saturation. However, in a general motor, the dq-axis inductance is as high as several tens of mH to several hundreds of mH, and the reluctance torque, which is the ratio of the dq-axis inductance, is also positively greater than 2. It is often a motor used for. Further, the q-axis inductance fluctuation due to magnetic saturation is as large as several tens of mH, and the d-axis inductance fluctuation is less than 1/10 of that, so that it is ignored.

On the other hand, for example, as a motor for an electric power steering device, there is a motor having a dq-axis inductance as small as about 100 μH to 700 μH, a salient pole ratio as large as about 1.2, and a motor mainly composed of magnet torque. .. In such a motor, the inductance fluctuation of each of the d-axis and the q-axis due to magnetic saturation is as small as about 10 to 20 μH, and these fluctuations show substantially similar behavior. Therefore, as described above, it can be assumed that the dq-axis inductance difference Lq-Ld is constant with respect to the dq-axis current as the parameter fluctuation due to magnetic saturation. This makes it possible to simplify complicated calculations when calculating the declination δ.

Further, the motor control device 320 calculates the angle δ'using the above equation (5) obtained by differentiating the d-axis current id with the q-axis current iq, and as shown in the above equation (6), the calculated angle δ'. Is subtracted from π / 2 to calculate the declination δ. That is, in the present embodiment, the slope did / diq seen from the q-axis of the tangent line C (straight line in the f-axis direction) shown in FIG. 4 is calculated, and the declination δ is calculated based on this slope did / diq. Therefore, the declination can be calculated by a simple calculation.
When trying to directly calculate the slope diq / did seen from the d-axis of the tangent line C (straight line in the f-axis direction) shown in FIG. 4, the torque formula of the motor 310 represented by the above equation (2) is the q-axis current iq. Need to be solved. At this time, if the permanent magnet interlinkage magnetic flux Ψa is considered as a quadratic function of the q-axis current iq, it is necessary to solve a cubic equation, which complicates the calculation.
On the other hand, in the present embodiment, even when the permanent magnet interlinkage magnetic flux Ψa is considered as a quadratic function of the q-axis current iq, the declination can be calculated accurately by a simple calculation.

Further, as described above, the motor control device 320 realizes driving with the maximum torque by controlling the f-axis current to 0 by using the maximum torque control coordinate system. Therefore, the t-axis current and the motor output torque at this time have a one-to-one correspondence. Therefore, converted from the torque command value tau ^* to t-axis current command value it ^*, and converted t-axis current command value it ^*, based on the f-axis current command value is set to 0 an if ^*, declination δ By performing the rotating coordinate conversion using the above, it is possible to generate the ^{dq axis current command values id *} and iq ^{* that can easily realize the maximum torque control.}
As described above, the motor control device 320 in the present embodiment can easily and appropriately realize the maximum torque control in consideration of the parameter fluctuation due to the magnetic saturation in the drive control of the permanent magnet synchronous motor.

(Second embodiment)
Next, a second embodiment of the present invention will be described.
In the first embodiment described above, a case where the permanent magnet interlinkage magnetic flux Ψa is approximated as a quadratic function of the q-axis current iq has been described. In this embodiment, a case where the permanent magnet interlinkage magnetic flux Ψa is approximated as a linear function of the q-axis current iq will be described.

The configuration of the motor control device in this embodiment is the same as that of the motor control device 320 of the first embodiment shown in FIGS. 1 to 3. However, the method of calculating the declination δ in the declination calculation unit 304a of FIG. 3 is different from that of the first embodiment.
In the declination calculation unit 304a, the assumption 2 in the first embodiment described above is changed as follows.
Assumption 2': Permanent magnet interlinkage flux Ψa can be approximated as a linear function of the q-axis current iq.
Ψa = b _Ψ iq + c _Ψ (iq> 0) ……… (8)
Then, the declination calculation unit 304a calculates the declination δ by the same method as in the case where _{a Ψ = 0 is set in the first embodiment described above.}

Thereby, also in the present embodiment, the maximum torque control of the motor 310 can be easily realized by using the maximum torque control coordinate system (ft coordinate system) as in the first embodiment described above. Further, since the parameter fluctuation due to magnetic saturation is taken into consideration when calculating the declination δ of the maximum torque control coordinate system, appropriate maximum torque control can be realized. In particular, in the present embodiment, since the permanent magnet interlinkage magnetic flux Ψa, which is the above parameter, is approximated as a linear function of the q-axis current iq, the declination δ can be calculated using a simpler derivation formula.

(Third embodiment)
Next, a third embodiment of the present invention will be described.
In the first embodiment and the second embodiment described above, the angle δ'from the q-axis of the f-axis is calculated by calculating did / diq obtained by differentiating the d-axis current id with the q-axis current iq, and the angle. The case of calculating the declination δ from δ'has been described. In the present embodiment, a case where the declination δ is directly calculated by calculating diq / did obtained by differentiating the q-axis current iq with the d-axis current id will be described.

The configuration of the motor control device in this embodiment is the same as that of the motor control device 320 of the first embodiment shown in FIGS. 1 to 3. However, the method of calculating the declination δ in the declination calculation unit 304a of FIG. 3 is different from that of the first embodiment.
The declination calculation unit 304a calculates the declination δ of the maximum torque control coordinate system under the same assumption as in the second embodiment described above.
Assumption 1: The dq-axis inductance difference Lq-Ld is constant with respect to the dq-axis current.
Assumption 2': Permanent magnet interlinkage flux Ψa can be approximated as a linear function of the q-axis current iq.
Ψa = b _Ψ iq + c _Ψ (iq> 0) ……… (9)

In this case, the torque equation can be expressed by the following equation.
T = P {Ψa + (Ld-Lq) id} iq
= P {b _Ψ iq + c _Ψ + (Ld-Lq) id} iq
b _Ψ iq ² + {c _Ψ + (Ld-Lq) id} iq-T / P = 0 ……… (10)
In the above torque equation, assuming that the torque is constant (thinking on the constant torque curve), solving the above equation (10) with respect to the q-axis current iq gives the following equation (11).

When the q-axis current iq is 0, the motor output torque is also 0, so the equation of the q-axis current iq is as follows.

When the q-axis current iq represented by the above equation (12) is differentiated by the d-axis current id, the following equation (13) is obtained.

Then, by substituting the torque equation represented by the equation (10) into the torque T of the equation (13), the following equation (14) is obtained.

The diq / did represented by the above equation (14) is an inclination seen from the d-axis of the tangent line C (straight line in the f-axis direction) of the constant torque curve A at the point B on the constant torque curve A shown in FIG. .. Therefore, the declination δ from the dq axis of the ft axis can be calculated by taking the inverse tangent of did / diq.

As described above, the declination calculation unit 304a uses the above equation (14) obtained by differentiating the q-axis current iq by the d-axis current id, which is derived from the torque equation of the motor 310 represented by the above equation (2). , The declination δ is calculated as shown in the above equation (15).
By using the declination δ calculated by the above equation (15) and performing the rotation coordinate conversion by the declination δ, it is possible to convert from the dq coordinate to the maximum torque control coordinate (ft coordinate) or vice versa. Become.

As a result, also in the present embodiment, the maximum torque control of the motor 310 can be easily performed by using the maximum torque control coordinate system (ft coordinate system) as in the first embodiment and the second embodiment described above. It can be realized. Further, since the parameter fluctuation due to magnetic saturation is taken into consideration when calculating the declination δ of the maximum torque control coordinate system, appropriate maximum torque control can be realized.
Further, in the present embodiment, the declination δ can be directly calculated by using the did / diq equation derived from the torque equation. At this time, by approximating the permanent magnet interlinkage magnetic flux Ψa, which is the above parameter, as a linear function of the q-axis current iq, the deviation angle δ can be calculated using a relatively simple derivation formula.

(Fourth Embodiment)
Next, a third embodiment of the present invention will be described with reference to FIG.
In the first to third embodiments described above, the fourth embodiment suppresses the torque ripple generated when the motor is driven.
The configuration of the motor control device in this embodiment is the same as that of the motor control device 320 shown in FIGS. 1 and 2. However, the configurations of the command generation unit 321 of FIG. 1 and the current command generation unit 304 of FIG. 2 are different from those of the above-described embodiments.

FIG. 6 is a block diagram showing the configuration of the command generation unit 321 of FIG. 1 in the present embodiment.
As shown in FIG. 6, the command generation unit 321 includes a torque ripple estimator 321a, a subtractor 321b, and a command value converter 321c.
The torque ripple estimator 321a estimates the torque ripple estimated value τr by inputting, for example, the motor angular velocity ω and the motor current i. The configuration of the torque ripple estimator 321a is not limited to the above, and any method may be used as long as the torque ripple can be estimated.

Subtractor 321b is a torque command value tau ^* from the torque ripple estimate τr given by subtracting, to output a torque command value after correction tau ^* '. As described above, in the present embodiment, the torque ripple estimated value τr is used as the torque ripple correction value for correcting the ^{torque command value τ *.}
The command value converter 321c converts the corrected torque command value τ ^* 'to the current command value i ^* and outputs it.
In FIG. 6, the torque ripple estimator 321a corresponds to the torque ripple estimation unit, and the subtractor 321b corresponds to the torque command correction unit.

FIG. 7 is a functional block diagram showing a specific configuration of the current command generation unit 304 of FIG. 2 in the present embodiment. In FIG. 7, the portions having the same configuration as that of FIG. 3 are designated by the same reference numerals as those in FIG. 3, and the portions having different configurations will be mainly described below.
The torque command correction unit 304d corrects the torque command value τ ^* with the torque ripple correction value τr, ^{and outputs the corrected torque command value τ *} 'to the t-axis current command generation unit 304b. Here, the torque ripple correction value τr is a torque ripple estimated value estimated by the torque ripple estimator 321a of FIG. 6 described above.

The t-axis current command generation unit 304b generates the t-axis current command value it ^{* based on the torque command value τ *} 'corrected by the torque ripple correction value τr. t-axis current command value it ^* generation method is the same as the method of generating the torque command value tau ^* from t-axis current command value it ^* in the above embodiments.
In this way, the motor 310 is driven based on ^{the corrected torque command value τ *} 'corrected by the torque ripple correction value (torque ripple estimated value) τr. As a result, the torque ripple generated when the motor 310 is driven can be appropriately suppressed. Further, by suppressing the torque ripple, it is possible to reduce the noise caused by the torque ripple. Therefore, the quietness of the motor system 300 can be improved.

As described above, the motor control device 320 in the present embodiment easily realizes the maximum torque control of the motor 310 by using the maximum torque control coordinate system (ft coordinate system) as in each of the above-described embodiments. be able to. Since the maximum torque control is realized by controlling the f-axis current to 0 using the maximum torque control coordinate system, the t-axis current and the motor output torque at this time have a one-to-one correspondence.
Therefore, by correcting the torque command value tau ^* by superimposing the torque ripple correction value τr to the torque command value tau ^*, to generate a t-axis current command value correction is reflected in the torque ripple it ^* It is possible to suppress torque ripple appropriately. Further, since the torque ripple estimated value obtained by some method can be used as it is for the torque ripple correction value τr for reducing the torque ripple, the torque ripple can be easily suppressed.

(Application example)
The motor system 300 in each of the above-described embodiments can be applied to the electric power steering device 10 mounted on the vehicle 1, for example, as shown in FIG.
FIG. 9 is a schematic view of the electric power steering device 10.
As shown in FIG. 9, the electric power steering device 10 is a column type EPS (electric power steering device), and the steering wheel 11, the steering shaft 12, and the steering assist mechanism 13 are arranged in the order in which the force given by the driver is transmitted. It includes a universal joint 14a, an intermediate shaft 15, a universal joint 14b, a pinion shaft 16, a steering gear 17, tie rods 18a and 18b, hub units 19a and 19b, and steering wheels 20L and 20R.

The steering shaft 12 includes an input shaft 12a and an output shaft 12b. One end of the input shaft 12a is connected to the steering wheel 11, and the other end of the input shaft 12a is connected to one end of the output shaft 12b via the torque sensor 31. The other end of the output shaft 12b is connected to a universal joint 14a. A torsion bar (not shown) is interposed between the input shaft 12a and the output shaft 12b.

The steering force acted on by the driver is transmitted from the steering wheel 11 to the steering shaft 12 having the input shaft 12a and the output shaft 12b. Then, the steering force transmitted to the output shaft 12b is transmitted to the intermediate shaft 15 via the universal joint 14a, and further transmitted to the pinion shaft 16 via the universal joint 14b. The steering force transmitted to the pinion shaft 16 is transmitted to the tie rods 18a and 18b via the steering gear 17, respectively, and steers the steering wheels 20L and 20R via the hub units 19a and 19b, respectively.
Here, the steering gear 17 includes a pinion 17a connected to the pinion shaft 16 and a rack 17b that meshes with the pinion 17a, and the rotational motion transmitted to the pinion 17a is converted into a straight motion by the rack 17b. There is.

The steering assist mechanism 13 is connected to the output shaft 12b of the steering shaft 12 and transmits the steering assist force to the output shaft 12b. The steering assist mechanism 13 includes a reduction gear 21 connected to the output shaft 12b, and an electric motor 22 connected to the reduction gear 21 to generate an auxiliary steering force with respect to the steering system.
Here, the electric motor 22 can be, for example, a three-phase brushless motor. Further, the reduction gear 21 can be, for example, a worm gear. In this case, the torque generated by the electric motor 22 is transmitted to the worm wheel via the worm inside the reduction gear 21 to rotate the worm wheel. As a result, the reduction gear 21 increases the torque generated by the electric motor 22 and applies the auxiliary steering torque to the output shaft 12b.

The torque sensor 31 is for detecting the steering torque applied to the steering wheel 11 and transmitted to the input shaft 12a, and is a relative displacement between the input shaft 12a and the output shaft 12b connected by a torsion bar (not shown). (Rotary displacement) is detected in correspondence with the change in the impedance of the coil pair. The torque detection value output from the torque sensor 31 is input to the control unit (ECU) 30.
The ECU 30 operates by being supplied with power from the battery 38, which is an in-vehicle power source. The negative electrode of the battery 38 is grounded, and the positive electrode thereof is connected to the ECU 30 via the ignition switch 39 that starts the engine, and is directly connected to the ECU 30 without the ignition switch 39.

The ECU 30 includes a CPU that realizes the functions of the motor control device 320 described above by executing a predetermined program.
The ECU 30 acquires the vehicle speed detection value detected by the vehicle speed sensor 32 in addition to the torque detection value detected by the torque sensor 31. Then, the ECU 30 performs steering assist control (steering assist) that applies steering assist force corresponding to these to the steering system. Specifically, the ECU 30 calculates a steering assist command value (steering assist torque command value) based on the torque detection value and the vehicle speed detection value. Then, the ECU 30 controls the drive current supplied to the electric motor 22 based on the calculated steering auxiliary torque command value.
Here, the ECU 30 uses the steering auxiliary torque command value as the torque command value τ ^* in FIG. 2 and the electric motor 22 as the motor 310 in FIG. 2, and has the same function as the motor control device 320 shown in FIG. It can be realized.

Therefore, it is possible to easily and appropriately realize the maximum torque control of the in-vehicle permanent magnet synchronous motor for performing steering assist control (steering assist). Therefore, it is possible to obtain an electric power steering device in which high efficiency is realized.
Further, when the current command generation unit 304 of FIG. 2 has the configuration shown in FIG. 7, as in the third embodiment described above, torque ripple during driving of the electric motor 22 can be appropriately suppressed. ..

In the case of the column type EPS as shown in FIG. 9, since the electric motor 22 is close to the driver (steering wheel 11), the vibration of the electric motor 22 is easily transmitted to the driver through the mechanical element. In addition, noise is generated due to the vibration of the electric motor 22, which is transmitted to the passenger compartment and causes discomfort to the passengers. If the torque ripple of the electric motor 22 can be appropriately suppressed as described above, the noise caused by the torque ripple can be reduced, and the quietness in the vehicle interior can be improved.

(Modification example)
In each of the above embodiments, the electric power steering device 10 is shown as an application example of the motor system 300, but the present invention is not limited to the above. The motor system 300 converts, for example, an electric brake (brake-by-wire) that converts a brake pedal operation into an electric signal and differentials a friction brake via an actuator, or converts a steering operation into an electric signal and uses an actuator. It can also be applied to steering by wire that steers wheels. The motor to be controlled is not limited to the in-vehicle motor as described above.

In the present invention, a program that realizes one or more functions of the above-described embodiment is supplied to a system or device via a network or a storage medium, and one or more processors in the computer of the system or device program the program. It can also be realized by reading and executing. In this case, the program itself read from the recording medium realizes the function of the embodiment. In addition, the recording medium on which the program is recorded can constitute the present invention.
Further, by executing the program read by the computer, not only the function of the embodiment is realized, but also the operating system (OS) running on the computer is one of the actual processes based on the instruction of the program. The function of the above-described embodiment may be realized by performing a part or all of the processing.

10 ... Electric power steering device, 300 ... Motor system, 304 ... Current command generation unit, 304a ... Deviation angle calculation unit, 304b ... t-axis current command generation unit, 304c ... f-axis current command generation unit, 304d ... dq-axis current command Generation unit, 304e ... Torque command correction unit, 320 ... Motor control device

Claims (10)

A motor control device that drives and controls a permanent magnet synchronous motor.
A deviation that calculates the deviation angle between the dq coordinate system, which is a rotating coordinate system, and the ft coordinate system, which is a maximum torque control coordinate system, based on the dq-axis current of the permanent magnet synchronous motor and parameters that can fluctuate due to magnetic saturation. Angle calculation unit and
A t-axis current command generator that generates a t-axis current command value in the ft coordinate system based on the torque command value of the permanent magnet synchronous motor.
An f-axis current command generator that generates the f-axis current command value in the ft coordinate system as zero, and an f-axis current command generator.
The deviation angle calculated by the deviation angle calculation unit, the t-axis current command value generated by the t-axis current command generation unit, and the f-axis current command value generated by the f-axis current command generation unit. Based on this, a motor control device including a dq-axis current command generator that generates a dq-axis current command value by rotational coordinate conversion. The motor control device according to claim 1, wherein the parameters are a permanent magnet interlinkage magnetic flux, a d-axis inductance, and a q-axis inductance. The declination calculation unit
A claim characterized in that the difference between the d-axis inductance and the q-axis inductance is made constant with respect to the dq-axis current, and the deviation angle is calculated by using the permanent magnet interlinkage magnetic flux as a quadratic function of the q-axis current. Item 2. The motor control device according to Item 2. The declination calculation unit
A claim characterized in that the difference between the d-axis inductance and the q-axis inductance is made constant with respect to the dq-axis current, and the deviation angle is calculated by using the permanent magnet interlinkage magnetic flux as a linear function of the q-axis current. Item 2. The motor control device according to Item 2. The declination calculation unit
The angle obtained by subtracting the deviation angle from π / 2 is calculated according to the equation obtained by differentiating the d-axis current with the q-axis current derived from the torque equation of the permanent magnet synchronous motor, and the deviation angle is calculated from the angle. The motor control device according to any one of claims 1 to 4, wherein the motor control device is characterized. The declination calculation unit
The invention according to any one of claims 1 to 4, wherein the declination is calculated according to an equation obtained by differentiating the q-axis current with the d-axis current, which is derived from the torque equation of the permanent magnet synchronous motor. Motor control device. A torque ripple estimation unit that estimates the torque ripple of the permanent magnet synchronous motor,
A torque command correction unit that corrects the torque command value based on the torque ripple estimated by the torque ripple estimation unit is further provided.
The t-axis current command generator
The motor control device according to any one of claims 1 to 6, wherein the t-axis current command value is generated based on the torque command value corrected by the torque command correction unit. It is a motor control method that drives and controls a permanent magnet synchronous motor.
A step of calculating the deviation angle between the dq coordinate system, which is a rotating coordinate system, and the ft coordinate system, which is a maximum torque control coordinate system, based on the dq-axis current of the permanent magnet synchronous motor and parameters that can fluctuate due to magnetic saturation. When,
A step of generating a t-axis current command value in the ft coordinate system based on the torque command value of the permanent magnet synchronous motor, and
The step of generating the f-axis current command value in the ft coordinate system as zero, and
A motor control method including a step of generating a dq-axis current command value by rotational coordinate conversion based on the declination angle, the t-axis current command value, and the f-axis current command value. .. An electric power steering device comprising the motor control device according to any one of claims 1 to 7. A program for causing a computer to function as each part of the motor control device according to any one of claims 1 to 7.

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