JP2020134147A - Radar device and target distance detection method - Google Patents

Abstract

To enhance accuracy of target distance detection in a radar device.SOLUTION: A radar device includes: transmission sections (111, 112, 21) which transmit a reference chirp by a combination of increase and decrease in frequency as a transmission signal repeatedly; reception sections (31, 33, 34) which receive a reflection wave by a target of the transmission signal as a reception signal; and an inverted chirp generation section (115) which generates an inverted chirp (Sr) obtained by inverting the reference chirp on a time base. The radar device detects a distance between the radar device and the target on the basis of a result of mixing the reception signal and the inverted chirp.SELECTED DRAWING: Figure 6

Description

The present invention relates to a radar device and a target distance detection method.

In the radar device, a chirp signal whose frequency fluctuates periodically is transmitted toward the target, and a beat signal is obtained by mixing the received signal of the reflected wave by the target with the transmission signal. The beat signal has a beat frequency indicating a frequency difference between the transmitted signal and the received signal. This beat frequency depends on the distance between the radar device and the target. Therefore, it is possible to generate the frequency spectrum of the beat signal and obtain the distance between the radar device and the target from the frequency having the peak in the frequency spectrum (see, for example, Patent Document 1 below).

JP-A-2015-210157

Ideally, the frequency difference between the transmitted signal and the received signal becomes constant, and a frequency spectrum having power only at a frequency corresponding to the distance between the radar device and the target is obtained. However, in reality, various error factors including signal linearity error are entangled, and a peak waveform having a corresponding spread is generated in the frequency spectrum. Therefore, the detection accuracy when detecting the distance depending on the peak of the frequency spectrum may not be sufficiently high.

An object of the present invention is to provide a radar device and a target distance detection method that contribute to improving the accuracy of target distance detection.

The radar device according to the present invention includes a transmission unit that transmits a reference chirp having a section in which the frequency changes monotonically as a transmission signal, a reception unit that receives a reflected wave from a target of the transmission signal as a reception signal, and the reference chirp. It has an inverted chirp generator that generates an inverted chirp that is inverted on the time axis, and a ranging unit that detects the distance between the radar device and the target based on the mixing result of the received signal and the inverted chirp. It is a configuration (first configuration) including a signal processing unit.

In the radar device according to the first configuration, the receiving unit has a mixer, and by inputting the received signal and the inverted chirp to the mixer, a mixing result between the received signal and the inverted chirp can be obtained. It may be the configuration (second configuration).

In the radar device according to the second configuration, the mixer is used to generate a beat signal indicating the mixing result of the received signal and the inverted chirp, and the distance measuring unit is a chirp section to which the reference chirp is transmitted. On the other hand, a conversion target section including the center of the chirp section is set, and the frequency spectrum of the target signal is generated by converting the signal in the conversion target section of the beat signal into a signal on the frequency region as the target signal. It may have a configuration (third configuration) including a spectrum generation unit and a distance derivation unit that derives the distance based on the frequency spectrum of the target signal.

In the radar device according to the third configuration, the distance derivation unit has a configuration (fourth configuration) for deriving the distance based on the maximum frequency having a power equal to or higher than a predetermined value in the frequency spectrum. Is also good.

In the radar device according to the second configuration, the mixer is used to generate a beat signal indicating the mixing result of the received signal and the inverted chirp, and the distance measuring unit is a chirp section to which the reference chirp is transmitted. By setting a plurality of conversion target sections and converting the signal in the conversion target section of the beat signal into a signal on the frequency region for each conversion target section, the target is converted for each conversion target section. Even in a configuration (fifth configuration) having a spectrum generation unit that generates a frequency spectrum of a signal and a distance derivation unit that derives the distance based on the frequency spectrum generated for each conversion target section. good.

In the radar device according to the fifth configuration, the distance derivation unit extracts the maximum frequency having a power equal to or higher than a predetermined value in the frequency spectrum for each conversion target section, and sets the extraction frequency for each conversion target section. The configuration may be such that the distance is derived based on the above (sixth configuration).

In the radar device according to the fifth configuration, the distance derivation unit extracts the minimum frequency having a power equal to or higher than a predetermined value in the frequency spectrum for each conversion target section, and sets the extraction frequency for each conversion target section. The configuration may be such that the distance is derived based on the above (seventh configuration).

In the radar device according to any one of the second to seventh configurations, the operation mode of the radar device is set by the first mode in which the received signal and the reference chirp are mixed by the mixer, or by the mixer. A configuration (eighth configuration) may further include a mode setting unit capable of switching to and setting a second mode for mixing the received signal and the inverted chirp.

In the radar device according to the eighth configuration, the reference chirp includes a first section in which the frequency increases and a second section in which the frequency decreases, and in at least one of the first section and the second section, the reference chirp is described. When the frequency of the reference chirp changes monotonically, the transmission unit repeatedly transmits the reference chirp as the transmission signal, and the operation mode is set to the second mode, the reception signal and the reception signal are transmitted to the mixer. An inverted chirp is input, and the distance is detected by the ranging unit based on the mixing result of the received signal obtained from the mixer and the inverted chirp, and the operation mode is set to the first mode. At that time, the received signal and the reference chirp are input to the mixer, and the target related to the target by the signal processing unit based on the mixing result of the received signal obtained from the mixer and the reference chirp. The target data for which data is generated and generated in the first mode may have a configuration (nineth configuration) including information other than the distance.

The target distance detection method according to the present invention is a target distance detection method used in a radar device, and includes a transmission step of transmitting a reference chirp having a section where a frequency changes monotonically as a transmission signal, and a transmission signal of the transmission signal. Based on the reception process of receiving the reflected wave from the target as a reception signal, the inversion chirp generation step of generating the inverted chirp in which the reference chirp is inverted on the time axis, and the mixing result of the received signal and the inverted chirp. It is a configuration (tenth configuration) including a distance measuring step for detecting the distance between the radar device and the target.

According to the present invention, it is possible to provide a radar device and a target distance detection method that contribute to improving the accuracy of target distance detection.

It is a schematic block diagram of the radar apparatus which concerns on embodiment of this invention. FIG. 5 is a waveform diagram of a transmission signal by a radar device according to an embodiment of the present invention. FIG. 5 is a modified waveform diagram of a transmission signal by a radar device according to an embodiment of the present invention. It is an internal block diagram of the signal processing unit of FIG. It is explanatory drawing of the azimuth angle of a target. FIG. 5 is a configuration diagram of a transmission / reception block of a radar device according to the first embodiment of the present invention. It is a figure which shows the reference chirp and the inversion chirp according to 1st Example of this invention. It is a figure which shows the relationship of the reference chirp, the inversion chirp and the reception chirp according to 1st Example of this invention. FIG. 5 is a diagram showing a frequency spectrum of a beat signal generated based on an inverted chirp and a received chirp according to the first embodiment of the present invention. FIG. 5 is a diagram showing a frequency spectrum of a beat signal generated based on a reference chirp and a received chirp according to the first embodiment of the present invention. It is a figure which shows the relationship of the reference chirp, the inversion chirp and the reception chirp according to the 2nd Example of this invention. FIG. 5 is a diagram showing a frequency spectrum of a beat signal generated based on an inverted chirp and a received chirp under a zero distance situation according to a second embodiment of the present invention. FIG. 5 is a diagram showing a frequency spectrum of a beat signal generated based on an inverted chirp and a received chirp under an actual measurement situation according to a second embodiment of the present invention. It is a figure for demonstrating the derivation method of the distance of a target, which concerns on 2nd Example of this invention. It is a figure which shows the example of the relationship between the 1st FFT section and the 2nd FFT section which concerns on 2nd Example of this invention. It is a figure which shows the appearance that the intersection between the receiving chirp and the inverted chirp is not included in both the 1st and 2nd FFT sections in relation to the modified structure of 2nd Example of this invention. FIG. 5 is a diagram showing a frequency spectrum of a beat signal generated based on an inverted chirp and a received chirp under an actual measurement situation according to a modified configuration of the second embodiment of the present invention. FIG. 3 is a configuration diagram of a transmission / reception block of a radar device according to a third embodiment of the present invention. It is a block diagram of the modulation signal generation part of FIG. FIG. 5 is a configuration diagram of a transmission / reception block of a radar device according to a fourth embodiment of the present invention.

Hereinafter, examples of embodiments of the present invention will be specifically described with reference to the drawings. In each of the referenced figures, the same parts are designated by the same reference numerals, and duplicate explanations regarding the same parts will be omitted in principle. In this specification, for the sake of simplification of description, by describing a symbol or a code that refers to an information, a signal, a physical quantity, a member, etc., the name of the information, a signal, a physical quantity, a member, etc. corresponding to the symbol or the code is given. It may be omitted or abbreviated.

FIG. 1 is a diagram showing a schematic configuration of a radar device 1 according to an embodiment of the present invention. The radar device 1 can be mounted on any device, and the radar device 1 may be installed at a predetermined position of a vehicle such as an automobile. In the following, it is assumed that the radar device 1 is installed at a predetermined position of the vehicle, and the vehicle on which the radar device 1 is installed is referred to as a own vehicle.

A predetermined transmitted wave is transmitted from the transmitting antenna of the radar device 1, and the transmitted wave is reflected by a target existing around the own vehicle. The reflected wave due to this reflection is received by the receiving antenna of the radar device 1, and the radar device 1 can acquire the target data related to the target based on the received signal by the receiving antenna. The target existing around the own vehicle is, for example, a vehicle different from the own vehicle, a human being, or an installation on the road surface. The target data includes a plurality of parameters, and the plurality of parameters include the distance between the own vehicle and the target, the relative speed of the target with respect to the own vehicle, and the distance between the own vehicle and the target along the front-rear direction of the own vehicle. The distance, the distance between the own vehicle and the target along the left-right direction of the own vehicle, and the like may be included. Since the radar device 1 is fixed to the own vehicle, it can be said that the distance between the own vehicle and the target represents the distance between the radar device 1 and the target. The same applies to the relative speed of the target with respect to the own vehicle.

As shown in FIG. 1, the radar device 1 mainly includes a transmitting unit 2, a receiving unit 3, a chirp processing unit 4, and a signal processing unit 5. Here, a configuration using FMCW (Frequency Modulated Continuous Wave), which is a frequency-modulated continuous wave, will be described as an example. In general, the FMCM method may refer to a method of performing triangular wave-shaped frequency modulation in which an up-chirp section and a down-chirp section that are symmetrical to each other are combined, but once, here, simply frequency-modulated continuous. It will be described as referring to a method using waves.

The transmission unit 2 includes a transmission antenna 21 and a transmission processing unit 22. The transmission processing unit 22 generates a predetermined chirp signal as a transmission signal under the control of the signal processing unit 5. A chirp signal is a signal whose frequency increases or decreases with the passage of time. For example, the transmission processing unit 22 generates a modulated signal whose voltage changes in a sawtooth shape under the control of the signal processing unit 5, and frequency-modulates the continuous wave signal with the modulated signal, so that the frequency changes over time. Generates a changing chirp signal as a transmission signal. The transmission signal generated by the transmission processing unit 22 is sent to the transmission antenna 21. The transmission antenna 21 outputs the transmission signal from the transmission processing unit 22 as a transmission wave TW in a predetermined direction (that is, radiates it into space). The transmitted wave TW output from the transmitting antenna 21 is an FMCW whose frequency fluctuates in a predetermined cycle. When the transmitted wave TW is reflected by the target, the reflected wave RW based on the transmitted wave TW is directed from the target to the own vehicle. In the present embodiment, various chirp signals may be referred to as chirps, but chirp signals and chirps are synonymous with each other.

FIG. 2 shows an example of the waveform of the transmission signal by the transmission unit 2. Here, the reference chirp signal is referred to as a reference chirp for convenience. The transmitted signal from the transmitting antenna 21 consists of repeating reference chirps. The frequency of the reference chirp changes (increases or decreases) with the passage of time, and the cycle of the change is called the chirp cycle, and the section having the time length of one chirp cycle in the transmission signal is called the chirp section. The reference chirp has a period in which the frequency changes monotonically within the chirp period. That is, the reference chirp contains at least one chirp signal within the chirp period. In any two chirp sections that differ from each other, the frequency of the reference chirp changes in the same way. From the transmission unit 2, a reference chirp composed of a combination of frequency increase and decrease is repeatedly transmitted as a transmission signal toward the target. The chirp period is, for example, 10 microseconds to several milliseconds. The chirp cycle should be determined as appropriate. As will be described later, when switching between the normal mode and the high precision mode, it is preferable to match the chirp cycle of the high precision mode with the chirp cycle of the normal mode.

The characteristics of the reference chirp shown in FIG. 2 will be described. In the waveform of FIG. 2, the reference chirp has two chirp signals. Specifically, in the waveform of FIG. 2, the reference chirp is a first chirp signal that increases with a gradient θ ₁ from the frequency f _L to a frequency f _H, and a first chirp signal that decreases with a gradient θ ₂ from the frequency f _H to the frequency f _L. It has two chirp signals. In each chirp interval according to the waveform example of Fig. 2, the frequency of the reference chirp at the start timing of the chirp period is a predetermined frequency f _L, and the start timing as a starting point, the frequency of the reference chirp predetermined frequency f _L With the passage of time, the slope increases with θ ₁ . When the frequency of the reference chirp reaches a predetermined frequency f _H , the frequency of the reference chirp in turn decreases with a slope θ ₂ with the passage of time. Then, the frequency of the reference chirp finally returns to the predetermined frequency f _L. The frequencies f _L and f _H are arbitrary frequencies different from each other, and the frequency f _H is higher than the frequency f _L. In the following, for the sake of simplification and materialization of the description, a case where the center frequency is 76.5 GHz (gigahertz) and ± 0.5 GHz, that is, the modulation width is 1 GHz will be described as an example. That is, it is assumed that the frequencies f _L and f _H are 76.0 GHz and 77.0 GHz, respectively. The center frequency and modulation width may be appropriately set based on the target band, applicable regulations, characteristics of the amplifier, design likelihood, and the like.

Incidentally, as shown in FIG. 3 (a), the timing of frequency of the reference chirp is consistent with the predetermined frequency f _H may be set to the start timing of the chirp period. FIG. 3A shows an example of a first modified waveform of the transmitted signal. In the first modified waveform example, in each chirp section, the frequency of the reference chirp first decreases from the frequency f _H with a slope θ ₂ with the passage of time, reaches the frequency f _L , and then with the passage of time. It increases with the slope θ ₁ and returns to the frequency f _H.

The slope θ ₁ represents the rate of increase of the frequency of the reference chirp with respect to time. The slope θ ₂ represents the rate of decrease of the frequency of the reference chirp with respect to time. The specific values of the slopes θ ₁ and θ ₂ are arbitrary. On a graph in which time is taken on the horizontal axis and frequency is taken on the vertical axis, _one of the slopes θ ₁ and θ ₂ may be 90 degrees.

That is, for example, as shown in FIG. 3 (b), in each chirp section, the frequency of the reference chirp increases instantaneously from the frequency f _L to the frequency f _H (that is, increases with a slope θ _{1 of} 90 degrees), and then. , The frequency of the reference chirp may decrease with a slope θ ₂ with the passage of time from the frequency f _H to the frequency f _L. FIG. 3B shows an example of a second modified waveform of the transmitted signal. Alternatively, for example, as shown in FIG. 3C, in each chirp section, the frequency of the reference chirp instantly decreases from the frequency f _H to the frequency f _L , and then the frequency of the reference chirp changes from the frequency f _L to the frequency f _H. It is also possible to increase the inclination θ ₁ with the passage of time. FIG. 3C shows an example of a third modified waveform of the transmitted signal.

In a waveform in which the slope θ ₁ or θ ₂ is 90 degrees, a semi-discontinuous signal is input to the modulator in the section where the slope (θ ₁ , θ ₂ ) is 90 degrees. The actual signal in the section often has a waveform that can no longer be called a chirp signal. That is, it is preferable to consider that the transmission signal as shown in FIG. 3 (b) or (c) is composed of a reference chirp containing only one chirp signal in each chirp section.

In the following, unless otherwise specified, it is assumed that the reference chirp of the aspect shown in FIG. 2 is transmitted as a transmission signal, and the slope θ ₁ is larger than the slope θ ₂ .

The receiving unit 3 includes a plurality of receiving antennas 31 forming an array antenna, and a plurality of receiving processing units 32 connected to the plurality of receiving antennas 31. One reception processing unit 32 is individually assigned to each reception antenna 31. Here, as an example, it is assumed that four receiving blocks including one receiving antenna 31 and one receiving processing unit 32 are provided in the receiving unit 3. For each reception block, the reception antenna 31 receives the reflected wave RW from the target to acquire the reception signal, and the reception processing unit 32 performs predetermined processing on the reception signal obtained by the corresponding reception antenna 31.

The configurations of the plurality of reception processing units 32 are the same as each other, and each reception processing unit 32 includes a mixer 33 and an A / D converter 34. In each reception processing unit 32, the received signal obtained by the corresponding receiving antenna 31 is amplified by a low noise amplifier (not shown) and then sent to the mixer 33, and the mixer 33 supplies the received signal via the low noise amplifier. And the mixing signal provided by the chirp processing unit 4 are mixed to generate a beat signal. The beat signal has a beat frequency which is the difference between the frequency of the received signal and the frequency of the mixing signal. In each reception processing unit 32, the beat signal generated by the mixer 33 is converted into a digital signal by the A / D converter 34 after adjusting the timing between the reception antennas 31 by a synchronization circuit (not shown). , Is output to the signal processing unit 5.

The chirp processing unit 4 generates the above mixing signal based on the transmission signal generated by the transmission processing unit 22, and sends the mixing signal to the mixer 33 of each reception processing unit 32. The mixing signal has a form of a chirp signal, and hereinafter, the mixing signal may be referred to as a mixing chirp. Although the details will be described later, the mixing chirp may be an inverted chirp obtained by inverting the reference chirp on the time axis. When an inverted chirp is generated as a mixing chirp, the chirp processing unit 4 functions as an inverted chirp generation unit.

The signal processing unit 5 includes a microcomputer including a CPU (Central Processing Unit) and a memory, and is based on the signals supplied from each A / D converter 34 (hence, based on the received signals of the four receiving antennas 31). Target data can be generated.

FIG. 4 shows an internal block diagram of the signal processing unit 5. The signal processing unit 5 includes a Fourier transform unit 51 as an example of a spectrum generation unit, a target data generation unit 52, and a mode setting unit 56. The functions realized by the Fourier transform unit 51, the target data generation unit 52, and the mode setting unit 56 are described by a program, the program is stored in a memory that can be mounted on the signal processing unit 5, and the signal processing unit is stored. By executing the program on the microcomputer in 5, the functions of the Fourier transform unit 51, the target data generation unit 52, and the mode setting unit 56 may be realized. The configuration shown in FIG. 4 is a configuration for realizing a part of the functions of the signal processing unit 5, and a processing unit (not shown) for realizing other functions is further provided in the signal processing unit 5. It's okay.

The Fourier transform unit 51 generates a frequency spectrum of the beat signal by converting the beat signal expressed in the digital signal format supplied from each A / D converter 34 into a signal on the frequency domain. A fast Fourier transform may be used as this transform. Since the beat signals are individually generated by the four reception processing units 32, the four beat signals as a whole are sent from the reception unit 3 to the Fourier transform unit 51. The Fourier transform unit 51 generates a frequency spectrum for each beat signal supplied. The generated frequency spectrum is sent to the target data generation unit 52.

The target data generation unit 52 includes a distance calculation unit 53, an azimuth calculation unit 54, and a relative speed calculation unit 55. The distance calculation unit 53 functions as a distance derivation unit that derives the distance between the target and the radar device 1 based on the frequency spectrum provided by the Fourier transform unit 51. The distance between the target and the radar device 1 is hereinafter referred to as the distance d of the target or simply the distance d. Since the radar device 1 is fixed to the own vehicle, it can be said that the distance d of the target represents the distance between the own vehicle and the target. The azimuth calculation unit 54 functions as an azimuth derivation unit that derives the azimuth angle φ of the target based on the frequency spectrum provided by the Fourier transform unit 51. With reference to FIG. 5, the azimuth angle φ of the target is a coordinate plane parallel to the road surface on which the own vehicle travels, and the origin is on a two-dimensional coordinate plane with the installation position of the radar 1 as the origin. Represents the angle formed by a predetermined axis passing through the above and a line segment connecting the radar device 1 and the target. The predetermined axis is, for example, an axis parallel to the traveling direction of the own vehicle when the own vehicle goes straight. The relative velocity calculation unit 55 functions as a relative velocity derivation unit that derives the relative velocity v of the target based on the frequency spectrum provided by the Fourier transform unit 51. The relative speed v of the target represents the relative speed of the target with respect to the radar device 1, and also represents the relative speed of the target with respect to the own vehicle.

There are a plurality of candidate modes as candidates for the operation mode of the radar device 1, and the mode setting unit 56 sets any one of the plurality of candidate modes to the operation mode of the radar device 1. The plurality of candidate modes include a normal mode and a high-precision mode. Modes other than the normal mode and the high-precision mode may be included in the plurality of candidate modes, but in the following, only the normal mode and the high-precision mode will be focused on. In this case, the mode setting unit 56 switches and sets the operation mode of the radar device 1 to the normal mode or the high precision mode.

Among the plurality of examples shown below, the details of the chirp processing unit 4, the operation in the normal mode and the high precision mode, the specific applied technique related to the radar device 1, and the like will be described. Unless otherwise specified and without contradiction, the above-mentioned matters apply to each embodiment described later. In each of the examples described later, with respect to matters that contradict the above-mentioned contents, the description in each example may take precedence. Further, as long as there is no contradiction, the matters described in any of the plurality of examples described below may be applied to any other example (that is, any two or more of the plurality of examples). It is also possible to combine the examples of).

In the following, for the sake of simplification of the description, the block in which the transmission unit 2, the reception unit 3 and the chirp processing unit 4 are combined is referred to as a transmission / reception block. Further, as described above, the receiving unit 3 is provided with a total of four sets of receiving blocks including the receiving antenna 31 and the receiving processing unit 32, but the configuration and operation of the receiving blocks are common among the plurality of receiving blocks. Therefore, in the following, only one receiving block will be focused on unless otherwise required. Further, in the following description, unless otherwise specified, the transmission signal refers to the transmission signal for the transmission antenna 21, and the reception signal refers to the reception signal for the reception antenna 31.

<< First Example >>
A first embodiment of the radar device 1 will be described. FIG. 6 is a configuration diagram of a transmission / reception block according to the first embodiment. The transmission / reception block according to the first embodiment includes a transmission antenna 21, a reception antenna 31, a mixer 33, an A / D conversion unit 34, a modulation signal generation unit 111, an oscillator 112, a coupler 113, and a chirp processing unit 114.

The transmitting antenna 21, the receiving antenna 31, the mixer 33, and the A / D converter 34 are as described above. The receiving antenna 31, the mixer 33, and the A / D conversion unit 34 in FIG. 6 represent the receiving antenna 31, the mixer 33, and the A / D conversion unit 34 provided in one reception block. The modulation signal generation unit 111 and the oscillator 112 form the transmission processing unit 22 of FIG. 1, and the chirp processing unit 114 forms the chirp processing unit 4 of FIG. 1 including the inverting chirp generation unit. It may be considered that the coupler 113 is included in any of the components of the transmission unit 2 and the chirp processing unit 4 of FIG. 1, and it is considered that the coupler 113 is provided separately from the transmission unit 2 and the chirp processing unit 4 of FIG. You may think.

The modulation signal generation unit 111 generates a modulation signal which is a voltage signal whose voltage changes in a sawtooth shape, and supplies the modulation signal to the oscillator 112. The oscillator 112 frequency-modulates a continuous wave signal based on the modulation signal supplied by the modulation signal generation unit 111 to generate a chirp signal whose frequency changes with the passage of time, and outputs the generated chirp signal. The chirp signal output from the oscillator 112 corresponds to the reference chirp described above. The reference chirp output from the oscillator 112 is supplied to the transmission antenna 21 through the coupler 113, and the transmission antenna 21 outputs a transmission signal consisting of repetitions of the supplied reference chirp as a transmission wave TW in a predetermined direction (that is, in space). Radiate). Therefore, the frequency of the transmitted signal changes between the frequencies f _L and f _H over time.

The coupler 113 takes out a part of the output signal of the oscillator 112 and supplies the taken out signal to the chirp processing unit 114. The signal sent from the oscillator 112 to the transmitting antenna 21 and the signal sent to the chirp processing unit 114 are the same signals except that the powers of the signals are different. Therefore, the reference chirp is also supplied to the chirp processing unit 114. The coupler 113 may be a circuit called a divider (the same applies to any other coupler described later).

The chirp processing unit 114 includes a chirp inversion unit (reversal chirp generation unit) 115. The chirp inverting unit 115 generates an inverting chirp, which is a signal obtained by inverting the reference chirp supplied from the oscillator 112 through the coupler 113 on the time axis. That is, the inverted chirp is a signal in which the mode of frequency increase and decrease in the reference chirp is inverted on the time axis. In FIG. 7, the solid line waveform 611 represents the signal waveform of the reference chirp in the two chirp sections, and the broken line waveform 612 represents the signal waveform of the inverted chirp in the two chirp sections. An inverted chirp can be obtained by inverting the reference chirp on the time axis for each chirp section. For example, if the transmitted signal according to the second modified waveform example shown in FIG. 3B corresponds to the reference chirp, the transmitted signal according to the third modified waveform example shown in FIG. 3C corresponds to the inverted chirp. Will be done.

In FIG. 7, the inverted chirp is generated by inverting the reference chirp on the time axis so that the cycle start time and the cycle end time of the reference chirp are exchanged by inversion. Therefore, in FIG. 7, the time (timing) when the frequency of the reference chirp becomes the frequency f _L and the time (timing) when the frequency of the inverted chirp becomes the frequency f _L are equal. On the other hand, the inverted chirp may be generated so that the start time and the end time of the chirp signal included in the reference chirp are exchanged by inversion. In this case, the time (timing) when the frequency of the reference chirp becomes the frequency f _L and the time (timing) when the frequency of the inverted chirp becomes the frequency f _H become equal. On the contrary, the time (timing) when the frequency of the reference chirp becomes the frequency f _H and the time (timing) when the frequency of the inverted chirp becomes the frequency f _L become equal.

The chirp processing unit 114 is a mixing signal Sn (corresponding to the signal according to the waveform 611 in FIG. 7) corresponding to the signal itself supplied from the oscillator 112 through the coupler 113, or a mixing signal Sr output from the chirp inversion unit 115. (Corresponding to the signal according to the waveform 612 in FIG. 7) is selectively output to the mixer 33. The mixing signal Sn consists of repeating the reference chirp like the transmission signal, and the frequency of the reference chirp in the mixing signal Sn changes between the frequencies f _L and f _H with the passage of time. On the other hand, the mixing signal Sn consists of repeating inversion chirps. The frequency of the inverted chirp also changes between frequencies f _L and f _H with the passage of time, but the mode of the change is inverted on the time axis in comparison with the reference chirp.

The description of the relationship between the reference chirp that constitutes the mixing signal Sn and the inverted chirp that constitutes the mixing signal Sr will be supplemented. Here, since it is assumed that the transmission signal of the aspect shown in FIG. 2 is generated, the frequency of the reference chirp (corresponding to the solid line waveform 611 in FIG. 7) constituting the mixing signal Sn is set to each chirp of the transmission signal. In the section, the slope f _L increases from the frequency f _L to the frequency f _H with a slope θ ₁ with the passage of time, and when the frequency f _H is reached, the slope f _L decreases with the slope θ ₂ with the passage of time. Return to. On the other hand, the frequency of the inverted chirps constituting the mixing signal Sr (corresponding to the broken line waveform 612 in FIG. 7) changes with the passage of time from the frequency f _L to the frequency f _H in each chirp section of the transmission signal. It increases with a slope θ ₂ , and when it reaches the frequency f _H , it decreases with a slope θ ₁ and returns to the frequency f _L with the passage of time.

The received signal obtained by the receiving antenna 31 is amplified by a low noise amplifier (not shown) and then sent to the mixer 33, and the mixer 33 is supplied from the reception signal supplied via the low noise amplifier and the charp processing unit 114. A beat signal is generated by mixing with the mixing signal Sn or Sr.

The chirp processing unit 114 selectively supplies the mixing signal Sn or Sr to the mixer 33 under the control of the mode setting unit 56 (see FIG. 4). When the operation mode of the radar device 1 is set to the normal mode, the mixing signal Sn is supplied from the chirp processing unit 114 to the mixer 33, and as a result, the beat signal is the frequency of the received signal and the frequency of the mixing signal Sn. It has a beat frequency that is the difference from. On the other hand, when the operation mode of the radar device 1 is set to the high precision mode, the mixing signal Sr is supplied from the chirp processing unit 114 to the mixer 33, and as a result, the beat signal is mixed with the frequency of the received signal. It has a beat frequency that is the difference from the frequency of the signal Sr. The beat signal generated by the mixer 33 is converted into a digital signal by the A / D conversion unit 34, and then output to the signal processing unit 5. In order to clearly distinguish between the beat signal in the normal mode and the beat signal in the high precision mode, the beat signal in the normal mode may be referred to as a normal beat signal as necessary in the following, and the beat signal in the high precision mode may be referred to. May be referred to as a special beat signal.

Although not shown in particular, a bandpass filter (not shown) is provided between the mixer 33 and the A / D conversion unit 34. It is preferable that filtering for removing unnecessary frequency components from the mixing result by the mixer 33 is executed by the bandpass filter, and a beat signal indicating the mixing result after the filtering is sent to the A / D conversion unit 34 (others described later). The same applies to the examples).

――― High precision mode ―――
The operation in the high precision mode will be described. In the high-precision mode, the distance calculation unit 53 can obtain the target distance d based on the special beat signal indicating the mixing result (multiplication result) of the reception signal by the receiving antenna 31 and the mixing signal Sr. This principle will be described with reference to FIGS. 8A and 8B. In addition, in FIGS. 8 (a) and 8 (b), it is assumed that the transmission signal has the waveform of the aspect shown in FIG. 3 (b) for convenience of illustration (FIG. 11 (a) described later). And (b) as well).

A situation in which a received signal based on the reflected wave RW from a target at which the distance d is substantially zero is acquired by the receiving antenna 31 is referred to as a zero distance situation. On the other hand, a situation in which a received signal based on the reflected wave RW from a target whose distance d is not substantially zero is acquired by the receiving antenna 31 is referred to as an actual measurement situation. The zero-distance situation is a situation referred to for convenience of explanation, and it is understood that all the situations when the radar device 1 is actually driven belong to the actual measurement situation.

In FIG. 8B, the alternate long and short dash waveform 661 represents the transmission signal for one chirp section under the actual measurement condition (that is, the reference chirp in the transmission signal), and the dashed line waveform 663 is 1 under the actual measurement condition. It represents the inverted chirp for the chirp section. Further, in FIG. 8B, the solid line waveform 662 represents the received signal for one chirp section under the actual measurement situation. The reception signal by the reception antenna 31 is a signal obtained by delaying the transmission signal by a time corresponding to the distance d. Therefore, the received signal by the receiving antenna 31 consists of repeating a chirp signal (hereinafter referred to as a received chirp) having the same waveform as the reference chirp in the transmitted signal, and the received chirp in the received signal is a transmitted signal under actual measurement conditions. Delayed with respect to the reference chirp in.

In FIG. 8A, the broken line waveform 653 represents the inverted chirp for one chirp section under the zero distance situation. In FIG. 8A, the solid line waveform 652 represents a reception signal for one chirp section under a zero distance situation and a transmission signal for one chirp section. In the zero distance situation, the received signal overlaps with the transmitted signal because the delay can be regarded as zero.

The relationship between the inverted chirp and the reference chirp in the transmission signal is, of course, constant regardless of the distance d of the target. Since the inverted chirp is the inverted chirp on the time axis, the frequency of the reference chirp and the frequency of the inverted chirp in the transmission signal do not depend on the distance d of the target in each chirp section. It intersects (ie, matches) at the central timing t _CNT of the chirp section. On the other hand, the received chirp in the received signal is delayed by the time corresponding to the distance d with respect to the reference chirp in the transmitted signal, so that the frequency relationship between the received chirp and the inverted chirp changes according to the distance d. become. Since the amount of this change includes the information of the distance d, it is possible to obtain the distance d by detecting the amount of the change. That is, it is possible to detect the distance d by finding the intersection of the received chirp and the inverted chirp. The intersection of the received chirp and the inverted chirp refers to the intersection between the received chirp and the inverted chirp on a plane having time on the horizontal axis and frequency on the vertical axis. Determining the intersection of the received chirp and inverted chirp can be considered equal to obtaining a central timing t _CNT, a time interval between the timing of frequency matches the received chirp and inverted chirp.

The Fourier transform unit 51 (see FIG. 4) sets a section that is a part of the charp section and includes the center of the charp section as an FFT section for each charp section in the transmission signal, and within the FFT section in the special beat signal. Is treated as a target signal, and the target signal is converted into a signal on the frequency region to generate a frequency spectrum of the target signal (in other words, a frequency spectrum of a special beat signal in the FFT section). The frequency spectrum of the target signal is data indicating the power of a plurality of frequency components included in the target signal for each frequency component. In each chirp interval time from the start timing _{t A1} of the FFT interval to the central timing _{t CNT} chirp interval, the time from the central timing _{t CNT} chirp section to the end timing _{t A2} of the FFT interval, and equal to each other To do.

Under zero distance conditions, the frequency difference between the received chirp and the inverted chirp at timing t _A1 and the frequency difference between the received chirp and the inverted chirp at timing t _A2 are equal to each other, and the frequency difference between them is the predetermined frequency f. It becomes _REF . The frequency f _REF is determined according to the characteristics of the reference chirp. On the other hand, under the actual measurement situation, due to the above delay, the frequency difference between the reception chirp and the inversion chirp at the timing t _A1 becomes a frequency f1 larger than the frequency f _REF , and the reception at the timing t _A2. The frequency difference between the chirp and the inverted chirp is a frequency f2 smaller than the frequency f _REF .

Therefore, when the signal in the FFT section of the special beat signal is used as the target signal and the target signal is converted into a signal on the frequency domain, the obtained frequency spectrum differs between the zero distance situation and the actual measurement situation.

FIG. 9A shows a schematic frequency spectrum 655 obtained by the Fourier transform unit 51 in a high precision mode and in a zero distance situation. In the high-precision mode, a special beat signal is obtained by mixing the received signal and the mixing signal Sr, so that the special beat signal has a beat frequency corresponding to the frequency difference between the received chirp and the inverted chirp. .. In the zero distance situation, the frequency difference between the received chirp and the inverted chirp in the FFT section (corresponding to the frequency difference between the waveforms 652 and 653 in FIG. 8A) has a spread from zero to the frequency f _REF . The frequency spectrum 655 continuously has a corresponding power (for example, a power of a predetermined value or more) in the range from the DC component to the frequency f _REF .

FIG. 9B shows a schematic frequency spectrum 665 obtained by the Fourier transform unit 51 in a high precision mode and under actual measurement conditions. Under actual measurement conditions, the frequency difference between the received chirp and the inverted chirp in the FFT section (corresponding to the frequency difference between the waveforms 662 and 663 in FIG. 8 (b)) has a range from zero to frequency f1, and thus has a frequency. The spectrum 665 has a corresponding power (for example, a power of a predetermined value or more) continuously in the range from the DC component to the frequency f1. The frequency f1 is information that depends on the delay time of the received signal as seen from the transmitted signal, and therefore information that depends on the distance d. Therefore, in the high-precision mode, the distance calculation unit 53 can obtain the target distance d by referring to the frequency spectrum 665 and specifying the frequency f1 in the frequency spectrum 665.

In summary, in high precision mode, the following operations are performed: In the high-precision mode, a special beat signal which is a mixing result of the received signal and the inverted chirp is generated by inputting the received signal and the mixing signal Sr (hence, the inverted chirp) to the mixer 33. The Fourier transform unit 51 sets an FFT section (conversion target section) including the center of the charp section with respect to the charp section to which the reference chapter is transmitted as a transmission signal, and handles the signal in the FFT section in the special beat signal as the target signal. By converting the target signal into a signal on the frequency region, the frequency spectrum 665 of the target signal (in other words, the frequency spectrum of the special beat signal in the FFT section) is generated. The distance calculation unit 53 derives the distance d of the target based on the frequency spectrum 665.

Specifically, for example, the distance calculation unit 53 extracts the maximum frequency having a power equal to or higher than a predetermined value in the frequency spectrum 665 (in other words, extracts the maximum value of the frequency component having a power equal to or higher than a predetermined value). ), The distance d is obtained based on the extracted frequency. The frequency extracted here corresponds to the frequency f1. Since the extracted frequency f1 depends on the distance d, a relational expression showing the relationship between the extracted frequency f1 and the distance d, or table data corresponding to the relationship is given to the distance calculation unit 53 in advance. The distance calculation unit 53 may obtain the distance d from the frequency f1 by using the relational expression or the table data.

In the high-precision mode, the distance d may be obtained based on a plurality of special beat signals obtained by the plurality of reception blocks. At this time, for example, the target distance d is obtained as a candidate distance based on the special beat signal for each reception block, and the final distance d is determined based on the plurality of candidate distances obtained for the plurality of reception blocks. (For example, the average of a plurality of candidate distances may be determined as the distance d). However, the special beat signal may be generated only in one of the plurality of reception blocks provided in the radar device 1, and the distance d may be obtained based on the one special beat signal.

――― Normal mode ―――
The operation in the normal mode will be described. In the normal mode, the distance d of the target can be obtained based on the normal beat signal indicating the mixing result (multiplication result) between the reception signal by the receiving antenna 31 and the mixing signal Sn. As a method for deriving the distance d in the normal mode, a known method can be used. That is, in the normal mode, the Fourier transform unit 51 converts the normal beat signal into a signal on the frequency domain to generate the frequency spectrum 675 as shown in FIG. The FFT section described above is also set during this conversion, and the frequency spectrum 675 is generated based on the normal beat signal in the FFT section. Since the received signal by the receiving antenna 31 is a signal obtained by delaying the transmitted signal by a time corresponding to the distance d, the frequency spectrum 675 has a frequency at which the power has a maximum value as a peak frequency. The frequency depends on the distance d. Therefore, the distance calculation unit 53 specifies the peak frequency in the frequency spectrum 675 and obtains the distance d based on the peak frequency.

Further, in the normal mode, the azimuth angle φ of the target is obtained by the azimuth calculation unit 54 and the relative speed calculation unit 55 of the target using a known method based on the frequency spectrum based on the normal beat signal. The velocity v is required. However, since a plurality of receiving antennas 31 are indispensable for the calculation of the azimuth angle φ, the azimuth angle calculation unit 54 uses a plurality of frequency spectra based on a plurality of normal beat signals obtained by the plurality of reception blocks. Find the azimuth φ of the target.

In the normal mode, a plurality of receiving antennas 31 may be used for deriving the target distance d and the relative velocity v. That is, in the normal mode, the target distance d and the relative velocity v may be derived using a plurality of frequency spectra based on a plurality of normal beat signals obtained by the plurality of reception blocks. However, in normal mode, a single frequency spectrum based on a single normal beat signal obtained in a single reception block can also be used to derive the target distance d and relative velocity v.

In the normal mode, it is preferable to perform triangular wave-shaped frequency modulation in which an up-chirp section and a down-chirp section that are symmetrical to each other are combined, as used in a general FMCW method. That is, in the normal mode, the reference chirp preferably includes a first chirp signal corresponding to the up chirp section and a second chirp signal corresponding to the down chirp section. In this case, the radar device 1 according to the present embodiment can behave as a general FMCW radar in the normal mode, and therefore is also known for calculating the distance d, the relative speed v, and the azimuth angle φ. The method in radar can be used. In such a case, it is preferable to use the same reference chirp as in the normal mode even in the high precision mode. As a result, regarding the output characteristics of the modulation waveform input to the VCO (voltage controlled oscillator) and the modulation characteristics of the VCO, the same design as in the normal mode can be used in the high precision mode, and labor saving in the design can be achieved.

――― Comparison between high-precision mode and normal mode ―――
As can be understood from the above description, in the high precision mode, the target distance d is detected based on the mixing result of the received signal and the inverted chirp (in other words, the mixing result of the received chirp and the inverted chirp). Therefore, the distance measuring unit responsible for the detection includes the Fourier transform unit 51 (spectrum generation unit) and the distance calculation unit 53 (distance derivation unit) in FIG.

In normal mode, ideally, the frequency difference between the transmit and receive signals in the FFT section is constant over the entire FFT section, resulting in a frequency spectrum with power only at frequencies corresponding to the distance d. Be done. However, in reality, various error factors including the linearity error of the reference chirp are entwined, and in the frequency spectrum based on the normal beat signal, a peak waveform having a corresponding spread is generated (see FIG. 10). Therefore, when the frequency spectrum based on the normal beat signal is used, it becomes difficult to improve the detection accuracy of the distance d by a certain amount or more. On the other hand, in the high-precision mode, the frequency f1 is not easily affected by the linearity error of the reference chirp, and by specifying the frequency f1 (in other words, by obtaining the difference between the known frequency f _REF and the frequency f1). Since the method of detecting the distance d is adopted, it is possible to obtain the distance d with high accuracy.

On the other hand, the radar device 1 is provided with a mode setting unit 56 (see FIG. 4) capable of switching the operation mode of the radar device 1 between a normal mode and a high-precision mode, and the mode setting unit 56 is a target. The operation mode of the radar device 1 is set to the high-precision mode when the distance d is detected, and the operation mode of the radar device 1 is set to the normal mode when the azimuth angle φ and the relative velocity v of the target are detected. can do. As described above, in the normal mode, the received signal and the mixing signal Sn (that is, the reference chirp) are mixed by the mixer 33, and in the high precision mode, the received signal and the mixing signal Sr (that is, the inverted chirp) are mixed by the mixer 33. Be mixed.

By making it possible to switch between the normal mode and the high-precision mode in this way, it is possible to easily obtain information other than the target distance d by using an existing method.

Specifically, when the operation mode of the radar device 1 is set to the high-precision mode, a received signal and a mixing signal Sr (that is, an inverted chirp) are input to the mixer 33, thereby obtaining the mixer 33. The distance d is detected by the ranging unit based on the special beat signal that is the mixing result. When the operation mode of the radar device 1 is set to the normal mode, a received signal and a mixing signal Sn (that is, a reference chirp) are input to the mixer 33, and a normal beat which is a mixing result obtained from the mixer 33 by this is input. Based on the signal, the signal processing unit 4 (target data generation unit 52) generates target data related to the target. The target data generated in the normal mode includes information other than the target distance d. Here, the azimuth angle φ and the relative velocity v of the target are obtained as information other than the distance d of the target, but even if only one of them is obtained as information other than the distance d of the target. good.

By collating the distance d derived in the high-precision mode with the distance d derived in the normal mode, it is possible to improve the reliability of the distance d required by the radar device 1. However, the distance d may not be derived in the normal mode.

The mode setting unit 56 is, for example, a radar device so that a section for setting the operation mode of the radar device 1 to the high precision mode and a section for setting the operation mode of the radar device 1 to the normal mode alternately and periodically visit. The operation mode of 1 can be switched and set.

<< Second Example >>
A second embodiment of the radar device 1 will be described. The second embodiment is an embodiment obtained by modifying a part of the first embodiment, and the description of the first embodiment is applied to the second embodiment as long as there is no contradiction regarding matters not particularly described in the second embodiment. May be done.

In the first embodiment described above, a single FFT section is set for each chirp section, but a plurality of FFT sections may be set for each chirp section. In this case, the frequency spectrum of the beat signal can be generated for each FFT section, and the distance d of the target can be obtained based on the plurality of frequency spectra generated for the plurality of FFT sections. The plurality of FFT sections to be set may be three or more FFT sections, but assuming that the first FFT section and the second FFT section are set as the plurality of FFT sections, the present method will be described in detail below. explain.

See FIGS. 11 (a) and 11 (b). The waveforms 652, 653 and 661-663 in FIGS. 11 (a) and 11 (b) are the same as those shown in FIGS. 8 (a) and 8 (b).

The Fourier transform unit 51 (see FIG. 4) according to the second embodiment sets a first FFT section and a second FFT section that are different from each other for each chirp section in the transmission signal. Here, in each chirp section, it is assumed that the first FFT section is a section from timing t _B1 to timing t _CNT , and the second FFT section is a section from timing t _CNT to timing t _B2 . In each chirp section, the timing t _CNT is the central timing of the chirp section. In each chirp section, the timing t _B1 is the timing after the start timing of the chirp section and the timing before the central timing t _{CNT of the} chirp section, and the timing t _B2 is the timing before the end timing of the chirp section. It is the timing of and after the central timing t _{CNT of the} chirp section.

Then, the interval between the timing t _B1 and t _CNT and the interval between the timing t _CNT and t _B2 shall be equal to each other (thus, the length of the first FFT section and the length of the second FFT section shall be equal to each other). .. Then, the timings t _B1 and t _B2 correspond to the above-mentioned timings t _A1 and t _A2 , respectively, and therefore, under the zero distance situation, the frequency difference between the received chirp and the inverted chirp at the timing t _B1 is also the timing t. The frequency difference between the received chirp and the inverted chirp at _B2 is also a predetermined frequency f _REF . The frequency f _REF is determined according to the characteristics of the reference chirp. On the other hand, under the actual measurement situation, the frequency difference between the received chirp and the inverted chirp at the timing t _B1 is a frequency f1 larger than the frequency f _{REF due} to the delay based on the distance d, and the timing t _B2. The frequency difference between the received chirp and the inverted chirp at is a frequency f2 smaller than the frequency f _REF .

The Fourier transform unit 51 converts the special beat signal into a signal on the frequency domain for each FFT section to generate the frequency spectrum of the special beat signal for each FFT section. That is, the Fourier conversion unit 51 treats the signal in the first FFT section of the special beat signal as the first target signal and converts the first target signal into a signal on the frequency domain to generate the frequency spectrum of the first FFT section. At the same time, the frequency spectrum of the second FFT section is generated by treating the signal in the second FFT section of the special beat signal as the second target signal and converting the second target signal into a signal on the frequency domain.

The frequency spectrum of the first FFT section is the frequency spectrum of the first target signal (data showing the power of a plurality of frequency components included in the first target signal for each frequency component), in other words, a special within the first FFT section. The frequency spectrum of the beat signal. The frequency spectrum of the second FFT section is the frequency spectrum of the second target signal (data indicating the power of a plurality of frequency components included in the second target signal for each frequency component), in other words, a special within the second FFT section. The frequency spectrum of the beat signal.

12 (a) and 12 (b) schematically show the frequency spectrum 671 of the first FFT section and the frequency spectrum 672 of the second FFT section obtained by the Fourier transform unit 51 in the high precision mode and in the zero distance situation. In the zero distance situation, the frequency difference between the received chirp and the inverted chirp in the first FFT section has a range from zero to the frequency f _REF, so that the frequency spectrum 671 is appropriate in the range from the DC component to the frequency f _REF . It will have power (for example, power equal to or higher than a predetermined value). The same applies to the frequency spectrum 672.

13 (a) and 13 (b) schematically show the frequency spectrum 681 of the first FFT section and the frequency spectrum 682 of the second FFT section obtained by the Fourier transform unit 51 under the high precision mode and the actual measurement situation. Under the actual measurement situation, the frequency difference between the received chirp and the inverted chirp in the first FFT section has a spread from the frequency f1a to the frequency f1 larger than the frequency f1a, so that the frequency spectrum 681 is in the range from the low frequency component to the frequency f1. It will have a corresponding power (for example, a power of a predetermined value or more) (f1a> 0). On the other hand, under the actual measurement situation, the frequency difference between the received chirp and the inverted chirp in the second FFT section has a range from zero to the frequency f2. This is because the frequency difference becomes zero at the intersection of the receiving chirp and the inverted chirp. Therefore, the frequency spectrum 682 has a corresponding power (for example, a power of a predetermined value or more) in the range from the DC component to the frequency f2. In the high-precision mode, the distance calculation unit 53 can obtain the target distance d by referring to the frequency spectra 681 and 682 and specifying the frequency f1 in the frequency spectrum 681 and the frequency f2 in the frequency spectrum 682.

In summary, in the high precision mode according to the second embodiment, the following operations are executed. In the high-precision mode, a special beat signal which is a mixing result of the received signal and the inverted chirp is generated by inputting the received signal and the mixing signal Sr (hence, the inverted chirp) to the mixer 33. The Fourier transform unit 51 sets a plurality of FFT sections (conversion target sections) for the charp section in which the reference chapter is transmitted as the transmission signal, and handles the signal in the FFT section in the special beat signal as the target signal for each FFT section. By converting the target signal into a signal on the frequency region, the frequency spectrum (for example, 681, 682) of the target signal is generated for each FFT section. The distance calculation unit 53 derives the target distance d based on the frequency spectrum (for example, 681, 682) generated for each FFT section.

Specifically, for example, the distance calculation unit 53 extracts the maximum frequency having a power of a predetermined value or more in the frequency spectrum for each FFT section (in other words, the maximum value of the frequency component having a power of a predetermined value or more). Is extracted), and the distance d is obtained based on the extraction frequency for each FFT section. In the examples of FIGS. 11B and 13A and 13B, the frequency f1 is extracted from the frequency spectrum of the first FFT section and the frequency f2 is extracted from the frequency spectrum of the second FFT section. Since the extracted frequencies f1 and f2 depend on the distance d, the distance calculation unit 53 uses a predetermined calculation formula or table data that defines the relationship between the frequencies f1 and f2 and the distance d, and uses the frequencies. The distance d may be obtained from f1 and f2. At this time, since the difference between the frequencies f1 and f2 also depends on the distance d, the distance d may be obtained based on the difference between the frequencies f1 and f2.

The description of the method of calculating the distance d based on the frequencies f1 and f2 will be supplemented with reference to FIG. At the timing t _CNT , the frequency of the transmitting chirp and the frequency of the inverted chirp coincide with their center frequencies (here, 76.5 GHz). Now, in the actual measurement situation, the difference between the center frequency and the frequency of the received chirp at the timing t _CNT is represented by the symbol “fb”. Then, since “f1 = f _REF + fb” and “f2 = f _REF −fb” are established, the frequency fb is represented by the following equation (1).
fb = (f1-f2) / 2 ... (1)
On the other hand, the delay time t _DELAY of the received signal (received chirp) with respect to the transmitted signal (transmitted chirp) is expressed by the following equation (2) because the slope of the reference chirp between the timings t _B1 and t _B2 is θ _2. To. However, the slope θ ₂ here represents the rate of change of the frequency of the reference chirp per unit time, and the unit of “tan θ ₂ ” is [Hz / s].
t _DELAY = fb / | tanθ ₂ | ... (2)
Since the delay time t _DELAY is the total time of the signal going and returning, the distance d to the target is calculated by the following equation (the speed of light in the space where the transmission and reception signals propagate) using the speed of light c. It is represented by 3).
d = (t _DELAY / 2) × c ・ ・ ・ (3)
Summarizing these equations, the distance d can be obtained by the following equation (4). Since the value of tan θ _{2 and} the value of the speed of light c are known to the distance calculation unit 53, the distance d is derived from the frequencies f1 and f2.
d = (f1-f2) × c / (4 × tanθ ₂ ) ・ ・ ・ (4)

In the method of using the frequency spectrum for a single FFT section as in the first embodiment, it is often difficult to obtain the information of the frequency f2 from the frequency spectrum. In the method of the second embodiment, the information of the frequency f2 can be obtained in addition to the information of the frequency f1, and the information of the frequency f1 can be obtained by using the information of both the frequencies f1 and f2. There is a possibility that the detection accuracy of the distance d (accuracy of the derived distance d) can be improved as compared with the method that can only be used.

The first FFT section and the second FFT section according to the examples of FIGS. 11A and 11B are two FFT sections separated by the central timing t _CNT of the chirp section, but a part of the first FFT section. And a part of the second FFT section may overlap each other. That is, for example, as shown in FIG. 15A, in each chirp section, the first FFT section is a section from timing t _B1 to timing t _B3 , and the second FFT section is a section from timing t _B4 to timing t _B2. There may be. Here, in each chirp section, the timing t _B3 is after the timing t _CNT and before the timing t _B2 , and the timing t _B4 is before the timing t _CNT and before the timing t _B1. It is a later timing. Therefore, in the example of FIG. 15A, the first FFT section and the second FFT section overlap each other between the timings t _B4 and t _B3 . At this time, it is preferable that the interval between the timing t _B4 and t _CNT and the interval between the timing t _CNT and t _B3 are equal to each other, whereby the length of the first FFT section and the length of the second FFT section are mutually equal. Become equal.

Alternatively, for example, as shown in FIG. 15B, in each chirp section, the section from timing t _B1 to timing t _B4 is set as the first FFT section, and the section from timing t _B3 to timing t _B2 is set as the second FFT section. You may set it. Also in this case, it is preferable that the interval between the timing t _B4 and t _CNT and the interval between the timing t _CNT and t _B3 are equal to each other, whereby the length of the first FFT section and the length of the second FFT section are equal to each other. Are equal to each other.

Further, when the first and second FFT sections are set as shown in FIG. 15B, both the first and second FFT sections do not include the intersection of the receiving chirp and the inverted chirp as shown in FIG. A configuration (hereinafter referred to as a modified configuration) can also be adopted. In the modified configuration, the minimum frequency in the frequency spectrum for each FFT section can be detected, and the distance d can be obtained using the minimum frequency. This will be described.

17 (a) and 17 (b) show schematicly the frequency spectrum 691 of the first FFT section and the frequency spectrum 692 of the second FFT section obtained by the Fourier transform unit 51 under the high-precision mode and the actual measurement situation related to the deformation configuration. Shown in. In the modified configuration, since the first FFT section does not include the above intersection, the frequency spectrum 691 has no power in the DC component and has a spread from frequency f1'greater than zero to frequency f1 (f1'). <F1). Similarly, since the second FFT section does not include the above intersection, the frequency spectrum 692 has no power in the DC component and has a spread from frequency f2'greater than zero to frequency f2 (f2'<. f2).

Then, the following operations can be executed in the modified configuration. That is, in the high-precision mode, the distance calculation unit 53 related to the modified configuration extracts the minimum frequency having a power equal to or higher than a predetermined value in the frequency spectrum for each FFT section (in other words, a frequency having a power equal to or higher than a predetermined value). The minimum value of the component is extracted), and the distance d may be obtained based on the extraction frequency for each FFT section. In the examples of FIGS. 17A and 17B, the frequency f1'is extracted as the minimum frequency from the frequency spectrum 691 in the first FFT section, and the frequency f2'is extracted as the minimum frequency from the frequency spectrum 692 in the second FFT section. To. Since the extracted frequencies f1'and f2'depend on the distance d, the distance calculation unit 53 uses a predetermined calculation formula or table data that defines the relationship between the frequencies f1'and f2'and the distance d. The distance d may be obtained from the frequencies f1'and f2'. At this time, since the difference between the frequencies f1'and f2' also depends on the distance d, the distance d may be obtained based on the difference between the frequencies f1'and f2'. The distance d can be derived from between the minimum frequencies f1'and f2' by a principle similar to the above-mentioned principle of deriving the distance d based on the maximum frequencies f1 and f2.

The distance d may be obtained by using both the maximum frequency and the minimum frequency. That is, for example, the distance calculation unit 53 related to the modified configuration extracts frequencies f1'and f1 as the minimum frequency and the maximum frequency having a power equal to or higher than a predetermined value from the frequency spectrum 691 of the first FFT section, respectively, and the second FFT section. The frequencies f2'and f2 are extracted as the minimum frequency and the maximum frequency having a power equal to or more than a predetermined value from the frequency spectrum 692 of the above, and the distance d is obtained based on the extraction frequencies f1', f1, f2' and f2, respectively. You may. For example, the distance d calculated based on the maximum frequencies f1 and f2 is treated as the first candidate distance, and the distance d calculated based on the minimum frequencies f1'and f2' is treated as the second candidate distance. The final distance d may be determined based on the second candidate distance (for example, the average of the first and second candidate distances may be determined as the distance d).

In the method of the second embodiment, the total amount of power of the received signal belonging to each FFT section tends to be smaller than that of the method of the first embodiment. When the total amount of power of the received signal that is the source of the frequency spectrum is reduced, the signal-to-noise ratio in the frequency spectrum decreases, and when the signal-to-noise ratio above a certain value cannot be obtained, the distance d can be significantly detected. Can not. On the other hand, the strength of the received signal tends to decrease as the distance d increases. From these facts, in the method of the second embodiment, it is significantly difficult to obtain the distance d for a relatively distant target (that is, the maximum detection distance is reduced) as compared with the method of the first embodiment. Can be considered.

In consideration of this, when the method of the second embodiment is used in the high precision mode, the following may be performed. That is, the high-precision mode may be used to detect the distance d of the target located within the predetermined distance range, and the normal mode may be used to detect the distance d of the target located outside the predetermined distance range. Here, the predetermined distance range refers to a range in which the distance from the radar device 1 is equal to or less than a predetermined high-precision detectable distance.

Actually, for example, the mode setting unit 56 visits the section in which the operation mode of the radar device 1 is set to the high-precision mode and the section in which the operation mode of the radar device 1 is set to the normal mode alternately and periodically. The operation mode of the radar device 1 is switched and set, and in the high-precision mode, the distance d of the target is derived only for the target located within the predetermined distance range, and in the normal mode, the target located outside the predetermined distance range is derived. It is sufficient to derive the distance d of the target only for the target. Since high accuracy is often not required for detecting the distance to a distant target, there is also an idea that the normal mode is sufficient for detecting the distance of a target outside the predetermined distance range.

<< Third Example >>
A third embodiment of the radar device 1 will be described. In the third embodiment, an example of a configuration for selectively supplying the above-mentioned mixing signal Sn or Sr (see FIG. 6) to the mixer 33 will be described.

FIG. 18 is a configuration diagram of a transmission / reception block according to the third embodiment. The transmission / reception block according to the third embodiment includes a transmission antenna 21, a reception antenna 31, a mixer 33, an A / D conversion unit 34, a continuous wave oscillator 211, a coupler 212, a modulator 213, a VCO 214, a modulation signal generation unit 215, and a VCO 216. And a modulator 217.

The transmitting antenna 21, the receiving antenna 31, the mixer 33, and the A / D converter 34 are as described above. The receiving antenna 31, the mixer 33, and the A / D conversion unit 34 in FIG. 18 represent the receiving antenna 31, the mixer 33, and the A / D conversion unit 34 provided in one reception block. The transmission processing unit 22 of FIG. 1 is mainly composed of a continuous wave oscillator 211, a modulator 213 and a VCO 214, and the chirp processing unit 4 of FIG. 1 including an inverting chirp generation unit mainly includes a VCO 216 and a modulator 217. It is composed. It can be considered that each of the modulation signal generation unit 215 and the coupler 212 is also used as a component of each of the transmission processing unit 22 and the chirp processing unit 4.

Continuous wave oscillator 211 generates and outputs a continuous wave signal 230 having a fixed frequency f _O. The frequency f _O has a frequency similar to that of the reference chirp. The frequency f _O is set, for example, to the center frequency between the frequencies f _H and f _L (here, 76.5 GHz), or to the frequency f _H or f _L. The signal 230 is supplied to the modulator 213 through the coupler 212.

The modulation signal generation unit 215 generates and outputs a modulation signal which is a voltage signal whose voltage changes in a sawtooth shape. However, the modulation signal generation unit 215 selectively outputs the reference modulation signal 231 or the inverting modulation signal 232 as the modulation signal. The reference modulation signal 231 is a modulation signal for generating a reference chirp, and the inverting modulation signal 232 is a modulation signal for generating an inverting chirp. In each cycle of the reference modulation signal 231, the inversion modulation signal 232 is obtained by inverting the reference modulation signal 231 on the time axis. As shown in FIG. 19, for example, the modulation signal generation unit 215 generates a digital waveform generation unit 215a that generates a reference digital waveform corresponding to the reference modulation signal 231 and an inverted digital waveform obtained by inverting the reference digital waveform on the time axis. It has an inversion portion 215b and the like. Then, the modulation signal generation unit 215 can generate the reference modulation signal 231 based on the reference digital waveform and can generate the inverting modulation signal 232 based on the inverting digital waveform.

The modulation signal generation unit 215 always supplies the reference modulation signal 231 to the VCO 214. The VCO 214 is a voltage-controlled oscillator that controls its own oscillation frequency based on the supplied reference modulation signal 231 and generates a chirp signal 233 whose frequency changes with the passage of time based on the reference modulation signal 231. The frequency change width Δf in the chirp signal 233 coincides with the frequency change width Δf (that is, the difference between the frequencies f _L and f _H ) of the reference chirp signal. The period of the chirp signal 233 is the same as the period of the reference chirp. The VCO 214 may be a voltage controlled oscillator capable of generating a signal having a frequency of about Δf (here, 1 GHz) (the same applies to the VCO 216). The chirp signal 233 generated by the VCO 214 is supplied to the modulator 213.

The modulator 213 generates a chirp signal 234 by frequency-modulating the continuous wave signal 230 supplied from the continuous wave oscillator 211 through the coupler 212 using the chirp signal 233 supplied from the VCO 214. This chirp signal 234 corresponds to the above-mentioned reference chirp. By multiplying the continuous wave signal 230 by the chirp signal 233, the frequency of the chirp signal 234 changes between the frequencies f _L and f _H with the passage of time. The chirp signal 234 is supplied to the transmitting antenna 21 as a transmitting signal. The transmitting antenna 21 outputs a transmission signal composed of repetitions of a reference chirp (chirp signal 234) supplied from the modulator 213 as a transmission wave TW in a predetermined direction (that is, radiates into space).

On the other hand, the modulation signal generation unit 215 selectively supplies the reference modulation signal 231 or the inverting modulation signal 232 to the VCO 216. Specifically, the modulation signal generation unit 215 sets the reference modulation signal 231 to the VCO 216 when the operation mode of the radar device 1 is set to the normal mode under the control of the mode setting unit 56 (see FIG. 4). When the operation mode of the radar device 1 is set to the high precision mode, the inverting modulation signal 232 is supplied to the VCO 216.

The VCO 216 is a voltage controlled oscillator that controls its own oscillation frequency based on the supplied modulated signal. The VCO 216 generates a chirp signal 235 and supplies the chirp signal 235 to the modulator 217 when the reference modulation signal 231 is being supplied, while generating a chirp signal 236 when the reference modulation signal 232 is being supplied. The chirp signal 236 is supplied to the modulator 217. The chirp signal 235 is the same as the chirp signal 233 generated by the VCO 214. Since the inverting modulation signal 232 is obtained by inverting the reference modulation signal 231 on the time axis in each cycle of the reference modulation signal 231, the chirp signal 235 is inverted on the time axis in each cycle of the chirp signal 235. Corresponds to the chirp signal 236. Both the frequency change width Δf in the chirp signal 235 and the frequency change width Δf in the chirp signal 236 coincide with the frequency change width Δf (that is, the difference between the frequencies f _L and f _H ) of the reference chirp signal. The respective cycles of the chirp signals 235 and 236 are also the same as the cycle of the reference chirp.

Further, the coupler 212 takes out a part of the output signal 230 of the continuous wave oscillator 211 and supplies the taken out signal to the modulator 217.

The modulator 217 generated and generated a mixing signal 237 which is a chirp signal by frequency-modulating the continuous wave signal 230 supplied through the coupler 212 using the chirp signal 235 or 236 supplied from the VCO 216. The mixing signal 237 is output to the mixer 33. The mixing signal 237 is any of the above-mentioned mixing signals Sn and Sr. As described above, the mixing signal Sn consists of repeating the reference chirp like the transmission signal, and the frequency of the reference chirp in the mixing signal Sn changes between the frequencies f _L and f _H with the passage of time. On the other hand, the mixing signal Sn consists of repeating inversion chirps. The frequency of the inverted chirp also changes between frequencies f _L and f _H with the passage of time, but the mode of the change is inverted on the time axis in comparison with the reference chirp.

When the operation mode of the radar device 1 is set to the normal mode, the chirp signal 235 is supplied from the VCO 216 to the modulator 217 through the reference modulation signal 231 being supplied to the VCO 216, and the modulator 217 is a continuous wave. By multiplying the signal 230 of the above by the chirp signal 235, the mixing signal Sn is generated as the mixing signal 237 and output to the mixer 33.

When the operation mode of the radar device 1 is set to the high precision mode, the chirp signal 236 is supplied from the VCO 216 to the modulator 217 through the inverting modulation signal 232 being supplied to the VCO 216, and the modulator 217 is continuous. By multiplying the wave signal 230 by the chirp signal 236, the mixing signal Sr is generated as the mixing signal 237 and output to the mixer 33.

The received signal obtained by the receiving antenna 31 is amplified by a low noise amplifier (not shown) and then sent to the mixer 33. The mixer 33 is supplied from the received signal supplied through the low noise amplifier and the modulator 217. A beat signal is generated by mixing with the mixing signal 237. Therefore, when the operation mode of the radar device 1 is set to the normal mode, the normal beat signal indicating the mixing result of the received signal and the mixing signal Sn (in other words, the mixing result of the received chirp and the reference chirp) is displayed. It is generated by the mixer 33. When the operation mode of the radar device 1 is set to the high precision mode, the special beat signal indicating the mixing result of the received signal and the mixing signal Sr (in other words, the mixing result of the received chirp and the inverted chirp) is the mixer. Generated at 33.

The operations of the A / D conversion unit 34 and the signal processing unit 5 based on the generated beat signal are as described above, and any of the first and second embodiments is used for the detection method (derivation method) of the distance d. You may.

<< Fourth Example >>
A fourth embodiment of the radar device 1 will be described. In the third embodiment, a specific configuration example for generating the reference chirp and the inverted chirp has been given, but as long as the reference chirp and the inverted chirp can be generated in the transmission / reception block, the configuration of the transmission / reception block (particularly, the chirp processing unit 4 Configuration) can be modified in various ways. For example, the transmission / reception block shown in FIG. 20 may be configured.

The transmission / reception block of FIG. 20 includes a transmission antenna 21, a reception antenna 31, a mixer 33, and an A / D converter 34, a continuous wave oscillator 311, a coupler 312, a modulator 313, a coupler 314, a continuous wave selection output unit 315, and a mixer 316. And a modulator 317.

The transmitting antenna 21, the receiving antenna 31, the mixer 33, and the A / D converter 34 are as described above. The receiving antenna 31, the mixer 33, and the A / D conversion unit 34 in FIG. 20 represent the receiving antenna 31, the mixer 33, and the A / D conversion unit 34 provided in one reception block. The transmission processing unit 22 of FIG. 1 is mainly composed of a continuous wave oscillator 311 and a modulator 313, and the chirp processing unit 4 of FIG. 1 including an inverting chirp generation unit mainly includes a continuous wave selection output unit 315, a mixer 316 and It is configured to include a modulator 317. It can be considered that the couplers 312 and 314 are also used as the respective components of the transmission processing unit 22 and the chirp processing unit 4.

Continuous wave oscillator 311 generates and outputs a signal 330 of a continuous wave having a fixed frequency f _O. The frequency f _O has a frequency similar to that of the reference chirp. The frequency f _O is set, for example, to the center frequency between the frequencies f _H and f _L (here, 76.5 GHz), or to the frequency f _H or f _L. The signal 330 is supplied to the modulator 313 through the coupler 312.

The modulator 313 generates a chirp signal 331 by frequency-modulating the continuous wave signal 330 supplied from the continuous wave oscillator 311 through the coupler 312. The modulator 313 generates the chirp signal 233 (see FIG. 18) shown in the third embodiment by itself or receives the supply of the chirp signal 233 from the outside, and uses the chirp signal 233 to frequency-modulate the continuous wave signal 330. It is good to realize. The chirp signal 331 generated by the modulator 313 corresponds to the reference chirp described above, and therefore the frequency of the chirp signal 331 will change between frequencies f _L and f _H over time. The chirp signal 331 generated by the modulator 313 is supplied to the transmitting antenna 21 as a transmission signal through the coupler 314. The transmitting antenna 21 outputs a transmission signal composed of repetitions of a reference chirp (chirp signal 331) supplied from the modulator 313 as a transmission wave TW in a predetermined direction (that is, radiates into space).

The continuous wave selection output unit 315 generates a continuous wave signal 332 having a frequency f _H and a continuous wave signal 333 having a frequency f _L, and selectively transmits the continuous wave signal 332 or 333 to the mixer 316. Output. The continuous wave signal may be appropriately referred to as a continuous wave signal below. Specifically, the continuous wave selection output unit 315 is a continuous wave having a frequency f _L when the operation mode of the radar device 1 is set to the normal mode under the control of the mode setting unit 56 (see FIG. 4). provides a signal 332 to the mixer 316 supplies a continuous wave signal 333 of frequency f _H in when the operation mode of the radar apparatus 1 is set to the high mode to the mixer 316.

The coupler 314 takes out a part of the output signal (that is, the chirp signal 331) of the modulator 313 and supplies the taken out signal to the mixer 316. The mixer 316 has an output signal of the modulator 313 (that is, a chirp signal 331) supplied through the coupler 314 and a continuous wave signal (that is, a continuous wave signal 332 of a frequency f _L or a frequency f) supplied from the continuous wave selection output unit 315. The continuous wave signal 333) of _H is mixed, and a signal indicating the mixing result is output.

The output signal of the mixer 316 is a chirp signal, and the chirp signal output from the mixer 316 is supplied to the modulator 317. When the continuous wave selection output unit 315 outputs the continuous wave signal 332 of the frequency f _L , the mixing result of the mixer 316 is the output signal of the modulator 313 (that is, the chirp signal 331) and the continuous wave signal 332 of the frequency f _L. As a result, the output signal of the mixer 316 becomes a chirp signal 334 whose frequency changes by the frequency difference Δf between the frequencies f _L and f _H. When the continuous wave selection output unit 315 outputs the continuous wave signal 333 of the frequency f _H , the mixing result of the mixer 316 is the output signal of the modulator 313 (that is, the chirp signal 331) and the continuous wave signal 333 of the frequency f _H. As a result, the output signal of the mixer 316 becomes a chirp signal 335 whose frequency changes by the frequency difference Δf between the frequencies f _L and f _H.

The frequency change widths (Δf) in the chirp signals 334 and 335 are common to each other, but the mode of the change differs between the chirp signals 334 and 335. That is, since it is assumed here that the transmission signal of the aspect shown in FIG. 2 is generated as a chirp signal 331 and the frequencies f _L and f _H are 76.0 GHz and 77.0 GHz, respectively. The frequency of the chirp signal 334 increases with a gradient θ ₁ from 0 GHz to 1 GHz with the passage of time in each chirp section of the transmission signal as the chirp signal 331, and when it reaches 1 GHz, it is tilted with the passage of time. It decreases at θ ₂ and returns to 0 GHz. In contrast, the frequency of the chirp signal 335 in each chirp interval of the transmission signal as a chirp signal 331, over time toward the 1GHz so on are increased in inclination theta ₂ from 0 GHz, the time elapsed reaches the 1GHz Along with this, the slope decreases at θ ₁ and returns to 0 GHz. That is, in each cycle of the chirp signal 334, the chirp signal 334 inverted on the time axis corresponds to the chirp signal 335.

Further, the coupler 312 takes out a part of the output signal 330 of the continuous wave oscillator 311 and supplies the taken out signal to the modulator 317.

The modulator 317 generates and generates a mixing signal 336, which is a chirp signal, by frequency-modulating the continuous wave signal 330 supplied through the coupler 312 using the chirp signal 334 or 335 supplied from the mixer 316. The mixing signal 336 is output to the mixer 33. The mixing signal 336 is any of the above-mentioned mixing signals Sn and Sr. As described above, the mixing signal Sn consists of repeating the reference chirp like the transmission signal, and the frequency of the reference chirp in the mixing signal Sn changes between the frequencies f _L and f _H with the passage of time. On the other hand, the mixing signal Sn consists of repeating inversion chirps. The frequency of the inverted chirp also changes between frequencies f _L and f _H with the passage of time, but the mode of the change is inverted on the time axis in comparison with the reference chirp.

When the operation mode of the radar device 1 is set to the normal mode, the chirp signal 334 is supplied from the mixer 316 to the modulator 317 and modulated by supplying the continuous wave signal 332 of the frequency f _{L to} the mixer 316. The device 317 generates a mixing signal Sn as a mixing signal 336 by multiplying the signal 330 from the continuous wave oscillator 311 by the chirp signal 334, and outputs the mixing signal Sn to the mixer 33.

When the operation mode of the radar device 1 is set to the high precision mode, the chirp signal 335 is supplied from the mixer 316 to the modulator 317 through the continuous wave signal 333 having a frequency f _H being supplied to the mixer 316. The modulator 317 generates a mixing signal Sr as a mixing signal 336 by multiplying the signal 330 from the continuous wave oscillator 311 by the chirp signal 335, and outputs the mixing signal Sr to the mixer 33.

The received signal obtained by the receiving antenna 31 is amplified by a low noise amplifier (not shown) and then sent to the mixer 33, and the mixer 33 is supplied from the received signal supplied via the low noise amplifier and the modulator 317. A beat signal is generated by mixing with the mixing signal 336. Therefore, when the operation mode of the radar device 1 is set to the normal mode, the normal beat signal indicating the mixing result of the received signal and the mixing signal Sn (in other words, the mixing result of the received chirp and the reference chirp) is displayed. It is generated by the mixer 33. When the operation mode of the radar device 1 is set to the high precision mode, the special beat signal indicating the mixing result of the received signal and the mixing signal Sr (in other words, the mixing result of the received chirp and the inverted chirp) is the mixer. Generated at 33.

The operations of the A / D conversion unit 34 and the signal processing unit 5 based on the generated beat signal are as described above, and any of the first and second embodiments is used for the detection method (derivation method) of the distance d. You may.

<< Fifth Example >>
A fifth embodiment of the radar device 1 will be described. In the fifth embodiment, some deformation techniques and the like for the above-mentioned rotor device 1 will be described.

An example is given in which four reception blocks including the reception antenna 31 and the reception processing unit 32 are provided for the radar device 1, but the number of reception blocks provided in the radar device 1 is arbitrary as long as it is two or more. It may be 1. It is possible to obtain the distance d even if there is only one receiving block.

When focusing only on detecting the distance d of the target with high accuracy, the mixing signal Sr composed of the inverted chirp may always be supplied to the mixer 33 as the mixing signal. In this case, it is not necessary to switch the operation mode.

Of the plurality of reception blocks provided in the radar device 1, a special beat signal is generated only in a part of the reception blocks (for example, one reception block), and the other reception blocks generate only a normal beat signal as a beat signal. You can do it.

The configuration of the radar device 1 has been described by taking the FMCW method as an example, but the radar device 1 may be a radar device that employs the FCM (Fast Chirp Modulation) method.

In the above description, it is assumed that the transmission signal from the transmission antenna 21 consists of repeating reference chirps. However, in the high precision mode, it is not essential that the reference chirp is transmitted repeatedly, and a single reference chirp may be transmitted.

Further, the reference chirp according to an example of the present invention as represented by the example of FIG. 2 includes a first section in which the frequency increases and a second section in which the frequency decreases. That is, the chirp section, which is the cycle of the reference chirp, is composed of a combination of the first section and the second section. Then, in the example of FIG. 2, the frequency of the reference chirp changes monotonically in each of the first section and the second section (monotonically increases in the first section and monotonically decreases in the second section). However, in the present invention, the frequency of the reference chirp according to the high-precision mode may be monotonically changed (monotonically increasing or monotonically decreasing) only in one of the first section and the second section. That is, when detecting the distance d in the high-precision mode, it is sufficient that the reference chirp has at least one section in which the frequency changes monotonically.

The embodiments of the present invention can be appropriately modified in various ways within the scope of the technical idea shown in the claims. The above embodiments are merely examples of the embodiments of the present invention, and the meanings of the terms of the present invention and the respective constituent requirements are not limited to those described in the above embodiments. The specific numerical values shown in the above description are merely examples, and as a matter of course, they can be changed to various numerical values.

1 Radar device 2 Transmitter 3 Receiver 4 Chirp processing 5 Signal processing 21 Transmitting antenna 22 Transmitting processing 31 Receiving antenna 32 Receiving processing 33 Mixer 34 A / D converter

Claims (10)

A transmitter that transmits a reference chirp having a section where the frequency changes monotonically as a transmission signal,
A receiver that receives the reflected wave from the target of the transmission signal as a reception signal, and
An inverted chirp generator that generates an inverted chirp by inverting the reference chirp on the time axis.
A radar device including a radar device and a signal processing unit having a distance measuring unit that detects a distance between the targets based on a mixing result of the received signal and the inverted chirp. The radar device according to claim 1, wherein the receiving unit has a mixer, and by inputting the received signal and the inverted chirp to the mixer, a mixing result of the received signal and the inverted chirp can be obtained. Using the mixer, a beat signal indicating the mixing result of the received signal and the inverted chirp is generated.
The ranging unit
By setting a conversion target section including the center of the chirp section with respect to the chirp section to which the reference chirp is transmitted, and converting the signal in the conversion target section of the beat signal into a signal on the frequency region as the target signal. A spectrum generator that generates the frequency spectrum of the target signal,
The radar device according to claim 2, further comprising a distance deriving unit that derives the distance based on the frequency spectrum of the target signal. The radar device according to claim 3, wherein the distance derivation unit derives the distance based on the maximum frequency having a power equal to or higher than a predetermined value in the frequency spectrum. Using the mixer, a beat signal indicating the mixing result of the received signal and the inverted chirp is generated.
The ranging unit
A plurality of conversion target sections are set for the chirp section to which the reference chirp is transmitted, and the signal in the conversion target section in the beat signal is converted into a signal on the frequency domain for each conversion target section. A spectrum generator that generates the frequency spectrum of the target signal for each conversion target section,
The radar device according to claim 2, further comprising a distance deriving unit that derives the distance based on the frequency spectrum generated for each conversion target section. 5. The distance derivation unit extracts the maximum frequency having a power equal to or higher than a predetermined value in the frequency spectrum for each conversion target section, and derives the distance based on the extraction frequency for each conversion target section. Radar device described in. 5. The distance derivation unit extracts the minimum frequency having a power equal to or higher than a predetermined value in the frequency spectrum for each conversion target section, and derives the distance based on the extraction frequency for each conversion target section. Radar device described in. The operation mode of the radar device is set to be switched to the first mode in which the received signal and the reference chirp are mixed by the mixer, or the second mode in which the received signal and the inverted chirp are mixed by the mixer. The radar device according to any one of claims 2 to 7, further comprising a possible mode setting unit. The reference chirp includes a first section in which the frequency increases and a second section in which the frequency decreases.
The frequency of the reference chirp changes monotonically in at least one of the first section and the second section.
The transmitter repeatedly transmits the reference chirp as the transmission signal,
When the operation mode is set to the second mode, the received signal and the inverted chirp are input to the mixer, and the mixing result of the received signal obtained from the mixer and the inverted chirp is obtained. Based on this, the distance is detected by the distance measuring unit.
When the operation mode is set to the first mode, the received signal and the reference chirp are input to the mixer, and the mixing result of the received signal obtained from the mixer and the reference chirp is obtained. Based on this, the signal processing unit generates target data related to the target, and the target data is generated.
The radar device according to claim 8, wherein the target data generated in the first mode includes information other than the distance. This is a target distance detection method used in radar equipment.
A transmission process in which a reference chirp having a section in which the frequency changes monotonically is transmitted as a transmission signal, and
The receiving process of receiving the reflected wave by the target of the transmission signal as a reception signal, and
An inverted chirp generation step of generating an inverted chirp by inverting the reference chirp on the time axis, and
A target distance detection method comprising a distance measuring step of detecting a distance between the radar device and the target based on a mixing result of the received signal and the inverted chirp.

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