JP2020056772A - Radar device, and radar method - Google Patents

Abstract

To provide a radar device capable of reducing the ambiguity of Doppler frequency.SOLUTION: The radar device includes: multiple transmission parts; a control part configured to select a transmission part that transmits a transmission signal out of the multiple transmission parts for each transmission period of the transmission signal and set a transmission gap period which is a period when the transmission signal is not transmitted between the first period in which each of the multiple transmission parts is selected at least once and a second period after the first period in which each of the multiple transmission parts is selected at least once.SELECTED DRAWING: Figure 14

Description

  The present invention relates to a radar device and a radar method.

  2. Description of the Related Art In recent years, a radar device using a short-wavelength signal including a microwave or a millimeter wave capable of obtaining high resolution has been studied. Further, in order to improve outdoor safety, there is a demand for the development of a radar device (wide-angle radar device) that detects an object (target) including a pedestrian in a wide-angle range in addition to a vehicle.

  As a configuration of a radar apparatus having a wide-angle detection range, a method of receiving with a plurality of antennas (array antennas) and estimating a direction (arrival angle) at which a reflected wave arrives from a target based on a reception phase difference with respect to an antenna interval ( Angle of arrival estimation method). As the arrival angle estimation method, for example, an FFT (Fast Fourier Transform) method is used. As the arrival angle estimation method, for example, a Capon method, MUSIC (Multiple Signal Classification), ESPRIT (Estimation of Signal Parameters via Rotational Invariance Techniques), or the like is used as a method for obtaining high angle resolution.

  Further, there has been proposed a configuration in which a plurality of transmission antennas (array antennas) are provided not only on the reception side but also on the transmission side, and beam scanning is performed by signal processing using a transmission / reception array (MIMO radar) (for example, non-transmission type). Patent Document 1).

  The MIMO radar transmits signals multiplexed from a plurality of transmission antennas using time division, frequency division, or code division from the plurality of transmission antennas. The MIMO radar receives a signal reflected by an object (target) located in the vicinity by a plurality of receiving antennas, and separates a multiplexed transmission signal from each of the received signals. This allows the MIMO radar to extract a channel response indicated by the product of the number of transmitting antennas and the number of receiving antennas. Further, in the MIMO radar, by appropriately arranging the intervals between the transmitting and receiving antennas, the antenna aperture can be virtually enlarged and the angular resolution can be improved.

For example, Patent Literature 1 discloses a MIMO radar using time division multiplex transmission in which a signal is transmitted with a transmission time being shifted for each transmission antenna (hereinafter, referred to as “time division multiplex MIMO radar”). Is disclosed. Time division multiplex transmission can be realized with a simpler configuration than frequency multiplex transmission or code multiplex transmission. In time division multiplex transmission, orthogonality between transmission signals can be favorably maintained by sufficiently widening transmission time intervals. The time division multiplexed MIMO radar outputs a transmission pulse, which is an example of a transmission signal, while sequentially switching the transmission antenna at a predetermined period _Tr . Then, the time-division multiplexing MIMO radar receives a signal in which a transmission pulse is reflected by an object by a plurality of reception antennas, performs a correlation process between the reception signal and the transmission pulse, and performs a spatial FFT process (the arrival direction of the reflected wave). Estimation processing).

JP 2008-304417 A JP 2011-526371 A JP-A-2006-50778

J. Li, P. Stoica, "MIMO Radar with Colocated Antennas," Signal Processing Magazine, IEEE Vol. 24, Issue: 5, pp. 106-114, 2007 M. Kronauge, H. Rohling, "Fast two-dimensional CFAR procedure", IEEE Trans. Aerosp. Electron. Syst., 2013, 49, (3), pp. 1817-1823 Direction-of-arrival estimation using signal subspace modeling Cadzow, J.A .; Aerospace and Electronic Systems, IEEE Transactions on Volume: 28, Issue: 1 Publication Year: 1992, Page (s): 64-79

As described above, the time division multiplexed MIMO radar sequentially switches the transmission antenna that transmits a transmission signal (for example, a transmission pulse or a radar transmission wave) at a predetermined period _Tr . Therefore, time-division multiplexing may require a longer time to complete transmission of transmission signals from all transmission antennas than frequency-division transmission or code-division transmission. For this reason, for example, when a transmission signal is transmitted from each transmission antenna and the Doppler frequency (that is, the relative speed of the target) is detected from a change in the reception phase of the transmission signal as in Patent Literature 2, it is necessary to detect the Doppler frequency. In applying the Fourier frequency analysis to the above, the time interval of observation of the change in the reception phase becomes longer. Therefore, the Doppler frequency range in which the Doppler frequency can be detected without aliasing (that is, the range of the target relative velocity that can be detected) is reduced.

Also, when a reflected signal from a target exceeding the Doppler frequency range in which Doppler frequency can be detected without aliasing is assumed, it is not possible to specify whether or not the aliasing component is present, and the ambiguity of the Doppler frequency (that is, the relative velocity of the target) ( Uncertainty, Ambiguity) occurs. For example, when sending the transmission signal (transmission pulse) while sequentially switching the N _t transmit antennas in a predetermined cycle T _r, the transmission time of T _r N _t is required. When such time division multiplexing transmission is repeated _Nc times and Fourier frequency analysis is applied to detect the Doppler frequency, the Doppler frequency range in which the Doppler frequency can be detected without aliasing is ± 1 / (2T _r N _t ). Therefore, the Doppler frequency range capable of detecting the Doppler frequency without aliasing is reduced as the number of transmit antennas N _t increases, ambiguity of the Doppler frequency is likely to occur at slower relative velocity.

  An object of one embodiment of the present disclosure is to provide a radar device that can reduce ambiguity of Doppler frequency.

  The radar device according to an aspect of the present disclosure, a plurality of transmission units, and a transmission unit that transmits the transmission signal among the plurality of transmission units for each transmission cycle of the transmission signal, and each of the plurality of transmission units A period in which the transmission signal is not transmitted between a first period in which the transmission signal is selected at least once and a second period after the first period in which each of the plurality of transmission units is selected at least one time. And a control unit for providing a transmission gap period.

  Further, the radar apparatus according to an aspect of the present disclosure includes a plurality of transmission units, and a code multiplexing unit that generates a code-multiplexed transmission signal by cyclically multiplexing code elements of orthogonal codes into transmission signals for each transmission cycle. Wherein each of the plurality of transmission units code-multiplexes the code element for each transmission cycle of the transmission signal during a first period in which a cyclically generated orthogonal code element is transmitted at least once. After the first period, a predetermined transmission gap period, does not transmit a transmission signal code-multiplexed the code element, after the transmission gap period, at least code elements of the cyclically generated orthogonal code During a second period of one-cycle transmission, each transmission signal obtained by code-multiplexing the code element is transmitted for each transmission cycle.

  Note that these comprehensive or specific aspects may be realized by an apparatus, a method, an integrated circuit, a computer program, or a recording medium, and any of a system, an apparatus, a method, an integrated circuit, a computer program, and a recording medium. It may be realized by any combination.

  According to an embodiment of the present disclosure, ambiguity of Doppler frequency can be reduced.

FIG. 2 is a diagram illustrating a configuration example of a radar device according to Embodiment 1. FIG. 4 is a diagram illustrating an example of a transmission signal generated by a radar transmission signal generation unit. FIG. 7 is a diagram for explaining timings at which transmission RF sections # 1 to # 3 according to Embodiment 1 transmit transmission signals when the number of transmission antennas N _t = 3. FIG. 5 is a diagram for explaining timings at which transmission RF sections # 1 to # 4 according to Embodiment 1 output transmission signals when the number of transmission antennas N _t = 4. FIG. 7 is a diagram for explaining timings at which transmission RF sections # 1 to # 5 according to Embodiment 1 output transmission signals when the number of transmission antennas N _t = 5. FIG. 4 is a diagram illustrating an example in which a transmission delay is provided at a transmission start time of a transmission signal of a transmission RF unit. It is a figure showing the modification of a radar transmission signal generation part. FIG. 3 is a diagram for explaining a transmission signal timing and a measurement range of discrete times. FIG. 3 is a diagram for explaining a relationship among a transmitting antenna, a receiving antenna, and a virtual receiving antenna. It is a figure showing a modification of a radar transmitting part. FIG. 9 is a diagram illustrating an example of a spatial profile result when a beamformer method is used in a direction estimating unit. FIG. 9 is a diagram illustrating a configuration example of a radar device according to a second embodiment. FIG. 9 is a diagram showing a transmission chirp pulse signal and a reflected wave signal according to Embodiment 2. FIG. 13 is a diagram illustrating a configuration example of a radar device according to a third embodiment. Transmission RF section # 1~ # N _t according to the third embodiment is a diagram for explaining the timing for transmitting a transmission signal. FIG. 13 is a diagram illustrating a configuration example of a radar device according to a fourth embodiment. FIG. 15 is a diagram for explaining timings at which transmission RF sections # 1 to # 3 according to Embodiment 5 output transmission signals when the number of transmission antennas N _t = 3. FIG. 17 is a diagram illustrating a configuration example of a radar device according to a sixth embodiment. FIG. 19 is a diagram for explaining an example of transmission timing of the radar device according to Embodiment 6. FIG. 19 is a diagram for explaining an example of transmission timing of the radar device according to Embodiment 6. FIG. 21 is a diagram illustrating a configuration example of a radar device according to a seventh embodiment.

  Hereinafter, embodiments of the present invention will be described in detail with reference to the drawings as appropriate. However, an unnecessary detailed description may be omitted. For example, a detailed description of a well-known item or a redundant description of substantially the same configuration may be omitted. This is to prevent the following description from being unnecessarily redundant and to facilitate understanding by those skilled in the art.

  The accompanying drawings and the following description are provided to enable those skilled in the art to fully understand the present disclosure, and are not intended to limit the claimed subject matter.

(Embodiment 1)
FIG. 1 shows a configuration example of a time division multiplexed MIMO radar device (hereinafter, simply referred to as “radar device”) according to Embodiment 1. The radar device 1 has a radar transmitting unit 100 and a radar receiving unit 200. Radar transmitter 100 transmits the transmission signal is switched in a time division multiple transmission antennas Tx # 1~Tx # _{N t.} The radar receiving unit 200 receives a reflection signal obtained by reflecting a transmission signal transmitted from the radar transmission unit 100 from a target (object), and estimates a direction of the target.

<Radar transmitter 100>
Next, the radar transmitting unit 100 will be described. Radar transmitter 100 includes a plurality of radar transmission signal generation unit 101, a switching control unit 105, transmits an RF switching unit 106, and the _{N t} transmission RF unit 107 #. 1 to # _{N t, N t} pieces of transmission and a antenna Tx # 1~ # _{N t.} The transmission antenna Tx # 1~ # _{N t,} may be referred to as a transmission array antenna unit.

  The radar transmission signal generator 101 includes a code generator 102, a modulator 103, and a bandpass filter (LPF: Low Pass Filter) 104.

  Transmission RF switching section 106 selects one of transmission RF sections 107 based on the switching control signal output from switching control section 105. Then, transmission RF switching section 106 outputs the baseband transmission signal output from the radar transmission signal generation section to selected transmission RF section 107.

  The transmission RF unit 107 selected by the transmission RF switching unit 106 converts the frequency of the baseband transmission signal output from the transmission RF switching unit 106 into a predetermined radio frequency band, and is connected to the transmission RF unit 107. To the transmitting antenna Tx.

Transmission antenna Tx # 1~ # _{N t} are respectively connected to the transmission RF section 107 # 1~ # _{N t.} The transmission antenna Tx radiates the transmission signal output from the transmission RF unit 107 into space.

  Next, the operation of the radar transmitting unit 100 will be described in detail.

The radar transmission signal generation unit 101 generates a timing clock obtained by multiplying the reference signal output from the reference signal generator Lo by a predetermined number, and generates a transmission signal based on the generated timing clock. Then, the radar transmission signal generation unit 101 outputs a transmission signal at every predetermined transmission cycle _Tr . Transmission _{signal, y (k t, M)} = I (k t, M) + jQ (k t, M) is represented by. Here, j denotes an imaginary unit, k _t denotes discrete time, M represents the ordinal number of the transmission cycle. Further, I _(k t, M) and Q _(k t, M) is at the discrete time _{k t} of the M-th transmission cycle _{T r,} the transmission signal y _(k t, M) phase component (an In-Phase Component) and a quadrature component (Quadrature component).

Code generator 102 generates the sign of the code sequence of the code length L _a n (M) in the M-th transmission cycle _{T r (n = 1, ...} , L). The code a _{n (M),} using a pulse code the low range side lobe properties. Examples of the code sequence include a Barker code, an M-sequence code, and a Gold code.

Modulator 103, a pulse modulation to the code output from the code generation unit _a n (M) (amplitude modulation, ASK (Amplitude Shift Keying), pulse shift keying) or phase modulation of (PSK (Phase Shift Keying)) Apply. Then, the modulation unit outputs a signal (modulated signal) on which pulse modulation has been performed to the LPF 104.

  LPF 104 extracts a signal component equal to or less than a predetermined limited band from the modulated signal output from modulating section 103 and outputs the signal component to transmission RF switching section 106 as a baseband transmission signal.

  FIG. 2 shows a transmission signal generated by the radar transmission signal generation unit 101.

Of section of transmission cycle _{T r,} code transmission period _{T w} is present signal, the signal is absent remaining _(T r -T _w) section. That is, the ( _Tr - _Tw ) section is a non-signal section. In the code transmission period T _w, include pulse code pulse code length L. The one pulse code includes L number of sub-pulses per sub-pulses, the pulse modulation using N _o samples are subjected. Thus, in the code transmission period _{T w,} include signal _N r ₍₌ N o L) samples. That is, the sampling rate in the modulation unit is ( _NoL ) / _Tw . Further, in the no signal section _(T r -T _w), include _{N u} samples.

  The switching control unit 105 outputs a switching control signal for instructing the transmission RF switching unit 106 of the radar transmitting unit 100 and the output switching unit 211 of the radar receiving unit 200 to switch the output destination. Note that an instruction to switch the output destination to the output switching unit 211 will be described later (see the description of the operation of the radar receiving unit 200). Hereinafter, an instruction to switch the output destination to the transmission RF switching unit 106 will be described.

Switching control unit 105, for each transmission cycle _{T r,} from the transmission RF unit 107 #. 1 to # _{N t,} the transmission RF unit 107 selects one for use in the transmission of the transmission signal. Then, switching control section 105 outputs a switching control signal instructing transmission RF switching section 106 to switch the output destination to selected transmission RF section 107.

Transmitting RF switching unit 106, based on the switching control signal outputted from the switching control unit 105, an output destination is switched to one of the transmission RF section 107 # 1~ # _{N t.} Then, transmission RF switching section 106 outputs the transmission signal output from radar transmission signal generation section 101 to transmission RF section 107 at the switching destination.

Here, the switching control unit 105 determines that the transmission interval of the transmission signal of at least one of the transmission RF units 107 among the N _t transmission RF units 107 is longer than the transmission interval of the transmission signal of each of the other transmission RF units 107. A short switching control signal is output to transmission RF switching section 106. Note that the transmission intervals of the at least one transmission RF section 107 may be equal. In other words, the switching control unit 105 selects the at least one transmission RF unit 107 in a shorter cycle than each of the other transmission RF units 107. Hereinafter, the transmission RF unit 107 selected in the short cycle may be referred to as a “short-cycle transmission RF unit”. A transmission signal transmitted by the short-period transmission RF unit may be referred to as a “short-period transmission signal”.

  Hereinafter, a specific example will be described with reference to FIGS. 3, 4, and 5. FIG.

FIG. 3 is a diagram for explaining the timing at which the transmission RF sections 107 # 1 to # 3 transmit transmission signals when the number of transmission antennas _Nt = 3. FIG. 3 is an example in which transmission RF section 107 # 2 is a short-period transmission RF section.

In this case, transmission RF section 107 # 2 outputs a transmission signal every 2Tr cycles. Transmission RF sections 107 # 1 and # 3 sequentially output transmission signals in each _Tr period in which transmission RF section 107 # 2 does not output transmission signals. In other words, transmission RF sections 107 # 1 and # 3 each output a transmission signal every N _p = 4T _r = 2 (N _t −1) T _r periods.

FIG. 4 is a diagram for explaining the timing at which transmission RF sections 107 # 1 to # 4 output a transmission signal when the number of transmission antennas _Nt = 4. FIG. 4 shows an example in which transmission RF section 107 # 2 is a short-period transmission RF section.

In this case, the transmission RF section 107 # 2 outputs the transmission signal every 2T _r cycle. Transmission RF sections 107 # 1, # 3, and # 4 sequentially output transmission signals in each _Tr period in which transmission RF section 107 # 2 does not output transmission signals. That is, transmission RF section 107 # 1, # 3, # 4, _{respectively, N p = 6T r = 2} (N t -1) every _{T r} period, and outputs a transmission signal.

FIG. 5 is a diagram for explaining the timing at which the transmission RF units 107 # 1 to # 5 output transmission signals when the number of transmission antennas _Nt = 5. FIG. 5 is an example in which transmission RF section 107 # 2 is a short-period transmission RF section.

In this case, the transmission RF section 107 # 2 outputs the transmission signal every 2T _r cycle. Transmission RF sections 107 # 1, # 3, # 4, and # 5 sequentially output transmission signals in each _Tr period in which transmission RF section 107 # 2 does not output transmission signals. That is, transmission RF sections 107 # 1, # 3, # 4, and # 5 each output a transmission signal every N _p = 8T _r = 2 (N _t −1) T _r periods.

Switching control unit 105, the switching process of the above-described output _{destination, N p = 2 (N t} -1) T r period _{N c} times, is repeated. In this _N p _{N c} period, transmission RF section 107 # 2 (short-cycle RF transmitter), since the 2T _{r period,} (N t -1) _{N c} times, and outputs a transmission signal. The transmission RF section 107 # the transmission RF unit 107 other than 2, for _{N p} cycles, _{N c} times, and outputs a transmission signal.

  The transmission RF section 107 having received the transmission signal from the transmission RF switching section 106 outputs the transmission signal to the transmission antenna Tx connected to the transmission RF section 107. For example, the transmission RF unit 107 performs frequency conversion on a baseband transmission signal output from the radar transmission signal generation unit 101 to generate a carrier signal (RF (Radio Frequency)) band transmission signal, and transmits the signal. The signal is amplified to a predetermined transmission power P [dB] by an amplifier and output to the transmission antenna Tx.

  The transmission antenna Tx radiates a transmission signal output from the transmission RF unit 107 connected to the transmission antenna Tx to space.

Note that the transmission start time of the transmission signal of each transmission RF section 107 does not necessarily have to be synchronized with the cycle _Tr . For example, as shown in FIG. 6, to the transmission start time of each transmission RF section 107, transmission delay delta _1, delta _2, ..., it may be provided .DELTA.N _t. That is, in each of the transmission RF units 107, the delay of the output timing of the transmission signal may be different. Next, further description will be made with reference to FIG.

6, transmission start time of the transmission signal of the transmission RF section 107 # 1 is after the transmission delay delta ₁ elapsed from the start time of T _r period. Similarly, transmission start time of the transmission signal of the transmission RF section 107 # 2 is after the transmission delay delta ₂ elapsed from the start time of T _r period. Transmission start time of the transmission signal of the transmission RF section 107 # 3 is after the transmission delay delta ₃ elapsed from the start time of T _r period.

When the transmission delays Δ ₁ , Δ ₂ ,..., _ΔNt are provided, as described later, in the processing of the radar receiving unit 200, the transmission delays Δ ₁ , Δ _{2 ,.} A corrected coefficient may be introduced. With this, it is possible to remove the influence of different phase rotation when the Doppler frequency is different (details will be described later).

Further, each time the target is measured, the transmission delays Δ ₁ , Δ ₂ ,..., _ΔNt may be changed. Thereby, when receiving interference from another radar or giving interference to another radar, it is possible to randomize the influence of interference with other radars.

  Next, a modified example of the radar transmission signal generation unit 101 will be described with reference to FIG.

  As shown in FIG. 7, the radar transmission signal generation unit 101 may have a configuration including a code storage unit 111 and a D / A conversion unit 112. The code storage unit 111 previously stores the code sequence generated by the code generation unit 102, and reads out the code sequence cyclically. The D / A converter 112 converts a digital signal into an analog signal. That is, according to the configuration shown in FIG. 7, radar transmission signal generation section 101 converts the output of code storage section 111 into an analog baseband transmission signal and outputs the signal to transmission RF section 107.

<Radar receiver>
Next, the radar receiving unit 200 will be described. Radar receiver 200, a _{N a} number of receiving antennas Rx #. 1 to # _{N a,} and _{N a} number of antenna systems processing unit 201 #. 1 to # _{N a,} a CFAR section 215, the direction estimating section 214, the Have. The receiving antennas Rx # 1~ # _{N a,} may also be referred to as the receiving array antenna unit. One receiving antenna Rx is associated with one antenna system processing unit 201. That is, the reception antenna Rx # z (z = 1, ..., N a) relative to the antenna system processing unit 201 # z is associated. Each antenna system processing unit 201 includes a reception radio unit 203 and a signal processing unit 207.

  Each receiving antenna Rx receives a reflected signal obtained by reflecting a transmission signal transmitted from the radar transmission unit 100 from a target. The reception antenna Rx outputs the received signal to the reception radio unit 203 of the antenna system processing unit 201 associated with the reception antenna Rx. Receiving radio section 203 outputs the received signal to signal processing section 207 belonging to the same antenna system processing section 201.

  The reception radio section 203 includes an amplifier 204, a frequency conversion section 205, and a quadrature detection section 206. The reception radio section 203 performs signal amplification by the amplifier 204 on the reception signal output from the reception antenna Rx. Then, the reception radio unit 203 converts the received signal into a baseband reception signal including an I signal component (In-Phase signal component) and a Q signal component (Quadrature signal component) by the frequency conversion unit 205 and the quadrature detection unit 206. And a baseband received signal including

The signal processing unit 207, an A / D converter 208, an A / D conversion unit 209, a correlation calculation unit 210, an output switching unit 211, and an _{N t} pieces of the Doppler analysis unit 213 #. 1 to # _{N t} Have. Next, each functional block will be described.

A / D conversion section 208 performs sampling at discrete times on the baseband reception signal including the I signal component output from reception radio section 203, and converts it into digital data. Further, the A / D converter 209 performs sampling at discrete times on the baseband received signal including the Q signal component, and converts the signal into digital data. Here, the sampling rate of the A / D converter 208 and 209, performs sub-pulse time _{T p (= T w / L} ) per the transmit signal, the _{N s} number of discrete samples. In other words, the number of over-samples per sub-pulses becomes N _s number.

In the following, the baseband reception signal _Iz (k, M) including the I signal component and the Q signal component received by the reception antenna Rx # z at the discrete time k in the M-th transmission cycle _Tr are described. baseband received signal _Q z (k, M) comprising a, using a complex _number, representing _{x z (k, M) =} I z (k, M) + jQ z (k, M) and. Here, j is an imaginary unit.

In the following, the discrete time k is based on the start timing of the transmission cycle _Tr (k = 1). Then, the signal processing unit, the transmission cycle _{T r} until sampling points is _{k = (N r + N u} ) N s / N o and before ending, periodically operate. In other words, k = _1, ..., a _{(N r + N u) N} s / N o.

  Note that the reference clock signal of the reception radio section 203 and the signal processing section 207 may be a signal obtained by multiplying a reference signal from the same reference signal generator Lo as the radar transmission signal generation section 101 by a predetermined number. Thereby, the operations of the radar transmission signal generation unit 101 and the reception radio unit 203 and the signal processing unit 207 of the radar reception unit 200 are synchronized.

The correlation calculation section 210 in the antenna system processing section 201 # z determines whether the discrete sample values x _z (k, M) output from the A / D conversion sections 208 and 209 and the radar transmission section 100 for each transmission cycle _Tr. performing a pulse code a _n of the transmitted code length L _(M), a correlation calculation. Here, z = 1, ..., a _{N a, n = 1, ...} , a L. For example, the correlation calculator 210 calculates the sliding correlation between the discrete sample value x _z (k, M) and the transmission pulse code a _n (M) in the M-th transmission cycle _Tr based on the following equation (1). Is calculated. In Equation (1), AC _z (k, M) indicates a correlation operation value at discrete time k. An asterisk (*) represents a complex conjugate operator. _Here, the calculation of AC z (k, M) is, k = _1, ..., occurs over a period of _{(N r + N u) N} s / N o.

Incidentally, the correlation calculation section 210, k = _1, ..., without being limited to the case of performing the correlation operation for _{(N r + N u) N} s / N o, in accordance with the existing range of the target, the measurement range (i.e. , K) may be limited. Thereby, the amount of calculation processing in correlation calculation section 210 can be reduced. For example, correlation calculation section _{210, k = N s (L +} 1), ..., may be limited measurement range to _{(N r + N u) N} s / N o -N s L. In this case, as shown in FIG. 11, the radar device 1 is the time interval corresponding to a code transmission period T _w is not measured.

  Thus, even in the case where the transmission signal goes directly to the radar receiving unit 200, the processing performed by the correlation calculation unit is not performed in the period in which the transmission signal goes around (at least the period less than τ1 in FIG. 8). Therefore, it is possible to perform measurement without the influence of the wraparound. Further, when the measurement range (range of k) is limited, the process of limiting the measurement range (range of k) is similarly applied to the processes of the Doppler analysis unit 213 and the direction estimation unit 214 described below. do it. Thereby, the processing amount in each block can be reduced, and the power consumption in the radar receiving unit 200 can be reduced.

Output switching unit 211, based on the switching control signal outputted from the switching control unit 105, for each transmission cycle T _r, selects one of the N _t Doppler analysis unit 213. Then, output switching section 211 outputs the correlation operation result output from correlation operation section 210 for each transmission cycle _Tr to selected Doppler analysis section 213.

Switching control signal in the M-th transmission cycle _{T r} is, _{N t} bits _{[bit 1 (M), bit} 2 (M), ..., bit Nt (M)] may be constituted by. In this case, when the ND bit is 1 in the switching control signal of the M-th transmission cycle _Tr , the output switching unit 211 selects the ND-th Doppler analyzer 213 as the output destination, and sets the ND bit to In the case of 0, the ND-th Doppler analyzer 213 is not selected as the output destination (it is not selected). It should be noted, ND = 1, ..., a _{N t.}

When the number of transmission antennas N _t = 3, the switching control unit 105 outputs, for example, a 3-bit switching control signal shown in the following (A1) to correspond to the output pattern of the transmission signal shown in FIG. Output to 211.
(A1)
[Bit ₁ (1), bit ₂ (1), bit ₃ (1)] = [0, 1, 0]
[Bit ₁ (2), bit ₂ (2), bit ₃ (2)] = [1, 0, 0]
[Bit ₁ (3), bit ₂ (3), bit ₃ (3)] = [0, 1, 0]
[Bit ₁ (4), bit ₂ (4), bit ₃ (4)] = [0, 0, 1]

That is, the switching control unit 105, 1 (ON) becomes bit ₂ (M) is in each 2T _r period, bit ₂ (M) other than the bit ₁ (M) and bit ₃ (M), _{respectively, N} p = _{4T r = 2 (N t -1} ) sequentially output 1 to become a switching control signal for each _{T r} period. The switching control unit 105 repeats one set shown in the above (A1) _Nc times.

When the number of transmission antennas N _t = 4, the switching control unit 105 outputs, for example, a 4-bit switching control signal shown in the following (A2) so as to correspond to the output pattern of the transmission signal shown in FIG. Output to 211.
(A2)
[Bit ₁ (1), bit ₂ (1), bit ₃ (1), bit ₄ (1)] = [0, 1, 0, 0]
[Bit ₁ (2), bit ₂ (2), bit ₃ (2), bit ₄ (2)] = [1, 0, 0, 0]
[Bit ₁ (3), bit ₂ (3), bit ₃ (3), bit ₄ (3)] = [0, 1, 0, 0]
[Bit ₁ (4), bit ₂ (4), bit ₃ (4), bit ₄ (4)] = [0, 0, 1, 0]
[Bit ₁ (5), bit ₂ (5), bit ₃ (5), bit ₄ (5)] = [0, 1, 0, 0]
[Bit ₁ (6), bit ₂ (6), bit ₃ (6), bit ₄ (6)] = [0, 0, 0, 1]

That is, the switching control unit 105, bit ₂ (M) becomes 1 every 2T _r period, bit ₂ (M) other than the _{bit 1 (M), bit 3} (M) and bit ₄ (M), respectively, _{N p = 6T r = 2 (} N t -1) sequentially output 1 to become a switching control signal for each _{T r} period. The switching control unit 105 repeats one set shown in the above (A2) _Nc times.

When the number of transmitting antennas is _Nt = 5, the switching control signal is, for example, a 5-bit switching control signal shown in the following (A3), which corresponds to the output pattern of the transmission signal shown in FIG. Output to
(A3)
[Bit ₁ (1), bit ₂ (1), bit ₃ (1), bit ₄ (1), bit ₅ (1)] = [0,1,0,0,0]
[Bit ₁ (2), bit ₂ (2), bit ₃ (2), bit ₄ (2), bit ₅ (2)] = [1, 0, 0, 0, 0]
[Bit ₁ (3), bit ₂ (3), bit ₃ (3), bit ₄ (3), bit ₅ (3)] = [0,1,0,0,0]
[Bit ₁ (4), bit ₂ (4), bit ₃ (4), bit ₄ (4), bit ₅ (4)] = [0,0,1,0,0]
[Bit ₁ (5), bit ₂ (5), bit ₃ (5), bit ₄ (5), bit ₅ (5)] = [0,1,0,0,0]
[Bit ₁ (6), bit ₂ (6), bit ₃ (6), bit ₄ (6), bit ₅ (6)] = [0,0,0,1,0]
[Bit ₁ (7), bit ₂ (7), bit ₃ (7), bit ₄ (7), bit ₅ (7)] = [0,1,0,0,0]
[Bit ₁ (8), bit ₂ (8), bit ₃ (8), bit ₄ (8), bit ₅ (8)] = [0,0,0,0,1]

That is, the switching control unit 105, bit ₂ (M) becomes 1 every 2T _r period, bit ₂ (M) other than the _{bit 1 (M), bit 3} (M), bit 4 (M) and bit ₅ ( M), _{respectively, N p = 8T r = 2} (N t -1) sequentially output 1 to become a switching control signal for each _{T r} period. The switching control unit 105 repeats one set shown in the above (A3) _Nc times.

Antenna system processor 201 # z of the signal processing unit 207 includes a Doppler analysis unit 213 # 1~ # _{N t.} The Doppler analysis unit 213 performs Doppler analysis on the correlation operation result output from the output switching unit 211 at each discrete time k. That is, the Doppler analyzer 213 analyzes the Doppler frequency component of each received signal corresponding to each transmitted signal. In Doppler analysis, if _Nc is a power of 2, FFT processing as shown in Expressions (2) and (3) can be applied.

^{_{Here, FT_CI z ND (k, f}} s, w) is an antenna system processing unit 201 # z (i.e., receiving antennas Rx # corresponding to z) w from the Doppler analysis unit 213 # ND in the signal processing unit 207 a second output, indicating a Doppler frequency response of the Doppler frequency index f _s at the discrete time k. It should be noted, is a _{ND = 1~N t, k = 1} , ..., a _{(N r + N u) N} s / N o, z = 1, ..., a _{N a.} W is a natural number.

  In the FFT processing, a window function coefficient such as a Han window or a Hamming window may be multiplied. By applying the window function, side lobes generated around the beat frequency peak can be suppressed.

When ND = 2 (in the case of a short-period received signal), the FFT size of the Doppler analysis is (N _t −1) N _c , and the maximum Doppler frequency at which aliasing derived from the sampling theorem does not occur is ± 1/1/2. (4T _r ). Also, Doppler frequency interval of the Doppler frequency index _{f s} is _{1 / {2 (N t -1} ) N c T r}, a range of Doppler frequency index _{f s} is _{f s = - (N t -1} ) N c .., 0,..., (N _t −1) N _c / 2.

When ND の 2 (when the signal is not a short-period reception signal), the FFT size of the Doppler analysis is _Nc , and the maximum Doppler frequency at which aliasing derived from the sampling theorem does not occur is ± 1 / ｛4 (N _t −1). ) T _r }. Also, Doppler frequency interval of the Doppler frequency index _{f u} is _{1 / {2 (N t -1} ) N c T r}, a range of Doppler frequency index _{f u} is _{f u = -N c / 2 +} 1, ..., 0 ,..., N _c / 2.

Comparing the outputs from the Doppler analyzer 213 in the case of ND = 2 and the case of ND ≠ 2, the Doppler frequency intervals of both are the same. However, the maximum Doppler frequency at which aliasing does not occur when ND = 2 is multiplied by ± (N _t −1) compared to the case where ND ≠ 2, and the Doppler frequency range is expanded to (N _t −1) times. Have been.

Accordingly, a transmission antenna for outputting the transmission signal Tx # 1, Tx # 2, ..., compared to the case of switching sequence as Tx # N _t, the configuration of setting the short period transmission antenna Tx # 2 as described above If the maximum Doppler frequency wrapping ND = 2 does not occur, in the case of 3 or more the number of transmission antennas N _t, expanded to N t _{/ 2} times. In other words, in proportion to the number of transmit antennas N _t, the Doppler frequency range is extended wrapping does not occur.

In the case of ND ≠ 2, if there is no output from the output switching unit 211, the FFT size of the Doppler analysis is set to (N _t −1) N _c, and virtually zero output is obtained using Expression (4). May be sampled. Equation (4) is the same as equation (2). As a result, the FFT size increases, so that the processing amount increases. However, since the Doppler frequency index matches the case of ND = 2, the later-described Doppler frequency index conversion processing becomes unnecessary.

The CFAR unit 215 adaptively sets (adjusts) a threshold using the short-cycle received signal, and performs a peak signal detection process. That is, the CFAR unit 215 detects a peak signal by CFAR (Constant False Alarm Rate) processing. Thus, CFAR unit 215 detects the discrete time index k_ _CFAR and Doppler frequency index _{f S_cfar} gives a peak signal. In this embodiment, transmission RF section 107 # 2 is an example of outputting the short-period transmission signal with a period 2T _r. Therefore, CFAR unit 215, FT_CI ₁ is a w-th output from the Doppler analysis unit 213 # 2 of the antenna system processing unit ^{_{201 # 1~ # N a (2}} ) (k, f s, w), ..., ^{_{FT_CI Na (2) (k,}} f s, w) using, performs CFAR processing.

CFAR unit 215, as shown in Equation (5), w th output FT_CI ₁ ^{(2) (k} from the Doppler analysis unit 213 # 2 of the antenna system processing unit _{201 # 1~ # N a, f} s, ^{_{w), ..., FT_CI Na (}} 2) (k, f s, w) to the power addition. However, ND = 2 in equation (5). Then, the CFAR unit 215 performs, for example, a CFAR process combining a one-dimensional CFAR process or a two-dimensional CFAR process on the power addition result. The processing disclosed in Non-Patent Document 2 may be applied to this CFAR processing. Here, in the two-dimensional CFAR processing, an axis of discrete time (corresponding to the distance to the target) and an axis of Doppler frequency (corresponding to the relative speed of the target) may be used.

Alternatively, CFAR unit 215, as shown in equation (6), on the received signal from the discrete time k and Doppler frequency index _{f s} common reception antenna Rx # 1~ # _{N a,} the main beam direction θ Multiply the directional weight W (θ) = [w ₁ (θ), w ₂ (θ),..., W _Na (θ)]. However, ND = 2 in the equation (6). Then, the CFAR unit 215 performs CFAR processing combining one-dimensional CFAR processing or two-dimensional CFAR processing for each of a plurality of directional beam directions. Here, the axis of the discrete time k and the axis of the Doppler frequency may be used for the two-dimensional CFAR processing.

The CFAR unit 215 adaptively sets a threshold, and outputs the ND = 2 discrete time index _{k_cfar} and the Doppler frequency index f _{s_cfar} that have received power greater than the threshold to the direction estimation unit 214. Further, CFAR unit 215, a wide ND = 2 Doppler frequency index _{f S_cfar} the Doppler frequency range, Doppler analysis unit 213 # each Doppler analysis unit 213 # 1 other than 2, # 3, ..., w from # _{N t} th output ^{_{FT_CI 1 (ND ≠ 2) (}} k, f u, w), ..., FT_CI Na (ND ≠ 2) (k, f u, w) in order to correspond to the Doppler frequency index _{f u} of index conversion I do. The index conversion may be performed by Expressions (7) and (8). The CFAR unit 215 outputs the Doppler frequency index _{fu_cfar} after the index conversion to the direction estimating unit 214. That is, the CFAR unit 215 detects a peak Doppler frequency component, which is a frequency component whose received power is greater than the threshold, from the Doppler frequency component of the received signal.

_{Here, f s_cfar = - (N t} -1) N c / 2 + 1, ..., 0, ..., (N t -1) is _{N c / 2, f u_cfar =} -N c / 2 + 1, ..., 0, ..., _Nc / 2.

Hereinafter, in the present embodiment, the Doppler frequency index f _{s_cfar} having a wide Doppler frequency range in which ND = 2 is _referred to as a wide range Doppler frequency index _{fs_cfar} . In the present embodiment, a Doppler frequency index f _u having a narrow Doppler frequency range of NDＮ2 is expressed as a narrow range Doppler frequency index f _u . When the wide-range Doppler frequency index f _{s_cfar corresponds} to the narrow-range Doppler frequency index f _u , an overlap may be included.

For example, when the wide-range Doppler frequency index f _{s_cfar} includes a Doppler frequency index α in the range of 0 ≦ α ≦ N _c / 2, it is converted to α by an index conversion corresponding to the narrow-range Doppler frequency index f _u . Here, a wide range Doppler frequency index _{f S_cfar,} when also includes β = _{α-N c,} β, since it is within the scope of _{-N c ≦ β ≦ -N c /} 2, the narrow range Doppler frequency index f _{By the} index conversion corresponding to _u , β + N _c = α is converted. Therefore, in the index conversion that makes the wide-range Doppler frequency index _{fs_cfar correspond} to the narrow-range Doppler frequency index f _u , overlap occurs.

Similarly, a wide range Doppler frequency index _{f s_cfar,} β = _α + _N when _c is also included, beta, since it is within the scope of _{N c ≦ β ≦ 3N c /} 2, to correspond to a narrow range Doppler frequency index _{f u} By the index conversion, β + N _c = α is converted. Therefore, the index conversion to correspond to a narrow range Doppler frequency index f _u, overlapping occurs.

Thus, a wide range Doppler frequency index _f s_cfar, | α-β | When alpha relationship becomes an integral multiple of N _c, include beta is, when made to correspond to a narrow range Doppler frequency index f _u, duplicates Occur.

When duplicate narrow range Doppler frequency index f _u is occurring, the signal component of the narrow range Doppler frequency index f _u is in a state in which signals of different Doppler frequency components have been mixed. The closer the power of the mixed signal is, the more the amplitude / phase component fluctuates, and the accuracy of the angle measurement in the subsequent direction estimating unit 214 may deteriorate. Therefore, in the present embodiment, an overlap determination process is introduced. This suppresses the influence of the direction estimating unit 214 causing deterioration in the accuracy of the side angle. Next, the overlap determination processing will be described.

<Duplicate determination process>
The Doppler frequency index α and the Doppler frequency index β of the wide-range Doppler frequency index f _{s_cfar} extracted by the CFAR processing are _converted into the w-th output FT_CI ₁ ^{(ND ≠ 2)} from the Doppler analyzer 213 other than the Doppler analyzer 213 # ^{2. _{) (k, f u, w}} ), ..., to index conversion correspond to ^{_{FT_CI Na (ND ≠ 2) (}} k, f u, w Doppler frequency index _{f u)} of. When the converted Doppler frequency index _{fu_cfar} overlaps, the following processes (B1) to (B3) are performed.

(B1) The CFAR unit 215 outputs FT_CI ₁ ^{(ND = 2)} (k, α, w),..., FT_CI _Na ^{(ND = 2)} (k, α), which is the w-th output from the Doppler analysis unit 213 # 2. , W) and the power sum of FT_CI ₁ ^{(ND = 2)} (k, β, w),..., FT_CI _Na ^{(ND = 2)} (k, β, w).

  (B2) As a result of the comparison of the power sum in (B1), if there is a power difference equal to or more than a predetermined value (for example, set to about 6 to 10 dB), the CFAR unit 215 has a larger power among the Doppler frequency indexes α and β. The lower Doppler frequency index is made valid, and the lower Doppler frequency index is excluded from the output target to the direction estimation unit 214.

  (B3) The CFAR unit 215 excludes both the Doppler frequency indexes α and β from the output target to the direction estimating unit 214 when there is no power difference equal to or more than the predetermined value as a result of the power sum comparison of (B1). .

Based on the discrete time index _{k_cfar} , the Doppler frequency index f _{s_cfar} , and the Doppler frequency index _{fu_cfar} output from the CFAR unit 215, the direction estimation unit 214 estimates the direction of the target using the output from each Doppler analysis unit 213. Perform processing. Specifically, the direction estimating unit 214, the virtual reception array correlation vector h shown in equation (9) (k, f s , w) generates, performs direction estimation process.

In, it summarizes the w-th output from the antenna system processing unit 201 # 1 to # _{N a} Doppler analysis unit 213 # 1 obtained by performing similar processing in the signal processing units 207 of # _{N t} or less the formula contains a _N t _{N a} number of factors, such as the product of number of transmit antennas _{N t} and the number of reception antennas _{N a} as shown in (9), the virtual reception array correlation vector _{h (k _cfar, f s_cfar,} w) Notation as The virtual reception array correlation vector h ( _{k_cfar} , _{fs_cfar} , w) is used for processing for estimating the direction of the reflection signal from the target based on the phase difference between the reception antennas Rx. Here, z = 1, ..., a _{N a, ND = 1, ...} , a _{N t.}

h _{cal [b]} is an array correction value for correcting a phase deviation and an amplitude deviation between the transmission array antennas and between the reception array antennas. b = _1, ..., a _N t N _a.

Further, since the transmission antenna Tx is switched in a time division manner, different phase rotations occur at different Doppler frequencies f. TxCAL ⁽¹⁾ (f),..., TxCAL ^(Nt) (f) are transmission phase correction coefficients for correcting the phase rotation to match the phase of the reference transmission antenna. For example, the number of transmit antennas N _{t =} 3 as shown in FIG. 3, when the transmitting antenna Tx # 2 as a reference transmission antenna, transmission phase correction coefficient becomes equation (10).

Further, the number of transmit antennas N _{t =} 4, as shown in FIG. 4, or, if the number of transmit antennas N _{t =} 5, as shown in FIG. 5, and the transmission antenna Tx # 2 as a reference transmission antenna, transmission phase correction factor Becomes the equation (11).

Incidentally, the transmission delay delta ₁ which different transmission start time of the transmission signal of each transmission RF unit 107, delta _2, ..., the case of providing the delta _Nt, transmission phase correction coefficient ^{TxCAL (ND) (f),} the formula (12) _{May be} multiplied by a correction coefficient Δ _TxCAL ^(ND) (f) as a new transmission phase correction coefficient TxCAL ^(ND) (f). As a result, the influence of the phase rotation that differs depending on the Doppler frequency can be removed. Here, the delta _ref, represent the transmission delay of the reference transmitting antenna number to a phase reference, in the present embodiment, since the reference transmitting antenna is a Tx # 2, the Δ _{ref =} Δ _2.

Virtual reception array correlation vector _{h (k _cfar, f s_cfar,} w) _is a column vector comprised of N a _{N r} number of elements.

DOA estimation is the direction estimation evaluation function value _{P H (θ, k _cfar,} f s_cfar, w) to calculate the spatial profile of the azimuth direction theta in a variable within a predetermined angular range. In the direction of arrival estimation, a predetermined number of maximal peaks in the spatial profile are extracted in descending order, and the elevation direction of each maximal peak is output as an estimated direction of arrival.

The direction estimation evaluation function value P _H (θ, _{k_cfar} , _{fs_cfar} , w) may be calculated by an arrival direction estimation algorithm. The direction-of-arrival estimation algorithm includes various methods such as a beamformer method, Capon or MUSIC. For example, an estimation method using an array antenna disclosed in Non-Patent Document 3 may be used.

As illustrated in FIG. _9, N t _{N if a} number of virtual reception array are arranged linearly at equal intervals _{d H (N t = 3,} N a = 4), the beam former method, the formula (13 ) And Equation (14). Here, the superscript H is a Hermitian transpose operator. a (θ _u ) indicates the direction vector of the virtual reception array with respect to the arriving wave in the azimuth direction θ. θ _u is obtained by changing the azimuth range for estimating the direction of arrival at a predetermined azimuth interval β ₁ . For example, θ _u is set as follows.
θ _u = θ _min + uβ ₁
Here, u = 0,..., NU, and NU = floor [(θ _max −θ _min ) / β ₁ ] +1. Floor (x) is a function that returns the largest integer value that does not exceed the real number x.

The time information (discrete time) _{k_cfar} may be converted into distance information and output. For example, the time information _{k_cfar} is _converted into distance information R ( _{k_cfar} ) using Expression (15). Here, T _w is the code transmission period, L is a pulse code length, C ₀ denotes the speed of light.

The Doppler frequency information may be converted into a relative velocity component and output. For example, the Doppler frequency index f s — _cfar may be converted to a relative velocity component v _{d according} to equation (16). Here, _{d f} is the Doppler frequency interval in the FFT processing in the Doppler analysis unit 213, in this _embodiment, a _{d f = 1 / {2 (} N t -1) N c T r}. Λ is the wavelength of the carrier frequency of the RF signal output from the transmission RF unit 107.

As described above, the radar apparatus 1 according to the first embodiment, (in the present embodiment the transmitting antenna Tx # 2) short-period transmission antenna to the transmission period of the 2T _r, other than short-period transmission antennas of each transmission antenna The transmission cycle is 2 (N _t −1) _Tr . Thus, short-period received signal, compared to the case sequentially switching the N _t transmit antennas, increases the maximum Doppler frequency aliasing does not occur (relative speed) to N _{t /} 2 times, the Doppler frequency range aliasing does not occur Expands N _t / 2 times.

In the present embodiment, radar receiving section 200 _converts the Doppler frequency index _{fs_cfar} extracted by performing the CFAR processing on the short-period received signal so as to be applied to received signals other than the short-period received signal. Then, a Doppler frequency index f _{S_cfar} for short period received signal, the received signal other than the short-period received signal using the Doppler frequency index f _{U_cfar} that the conversion, performs direction estimation process. As a result, direction estimation processing using all virtual reception arrays becomes possible.

Further, in the present embodiment, in the CFAR processing, not all received signals but short-period received signals are used, but the FFT size of Doppler analyzer 213 is multiplied by (N _t −1), so that (N A coherent addition gain of _t- 1) times is obtained. Therefore, it is possible to compensate for the SNR corresponding to the decrease in the number of receiving antennas used for CFAR processing. Specifically, as a conventional technique, sequentially switching the transmission antenna Tx # 1~ # N _t, compared with the case of CFAR processing the output of the Doppler analysis unit 213 of all virtual reception antennas and power combining, in this embodiment The reception SNR at the time of the CFAR processing is about 0.5 (N _t ) ^1/2 (however, N _t ≧ 3). That is, the reception SNR at CFAR processing _is about 0.9 times in case of _N t = _3, becomes equal to or more _N t = 4 above, the present embodiment, as compared with conventional techniques, special degradation Does not occur.

  In the present embodiment, as shown in FIG. 10, in radar transmitting section 100, transmission antenna switching section 121 selectively switches the output from transmission RF section 107 to one of a plurality of transmission antennas Tx. It may be. In this case, the same effect as described above can be obtained.

  As shown in FIG. 9, a transmission antenna forming a virtual reception antenna (for example, within a dotted line in FIG. 9) located near the center in a virtual array of a plurality of virtual reception antennas is selected as the short-period transmission antenna Tx May be. Thereby, the effect of reducing the side lobe on the angle profile in the direction estimation processing can be obtained. Next, a specific example will be described.

Figure 9 shows an example of an antenna arrangement of the MIMO radar when the number of transmit antennas _N t = 3, the number of reception antennas _N a = 4. FIG. 11 shows an example of the spatial profile result (true direction of the target direction of 0 °) when the beamformer method is used in the direction estimating section 214.

  FIG. 11A shows a spatial profile result when the transmission antennas Tx # 1, Tx # 2, and Tx # 3 are sequentially switched according to the conventional method. FIG. 11 (b) shows a spatial profile result when transmitting antenna Tx # 2 is a short-period transmitting antenna as in the present embodiment. As shown in FIGS. 11A and 11B, both can correctly estimate the target in the front direction.

  Also, comparing FIGS. 11A and 11B, in FIG. 11B corresponding to the present embodiment, the effect (about 3 dB) of reducing the side lobe of the beam former method is obtained. You can check. This is for the following reason. That is, in virtual receiving antennas (arrangement of MIMO VAs # 1 to # 12 in FIG. 9), MIMO VAs # 5 to # 8 arranged near the center receive short-period transmission signals from Tx # 2. Therefore, the received signal levels of MIMO VAs # 5 to # 8 are higher than the received signal levels of other MIMO VAs # 1 to # 4 and # 9 to # 12, so that the side lobe reduction effect of the spatial profile can be obtained. .

(Embodiment 2)
In the first embodiment, a case has been described where the radar transmitting section 100 uses a pulse compression radar that transmits a pulse train by phase modulation or amplitude modulation. In the second embodiment, a radar system using a pulse compression wave that has been frequency-modulated (fast chirp modulation) such as a chirped pulse will be described. In the second embodiment, description of the same contents as in the first embodiment will be omitted.

  FIG. 12 shows a configuration example of a radar apparatus 1 using a chirp pulse in radar transmission.

Radar transmitter 100, a radar transmission signal generation unit 101, a directional coupler 124, a transmission RF section 107, a transmission antenna switching unit 121, and a plurality of transmitting antennas Tx # 1~ # _{N t,} the switching control unit 105. The radar transmission signal generator 101 includes a modulation signal generator 122 and a VCO (Voltage Controlled Oscillator) 123.

As shown in FIG. 13A, the modulation signal generator 122 periodically generates a sawtooth modulation signal. Here, the transmission cycle is assumed to be _Tr .

  VCO 123 performs frequency modulation on the transmission signal based on the output from modulation signal generation section 122, and generates a frequency modulation signal (frequency chirp signal). Then, the VCO 123 outputs the frequency modulation signal to the directional coupling unit 124.

  The directional coupling unit 124 outputs the frequency modulation signal output from the VCO 123 to the transmission RF unit 107, extracts a part of the frequency modulation signal, and outputs it to each mixer unit 224 of the radar reception unit 200.

  Transmission RF section 107 amplifies the frequency modulation signal output from directional coupling section 124 and outputs the amplified signal to transmission antenna switching section 121.

  Transmission antenna switching section 121 outputs the frequency modulation signal output from transmission RF section 107 to transmission antenna Tx switched by switching control section 105. The transmission antenna Tx radiates the transmission signal output from the transmission antenna switching unit 121 into space.

Radar receiver 200 includes a plurality of receiving antennas Rx # 1 to # _{N a,} the receiving antenna Rx # 1 to # _{N a} corresponding antenna system processing unit 201 # 1 to # _{N a,} a CFAR section 215, A direction estimating unit 214. Each antenna system processing unit 201 includes a reception radio unit 203 and a signal processing unit 207. Receiving radio section 203 has mixer section 224 and LPF 226. The signal processing unit 207 includes an A / D conversion unit 228, an R-FFT unit 220, an output switching unit 211, and a Doppler analysis unit 213.

  The radar receiving unit 200 mixes the received signal obtained by receiving the reflected signal with the receiving antenna Rx with the frequency modulation signal that is the transmission signal in the mixer unit 224 and passes the mixed signal through the LPF 226, so that the transmission signal and the received signal are A beat signal having a frequency corresponding to the delay time of the signal is extracted. For example, as shown in FIG. 13B, the difference frequency between the frequency of the transmission frequency modulation wave (radar transmission wave) and the frequency of the reception frequency modulation wave (radar reflected wave reception signal) is extracted as the beat frequency.

  The A / D conversion unit 228 of the signal processing unit 207 converts the beat signal output from the reception radio unit 203 into discrete sampling data.

The R-FFT unit 220 performs an FFT process on N _data discrete sampling data obtained in a predetermined time range (range gate) for each _Tr cycle. As a result, a frequency spectrum having a peak at a beat frequency corresponding to the delay time of the received signal is output. In the FFT processing, a window function coefficient such as a Han window or a Hamming window may be multiplied. By applying the window function, side lobes generated around the beat frequency peak can be suppressed.

Here obtained by the M-th chirped pulse transmission, the beat frequency spectrum response outputted from the R-FFT unit 220 in the antenna system processing unit 201 # z of the signal processing unit _{207, AC_RFT z (f b,} M) Expressed by Here, f _b is an index number of the FFT, and f _b = 0,..., N _data / 2. Frequency index f _b, the more the index number is small, the delay time of the received signal (reflected signal) is small represents a (i.e., a short distance from the target) beat frequency.

The output switching unit 211 performs the same operation as the output switching unit 211 of the first embodiment. That is, the output switching unit 211, based on the switching control signal from the switching control unit 105 selects one of the _{N t} Doppler analysis unit 213 #. 1 to # _{N t} (switch). Then, the output switching unit 211 outputs the frequency spectrum output from the R-FFT unit 220 to the selected Doppler analysis unit 213 for each _Tr period.

Switching control signal in the M-th transmission cycle _{T r} is, _{N t} bits _{[bit 1 (M), bit} 2 (M), ..., bit Nt (M)] may be constituted by. In this case, when the ND bit is 1 in the switching control signal of the M-th transmission cycle _Tr , the output switching unit 211 selects the Doppler analysis unit 213 # ND as the output destination, and sets the ND bit to 0. In this case, the Doppler analyzer 213 # ND is not selected as the output destination (it is not selected). It should be noted, ND = 1, ..., a _{N t.}

Switching control section 105 performs the same operation as switching control section 105 of the first embodiment. For example, if the transmission antenna Tx # 2 is a short period transmitting antennas, the transmitting antenna Tx # 2 was selected for each 2T _r period, transmission antenna Tx # each transmission antenna Tx # 1 other than 2, # 2, ..., # the _{N t N p = 2 (N} t -1) is selected for each _{T r} period.

Note that, as described in Embodiment 1, the transmission start time of the transmission signal from each transmission antenna Tx does not necessarily have to be synchronized with the cycle _Tr . For example, as shown in FIG. 6, transmission delays Δ ₁ , Δ ₂ ,..., _ΔNt may be provided for the transmission start time from each transmission antenna.

Switching control unit _105, a set of _{N p = 2 (N t -1} ) T r period, repeated _{N c} times. _Thus, in the N p _{N c} period, from the transmission antenna Tx # 2 is a short period transmitting _{antennas, (N} t -1) N _c times, the transmission signal is transmitted, the short-cycle transmit each transmit antenna other than the antenna Tx # 1, # 3, ..., from the # _{N t, N c} times, the transmission signal is transmitted.

In radar receiver 200, a signal processing unit 207 of the antenna system processing unit 201 # z has a Doppler analysis unit 213 # 1~ # _{N t.} Doppler analysis unit 213, the received signal output from the output switching unit 211, performs the Doppler analysis for each beat frequency index f _b. In the Doppler analysis, if _Nc is a power of 2, FFT processing as shown in Expressions (17) and (18) can be applied.

Here, FT_CI _z ^ND (f _b , f _s , w) is the w-th output by the Doppler analysis unit 213 # ND in the signal processing unit 207 of the antenna system processing unit 201 # z, and is represented by the beat frequency index f _b shows the Doppler frequency response of the Doppler frequency index f _s. It should be noted, is a _{ND = 1~N t, k = 1} , ..., a _{(N r + N u) N} s / N o, z = 1, ..., a _{N a.} W is a natural number.

When ND = 2 (in the case of a short-period received signal), the FFT size of the Doppler analysis is (N _t −1) N _c , and the maximum Doppler frequency at which aliasing derived from the sampling theorem does not occur is ± 1/1/2. (2T _r ). Also, Doppler frequency interval of the Doppler frequency index _{f s} is _{2 / {2 (N t -1} ) N c T r}, a range of Doppler frequency index _{f s} is _{f s = - (N t -1} ) N c .., 0,..., (N _t −1) N _c / 2.

When ND ≠ 2 (when the signal is not a short-period received signal), the FFT size of the Doppler analysis is _Nc , and the maximum Doppler frequency at which aliasing derived from the sampling theorem does not occur is ± 1 / ｛2 (N _t −1). ) T _r }. Also, Doppler frequency index _{f u} Doppler frequency interval _{2 /} of a _{{2 (N t -1) N} c T r}, the Doppler frequency index _f range _u is _{f u = -N c / 2 +} 1 ..., 0, ..., _Nc / 2.

Comparing the outputs from the Doppler analyzer 213 when ND = 2 and when ND ≠ 2, the two Doppler frequency intervals are the same. However, the maximum Doppler frequency at which aliasing does not occur when ND = 2 is multiplied by ± (N _t −1) compared to the case where ND ≠ 2, and the Doppler frequency range is expanded to (N _t −1) times. Is output.

Therefore, the transmission antenna Tx # 1, Tx # 2, ..., Tx # compared with the case where as sequentially switched in N _t, the arrangement according to this embodiment, the maximum Doppler frequency wrapping ND = 2 does not occur Expands to N _t / 2 times when the number of transmission antennas N _t is 3 or more. In other words, in proportion to the number of transmit antennas N _t, the Doppler frequency range is extended wrapping does not occur.

In the case of ND ≠ 2, if there is no output from the output switching unit 211, the FFT size of the Doppler analysis is set to (N _t −1) N _c, and virtually zero output is obtained using Expression (19). May be sampled. Equation (19) is the same as equation (17). As a result, the FFT size increases, so that the processing amount increases. However, since the Doppler frequency index matches the case of ND = 2, the later-described Doppler frequency index conversion processing becomes unnecessary.

Processing in the CFAR section 215 and the direction estimation unit 214 of the later, since the discrete time k be the same as the process was replaced with the beat frequency index f _b used in the first embodiment, and a description thereof will be omitted. With the above configuration and processing, the second embodiment can obtain the same effects as those of the first embodiment.

  Similarly, in the following embodiments, a frequency chirp signal can be applied as a transmission signal, and the same effect as in the case of using an encoded pulse signal can be obtained.

Note that the beat frequency index f _b can be converted into distance information R (f _b ) using Expression (20). Here, B _w is a frequency modulation bandwidth in the frequency chirp signals generated by frequency modulation, C ₀ is the speed of light.

(Embodiment 3)
FIG. 14 shows a configuration example of a radar device 1 according to the third embodiment. The radar device 1 according to the third embodiment further includes a loopback determination unit 216 as compared with the radar device 1 according to the first embodiment. Further, the third embodiment differs from the first embodiment in the operations of the switching control unit 105, the Doppler analysis unit 213, the CFAR unit 215, and the direction estimation unit 214. Hereinafter, in the third embodiment, description will be mainly given of contents different from the first embodiment, and description of the same contents as the first embodiment will be omitted.

  The switching control unit 105 outputs a switching control signal for instructing the transmission RF switching unit 106 of the radar transmitting unit 100 and the output switching unit 211 of the radar receiving unit 200 to switch the output destination. The instruction to switch the output destination to the output switching unit 211 will be described later. Hereinafter, an instruction to switch the output destination to the transmission RF switching unit 106 will be described.

Switching control unit 105, for each transmission cycle _{T r,} sequentially selects one of the transmission RF section 107 # 1~ # _{N t.} Then, switching control section 105 outputs a switching control signal instructing the selected transmission RF section 107 to switch the output destination to transmission RF switching section 106.

Transmitting RF switching unit 106, based on the switching control signal outputted from the switching control unit 105, an output destination is switched to one of the transmission RF section 107 # 1~ # _{N t.} Then, transmission RF switching section 106 outputs the transmission signal output from radar transmission signal generation section 101 to switching transmission RF section 107.

Figure 15 is a diagram for transmitting RF unit 107 # 1~ # _{N t} is described a timing of transmitting a transmission signal.

Switching control unit 105, as shown in FIG. 15, for each transmission cycle _{T r,} the transmission RF unit 107 # 1 to # _{N t} sequentially selects the output destination. Then, switching control section 105 outputs a switching control signal for instructing switching to selected transmission RF section 107 to transmission RF switching section 106 for each transmission cycle _Tr . Thus, the transmission RF switching unit 106, the respective transmission RF section 107 # 1~ # _{N t,} is selected in the period _N p _{= N} t _{T r.} With other words, the transmission RF unit 107 # 1~ # _{N t,} respectively, and transmits a transmission signal at a period _N p _{= N} t _{T r.}

Switching control unit 105, as shown in FIG. 15, the period _N p _{= N} t _T a set of process _{r, N} After repeating c / 2 times, transmission gap period _{T GAP} is a period that does not sends the transmission signal Is provided. Then, after the transmission gap period T _GAP has elapsed, the switching control unit 105 repeats one set of processing of the period N _p = N _{t Tr} again N _c / 2 times. By this processing, each transmission RF section 107 transmits a transmission signal _Nc times. The transmission gap period T _GAP may be set to の of the sampling period (N _p = N _{t Tr} ) of the Doppler analysis unit 213. In other words, T _GAP = N _p / 2 = N _{t Tr} / 2.

In the radar receiving unit 200, the output switching unit 211 of the signal processing unit 207 of the antenna system processing unit 201 # z outputs a Doppler analysis unit 213 for each _Tr cycle based on the switching control signal output from the switching control unit 105. # selects 1 one of # _{N t.} Then, output switching section 211 outputs the correlation operation result output from correlation operation section 210 for each _Tr cycle to selected Doppler analysis section 213.

Switching control signal in the M-th transmission cycle _{T r} is, _{N t} bits _{[bit 1 (M), bit} 2 (M), ..., bit Nt (M)] may be constituted by. In this case, when the ND bit of the switching control signal is 1 in the M-th transmission cycle _Tr , the output switching unit 211 selects the Doppler analyzer 213 # ND as the output destination, and outputs the ND-th switching control signal. When the bit is 0, the Doppler analyzer 213 # ND is not selected as an output destination (not selected). It should be noted, ND = 1, ..., a _{N t.}

Switching control unit 105, before starting the transmission gap period _{T GAP,} every N _{p =} N t _{T r} period, switches sequentially 1 each bit, and outputs a switching control signal. Next, a specific example will be described.

First, the switching control unit 105, a switching control signal below a set shown in (C1) _{(N p period),} and outputs _N c / 2 times.
(C1)
[Bit ₁ (1), bit ₂ (1), ..., bit _Nt (1)] = [1, 0, ..., 0]
[Bit ₁ (2), bit ₂ (2), ..., bit _Nt (2)] = [0, 1, ..., 0]
…
[Bit ₁ (N _t ), bit ₂ (N _t ),..., Bit _Nt (N _t )] = [0, 0,..., 1]

Switching control unit 105, after the switching control signal of the set shown in (C1) _{(N p} period) and outputs _N c / 2 times, the transmission gap period _{T GAP,} the total bit shown below (C2) It outputs a zero switching control signal.
(C2)
_{[Bit 1, bit 2, ...} , bit Nt] = [0,0, ..., 0]

Switching control unit 105, after the end of the transmission gap period _{T GAP,} the switching control signal below a set shown in (C3) _{(N p period),} and outputs _N c / 2 times.
(C3)
_{[Bit 1 (N t N c} / 2 + 1), bit 2 (N t N c / 2 + 1), ..., bit Nt (N t N c / 2 + 1)] = [1,0, ..., 0]
_{[Bit 1 (N t N c} / 2 + 2), bit 2 (N t N c / 2 + 2), ..., bit Nt (N t N c / 2 + 2)] = [0,1, ..., 0]
…
_{[Bit 1 (N t N c} ), bit 2 (N t N c), ..., bit Nt (N t N c)] = [0,0, ..., 1]

Antenna system processor 201 # z of the signal processing unit 207 includes a Doppler analysis unit 213 # 1~ # _{N t.} Doppler analysis unit 213 # 1~ # _{N t,} respectively, and the correlation operation result of _N c / 2 times prior to the start of the transmission gap period _{T GAP} (half period), after the end of the transmission gap period _{T GAP N} c / The Doppler analysis is performed separately for each discrete time k with the correlation calculation results for the two times (the latter half period). In the Doppler analysis, if _Nc is a power of 2, FFT processing as shown in Expressions (21) and (22) can be applied.

FT_FH_CI _z ^ND of formula _{(21) (k, f s} , w) is the first w-th output by Doppler analysis unit 213 # ND in the antenna system processing unit 201 # z of the signal processing section 207, at the discrete time k Doppler frequency index _{f s,} indicating a Doppler frequency response of the first half period _N c / 2 times.

FT_SH_CI _z ^ND of formula _{(22) (k, f s} , w) is the first w-th output by Doppler analysis unit 213 # ND in the antenna system processing unit 201 # z of the signal processing section 207, at the discrete time k 9 shows the Doppler frequency response of the Doppler frequency index f _s for the second half period N _c / 2 times. Here is a _{ND = 1~N t, k = 1} , ..., a _{(N r + N u) N} s / N o, z = 1, ..., a _{N a.} W is a natural number.

^{_{FT_FH_CI z ND (k, f s}} , w) is, FFT size is _{N c,} in which the Nc / 2 pieces of data in the latter half portion and zero-padding. ^{_{Further, FT_SH_CI z ND (k, f}} s, w) is one in which FFT size is _{N c,} the _N c / 2 pieces of data of the first half portion and zero-padding.

Therefore, the maximum Doppler frequency at which aliasing does not occur, derived from the sampling theorem, is ± 1 / (2N _{t Tr} ). Also, Doppler frequency interval of the Doppler frequency index _{f s is 1 / {N t N c T} r}, a range of Doppler frequency index _{f s is, f s = -N c / 2} + 1, ..., 0, ..., N _c / 2.

In the FFT processing, a window function coefficient such as a Han window or a Hamming window may be multiplied. By applying the window function, side lobes generated around the beat frequency peak can be suppressed. Incidentally, window function coefficients, FFT size is applied to one of the _{N c,} using _N c / 2 pieces of window function coefficients of the first half ^{_{FT_FH_CI z ND (k, f s}} , w) at the time of calculation of the second half of the N _c / 2 coefficients are used when calculating FT_SH_CI _z ^ND (k, f _s , w).

CFAR unit 215, w th output from the Doppler analysis unit 213 # 1~ # _{N t} antennas system processing unit _{201 # 1~ # N a FT_FH_CI z} ND (k, f s, w) and FT_SH_CI _z ^ND (k , F _s , w) to perform CFAR processing. The CFAR processing is performed on a two-dimensional input signal of a discrete time k (corresponding to a distance to a target) and a Doppler frequency index f _s (corresponding to a relative speed of a target).

The CFAR processing, for example, as shown in equation (22a), w th output FT_FH_CI _z ^ND _{(k, f} s from the Doppler analysis unit 213 # 2 of the antenna system processing unit 201 #. 1 to # _{N a,} w ) and ^{_{FT_SH_CI z ND (k, f s}} , w) to the power addition. Then, the CFAR unit 215 performs a CFAR process combining a one-dimensional CFAR process or a two-dimensional CFAR process on the power addition result. The processing disclosed in Non-Patent Document 2 may be applied to this CFAR processing. Here, in the two-dimensional CFAR processing, an axis of discrete time (corresponding to the distance to the target) and an axis of Doppler frequency (corresponding to the relative speed of the target) may be used. Then, CFAR unit 215, a large power sum than the threshold, the discrete time index _{k _Cfar,} and the Doppler frequency index _{f S_cfar,} and outputs the direction estimation unit 214 and the folded-back determination unit 216.

The return determination unit 216 determines whether or not the output from the Doppler analysis unit 213 includes a return signal based on the discrete time index _{k_cfar} output from the CFAR unit 215 and the Doppler frequency index _{fs_cfar} . For example, the return determination unit 216 makes the determination based on Expressions (23) and (24).

である。 Note that, in Expressions (23) and (24),

It is.

Here, FT_CAL _z ^ND shown in equation _{(25) (k, f s} , w) , when the signal of the Doppler frequency index _{f S_cfar} is assumed to be free of aliasing ^{_{signal, FT_FH_CI z ND (k, f}} s , w) and ^{_{FT_SH_CI z ND (k, f s}} , w) to an expression for phase addition. In the equation (25), since the signal of the Doppler frequency index f _{s_cfar undergoes} a phase change (phase rotation) during the transmission gap period T _GAP , the term of the equation (25a) is introduced to correct this phase rotation. ing. Here, since the transmission gap period T _GAP is set to _の of the sampling period (N _p = N _{t Tr} ) of the Doppler analysis unit 213, the transmission gap period T _GAP is the sampling period period of the Doppler frequency index ( _{fs_cfar} ). The phase rotation corresponding to half (1/2) of the phase change (phase rotation) is corrected.

On the other hand, FT_ALIAS _z ^ND shown in equation _{(26) (k, f s} , w) , when the signal of the Doppler frequency index _{f S_cfar} is assumed to include a folded ^{_{signal, FT_FH_CI z ND (k, f}} s, w ) and ^{_{FT_SH_CI z ND (k, f s}} , w) to an expression for phase addition. When the signal of the Doppler frequency index _{fs_cfar includes} a (primary) loopback signal, when the Doppler frequency index _{fs_cfar} ≧ 0, the phase change by the Doppler frequency index of ( _{fs_cfar− Nc} ) during the transmission gap period T _GAP. (Phase rotation) occurs. Further, when the Doppler frequency index _{f s_cfar} <0, during the transmission gap period _{T GAP,} resulting Doppler frequency index of the phase change of _{(f s_cfar} + _{N c) (phase} rotation). Therefore, Expression (26a) is introduced into Expression (26) in order to correct the phase rotation occurring during the transmission gap period T _GAP . Equation (26a) is obtained by substituting _{(f s_cfar} ± _{N c)} to _{f S_cfar} of formula (25a), it becomes equation (25a) and the phase inverted expression. Therefore, one of FT_CAL _z ^ND (k, f _s , w) and FT_ALIAS _z ^ND (k, f _s , w) is added in the same phase and the other is added in the opposite phase. The relationship clearly appears, and it is possible to determine the presence or absence of a loopback signal even when the SNR of the received signal is low.

That is, from the above, when a signal of the Doppler frequency index _{f S_cfar} comprises a folded ^{_{signal, FT_CAL z ND (k, f}} s, w) ^{_{is, FT_ALIAS z ND (k, f}} s, w) smaller in terms of electric power than . On the other hand, _when the signal of the Doppler frequency index f _{s_cfar} does not include a folded signal, FT_ALIAS _z ^ND (k, f _s , w) is smaller in power than FT_CAL _z ^ND (k, f _s , w). For such a reason, the determination methods of Expressions (23) and (24) can be applied.

The return determination unit 216 _converts the Doppler frequency index _{fs_cfar} of the signal determined to include the (primary) return signal, and _outputs the signal as described below.
If Doppler frequency index _{fs_cfar} ≧ 0, convert DopConv ( _{fs_cfar} ) = _{fs_cfar− Nc} and output.
If the Doppler frequency index _{fs_cfar} <0, convert to DopConv ( _{fs_cfar} ) = _{fs_cfar} + _Nc and output.
Here, DopConv (f) represents the result of conversion of the Doppler frequency index to the Doppler frequency index f based on the determination of the aliasing signal.

The return determination unit 216 outputs the signal of the Doppler frequency index _{fs_cfar} determined not to include the return signal without converting the Doppler frequency index as described below.
DopConv ( _{fs_cfar} ) = _{fs_cfar}

Direction estimating unit 214, based on the output from the folded determination unit, the virtual reception array correlation vector h shown in equation (27) (k, f s , w) generates, performs direction estimation process.

In, it summarizes the w-th output from the antenna system processing unit 201 # 1 to # _{N a} Doppler analysis unit 213 # 1 obtained by performing similar processing in the signal processing units 207 of # _{N t} or less and including _N t _{N a} number of elements which is the product of the number of transmit antennas _{N t} as shown in equation (27) and the number of reception antennas _{N a,} the virtual reception array correlation vector _{h (k _cfar, f s_cfar,} w ). The virtual reception array correlation vector h ( _{k_cfar} , _{fs_cfar} , w) is used for processing for estimating the direction of the reflected wave from the target based on the phase difference between the reception antennas Rx. Here, z = 1, ..., a _{N a, ND = 1, ...} , a _{N t.}

h _{cal [b]} is an array correction value for correcting a phase deviation and an amplitude deviation between the transmission array antennas and between the reception array antennas. _Here, b = 1, ..., a _N t N _a.

Further, since the transmission antenna Tx is switched in a time division manner, different phase rotations occur at different Doppler frequencies f. TxCAL ⁽¹⁾ (f),..., TxCAL ^(Nt) (f) are transmission phase correction coefficients for correcting the phase rotation to match the phase of the reference transmission antenna. For example, when Tx # 1 is used as the reference transmission antenna, the transmission phase correction coefficient is given by Expression (29).

In this case, the virtual reception array correlation vector _h of the formula _{(27) (k _cfar, f} s_cfar, w) _is a column vector comprised of N a _{N r} number of elements.

The arrival direction estimation calculates a spatial profile by changing the azimuth direction θ in the direction estimation evaluation function value P _H (θ, _{k_cfar} , _{fs_cfar} , w) within a predetermined angle range. Then, in the arrival direction estimation, a predetermined number of the maximum peak directions of the spatial profile are extracted in descending order, and the elevation angle direction of each maximum peak is output as an arrival direction estimation value.

Radar apparatus 1 according to Embodiment 3 switches a plurality of transmission antennas Tx in a time-division manner and transmits a transmission signal _Nc times from each transmission antenna Tx. At this time, the radar apparatus 1 provides a transmission gap period T _GAP after transmitting a transmission signal _Nc / 2 times from each transmission antenna Tx. Then, in the radar apparatus 1, the loopback determining unit 216 determines whether or not the output signal from the Doppler analyzer 213 includes a loopback signal based on the phase change occurring in the transmission gap period T _GAP . Thereby, the Doppler frequency range in which ambiguity does not occur can be doubled as compared with the case where the transmission gap period T _GAP is not provided.

When the transmission gap period T _GAP is set to N _{t Tr} / 2, the performance (accuracy) of determining whether or not the signal is a loopback signal is the highest. However, the transmission gap period T _GAP is not limited to this, and may be about N _{t Tr} / 2 or a period before and after that.

Also, in transmitting a transmission signal _Nc times from each transmission antenna Tx, _Nc / 2 times transmission signal transmission from each transmission antenna Tx, and then providing a transmission gap period T _GAP after transmitting a transmission signal to determine whether a folded signal is included. The performance (accuracy) of determining whether or not to perform the determination is highest. However, the timing at which the transmission gap period T _GAP is provided is not limited to this, and may be after transmitting about N _c / 2 times, or after transmitting before and after that number of times.

(Embodiment 4)
In the first embodiment, a configuration has been described in which at least one of the plurality of transmission antennas Tx is a short-period transmission antenna. Embodiment 3 has described the configuration in which the transmission gap period T _GAP is provided. In the fourth embodiment, an example of a combination of the configurations of the first and third embodiments will be described. Thereby, the detection range of the Doppler frequency can be further expanded as compared with the first embodiment. Hereinafter, in the fourth embodiment, description will be mainly given of contents different from the first and third embodiments, and description of the same contents as the first and third embodiments will be omitted.

  FIG. 16 shows a configuration example of a radar device 1 according to the fourth embodiment.

  The switching control unit 105 outputs a switching control signal for instructing the transmission RF switching unit 106 of the radar transmitting unit 100 and the output switching unit 211 of the radar receiving unit 200 to switch the output destination. The instruction to switch the output destination to the output switching unit 211 will be described later. Hereinafter, an instruction to switch the output destination to the transmission RF switching unit 106 will be described.

Switching control unit 105, for each transmission cycle _{T r,} from the transmission RF unit 107 #. 1 to # _{N t,} the transmission RF unit 107 selects one for use in the transmission of the transmission signal. Then, switching control section 105 outputs a switching control signal instructing transmission RF switching section 106 to switch the output destination to selected transmission RF section 107.

Transmitting RF switching unit 106, based on the switching control signal outputted from the switching control unit 105, an output destination is switched to one of the transmission RF section 107 # 1~ # _{N t.} Then, transmission RF switching section 106 outputs the transmission signal output from radar transmission signal generation section 101 to switching transmission RF section 107.

FIG. 17 is a diagram for explaining the timing at which the transmission RF units 107 # 1 to # 3 output transmission signals when the number of transmission antennas N _t = 3. FIG. 17 is an example in which transmission RF section 107 # 2 is a short-period transmission RF section.

In this case, the switching control unit 105, every 2T _r period, the transmission RF section 107 # 2, selects the output destination of the transmission signal. Then, in each _Tr period during which the transmission RF section 107 # 2 does not output a transmission signal, the switching control section 105 sequentially selects the transmission RF sections 107 # 1 and # 3 as the output destination of the transmission signal. That is, the switching control unit 105, transmission RF section 107 # 1, and # 3, _{respectively, N p = 4T r = 2} (N t -1) every _{T r} period, selects the output destination.

Here, the switching control unit 105, as shown in FIG. 17, after a set of processing period _{N p = 4T r = 2 (} N t -1) T r, repeated _N c / 2 times, the transmission signal _Is provided in the transmission gap period T _GAP which does not output. Then, switching control unit 105, after the transmission gap period _{T GAP,} again, a set of processing period _{N p = 4T r = 2 (} N t -1) T r, repeated _N c / 2 times. By this processing, transmission RF section 107 # 2, which is a short-period transmission RF section, transmits a transmission signal ( _Nt- 1) _Nc times. Further, transmission RF sections 107 # 1 and # 3 other than transmission RF section 107 # 2 each transmit a transmission signal _Nc times.

Transmission gap period _{T GAP} may be set to 1/2 of the transmission cycle 2T _r transmit RF section 107 # 2 (short-cycle transmission RF unit). In other words, T _GAP = _Tr .

Note that the transmission start time of the transmission signal of each transmission RF section 107 does not necessarily have to be synchronized with the cycle _Tr . For example, as shown in FIG. 6, to the transmission start time of each transmission RF section 107 # 1~ # _{N t,} the transmission delay delta _1, delta _2, ..., it may be provided delta _Nt.

In the radar receiving unit 200, the output switching unit 211 of the signal processing unit 207 of the antenna system processing unit 201 # z outputs a Doppler analysis unit 213 # every _Tr period based on the switching control signal output from the switching control unit 105. 1 one selects one of # _{N t} (switch). Then, output switching section 211 outputs the correlation operation result output from correlation operation section 210 for each _Tr cycle to selected Doppler analysis section 213.

Switching control signal in the M-th transmission cycle _{T r} is, _{N t} bits _{[bit 1 (M), bit} 2 (M), ..., bit Nt (M)] may be constituted by. In this case, when the ND bit is 1 in the switching control signal of the M-th transmission cycle _Tr , the output switching unit 211 selects the Doppler analysis unit 213 # ND as the output destination, and sets the ND bit to 0. In this case, the Doppler analyzer 213 # ND is not selected as the output destination. Here, ND = 1, ..., a _{N t.}

When the number of transmitting antennas N _t = 3, the switching control signal outputs a 3-bit switching control signal to the output switching unit 211 so as to correspond to the output pattern of the transmission signal shown in FIG. Next, a specific example will be described.

First, the switching control unit 105 _causes the _{N p} N c / 2 period (half period), every _{T r} period, each switching control signal shown in the following (D1), is repeatedly output. Note _that it is _{N p = 4T r = 2 (} N t -1) T r.
(D1)
[Bit ₁ (1), bit ₂ (1), bit ₃ (1)] = [0, 1, 0]
[Bit ₁ (2), bit ₂ (2), bit ₃ (2)] = [1, 0, 0]
[Bit ₁ (3), bit ₂ (3), bit ₃ (3)] = [0, 1, 0]
[Bit ₁ (4), bit ₂ (4), bit ₃ (4)] = [0, 0, 1]

After the above-described first half period, the switching control unit 105 outputs a switching control signal in which all bits shown in (D2) below are zero in the transmission gap period T _GAP .
(D2)
[Bit ₁ , bit ₂ ,..., Bit _Nt ] = [0, 0, 0]

After the transmission gap period T _GAP ends, the switching control unit 105 outputs each switching control signal shown in the following (D3) for each _Tr cycle in the N _{pn c} / 2 period (second half period).
(D3)
_{[Bit 1 (2 (N t} -1) N c / 2 + 1), bit 2 (2 (N t -1) N c / 2 + 1), bit 3 (2 (N t -1) N c / 2 + 1)] = [0,1,0]
_{[Bit 1 (2 (N t} -1) N c / 2 + 2), bit 2 (2 (N t -1) N c / 2 + 2), bit 3 (2 (N t -1) N c / 2 + 2)] = [1,0,0]
_{[Bit 1 (2 (N t} -1) N c / 2 + 3), bit 2 (2 (N t -1) N c / 2 + 3), bit 3 (2 (N t -1) N c / 2 + 3)] = [0,1,0]
[Bit ₁ (2 (N _t −1) (N _c / 2 + 1)), bit ₂ (2 (N _t −1) (N _c / 2 + 1)), bit ₃ (2 (N _t −1) (N _c / 2 + 1))] = [0,0,1]
…
[Bit ₁ (2 (N _t -1) (N _c -1) +1), bit ₂ (2 (N _t -1) (N _c -1) +1), bit ₃ (2 (N _t -1) ( _Nc- 1) +1)] = [0,1,0]
[Bit ₁ (2 (N _t -1) (N _c -1) +2), bit ₂ (2 (N _t -1) (N _c -1) +2), bit ₃ (2 (N _t -1) ( _Nc- 1) +2)] = [1,0,0]
[Bit ₁ (2 (N _t -1) (N _c -1) +3), bit ₂ (2 (N _t -1) (N _c -1) +3), bit ₃ (2 (N _t -1) ( _Nc- 1) +3)] = [0,1,0]
[Bit ₁ (2 (N _t -1) N _c ), bit ₂ (2 (N _t -1) N _c ), bit ₃ (2 (N _t -1) N _c )] = [0,0,1 ]

Antenna system processor 201 # z of the signal processing unit 207 includes a Doppler analysis unit 213 # 1~ # _{N t.} Doppler analysis unit 213, the correlation calculation result output from the output switching unit 211, a correlation calculation result of _N c / 2 times prior to the start of the transmission gap period _{T GAP} (half period), the end of the transmission gap period _{T GAP} The Doppler analysis is performed separately for each discrete time k with the correlation calculation results for the later N _c / 2 times (second half period). In the Doppler analysis, if _Nc is a power of 2, for example, FFT processing as shown in Expressions (30) to (34) can be applied.

FT_FH_CI _z ^ND of formula (30) and equation (31) _{(k, f} s, w) is located at the w-th output by Doppler analysis unit 213 # ND in the antenna system processing unit 201 # z of the signal processing unit 207 shows Doppler frequency index f _s at discrete time k, the first half period N _{c /} 2 times the Doppler frequency response.

FT_SH_CI _z ^ND of formula (32) and formula (33) _{(k, f} s, w) is located at the w-th output by Doppler analysis unit 213 # ND in the antenna system processing unit 201 # z of the signal processing unit 207 , Shows the Doppler frequency response of the Doppler frequency index f _{s at} the discrete time k for the latter half period N _c / 2 times. It should be noted, is a _{ND = 1~N t, k = 1} , ..., a _{(N r + N u) N} s / N o, z = 1, ..., a _{N a.} W is a natural number.

Hereinafter, a specific example the second transmission RF section 107 # 2 as short-cycle transmission RF unit 107 of the 2T _r period.

ND = 2 If the (a short period received ^{_{signal), FT_FH_CI z ND (k,}} f s, w) is the FFT size _(N t -1) N _c, the subsequent portion _(N t -1) It is the result of N _c / 2 data padded with zeros. Also, in the case of ^{_{ND = 2, FT_SH_CI z ND (}} k, f s, w) is the FFT size _(N t -1) N _c, the front portion _{(N t -1) N c /} 2 pieces of Data is zero-filled. The maximum Doppler frequency at which aliasing does not occur, derived from the sampling theorem, is both ± 1 / (4T _r ). Further, both the Doppler frequency interval of the Doppler frequency index _{f s} is _{1 / {2 (N t -1} ) N c T r}, a range of Doppler frequency index _{f s is} f s = - _{(N t} -1) _Nc / 2 + 1, ..., 0, ..., ( _Nt- 1) _Nc / 2.

For ND ≠ 2 (not short-period received ^{_{signal), FT_FH_CI z ND (k,}} f s, w) is, FFT size is _{N c,} obtained by zero filling the _N c / 2 pieces of data subsequent portion is there. Further, when the ^{_{ND ≠ 2, FT_SH_CI z ND (}} k, f s, w) is, FFT size is _N c / 2, zero fill the _{(N t -1) N c /} 2 pieces of data of the previous portion It was done. The maximum Doppler frequency at which aliasing does not occur, which is derived from the sampling theorem, is ± 1 / {4 (N _t −1) T _r }. Further, both the Doppler frequency interval of the Doppler frequency index _{f u} is _{1 / {2 (N t -1} ) N c T r}, a range of Doppler frequency index _{f u is,} f u = _-N c / , 0, ..., _Nc / 2.

Comparing the outputs from the Doppler analyzer 213 in the case of ND = 2 and the case of ND ≠ 2, both Doppler frequency intervals are the same. Also, the maximum Doppler frequency at which no aliasing occurs when ND = 2 is ± (N _t −1) times that in the case of ND ≠ 2, and the Doppler frequency range is expanded to (N _t −1) times. Is output.

Therefore, the transmission antenna Tx # 1, Tx # 2, ..., Tx # compared with the case where as sequentially switched in N _t, the arrangement according to this embodiment, the maximum Doppler frequency wrapping ND = 2 does not occur Expands to N _t / 2 times when the number of transmission antennas N _t is 3 or more. In other words, in proportion to the number of transmit antennas N _t, the Doppler frequency range is extended wrapping does not occur.

Equations (30) and (31) are applied to N _c / 2 times before the start of the transmission gap period T _GAP (first half period).

Equations (32) and (33) are applied to N _c / 2 times after the end of the transmission gap period T _GAP (second half period).

Note that, in the case of ND ≠ 2, if there is no output from the output switching unit 211, the FFT size is set to (N _t −1) N _c, and virtually using Expressions (34) and (35). The output may be sampled as zero. Expression (34) is the same as Expression (30), and Expression (35) is the same as Expression (32). As a result, the FFT size increases, so that the processing amount increases. However, since the Doppler frequency index matches the case of ND = 2, the later-described Doppler frequency index conversion processing becomes unnecessary.

In the FFT processing, a window function coefficient such as a Han window or a Hamming window may be multiplied. By applying the window function, side lobes generated around the beat frequency peak can be suppressed. For ND = 2, FFT size applies a window function coefficient of _(N t -1) N _c, the _(N t -1) of _{N c,} the first half period _{(N t -1) N c /} 2 the number of window function ^{_{coefficients, FT_FH_CI z ND (k, f}} s, w) used in calculating of the _N c / 2 pieces of window function coefficients of the second half period, the calculation of ^{_{FT_SH_CI z ND (k, f s}} , w) Sometimes used.

Also, ND ≠ case of 2, FFT size applies a window function coefficient of the _{N c,} among the _{N c,} the _N c / 2 pieces of window function coefficients of the first half ^{_{period, FT_FH_CI z ND (k, f}} s, used in calculating the w), the _N c / 2 pieces of window function coefficients of the second half period is used when calculating the ^{_{FT_SH_CI z ND (k, f s}} , w).

The CFAR unit 215 adaptively sets (adjusts) a threshold using the short-cycle received signal and performs a peak signal detection process (CFAR process). In this embodiment, transmission RF section 107 # 2 transmits a transmission signal with 2T _r period. Therefore, CFAR unit 215, FT_FH_CI ₁ is a w-th output from the Doppler analysis unit _{^{213 # 2 (2) (k}} , f s, w), ..., FT_FH_CI Na (2) (k, f s, w) ^{_{If, FT_SH_CI 1 (2) (k}} , f s, w), ..., performs CFAR processing using _{^{FT_SH_CI Na (2) (k,}} f s, w) and, the.

CFAR unit 215, the CFAR processing, set the adaptive threshold, the larger received power than the threshold value, in the case of ND = 2, the discrete time index _{k _Cfar} and Doppler frequency index _{f S_cfar,} folded determination unit 216 Output to

Folded determination unit 216, based on the discrete time index _{k _Cfar} and Doppler frequency index _{f S_cfar} output from CFAR section 215 determines the output of the Doppler analysis unit 213 # 2 whether includes aliasing signal. In the case of the present embodiment, loopback determining section 216 applies equations (36) and (37) to the output from Doppler analyzer 213 # 2, and determines whether or not a loopback signal is included. Note that ND is the number of the short-period transmission antenna Tx, and in the present embodiment, ND = 2.

である。 Note that in equations (36) and (37),

It is.

Here, FT_CAL _z ^ND shown in equation _{(38) (k, f s} , w) , when the signal of the Doppler frequency index _{f S_cfar} is assumed to be free of aliasing ^{_{signal, FT_FH_CI z ND (k, f}} s , w) and ^{_{FT_SH_CI z ND (k, f s}} , w) to an expression for phase addition. In equation (38), since the signal of the Doppler frequency index f _{s_cfar undergoes} a phase change (phase rotation) during the transmission gap period T _GAP , the term of equation (38a) is introduced to correct this phase rotation. ing. Here, the transmission gap period _{T GAP} as half the transmission cycle 2T _r transmit RF section 107 # 2 (short-cycle RF transmitter), that is, from that set to _T GAP = _{T r,} Doppler frequency The phase rotation corresponding to half (1/2) of the phase change (phase rotation) of the index ( _{fs_cfar} ) during the sampling cycle period is corrected.

On the other hand, FT_ALIAS _z ^ND shown in equation (39) _{(k, f} s, w), when the signal of the Doppler frequency index _{f S_cfar} is assumed to include a (primary) loop signal, FT_FH_CI _z ^ND (k, f _s, w) and ^{_{FT_SH_CI z ND (k, f s}} , w) to an expression for phase addition. If the signal of the Doppler frequency index _{fs_cfar includes} a (primary) loopback signal, and if the Doppler frequency index _{fs_cfar} ≧ 0, the Doppler of ( _{fs_cfar−} (N _t −1) N _c ) during the transmission gap period T _GAP. A phase change (phase rotation) corresponding to the frequency index occurs. Further, when the Doppler frequency index _{f s_cfar} <0, during the transmission gap period _{T GAP,} occurs a phase rotation of the Doppler frequency index component of _{(f s_cfar + (N t -1} ) N c). Therefore, the following equation (39a) obtained by inverting the equation (38a) is introduced into the equation (39) in order to correct the phase rotation. Equation (39a) is obtained by substituting ( _{fs_cfar} ± (N _t −1) N _c ) into f _{s_cfar} of equation (38a), and is an equation obtained by inverting equation (25a). Therefore, one of FT_CAL _z ^ND (k, f _s , w) and FT_ALIAS _z ^ND (k, f _s , w) is added in the same phase and the other is added in the opposite phase. The relationship clearly appears, and it is possible to determine the presence or absence of a loopback signal even when the SNR of the received signal is low.

That is, from the above, when a signal of the Doppler frequency index _{f S_cfar} comprises a folded ^{_{signal, FT_CAL z ND (k, f}} s, w) ^{_{is, FT_ALIAS z ND (k, f}} s, w) smaller in terms of electric power than . On the other hand, _when the signal of the Doppler frequency index f _{s_cfar} does not include a folded signal, FT_ALIAS _z ^ND (k, f _s , w) is smaller in power than FT_CAL _z ^ND (k, f _s , w). For such a reason, the determination methods of Expressions (36) and (37) can be applied.

The return determination unit 216 _converts the Doppler frequency index _{fs_cfar} of the signal determined to include the (primary) return signal to the Doppler frequency index as _follows , and outputs the signal to the direction estimation unit 214 together with the discrete time index _{k_cfar.} I do.
For Doppler frequency index _{f s_cfar ≧ 0, DopConv (f} s_cfar) = f s_cfar - converting the _(N t -1) N _c, and outputs.
For Doppler frequency index _{f s_cfar} <0, then converted _{DopConv (f s_cfar) = f s_cfar} + (N t -1) N c, and outputs.

For the signal of the Doppler frequency index f _{s_cfar} determined not to include the return signal, the return determination unit 216 _outputs the signal to the direction estimation unit 214 together with the discrete time index _{k_cfar} without converting the Doppler frequency index as follows. I do.
DopConv ( _{fs_cfar} ) = _{fs_cfar}

In addition, the return determination unit 216 converts DopConv ( _{fs_cfar} ) _obtained by converting the wide-range Doppler frequency index _{fs_cfar} of ND = 2 into Doppler analysis units 213 # 1, # 3 other than the Doppler analysis unit 213 # 2. ..., to index conversion used to correspond to a narrow range Doppler frequency index f _u of w-th output from the # N _t, the following equation (40) and (41). Then, the aliasing determination unit 216 outputs the narrow range Doppler frequency index _{fu_cfar} after the index conversion to the direction estimation unit 214.

The direction estimating unit 214 is based on the discrete time index _{k_cfar} , the Doppler frequency index _{fs_cfar} , the Doppler frequency index DopConv ( _{fs_cfar} ), and the Doppler frequency index f _{u_cfar} output from the loopback determining unit 216. from the output of 213, the virtual reception array correlation vector h shown in equation _{(42) (k, f s} , w) generates, performs direction estimation process.

In the following, a summary of the w-th output from the antenna system processing unit 201 # 1~ # _{N a} Doppler analysis unit 213 obtained by performing the same processing in the signal processing unit 207 of the # 1~ # _{N t} formula including _N t _{N a} number of elements which is the product of the number of transmit antennas _{N t} as shown in (42) and the number of reception antennas _{N a,} the virtual reception array correlation vector _{h (k _cfar, f s_cfar,} w) Notation as The virtual reception array correlation vector h ( _{k_cfar} , _{fs_cfar} , w) is used for processing for estimating the direction of the reflected wave from the target based on the phase difference between the reception antennas Rx. Here, z = 1, ..., a _{N a, ND = 1, ...} , a _{N t.}

h _{cal [b]} is an array correction value for correcting a phase deviation and an amplitude deviation between the transmission array antennas and between the reception array antennas. _Here, b = 1, ..., a _N t N _a.

Further, since the transmission antenna Tx is switched in a time division manner, different phase rotations occur at different Doppler frequencies f. TxCAL ⁽¹⁾ (f),..., TxCAL ^(Nt) (f) are transmission phase correction coefficients for correcting the phase rotation to match the phase of the reference transmission antenna. For example, when the transmission antenna Tx # 2 is used as a reference transmission antenna, the transmission phase correction coefficient is represented by Expression (44).

In this case, the virtual reception array correlation vector _h of the formula _{(42) (k _cfar, f} s_cfar, w) _is a column vector comprised of N a _{N r} number of elements.

Radar apparatus 1 according to the fourth embodiment, of the transmission antenna Tx # 1~ # _{N t,} the transmission period of the transmission signal from the transmitting antenna (short-period transmission antenna) Tx # 2 is 2T _r, otherwise each transmission antenna Tx # 1 of # 3, ..., the transmission period of the transmission signal from the # _{N t} is _{2 (N t -1) T r} . Thus, compared with the case of transmitting a transmission signal by sequentially switching the transmission antenna Tx # 1~ # N _t, in the short period reception signal corresponding to the short period transmission signal from the short-period transmission antenna Tx # 2, aliasing occurs maximum Doppler frequency without (relative speed) is increased to N _{t /} 2 times, the Doppler frequency range aliasing does not occur is increased to N _{t /} 2 times (E1 effects of).

Moreover, the radar apparatus 1 according to the fourth embodiment, the transmission antenna Tx # 1, # 3, ... , N c times from # _{N t,} and transmits the transmission signal. At this time, the radar device 1, the transmission antennas Tx # 1, # 3, ... , N c / 2 times the # _{N t,} after transmitting the transmission signal, providing the transmission gap period _{T GAP.} The radar device 1 judges the folded determination unit 216, based on the phase rotation caused in transmission gap period T _GAP, the result of the Doppler analysis from Doppler analysis unit 213 # 2 whether includes aliasing signal. As a result, the Doppler frequency range in which ambiguity does not occur can be further doubled compared to the case where the transmission gap period T _GAP is not provided (E2 effect).

Therefore, the radar apparatus 1 according to the fourth embodiment, the two effects described above and (E1) (E2), transmission antenna Tx #. 1 to # _{N t} compared to the case of successively switching the, the Doppler frequency range _{N t} It can be enlarged twice (= N _t / 2 times × 2 times).

When the transmission gap period T _GAP is set to _Tr , the performance (accuracy) of determining the presence or absence of a loopback signal is the highest. However, the transmission gap period T _GAP is not limited to this, and may be about _Tr , or a period before and after _Tr .

Further, the transmission antennas # 1, # 3, ..., N c times from # _{N t,} in transmitting the transmission signal, the transmission antennas # 1, # 3, ..., N c / 2 times the transmission signal from the # _{N t} In the case where the transmission gap period T _GAP is provided after the transmission of the transmission signal, the determination performance (accuracy) of the presence / absence of a loopback signal is highest. However, the timing at which the transmission gap period T _GAP is provided is not limited to this, and may be after transmitting about N _c / 2 times, or after transmitting before and after that number of times.

(Embodiment 5)
Embodiment 3 has described an example in which one transmission gap period T _GAP is provided. Embodiment 5 describes an example in which transmission gap period T _GAP is provided N _GAP times. The configuration of the radar device 1 is the same as that of the third embodiment shown in FIG. However, since some operations are different, the different operations will be mainly described below.

  Transmission RF switching section 106 outputs the output from radar transmission signal generation section 101 to transmission RF section 107 at the designated switching destination based on the instruction of the switching control signal output from switching control section 105.

Switching control unit 105, for each transmission cycle _{T r,} sequentially selects one of the transmission RF section 107 # 1~ # _{N t.} Then, switching control section 105 outputs a switching control signal instructing the selected transmission RF section 107 to switch the output destination to transmission RF switching section 106. Thus, the transmission RF switching unit 106, the respective transmission RF section 107 # 1~ # _{N t,} in the cycle _N t _{T r,} selects sequentially output destination. With other words, each of the transmission RF unit 107, in the cycle _N t _{T r,} and transmits the transmission signal.

Switching control unit 105, the processing period _N p _{= N} t _{T r,} repeated _{N c / (N} GAP +1) times. After that, the switching control unit 105 provides the first transmission gap period T _GAP # 1.

Then, after the transmission gap period T _GAP # 1 has elapsed, the switching control unit 105 repeats the processing of the period N _p = N _{t Tr} again _Nc / (N _GAP +1) times. After that, the switching control unit 105 provides a second transmission gap period T _GAP # 2.

Then, after the transmission gap period T _GAP # 2 has elapsed, the switching control unit 105 repeats the process of the period N _p = N _{t Tr} again N _c / (N _GAP +1) times.

According to the above processing, the transmission gap period _{T GAP} is provided _{N GAP} times, #. 1 to # _{N t} is the transmission RF section 107, _{N c} times, and transmits a transmission signal.

If N _c / (N _GAP +1) is not an integer, the decimal part may be rounded down or rounded up to an integer.

The transmission gap period T _GAP may be 1 / (N _GAP +1) times the sampling period of Doppler analysis (period for selecting the transmission RF units 107 # 1 to N _{t one} cycle) N _p = N _{t Tr} . That is, the transmission gap period T _GAP = N _p / (N _GAP +1) = N _{t Tr} / (N _GAP +1).

Output switching unit 211, for each transmission cycle _{T r,} based on the switching control signal outputted from the switching control unit 105 sequentially selects the Doppler analysis unit 213 # 1~ # _{N t.} Then, the output switching unit 211 outputs the correlation operation result output from the correlation operation unit 210 to the selected Doppler analysis unit 213 for each transmission cycle _Tr .

Switching control signal in the M-th transmission cycle _{T r} is, _{N t} bits _{[bit 1 (M), bit} 2 (M), ..., bit Nt (M)] may be constituted by. In this case, when the ND bit of the switching control signal is 1 in the M-th transmission cycle _Tr , the output switching unit 211 selects the Doppler analyzer 213 # ND as the output destination, and outputs the ND-th switching control signal. When the bit is 0, the Doppler analyzer 213 # ND is not selected as an output destination (not selected). It should be noted, ND = 1, ..., a _{N t.}

Before the start of the transmission gap period T _GAP # 1, the switching control unit 105 transmits one set (N _p = N _{t Tr} period) of switching control signals shown in (F1) below to N _c / (N _GAP +1). ) Times output.
(F1)
[Bit ₁ (1), bit ₂ (1), ..., bit _Nt (1)] = [1, 0, ..., 0]
[Bit ₁ (2), bit ₂ (2), ..., bit _Nt (2)] = [0, 1, ..., 0]
…
[Bit ₁ (N _t ), bit ₂ (N _t ),..., Bit _Nt (N _t )] = [0, 0,..., 1]

Then, the switching control unit 105 outputs one set of switching control signals shown in the above (F1) N _c / (N _GAP +1) times, and then, in the transmission gap period T _GAP # 1, shows the following (F2) A switching control signal in which all bits are zero is output.
(F2)
_{[Bit 1, bit 2, ...} , bit Nt] = [0,0, ..., 0]

Switching control unit 105, after the end of the transmission gap period _{T GAP} # 1, before the start of the transmission gap period _{T GAP} # 2, switching of the following set shown in _{(F3) (N p = N} t T r period) The control signal is output _Nc / ( _NGAP + 1) times.
(F3)
_{[Bit 1 (N t N c} / (N GAP +1) +1), bit 2 (N t N c / (N GAP +1) +1), ..., bit Nt (N t N c / (N GAP +1) +1) ] = [1,0, ..., 0]
_{[Bit 1 (N t N c} / (N GAP +1) +2), bit 2 (N t N c / (N GAP +1) +2), ..., bit Nt (N t N c / (N GAP +1) +2) ] = [0,1, ..., 0]
…
_{[Bit 1 (2N t N c} / (N GAP +1)), bit 2 (2N t N c / (N GAP +1)), ..., bit Nt (2N t N c / (N GAP +1))] = [ 0,0, ..., 1]

After outputting one set of the switching control signal shown in (F3) _Nc / (N _GAP +1) times, the switching control unit 105 outputs all the bits shown in the following (F4) in the transmission gap period T _GAP # 2. Outputs a switching control signal of zero.
(F4)
_{[Bit 1, bit 2, ...} , bit Nt] = [0,0, ..., 0]

Thereafter, the same processing is repeated, and after the transmission gap period T _GAP #N _GAP ends, the switching control unit 105 transmits one set (N _p = N _{t Tr} period) of switching control signals shown in (F5) below. It outputs _Nc / ( _NGAP + 1) times.
(F5)
[Bit ₁ (N _GAP N _t N _c / (N _GAP +1) +1), bit ₂ (N _GAP N _t N _c / (N _GAP +1) +1),..., Bit _Nt (N _GAP N _t N _c / ( N _GAP +1) +1)] = [1,0,..., 0]
[Bit ₁ (N _GAP N _t N _c / (N _GAP +1) +2), bit ₂ (N _GAP N _t N _c / (N _GAP +1) +2), ..., bit _Nt (N _GAP N _t N _c / ( N _GAP +1) +2)] = [0,1,..., 0]
…,
_{[Bit 1 (N t N c} ), bit 2 (N t N c), ..., bit Nt (N t N c)] = [0,0, ..., 1]

Antenna system processor 201 # z of the signal processing unit 207 includes a Doppler analysis unit 213 # 1~ # _{N t.} Doppler analysis unit 213 # 1~ # _{N t,} respectively, the _N c _{/ (N} GAP +1) times of the correlation calculation result to the start of each transmission gap period _{T GAP,} separately, _{(i.e., (N} GAP +1 A) Doppler analysis is performed at each discrete time k. In the Doppler analysis, if _Nc is a power of 2, an FFT process as shown in Expression (45) can be applied.

Formula FT_GAP_CI _z ^ND of _{(45) (n g, k} , f s, w) is the first w-th output by Doppler analysis unit 213 # ND in the antenna system processing unit 201 # z of the signal processing unit 207, a discrete Doppler frequency index _{f s} at time k, indicating the Doppler frequency response for _N c _{/ (N} gAP +1) times the correlation calculation result, separated by transmission gap period. Here, _ng = 0,..., _NGAP , and in the case of _ng = 0, it is a Doppler frequency response to the first _Nc / ( _NGAP + 1) correlation operation results, where 0 < _ng <N for _gAP, Doppler frequency for _n c _{/ (n} gAP +1) times the correlation calculation result between the after end of the transmission gap period _T gAP #n _g before the start of the transmission gap period _T gAP _{# (n} g +1) Indicates a response. If _ng = N _GAP , it is the Doppler frequency response to the last N _c / (N _GAP +1) correlation operation results. In addition, a _{ND = 1~N t, k = 1} , ..., a _{(N r + N u) N} s / N o, z = 1, ..., a _{N a.} W is a natural number.

^{_{FT_GAP_CI z ND (n g, k}} , f s, w) is, FFT size of the Doppler analysis is _{N c,} which was zero padded data _{N c / (N} GAP +1) times the portion other than the correlation calculation output It is.

Therefore, the maximum Doppler frequency at which aliasing does not occur, derived from the sampling theorem, is ± 1 / (2N _{t Tr} ). Also, Doppler frequency interval of the Doppler frequency index _{f s is 1 / {N t N c T} r}, a range of Doppler frequency index _{f s is, f s = -N c / 2} + 1, ..., 0, ..., N _c / 2.

In the FFT processing, a window function coefficient such as a Han window or a Hamming window may be multiplied. By applying the window function, side lobes generated around the beat frequency peak can be suppressed. For example, as shown in Expression (46), a window function coefficient having an FFT size of _Nc is applied. Here, winf (x) represents a window function coefficient, and x represents an index of the window function (x = 1,..., N _c ).

CFAR section 215, with respect to w-th output from the Doppler analysis unit 213 # 1~ # _{N t} antennas system processing unit _{201 # 1~ # N a, FT_GAP_CI} z ND (n g, k, f s, w ) To perform the CFAR processing. The CFAR processing is performed on a two-dimensional input signal of a discrete time k (corresponding to a distance to a target) and a Doppler frequency index f _s (corresponding to a relative speed of a target).

The CFAR processing, for example, as shown in equation (46a), w th output FT_GAP_CI _z ^ND from Doppler analysis unit 213 # 2 of the antenna system processing unit _{201 # 1~ # N a (0} , k, f s ^{_{, w), FT_GAP_CI z ND (}} 1, k, f s, w), ..., FT_GAP_CI z ND (N GAP, k, f s, w) to the power addition. Then, the CFAR unit 215 performs a CFAR process combining a one-dimensional CFAR process or a two-dimensional CFAR process on the power addition result. The processing disclosed in Non-Patent Document 2 may be applied to this CFAR processing. Here, in the two-dimensional CFAR processing, an axis of discrete time (corresponding to the distance to the target) and an axis of Doppler frequency (corresponding to the relative speed of the target) may be used.

For example, the CFAR unit 215 may set an adaptive threshold as disclosed in Non-Patent Document 2. Then, CFAR section 215 becomes larger received power than the threshold, the discrete time index _{k _Cfar,} and the Doppler frequency index _{f S_cfar,} and outputs the direction estimation unit 214 and the folded-back determination unit 216.

The return determination unit 216 determines whether or not the output from the Doppler analysis unit 213 includes a return signal based on the discrete time index _{k_cfar} output from the CFAR unit 215 and the Doppler frequency index _{fs_cfar} . For example, the return determination unit 216 makes the determination based on Expressions (47) and (48).

である。 Note that in equations (47) and (48),

It is.

  Here, sign (x) is a function that returns 1 when x is positive and -1 when x is negative.

Here, FT_CAL _z ^ND (k, f _s , w) shown in Expression (49) is FT_GAP_CI _z ^ND (0, k, k), assuming that the signal of the Doppler frequency index f s — _cfar does not include a folded signal. ^{_{f s, w), FT_GAP_CI z}} ND (1, k, f s, w), ..., an equation ^{_{FT_GAP_CI z ND (N GAP, k}} , f s, w) a phase addition. In the equation (49), the signal of the Doppler frequency index f _{s_cfar undergoes} a phase change (phase rotation) during the transmission gap period T _GAP . Therefore, to correct this phase rotation, the term of the equation (49a) is introduced. ing. Here, since the transmission gap period T _GAP is set as T _GAP = N _p / (N _GAP +1) = N _{t Tr} / (N _GAP +1), FT_GAP_CI _z ^ND (1, k, f _s) , W), the phase rotation corresponding to _ng / (N _GAP +1) of the phase change (phase rotation) of the Doppler frequency index ( _{fs_cfar} ) during the sampling period is corrected.

On the other hand, FT_ALIAS _z ^ND (k, f _s , w) shown in equation (50) is FT_GAP_CI _z ^ND (0, k), assuming that the signal of the Doppler frequency index f _{s_cfar includes} a (primary) folded signal. ^{_{, f s, w), FT_GAP_CI}} z ND (1, k, f s, w), ..., FT_GAP_CI z ND (N GAP, k, f s, w) to an expression for phase addition. When the signal of the Doppler frequency index f _{s_cfar includes} a (primary) loopback signal, when the Doppler frequency index f _{s_cfar} ≧ 0, the phase change by the Doppler frequency index of ( _{fs_cfar−} N _c ) during the transmission gap period T _GAP (Phase rotation) occurs. When the Doppler frequency index f _{s_cfar} <0, a phase change _{corresponding} to ( _{fs_cfar} + N _c ) Doppler frequency index occurs during the transmission gap period T _GAP . Therefore, equation (50a) is introduced into equation (50) to correct this phase rotation. Formula (50a) to _{f S_cfar} of formula _{(49a) (f s_cfar -sign (} f s_cfar) N c) obtained by substituting equation (49a) phase rotation 2π × _n g _/ in _(N GAP +1) Is added. Here, when N _GAP = 2, the phase rotation 2π × _ng / (N _GAP +1) is {0, 2π / 3, 4π / 3}. When N _GAP = 3, the phase rotation 2π × ng / (N _GAP +1) is {0, 2π / 4, 4π / 4, 6π / 4}. In this manner, the phase rotation 2π × _ng / (N _GAP +1) is canceled out to zero by adding the phase rotation 2π × _ng / (N _GAP +1) with _ng = 0,..., N _GAP. Give rotation. _Therefore, when _{^{FT_CAL z ND (k, f s}} , w) and ^{_{FT_ALIAS z ND (k, f s}} , w) is the either the in-phase addition, the other ^{_{FT_GAP_CI z ND (n g, k}} , f s , W) are mutually canceled and added, so that a signal level difference clearly occurs, and it is possible to determine the presence or absence of a folded signal even when the SNR of the received signal is low.

Therefore, _when the signal of the Doppler frequency index f s — _{cfar includes} a (primary) folded signal, FT_CAL _z ^ND (k, f _s , w) is smaller in power than FT_ALIAS _z ^ND (k, f _s , w). . On the other hand, _when the signal of the Doppler frequency index f _{s_cfar} does not include a folded signal, FT_ALIAS _z ^ND (k, f _s , w) is smaller in power than FT_CAL _z ^ND (k, f _s , w).
For such a reason, the determination methods of Expression (49) and Expression (50) can be applied.

By setting the number of _NGAPs to be plural, there is an additional effect that determination can be performed even when a higher-order return signal is included. For example, when a secondary aliasing signal is included, the aliasing determination unit 216 makes the determination using Equation (50b), Equation (50c), and Equation (50d).

である。ここで、式（５０ｅ）に示すＦＴ＿２ｎｄＡＬＩＡＳ_ｚ ^ＮＤ（ｋ，ｆ_ｓ，ｗ）は、ドップラ周波数インデックスｆ_{ｓ＿ｃｆａｒ}の信号が（二次）折り返し信号を含むものと仮定した場合に、ＦＴ＿ＧＡＰ＿ＣＩ_ｚ ^ＮＤ（０，ｋ，ｆ_ｓ，ｗ）、ＦＴ＿ＧＡＰ＿ＣＩ_ｚ ^ＮＤ（１，ｋ，ｆ_ｓ，ｗ）、…、ＦＴ＿ＧＡＰ＿ＣＩ_ｚ ^ＮＤ（Ｎ_ＧＡＰ，ｋ，ｆ_ｓ，ｗ）を同相加算する式である。ドップラ周波数インデックスｆ_{ｓ＿ｃｆａｒ}の信号が折り返し信号を含む場合、ドップラ周波数インデックスｆ_{ｓ＿ｃｆａｒ}≧０のとき、送信ギャップ期間Ｔ_ＧＡＰ中に、（ｆ_{ｓ＿ｃｆａｒ}＋２Ｎ_ｃ）のドップラ周波数インデックス分の位相変化（位相回転）が生じる。ドップラ周波数インデックスｆ_{ｓ＿ｃｆａｒ}＜０のとき、送信ギャップ期間Ｔ_ＧＡＰ中に（ｆ_{ｓ＿ｃｆａｒ}−２Ｎ_ｃ）ドップラ周波数インデックス分の位相変化が生じる。そこで、式（５０ｅ）には、この位相回転を補正するために、式（５０ｆ）を導入している。式（５０ｆ）は式（４９ａ）のｆ_{ｓ＿ｃｆａｒ}に（ｆ_{ｓ＿ｃｆａｒ}＋ｓｉｇｎ（ｆ_{ｓ＿ｃｆａｒ}）×２Ｎ_ｃ）を代入することで得られ、式（４９ａ）に位相回転４π×ｎ_ｇ／（Ｎ_ＧＡＰ＋１）が付与された式となる。例えば、Ｎ_ＧＡＰ＝２のとき、位相回転４π×ｎ_ｇ／（Ｎ_ＧＡＰ＋１）は、｛０、４π／３、８π／３｝である。また、Ｎ_ＧＡＰ＝３のとき、位相回転４π×ｎ_ｇ／（Ｎ_ＧＡＰ＋１）は、｛０、４π／４、８π／４、１２π／４｝ある。このように、位相回転４π×ｎ_ｇ／（Ｎ_ＧＡＰ＋１）は、ｎ_ｇ＝０、…、Ｎ_ＧＡＰでの位相回転４π×ｎ_ｇ／（Ｎ_ＧＡＰ＋１）を付与して加算すると互いに打ち消してゼロとなる性質を有する。従って、ＦＴ＿ＣＡＬ_ｚ ^ＮＤ（ｋ，ｆ_ｓ，ｗ）、ＦＴ＿ＡＬＩＡＳ_ｚ ^ＮＤ（ｋ，ｆ_ｓ，ｗ）、およびＦＴ＿２ｎｄＡＬＩＡＳ_ｚ ^ＮＤ（ｋ，ｆ_ｓ，ｗ）は、いずれか一つが同相加算され、残りの二つはＦＴ＿ＧＡＰ＿ＣＩ_ｚ ^ＮＤ（ｎ_ｇ，ｋ，ｆ_ｓ，ｗ）の各項が互いに打ち消され、信号レベル差が明確に生じる関係となる。そのため、受信信号のＳＮＲが低い場合でも、折り返し信号の有無の判定が可能となり、さらに（一次）あるいは（二次）折り返し信号が含まれるかの判定も可能となる。 In addition, in Formula (50b), Formula (50c), and Formula (50d),

It is. Here, FT — 2ndALIAS _z ^ND (k, f _s , w) shown in equation (50e) is FT_GAP_CI _z ^ND (0, assuming that the signal of the Doppler frequency index f s — _{cfar includes} a (secondary) folded signal. ^{_{, k, f s, w)}} , FT_GAP_CI z ND (1, k, f s, w), ..., FT_GAP_CI z ND (N GAP, k, f s, w) to an expression for phase addition. When the signal of the Doppler frequency index f _{s_cfar} includes a loopback signal, when the Doppler frequency index f _{s_cfar} ≧ 0, the phase change (phase rotation) by the Doppler frequency index of ( _{fs_cfar} + 2N _c ) during the transmission gap period T _GAP. Occurs. When the Doppler frequency index f _{s_cfar} <0, a phase change _{corresponding} to ( _{fs_cfar−2N c} ) Doppler frequency index occurs during the transmission gap period T _GAP . Therefore, equation (50f) is introduced into equation (50e) to correct this phase rotation. Equation (50f) to _{f S_cfar} of formula _{(49a) (f s_cfar + sign} (f s_cfar) × 2N c) obtained by substituting equation (49a) phase rotation 4π × _n g _/ in _(N GAP +1) Is given by the expression. For example, when N _GAP = 2, the phase rotation 4π × _ng / (N _GAP +1) is {0, 4π / 3, 8π / 3}. When N _GAP = 3, the phase rotation 4π × _ng / (N _GAP +1) is {0, 4π / 4, 8π / 4, 12π / 4}. Thus, phase rotation _{4π × n g / (N GAP} +1) _is, n g = 0, _..., and cancel each other when added to impart phase rotation _{4π × n g / (N GAP} +1) in _{N GAP} It has the property of being zero. Therefore, one of FT_CAL _z ^ND (k, f _s , w), FT_ALIAS _z ^ND (k, f _s , w), and FT_2ndALIAS _z ^ND (k, f _s , w) is added in-phase and the remaining one is added. two are ^{_{FT_GAP_CI z ND (n g, k}} , f s, w) terms of canceled each other, the signal level difference is clearly caused relationship. Therefore, even when the SNR of the received signal is low, it is possible to determine the presence or absence of a loopback signal, and it is also possible to determine whether a (primary) or (secondary) loopback signal is included.

For the signal of the Doppler frequency index f _{s_cfar} determined to be a (primary) return signal, the return determination unit 216 converts the Doppler frequency index as described below and _outputs the signal to the direction estimation unit 214 together with the discrete time index _{k_cfar.} I do.
If Doppler frequency index _{fs_cfar} ≧ 0, convert DopConv ( _{fs_cfar} ) = _{fs_cfar− Nc} and output.
If the Doppler frequency index _{fs_cfar} <0, convert to DopConv ( _{fs_cfar} ) = _{fs_cfar} + _Nc and output.

The return determining unit 216 _converts the Doppler frequency index of the signal of the Doppler frequency index _{fs_cfar} determined to be the (secondary) return signal as follows, and _sends the signal to the direction estimating unit 214 together with the discrete time index _{k_cfar} . Output.
For Doppler frequency index _{f s_cfar} ≧ 0, then convert _{DopConv (f s_cfar) = f s_cfar} + 2N c, outputs.
For Doppler frequency index _{f s_cfar} <0, then converted _{DopConv (f s_cfar) = f s_cfar} -2N c, and outputs.

The return determination unit 216 outputs the signal of the Doppler frequency index f _{s_cfar} determined not to be a return signal to the direction estimation unit 214 together with the discrete time index _{k_cfar} without converting the Doppler frequency index as described below.
DopConv ( _{fs_cfar} ) = _{fs_cfar}

Direction estimating unit 214, based on the output from the folded determination unit 216, the output from the Doppler analysis unit 213 generates the equation virtual reception array correlation vector h shown in _{(51) (k, f s} , w), the direction Perform estimation processing.

In, it summarizes the w-th output from the antenna system processing unit 201 # 1 to # _{N a} Doppler analysis unit 213 # 1 obtained by performing similar processing in the signal processing units 207 of # _{N t} or less and including _N t _{N a} number of elements which is the product of the number of transmit antennas _{N t} as shown in equation (51) and the number of reception antennas _{N a,} the virtual reception array correlation vector _{h (k _cfar, f s_cfar,} w ). The virtual reception array correlation vector h ( _{k_cfar} , _{fs_cfar} , w) is used for processing for estimating the direction of the reflected wave from the target based on the phase difference between the reception antennas Rx. Here, z = 1, ..., a _{N a, ND = 1, ...} , a _{N t.}

h _{cal [b]} is an array correction value for correcting a phase deviation and an amplitude deviation between the transmission array antennas and between the reception array antennas. _Here, b = 1, ..., a _N t N _a.

Further, since the transmission antenna Tx is switched in a time division manner, different phase rotations occur at different Doppler frequencies f. TxCAL ⁽¹⁾ (f),..., TxCAL ^(Nt) (f) are transmission phase correction coefficients for correcting the phase rotation to match the phase of the reference transmission antenna. For example, when Tx # 1 is used as the reference transmission antenna, the transmission phase correction coefficient is given by Expression (53).

In this case, the virtual reception array correlation vector _h of the formula _{(53) (k _cfar, f} s_cfar, w) _is a column vector comprised of N a _{N r} number of elements.

The arrival direction estimation calculates a spatial profile by changing the azimuth direction θ in the direction estimation evaluation function value P _H (θ, _{k_cfar} , _{fs_cfar} , w) within a predetermined angle range. Then, in the arrival direction estimation, a predetermined number of the maximum peak directions of the spatial profile are extracted in descending order, and the elevation angle direction of each maximum peak is output as an arrival direction estimation value.

Radar apparatus 1 according to Embodiment 5 switches a plurality of transmission antennas Tx in a time-division manner, and transmits a transmission signal _Nc times from each transmission antenna Tx. At this time, the radar apparatus 1 provides a transmission gap period T _GAP every time a transmission signal is transmitted N _c / (N _GAP +1) times from each transmission antenna Tx. That is, the transmission gap period T _GAP is provided N _GAP times. In addition, the radar device 1 includes a return determination unit 216. Then, in the radar apparatus 1, the loopback determining unit 216 determines whether or not the output signal from the Doppler analyzer 213 includes a loopback signal based on the phase rotation occurring during the transmission gap period T _GAP . As a result, the Doppler frequency range in which ambiguity does not occur can be expanded to twice or more as compared with the case where the transmission gap period T _GAP is not provided.

When the transmission gap period T _{GAP is set} to N _{t Tr} / (N _GAP +1), the performance (accuracy) of determining whether or not the signal is a loopback signal is the highest. However, the transmission gap period T _GAP is not limited to this, and may be about N _{t Tr} / (N _GAP +1), or a period before and after that.

Further, when transmitting a transmission signal _Nc times from each transmission antenna Tx, and transmitting a transmission signal _Nc / (N _GAP +1) times from each transmission antenna Tx, a transmission gap period T _GAP is provided, thereby returning the transmission signal. The performance (accuracy) of determining whether to include a signal is highest. However, the timing at which the transmission gap period T _GAP is provided is not limited to this, and may be about N _c / (N _GAP +1) times, or after transmission about the number of times.

(Embodiment 6)
The transmission gap period described above is not limited to the time-division multiplexed MIMO radar device described above. For example, a MIMO radar device that transmits signals simultaneously from a plurality of transmission antennas Tx using code multiplexing (hereinafter, “code multiplexed MIMO radar device”) "May be referred to as").

  A MIMO radar apparatus using code multiplex transmission is described in, for example, Patent Document 3 (see, for example, FIG. 1). In Patent Literature 3, phase modulation (0 ° or 180 °) based on a different code sequence is applied to each transmission antenna for each repeated transmission of a transmission signal (chirp signal), and code multiplex transmission is performed from a plurality of transmission antennas. .

By detecting signals received by a plurality of receiving antennas, distance information of the code-multiplexed received signal is extracted. The distance information obtained for each repetitive transmission of the transmission signal is multiplied by an inverse code string for each transmission antenna to separate the code-multiplexed reception signal, and subjected to velocity Fourier transform processing to perform velocity (Doppler) information. Is extracted. There was thus obtained, performs azimuth Fourier transform process using a velocity (Doppler) information of the number of reception antennas N _a and multiple code multiplex number N _t (N _a × N _t) strains.

  In this configuration, since the transmission is performed simultaneously from a plurality of transmission antennas every time the transmission signal is repeatedly transmitted, the reception signal can be sampled every time the transmission signal is repeatedly transmitted. For this reason, the Doppler velocity range that satisfies the sampling theorem (in other words, does not cause frequency aliasing and does not generate Ambiguity) can be expanded as compared with the time division multiplexing method.

  However, before the velocity direction Fourier transform process, the received signal contains Doppler fluctuations due to the movement of the target or the radar device, because the signals multiplexed by the code are separated by multiplying the inverse code sequence for each transmitting antenna. And the orthogonality between codes is reduced, and intersymbol interference occurs.

  Since the code sequence is superimposed every time the transmission signal is repeatedly transmitted, if intersymbol interference occurs, the peak sidelobe ratio in the velocity direction obtained by the velocity direction Fourier transform is determined by the cross-correlation characteristic between the code sequences used in code multiplex transmission. It becomes smaller than the ideal peak side lobe ratio determined.

  Therefore, when a plurality of targets are present at the same distance and the received power level difference between the plurality of target reflected waves is larger than the peak side lobe ratio in the velocity direction, the reflected wave from the target having the smaller received power is It becomes lower than the side lobe level in the velocity direction, and the possibility of no detection increases.

  The greater the Doppler variation associated with the movement of the target or radar device, the greater the intersymbol interference, the smaller the peak sidelobe ratio, and the greater the probability of undetection when multiple targets are present at the same distance. Will be.

In the sixth embodiment, the code multiplexing MIMO radar apparatus performs transmission with the transmission gap period T _GAP described in the third embodiment. Thus, similarly to Embodiment 3 described above, the detection range of the Doppler frequency (relative speed) in which ambiguity (Ambiguity) does not occur can be expanded. In addition, even when the received signal includes Doppler fluctuation due to movement of the target or the radar apparatus 1, occurrence of intersymbol interference can be suppressed.

FIG. 18 is a diagram illustrating a configuration example of a radar device 1 according to Embodiment 6. The configuration illustrated in FIG. 18 corresponds to a configuration in which transmission with the transmission gap period T _GAP described in the third embodiment (FIGS. 14 and 15) is performed in the code multiplexing MIMO radar device.

For example, the code multiplexing MIMO radar device 1 shown in FIG. 18 differs from the configuration illustrated in FIG. 14 in that the orthogonal transmission unit 108 in the radar transmission unit 100 is replaced with the orthogonal code generation unit 108 instead of the switching control unit 105 and the transmission RF switching unit 106. When, a code multiplexing unit 109 including the first to code multiplication section 191 # 1~191 # _{N t} of the _{N t,} that it comprises the different.

  Also, the radar receiving section 200 shown in FIG. 18 is different from the configuration illustrated in FIG. 14 in that a code multiplexing / demultiplexing section 217 is provided between the loopback determining section 216 and the direction estimating section 214. The difference is that the output of the orthogonal code generation unit 108 is input to the output switching unit 211. The code multiplexing / demultiplexing unit 217 separates the code-multiplexed received signal based on, for example, the result of the determination (or detection) of the presence / absence of the return of the Doppler frequency in the return determination unit 216.

  By using the configuration illustrated in FIG. 18, even when the received signal includes Doppler fluctuation due to the movement of the target or the radar apparatus 1, the code demultiplexing is performed after correcting the phase fluctuation caused by the Doppler fluctuation. It becomes possible.

  Hereinafter, an operation of the code multiplexing MIMO radar apparatus 1 according to the sixth embodiment will be described while focusing on differences from the third embodiment.

The radar transmitting section 100 performs MIMO radar transmission using code multiplexing. For example, the orthogonal code generator 108 is comprised of an orthogonal code length _{L OC N t} pieces of orthogonal code sequences _{OCS ND = {OC ND (1} ), OC ND (2), ..., OC ND (L OC)} generate I do. Here, ND = 1, ..., a _{N t.}

For example, the orthogonal code generation unit 108 cyclically varies the orthogonal code element index OC_INDEX indicating the elements of the orthogonal code sequences OCS _{1 to} OCS _Nt for each radar transmission cycle (T _r ), so that the orthogonal code sequence OCS outputs _{1 ~OCS Nt} elements _OC 1 of the _{(OC_INDEX)} ~OC Nt _{(OC_INDEX)} to the first to _{N t} code multiplication section 191 # 1 to 191 of # _{N t.} Further, the orthogonal code generation unit 108 outputs the element index OC_INDEX to the output switching unit 211 of the radar reception unit 200, for example, for each radar transmission cycle (T _r ).

_Here, OC_INDEX = 1,2, ..., a _{L OC,} in M-th transmission period, a _{OC_INDEX = MOD (M-1,} L OC) +1. MOD (x, y) is a modulo operator, and is a function that outputs the remainder after dividing x by y.

  For the orthogonal code sequence generated by the orthogonal code generation unit 108, for example, codes that are mutually uncorrelated are used. For example, the orthogonal code generation unit 108 uses a Walsh-Hadamard-code for an orthogonal code sequence.

When N _t = 2, since the orthogonal code length L _OC = 2 of the Walsh-Hadamard-code, the orthogonal code generation unit 108 sets OCS ₁ = {1,1} and OCS ₂ = {1, −1}. Is generated.

Also, when N _t = 4, the orthogonal code length L _OC = 4, so that the orthogonal code generation unit 108 sets the OCS ₁ = {1,1,1,1} and the OCS ₂ = {1, −1,1 , −1}, OCS ₃ = {1,1, −1, −1}, and OCS ₄ = {1, −1, −1,1}.

Note that elements constituting the orthogonal code sequence are not limited to real numbers, and may include complex values. For example, an orthogonal code using phase rotation represented by the following equation (6-1) may be used.

In this case, when N _t = 3, since the orthogonal code length L _OC = N _t , the orthogonal code generation unit 108 sets the OCS ₁ = {1,1,1}, OCS ₂ = ｛1, exp (j2π / 3), exp (j4π / 3)}, and generates an orthogonal code sequence in which OCS ₃ = {1, exp (-j2π / 3), exp (-j4π / 3)}.

Also, in the case of Nt = 4, because it is orthogonal code length _L OC = _{N t,} orthogonal code generator _{108, OCS 1 = {1,1,1,1},} OCS 2 = {1, j, -1 , -J}, OCS ₃ = {1, -1, 1, -1} and OCS ₄ = {1, -j, -1, j}.

Code multiplication section 191 # 1-191 # _{N t} of the first to _{N t,} the elements OC ₁ of the radar transmission cycle _{(T r)} orthogonal generated by the orthogonal code generator 108 for every code sequence _OCS 1 _{~OCS Nt (OC_INDEX)} ~OC Nt and (OC_INDEX), as illustrated in FIGS. 19A and 19B, is multiplied to the radar transmission signal of a base band, respectively, _{N t} pieces of transmission RF section 107 # 1~17 # _{N t} Output to

Further, as illustrated in FIG. 19A, the transmission RF unit 107 # 1~107 # _{N t is} the operation of transmitting a transmission signal _{L OC} once in N p _{= L OC} × _{T r period,} over _N c / 2 times After the repetition, the transmission signal is not transmitted for the transmission gap period T _GAP .

With other words, each of the transmission RF unit 107 # 1~107 # _{N t} is a first time period at least one round send orthogonal codes that are cyclically generated _{(N p = L OC × T} r period), the transmission signal Each code-multiplexed transmission signal is transmitted for each transmission period _Tr , and the code-multiplexed transmission signal is not transmitted for a predetermined transmission gap period T _GAP thereafter.

After the transmission gap period _{T GAP} has elapsed, as illustrated in FIG. 19B, transmission RF section 107 # 1~107 # _{N t} is _again a transmission signal _{L OC} times transmitted in N p _{= L OC} × _{T r} period Is repeated _Nc / 2 times.

The transmission operation of FIGS. 19A and transmission RF section 107 # 1 to 107 as illustrated in FIG. 19B # _{N t,} transmitted from the first transmit RF section 107 # 1 transmit RF section to 107 # _{N t} of the _{N t} The signal will be transmitted L _OC × N _c times.

With other words, each of the transmission RF unit 107 # 1~107 # _{N t} after the transmission gap period _{T GAP,} cyclically generated orthogonal code and the second time period at least one round send _(N p _{= L OC} × T _r period), each transmission cycle T _r, transmitting each transmission signal code-multiplexed.

Here, the transmission gap period T _GAP is set to N _p / 2, which is equivalent to a half period of the transmission period N _p = L _OC × _Tr period of the orthogonal code, which is the sampling period in the Doppler analysis unit 213. That is, T _GAP = L _OC × T _r / 2.

Next, the operation of the radar receiving unit 200 illustrated in FIG. 18 will be described.
Output switching unit 211 in the z-th signal processing unit 207, based on an orthogonal code element index OC_INDEX from the orthogonal code generator 108, the output of the correlation calculation unit 210 of each transmission _{cycle, L OC} number of Doppler analysis unit 213 is selectively switched to the OC_INDEX-th Doppler analyzer 213 for output.

That is, the output switching unit 211, the transmission cycle _{T r} of the M-th selects _{OC_INDEX = MOD (M-1,} L OC) +1 th Doppler analysis unit 213. Further, the output switching unit 211 does not select all the Doppler analysis units 213 during the transmission gap period T _GAP .

In the z-th signal processing unit 207, a plurality of (L _OC ) Doppler analysis units 213 output the first half of N _c / 2 times until the transmission gap period T _GAP starts, and the transmission gap period T The output of Nc / 2 times in the latter half after the end of the _GAP and the Doppler analysis are separately performed twice. When _Nc is a power of 2, an FFT (fast Fourier transform) process as shown in Expressions (6-2) and (6-3) can be applied to Doppler analysis.

For example, the FFT processing on the output of N _c / 2 times in the first half until the transmission gap period T _GAP is started is represented by Expression (6-2).

Further, the FFT processing on the output of N _c / 2 times in the latter half after the transmission gap period T _GAP ends is represented by Expression (6-3).

^{_{Here, FT_FH_CI z (OC_INDEX) (k}} , f s, w) represents the output of the w th according OC_INDEX th Doppler analysis unit 213 in the z-th signal processing unit 207, the Doppler in the discrete time k 4 shows the Doppler frequency response of the frequency index f _s to the first half N _c / 2 outputs before the start of the transmission gap period T _GAP .

FT_SH_CI _z ^(OC_INDEX) (k, f _s , w) represents the w-th output of the OC_INDEX-th Doppler analyzer 213 in the z-th signal processor, and the Doppler frequency index f at the discrete time k _FIG . 14 shows a Doppler frequency response to N _c / 2 outputs of the second half after the transmission gap period T _GAP ends in _s .

It should be noted, is a _{OC_INDEX = 1~L OC, k = 1} , ..., a _{(N r + N u) N} s / N o, w is an integer of 1 or more. j is an imaginary unit. In addition, z = 1, ..., a _{N a.}

^{_{Further, FT_FH_CI z (OC_INDEX) (k}} , f s, w) is the FFT size of Nc, is obtained by the _N c / 2 pieces of data in the latter half portion padded zero (zero padding). ^{_{Further, FT_SH_CI z (OC_INDEX) (k}} , f s, w) is the FFT size of _{N c,} in which the _N c / 2 pieces of data of the first half portion and zero-padding.

Therefore, the maximum Doppler frequency without aliasing derived from the sampling theorem is ± 1 / (2L _OC × T _r ). Also, Doppler frequency interval of the Doppler frequency index _{f s} is _{1 / {L OC × N c} × T r}, a range of Doppler frequency index _{f s is, f s = -N c / 2} + 1, ..., 0, ... , N _c / 2.

In the FFT processing, a window function coefficient such as a Han window or a Hamming window may be multiplied. By applying the window function, side lobes generated around the beat frequency peak can be suppressed. A coefficient having an FFT size of _Nc may be applied to the window function coefficient. For example, the first half N _c / 2 window function coefficients are used in calculating FT_FH_CI _z ^(OC_INDEX) (k, f _s , w), and the second half N _c / 2 coefficients are used in FT_SH_CI _z ^(OC_INDEX) ( _{k, f s,} used in the calculation of w).

CFAR unit _215, the output of the w th from _{L oc} number of Doppler analysis unit ^{_{213, FT_FH_CI z (OC_INDEX) (}} k, f s, w), ^{_{and, FT_SH_CI z (OC_INDEX) (k}} , f s, w ) To perform CFAR processing.

For example, the CFAR unit 215 calculates a power addition value represented by Expression (6-4), and calculates a two-dimensional CFAR of a discrete time axis (corresponding to a distance axis) and a Doppler frequency axis (corresponding to a relative velocity). Processing or CFAR processing combining one-dimensional CFAR processing is performed. For the CFAR processing combining two-dimensional CFAR processing or one-dimensional CFAR processing, for example, the processing described in Non-Patent Document 2 may be applied.

CFAR unit 215 sets the adaptive threshold using the CFAR processing, discrete time index k _{_Cfar} larger received power than the threshold value, the Doppler frequency index _{f S_cfar,} instructs the direction estimation unit 214 and the folded-back determination unit 216 .

The loopback determination unit 216 extracts the output of the Doppler analysis unit 213 based on the discrete time index _{k_cfar} and the Doppler frequency index _{fs_cfar} instructed by the CFAR unit 215, and _obtains the following equations (6-5) and (6). A determination process for determining whether or not the signal is a loopback signal is performed by the determination method using -6). For example, the return determination unit 216 determines that the signal is not a return signal when Expression (6-5) is satisfied, and determines that the signal is a return signal when Expression (6-6) is satisfied.

である。 In addition, in Formula (6-5) and Formula (6-6),

It is.

の項は、ドップラ周波数インデックスｆ_{ｓ_cfar}の信号に対する送信ギャップ期間中の位相回転を補正するために導入されている。 here,

Is introduced to correct the phase rotation during the transmission gap for the signal with the Doppler frequency index _{fs_cfar} .

の項は位相反転された出力、すなわち

となる。 At this time, when the signal of the Doppler frequency index f _{s_cfar} is a loopback signal, when Doppler frequency index f _{s_cfar} ≧ 0, a phase change by the Doppler frequency index of ( _{fs_cfar−} Nc) _occurs during the transmission gap period, When the Doppler frequency index _{fs_cfar} <0, a phase change corresponding to the Doppler frequency index of ( _{fs_cfar} + _Nc ) _occurs during the transmission gap period.

Is a phase-inverted output, that is,

Becomes

は、

よりも電力的に小さくなる。 Therefore, _when the signal of the Doppler frequency index _{fs_cfar} is a loopback signal,

Is

Power.

は、

よりも電力的に小さくなる。 On the other hand, _when the signal of the Doppler frequency index _{fs_cfar} is not a return signal,

Is

Power.

For such a reason, it is possible to apply the return determination method as described above. As a result of the determination, when it is determined that the signal of the Doppler frequency index f _{s_cfar} is a loopback signal, the loopback determination unit 216 determines the conversion result of the Doppler frequency index as illustrated in the following (1) and (2). Is output.

(1) If Doppler frequency index f _{s_cfar} ≧ 0, DopConv ( _{fs_cfar} ) = _{fs_cfar} −Nc
(2) When Doppler frequency index _{fs_cfar} <0, DopConv ( _{fs_cfar} ) = _{fs_cfar} + Nc

  DopConv (f) represents the result of conversion of the Doppler frequency index to the Doppler frequency index f based on the determination result of the aliasing signal.

On the other hand, when it is determined that the signal of the Doppler frequency index f _{s_cfar} is not a loopback signal, the loopback determination unit 216 outputs a conversion result of the Doppler frequency index as follows.
DopConv ( _{fs_cfar} ) = _{fs_cfar}

The code multiplex / separation unit 217 separates a multiplexed signal using orthogonal codes based on the output of the loopback determination unit 216. For example, the signal multiplex-transmitted from the ND-th transmission antenna Tx # ND has a complex conjugate of the orthogonal code element used at the time of transmission, as shown in Expressions (6-9) and (6-10). *) And then multiplied by the Doppler analysis result for each code element index and added to separate them. It should be noted, ND = 1, ..., a _{N t.} The term exp in the equation (6-9) is provided to correct a phase fluctuation caused by a transmission time delay of the orthogonal code.

Direction estimating unit 214, based on the output of the code demultiplexing section 217, generates a virtual reception array correlation vector _{h (k, f s, w} ), performs the direction estimation process based on the vector. For example, including _{N t} × _{N a} number of elements which is the product of the number of transmit antennas _{N t} of the formula (6-11) and the number of reception antennas _{N a,} the virtual reception array correlation vector _h (k _cfar, f _{Using s_cfar} , w), direction estimation is performed on the reflected wave from the target based on the phase difference between the receiving antennas Rx.

Here, the virtual reception array correlation vector _{h (k _cfar, f s_cfar,} w) is the w from Doppler analysis unit 213 obtained in each of the first signal processing unit 207 from the signal processing unit 207 of the _{N a} It is equivalent to a vector in which the output of the number is put together. Incidentally, z = 1, ..., a _{N a, ND = 1, ...} , a _{N t.}

Here, h_cal _[b] is an array correction value for correcting a phase deviation and an amplitude deviation between the transmission array antennas and between the reception array antennas. Also, b = 1,..., (N _t × N _a ). Virtual reception array correlation vector _{h (k _cfar, f s_cfar,} w) is a column vector of _{N t} × _{N a} number of elements.

The direction estimating unit 214 calculates the spatial profile by changing the azimuth direction θ in the direction estimation evaluation function value P _H (θ, _{k_cfar} , _{fs_cfar} , w) within a predetermined angle range, and increases the local maximum peak direction. A predetermined number is sequentially extracted, and the azimuth direction of each local maximum peak is output as an arrival direction estimation value. Further, as the positioning result of the radar reflected wave, information on the maximum peak level may be output together with the azimuth direction. Also, as the positioning result of the radar reflected wave, it _outputs arrival time information (distance information) based on _{k_cfar} and DopConv ( _{fs_cfar} ) after the return determination as Doppler frequency information (relative speed information).

As described above, in code multiplexing MIMO radar apparatus 1 according to Embodiment 6, radar transmitting section 100 uses a plurality of transmission antennas 107 and code multiplexing to transmit (L _OC × Nc) times from each transmission antenna 107. , A transmission gap period T _GAP is provided after transmission from each transmission antenna 107 _LOC × Nc / 2 times.

In the radar receiving section 200, the return determination section 216 determines whether or not the output of the Doppler analysis section 213 includes a return signal, thereby expanding the Doppler frequency range in which the ambiguity of the Doppler frequency does not occur. For example, a larger L _OC to twice the Doppler frequency range in the case where the sampling period.

  Also, due to the expansion of the Doppler frequency range in which the ambiguity of the Doppler frequency does not occur, the code demultiplexing unit 217 may add the result of performing the Doppler analysis for each orthogonal code element by multiplying the result by the complex conjugate of the orthogonal code element. In addition, the orthogonal code separation processing can be performed using the determination result of whether or not the signal is a folded signal.

  This makes it possible to separate code-multiplexed signals while suppressing orthogonal intersymbol interference. Therefore, the side lobe in the time direction or the frequency direction can be reduced. In principle, when there is no noise component or when it can be ignored, the side lobe can be made substantially zero.

Note that, by using L _OC × T _r / 2 for the transmission gap period T _GAP , the loopback determination performance can be maximized, but the present invention is not limited to this. For example, about L _OC × T _r / 2, or a period before and after that may be set.

Also, when a plurality of (L _OC × N _c ) radar transmissions are performed from each transmission antenna 107, a transmission gap period T _GAP is provided after each transmission antenna 107 transmits L _OC × N _c / 2 times. The return determination performance can be maximized, but is not limited to this.

For example, the transmission gap period T _GAP may be provided after transmitting about L _OC × N _c / 2, or the number of times before and after that from each transmitting antenna 107. For example, they may be set at unequal intervals in a range in which a bias of SNR (signal-to-noise ratio) does not occur.

Although an example in which one transmission gap period is provided has been described in the sixth embodiment described above, a plurality of (N _GAP ) transmission gap periods T _GAP may be provided as described in the fifth embodiment. . By providing a plurality of transmission gap periods T _GAP , it is possible to determine whether or not a higher-order aliasing signal is included in the received signal. Therefore, the effect of further expanding the Doppler frequency range in which Doppler frequency ambiguity does not occur is obtained. Can be

(Embodiment 7)
In the sixth embodiment described above, the code multiplexing MIMO radar apparatus 1 in which a pulse train is phase-modulated or amplitude-modulated in the radar transmitting section 100 and transmitted is described. In the seventh embodiment, a code multiplexing MIMO radar device 1 using a frequency-compressed pulse compression wave such as a chirp pulse in a radar transmission unit 100 will be described.

FIG. 20 is a diagram illustrating a configuration example of the code multiplexing MIMO radar device 1 using a frequency-modulated chirp pulse for a radar transmission signal. The code multiplexing MIMO radar apparatus 1 illustrated in FIG. 20 differs from the configuration illustrated in Embodiment 2 (FIG. 12) in the radar transmitting section 100 in the switching control section 105, the transmitting RF section 107, and the transmitting antenna. instead of the switching unit 121, an orthogonal code generator 108, a transmission RF section 107 # 1~107 # _{N t} of the first to _{N t,} code multiplication section 191 of the first to _{N t} # 1-191 # a code multiplexing unit 109 including N _t, that it comprises the different.

  In addition, the radar receiving unit 200 shown in FIG. 20 includes a loopback determining unit 216 and a code multiplexing / demultiplexing unit 217 between the signal processing unit 207 and the direction estimating unit 214 as compared with the configuration illustrated in FIG. The difference is that the output of the orthogonal code generation unit 108 is input to the output switching unit 211. The code multiplexing / demultiplexing unit 217 separates the code-multiplexed received signal based on, for example, the result of the determination (or detection) of the presence / absence of the return of the Doppler frequency in the return determination unit 216.

  Hereinafter, the operation of the code multiplexing MIMO radar apparatus 1 according to the seventh embodiment will be described while focusing on differences from the second embodiment.

  In radar transmission section 100, radar transmission signal generation section 101 generates a frequency modulation signal (frequency chirp signal) by modulation signal generation section 122 and VCO 123 as described in the second embodiment. The generated frequency chirp signal is input to code multiplexing section 109 and mixer section 224 of reception RF section 203 via directional coupling section 124.

Orthogonal code generator 108, as in the sixth embodiment, made of an orthogonal code length _{L OC N t} pieces of orthogonal code sequences _{OCS ND = {OC ND (1} ), OC ND (2), ..., OC ND ( L _OC )}. Here, ND = 1, ..., a _{N t.}

For example, the orthogonal code generation unit 108 cyclically varies the orthogonal code element index OC_INDEX indicating the elements of the orthogonal code sequences OCS _{1 to} OCS _Nt for each radar transmission cycle (T _r ), so that the orthogonal code sequence OCS outputs _{1 ~OCS Nt} elements _OC 1 of the _{(OC_INDEX)} ~OC Nt _{(OC_INDEX)} to the first to _{N t} code multiplication section 191 # 1 to 191 of # _{N t.} Further, the orthogonal code generation unit 108 outputs the element index OC_INDEX to the output switching unit 211 in the signal processing unit 207 of the radar reception unit 200 for each radar transmission cycle (T _r ).

_Here, OC_INDEX = 1,2, ..., a _{L OC,} in M-th transmission period, a _{OC_INDEX = MOD (M-1,} L OC) +1. MOD (x, y) is a modulo operator, and is a function that outputs the remainder after dividing x by y.

The first to the code multiplication section 191 of the _{N t,} as in the sixth embodiment, elements OC orthogonal code sequence _OCS 1 _{~OCS Nt} produced by the radar transmission cycle _{(T r)} orthogonal code generator 108 for each _{1 (OC_INDEX) ~OC Nt (OC_INDEX)} , radar transmission signal of a base band (in this case, the frequency chirp signal) is multiplied to each, _{N t} pieces of transmission RF section 107 # 1~107 # _{N t} output I do.

Also, as in the sixth embodiment, the transmission RF unit 107 # 1~107 # _{N t is} the operation of transmitting a transmission signal _{L OC} once in N p _{= L OC} × _{T r period,} over _N c / 2 times After the repetition, the transmission signal is not transmitted over the transmission gap period T _GAP .

After a transmission gap period _{T GAP,} the RF transmission unit 107 # 1~107 # _{N t} is _again an operation of transmitting a transmission signal _{L OC} once in N p _{= L OC} × _{T r period, N} c / 2 times Repeat over.

The transmission operation of the transmission RF unit 107 # one to one hundred and seven # _{N t,} the transmission signal of the transmission RF section 107 # _{N t} of the first transmit RF section 107 # 1 from the _{N t is, L OC} × _{N c} Will be sent twice.

Here, the transmission gap period T _GAP is set to Np / 2 corresponding to a half period of the orthogonal code transmission period N _p = L _OC × T _r period which is the sampling period in the Doppler analysis unit 213. That is, T _GAP = L _OC × T _r / 2.

The output of the code multiplication section 191 of the first to _{N t} is amplified to a predetermined transmission power by the transmission RF unit 107, it is radiated into space from the transmission antenna Tx # 1 to TX # _{N t} constituting a transmission array antenna unit You.

Next, the operation of the radar receiving unit 200 illustrated in FIG. 20 will be described. In radar receiver 200, the operation or processing of the signals received by the receiving antennas Rx # 1~Rx # _{N a} which forms a receiving array antenna unit up to the signal output of the R-FFT section 220 in Embodiment 2 This is the same as the operation or processing described.

Here, the beat frequency spectrum response output from the z-th R-FFT unit 220 in the z-th signal processing unit 207 obtained by the M-th chirp pulse transmission is represented by AC_RFT _z (f _b , M). f _b represents the index number of the beat frequency output from the R-FFT section 220, and f _b = 0,..., N _data / 2. Frequency index f _b is the more delay time of the reflected wave is small small (in other words, close distance to the target) represents the beat frequency.

The output switching unit 211 in the z-th signal processing unit 207 converts the output from the R-FFT unit 220 for each transmission cycle into L _OC Doppler analysis based on the orthogonal code element index OC_INDEX from the orthogonal code generation unit 108. And selectively switches to the OC_INDEX-th Doppler analyzer 213 in the section 213 for output.

For example, the output switching unit 211, the M-th transmission cycle _{T r,} selects _{OC_INDEX = MOD (M-1,} L OC) +1 th Doppler analysis unit 213. Further, the output switching unit 211 does not select all the Doppler analysis units 213 during the transmission gap period T _GAP .

The plurality of (L _OC ) Doppler analyzers 213 in the z-th signal processing unit 207 output N _c / times in the first half until the transmission gap period T _GAP starts, and the transmission gap period T _GAP ends. Doppler analysis is separately performed by dividing the output into N _c / 2 outputs in the latter half of the process. When _Nc is a power of 2, an FFT process represented by Expressions (6-12) and (6-13) can be applied to Doppler analysis.

For example, the FFT processing on the output of N _c / 2 times in the first half until the transmission gap period T _GAP is started is represented by Expression (6-12).

On the other hand, the FFT processing for the output of N _c / 2 times in the latter half after the transmission gap period T _GAP ends is represented by Expression (6-13).

Here, FT_FH_CI _z ^(OC_INDEX) (f _b , f _s , w) represents the w-th output by the OC-INDEX-th Doppler analyzer 213 in the z-th signal processor 207, and is represented by the frequency index f _b . 2 shows the Doppler frequency response of the Doppler frequency index f _s to the output of N _c / 2 times in the first half before the transmission gap period T _GAP is started.

FT_SH_CI _z ^(OC_INDEX) (f _b , f _s , w) represents the w-th output by the OC-INDEX-th Doppler analyzer 213 in the z-th signal processor 207, and represents the w-th output at the frequency index f _b FIG. 14 shows a Doppler frequency response of the Doppler frequency index f _s to the output of N _c / 2 times in the latter half after the transmission gap period T _GAP ends. Note that OC_INDEX = 1 to L _OC , f _b = 0,..., N _data / 2, and w is an integer of 1 or more. j is an imaginary unit. In addition, z = 1, ..., a _{N a.}

^{_{Further, FT_FH_CI z (OC_INDEX) (f}} b, f s, w) is the FFT size of _{N c,} in which the _N c / 2 pieces of data in the latter half portion and zero-padding. ^{_{FT_SH_CI z (OC_INDEX) (f b}} , f s, w) is the FFT size of _{N c,} in which the _N c / 2 pieces of data of the first half section and zero-padding.

Therefore, the maximum Doppler frequency at which aliasing does not occur, derived from the sampling theorem, is ± 1 / (2L _OC × T _r ). Also, Doppler frequency interval of the Doppler frequency index _{f s} is _{1 / {L OC × N c} × T r}, a range of Doppler frequency index _{f s} is _{f s = -N c / 2 +} 1, ..., 0, ..., N _c / 2.

At the time of the FFT processing, a window function coefficient such as a Han window or a Hamming window may be multiplied. By applying the window function, side lobes generated around the beat frequency peak can be suppressed. As the window function coefficient, an FFT size of Nc is applied, and the first half N _c / 2 window function coefficients are used when calculating FT_FH_CI _z ^(OC_INDEX) (f _b , f _s , w). using the coefficients of ^{_{N c / 2 FT_SH_CIz (OC_INDEX)}} (f b, f s, w) when calculating the.

Subsequent CFAR section 215, folded determination unit 216, code demultiplexing section 217, and the process in the direction estimating unit 214, the processing by replacing the discrete time k used in the sixth embodiment the frequency index _{f b} of the beat frequency Equivalent to.

  With the above configuration and operation, in addition to the effects or advantages described in the second embodiment, the same effects or advantages as in the sixth embodiment can be obtained.

  As described above, a plurality of embodiments according to the present disclosure have been described.

  Each functional block used in the description of the above embodiment is typically realized as an LSI which is an integrated circuit. These may be individually integrated into one chip, or may be integrated into one chip so as to include some or all of them. Although an LSI is used here, it may be called an IC, a system LSI, a super LSI, or an ultra LSI depending on the degree of integration.

  Further, the method of circuit integration is not limited to LSI, and may be realized by a dedicated circuit or a general-purpose processor. After the LSI is manufactured, a field programmable gate array (FPGA) that can be programmed, or a reconfigurable processor that can reconfigure connection and settings of circuit cells inside the LSI may be used.

  Furthermore, if an integrated circuit technology that replaces the LSI appears due to the progress of the semiconductor technology or another technology derived therefrom, the functional blocks may be naturally integrated using the technology. Application of biotechnology, etc. is possible.

  One aspect of the present disclosure is useful for radar systems.

REFERENCE SIGNS LIST 1 radar device 100 radar transmission unit 101 radar transmission signal generation unit 102 code generation unit 103 modulation unit 104 LPF
Reference Signs List 105 switching control unit 106 transmission RF switching unit 107 transmission RF unit 108 orthogonal code generation unit 109 code multiplexing unit 191 code multiplication unit 111 code storage unit 112 D / A conversion unit 121 transmission antenna switching unit 122 modulation signal generation unit 123 voltage controlled oscillator 124 Directional coupling unit 200 Radar receiving unit 201 Antenna system processing unit 203 Reception radio unit 204 Amplifier 205 Frequency conversion unit 206 Quadrature detection unit 207 Signal processing unit 208, 209, 228 A / D conversion unit 210 Correlation calculation unit 211 Output switching unit 213 Doppler analysis unit 214 Direction estimation unit 215 CFAR unit 216 Fold determination unit 217 Code demultiplexing unit 220 R-FFT unit 224 Mixer unit 226 LPF

Claims (15)

A plurality of transmission units,
A first period in which a transmission unit transmitting the transmission signal is selected from the plurality of transmission units for each transmission cycle of the transmission signal, and a first period in which each of the plurality of transmission units is selected at least once, A control unit that provides a transmission gap period, which is a period during which the transmission signal is not transmitted, between a second period after which the plurality of transmission units are selected at least one time,
Comprising,
Radar equipment. A receiving unit that receives a signal reflected by the object at each of the transmission signals,
A Doppler analyzer for analyzing the Doppler frequency component of each received signal corresponding to each of the transmitted signals,
From the Doppler frequency component, a detection unit that detects a peak Doppler frequency component that is a frequency component whose received power is greater than a threshold,
The peak Doppler frequency component of the reception signal corresponding to the transmission signal in the first period and the peak Doppler frequency component of the reception signal corresponding to the transmission signal in the second period are compared, and the peak Doppler frequency component is A determining unit for determining whether or not a return signal is included;
Based on the peak Doppler frequency component of each of the received signals, a direction estimating unit that estimates the direction of the object,
Comprising,
The radar device according to claim 1. The determination unit includes:
When it is determined that the aliasing signal is included, the peak Doppler frequency component is converted and output to the direction estimating unit,
If it is determined that the return signal is not included, output to the direction estimating unit without converting the peak Doppler frequency component,
The radar device according to claim 2. The control unit sequentially selects the plurality of transmission units,
The transmission gap period is の of a cycle in which the control unit selects one of the plurality of transmission units.
The radar device according to claim 1. The control unit selects a first transmission unit of the plurality of transmission units at a first cycle, and sets each second transmission unit other than the first transmission unit to be longer than the first cycle. Select in the second cycle,
The detection unit detects the peak Doppler frequency component from a Doppler frequency component of a reception signal corresponding to the transmission signal from the first transmission unit,
The determination unit determines whether or not the return signal is included in the peak Doppler frequency component of the reception signal corresponding to the transmission signal from the first transmission unit,
The radar device according to claim 2. The transmission gap period is の of the first cycle,
The radar device according to claim 5. The control unit sets the transmission gap period N _GAP times,
The transmission gap period is 1 / (N _GAP +1) of a cycle in which the control unit makes one round selection of the plurality of transmission units.
The radar device according to claim 1. A plurality of transmission units,
A code multiplexing unit that cyclically multiplexes code elements of orthogonal codes into transmission signals for each transmission cycle to generate code-multiplexed transmission signals,
The plurality of transmission units,
A first period for transmitting the code element of the cyclically generated orthogonal code at least once, for each transmission cycle of the transmission signal, transmitting each transmission signal obtained by code-multiplexing the code element;
After the first period, a predetermined transmission gap period, without transmitting a transmission signal code-multiplexed the code element,
After the transmission gap period, a second period for transmitting at least one cycle of cyclically generated code elements of orthogonal codes, for each transmission cycle, transmit each transmission signal code-multiplexed code elements,
Radar equipment. A receiving unit that receives the code-multiplexed transmission signal reflected by the object,
A Doppler analyzer for analyzing the Doppler frequency component of each received signal corresponding to the code element of the code-multiplexed transmission signal,
From the Doppler frequency component, a detection unit that detects a peak Doppler frequency component that is a frequency component whose received power is greater than a threshold,
A peak Doppler frequency component of the received signal corresponding to the code element of the code-multiplexed transmission signal in the first period, and a peak Doppler frequency component of the reception signal corresponding to the code element of the code-multiplexed transmission signal in the second period And a determination unit that determines whether a folded signal is included in the peak Doppler frequency component,
Comprising,
The radar device according to claim 8. Based on the determination result by the determination unit and the orthogonal code, a code demultiplexing unit that separates the received signal in which the orthogonal code is multiplexed from the output of the Doppler analysis unit,
A direction estimating unit that estimates a direction of the object based on an output of the code demultiplexing unit;
The radar device according to claim 9, comprising: The transmission gap period is の of a cycle of transmitting the code element of the orthogonal code in a loop.
The radar device according to claim 8. The transmission gap period is provided N _GAP times,
The transmission gap period is 1 / (N _GAP +1) of a cycle of transmitting the code element of the orthogonal code in one cycle.
The radar device according to claim 8. The transmission signal is a chirp pulse signal,
The radar device according to claim 8. A first period in which each of the plurality of transmission units is selected at least once, for each transmission cycle of the transmission signal, each of the transmission units transmits the transmission signal;
After the first period, a predetermined transmission gap period, the transmission signal is not transmitted,
After the transmission gap period, a second period in which each of the plurality of transmission units is selected at least once, for each transmission cycle, each transmission unit transmits the transmission signal,
Radar method. A first period for transmitting at least one cycle of cyclically generated code elements of orthogonal codes, for each transmission cycle of a transmission signal, each transmission unit transmits a transmission signal obtained by code-multiplexing the code element,
After the first period, during a predetermined transmission gap period, the transmitting units do not transmit a transmission signal obtained by code-multiplexing the code element,
After the transmission gap period, the transmission unit transmits each transmission signal in which the transmission unit code-multiplexes the code element for each of the transmission periods during a second period in which the code element of the cyclic code generated cyclically is transmitted at least once. Do
Radar method.

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