JP2020035028A - Antenna coil design method and program therefor - Google Patents

Abstract

To provide a technique for reducing trial and error design effort when designing an antenna coil.SOLUTION: An ambient environment including an antenna coil to be paired with an antenna coil to be designed is modeled (S100). A space in which an antenna to be designed is to be placed is segmented into a plurality of meshes and the optimum current amount for each mesh is calculated (S102). A shape of the antenna coil to be designed is obtained based on a current distribution of the plurality of meshes (S104).SELECTED DRAWING: Figure 4

Description

  The present invention relates to wireless power supply, and more particularly to a technique for designing an antenna coil.

  Wireless power transmission can be broadly classified into near-field type and far-field type. The near-field type is a method of transmitting power by connecting a power transmission side and a power reception side via a magnetic field or an electric field, and can transmit large power with high efficiency, but has a short transmission distance. On the other hand, the far-field type is a method of receiving power by capturing and rectifying energy emitted as an electromagnetic wave by an antenna, and is capable of long-distance transmission, but has low transmission efficiency.

  When power is supplied to home electric appliances, smartphones, and electric vehicles, a relatively large power of several tens of watts to several kW is required. Therefore, a near-field electromagnetic induction type or a magnetic resonance coupling type is used.

  Regarding the electromagnetic induction type, the Wireless Power Consortium (WPC) was established in 2008, an industry group that established international standards, and in 2010, Qi, the first international standard for wireless power supply, was formulated. Initially, it was designed to support low power of 5 W or less, but in 2015 a standard was established that supports up to 15 W. At present, the AirFuelAlliance, a standardization organization following WPC, has established a standard of up to 50W. However, the electromagnetic induction type has a feature that the transmission distance is limited to several centimeters and is susceptible to misalignment. Therefore, it has more significance as a non-contact power supply terminal than wireless power transmission.

  On the other hand, the magnetic resonance coupling type solves the problem of the electromagnetic induction type because it has features such as medium-range and high-efficiency transmission, resistance to misalignment, and simple expansion of the transmission distance by a relay coil. .

  The magnetic resonance coupled wireless power transmission system was announced by a research group at the Massachusetts Institute of Technology in 2007, and has achieved a transmission efficiency of 50% or more at a distance of 1 to 2 m (Non-Patent Document 1). Here, the basic principle of the magnetic resonance coupling type will be described. FIG. 1 is a block diagram of a magnetic resonance coupling type wireless power supply system. The wireless power supply system 2 includes a power transmitter (power transmitter) 10 and a power receiver (power receiver) 20. In some cases, a repeater that relays electric power is provided between the power transmitting device 10 and the power receiving device 20.

The power transmission device 10 includes a power transmission coil L _TX , a resonance capacitor C _TX, and a drive circuit 12. The drive circuit 12 generates a magnetic field by passing an alternating current through the coil L _TX on the power transmission side and exciting the coil, as in the principle of the electromagnetic induction method, and induces a current by capturing the magnetic field with the power receiving coil L _RX . Transmit power.

The power receiving device 20 includes a power receiving coil L _RX , a resonance capacitor C _RX, and a load 22. Load 22 including a rectifier circuit for rectifying an alternating current to flow in the receiving coil L _RX.

  In the resonance-coupled wireless power transmission, energy is transmitted by resonating a near electromagnetic field with a transmitting / receiving resonator. In particular, the magnetic resonance coupling type, which resonates the magnetic field, has features such as medium-distance and high-efficiency transmission, resistance to misalignment, and easy expansion of the transmission distance using relay coils. It is being actively performed. The magnetic resonance coupling type is expected to be applied to various applications.

FIGS. 2A and 2B are equivalent circuit diagrams of the wireless power supply system 2. As shown in FIG. 2 (a), the power transmission coil _{L TX} and receiving coil _{L RX} is coupled with mutual inductance _{L M.} It is known that such a pair of power transmission and reception coils is represented by a T-type equivalent circuit as shown in FIG. From the equivalent circuit analysis of FIG. 2B, the maximum transmission efficiency η _max of wireless power is expressed by Expression (1) using the product of the coupling coefficient k and the quality coefficient Q.

FIG. 3 is a diagram showing the maximum efficiency with k ² Q ₁ Q ₂ as a parameter. As can be seen from FIG. 3, since the maximum efficiency monotonically increases with kQ, the coupling coefficient k or the quality coefficient Q is used as a performance index of the resonator. In the magnetic field resonance method, it can be understood that the resonance improves the efficiency by increasing the quality factor Q value.

A. Kurs, et.al., "Wireless power transfer via strongly coupled magnetic resonances", science, vol.317, no.5834, pp.83-86, July 2007. H. D. Lang et.al., "Convex optimization of wireless power transfer systems with multiple transmitters." IEEE transactions on Antennas and Propagation, vol.62, no.9, pp.4623-4636, Sept. 2014 Y. Narusue et.al., "Distributed reactance compensation for printed spiral coils in wileress power transfer," Wireless Power Transfer Conference (WPTC) IEEE, pp.1-4 May 2017.

  Conventionally, the design of a resonator used in an actual environment relies on experience, and optimization by cut and try is required. This is because the transmission efficiency depends on innumerable parameters such as the shape, size, number of turns, and pitch of the resonator, and metal pieces and electronic components of the power receiving device exist around the resonator, and loss in surrounding materials is reduced. This is because it occurs. Therefore, when designing the resonator, the parameters are adjusted by paying sufficient attention to these operating environments, and there is a problem that the design cost increases.

  The present invention has been made in such a situation, and an exemplary object of one embodiment of the present invention is to provide a technique for reducing the effort of trial and error design when designing an antenna coil.

  One embodiment of the present invention relates to a method for designing an antenna coil. The antenna coil design method includes a step of modeling a surrounding environment including a pair of antenna coils, a step of dividing a space in which an antenna to be designed is to be arranged into a plurality of meshes, and an optimization of each mesh in the modeled surrounding environment. And calculating the shape of the antenna coil to be designed from the current distribution of the plurality of meshes.

  Another embodiment of the present invention is a method for designing an antenna coil. The antenna coil design method includes a step of modeling a surrounding environment including a pair of antenna coils, a step of dividing a space in which an antenna to be designed is to be arranged into a plurality of meshes, and an optimization of each mesh in the modeled surrounding environment. Calculating the magnetic dipole moment of the plurality of meshes, obtaining contour lines of the distribution of magnetic dipole moments of the plurality of meshes, and obtaining the shape of the antenna coil based on the contour lines.

  It is to be noted that any combination of the above-described components and any conversion of the expression of the present invention between an apparatus, a method, a system, a recording medium, a computer program, and the like are also effective as embodiments of the present invention.

According to the present invention, antenna design can be simplified.
Can be provided.

1 is a block diagram of a magnetic resonance coupling type wireless power feeding system. 2A and 2B are equivalent circuit diagrams of the wireless power supply system. The k _² Q ₁ Q ² is a diagram showing the maximum efficiency as parameters. It is a flowchart of the automatic design method of an antenna coil. It is a figure explaining a subdivided conductor mesh. FIG. 2 is a diagram illustrating a model of a wireless power supply system to be designed. It is a figure explaining the influence to Z matrix by sharing a conductor. It is a figure showing an example of the derived current distribution. FIG. 4 is a diagram illustrating a magnetic dipole moment in a minute region. FIG. 9 is a diagram illustrating an example of a magnetic dipole moment distribution derived from the current distribution in FIG. 8. FIGS. 11A and 11B are diagrams showing an example of the shape of the antenna coil obtained from the distribution of the magnetic dipole moment in FIG. FIG. 3 is an equivalent circuit diagram of a wireless power supply system including a plurality of transmission-side resonators. It is an equivalent circuit diagram of a transmission circuit. FIG. 4 is an equivalent circuit diagram illustrating dispersion reactance compensation. FIG. 3 is a diagram showing uniform current distribution by dispersion reactance compensation.

  Hereinafter, the present invention will be described based on preferred embodiments with reference to the drawings. The same or equivalent components, members, and processes shown in each drawing are denoted by the same reference numerals, and the repeated description will be omitted as appropriate. In addition, the embodiments do not limit the invention, but are exemplifications, and all features and combinations thereof described in the embodiments are not necessarily essential to the invention.

1. Overview As shown in FIG. 1, the wireless power supply system 2 forms a resonator including a power transmission device 10 and a power reception device 20. In the present embodiment, an automatic design method of an antenna coil (hereinafter, simply referred to as a coil) forming the resonator will be described. Coil L1 of automatic design object, the power transmission is one of a coil L _TX and receiving coil L _RX, the information such as the shape of the other (antenna coil L2 of the pair) to be known.

  FIG. 4 is a flowchart of an automatic antenna coil design method. This design method is embodied by a software program, which is loaded into a memory of a computer and causes a computer (specifically, a CPU) to execute each processing shown in the flowchart.

  First, a surrounding environment including the coil L2 forming a pair is modeled (S100). The surrounding environment includes, in addition to the antenna coil L2, various metal and dielectric components existing around the antenna coil L2 and the coil L1 to be designed. For example, it is rare that the antenna coils L1 and L2 are exposed to space, and are usually covered with a dielectric cover. In this case, the surrounding environment includes the cover.

  Subsequently, the space in which the coil L1 to be designed is to be arranged is divided into a plurality of meshes, and the optimal current amount of each mesh in the modeled surrounding environment is calculated (S102). The space may be a two-dimensional plane or a three-dimensional space.

  Subsequently, the shape of the coil L1 to be designed is obtained from the current distributions of the plurality of meshes (S104).

  The above is the outline of the antenna coil design method according to the embodiment. Hereinafter, this design method will be described more specifically.

2. Modeling of the surrounding environment and introduction of a conductor mesh When performing automatic design of an antenna coil, the physical position at which the antenna coil is to be installed, the three-dimensional shape and material of the substance around the antenna coil, and the antenna coil to be paired Is given, and the current distribution that maximizes the efficiency at the physical position where the antenna coil is installed is derived. Here, the paired resonator indicates a power receiving coil (power transmitting coil) if the design target is a power transmitting coil (power receiving coil).

  In the present embodiment, the efficiency is defined by the power efficiency. However, the loss is not considered in the RF inverter and the rectifier circuit, but is analyzed based on the coupling between the coils and the loss in the surrounding material.

  FIG. 5 is a diagram illustrating a subdivided conductor mesh. The antenna coil L1 to be designed is provided at a position facing the pair of antenna coils L2. Here, the antenna coil L1 forming a pair is simply shown as a loop coil having one turn. The space 30 in which the antenna coil L1 to be designed is to be incorporated is divided into a plurality of meshes. In this example, the space 30 is a plane with a negligible thickness, and in the drawing, it is assumed that there is no current component in a direction orthogonal to the coil surface of the antenna. When deriving the current distribution that maximizes the efficiency, it is assumed that the plane 30 has a conductor mesh 32 that is finely divided so as to approximate a conductor surface. The size of the conductor mesh 32 may be determined in consideration of the amount of calculation, required calculation accuracy, and the like. For example, the size is arranged in a 100 × 100, 50 × 50, 10 × 10 matrix. The mesh is not limited to a square, but may be rectangular according to the shape of the area where the antenna coil is set. Therefore, a matrix-like conductor mesh such as 50 × 100 may be employed.

3. Upon derivation of deriving the current distribution of the current distribution, it considers a circular current I _i flowing through each loop of the conductor mesh 32. Then, the efficiency is to calculate the value of each circle current I _i that maximizes the resulting current distribution efficiency is maximized.

FIG. 6 is a diagram illustrating a model of the wireless power supply system to be designed. When the number of loops and N, N pieces of circular current I ₁ ~I _N is regarded as the N loop coil. At this time, if a paired resonator is included, it can be considered as an N + 1 port circuit network. Therefore, the Z matrix of the (N + 1) th port is calculated using an electromagnetic field simulation or an equivalent circuit analysis, and the optimum current distribution is calculated using the Z matrix.

V = Z · I
V = {v ₁ , v ₂ ,... V _{N + 1} }
I = {I ₁ , I ₂ ,... I _{N + 1} }

  Here, when an interaction occurs between the electromagnetic field generated by the loop current and the surrounding material, it can be observed as a change in impedance in the conductor mesh, that is, a change in the Z matrix.

  In N independent power transmitting side antenna coils and a pair of power receiving side antenna coils, the current distribution of the N loop coils that maximizes the efficiency is the same as that of a MISO (Multiple-Input Single-Output) system. It can be treated as an efficiency maximization problem, and its solution is known (Non-Patent Document 2). Since the matrix R (R = Re [Z]) of the real components of the Z matrix is a semi-definite value, the derivation of the current that maximizes the efficiency is reduced to a convex optimization problem.

Also for the conductor mesh in the present embodiment, if the matrix R is a semi-positive definite value, it is possible to derive a current distribution that maximizes the efficiency as in the MISO system. The difference between the conductor mesh and the MISO system is that the former shares the conductor between adjacent loops, while the latter does not. FIG. 7 is a diagram for explaining the effect on the Z matrix by sharing a conductor. The impedance z included in both the loop i and the adjacent loop j is incorporated in Z as -z because the direction of the current flowing through the shared side is opposite. As a result, the matrix R of the conductor mesh is represented by the following equation.
Here, r _ij is a resistance component of a conductor portion shared by adjacent loops i and j. Since the conductor mesh, the pair of antenna coils and the surrounding environment can be regarded as a passive network that does not include a power supply therein, the power consumption caused by an arbitrary current vector I is always non-negative. Therefore, the matrix R satisfies Equation (3) for any real vector I.

  Since the equation (4) holds for an arbitrary real vector I, the matrix R is a semi-definite value. That is, the current distribution of the loop coil that maximizes the efficiency in the conductor mesh can be obtained in the same manner as the efficiency maximization problem in the MISO system.

4. Example of Derivation of Current Distribution As an initial study, a conductor mesh of 100 × 100 is assumed, the resistance value of one side constituting each loop is r, and the current is under the condition that the mutual inductance between each loop and the receiving resonator is constant at M. Derive the distribution. Here, the fact that the mutual inductance is constant means that the magnetic field distribution generated by the power receiving resonator is uniform.

  FIG. 8 is a diagram illustrating an example of the derived current distribution. Here, in the horizontal direction, a positive (negative) value indicates a right (left) direction, and a positive (negative) value in a vertical direction indicates an upward (down) direction.

5. Derivation of Antenna Coil Shape Based on Magnetic Dipole Moment Derivation of the antenna coil shape that reproduces the current distribution as shown in FIG. 8 will be described. Since the derived current distribution of each current loop takes a discrete value between each loop, it is difficult to directly reproduce a given current distribution with one coil. Therefore, paying attention to the fact that the basic principle of the magnetic resonance coupling type is through the magnetic field generated by the current similarly to the electromagnetic induction type, the resonator is designed so as to reproduce the magnetic field generated by each loop.

The magnetic field at position r can be represented using a magnetic dipole moment. The magnetic dipole moment m is a vector d having a magnitude of the distance d between the magnetic poles, assuming a pair of positive and negative magnetic poles in which the magnitude of the magnetic charge is equal to q _m and the positive magnetic charge is turned from the negative magnetic charge. And is defined as equation (4).

The magnetic field B (r) is represented by Expression (5) using the magnetic dipole moment.

In practice, no single magnetic pole has been found, and the magnetic dipole moment m is defined by equation (6) by the current density J _e (r) at position r, and is equivalent to equation (4). It is.

  Here, r 'is a position vector from the origin to a region where the current exists, and V is a region where the current is distributed. From the above, the magnetic dipole moment generated by each loop of the conductor mesh is obtained and reproduced to generate a magnetic field due to the current distribution. The magnetic dipole moment of the region surrounded by the current loop is defined by Expression (6). To reproduce each magnetic dipole moment, the magnetic field of each minute region when the loop is further divided into smaller loops Consider the dipole moment.

FIG. 9 is a diagram illustrating a magnetic dipole moment in a minute region. It is assumed that the surface surrounded by the closed loop C in which the conductive current I flows is divided into an infinite number of small areas having an area ΔS _i , and the same current I is virtually supplied to all the small loops. Each minute loop can be replaced by a magnetic dipole with a moment _{m i} = _{μ 0 IΔS i.}

  The currents adjacent to each other in the micro loop cancel each other out, and only the current flowing in the closed loop C remains as a whole, and the sum of the magnetic dipole moments of each micro loop matches the magnetic dipole moment in the original loop C.

In other words, loops with the same magnitude of magnetic dipole moment can be reproduced collectively by passing an appropriate current through a single loop surrounding them, and the current at that time is the area of the region Equation (7) can be obtained from the relationship between the magnitude of the magnetic dipole moment of each loop and the magnitude.

Here, S _i is the area of the region surrounding loop, m _i is the magnetic dipole moment magnitude to produce a loop in the enclosed area, mu ₀ denotes the permeability of vacuum.

  FIG. 10 is a diagram showing an example of the distribution of the magnetic dipole moment derived from the current distribution in FIG.

6. Derivation of Antenna Coil Shape In the present embodiment, the contour of the resonator is presented by using the contour of the distribution of the magnetic dipole moment. Note that contour lines are drawn for each | m _max | / n _{number of turns} . m _max is the maximum value of the magnetic dipole moment.

  FIGS. 11A and 11B are diagrams showing an example of the shape of the antenna coil obtained from the distribution of the magnetic dipole moment in FIG. Here, regarding the current distribution of the 100 × 100 conductor mesh, the outline of the antenna coil is determined using the contour line of the magnetic dipole moment, and the number of turns of the coil is set to five. As shown in FIG. 10A, the shape of the antenna coil may be defined so as to follow the contour line. However, since it is assumed that the current flowing through all the antenna coils is the same, the antenna coil is finally formed. Can be designed as a single conductor as shown in FIG.

7. Verification of Transmission Efficiency The shape of the antenna coil derived by this design method is compared with the transmission efficiency of a coil shape that is generally considered to have high efficiency. Here, the transmission efficiency is compared by the kQ product.

A kQ product is obtained from an equivalent circuit of the power transmitting device 10 and the power receiving device 20, and improvement in efficiency is confirmed by the kQ product. FIGS. 12A and 12B are equivalent circuit diagrams of the wireless power supply system 2 including a plurality of transmission-side resonators. When obtaining the kQ product, as shown in FIG. 12B, a number of resonators on the transmission side are combined as one resonator. Assuming that the ratio of the current in each micro loop is defined as S _i = I _i / I _max , the resistance component r _total when _integrated into one is based on the loss component generated by the resonator array, and is expressed by equation (8). expressed.
Here, i and j are loops sharing a side. Voltage generated in the power receiving side of the coil, since the equation (9) using a resonance angular frequency ω of the _resonator, can be defined as _{M total} = _{MΣ i} S _i.

As a result, the kQ product of the power-transmitting-side resonator and the power-receiving-side resonator combined into one is given by Expression (10).

  In order to confirm the usefulness of this method, a comparison was made between an antenna coil in which windings are concentrated on the outside, which is generally considered to be highly efficient, and an antenna coil presented by this method. As a result of taking the ratio of the kQ product when the number of windings is fixed to 5 and the number of conductor meshes is changed to 10 × 10, 50 × 50, and 100 × 100, they are 1.24, 1.93, and 3.30, respectively. Also, the kQ product was higher in the resonator according to the present design method than in the above. Also, it can be seen that as the number of meshes increases, the amount of improvement in the kQ product increases.

  Further, even when the number of meshes is fixed to 100 × 100 and the number of windings is changed to 3, 5, and 7, respectively, they are 5.18, 3.30, and 2.51, respectively. And the efficiency was improved.

8. Distributed Reactance Compensation The physical shape of an antenna is limited by frequency. This recognition is the same not only for the radiation type antenna but also for the resonator used in the magnetic resonance coupling type wireless power transmission. When the operating frequency is high and the resonator shape is large, the current distribution is biased, and sometimes the direction of the current is reversed inside the resonator. At this time, the magnetic field cancels out in the resonator, which leads directly to a decrease in the Q value, which is a performance index of the resonator.

  One of the reasons why the parameter adjustment of the resonator design is still performed by trial and error is the non-uniformity of the current distribution. Also in this design method, it is assumed that the current flowing through one conductor (antenna coil) is uniform. Therefore, the applicable resonator shape may be limited to a range that can be regarded as electrically small. If the non-uniformity of the current distribution can be improved, the applicable range of the present design method will be greatly expanded.

  The non-uniformity of the current distribution is caused by the fact that when a conductor as an antenna coil is used as a kind of transmission line, electric charges are stored in a parasitic capacitor existing between the lines. FIGS. 13A and 13B are equivalent circuit diagrams of lossy and lossless transmission circuits. Let L be the inductance per unit length, C be the capacitance per unit length, and R and G be the loss components associated with L and C, respectively. When R and G can be considered to be sufficiently small, a lossless line is obtained as shown in FIG.

  The present inventors have proposed distributed reactance compensation (DRC) (Non-Patent Document 3). This distributed reactance compensation was originally proposed as a technique for reducing the magnetic field strength around the antenna coil and reducing the dielectric loss by inserting a reactance compensation capacitor not only in the input port but also in the antenna coil.

  Unlike conventional Concentrated Reacquantance Compensation (CRC), distributed reactance compensation realizes a resonance state in the coil by inserting a large number of capacitances in the antenna coil and reduces the voltage difference between lines. As a result, the charge stored in the parasitic capacitor is reduced.

  The present inventors have found that the distributed reactance compensation not only reduces the dielectric loss but also makes the current distribution of the antenna coil uniform. This phenomenon should not be regarded as a known technique.

  FIG. 14 is an equivalent circuit diagram illustrating the dispersion reactance compensation. The antenna coil is divided into a plurality of elements, and a compensating capacitor is inserted between them.

  FIGS. 15A to 15D are diagrams showing simulation results of current distribution by dispersion reactance compensation. The current distribution of the loop coil having a diameter of 5 m at an operating frequency of 6.78 MHz was calculated by an electromagnetic field simulation. The wire diameter of the loop coil was 1 mm, the material was copper, and FEKO manufactured by Altair Inc. was used for the simulator.

  FIG. 15A shows a case without compensation. No distributed reactance compensation is performed, and a reactance compensation capacitor is connected to the input port. The current distribution is normalized by the maximum value in the resonator. The current intensity is minimum at the input port portion, and is about 30% of the current intensity as compared with the portion farthest from the port. This result indicates that at an operating frequency of 6.78 MHz, even with an antenna having one turn, the current distribution can be greatly distorted, and the distortion of the current distribution further increases as the number of turns increases.

  FIGS. 15B to 15D show simulation results when dispersion reactance compensation is performed. The number N of capacitors used for dispersion reactance compensation is three, two, four, and eight. When the current distribution is uniform, the intensity and direction of the current in each capacitor and the current flowing through the input port are equal. On the basis of this condition, the capacitance at which the current distribution becomes uniform is considered.

For the derivation of the capacitance, an electromagnetic field simulation is performed using the portion where the capacitor is connected as a port, and the S parameter derived as a circuit network of the (N + 1) port is used. With Z parameter obtained by converting the S parameters, conditions the current of each port is equal I _{includes, -Z c I = Z '(} 1,1,1, ..., 1) can be described by ^T I. Here, Z ′ is an N × (N + 1) matrix in which a row corresponding to an input port is deleted from the impedance matrix Z, and _Zc is an N-th order row vector including the impedance of the connected capacitor. The capacitance to be connected is obtained by solving equation (11).
_{Z c = -Z '(1,1,1,} ..., 1) T (11)

  Capacitors were arranged at equal intervals in a loop coil having a diameter of 5 m, and the capacitance was calculated by equation (11). The value did not depend on the position due to symmetry. When N = 2, 56.7 pF and when N = 4 It was 179.4 pF when 98.1 pF and N = 8.

  In FIGS. 15B to 15D, it should be noted that the correspondence between the shading and the current value is different in each figure in order to show the current distribution in more detail. It can be confirmed that the current distributions at the input port and the capacitor insertion portion are equal, and the current distribution becomes more uniform as the number of capacitors used for the distributed reactance compensation increases. When N = 8, the difference in current intensity in the resonator is suppressed to 1% or less.

  The factor that makes the current distribution uniform by the dispersion reactance compensation can be considered by the transmission line theory. When the conductor in the resonator is viewed as a kind of transmission line, the current distribution becomes non-uniform because charges are stored in a parasitic capacitor existing between the lines. It is considered that the distributed reactance compensation reduces the potential difference between the lines by arranging a large number of capacitances in the resonator, so that the charge stored in the parasitic capacitor is reduced and the current distribution is made uniform.

  Thus, by performing the distributed reactance compensation, the current distribution of the antenna coil can be made uniform. Therefore, it is preferable to incorporate dispersion reactance compensation into a software program using this design method when designing an antenna used in a frequency band of several MHz or more.

  The shape of the antenna coil is determined by the above-described design method. Then, additional processing related to dispersion compensation is performed on the antenna coil. The number N of capacitors to be inserted may be specified manually, or the non-uniformity of the current without compensation is calculated as shown in FIG. The number N may be determined.

  Then, if the shape of the antenna coil is known, the S parameter (Z parameter) when it is regarded as a transmission line can be calculated. When N is determined, the capacitance value of the capacitor to be inserted can be calculated by solving equation (11) using the Z parameter.

  Although the present invention has been described using specific terms based on the embodiments, the embodiments are merely illustrative of the principles and applications of the present invention, and the embodiments are defined in the appended claims. Many modifications and changes in arrangement may be made without departing from the spirit of the present invention.

2 Wireless power supply system 10 Power transmission device 20 Power reception device L _TX power transmission coil C _TX resonance capacitor L _RX power reception coil C _RX resonance capacitor

Claims (8)

An antenna coil design method,
Modeling the surrounding environment including the paired antenna coils;
Dividing the space in which the antenna to be designed is to be placed into a plurality of meshes, and calculating an optimal current amount of each mesh in the modeled surrounding environment;
Acquiring the shape of the antenna coil to be designed from the current distribution of the plurality of meshes,
A method for designing an antenna coil, comprising: Calculating a magnetic dipole moment distribution from the plurality of mesh current distributions;
Obtaining contours of the distribution of the magnetic dipole moment;
The design method according to claim 1, further comprising: a shape of the antenna coil to be designed according to the contour line. An antenna coil design method,
Modeling the surrounding environment including the paired antenna coils;
Dividing the space in which the antenna to be designed is to be placed into a plurality of meshes, and calculating an optimal magnetic dipole moment of each mesh in the modeled surrounding environment;
Acquiring contours of the distribution of magnetic dipole moments of the plurality of meshes,
Obtaining the shape of the antenna coil based on the contour line,
A method for designing an antenna coil, comprising:   4. The antenna coil designing method according to claim 1, further comprising a step of dividing the antenna to be designed into a plurality of elements, and inserting a capacitor between the divided elements. . On the computer,
Modeling the surrounding environment including the paired antenna coils;
Dividing the space in which the antenna to be designed is to be arranged into a plurality of meshes, and calculating an optimal current amount of each mesh in the modeled surrounding environment;
Obtaining the shape of the antenna coil from the current distribution of the plurality of meshes,
A program for executing On the computer,
Calculating a magnetic dipole moment distribution from the plurality of mesh current distributions;
Obtaining contours of the distribution of the magnetic dipole moment;
Acquiring the shape of the antenna coil to be designed according to the contour line,
The program according to claim 5, wherein the program is further executed. On the computer,
Modeling the surrounding environment including the paired antenna coils;
Dividing the space in which the antenna to be designed is to be placed into a plurality of meshes, and calculating an optimal magnetic dipole moment of each mesh in the modeled surrounding environment;
Acquiring contours of the distribution of magnetic dipole moments of the plurality of meshes,
Obtaining a shape of the antenna coil to be designed based on the contour line,
A program characterized by executing On the computer,
The program according to any one of claims 5 to 7, further comprising a step of dividing the antenna to be designed into a plurality of elements and inserting a capacitor between the divided elements.