JP2019198206A - Control arrangement of dynamo-electric machine - Google Patents

Abstract

To provide a control arrangement of a dynamo-electric machine capable of restraining reduction of controllability of the control amount of the dynamo-electric machine.SOLUTION: A control arrangement includes a field control section for controlling a field current, flowing through a field winding, to a field command current If*, a synchronous rectification control section, a sine wave control section, and a transition control section. When switching is made from control of the sine wave control section to control of the synchronous rectification control section, the transition control section controls the field control section so that the field current, immediately after switching to control of the synchronous rectification control section, becomes larger than the field current immediately before switching to control of the synchronous rectification control section, and when switching is made from control of the synchronous rectification control section to control of the sine wave control section, controls the field control section so that the field current, immediately after switching to control of the sine wave control section, becomes smaller than the field current immediately before switching to control of the sine wave control section.SELECTED DRAWING: Figure 12

Description

  The present invention relates to a control device for a rotating electrical machine.

  2. Description of the Related Art Conventionally, as described in Patent Document 1, there is known a control device that controls a current flowing in a stator winding of a rotating electrical machine and controls a field current flowing in a field winding of the rotating electrical machine. The control device controls the field current when the rotation speed of the rotating electrical machine is equal to or less than a predetermined value, and causes the rotating electrical machine to generate power by flowing a current controlled by PWM so that the modulation factor becomes 1 or less through the stator winding. Implement the PWM control mode. When the rotational speed of the rotating electrical machine exceeds a predetermined value, the control device implements a field control mode in which the rotating electrical machine generates power by controlling the field current. When switching from one control mode to the other control mode of the PWM control mode and the field control mode, the control device performs switching via the overmodulation PWM control mode in which the modulation factor is greater than 1. Thereby, suppression of the fluctuation | variation of the controlled variable of the rotary electric machine which occurs with switching is aimed at.

Japanese Patent Laid-Open No. 2006-189698

  Here, the responsiveness of the control amount of the rotating electrical machine may be reduced. When the responsiveness decreases, the time during which the control amount deviates from the command control amount when the control mode is switched becomes longer, and the controllability of the control amount may be reduced.

  The main object of the present invention is to provide a control device for a rotating electrical machine that can suppress a decrease in controllability of the control amount of the rotating electrical machine.

  The present invention includes a rotating electrical machine having a field winding and a stator winding, and a series connection body of an upper arm switch and a lower arm switch. By switching the upper arm switch and the lower arm switch, a DC power source and the In a control device for a rotating electrical machine applied to a control system including an inverter that transmits power to and from a stator winding, a field current flowing through the field winding is changed to a field command current that is a command value thereof. In the case where the field control unit to be controlled and the main control amount are any one of the generated torque of the rotating electrical machine, the direct current supplied from the inverter to the DC power source, or the generated power of the rotating electrical machine, On the condition that the upper arm switch is turned on in at least a part of the period in which the generated voltage exceeds the voltage of the DC power supply, the main control variable Based on the command value for controlling the command control amount, the upper arm switch and the lower arm switch are turned on once each while sandwiching a dead time in one electrical angle cycle of the rotating electrical machine, thereby the stator winding. When the peak value of each phase voltage applied to the stator winding is equal to or less than the voltage of the DC power source, the main control amount is controlled to the command control amount. On the basis of the command value for turning on and off the upper arm switch and the lower arm switch by sine wave PWM control so that each phase voltage applied to the stator winding has a PWM voltage waveform, the stator winding A sine wave control unit that controls the current flowing in the wire, and the current flowing in the stator winding is controlled by the control of the synchronous rectification control unit A control system that controls as a controlled variable, a control system that controls the current flowing through the stator winding as a controlled variable by the control of the sine wave controller, and a control system that controls the field current as a controlled variable by the control of the field controller The control unit corresponding to the control system having the lowest control amount responsiveness is the field control unit, and when switching from the control of the sine wave control unit to the control of the synchronous rectification control unit, the synchronous rectification control unit Control the field control unit so that the field current immediately after switching to the control becomes larger than the field current immediately before switching to the control of the synchronous rectification control unit, and control of the synchronous rectification control unit Is switched to control of the sine wave control unit, the field current immediately after switching to control of the sine wave control unit is switched to control of the sine wave control unit. A transition control unit that performs a transition control process for controlling the field control unit so as to be smaller than the field current immediately before the start.

  When the rotating electrical machine is driven as a generator, if the field current is made continuous before and after switching from the control of the sine wave control unit to the control of the synchronous rectification control unit, the torque of the rotating electrical machine decreases before and after the control switching. Resulting in. This is because the loss generated in the current path including the stator winding and the inverter is smaller in the control of the synchronous rectification control unit than in the control of the sine wave control unit, and the stator winding is used to cover the required generated power. This is because the flowing current decreases before and after the control switching. The reason why the loss generated by the control of the synchronous rectification control unit is smaller than the loss generated by the control of the sine wave control unit is, for example, that the switching loss generated by the control of the synchronous rectification control unit is This is because it is smaller than the switching loss generated by the control.

  In view of this point, in the present invention, when switching from the control of the sine wave control unit to the control of the synchronous rectification control unit, the field current immediately after switching to the control of the synchronous rectification control unit is controlled by the control of the synchronous rectification control unit. The field control unit is controlled so as to be larger than the field current immediately before switching to. Thereby, the fall of the torque in the case of switching from the control of the sine wave control unit to the control of the synchronous rectification control unit can be suppressed. As a result, it is possible to suppress a deviation between the main control amount and the command control amount when the control is switched, and it is possible to suppress a decrease in controllability of the control amount.

  On the other hand, when the rotating electrical machine is driven as a generator, if the field current is made continuous before and after switching from the control of the synchronous rectification control unit to the control of the sine wave control unit, the torque of the rotating electrical machine before and after the control switching. Will increase. This is because the loss generated in the current path including the stator winding and the inverter is larger in the control of the sine wave control unit than in the control of the synchronous rectification control unit, and the stator winding is used to cover the required generated power. This is because the flowing current increases before and after the control switching.

  In view of this point, in the present invention, when switching from the control of the synchronous rectification control unit to the control of the sine wave control unit, the field current immediately after switching to the control of the sine wave control unit is controlled by the control of the sine wave control unit. The field control unit is controlled so as to be smaller than the field current immediately before switching to. Thereby, an increase in torque when switching from the control of the synchronous rectification control unit to the control of the sine wave control unit can be suppressed. As a result, it is possible to suppress a deviation between the main control amount and the command control amount when the control is switched, and it is possible to suppress a decrease in controllability of the control amount.

1 is an overall configuration diagram of an in-vehicle system according to a first embodiment. The figure which shows the drive aspect of a field electricity supply circuit. The block diagram of sine wave PWM control. The figure which shows the definition of a voltage vector. The time chart which shows transition of the drive mode of a switch, phase current, etc. at the time of sine wave PWM control. The block diagram of synchronous rectification control. The time chart which shows transition of the drive mode of a switch, phase current, etc. at the time of synchronous rectification control. The figure which shows the closed loop transfer function of the control system which controls a field current, and the control system which controls d-axis current. The figure which shows the frequency characteristic regarding the gain of a closed loop transfer function. The figure which shows the definition of a time constant. The flowchart which shows the procedure of a transfer control process. The figure which shows the setting aspect of the field command current in the case of switching to idle up control. The figure which shows the setting aspect of the field command current in the case of switching to normal time control. The figure which shows the setting aspect of the field command current in the case of switching to the idle-up control which concerns on a comparative example. The figure which shows the setting aspect of the field command current in the case of switching to the normal time control which concerns on a comparative example. The time chart which shows transition of the drive mode of a switch, phase current, etc. at the time of overmodulation PWM control concerning a 2nd embodiment. The time chart which shows transition of the drive mode of a switch, phase current, etc. at the time of rectangular wave control. The flowchart which shows the procedure of a transfer control process. The flowchart which shows the procedure of a 1st process. The flowchart which shows the procedure of a 2nd process. The figure which shows the voltage vector at the time of sine wave PWM control. The figure which shows the voltage vector at the time of overmodulation PWM control. The figure which shows the voltage vector at the time of rectangular wave control. The figure which shows the voltage vector at the time of synchronous rectification control. The flowchart which shows the procedure of the transfer control process which concerns on 3rd Embodiment. The block diagram of the sine wave PWM control which concerns on 4th Embodiment. The block diagram of synchronous rectification control. The flowchart which shows the procedure of a transfer control process. The figure which shows the setting aspect of the field command current which concerns on the modification of 4th Embodiment. The figure which shows the setting aspect of the field command current which concerns on the modification of 4th Embodiment. The figure which shows the setting aspect of the field command current which concerns on 5th Embodiment. The figure which shows the setting aspect of a field command current. The figure which shows the setting aspect of the field command current which concerns on 6th Embodiment. The figure which shows the setting aspect of a field command current.

<First Embodiment>
Hereinafter, a first embodiment in which a control device according to the present invention is mounted on a vehicle will be described with reference to the drawings.

  As shown in FIG. 1, the vehicle includes an engine 10 as an in-vehicle main machine. The engine 10 includes a fuel injection valve and the like, and generates power by combustion of fuel such as gasoline or light oil injected from the fuel injection valve. The generated power is output from the output shaft 10a of the engine 10.

  The vehicle includes a battery 20 as a direct current power source, a load 22, and a control system. The battery 20 is, for example, a lead storage battery having a rated voltage of 12V. The control system includes a rotating electrical machine 30 that is AC driven. In the present embodiment, a wound field type synchronous machine is used as the rotating electrical machine 30. In the present embodiment, an ISG (Integrated Starter Generator) that is a generator to which a function of an electric motor is added is used as the rotating electrical machine 30. The rotating electrical machine 30 is a salient pole machine, for example.

  The rotating electrical machine 30 includes a rotor 31. The rotor 31 includes a field winding 32. The rotation shaft of the rotor 31 can transmit power to the output shaft 10a of the engine 10 through a pulley or the like (not shown). When the rotating electrical machine 30 is driven as a generator, the rotor 31 is rotated by the rotational power supplied from the output shaft 10a, and the rotating electrical machine 30 generates power. The battery 20 is charged by the electric power generated by the rotating electrical machine 30. On the other hand, when the rotating electrical machine 30 is driven as an electric motor, the output shaft 10a rotates as the rotor 31 rotates, and a rotational force is applied to the output shaft 10a. Thereby, for example, traveling of the vehicle can be assisted. In addition, the drive wheel of the vehicle is connected to the output shaft 10a via a transmission or the like.

  The rotating electrical machine 30 includes a stator 33. The stator 33 includes a stator winding. The stator winding includes U, V, and W phase windings 34U, 34V, and 34W arranged in a state of being shifted from each other by 120 degrees in electrical angle.

  The control system includes a three-phase inverter 40, a field energization circuit 41, and a capacitor 21. The inverter 40 includes a series connection body of U, V, W phase upper arm switches SUp, SVp, SWp and U, V, W phase lower arm switches SUn, SVn, SWn. At the connection point between the U, V, W phase upper arm switches SUp, SVp, SWp and the U, V, W phase lower arm switches SUn, SVn, SWn, U, V, W phase windings 34U, 34V, 34W The 1st end of is connected. The second ends of the U, V, and W phase windings 34U, 34V, and 34W are connected at a neutral point. In other words, in this embodiment, the U, V, and W phase windings 34U, 34V, and 34W are star-connected. In the present embodiment, N-channel MOSFETs are used as the arm switches SUp to SWn. Each switch SUp to SWn includes body diodes DUp to DWn. When the N-channel MOSFET is turned on, current flow between the drain that is the high potential side terminal and the source that is the low potential side terminal is permitted. On the other hand, when the N-channel MOSFET is turned off, current flow from the drain to the source is blocked.

  The positive terminal of the battery 20 is connected to the collector, which is the high potential side terminal of the U, V, W phase upper arm switches SUp, SVp, SWp, via the high potential side electrical path Lp. The negative electrode terminal of the battery 20 is connected to the emitter which is the low potential side terminal of the U, V, W phase lower arm switches SUn, SVn, SWn via the low potential side electrical path Ln. Each electric path Lp, Ln is a conductive member such as a bus bar. The high potential side terminal of the capacitor 21 is connected to the positive terminal side of the battery 20 from the connection point with the collectors of the upper arm switches SUp, SVp, SWp in the high potential side electric path Lp. The low potential side terminal of the capacitor 21 is connected to the negative terminal side of the battery 20 from the connection point with the emitters of the lower arm switches SUn, SVn, SWn in the low potential side electrical path Ln.

  The field energization circuit 41 is a full bridge circuit, and includes a series connection body of the first upper arm switch SH1 and the first lower arm switch SL1, and a series connection body of the second upper arm switch SH2 and the second lower arm switch SL2. It has. A first end of the field winding 32 is connected to a connection point between the first upper arm switch SH1 and the first lower arm switch SL1 via a brush (not shown). A second end of the field winding 32 is connected to a connection point between the second upper arm switch SH2 and the second lower arm switch SL2 via a brush (not shown). In the present embodiment, IGBTs are used as the arm switches SH1, SL1, SH2, and SL2. Also, the respective diodes DH1, DL1, DH2, DL2 are connected in reverse parallel to the arm switches SH1, SL1, SH2, SL2.

  The collectors that are the high potential side terminals of the first and second upper arm switches SH1 and SH2 are connected to the inverter 40 side from the connection point of the high potential side electrical path Lp with the high potential side terminal of the capacitor 21. . The inverter which is the low potential side terminal of the first and second lower arm switches SL1 and SL2 is connected to the inverter 40 side of the low potential side electric path Ln than the connection point with the low potential side terminal of the capacitor 21. .

  The control system includes a voltage detection unit 50, a phase current detection unit 51, a field current detection unit 52, and an angle detection unit 53. Voltage detector 50 detects the terminal voltage of capacitor 21 as power supply voltage VDC. The phase current detector 51 detects the phase current flowing through the U, V, and W phase windings 34U, 34V, and 34W. The field current detection unit 52 detects a field current flowing through the field winding 32. The angle detector 53 outputs an angle signal that is a signal corresponding to the rotation angle of the rotor 31. Output signals of the detection units 50 to 53 are input to the control device 60. Incidentally, in the present embodiment, the rotating electrical machine 30, the inverter 40, the field energizing circuit 41, and the control device 60 are integrated to form an electro-mechanical integrated drive device DU.

  The control device 60 includes a memory 60a corresponding to a storage unit. The memory 60a corresponds to a non-transitory computer readable medium, and is, for example, a nonvolatile memory.

  Note that some or all of the functions of the control device 60 may be configured in hardware by, for example, one or a plurality of integrated circuits. Moreover, each function of the control apparatus 60 may be comprised by the software recorded on the non-transitional tangible recording medium and the computer which performs it, for example.

  The control device 60 generates drive signals for the switches constituting the inverter 40 and the field energization circuit 41.

  First, the inverter 40 will be described. The control device 60 acquires the angle signal of the angle detection unit 53 and generates a drive signal for turning on / off each of the switches SUp to SWn configuring the inverter 40 based on the acquired angle signal. The drive signal takes either an on command for instructing switching on of the switch or an off command for instructing switching on off. When driving the rotating electrical machine 30 as an electric motor, the control device 60 converts each DC power output from the battery 20 into AC power and supplies it to the U, V, W phase windings 34U, 34V, 34W. A drive signal for turning on and off the switches SUp to SWn is generated, and the generated drive signal is supplied to the gates of the arm switches SUp to SWn. On the other hand, when driving the rotating electrical machine 30 as a generator, the control device 60 converts the AC power output from the U, V, and W phase windings 34U, 34V, and 34W into DC power to convert the battery 20 and the load 22 A drive signal for turning on / off each of the arm switches SUp to SWn is generated so as to be supplied to at least one of them.

  Next, the field energizing circuit 41 will be described. The control device 60 turns on and off each switch constituting the field energizing circuit 41 in order to excite the field winding 32. FIG. 2 shows an example of the driving mode of each switch. The control device 60 turns each switch on and off so that the first state shown in FIG. 2A and the second state shown in FIG. 2B appear alternately. The first state is a state in which the first upper arm switch SH1 and the second lower arm switch SL2 are turned on, and the second upper arm switch SH2 and the first lower arm switch SL1 are turned off. The second state is a state in which the first upper arm switch SH1 and the second lower arm switch SL2 are turned off, and the second upper arm switch SH2 and the first lower arm switch SL1 are turned on. The driving mode shown in FIG. 2 is merely an example, and other driving modes such as a driving mode in which the second state does not appear may be used. The field energizing circuit 42 is not limited to a full bridge circuit, and may be a half bridge circuit, for example.

  The control device 60 calculates the electrical angle θe of the rotating electrical machine 30 and the rotational speed Nm of the rotor 31 based on the angle signal from the angle detection unit 53.

  The vehicle includes an engine ECU 11 that is a control device that performs combustion control of the engine 10 and a host ECU 12 that is a control device that controls the vehicle. The control device 60, the engine ECU 11 and the host ECU 12 can exchange information via a communication line such as CAN.

  The engine ECU 11 performs normal time control and idle-up control as combustion control during idling operation of the engine 10. The normal control is combustion control for controlling the engine rotation speed Ner, which is the rotation speed of the output shaft 10a, to the first command rotation speed Netgt1. The idle-up control is combustion control for controlling the engine rotation speed Ner to a second command rotation speed Netgt2 that is higher than the first command rotation speed Netgt1. Each command rotational speed Netgt1, Netgt2 is variably set according to the temperature of the cooling water of the engine 10 and the like. When the engine ECU 11 determines that the predetermined condition is satisfied, the engine ECU 11 switches from the normal time control to the idle up control. The predetermined condition is, for example, a condition that the power consumption of the in-vehicle device driven by the power of the output shaft 10a is equal to or higher than the predetermined power. The in-vehicle device in this case also includes the rotating electrical machine 30.

  Hereinafter, this embodiment demonstrates the case where the rotary electric machine 30 is driven as a generator. FIG. 3 shows a block diagram of the sine wave PWM control mode performed by the control device 60. In this embodiment, the structure which performs the process shown in FIG. 3 among the control apparatuses 60 corresponds to a sine wave control part.

  The voltage deviation calculation unit 61 calculates the voltage deviation ΔV by subtracting the power supply voltage VDC detected by the voltage detection unit 50 from the command generated voltage VD *. The command power generation voltage VD * is a command value of a DC voltage output from the inverter 40 to the battery 20. The command power generation voltage VD * is input from the host ECU 12 to the control device 60, for example.

  The torque calculation unit 62 calculates a command value for the control amount of the rotating electrical machine 30 as an operation amount for performing feedback control of the voltage deviation ΔV to zero. In the present embodiment, the control amount is torque, and the command value is the command torque Trq *. In the present embodiment, the feedback control used in the torque calculation unit 62 is proportional-integral control. The feedback control is not limited to proportional integral control, and may be proportional integral differential control, for example.

  The d-axis command setting unit 71 sets a d-axis command current Id * for setting the torque of the rotating electrical machine 30 to the command torque Trq * based on the command torque Trq *. Specifically, the d-axis command setting unit 71 sets the d-axis command current Id * based on map information in which the command torque Trq * and the d-axis command current Id * are related. Based on the command torque Trq *, the q-axis command setting unit 72 sets a q-axis command current Iq * for setting the torque of the rotating electrical machine 30 to the command torque Trq *. Specifically, the q-axis command setting unit 72 sets the q-axis command current Iq * based on map information in which the command torque Trq * and the q-axis command current Iq * are related. The map information is stored in the memory 60a.

  Based on the phase current detected by the phase current detector 51 and the electrical angle θe, the two-phase converter 70 converts the U, V, and W phase currents IU, IV, and IW in the three-phase fixed coordinate system of the rotating electrical machine 30 to Conversion into d and q axis currents Idr and Iqr in the dq coordinate system, which is a two-phase rotating coordinate system.

  The stator control unit 73 calculates a d-axis command voltage Vd * as an operation amount for performing feedback control of the d-axis current Idr to the d-axis command current Id *. Specifically, the stator control unit 73 calculates the d-axis current deviation ΔId as a value obtained by subtracting the d-axis current Idr from the d-axis command current Id *, and feedback-controls the calculated d-axis current deviation ΔId to 0. As a manipulated variable, a d-axis command voltage Vd * is calculated.

  Stator control unit 73 calculates q-axis command voltage Vq * as an operation amount for feedback control of q-axis current Iqr to q-axis command current Iq *. Specifically, the stator control unit 73 calculates the q-axis current deviation ΔIq as a value obtained by subtracting the q-axis current Iqr from the q-axis command current Iq *, and feedback-controls the calculated q-axis current deviation ΔIq to 0. Q-axis command voltage Vq * is calculated as the operation amount.

  In the present embodiment, the feedback control used in the stator control unit 73 is proportional-integral control. The feedback control is not limited to proportional integral control, and may be proportional integral differential control, for example.

  A command voltage vector which is a command value of a voltage vector in the dq coordinate system is determined by the d-axis command voltage Vd * and the q-axis command voltage Vq *. Here, as shown in FIG. 4, the voltage vector Vtr applied to the stator winding is such that the d-axis component becomes the d-axis voltage Vd and the q-axis component becomes the q-axis voltage Vq. In the present embodiment, the voltage phase δ, which is the phase of the voltage vector Vtr, is defined with the positive direction of the d axis as a reference, and the counterclockwise direction from this reference is defined as the positive direction.

  The three-phase converter 74 converts the d and q-axis command voltages Vd * and Vq * into the U, V, and W in the three-phase fixed coordinate system based on the d and q-axis command voltages Vd * and Vq * and the electrical angle θe. It converts into phase command voltage Vu *, Vv *, Vw *. In the present embodiment, the U, V, and W phase command voltages Vu *, Vv *, and Vw * are sinusoidal waveforms that are 120 degrees out of phase in electrical angle.

  The stator generator 75 drives each switch for turning on and off each switch SUp to SWn of the inverter 40 by sine wave PWM control based on the carrier signal, each phase command voltage Vu *, Vv *, Vw * and the power supply voltage VDC. Generate a signal. Specifically, when the peak value of each phase command voltage Vu *, Vv *, Vw * is equal to or lower than power supply voltage VDC, stator generation unit 75 sets each phase command voltage Vu *, Vv *, Vw * to “VDC / 2. The drive signals of the switches SUp to SWn of the inverter 40 are generated on the basis of the magnitude comparison between the value divided by “and the carrier signal. In the present embodiment, the carrier signal is a triangular wave signal. In the sine wave PWM control, the value obtained by dividing the amplitude of each phase command voltage Vu *, Vv *, Vw * by “VDC / 2” is equal to or smaller than the amplitude of the carrier signal.

  In the present embodiment, the two-phase conversion unit 70, the d and q-axis command setting units 71 and 72, the stator control unit 73, the three-phase conversion unit 74, and the stator generation unit 75 correspond to a sine wave control unit. According to the sine wave control unit, the actual voltage vector is controlled to the command voltage vector determined by the d and q axis command voltages Vd * and Vq *.

  Field command setting unit 80 sets field command current If * based on command torque Trq *, rotation speed Nm, and power supply voltage VDC. Specifically, the field command setting unit 80 is map information in which the command torque Trq *, the rotation speed Nm (corresponding to the rotation speed parameter), the power supply voltage VDC, and the field command current If * are related. The field command current If * is set based on sine wave map information (corresponding to the first information) used in the sine wave PWM control. In the sine wave map information, as shown in FIGS. 12B and 13B, the rotation speed Nm and the field command current are set such that the field command current If * increases as the rotation speed Nm of the rotor 31 increases. If * is related. The sine wave map information is stored in the memory 60a.

  The field current control unit 81 calculates the field command voltage Vf * as an operation amount for performing feedback control of the field current Ifr detected by the field current detection unit 52 to the field command current If *. Specifically, the field current control unit 81 calculates the field current deviation ΔIf as a value obtained by subtracting the field current Ifr from the field command current If *, and feedback-controls the calculated field current deviation ΔIf to 0. As a manipulated variable, field command voltage Vf * is calculated. In the present embodiment, the feedback control used in the field current control unit 81 is proportional-integral control. The feedback control is not limited to proportional integral control, and may be proportional integral differential control, for example.

  The field generation unit 82 controls each switch SH1 of the field energization circuit 41 for controlling the applied voltage of the field winding 32 to the field command voltage Vf * based on the field command voltage Vf * and the power supply voltage VDC. Each drive signal of -SL2 is generated.

  In the present embodiment, the field command setting unit 80, the field current control unit 81, and the field generation unit 82 correspond to the field control unit.

  FIG. 5 shows the transition of each waveform for one phase when the sine wave PWM control is executed. FIGS. 5A and 5B show transitions of the drive signals of the upper and lower arm switches of the inverter 40, and FIG. 5C shows transitions of the phase current and the phase voltage.

  Next, FIG. 6 shows a block diagram of synchronous rectification control performed by the control device 60. In the present embodiment, the configuration for performing the processing shown in FIG. 6 in the control device 60 corresponds to the synchronous rectification control unit.

  In the synchronous rectification control, the peak value of each phase command voltage Vu *, Vv *, Vw * exceeds the power supply voltage VDC, and the peak value of the power generation voltage of each phase winding 34U to 34W exceeds the power supply voltage VDC. To be implemented. In the synchronous rectification control, the switch connected in reverse parallel to the diode in which the current is to flow is turned on during the period in which current is flowing in the body diode connected in reverse parallel to the switch of the inverter 40. In each phase, the period in which the current tends to flow through the body diode is a period in which the peak value of the generated voltage in each phase exceeds the power supply voltage VDC. In the synchronous rectification control, in each phase, the upper arm switch is turned on once in at least a part of a period in which the generated voltage exceeds the power supply voltage VDC in one electrical angle cycle. Thereby, the alternating current output from each phase winding 34U-34W is converted into a direct current.

  Based on the electrical angle θe, the dead time DT of the upper and lower arm switches of the inverter 40, and the command voltage phase δ *, the synchronization generator 93 generates each drive signal for turning on and off the switches SUp to SWn of the inverter 40. Generate. The command voltage phase δ * is a command value of the voltage phase δ for setting the torque of the rotating electrical machine 30 to the command torque Trq *. The drive signal generated by the synchronization generator 93 is a signal that turns on each of the upper arm switch and the lower arm switch once in one electrical angle cycle of each phase. This drive signal is shifted by 120 ° in electrical angle in each phase.

  Field command setting unit 80 sets field command current If * based on command torque Trq *, rotation speed Nm, and power supply voltage VDC. Specifically, the field command setting unit 80 is map information in which the command torque Trq *, the rotation speed Nm, the power supply voltage VDC, and the field command current If * are related, and is used in the synchronous rectification control. The field command current If * is set based on the synchronous rectification map information (corresponding to the second information). In the synchronous rectification map information, as shown in FIGS. 12B and 13B, the rotation speed Nm and the field command current are set such that the field command current If * increases as the rotation speed Nm of the rotor 31 increases. If * is related. The synchronous rectification map information is stored in the memory 60a.

  In FIG. 6, the voltage deviation calculation unit 61, the torque calculation unit 62, the field current control unit 81, and the field generation unit 82 have the same configuration as that shown in FIG.

  FIG. 7 shows the transition of each waveform for one phase when synchronous rectification control is executed. FIGS. 7A to 7C correspond to FIGS. 5A to 5C. FIG. 7 shows the dead time set in the synchronous rectification control as DT1.

  Subsequently, the response of the control system will be described. A control system that controls the d and q axis currents Idr and Iqr to d and q axis command currents Id * and Iq * is referred to as a sine wave current control system, and a control system that controls the field current Ifr to the field command current If *. Is referred to as a field current control system. In the present embodiment, the responsiveness of the d and q axis currents Idr and Iqr by the sine wave current control system to the change in the d and q axis command currents Id * and Iq * is the field current to the change in the field command current If *. It is higher than the response of the field current Ifr by the control system. The responsiveness of the sinusoidal current control system is higher than the responsiveness of the field current control system, for example, even though the inductance of the field winding 32 is larger than the inductance of the stator winding, This is because the voltage that can be applied to the magnetic winding 32 is not sufficient.

  Incidentally, the level of responsiveness of the control system can be determined by, for example, the following (A) to (C).

(A) Closed Loop Transfer Function The closed loop transfer function Gcf (s) having the field command current If * as an input and the field current Ifr as an output will be described with reference to FIG. The first transfer function 101 represented by G1 (s) receives the field current deviation ΔIf as an input, and the applied voltage Vf of the field winding 32, which is an operation amount for performing feedback control of the field current deviation ΔIf to zero. Is output. The first transfer function 101 corresponds to the field current control unit 81.

  The second transfer function 102 represented by G2 (s) modeling the frequency characteristics of the field winding 32 corresponds to a plant. The second transfer function 102 receives the applied voltage Vf of the field winding 32 and outputs a field current Ifr.

  The closed loop transfer function Gcf (s) of the field current control system is expressed by the following equation (eq1).

続いて、図８（ｂ）を用いて、ｄｑ軸間の非干渉化がなされている場合において、ｄ軸指令電流Ｉｄ＊を入力とし、ｄ軸電流Ｉｄｒを出力とする閉ループ伝達関数Ｇｃｄ（ｓ）について説明する。Ｇ３（ｓ）で表される第３伝達関数２０１は、ｄ軸電流偏差ΔＩｄを入力とし、ｄ軸電流偏差ΔＩｄを０にフィードバック制御するための操作量であるｄ軸電圧Ｖｄを出力する。第３伝達関数２０１がステータ制御部７３に相当する。

Subsequently, with reference to FIG. 8B, when the dq axes are made non-interfering, a closed-loop transfer function Gcd (s) having the d-axis command current Id * as an input and the d-axis current Idr as an output is used. ). The third transfer function 201 represented by G3 (s) receives the d-axis current deviation ΔId and outputs a d-axis voltage Vd that is an operation amount for performing feedback control of the d-axis current deviation ΔId to zero. The third transfer function 201 corresponds to the stator control unit 73.

  A fourth transfer function 202 represented by G4 (s) modeling the frequency characteristics of the stator winding corresponds to a plant. The fourth transfer function 202 receives the d-axis voltage Vd and outputs a d-axis current Idr.

  The closed loop transfer function Gcd (s) of the d-axis sine wave current control system is expressed by the following equation (eq2).

なお、ｑ軸の正弦波電流制御系は、ｄ軸の正弦波電流制御系と同様であるため、その説明を省略する。

Since the q-axis sine wave current control system is the same as the d-axis sine wave current control system, the description thereof is omitted.

  As shown in FIG. 9, in the frequency characteristic regarding the gain of the closed loop transfer function of the control system, the frequency at which the gain becomes a predetermined value (eg, −3 dB) less than 0 dB is defined as the response frequency. In the present embodiment, the response frequency of the closed-loop transfer function of the d and q-axis sine wave current control system is higher than the response frequency of the closed-loop transfer function Gcf (s) of the field current control system. In the present embodiment, the response frequency of the rectangular wave current control system is higher than the response frequency of the field current control system.

  Incidentally, the response frequency may be calculated analytically based on, for example, a closed loop transfer function, or may be calculated based on a simulation. For example, the response frequency may be calculated experimentally using an actual control system and the control device 60.

(B) Open loop transfer function The higher the gain crossing angular frequency of the open loop transfer function, the higher the response. For this reason, the level of response may be determined by the gain crossover angular frequency. The field current control system will be described as an example. The open loop transfer function Gof (s) of this control system is expressed by the following equation (eq3).

（Ｃ）時定数
図１０に示すように、ステップ状に変化させられた指令値と実値とに基づいて規定される時定数τが短いほど、応答性が高くなる。このため、時定数τで応答性の高低を定めてもよい。

(C) Time Constant As shown in FIG. 10, the shorter the time constant τ defined based on the command value and the actual value changed in steps, the higher the response. For this reason, the level of responsiveness may be determined by the time constant τ.

  Subsequently, a control mode transition control process during idling operation of the engine 10 will be described. This process is executed by the control device 60, and when it is determined that the calculated rotation speed Nm of the rotor 31 is equal to or higher than the second rotor threshold Nth2, the sine wave PWM control is switched to the synchronous rectification control, and the rotation speed Nm is When it is determined that the first rotor threshold value Nth1 is smaller than the second rotor threshold value Nth2, it is a process of switching from synchronous rectification control to sine wave PWM control. Here, the first rotor threshold value Nth1 and the second rotor threshold value Nth2 of the present embodiment will be described.

  Referring to FIG. 12, the rotational speed of rotor 31 corresponding to first command rotational speed Netgt1 is defined as first rotor rotational speed Nm1. The first rotor rotational speed Nm1 is determined based on the transmission ratio from the output shaft 10a to the rotor 31 determined by the pulley ratio and the like, and the first command rotational speed Netgt1. For example, when the first command rotation speed Netgt1 is 700 rpm and the gear ratio is 3, the first rotor rotation speed Nm1 is 2100 rpm.

  The rotation speed of the rotor 31 corresponding to the second command rotation speed Netgt2 is set as the second rotor rotation speed Nm2 (> Nm1). The second rotor rotational speed Nm2 is determined based on the speed ratio from the output shaft 10a to the rotor 31 determined by the pulley ratio and the like, and the second command rotational speed Netgt2.

  When the normal control is performed, the maximum assumed fluctuation amount of the engine rotation speed Ner with respect to the first command rotation speed Netgt1 toward the high rotation side is set as the first engine fluctuation amount ΔNe1 (> 0), and the first engine fluctuation amount ΔNe1. The amount of fluctuation in the rotational speed of the rotor 31 corresponding to the first rotor fluctuation amount ΔNm1. In other words, the maximum value that the rotation speed Nm of the rotor 31 can take when the normal control is performed is “Nm1 + ΔNm1”. The first rotor fluctuation amount ΔNm1 is determined based on the speed ratio from the output shaft 10a to the rotor 31 determined by the pulley ratio and the like, and the first engine fluctuation amount ΔNe1. For example, when the first engine fluctuation amount ΔNe1 is 80 rpm and the gear ratio is 3, the first rotor fluctuation amount ΔNm1 is 240 rpm.

  Referring to FIG. 13, the maximum assumed fluctuation amount of the engine rotation speed Ner toward the low rotation side with respect to the second command rotation speed Netgt2 when the idle-up control is performed is defined as a second engine fluctuation amount ΔNe2 (> 0). The variation amount of the rotational speed of the rotor 31 corresponding to the second engine variation amount ΔNe2 is defined as a second rotor variation amount ΔNm2. That is, the minimum value that the rotational speed Nm of the rotor 31 can take when the idle-up control is performed is “Nm2−ΔNm2”.

  The first rotor threshold Nth1 is set to a value that is equal to or higher than the first rotor rotational speed Nm1 and smaller than the value obtained by subtracting the second rotor fluctuation amount ΔNm2 from the second rotor rotational speed Nm2. In the present embodiment, the first rotor threshold value Nth1 is set to a value higher than the added value of the first rotor rotational speed Nm1 and the first rotor fluctuation amount ΔNm1. The engine speed corresponding to the first rotor threshold value Nth1 is referred to as a first engine threshold value Eth1, and the engine speed corresponding to the added value of the first rotor speed Nm1 and the first rotor fluctuation amount ΔNm1 is defined as a first engine maximum value Ne1H. I will call it.

  The second rotor threshold value Nth2 is set to a value that is greater than each of the added value of the first rotor rotational speed Nm1 and the first rotor fluctuation amount ΔNm1 and the first rotor threshold value Nth1 and equal to or less than the second rotor rotational speed Nm2. Has been. In the present embodiment, the second rotor threshold value Nth2 is set to a value lower than the value obtained by subtracting the second rotor fluctuation amount ΔNm2 from the second rotor rotational speed Nm2. The engine speed corresponding to the second rotor threshold value Nth2 is referred to as a second engine threshold value Eth2, and the engine speed corresponding to a value obtained by subtracting the second rotor fluctuation amount ΔNm2 from the second rotor speed Nm2 is defined as a second engine minimum value Ne2L. I will call it.

  FIG. 11 shows the procedure of the transition control process during idling operation. This process is repeatedly executed by the control device 60, for example, every predetermined control cycle.

  In step S11, it is determined whether or not the calculated rotation speed Nm of the rotor 31 has become equal to or higher than the second rotor threshold Nth2 in the current control cycle. The affirmative determination is made in step S11 when, for example, the combustion control of the engine 10 is switched from the normal time control to the idle up control. If a negative determination is made in step S11, the process proceeds to step S12, and it is determined whether or not the calculated rotation speed Nm of the rotor 31 has become equal to or less than the first rotor threshold Nth1 in the current control cycle. An affirmative determination is made in step S12 when, for example, the combustion control of the engine 10 is switched from the idle up control to the normal time control.

  If an affirmative determination is made in step S12, the process proceeds to step S13, and the determination flag F is set to zero. The determination flag F is instructed to execute sine wave PWM control by 0, and instructed to execute synchronous rectification control by 1. In the present embodiment, the initial value of the determination flag F is 0.

  If an affirmative determination is made in step S11, the process proceeds to step S14, and the determination flag F is set to 1. If a negative determination is made in step S12, the currently executed control is continued.

  If the processes of steps S13 and S14 are completed, or if a negative determination is made in step S12, the process proceeds to step S15. In step S15, it is determined whether or not the determination flag F is 1. If it is determined in step S15 that the determination flag F is 0, the process proceeds to step S16 to execute the sine wave PWM control shown in FIG. On the other hand, if it is determined in step S16 that the determination flag F is 1, the process proceeds to step S17 to execute the synchronous rectification control shown in FIG.

  Next, a method for setting the field command current If * will be described with reference to FIGS. 12 and 13. In the present embodiment, the field command setting unit 80, as shown in FIGS. 12 and 13, before and after switching from one control to the other control among the sine wave PWM control and the synchronous rectification control. The current If * is set discontinuously. 12A and 13A show the DC current IDC flowing from the connection point of the high-potential side electric path Lp to the high-potential side terminal of the capacitor 21 to the positive terminal side of the battery 20, and the engine speed Ner. Shows the relationship. FIGS. 12B and 13B show the relationship between the field command current If * and the engine speed Ner, and FIGS. 12C and 13C show the power generation torque Trqr of the rotating electrical machine 30. And the engine speed Ner.

  When switching from the normal time control to the idle up control, as shown in FIG. 12, the sine wave PWM control is switched to the synchronous rectification control. In this case, the field command current If * set based on the synchronous rectification map information in the control cycle immediately after switching to the synchronous rectification control is based on the sine wave map information in the control cycle immediately before switching to the synchronous rectification control. Is larger than the field command current If *. That is, when switching from sine wave PWM control to synchronous rectification control, the maximum value of field command current If * set based on sine wave map information is the field command set based on synchronous rectification map information. It is smaller than the minimum value of the current If *. Hereinafter, the reason for this setting will be described in comparison with a comparative example.

  As shown in FIG. 14B, the field command current If * is set to be continuous before and after switching from the sine wave PWM control to the synchronous rectification control, and the actual field current is continuously set before and after the control switching. This configuration is used as a comparative example. In the comparative example, the power generation torque of the rotating electrical machine 30 is greatly reduced before and after the control switching. This is because the loss generated in the current path including the stator winding and the inverter 40 is smaller in the synchronous rectification control than in the sinusoidal PWM control, and the current flowing in the stator winding is controlled to obtain the command generated voltage VD *. This is because it decreases before and after switching. The reason why the loss generated by the synchronous rectification control is smaller than the loss generated by the sine wave PWM control is, for example, that the switching loss generated by the synchronous rectification control is smaller than the switching loss generated by the sine wave PWM control. It is.

  In view of this point, when switching from sine wave PWM control to synchronous rectification control, the actual field current immediately after switching to synchronous rectification control is greater than the actual field current immediately before switching to synchronous rectification control. Thus, the field command current If * is set in the manner shown in FIG. Thereby, the fall of the power generation torque when switching from the sine wave PWM control to the synchronous rectification control can be suppressed, and the deviation between the power generation torque and the command torque Trq * when the control is switched can be suppressed. As a result, fluctuations in power generation torque can be suppressed.

  On the other hand, when the control is switched from the idle-up control to the normal control, the synchronous rectification control is switched to the sine wave PWM control as shown in FIG. In this case, the field command current If * set based on the sine wave map information in the control cycle immediately after switching to the sine wave PWM control is the synchronous rectification map information in the control cycle immediately before switching to the sine wave PWM control. Is smaller than the field command current If * set based on Hereinafter, the reason for this setting will be described in comparison with a comparative example.

  As shown in FIG. 15B, the field command current If * is set to be continuous before and after switching from the synchronous rectification control to the sine wave PWM control, and the actual field current is continuously set before and after the control switching. This configuration is used as a comparative example. In the comparative example, the power generation torque of the rotating electrical machine 30 greatly increases before and after the control switching. This is because the loss generated in the current path including the stator winding and the inverter 40 is larger in the sine wave PWM control than in the synchronous rectification control, and the current flowing in the stator winding is controlled in order to obtain the command generated voltage VD *. This is because it increases before and after switching.

  In view of this point, when switching from synchronous rectification control to sine wave PWM control, the actual field current immediately after switching to sine wave PWM control is greater than the actual field current immediately before switching to sine wave PWM control. The field command current If * is set in the manner shown in FIG. Thereby, an increase in the power generation torque when switching from the synchronous rectification control to the sine wave PWM control can be suppressed, and a deviation between the power generation torque and the command torque Trq * when the control is switched can be suppressed. As a result, fluctuations in power generation torque can be suppressed.

<Modification of First Embodiment>
In the sine wave map information, instead of the rotation speed Nm of the rotor 31, the engine rotation speed Ner and the field command current If * are related so that the field command current If * increases as the engine rotation speed Ner increases. Information may be provided. In this case, the field command setting unit 80 may set the field command current If * based on the acquired engine rotation speed Ner, power supply voltage VDC, command torque Trq, and sine wave map information. Similarly, with respect to the synchronous rectification map information, instead of the rotational speed Nm of the rotor 31, the engine rotational speed Ner may be related to the field command current If *.

  The information related to the rotation speed Nm and the field command current If * is not limited to the map information, and for example, information on a mathematical formula related to the rotation speed Nm and the field command current If * It may be.

  The second rotor threshold value Nth2 may be set to the second rotor rotational speed Nm2, for example. Further, the first rotor threshold value Nth1 may be set to the first rotor rotational speed Nm1.

Second Embodiment
Hereinafter, the second embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In the present embodiment, the stator generator 75 shown in FIG. 3 performs sine wave PWM control or overmodulation PWM control based on the carrier signal, each phase command voltage Vu *, Vv *, Vw * and the power supply voltage VDC. Each drive signal for driving on / off the switches SUp to SWn of the inverter 40 is generated.

  Explaining the overmodulation PWM control, the stator generator 75 generates the phase command voltages Vu *, Vv *, Vw * when the peak value of each phase command voltage Vu *, Vv *, Vw * exceeds the power supply voltage VDC. A drive signal for each of the switches SUp to SWn of the inverter 40 is generated based on a comparison between the value divided by “VDC / 2” and the carrier signal. In overmodulation PWM control, the value obtained by dividing the amplitude of each phase command voltage Vu *, Vv *, Vw * by “VDC / 2” is larger than the amplitude of the carrier signal. FIG. 16 shows the transition of each waveform for one phase when overmodulation PWM control is executed. FIGS. 16A to 16C correspond to FIGS. 5A to 5C.

  On the other hand, the synchronization generator 93 in FIG. 6 performs rectangular wave control or synchronous rectification control. In the rectangular wave control, the peak value of each phase command voltage Vu *, Vv *, Vw * exceeds the power supply voltage VDC, and the peak value of the generated voltage (that is, counter electromotive voltage) of each phase winding 34U to 34W is This is performed when the power supply voltage is lower than VDC. FIG. 17 shows the transition of each waveform for one phase when rectangular wave control is executed. FIGS. 17A to 17C correspond to FIGS. 5A to 5C. FIG. 17 shows the dead time set in the rectangular wave control as DT2. In this embodiment, DT2 is shorter than DT1 shown in FIG. DT1 is set to a time longer than the maximum value of the dead time range that can be set, for example, in sine wave PWM control, overmodulation PWM control, and rectangular wave control.

  In this embodiment, in the transition control process, when shifting from sine wave PWM control to synchronous rectification control, switching is performed in the order of sine wave PWM control, overmodulation PWM control, rectangular wave control, and synchronous rectification control. On the other hand, when shifting from synchronous rectification control to sine wave PWM control, switching is performed in the order of synchronous rectification control, rectangular wave control, overmodulation PWM control, and sine wave PWM control.

  The procedure of the transition control process according to the present embodiment is shown using FIG. This process is repeatedly executed by the control device 60, for example, every predetermined control cycle. 18, for the sake of convenience, the same reference numerals are given to the processing shown in FIG. 11 or the processing corresponding to this processing.

  If it is determined in step S15 that the determination flag F is 1, the process proceeds to step S20 and the first process is performed. FIG. 19 shows the procedure of the first process.

  In step S201, it is determined whether or not the control in the previous control cycle was sine wave PWM control.

  When it is determined in step S201 that the sine wave PWM control has been performed, the process proceeds to step S202, and it is determined whether or not the timing for switching to the overmodulation PWM control is included in the current control cycle. As an example of this determination method, in this embodiment, when it is determined that the modulation rate Mr exceeds the first modulation rate Ma (for example, 1) in the current control cycle, it is determined that the switching timing is included in the current control cycle. . In the present embodiment, the modulation factor Mr is calculated as “Vn / VDC”, where Vn indicates a voltage amplitude that is the magnitude of the voltage vector. The voltage amplitude Vn may be calculated based on, for example, the d-axis command voltage Vd * and the q-axis command voltage Vq *.

  If a negative determination is made in step S202, the process proceeds to step S203, and sine wave PWM control is performed with the configuration shown in FIG. On the other hand, when an affirmative determination is made in step S202, the process proceeds to step S204, and overmodulation PWM control is performed with the configuration shown in FIG.

  If a negative determination is made in step S201, the process proceeds to step S205, in which it is determined whether or not the control in the previous control cycle was overmodulation PWM control.

  If it is determined in step S205 that the overmodulation PWM control has been performed, the process proceeds to step S206 to determine whether or not the timing for switching to the rectangular wave control is included in the current control cycle. As an example of this determination method, in this embodiment, when it is determined that the modulation rate Mr exceeds the second modulation rate Mb (eg, 1.27) larger than the first modulation rate Ma in the current control cycle, the switching timing Is included in the current control cycle.

  If a negative determination is made in step S206, the process proceeds to step S204. On the other hand, if an affirmative determination is made in step S206, the process proceeds to step S207 to switch from overmodulation PWM control to rectangular wave control. The rectangular wave control is performed by the configuration shown in FIG.

  If a negative determination is made in step S205, the process proceeds to step S208, and it is determined whether or not the control in the previous control cycle was rectangular wave control.

  If it is determined in step S208 that the control is rectangular wave control, the process advances to step S209 to determine whether or not the timing for switching to synchronous rectification control is included in the current control cycle. As an example of this determination method, in this embodiment, when it is determined that the rotational speed Nm of the rotor 31 is higher than the threshold speed, it is determined that the switching timing is included in the current control cycle. The threshold speed is set to a value by which it can be determined whether or not the generated voltage is higher than the output voltage of the battery 20.

  If a negative determination is made in step S209, the process proceeds to step S207. On the other hand, if an affirmative determination is made in step S209, the process proceeds to step S210 to switch from rectangular wave control to synchronous rectification control. The synchronous rectification control is performed by the configuration shown in FIG.

  Returning to the description of FIG. 18 above, when it is determined in step S15 that the determination flag F is 0, the process proceeds to step S30, and the second process is performed. FIG. 20 shows the procedure of the second process.

  In step S301, it is determined whether the control in the previous control cycle was synchronous rectification control.

  When it is determined in step S301 that the synchronous rectification control has been performed, the process proceeds to step S302, and it is determined whether or not the switching timing to the rectangular wave control is included in the current control cycle. As an example of this determination method, in the present embodiment, when it is determined that the rotational speed Nm of the rotor 31 is equal to or lower than the threshold speed, it is determined that the switching timing is included in the current control cycle.

  If a negative determination is made in step S302, the process proceeds to step S303, and synchronous rectification control is performed with the configuration shown in FIG. On the other hand, if an affirmative determination is made in step S302, the process proceeds to step S304, and rectangular wave control is performed with the configuration shown in FIG.

  If a negative determination is made in step S301, the process proceeds to step S305, and it is determined whether or not the control in the previous control cycle was rectangular wave control.

  If it is determined in step S305 that the rectangular wave control has been performed, the process proceeds to step S306, where it is determined whether the timing for switching to overmodulation PWM control is included in the current control cycle. As an example of this determination method, in this embodiment, when it is determined that the modulation rate Mr is equal to or less than the second modulation rate Mb in the current control cycle, it is determined that the switching timing is included in the current control cycle.

  If a negative determination is made in step S306, the process proceeds to step S304. On the other hand, if an affirmative determination is made in step S306, the process proceeds to step S307 to switch from rectangular wave control to overmodulation PWM control.

  If a negative determination is made in step S305, the process proceeds to step S308, in which it is determined whether or not the control in the previous control cycle was overmodulation PWM control.

  If it is determined in step S308 that the overmodulation PWM control has been performed, the process advances to step S309 to determine whether or not the timing for switching to the sine wave PWM control is included in the current control cycle. As an example of this determination method, in this embodiment, when it is determined that the modulation rate Mr is equal to or less than the first modulation rate Ma in the current control cycle, it is determined that the switching timing is included in the current control cycle.

  If a negative determination is made in step S309, the process proceeds to step S307. On the other hand, if an affirmative determination is made in step S309, the process proceeds to step S310, where overmodulation PWM control is switched to sine wave PWM control.

  21 to 24, voltage vectors applied to the stator windings (phase windings 34U, 34V, 34W) at the time of each control will be described. In the examples shown in FIGS. 21 to 24, the field current is the same for the sake of convenience in order to explain the influence of the current flowing in the stator winding. 21 to 24, VINV represents a command voltage vector determined from d and q axis voltages Vd * and Vq * commanded in each control, and VK is induced in the stator winding by the field current flowing. VR represents a voltage vector corresponding to the amount of voltage drop due to the resistance component of the stator winding. C1 represents a virtual circle drawn by the command voltage vector when the modulation factor Mr is 1, and C2 represents a virtual circle drawn by the command voltage vector when the modulation factor Mr is 1.27.

  FIG. 21 shows voltage vectors during sine wave PWM control. V1 represents a voltage vector finally applied to the stator winding, and is a combined vector of VINV, VK, and VR. FIG. 22 shows voltage vectors during overmodulation PWM control. V2 indicates a voltage vector finally applied to the stator winding, and is a combined vector of VINV, VK, and VR.

  FIG. 23 shows voltage vectors during rectangular wave control. V3 indicates a voltage vector finally applied to the stator winding, and is a combined vector of VINV, VK, and VR. FIG. 24 shows voltage vectors during synchronous rectification control. In FIG. 24, VDT indicates a voltage vector corresponding to the dead time DT1. V4 indicates a voltage vector finally applied to the stator winding, and is a combined vector of VINV, VK, VR, and VDT. In synchronous rectification control, since the dead time DT1 is long, the influence of VDT cannot be ignored.

  For example, when the overmodulation PWM control is directly switched to the synchronous rectification control, the difference between the voltage amplitude of the voltage vector V2 during the overmodulation PWM control and the voltage amplitude of the voltage vector V4 during the synchronous rectification control is large. The fluctuation of the power generation torque of the rotating electrical machine 30 becomes large. On the other hand, in this embodiment, after the sine wave PWM control, the overmodulation PWM control is switched to the synchronous rectification control via the rectangular wave control. Thus, the difference between the voltage amplitude of the voltage vector V2 during overmodulation PWM control and the voltage amplitude of the voltage vector V3 during rectangular wave control, and the voltage amplitude of the voltage vector V3 during rectangular wave control and the voltage during synchronous rectification control. Each difference from the voltage amplitude of vector V4 can be made smaller than the difference between the voltage amplitude of voltage vector V2 during overmodulation PWM control and the voltage amplitude of voltage vector V4 during synchronous rectification control. As a result, it is possible to suppress the fluctuation of the voltage amplitude that occurs when shifting from one to the other among the sine wave PWM control and the synchronous rectification control, and thus, the degree of suppression of the fluctuation of the power generation torque of the rotating electrical machine 30 can be increased.

<Third Embodiment>
Hereinafter, the third embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  The field command setting unit 80 performs synchronous rectification in a control cycle immediately after the actual field current Ifr when the rotational speed Nm of the rotor 31 is determined to be equal to or higher than the second rotor threshold Nth2 is switched to synchronous rectification control. When it is determined that the current is less than or equal to the field command current If set based on the map information, the field command current If * set based on the synchronous rectification map information in the control cycle immediately after switching to the synchronous rectification control is obtained. Increase the predetermined value ΔF (> 0). This is to suppress fluctuations in the power generation torque of the rotating electrical machine 30 when switching from sine wave PWM control to synchronous rectification control.

  That is, in order to suppress fluctuations in the power generation torque, as described in the first embodiment, the actual field current Ifr immediately after switching to the synchronous rectification control is changed to the actual field current Ifr immediately before switching to the synchronous rectification control. It is required to be larger than the field current Ifr. However, since the responsiveness of the field current Ifr is lower than the responsiveness of the current flowing through the stator winding, the actual field current Ifr immediately after switching to the synchronous rectification control is switched to the synchronous rectification control. Immediately after being generated, it may become equal to or less than the field command current If * set based on the synchronous rectification map information. In this case, the actual field current Ifr immediately after switching to the synchronous rectification control cannot be made larger than the actual field current Ifr immediately before switching to the synchronous rectification control, and fluctuations in power generation torque are suppressed. There is a risk that it will not be possible. In order to cope with this problem, the field command current If * is increased by a predetermined value ΔF.

  On the other hand, the field command setting unit 80 controls the control period immediately after the actual field current Ifr when the rotational speed Nm of the rotor 31 is determined to be equal to or lower than the first rotor threshold Nth1 is switched to the sine wave PWM control. If it is determined that the current is greater than or equal to the field command current If * set based on the sine wave map information, the field set based on the sine wave map information in the control cycle immediately after switching to the sine wave PWM control The command current If * is decreased by a predetermined value ΔF. This is to suppress fluctuations in the power generation torque of the rotating electrical machine 30 when switching from synchronous rectification control to sine wave PWM control.

  That is, in order to suppress fluctuations in power generation torque, as described in the first embodiment, the actual field current Ifr immediately after switching to sine wave PWM control is immediately before switching to sine wave PWM control. It is required to be smaller than the actual field current Ifr. However, since the responsiveness of the field current Ifr is lower than the responsiveness of the current flowing through the stator winding, the actual field current Ifr immediately after switching to the sine wave PWM control is the sine wave PWM control. Immediately after switching to, the field command current If * set based on the sine wave map information may be exceeded. In this case, the actual field current Ifr immediately after switching to the sine wave PWM control cannot be made smaller than the actual field current Ifr immediately before switching to the sine wave PWM control, and the fluctuation of the generated torque is reduced. There is a possibility that it cannot be suppressed. In order to cope with this problem, the field command current If * is decreased by a predetermined value ΔF.

  FIG. 25 shows the procedure of the migration control process. This process is repeatedly executed by the control device 60, for example, every predetermined control cycle.

  In step S40, it is determined whether or not the rotational speed Nm of the rotor 31 has become equal to or higher than the second rotor threshold Nth2 in the current control cycle.

  When an affirmative determination is made in step S40, the process proceeds to step S41, and the field current Ifr detected in the previous control cycle in which the sine wave PWM control was performed is synchronized in the current control cycle that is switched to the synchronous rectification control. It is determined whether or not the field command current If * is set based on the rectification map information.

  When it is determined in step S41 that the field current Ifr is larger than the field command current If *, the process proceeds to step S42, and the field command current If * set based on the synchronous rectification map information is set in the synchronous rectification control. The field current controller 81 uses it as it is.

  On the other hand, if it is determined in step S41 that the field current Ifr is equal to or less than the field command current If *, the process proceeds to step S43. In step S43, the field current control unit 81 uses a value obtained by adding a predetermined value ΔF to the field command current If * set based on the synchronous rectification map information in the synchronous rectification control.

  In addition, after the process of step S43, the field command current If * may be made asymptotic toward the field command current If * set based on the synchronous rectification map information for each control cycle.

  If a negative determination is made in step S40, the process proceeds to step S44, and it is determined whether or not the rotational speed Nm of the rotor 31 has become equal to or lower than the first rotor threshold Nth1 in the current control cycle.

  If an affirmative determination is made in step S44, the process proceeds to step S45, and the field current Ifr detected in the previous control cycle in which the synchronous rectification control was performed is sine in the current control cycle in which the sine wave PWM control is switched. It is determined whether or not the field command current If * is set based on the wave map information.

  When it is determined in step S45 that the field current Ifr is smaller than the field command current If *, the process proceeds to step S46, and the field command current If * set based on the sine wave map information in the sine wave PWM control. Are used as they are in the field current control unit 81.

  On the other hand, if it is determined in step S45 that the field current Ifr is greater than or equal to the field command current If *, the process proceeds to step S47. In step S47, the field current control unit 81 uses a value obtained by subtracting a predetermined value ΔF from the field command current If * set based on the sine wave map information in the sine wave PWM control.

  In addition, after the process of step S47, the field command current If * may be made asymptotic toward the field command current If * set based on the sine wave map information for each control cycle.

  According to the present embodiment described above, fluctuations in power generation torque when switching from one to the other among the sine wave PWM control and the synchronous rectification control can be suppressed.

<Modification of Third Embodiment>
The predetermined value ΔF in step S43 in FIG. 25 may be different from the predetermined value ΔF in step S47. In this case, the predetermined value ΔF in step S43 may be set as the first predetermined value, and the predetermined value ΔF in step S47 may be set as the second predetermined value.

  One of the configuration for increasing the field command current If * by the predetermined value ΔF and the configuration for decreasing the field command current If * by the predetermined value ΔF may not be implemented.

<Fourth embodiment>
Hereinafter, the fourth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment. In the present embodiment, as shown in FIGS. 26 and 27, the control device 60 is provided with a low-pass filter. 26 and 27, the same reference numerals are given for the sake of convenience to the configurations shown in FIG. 3 and FIG.

  The first filter unit 301 performs a low-pass filter process on the detected power supply voltage VDC. The power supply voltage VDC that has been subjected to the low-pass filter processing is used by the voltage deviation calculation unit 61.

  The second filter unit 302 performs a low-pass filter process on the command torque Trq * calculated by the torque calculation unit 62. The command torque Trq * subjected to the low-pass filter process is used in the d and q axis command setting units 71 and 72 and the field command setting unit 80.

  The third filter unit 303 performs a low-pass filter process on the calculated rotation speed Nm of the rotor 31. The rotation speed Nm that has been subjected to the low-pass filter process is used by the field command setting unit 80. Further, the rotation speed Nm subjected to the low-pass filter process is used for comparison with the first rotor threshold value Nth1 and the second rotor threshold value Nth2 for control switching.

  The fourth filter unit 304 performs low-pass filter processing on the field command current If * set by the field command setting unit 80. The field command current If * subjected to the low-pass filter process is used by the field current control unit 81.

  Here, since each of the filter units 301 to 304 as the annealing means is provided, there is a concern that the responsiveness of the field current Ifr to the field command current If * is greatly reduced. For this reason, it is necessary to reduce the time constant of each filter part 301-304 and to improve the responsiveness of the field current Ifr. However, in this case, for example, the set field command current Ifr may fluctuate greatly due to fluctuations in the rotational speed Nm, and the field current Ifr may fluctuate greatly.

  In view of this, the field command setting unit 80 of the present embodiment has the field command current If regardless of the power supply voltage VDC and the rotation speed Nm from the switching timing from the sine wave PWM control to the synchronous rectification control over the first determination period. Keep * at a constant value. Further, the field command setting unit 80 sets the field command current If * to a constant value regardless of the power supply voltage VDC and the rotation speed Nm from the switching timing from the synchronous rectification control to the sine wave PWM control over the second determination period. To maintain. Thereby, the fluctuation | variation of a field current is suppressed and the fluctuation | variation of the power generation torque of the rotary electric machine 30 is suppressed, improving the responsiveness of a field current.

  FIG. 28 shows the procedure of the migration control process. This process is repeatedly executed by the control device 60, for example, every predetermined control cycle. In FIG. 28, the process shown in FIG. 25 or the process corresponding to this process is denoted by the same reference numeral for convenience.

  In step S50, it is determined whether or not the first flag F1 is zero. In the present embodiment, the initial value of the first flag F1 is 0.

  When it is determined in step S50 that the first flag F1 is 0, the process proceeds to step S51, and it is determined whether or not the second flag F2 is 0. In the present embodiment, the initial value of the second flag F2 is 0.

  If it is determined in step S51 that the second flag F2 is 0, the process proceeds to step S40. If a positive determination is made in step S40, the process proceeds to step S52, and the first flag F1 is set to 1. In addition, using the timer, the counting of the elapsed time CNT1 after the affirmative determination in step S40 is started.

  When the process of step S52 is completed, or when it is determined in step S50 that the first flag F1 is 1, the process proceeds to step S53, and it is determined whether or not the elapsed time CNT1 is equal to or longer than the first determination period Cth1. To do.

  If a negative determination is made in step S53, the process proceeds to step S54, where the field command current If * is changed to the field command current If * set based on the synchronous rectification map information in the control cycle immediately after switching to the synchronous rectification control. maintain.

  If it is determined in step S53 that the elapsed time CNT1 has become equal to or longer than the first determination period Cth1, the process proceeds to step S55, and the first flag F1 and the elapsed time CNT1 are reset to 0. In addition, field command current If * is set based on command torque Trq *, power supply voltage VDC, rotation speed Nm, and synchronous rectification map information. If the absolute value of the difference between the field command current If * maintained in step S54 and the field command current If * set in step S53 is greater than a predetermined amount, the current field command current If * May be made asymptotic toward the field command current If * set based on the synchronous rectification map information for each control period.

  If a negative determination is made in step S40, the process proceeds to step S44. When an affirmative determination is made in step S44, the process proceeds to step S56, and the second flag F2 is set to 1. Further, using the timer, the counting of the elapsed time CNT2 after the affirmative determination is made in step S44 is started.

  When the process of step S56 is completed, or when it is determined in step S51 that the second flag F2 is 1, the process proceeds to step S57, where it is determined whether the elapsed time CNT2 is equal to or longer than the second determination period Cth2. To do.

  When a negative determination is made in step S57, the process proceeds to step S58, and the field command current If * is set based on the sine wave map information in the control cycle immediately after switching to the sine wave PWM control. To maintain.

  If it is determined in step S57 that the elapsed time CNT2 has become equal to or longer than the second determination period Cth2, the process proceeds to step S59, and the second flag F2 and the elapsed time CNT2 are reset to 0. Further, field command current If * is set based on command torque Trq *, power supply voltage VDC and rotation speed Nm, and sine wave map information. If the absolute value of the difference between the field command current If * maintained in step S58 and the field command current If * set in step S59 is greater than a predetermined amount, the current field command current If * May be made asymptotic toward the field command current If * set based on the sine wave map information for each control period.

<Modification of Fourth Embodiment>
The method of maintaining the field command current If * at a constant value over a predetermined period from the switching timing from the sine wave PWM control to the synchronous rectification control is not limited to the method using a timer. For example, as shown in FIG. 29, in the synchronous rectification map information, the field command current If * extends from the second rotor threshold value Nth2 to the added value of the second rotor threshold value Nth2 and the first specified value Δα (> 0). It may be a method of setting to a constant value. Here, the added value of the second rotor threshold value Nth2 and the first specified value Δα is a value smaller than the second rotor rotational speed Nm2.

  Further, the method of maintaining the field command current If * at a constant value over a predetermined period from the switching timing from the synchronous rectification control to the sine wave PWM control is not limited to the method using a timer. For example, as shown in FIG. 30, in the sine wave map information, the field command current If extends from the first rotor threshold value Nth1 to a value obtained by subtracting the second specified value Δβ (> 0) from the first rotor threshold value Nth1. A method of setting * to a constant value may be used. Here, a value obtained by subtracting the second specified value Δβ from the first rotor threshold value Nth1 is a value larger than the first rotor rotational speed Nm1.

  A structure in which the field command current If * is maintained at a constant value over a predetermined period from the switching timing from the sine wave PWM control to the synchronous rectification control, and from the switching timing from the synchronous rectification control to the sine wave PWM control within the predetermined period. Of the configurations that maintain the field command current If * at a constant value, one configuration may not be implemented.

<Fifth Embodiment>
Hereinafter, the fifth embodiment will be described with reference to the drawings with a focus on differences from the first embodiment.

  In the present embodiment, as shown in FIG. 31B, in the sine wave map information, the field command current If * in the transient region from “Nm1 + ΔNm1” to the second rotor threshold value Nth2 is indicated by a one-dot chain line. Is set to be larger than the field command current If *. The field command current If * immediately before switching from the sine wave PWM control to the synchronous rectification control is set to be larger than the field command current If * immediately after the switching. The field current Ifr immediately before switching to the synchronous rectification control is set to be the field command current If * immediately after switching. FIGS. 31A and 31B correspond to FIGS. 12A and 12B. This setting is made in consideration of the responsiveness of the field current Ifr and the increasing speed (engine rotational acceleration) of the engine rotational speed Ner when switching from normal time control to idle-up control.

  That is, the higher the engine rotational acceleration, the shorter the time from when the instruction to switch to the idle-up control is given until the rotational speed Nm of the rotor 31 reaches the second rotor threshold value Nth2. In this case, when the sine wave map information is set as shown in FIG. 12B, the actual field current immediately after switching to the synchronous rectification control is changed to the actual field current immediately before switching to the synchronous rectification control. There is a possibility that it cannot be made larger than the current. Therefore, by setting the sine wave map information in the mode shown in FIG. 31B, the actual field current immediately after switching is made larger than the actual field current immediately before switching.

  In the sine wave map information, the field command current If * in the transient region from “Nm1 + ΔNm1” to the second rotor threshold Nth2 corresponds to an average engine rotational acceleration assumed when switching to idle-up control. It only needs to be set to a value. If the field current Ifr immediately before switching to the synchronous rectification control is equal to or less than the field command current If * immediately after switching, the process of step S43 in FIG. 25 may be executed.

  On the other hand, in the present embodiment, as shown in FIG. 32B, in the synchronous rectification map information, the field command current If * in the transient region from “Nm2−ΔNm2” to the first rotor threshold Nth1 is indicated by a one-dot chain line. It is set smaller than the field command current If * of the seventh embodiment shown. The field command current If * immediately before switching from the synchronous rectification control to the sine wave PWM control is set to be smaller than the field command current If * immediately after the switching. The field current Ifr immediately before switching to the sine wave PWM control is set to be the field command current If * immediately after switching. FIGS. 32A and 32B correspond to FIGS. 13A and 13B. This setting is made in consideration of the responsiveness of the field current Ifr and the rate of decrease in the engine rotation speed Ner (engine rotation deceleration) when switching from idle-up control to normal control.

  That is, the higher the engine rotation deceleration, the shorter the time from when the instruction to switch to the normal control is given until the rotation speed Nm of the rotor 31 reaches the first rotor threshold value Nth1. In this case, when the synchronous rectification map information is set as shown in FIG. 13B, the actual field current immediately after switching to the sine wave PWM control is changed to the actual field current immediately before switching to the sine wave PWM control. There is a possibility that it cannot be made smaller than the field current. Therefore, by setting the synchronous rectification map information in the mode shown in FIG. 32B, the actual field current immediately after switching is made smaller than the actual field current immediately before switching.

  In the synchronous rectification map information, the field command current If * in the transient region from “Nm2−ΔNm2” to the first rotor threshold Nth1 is an average engine rotation deceleration that is assumed when switching to normal control. It is only necessary to set a value corresponding to. If the field current Ifr immediately before switching to the sine wave PWM control is equal to or greater than the field command current If * immediately after switching, the process of step S47 of FIG. 25 may be executed.

<Sixth Embodiment>
Hereinafter, the sixth embodiment will be described with reference to the drawings with a focus on differences from the fifth embodiment.

  In the present embodiment, the field command setting unit 80, as shown in FIG. 33 (b), the field command current If * until the rotational speed Nm of the rotor 31 reaches the second rotor threshold Nth2 from “Nm1 + ΔNm1”. Is set larger with respect to the field command current If * set based on the sine wave map information as the increase speed (rotational acceleration AP) of the rotational speed of the rotor 31 is larger. 33A and 33B correspond to the previous FIGS. 12A and 12B.

  Specifically, the field command setting unit 80 first sets the first coefficient K1 (≧ 1) to be larger as the rotational acceleration AP of the rotor 31 after the instruction to switch to idle-up control is larger. The rotational acceleration AP may be calculated based on the rotational speed Nm. Then, the field command setting unit 80 sets the field command current If * until the rotation speed Nm reaches “the second rotor threshold Nth2” from “Nm1 + ΔNm1” based on the sine wave map information. Is multiplied by the first coefficient K1.

  As a result, the actual field current immediately after switching to the synchronous rectification control is switched even when the engine rotational acceleration when switching from the normal control to the idle up control differs for each switching to the idle up control. It can be made larger than the actual field current immediately before.

  On the other hand, as shown in FIG. 34B, the field command setting unit 80 sets the field command current If * until the rotational speed Nm of the rotor 31 reaches “the first rotor threshold value Nth1” from “Nm2−ΔNm2”. The field command current If * set based on the synchronous rectification map information is set smaller as the rotational speed decrease speed (rotational deceleration DP) of the rotor 31 is larger. FIGS. 34A and 34B correspond to FIGS. 13A and 13B.

  Specifically, the field command setting unit 80 first sets the second coefficient K2 (0 <K2 <1) as the rotational deceleration DP of the rotor 31 after the instruction to switch to the normal control is increased. Set smaller. The rotation deceleration DP may be calculated based on the rotation speed Nm. Then, the field command setting unit 80 sets the field command current If * until the rotation speed Nm reaches “the first rotor threshold Nth1” from “Nm2−ΔNm2” based on the synchronous rectification map information. A value obtained by multiplying If * by the second coefficient K2 is set.

  As a result, even if the engine rotation deceleration in the case of switching from idle-up control to normal time control is different every time switching to normal time control, the actual field current immediately after switching to sine wave PWM control is reduced. The actual field current immediately before switching can be made smaller.

<Modification of Sixth Embodiment>
The larger the rotational acceleration AP of the rotor 31, the larger the setting relative to the field command current If * set based on the sine wave map information, and the larger the rotational deceleration DP of the rotor 31, the greater the synchronous rectification map information. Of the configurations that are set smaller than the field command current If * that is set on the basis, one configuration may not be implemented.

<Other embodiments>
Each of the above embodiments may be modified as follows.

  In the sine wave PWM control of each of the above embodiments, a process in which the third harmonic is superimposed on each phase command voltage or a process in which two-phase modulation is used may be performed.

  -Each command value is not restricted to the structure memorize | stored in the memory 60a. For example, each command value may be calculated for each control cycle based on the acquired command torque Trq *, rotation speed Nm, and power supply voltage VDC.

  -You may apply the structure of 2nd Embodiment to the structure of 3rd-6th embodiment.

  -The main control amount of the rotating electrical machine is not limited to torque. For example, as shown in FIG. 1, a DC current IDC that flows from a connection point of the high potential side electric path Lp to the high potential side terminal of the capacitor 21 to the positive terminal side of the battery 20 may be used. In this case, the DC current IDC may be defined as a negative value when the rotating electrical machine 30 is driven as an electric motor, and the DC current IDC may be defined as a positive value when the rotating electrical machine 30 is driven as a generator. Further, for example, the main control amount may be the power consumption or generated power of the rotating electrical machine 30.

  The stator 33 of the rotating electrical machine may include a first stator winding group and a second stator winding group. Each stator winding group includes three-phase windings, and an inverter is provided corresponding to each stator winding group. The spatial phase difference Δθ, which is an angle formed by the first stator winding group and the second stator winding group, may be, for example, 30 ° in electrical angle.

  The rotating electric machine is not limited to a star connection, and may be a Δ connection, for example.

  DESCRIPTION OF SYMBOLS 30 ... Rotary electric machine, 32 ... Field winding, 34U-34W ... U, V, W phase winding, 40 ... Inverter, 60 ... Control apparatus.

Claims (9)

A rotating electrical machine (30) having a field winding (32) and a stator winding (34U-34W);
An upper arm switch (SUp to SWp) and a lower arm switch (SUn to SWn) are connected in series. By switching the upper arm switch and the lower arm switch, a DC power source (20) and the stator winding are connected. In a control device (60) for a rotating electrical machine applied to a control system comprising an inverter (40) for transmitting power between
A field control unit (80 to 82) for controlling a field current flowing in the field winding to a field command current (If *) which is a command value thereof;
In the case where any one of the generated torque of the rotating electrical machine, the direct current supplied from the inverter to the DC power source, or the generated power of the rotating electrical machine is a main control amount, the generated voltage of the stator winding is the DC power source. On the basis of a command value for controlling the main control amount to a command control amount on the condition that the upper arm switch is turned on in at least a part of a period exceeding the voltage of A synchronous rectification control unit (93) for controlling a current flowing in the stator winding by turning on the upper arm switch and the lower arm switch once each while holding a dead time;
When the peak value of each phase voltage applied to the stator winding is equal to or less than the voltage of the DC power supply, the stator winding is applied to the stator winding based on a command value for controlling the main control amount to the command control amount. A sine wave control unit (70) that controls the current flowing in the stator winding by turning on and off the upper arm switch and the lower arm switch by sine wave PWM control so that each applied phase voltage has a PWM voltage waveform. ~ 75)
A control system for controlling the current flowing in the stator winding as a control amount by the control of the synchronous rectification control unit, a control system for controlling the current flowing in the stator winding as a control amount by the control of the sine wave control unit, and the field control Among the control systems that control the field current as a control amount by the control of the part, the control unit corresponding to the control system having the lowest responsiveness of the control amount is the field control unit,
When switching from the control of the sine wave control unit to the control of the synchronous rectification control unit, the field current immediately after switching to the control of the synchronous rectification control unit is immediately before being switched to the control of the synchronous rectification control unit. Immediately after switching to the control of the sine wave control unit, when the field control unit is controlled to be larger than the field current, and the control of the synchronous rectification control unit is switched to the control of the sine wave control unit. A control device for a rotating electrical machine comprising a transition control unit that performs a transition control process for controlling the field control unit so that the field current is smaller than the field current immediately before switching to the control of the sine wave control unit . The control system is mounted on a vehicle including an engine (10),
The vehicle includes a normal control for performing combustion control of the engine to control the rotational speed of the output shaft (10a) of the engine during the idling operation to a first command rotational speed (Netgt1), and the vehicle during the idling operation. An engine control device (11) is provided for performing idle-up control for performing combustion control of the engine so as to control the rotation speed of the output shaft to a second command rotation speed (Netgt2) higher than the first command rotation speed. ,
The rotating electrical machine has a function of generating power by receiving power supply from the output shaft,
The rotational speed of the rotor (31) of the rotating electrical machine corresponding to the first command rotational speed is a first rotor rotational speed (Nm1),
The rotation speed of the rotor corresponding to the second command rotation speed is defined as a second rotor rotation speed (Nm2), and the maximum assumed fluctuation amount of the rotation speed of the rotor toward the high rotation side with respect to the first rotor rotation speed is the first. The rotor fluctuation amount (ΔNm1)
The maximum assumed fluctuation amount of the rotation speed of the rotor toward the low rotation side with respect to the second rotor rotation speed is defined as a second rotor fluctuation amount (ΔNm2).
A value that is equal to or higher than the first rotor rotational speed and smaller than a value obtained by subtracting the second rotor fluctuation amount from the second rotor rotational speed is defined as a first rotor threshold (Nth1).
A value that is larger than each of the first rotor rotational speed plus the first rotor fluctuation amount and the first rotor threshold and equal to or less than the second rotor rotational speed is defined as a second rotor threshold (Nth2). The transition control process switches from the control of the sine wave control unit to the control of the synchronous rectification control unit when it is determined that the rotational speed of the rotor is equal to or higher than the second rotor threshold value, 2. The control device for a rotating electrical machine according to claim 1, wherein when the speed is determined to be equal to or less than the first rotor threshold value, the control is performed from the control of the synchronous rectification control unit to the control of the sine wave control unit.   The transition control process is performed immediately after switching from the control of the sine wave control unit to the control of the synchronous rectification control unit over a predetermined period (Cth1) to the control of the sine wave control unit. The control apparatus for a rotating electrical machine according to claim 2, which is a process of maintaining the magnetic command current at a constant value.   The transition control process is performed immediately after switching to the control of the synchronous rectification control unit over a predetermined period (Cth2) from the switching timing of the control of the synchronous rectification control unit to the control of the sine wave control unit. The control device for a rotating electrical machine according to claim 2 or 3, wherein the magnetic command current is maintained at a constant value.   In the transition control process, the field current when it is determined that the rotational speed of the rotor is equal to or greater than the second rotor threshold is immediately after switching from the control of the sine wave control unit to the control of the synchronous rectification control unit. And a step of increasing the field command current set immediately after switching to the control of the synchronous rectification control unit by a predetermined value (ΔF). The control apparatus of the rotary electric machine of any one of 2-4.   In the transition control process, the field current when it is determined that the rotational speed of the rotor is equal to or less than the first rotor threshold is immediately after switching from the control of the synchronous rectification control unit to the control of the sine wave control unit. The field command current that is set immediately after switching to the control of the sine wave control unit when it is determined that the field command current is equal to or greater than the field command current set to a predetermined value (ΔF). The control apparatus of the rotary electric machine of any one of 2-5. When the control of the sine wave control unit is performed, the field control unit is configured to increase the field command current as the rotation speed parameter, which is either the rotation speed of the rotor or a correlation value thereof, increases. Based on the first information in which the rotation speed parameter and the field command current are related to each other and the rotation speed parameter, the field command current is set, and the synchronous rectification control unit is controlled. In this case, based on the rotation speed parameter and the second information related to the rotation speed parameter and the field command current so that the field command current increases as the rotation speed parameter increases. Set field command current,
When switching from the normal time control to the idle-up control, the maximum value of the field command current set based on the first information is the field command current set based on the second information. Is smaller than the minimum value of
When the transition control process is switched from the normal control to the idle-up control, the rotational speed of the rotor is changed from the added value of the first rotor rotational speed and the first rotor fluctuation amount to the second rotor threshold value. The field command current in a period until the value is set to be larger with respect to the field command current set based on the first information as the rotational acceleration (AP) of the rotor is larger. The control apparatus of the rotary electric machine of any one of 2-6. When the control of the sine wave control unit is performed, the field control unit is configured to increase the field command current as the rotation speed parameter, which is either the rotation speed of the rotor or a correlation value thereof, increases. Based on the first information in which the rotation speed parameter and the field command current are related to each other and the rotation speed parameter, the field command current is set, and the synchronous rectification control unit is controlled. In this case, based on the rotation speed parameter and the second information related to the rotation speed parameter and the field command current so that the field command current increases as the rotation speed parameter increases. Set field command current,
When switching from the idle-up control to the normal time control, the maximum value of the field command current set based on the first information is the field command current set based on the second information. Is smaller than the minimum value of
When the transition control process is switched from the idle-up control to the normal time control, the rotational speed of the rotor is the first rotor threshold value obtained by subtracting the second rotor fluctuation amount from the second rotor rotational speed. In this process, the field command current in the period until the value becomes smaller with respect to the field command current set based on the second information as the rotational deceleration (DP) of the rotor is larger. The control apparatus of the rotary electric machine of any one of Claims 2-7. When the peak value of each phase voltage applied to the stator winding exceeds the voltage of the DC power supply, the upper arm switch so that the PWM voltage waveform has a higher modulation rate than the PWM waveform voltage by the sine wave control unit And an overmodulation control unit that drives the lower arm switch;
A rectangular wave control unit that turns on the upper arm switch and the lower arm switch once by rectangular wave control while sandwiching a dead time in one electrical angle cycle,
In the transition control process, when it is determined that the rotation speed of the rotor is equal to or higher than the second rotor threshold, the control of the sine wave control unit, the control of the overmodulation control unit, the control of the rectangular wave control unit, and the When switching is performed in the order of control of the synchronous rectification control unit and it is determined that the rotation speed of the rotating electrical machine has become equal to or lower than the first rotor threshold value, control of the synchronous rectification control unit, control of the rectangular wave control unit, and overmodulation control The control device for a rotating electrical machine according to any one of claims 2 to 8, wherein the control is performed in the order of control of a part and control of the sine wave control part.

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