JP2019162023A - Wireless power transmission device and power receiving side current detection circuit thereof - Google Patents

Abstract

To provide a wireless power transmission device capable of accurately adjusting a frequency to a resonance frequency without depending on a coupling coefficient or a circuit constant between a power transmission side coil and a power reception side coil.SOLUTION: A wireless power transmission device includes a dummy coil 4 connected in parallel to a power transmission side coil 3 with respect to a power supply circuit 6 and not magnetically coupled to a power reception side coil 11, and difference current detection means 7 for detecting a difference current between the dummy coil current iflowing through the dummy coil 4 and a primary current iflowing through the power transmission side coil 3, and the difference current detection means 7 detects a current difference (αi-i) for the ratio α=L/Lof inductance Lof the dummy coil 4 to the inductance Lof the power transmission side coil 3. The current difference (αi-i) is proportional to the power receiving coil current i, and the frequency can be adjusted to the exact series resonance frequency fby using the detected value of the current difference.SELECTED DRAWING: Figure 4

Description

  The present invention relates to a wireless power transmission technology that transmits power in a contactless manner without using a metal contact or a connector, and more particularly, a magnetic field coupling method that transmits power using electromagnetic induction between a power transmission side coil and a power reception side coil. Related to wireless power transmission technology.

  In recent years, wireless power transmission technology for transmitting power in a contactless manner has been used in a wide range of fields such as charging various electric devices and feeding electric vehicles. Wireless power transfer technology can be broadly classified into electric field coupling method using capacitance coupling (electric field coupling) between opposing electrode plates and magnetic field coupling method using mutual inductance coupling (magnetic field coupling) between parallel coils. However, when the air gap between the power transmission side and the power reception side is wide, a magnetic field coupling method that can generally increase the power transmission distance is used (see Non-Patent Documents 1 and 2). In the magnetic field coupling type wireless power transmission technology, a primary side coil (power transmission side coil) is provided on the power transmission side, and a secondary side coil (power reception side coil) is provided on the power reception side, and the primary side coil and the secondary side coil face each other. By arranging them together, magnetic field coupling is performed, an alternating current is passed through the primary side coil, and an induction current is generated in the secondary side coil to transmit power. At this time, in order to maximize power transmission efficiency, a resonance capacitor is provided in series with the secondary coil, and power transmission efficiency is maximized by transmitting power from the power transmission side at the resonance frequency. This method is called an electromagnetic induction method (see Patent Documents 1-2 and 1-2).

FIG. 25 shows a principle diagram of a typical magnetic field resonance type wireless power transmission. Magnetic resonance method has a presence or absence of the resonance capacitor C ₁ at the primary side (transmission side), the connection relationship between the resonant capacitor C ₂ load resistance R _L at the secondary side (power receiving side), (a) N-S-type (Non-resonant-Series type), (b) NP type (Non-resonant-Parallel type), (c) SS type (Series-Series type), (d) SP type (Series-Parallel) type), (e) N-L type (Non-resonant-Load type), (f) S-L type (Series-Load type), etc. (for example, refer nonpatent literatures 1 and 2). In the NL type and the SL type, the secondary power receiving side coil is separated into a resonance power receiving side coil L ₂ that causes resonance with the primary side and a power receiving side load coil L _L that extracts power. It is a thing. In addition to this, there is a method in which the primary power transmission side coil is separated into a resonance power transmission side coil that causes resonance with the secondary side and a power supply coil (power supply excite coil) that supplies power to the power supply. Yes (see FIG. 3 of Patent Document 7). In these magnetic field resonance systems, a “parallel resonance mode (antiresonance mode) in which the interlinkage magnetic flux of the primary coil L ₁ on the primary side and the interlinkage magnetic flux of the secondary coil L ₂ on the secondary side are opposite to each other. ) and "primary interlinkage magnetic flux of the coil L ₁ of the primary side and is the flux linkage of the secondary coil L ₂ of the secondary side becomes the same direction (the magnetic field compensator coupled state)" series resonant mode In the case of the NS type, for example, the resonance frequency of each resonance mode is expressed as follows (see FIG. 26 or page 60 of Non-Patent Document 3). 9).

Here, ω ₂ and ω ₀ are the angular frequencies of the parallel resonance mode and the series resonance mode, f ₂ and f ₀ are the frequencies of the parallel resonance mode and the series resonance mode, respectively, and L ₂ is the secondary coil. Inductance, C ₂ is the capacitance of the secondary side resonance capacitor, and k ₁₂ is a coupling constant between the primary side coil L ₁ and the secondary side coil L ₂ . Since the current flowing through the load resistance _RL on the secondary side becomes the maximum in the series resonance mode, the magnetic field resonance method generally uses magnetic field phase coupling in the series resonance mode.

However, although the resonance frequency ω ₀ of the series resonance mode depends on the coupling constant k ₁₂ as shown in the equation (1b), the coupling constant k ₁₂ is a position between the primary side coil L ₁ and the secondary side coil L _2. It varies greatly depending on the relationship. Further, as can be said at the both of the series resonance mode and the parallel resonance mode, the inductance L ₁ and the capacitance C ₂ of the secondary side has a large variation due to manufacturing error or aging deterioration, the resonance frequency by these variations omega ₂ , Ω ₀ fluctuates. Therefore, such following the variations in the resonance frequency omega ₂ and the resonance frequency omega _0, to adjust the frequency of the current supplied to the primary coil L _1, technically in performing power transmission with high efficiency is important.

Thus, as a technique for adjusting the frequency of the current supplied to the primary coil L _1, Patent Document 3 and Non-Patent Document 3, and is known as described in Patent Document 4-6. Patent Document 3 and Non-Patent Document 3, in the power receiving side device, the phase information of the resonance current flowing through the resonant capacitor C _2, is detected by the resonance current phase detecting means, phase information transmission means (phase information transmitting means and The phase information detected by the phase information receiving means) is transmitted to the power transmission side device without phase delay, and the drive circuit of the power transmission side device determines the drive frequency based on the phase information transmitted by the phase information transmission means. A wireless power transmission device that drives a side coil (power transmission side coil) is described (see FIG. 1 of Patent Document 3). The phase information transmission means wirelessly transmits information between the power receiving side device and the power transmission side device, and an optical transmission method using an LED and a phototransistor, an electromagnetic wave transmission method for high-frequency wireless transmission, and the like are described as examples. Has been. By adjusting the frequency of the current (primary current) that flows through the primary side coil (power transmission side coil) based on the phase information on the secondary side, the resonance frequency of the equation (1b) is adapted. Thus, the primary side frequency control is performed to achieve high transmission efficiency.

  Patent Document 4 discloses an apparatus for wirelessly feeding power from a power supply coil (L2) to a power reception side coil (L3) based on the magnetic field resonance phenomenon of the power supply coil (L2) and power reception side coil (L3). (L2), a power transmission control circuit (200) for supplying AC power from the power supply coil (L2) to the power receiving coil (L3) by supplying AC power to the power supply coil (L2) at the drive frequency, and AC A phase detection circuit (150) for detecting a phase difference between the voltage phase of the electric power and the phase of the current, and the phase detection circuit (150) outputs one of the voltage waveform of the AC power and the current waveform of the AC power by 90 Then, by detecting a shift between the first detection period in which the voltage level of the AC power is within a predetermined range and the second detection period in which the current level of the AC power is within the predetermined range, 0 degrees to 180 degrees. Detecting a phase difference in the range of, the power transmission control circuit (200), (see FIGS. 1 to 3 of 仝 document) the detected wireless power supply apparatus is described for adjusting the drive frequency in accordance with the phase difference. In this wireless power supply device, the phase of the current flowing through the power supply coil (L2) is compared with the phase of the voltage waveform output from the power transmission control circuit (200) so that the phases of both are shifted by 90 degrees. The drive frequency of the power transmission control circuit (200) is adjusted. Thereby, the power transmission efficiency is improved by causing the drive frequency to follow the resonance frequency. Patent Documents 5 and 6 also describe a wireless power supply apparatus having a similar principle configuration (however, Patent Document 5 describes the equation (1a) for the resonance frequency, and the parallel resonance (on the power-receiving side coil side). It seems to adjust to the resonance frequency of only the resonance)).

International Publication WO2007 / 008646 International Publication WO2008 / 118178 International Publication WO2015 / 173850 JP 2012-182975 A Japanese Patent Laid-Open No. 10-225129 JP 2010-166893 A International Publication WO2012 / 090700

Takemura Imura, “Wireless Power Transmission by Magnetic Resonance”, 1st Edition, Morikita Publishing Co., Ltd., February 2017, pp.2-11. Takehiro Imura, Yoichi Hori, “Unified Theory of Electromagnetic Induction and Magnetic Resonance Coupling”, IEEJ Transactions D (Industrial Application Division), The Institute of Electrical Engineers of Japan, 2015, Vol.135, No.6, pp.697-710. Masakazu Ushijima, “Improved Efficiency and Robustness by Correcting Problems in Magnetic Resonance Theory”, Green Electronics, CQ Publishing Co., Ltd., 2017, Vol.19, pp.52-69.

Wireless power transmission device described in Patent Document 3 and Non-Patent Document 3, the phase information of the resonance current flowing through the resonant capacitor C ₂ to be detected in the power receiving device, it is necessary to wirelessly transmitted to the side of the power transmitting device. Accordingly, it is necessary to provide phase information transmission means for wireless transmission in both the power transmission device and the power reception device, and there is a problem that the device becomes complicated. In addition, when wireless transmission is hindered due to some environmental change such as foreign matter or dirt, or disturbance such as disturbance, there is also a problem that the resonance frequency cannot be acquired on the power transmission device side.

  On the other hand, the wireless power supply device described in Patent Document 4 detects the resonance frequency only on the power transmission device side and performs frequency adjustment, and thus the above problem does not occur. However, since the current waveform of the power supply coil is easily distorted due to the effect of the current of the power reception coil, in the actual machine, the current waveform of the power supply coil is not stable, and stable phase detection and frequency adjustment are difficult. There was a problem.

Further, depending on the combination of circuit constants such as L ₁ , L ₂ , C ₁ , C ₂ , and R _L and the coupling coefficient k ₁₂ , the series resonance point and the parallel resonance point may be close to each other. In such a case, the phase of the current of the power supply side coil at the series resonance point is greatly shifted due to the influence of the parallel resonance point. Therefore, the wireless power supply device described in Patent Literature 4 cannot adjust the frequency to the series resonance frequency. There is also a problem. This problem will be described in detail with reference to FIG.

  In view of the above, an object of the present invention is to make it possible to perform the frequency adjustment to an accurate resonance frequency regardless of the coupling coefficient and circuit constant between the power transmission side coil and the power reception side coil. An object of the present invention is to provide a wireless power transmission device capable of detecting a current flowing in a power receiving coil of a side device. Another object of the present invention is to provide a wireless power transmission device capable of stably adjusting the frequency to the resonance frequency based on the detected current flowing in the power receiving side coil of the power receiving side device, and a power receiving side current detection circuit thereof. .

The first configuration of the power receiving side current detection circuit according to the present invention includes a power transmission side coil, a waveform generator that generates an AC waveform to be passed through the power transmission side coil, and the power transmission side according to the AC waveform generated by the waveform generator. A power receiving circuit for supplying AC power to the coil; and power receiving side current detection used in a wireless power transmission device that wirelessly transmits power from the power transmitting side coil to a power receiving side coil provided in an external power receiving device A circuit,
A power transmission current waveform detecting means for generating a power transmission current detection waveform w _i1 having a waveform similar to the waveform of the power transmission current i ₁ energized from the power feeding circuit to the power transmission side coil;
The waveform of the transmission current i ₁ ⁽⁰⁾ when the power transmission side coil is energized from the power feeding circuit in a state where the power transmission side coil is not magnetically coupled to the power reception side coil (hereinafter referred to as “non-coupled state”) A reference current similarity waveform generating means for generating a reference current similarity waveform w _ref which is a similar waveform;
Differential waveform generation means for generating and outputting a power receiving side current detection waveform w _out that is a waveform of a weighted difference between the transmission current detection waveform w _i1 and the reference current similarity waveform w _ref ,
In a non-coupled state, the differential weight generation value in the differential waveform generation means or the output amplitude of the reference current similarity waveform in the reference current similarity waveform generation means so that the amplitude of the power reception side current detection waveform w _out becomes substantially zero The value is adjusted or adjustable.

According to this configuration, a waveform similar to the waveform of the received current i ₂ energized in the power receiving coil can be obtained as the reference current similar waveform w _ref output from the differential waveform generating means (formula (16a described later) ), (See 16b)). That is, as shown in equation (15a) will be described later, when the power transmission coil and the receiver coil are attached at mutual inductance M _12, the transmission current i ₁ of the waveform of the power transmission coil, the power transmission coil receiving A similar waveform of the induced current induced by the current i ₂ of the power receiving coil is superimposed on the waveform of the power transmission current i ₁ ⁽⁰⁾ flowing through the power transmitting coil when the side coil is in a non-coupled state (M ₁₂ = 0). The resulting waveform. Therefore, the non-coupled transmission current i ₁ ⁽⁰⁾ is generated by a separate circuit independent of the power transmission side coil, and the generated non-coupled transmission current i ₁ ⁽⁰⁾ is weighted with an appropriate “weighting factor”. by subtracting from the power transmitting current i ₁ and can be extracted only the waveform of induced induced current by a current i ₂ of the power receiving coil. Since the waveform of the induced current induced by the current i ₂ of the power receiving coil is in phase with the current i ₂ and the amplitude is a constant multiple (see equation (15a)), the similar waveform of the current i ₂ can be extracted. Become. At this time, the “weighting coefficient” is determined so that the amplitude of the power reception side current detection waveform w _out becomes substantially 0 in a state where the power transmission side coil is not magnetically coupled to the power reception side coil, thereby transmitting power transmission current i _1. Can completely cancel the component of the transmission current i ₁ ^{(0) in} the non-coupled state. In the non-coupled state, the coupling with the power receiving coil is irrelevant. Therefore, the “weighting factor” can be determined in advance only by the circuit on the power transmitting side, and the circuit constant can be set in advance.

Here, the “transmission current detection waveform”, “reference current similarity waveform”, and “reception-side current detection waveform” are respectively the transmission current i ₁ and the reference current (the transmission-side coil and the reception-side coil are in a non-coupled state ( ⁽ Transmission current flowing through the power transmission side coil when M ₁₂ = 0)) i ₁ ⁽⁰⁾ , power reception side current (weighted difference i ₁ −c ₀ i ₁ ⁽ transmission current i ₁ and reference current i ₁ ^{(0)) 0)} (c ₀ is a weighting factor) current) i ₃ , not the current itself, but the time-varying wave shape created by the oscillation of each current. Therefore, in addition to each current waveform itself, if the shape of the current wave is maintained, it is a waveform obtained by converting each current into a voltage, or a waveform obtained by converting each current waveform by AD conversion and quantizing. Also good. “Magnetic coupling” refers to electrical signal coupling (transformer coupling) in which the mutual inductance between one coil and the other coil is not zero. The “non-coupling state” means a state where the power transmission side coil is not magnetically coupled to the power reception side coil, that is, a state where the mutual inductance between the power transmission side coil and the power reception side coil is zero. The “difference weight value in the difference waveform generation means” means the transmission current detection waveform w _i1 and / or the reference current similarity waveform w when calculating the weighted difference between the transmission current detection waveform w _i1 and the reference current similarity waveform w _{ref. A} “weighting factor” multiplied by _ref . The difference weight value or the output amplitude value is “adjusted” means that the difference weight value or the output amplitude value is adjusted in advance by a circuit parameter or the like so that the amplitude of the power receiving side current detection waveform w _out becomes substantially zero in the non-coupled state. Means that The difference weight value or the output amplitude value is “adjustable” means that the difference weight value or the output amplitude value is set afterwards so that the amplitude of the power receiving side current detection waveform w _out becomes substantially zero in the non-coupled state. Meaning that adjustment means (for example, semi-fixed resistor R9 in FIG. 17, semi-fixed resistor R ₆ in FIG. 20, semi-fixed resistors R ₆ and R ₉ ′ in FIG. 23, etc.) that can be adjusted is provided. To do.

According to a second configuration of the power receiving side current detection circuit of the present invention, in the first configuration, the reference current similar waveform generating means is connected in parallel with the power transmitting side coil, and the power receiving side coil is a magnetic field. comprising Unbound dummy coil, which generates the reference current waveform similar w _ref as the waveform of the dummy coil current i _d is the current flowing through the dummy coil,
Said differential waveform generating means outputs a dummy coil current i _d flowing through the dummy coil, the waveform of the weighted difference current between the power current i ₁ flowing through the power transmission coil as the power-receiving-side current detection waveform w _out Is,
When the ratio L _d / L ₁ of the inductance L _d of the dummy coil to the inductance L ₁ of the coil on the power transmission side is α _d1 , the difference waveform generation means uses the dummy coil current i as the waveform of the weighted difference current. A waveform of a weighted difference current (α _d1 i _d −i ₁ ) between α _d1 times _{d and} the primary current i ₁ is output.

With this configuration, the weighted difference current (α _d1 i _d −i ₁ ) detected by the differential waveform generation means is the current i ₂ flowing in the secondary power receiving coil (the power receiving side coupled to the power transmitting coil on the secondary side). In the case where there are a plurality of coils, this is a constant multiple of each current i _2k (k = 1,..., N) flowing through each power receiving coil. Therefore, by adjusting the frequency of the AC waveform output from the waveform generator using the detected value of the secondary current detected by the weighted difference current (α _d1 i _d −i ₁ ), the accurate resonance frequency is obtained. Frequency adjustment can be performed.

Further, by using the detected value of the weighted difference current (α _d1 i _d −i ₁ ), it is possible to detect a foreign object that is close to the power transmission side coil.

A third configuration of the power reception side current detection circuit of the present invention includes an adjustment resistor connected in parallel to the power transmission side coil and the dummy coil in the second configuration,
Said differential waveform generating means, the sum of said dummy coil current is a current flowing through the dummy coil i _d and adjustment current i _aj is a current flowing through the adjustment resistor, wherein a current flowing through the power transmission coil the primary A weighted differential current (α _d1 i _d + i _aj −i ₁ ) with respect to the side current i ₁ is detected,
The adjusting resistor is adjusted so that the weighted difference current (α _d1 i _d + i _aj −i ₁ ) becomes zero in a state where the power receiving coil is not coupled to the power transmitting coil. Features.

According to this configuration, when the internal resistance of the dummy coil cannot be ignored, the weighted difference current (α _d1 i _d + i _aj −i ₁ ) is detected based on the adjustment current i _aj , so that the internal resistance of the dummy coil The detection error of the secondary current i ₂ can be canceled.

According to a fourth configuration of the power receiving side current detection circuit of the present invention, in the second configuration, the first and second configurations are formed in which a magnetic core and a winding portion where the magnetic core is linked at least once are formed. 2 and a third conductor, and
The first conductive wire is connected in series with the power transmission side coil and in parallel with the dummy coil on the input / output wiring of the power transmission side coil,
The second conductive wire includes a current transformer connected in series with the dummy coil and in parallel with the power transmission side coil on the input / output wiring of the dummy coil,
The power transmission current waveform detection means and the difference waveform generation means are configured integrally with the current transformer,
When a current flows from the dummy coil toward the winding portion of the second conductor, and a current flows from the power transmission side coil toward the winding portion of the first conductor,
The direction of the magnetic field created in the magnetic core by the current flowing in the winding part of the second conducting wire is the direction of the magnetic field created in the magnetic core by the current flowing in the winding part of the first conducting wire. So that the opposite direction
The power transmission coil is connected to the first conductor, the dummy coil is connected to the second conductor,
In addition, the number of turns of the winding portion of the second conductor is α _d1 times the number of turns of the winding portion of the first conductor.

According to this configuration, the weighted difference current (α _d1 i _d −i ₁ ) is output to the third conductor of the current transformer. By using the current transformer as the differential waveform generating means, the circuit is greatly simplified. Further, when detecting the weighted difference current (α _d1 i _d −i ₁ ) between the dummy coil current i _d and the primary side current i ₁ , the dummy coil current i _d and the primary side current i ₁ are detected. In addition, since a common magnetic core is used, it is possible to detect the weighted difference current (α _d1 i _d −i ₁ ) with extremely high accuracy.

According to a fifth configuration of the power receiving side current detection circuit of the present invention, in the third configuration, the magnetic core and the first and second winding portions where the magnetic core is linked at least once are formed. 2, 3rd and 4th conductors, and
The first conductive wire is connected in series with the power transmission side coil and in parallel with the dummy coil and the adjustment resistor on the input / output wiring of the power transmission side coil,
The second conductive wire is connected in series with the dummy coil and in parallel with the power transmission side coil and the adjustment resistor on the input / output wiring of the dummy coil,
The fourth conductor includes a current transformer connected to the adjustment resistor in series with the adjustment resistor and in parallel with the power transmission side coil and the dummy coil on the input / output wiring of the adjustment resistor,
The power transmission current waveform detection means and the difference waveform generation means are configured integrally with the current transformer,
Current flows from the dummy coil toward the winding portion of the second conductor, and current flows from the power transmission side coil toward the winding portion of the first conductor, and from the adjustment resistor When a current is passed toward the winding part of the fourth conductor,
The direction of the magnetic field created in the magnetic core by the current flowing in the winding part of the second conducting wire is the direction of the magnetic field created in the magnetic core by the current flowing in the winding part of the first conducting wire. The direction of the magnetic field created in the magnetic core by the current flowing in the winding portion of the fourth conductor is opposite to that of the fourth conductor, and the current flowing in the winding portion of the first conductor is the magnetic body. In the opposite direction to the direction of the magnetic field created in the core,
The power transmission side coil is connected to the first conductor, the dummy coil is connected to the second conductor, and the adjustment resistor is connected to the fourth conductor;
In addition, the number of turns of the winding portion of the second conductor is α _d1 times the number of turns of the winding portion of the first conductor.

According to this configuration, the weighted difference current (α _d1 i _d + i _aj −i ₁ ) is output to the third conductor of the current transformer. By using the current transformer as the differential waveform generating means, the circuit is greatly simplified. Further, when detecting the dummy coil current _{i d} and adjustment current _{i aj} be weighted difference current between the sum and the primary current _{i 1 (α d1 i d +} i aj -i 1), detection of the dummy coil current _{i d} Since the common magnetic core is used for the detection of the adjustment current i _{aj and} the detection of the primary current i ₁ , the weighted difference current (α _d1 i _d + i _aj −i ₁ ) is extremely accurate. Can be detected.
According to a sixth configuration of the power receiving side current detection circuit of the present invention, in the first configuration, the reference current similar waveform generating means is a waveform of the power transmission voltage v ₁ which is a voltage applied to the power transmission side coil. Power transmission voltage integrating means for calculating an integral waveform of the reference current and outputting it as a reference current similarity waveform w _ref ,
The difference waveform generation means is a weighted difference (γ _a w _ref −) between the reference current similarity waveform w _ref output from the transmission voltage integration means and the transmission current detection waveform w _i1 output from the transmission current waveform detection means. a waveform of γ _b w _i1 ) is output as the power receiving side current detection waveform w _out ,
In a state where the power transmission side coil is not magnetically coupled to the power reception side coil, the differential weight generation value (γ _a , γ in the differential waveform generation means is set so that the amplitude of the power reception side current detection waveform w _out becomes substantially zero. _b ), or the amplitude of the reference current similar waveform w _ref in the transmission voltage integrating means is adjusted or adjustable.

According to this configuration, the integrated waveform of the waveform of the power transmission voltage v ₁ output from the power transmission voltage integrating means is the power transmission current i ₁ ⁽⁰ when the power transmission side coil that is not magnetically coupled to the power reception side coil is energized. ⁾ (Similar to the waveform (17a) described later). Therefore, a waveform similar to the waveform of the received current i ₂ energized in the power receiving coil can be obtained from the power receiving side current detection waveform w _out output from the differential waveform generating means (formulas (16a) and (16b described later). )).

A first configuration of a wireless power transmission device according to the present invention includes: a power transmission side coil; a waveform generator that generates an AC waveform energized to the power transmission side coil; and the power transmission side coil according to the AC waveform generated by the waveform generator A power supply circuit that feeds AC power to the wireless power transmission device that wirelessly transmits power from the power transmission side coil to a power reception side coil provided in an external power reception device,
A power-receiving-side current detection circuit configured as described in any one of 1 to 6;
Power transmission frequency adjusting means for adjusting the frequency of the power supply voltage so that the phase difference between the phase of the power reception side current detection waveform and the phase of the power supply voltage supplied to the power transmission side coil is in phase. .

According to this configuration, by adjusting the frequency of the AC waveform generated by the waveform generator so that the phase difference between the phase of the reference current similar waveform w _{ref and} the phase of the AC waveform generated by the waveform generator is in phase. The frequency can be accurately adjusted to the series resonance frequency.

According to a second configuration of the wireless power transmission device of the present invention, in the first configuration including the power receiving side current detection circuit of any one of the configurations 1 to 5, the power transmission frequency adjusting unit includes:
The weight difference current (α _{d1 i} d _{-i 1)} and the phase difference [Delta] [phi _2d between the dummy coil current i _d, or the weighted difference current output by said differential waveform generating means for outputting said difference waveform generating means ( phase difference detecting means for detecting a phase difference Δφ ₂₀ between the waveform of α _d1 i _d −i ₁ ) and the AC waveform generated by the waveform generator;
Based on the phase difference detected by the phase difference detection means,
The phase difference Δφ _2d between the weighted difference current and the dummy coil current is 90 degrees.
Alternatively, the phase difference Δφ ₂₀ between the weighted difference current and the AC waveform generated by the waveform generator is 0 degree.
Frequency control means for controlling the frequency of the AC waveform generated by the waveform generator.

According to this configuration, when the phase difference detection unit detects the phase difference Δφ _2d between the weighted difference current (α _d1 i _d −i ₁ ) and the dummy coil current i _d , at the series resonance point, Regardless of the circuit constant on the secondary side and the coupling coefficient between the power transmission side coil and the power reception side coil, the phase difference Δφ _2d is 90 degrees. Further, when the phase difference detecting means detects the phase difference Δφ ₂₀ between the weighted difference current (α _d1 i _d −i ₁ ) and the AC waveform generated by the waveform generator, at the series resonance point, 2 Regardless of the circuit constant on the next side or the coupling coefficient between the power transmission side coil and the power reception side coil, the phase difference Δφ ₂₀ is 0 degrees. Therefore, by controlling the frequency of the AC waveform generated by the waveform generator so that the phase difference Δφ _2d becomes 90 degrees or the phase difference Δφ ₂₀ becomes 0 degrees by the frequency control means, the power receiving side Regardless of the circuit constants of the device and the coupling coefficient between the power transmission side coil and the power reception side coil, it is possible to accurately adjust the frequency to the resonance frequency.

Here, at the parallel resonance point, the primary current i ₁ is minimal and causes a phase shift of 180 degrees before and after the parallel resonance point, whereas the secondary current i ₂ is at the parallel resonance point. Neither amplitude nor phase is affected by resonance, and no phase shift of 180 degrees occurs. Therefore, when the series resonance point approaches the parallel resonance point due to the combination of the coupling coefficient between the power transmission side coil and the power reception side coil and the circuit constant of each part, the primary side current i _{1 at the} series resonance point is affected by the parallel resonance. As a result, a phase shift occurs, and the phase φ ₁ of the primary current i ₁ with respect to the phase of the power supply voltage at the series resonance point deviates from 0 degrees. On the other hand, since the secondary current i ₂ is not affected by the parallel resonance, the phase φ ₂ of the secondary current i ₂ with respect to the phase of the power supply voltage at the series resonance point does not move from 0 degree. Since the weighted difference current (α _d1 i _d −i ₁ ) detected by the differential waveform generation means is proportional to the secondary current i ₂ , the weighted difference current (α _d1 i _d −i ₁ ) with respect to the phase of the power supply voltage. The phase of is a φ ₂ , and even when the series resonance point approaches the parallel resonance point, the phase of the weighted difference current (α _d1 i _d −i ₁ ) does not move from 0 degrees. Therefore, it is possible to accurately adjust the frequency to the resonance frequency regardless of the circuit constant of the power receiving side device, the coupling coefficient between the power transmitting side coil and the power receiving side coil, or the circuit constant.

  In the present invention, when “the phase difference is 0 degree”, it means that the phase difference is congruent with 0 degree modulo 180 degrees. Further, when “the phase difference is 90 degrees”, it means that the phase difference is congruent with 90 degrees by modulo 180 degrees.

According to a third configuration of the wireless power transmission device of the present invention, in the first configuration including the power receiving side current detection circuit of any one of the configurations 1 to 5, the power transmission circuit is provided in parallel. A primary side capacitor connected in series with the power transmission side coil and the dummy coil, between the connected power transmission side coil and the dummy coil, and the power feeding circuit;
Phase difference detection means for detecting a phase difference Δφ ₂₀ between the weighted difference current detected by the difference waveform generation means and the AC waveform generated by the waveform generator;
Based on the phase difference detected by the phase difference detection means,
The phase difference Δφ ₂₀ between the weighted difference current and the AC waveform generated by the waveform generator is 0 degree.
Frequency control means for controlling the frequency of the AC waveform generated by the waveform generator.

According to this configuration, the phase difference Δφ ₂₀ between the weighted difference current (α _d1 i _d −i ₁ ) detected by the phase difference detection means and the AC waveform generated by the waveform generator is the same at the series resonance point. Regardless of the circuit constants on the secondary side and the coupling coefficient between the power transmission side coil and the power reception side coil, it is 0 degrees. Therefore, by controlling the frequency of the AC waveform generated by the waveform generator so that the phase difference Δφ ₂₀ becomes 0 degrees by the frequency control means, the circuit constants of the power receiving side device and the power transmitting side coil and the power receiving side coil It is possible to perform frequency adjustment to an accurate resonance frequency regardless of the coupling coefficient.

  As described above, according to the present invention, in order to enable frequency adjustment to an accurate resonance frequency regardless of the coupling coefficient and circuit constant between the power transmission side coil and the power reception side coil, It is possible to provide a wireless power transmission device capable of detecting a current flowing in a power receiving side coil of a power receiving side device. In addition, it is possible to provide a wireless power transmission device that can stably adjust the frequency to the resonance frequency based on the detected current flowing in the power receiving coil of the power receiving device.

It is a figure explaining the detection principle of the receiving side coil (secondary coil) electric current of the wireless power transmission apparatus which concerns on this invention. It is a figure which shows the wireless electric power feeding circuit of NS type | mold magnetic resonance coupling system which added the dummy coil. It is a figure which shows the frequency change of each part current in the circuit of FIG. (A) current amplitude of each current, (b) a phase with respect to the power supply voltage _{v 0} of each current. It is a figure showing the basic composition of the wireless power transmission device concerning Example 1 of the present invention. It is a figure showing the basic composition of the wireless power transmission apparatus which concerns on Example 2 of this invention. It is a figure which shows the frequency change of each part current in the circuit of FIG. (A) current amplitude of each current, (b) a phase with respect to the power supply voltage _{v 0} of each current. It is a figure showing the basic composition of the wireless power transmission apparatus which concerns on Example 3 of this invention. It is a figure which shows the frequency change of each part current in the circuit of FIG. (A) current amplitude of each current, (b) a phase with respect to the power supply voltage _{v 0} of each current. It is a figure which shows the frequency change of each part current in the circuit of FIG. (A) current amplitude of each current, (b) a phase with respect to the power supply voltage _{v 0} of each current. It is a figure showing the basic composition of the wireless power transmission apparatus which concerns on Example 4 of this invention. It is a figure which shows the frequency change of each part current in the circuit of FIG. (A) current amplitude of each current, (b) a phase with respect to the power supply voltage _{v 0} of each current. It is a figure which shows the frequency change of each part current in the circuit of FIG. (A) current amplitude of each current, (b) a phase with respect to the power supply voltage _{v 0} of each current. It is a figure explaining general series resonance in case there is no primary side capacitor. It is a figure explaining general series resonance in case there is primary side capacitor. It is a figure showing the basic composition of the wireless power transmission apparatus which concerns on Example 5 of this invention. It is a figure showing the structure of current transformer CT1 of FIG. It is a figure showing the basic composition of the wireless power transmission apparatus which concerns on Example 6 of this invention. It is a figure showing the structure of current transformer CT1 of FIG. It is a figure showing the basic composition of the generalized wireless power transmission system. It is a figure showing the basic composition of the wireless power transmission apparatus which concerns on Example 7 of this invention. FIG. 21 is a diagram illustrating frequency changes in amplitude and phase of primary side current i ₁ , secondary side current i ₂ , and power reception side current detection voltage v _out in the wireless power transmission device of FIG. 20. Is a graph showing a frequency change of the power receiving side current detection voltage _{v out} and the secondary current _{i 2} ratio _v out / _{i 2} amplitude and phase. It is a figure showing the other example of the basic composition of the wireless power transmission apparatus which concerns on Example 7 of this invention. It is a figure which shows the example of the power transmission voltage detection circuit 10a. It is a principle figure of the wireless power transmission of a typical magnetic field resonance system. The schematic diagram which shows the state of the electric current and magnetic field in a parallel resonance state (anti-resonance state) and a series resonance state (magnetic field phase coupling state).

  Hereinafter, embodiments for carrying out the present invention will be described with reference to the drawings.

(1) Detection Principle of Power-receiving-side Coil (Secondary Coil) Current by Dummy Coil Method FIG. 1 is a diagram for explaining the principle of detection of power-receiving-side coil (secondary coil) current of the wireless power transmission device according to the present invention. . The method of detecting the power receiving coil (secondary coil) current according to the circuit topology of FIG. 1 is hereinafter referred to as “dummy coil method”. In FIG. 1, “primary side” represents the wireless power transmission device (power transmission side device) side, and “secondary side” represents the power reception device side. The wireless power transmission apparatus includes a dummy coil L ₃ connected in parallel to the power transmission coil L _1. r ₁ and r ₃ are internal resistances of the power transmission side coil L ₁ and the dummy coil L ₃ , respectively. The input voltage v ₀ of the AC power supply is applied to both ends of the power transmission side coil L ₁ and the dummy coil L ₃ . At this time, the current flowing to the power transmission coil _{L 1 i 1,} a current flowing through the dummy coil _{L 3} and _{i 3.} On the other hand, it is assumed that the power receiving device side includes N (N ≧ 1) power receiving side coils L ₂₁ ,..., L _2N that are magnetically coupled to the power transmitting side coil L ₁ . The dummy coil _{L 3,} each power receiving coil _L 21, _..., unbound magnetically to the _{L 2N.} Normally, each of the power receiving coils L ₂₁ ,..., L _2N is wound around the common magnetic core in the same direction coaxially, or wound around the central axis in the same direction so that the central axis is coaxial. Arranged closely. Here, the number N of power receiving coils may be any number. The coupling coefficients between the power receiving coils L ₂₁ ,..., L _2N and the power transmitting coils L ₁ are k ₂₁ ,..., K _2N , respectively, and the mutual inductance is M _2i = k _2i √ (L ₁ L _2i ). The currents induced in the power receiving coils L ₂₁ ,..., L _2N when the current i ₁ flows through the power transmitting coil L ₁ are i _{21 ,.} At this time, the relationship between the input voltage v ₀ , the primary coil current i ₁ , and the dummy coil current i ₃ is as follows.

Therefore,

The relationship is obtained. Here, it is assumed that the internal resistances r ₁ and r ₃ of the coils L ₁ and L ₃ are sufficiently small with respect to the total mutual inductance jωM ₁₂ and can be ignored.

In formula (3a), the left side represents the weighted sum current of the secondary currents i ₂₁ ,..., I _2N , and the right side represents the weighted difference current (α ₃₁ i between the dummy coil current i ₃ and the primary coil current i _{1. 3−} i ₁ ) represents a real number multiple. Here, the weighting factor alpha ₃₁ is the ratio of the inductance L ₁ of the power transmission coil of the inductance L ₃ of the dummy coil, the weighting factor beta _k each for mutual inductance sum Σ _{k = 1} ^N M _12k of the total receiver coil it is the ratio of the mutual inductance _{M 12k} of the power receiving coil _{L 2k.} From the equation (3a), the weighted difference current (α ₃₁ i ₃ −i ₁ ) between the dummy coil current i ₃ and the primary coil current i ₁ is the weighted sum current of the secondary side currents i _{21 ,.} Therefore, by detecting the weighted difference current (α ₃₁ i ₃ −i ₁ ), it can be seen that the weighted sum current of the secondary currents i ₂₁ ,..., I _2N on the power receiving side can be detected. In particular, when N = 1 (for example, in the case of FIGS. 25A to 25D), the induced current i ₂ itself induced in the single power receiving coil L ₂ can be directly detected.

(2) Principle of power supply frequency adjustment in the dummy coil system In FIG. 1, assuming that the internal resistances r ₁ and r ₃ of the coil are small enough to be ignored, the dummy coil current i _{3 is obtained} from the equations (2a) and (2b). Is expressed as:

Here, as a simple example, consider an NS magnetic resonance coupling type wireless power feeding circuit with a dummy coil added as shown in FIG. In FIG. 2, v ₀ is an AC voltage source, r ₀ is a power source internal resistance, L ₁ is a power transmission side coil, L ₃ is a dummy coil, L ₂ is a power reception side coil, C ₂ is a resonance capacitor, and R _L is a load resistance. Represents. The load resistance _RL is a resistance of a circuit (for example, a rectifier circuit using a diode bridge) that extracts power on the power receiving side. A mutual inductance between the power transmission side coil L ₁ and the power reception side coil L ₂ is M _12, and currents flowing through the power transmission side coil L ₁ and the power reception side coil L ₂ are i ₁ and i ₂ . At this time, the currents i ₁ and i ₂ are expressed as follows.

Further, the power transmission side primary input impedance Z _in1 viewed from both sides terminals of the coil L ₁ is as follows.

At the series resonance point, the primary side input impedance Z _in1 is minimum, that is, the reactance part of the molecule (the part where R _L = 0 of the molecule) is 0, and therefore the series resonance frequency ω ₀ is expressed by the equation (1b). It is expressed in Therefore, at the series resonance point ω = ω ₀ , the currents i ₁ and i ₂ are as follows. Here, Δ ₀ = Δ (ω ₀ ), r ₀ << ω ₀ L ₁ , r ₀ << ω ₀ L ₃ .

By substituting the equations (7a) and (7b) into the equation (4), the dummy coil current i ₃ becomes as follows.

Accordingly, the dummy coil current i ₃ is always in a state shifted by π / 2 with respect to the secondary side current i ₂ .

For even wireless power feeding circuit other than the N-S magnetic resonant coupling system, the input impedance Z _in1 viewed from both ends terminal at the series resonance point primary coil (transmitting coil) L ₁ is a minimum (i.e., the molecular The portion where R _L = 0 is 0). Therefore, from the series resonance condition Z _in1 (R _L = 0) = 0, generally, the dummy coil current i _d flowing through the dummy coil L _d connected in parallel to the primary coil L ₁ is
(I) state in the absence primary capacitor C ₁ is shifted in phase by [pi / 2 with respect to the secondary current i ₂ at the series resonance point,
(Ii) if there is a primary-side capacitor _{C 1} is always in a state of phase with the secondary current _{i 2.} Therefore, in the absence of the primary side capacitor C ₁ , the AC power supply v ₀ is set so that the phase of the secondary side current i ₂ is shifted by π / 2 with respect to the phase of the dummy coil current i _d . By controlling the angular frequency ω, the angular frequency ω of the AC power supply v ₀ can be accurately adjusted to the series resonance frequency ω ₀ . The case where the primary capacitor C ₁ is provided will be described in detail in the second and fourth embodiments.

FIG. 3 is a diagram showing a frequency change of each part current in the circuit of FIG. 3 (a) is current amplitude, FIG. 3 (b) shows the phase with respect to the power supply voltage v ₀ of each current. The circuit constants are calculated as L ₁ = 100 mH, L ₂ = 100 mH, k ₁₂ = 0.8, L ₃ = 500 mH, C ₂ = 2.53 nF, R _L = 50Ω, r ₀ = 1Ω. In FIG. 3, _{f 2} denotes a parallel resonance frequency, _{f 0} is the series resonance frequency. From FIG. 3, since the amplitude value of (α ₃₁ i ₃ -i ₁ ) / i ₂ is constant, the difference current (α ₃₁ i ₃ -i ₁ ) is proportional to the secondary current i ₂ and (α ₃₁ Since the phase of i ₃ -i ₁ ) / i ₂ is always 0 degree, it can be seen that the phases of both are also in agreement. Further, in the vicinity of the series resonance point f ₀ , the currents i ₁ and i ₂ cause a phase transition of −180 degrees, so that the power supply voltage is applied at the series resonance point f ₀ at the center of the phase transition. Phases φ ₁ and φ ₂ of currents i ₁ and i ₂ with respect to phase φ _{0 of} v ₀ are congruent modulo 180 degrees, that is, phases φ ₀ , φ ₁ , and φ ₂ are in phase. On the other hand, since the dummy coil 4 does not participate in this series resonance, the phase φ ₃ of the dummy coil current i ₃ is constant in the vicinity of the series resonance point f ₀ , and with respect to the phase φ ₀ of the power supply voltage v _0. The phase difference of 90 degrees is maintained. Therefore, at the series resonance point, the currents i ₁ and i ₂ and the difference current (α ₃₁ i ₃ -i ₁ ) are in phase, and the phase difference between the currents i ₃ and (α ₃₁ i ₃ -i ₁ ) is 90 degrees. It turns out that it becomes.

(3) Basic Configuration of Power Supply Frequency Control by Dummy Coil System FIG. 4 is a diagram showing the basic configuration of the wireless power transmission device according to the present invention. In FIG. 4, a wireless power transmission device 1 (hereinafter “power transmission device 1”) according to the present invention is a device that performs wireless power transmission to the power reception device 2. The power transmission device 1 includes a power transmission side coil 3, a dummy coil 4, a waveform generator 5, a power feeding circuit 6, a differential waveform generation unit 7, a phase difference detection unit 8, and a frequency control unit 9. The power receiving device 2 includes a power receiving side coil 11, a resonance capacitor 12, and a load resistor 13.

The power transmission side coil 3 is a coil for power transmission, and is disposed so as to face the power reception side coil 11 and enable magnetic field coupling. The dummy coil 4 is connected in parallel to the power transmission side coil 3. Even when the dummy coil 4 and the power receiving side coil 11 are opposed to the power transmitting side coil 3, the magnetic field coupling with the power receiving side coil 11 is not performed. Here, the inductances of the power transmission side coil 3, the power reception side coil 11, and the dummy coil 4 are denoted as L ₁ , L ₂ , and L ₃ , respectively, and the mutual inductance between the power transmission side coil 3 and the power reception side coil 11 is M _12. = K ₁₂ √ (L ₁ L ₂ ) (k ₁₂ is a coupling coefficient between the power transmission side coil 3 and the power reception side coil 11). The waveform generator 5 is a circuit that generates an AC waveform for energizing the power transmission side coil 3. The power supply circuit 6 is a power supply circuit that supplies AC power to the power transmission coil according to the AC waveform generated by the waveform generator 5. Here, the AC voltage output from the power feeding circuit 6 is _denoted as v ₀ , and the internal resistance of the power feeding circuit 6 is denoted as r ₀ . The differential waveform generation means 7 is a difference current (α ₃₁ i ₃ −i ₁ ) between a primary current i ₁ that is a current flowing through the power transmission side coil 3 and a dummy coil current i ₃ that is a current flowing through the dummy coil 4. Is detected. Here, the weighting coefficient α ₃₁ is α ₃₁ = L ₃ / L ₁ . Differential waveform generating means 7 functionally includes a current sensor 7a for detecting the primary current _{i 1,} a current sensor 7b for detecting the dummy coil current _{i 3,} and the difference current (α ₃₁ i ₃ -i ₁₎ The differential current calculation circuit 7c is configured to calculate. The difference current calculation circuit 7c may be configured as an analog circuit or a digital circuit. In addition, as will be described later, if a current transformer is used, the current sensors 7a and 7b and the difference current calculation circuit 7c can be integrally configured by one current transformer. The phase difference detection means 8 is a circuit that detects the phase difference Δφ between the difference current (α ₃₁ i ₃ −i ₁ ) detected by the difference waveform generation means 7 and the dummy coil current i ₃ . The frequency control unit 9 generates the waveform generator so that the phase difference Δφ between the difference current (α ₃₁ i ₃ -i ₁ ) output from the phase difference detection unit 8 and the dummy coil current i ₃ is 90 degrees. This circuit controls the frequency of the AC waveform.

The power reception side coil 11 is a coil that is magnetically coupled to the power transmission side coil 3 and receives power from the power transmission side coil 3. The resonant capacitor 12 and the load resistor 13 are connected in series with the power receiving side coil 11. The capacitance of the resonant capacitor 12 is C ₂ , and the resistance of the load resistor 13 is R _L.

In FIG. 1, the waveform generator 5 generates an AC waveform having a basic angular frequency ω and outputs it to the power feeding circuit 6. The power feeding circuit 6 generates an AC voltage v ₀ having a basic angular frequency ω based on the AC waveform and applies it to the power transmission side coil 3 and the dummy coil 4. Here, the waveform of the AC voltage v ₀ is preferably a sine wave from the viewpoint of power transmission efficiency, but may be a rectangular wave in order to facilitate the actual circuit configuration. Thereby, the primary side current i ₁ and the dummy coil current i ₃ flow through the power transmission side coil 3 and the dummy coil 4, respectively. Then, when the primary current i ₁ flows in the power transmission side coil 3, the secondary side current i ₂ is induced in the power reception side coil 11. The currents i ₁ , i ₂ and i _{3 at} this time are as shown in the equations (5a), (5a) and (4), respectively. Here, if the angular frequency ω of the AC voltage v ₀ is the series resonance angular frequency ω ₀ of the equation (1b), the input impedance Z _in1 viewed from both terminals of the power transmission side coil 3 is minimized, and the current i ₁ , I ₂ are in the same phase state (magnetic field phase coupling state), and the phase of the dummy coil current i ₃ is shifted by π / 2 with respect to the phase of the secondary current i ₂ .

The differential waveform generation means 7 detects the primary side current i ₁ , the dummy coil current i _3, and the difference current (α ₃₁ i ₃ −i ₁ ). This difference current (α ₃₁ i ₃ −i ₁ ) takes a value that is a real number multiple of the secondary current i ₂ , as shown in the equation (3a). That is, the phase of the difference current (α ₃₁ i ₃ −i ₁ ) matches the phase of the secondary current i ₂ .

The phase difference detection means 8 has a phase of the difference current (α ₃₁ i ₃ −i ₁ ) (ie, a phase of the secondary current i ₂ ) φ ₂ and a phase of the dummy coil current i ₃ detected by the current sensor 7 b. It calculates the difference (φ ₂ -φ ₃₎ and phi _3, and outputs a phase difference detection signal proportional to the phase difference (φ ₂ -φ _3).

The frequency control unit 9 feedback-controls the angular frequency ω of the AC waveform generated by the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection unit 8 corresponds to 90 degrees. As a result, the AC waveform output from the waveform generator 5 is adjusted to the series resonance angular frequency ω ₀ .

Here, the phase difference detection means 8 is the phase difference between the phase φ _{2 of the} difference current (α ₃₁ i ₃ −i ₁ ) detected by the difference waveform generation means 7 and the phase φ ₃ of the dummy coil current i ₃ ( (φ ₂ −φ ₃ ) is detected, but the phase difference detection means 8 detects the phase φ _{2 of the} difference current (α ₃₁ i ₃ −i ₁ ) detected by the difference waveform generation means 7 and the waveform generator 5. The phase difference (φ ₂ −φ ₀ ) with respect to the phase φ _{0 of the} AC waveform generated by can be detected. In this case, the frequency control unit 9 may control the frequency of the AC waveform generated by the waveform generator 5 so that the phase difference detected by the phase difference detection unit 8 becomes 0 degrees.

Further, the phase difference detection means 8 sets the phase φ _{2 of the} difference current (α ₃₁ i ₃ -i ₁ ) detected by the difference waveform generation means 7 and the phase φ _{0 of the} AC waveform generated by the waveform generator 5 to 90 degrees. The phase difference (φ ₂ −φ ₀ ± 90 °) of the shifted phase φ ₀ ± 90 ° can also be detected. In this case, the frequency control unit 9 may control the frequency of the AC waveform generated by the waveform generator 5 so that the phase difference detected by the phase difference detection unit 8 is 90 degrees.

FIG. 5 is a diagram illustrating a basic configuration of a wireless power transmission device according to the second embodiment of the present invention. In FIG. 5, the same components as those in FIG. 4 are denoted by the same reference numerals. This embodiment is different from the first embodiment (FIG. 4) in that a primary side capacitor 14 for phase adjustment is added to a circuit on the primary side (power transmission side). The capacitance of the primary side capacitor 14 is a _{C 1.} In this case, the parallel resonance angular frequency ω ₂ and the series resonance angular frequency ω ₀ are as follows.

Therefore, it can be seen that two parallel resonance points and two series resonance points occur on the low frequency side and the high frequency side. FIG. 6 is a diagram showing a frequency change of each part current in the circuit of FIG. FIG. 6A shows the current amplitude of each current, and FIG. 6B shows the phase of each current with respect to the power supply voltage v ₀ . In FIG. 6, the circuit constants are L ₁ = 100 mH, L ₂ = 100 mH, k ₁₂ = 0.8, L ₃ = 500 mH, C ₁ = 50 pF, C ₂ = 2.53 nF, R _L = 50Ω, r ₀ = Calculated as 1Ω. In FIG. 6, f ₂ ⁽ⁱ¹⁾ and f ₂ ⁽ⁱ³⁾ are parallel resonance frequencies, f _0L is a series resonance frequency on the low frequency side, and f _0H is a series resonance frequency on the high frequency side. From FIG. 6, since the amplitude value of (α ₃₁ i ₃ -i ₁ ) / i ₂ is constant, the difference current (α ₃₁ i ₃ -i ₁ ) is proportional to the secondary current i ₂ and (α ₃₁ Since the phase of i ₃ -i ₁ ) / i ₂ is always 0 degree, it can be seen that the phases of both are also in agreement. Also, the series resonance point _{f 0L,} since the current _i 1, _{i 2} is caused a phase transition of 180 ° at the vicinity of _{f 0H,} this phase is in the center of the transition series resonance point _{f 0L,} is In _{f 0H} The phases φ ₁ and φ ₂ of the currents i ₁ and i ₂ with respect to the phase φ ₀ of the power supply voltage v ₀ are congruent modulo 180 degrees, that is, the phases φ ₀ , φ ₁ , and φ ₂ are in phase. On the other hand, when the primary side capacitor C ₁ is added, the primary side capacitor C ₁ is also involved in this series resonance, so that the phase of the dummy coil current i ₃ is also in the vicinity of the series resonance points f _0L and f _0H . In this case, the currents i ₁ and i ₂ cause a phase transition of 180 degrees. Accordingly, at the series resonance points f _0L and f _0H at the center of this phase transition, the phase φ ₃ of the current i ₃ with respect to the phase φ ₀ of the power supply voltage v ₀ is congruent modulo 180 degrees, ie, the phase φ ₀ and φ ₃ are in phase.

Therefore, in the present embodiment, the phase difference detection means 8 has the difference between the phase φ _{2 of the} difference current (α ₃₁ i ₃ −i ₁ ) detected by the difference waveform generation means 7 and the phase φ ₀ of the power supply voltage v ₀ . The frequency control unit 9 is configured to detect the phase difference, and the frequency control unit 9 generates an angle of the AC waveform generated by the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection unit 8 corresponds to 0 degree. The frequency ω is feedback controlled. Alternatively, the phase difference detection means 8 has a phase φ _{2 of the} difference current (α ₃₁ i ₃ −i ₁ ) detected by the difference waveform generation means 7 and a phase shifted by 90 degrees from the phase φ ₀ of the power supply voltage v ₀ . The phase difference is detected, and the frequency control unit 9 feeds back the angular frequency ω of the AC waveform generated by the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection unit 8 corresponds to 90 degrees. Control. As a result, the AC waveform output from the waveform generator 5 is adjusted to the series resonance angular frequency ω ₀ .

FIG. 7 is a diagram illustrating a basic configuration of a wireless power transmission device according to the third embodiment of the present invention. In FIG. 7, the same components as those in FIG. In Example 1 (FIG. 4), the secondary power receiving side coil 11 is separated into a power receiving side coil 11a for resonance that causes resonance with the primary side and a power receiving side load coil 11b that extracts power. Is different. The secondary side capacitor 12 is connected to both ends of the resonance power receiving side coil 11 a, and an LC resonance circuit is formed by the resonance power receiving side coil 11 a and the secondary side capacitor 12. The resistor r ₂ represents the internal resistance of this LC resonance circuit. On the other hand, load resistors 13 are connected to both ends of the power receiving side load coil 11b. The inductances of the resonance power receiving side coil 11a and the power receiving side load coil 11b are L ₂ and L ₃ , respectively. The inductance of the dummy coil 4 and _{L 4.} The mutual inductance between the power transmission side coil 3 and the resonance power reception side coil 11a is M ₁₂ = k ₁₂ √ (L ₁ L ₂ ), and the mutual inductance between the resonance power reception side coil 11a and the power reception side load coil 11b is M ₁₃ = k. Let ₁₃ √ (L ₂ L ₃ ) and the mutual inductance of the power receiving side load coil 11b and the power transmitting side coil 3 be M ₃₁ = k ₃₁ √ (L ₃ L ₁ ). In this case, the parallel resonance angular frequency ω ₂ and the series resonance angular frequency ω ₀ are as follows.

8 and 9 are diagrams showing the frequency change of each part current in the circuit of FIG. FIG. 8 (a), the FIG. 9 (a) current amplitude of each current, FIG. 8 (b), the FIG. 9 (b) represents the phase with respect to the power supply voltage _{v 0} of each current. In FIG. 8, the circuit constants are L ₁ = 100 mH, L ₂ = 100 mH, L ₃ = 100 mH, k ₁₂ = 0.8, k ₂₃ = 0.8, k ₃₁ = 0.8, L ₄ = 500 mH, The calculation is performed with C ₂ = 2.53 nF, R _L = 50Ω, and r ₀ = r ₂ = 1Ω. In FIG. 9, the circuit constants are L ₁ = 46 μH, L ₂ = 50 μH, L ₃ = 5 μH, k ₁₂ = 0.8, k ₂₃ = 0.8, k ₃₁ = 0.8, L ₄ = 460 μH, The calculation is performed with C ₂ = 66 nF, R _L = 50Ω, and r ₀ = r ₂ = 1Ω. Further, FIG. 8, in FIG. 9, _{f 2} denotes a parallel resonance frequency, _{f 0} is the series resonance frequency. From FIG. 8, since the amplitude value of (α ₄₁ i ₄ −i ₁ ) / (β ₂ i ₂ + β ₃ i ₃ ) is constant, the difference current (α ₄₁ i ₄ −i ₁ ) is the secondary current i. _Since the phase of (α ₄₁ i ₄ −i ₁ ) / (β ₂ i ₂ + β ₃ i ₃ ) is always 0 degree, it can be seen that the phases of both are also in agreement. Further, to produce a phase shift of the current i _1, i ₂ is -180 degrees at the vicinity of the series resonance point f _0, is at the series resonance point f ₀ in the center of the phase transition, the power supply voltage v ₀ The phases φ ₁ and φ ₂ of the currents i ₁ and i ₂ with respect to the phase φ ₀ are congruent modulo 180 degrees, that is, the phases φ ₀ , φ ₁ and φ ₂ are in phase. On the other hand, since the dummy coil 4 does not participate in this series resonance, the phase φ ₃ of the dummy coil current i ₃ is constant in the vicinity of the series resonance point f ₀ , and with respect to the phase φ ₀ of the power supply voltage v _0. The phase difference of 90 degrees is maintained. Therefore, at the series resonance point, the currents i ₁ and i ₂ and the difference current (α ₄₁ i ₄ −i ₁ ) are in phase, and the phase difference between the currents i ₄ and (α ₄₁ i ₄ −i ₁ ) is 90 degrees. It turns out that it becomes.

Therefore, in the present embodiment, the frequency control means 9 uses the angular frequency ω of the AC waveform generated by the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection means 8 corresponds to 90 degrees. Feedback control. As a result, the AC waveform output from the waveform generator 5 is adjusted to the series resonance angular frequency ω ₀ .

Further, the phase difference detection means 8 is a position between the phase φ _{2 of the} difference current (α ₄₁ i ₄ −i ₁ ) detected by the difference waveform generation means 7 and the phase φ _{0 of the} AC waveform generated by the waveform generator 5. It can also be configured to detect the phase difference (φ ₂ −φ ₀ ). In this case, the frequency control unit 9 may control the frequency of the AC waveform generated by the waveform generator 5 so that the phase difference detected by the phase difference detection unit 8 becomes 0 degrees.

Further, the phase difference detection means 8 sets the phase φ _{2 of the} difference current (α ₄₁ i ₄ −i ₁ ) detected by the difference waveform generation means 7 and the phase φ _{0 of the} AC waveform generated by the waveform generator 5 to 90 degrees. The phase difference (φ ₂ −φ ₀ ± 90 °) of the shifted phase φ ₀ ± 90 ° may be detected. In this case, the frequency control unit 9 may control the frequency of the AC waveform generated by the waveform generator 5 so that the phase difference detected by the phase difference detection unit 8 is 90 degrees.

FIG. 10 is a diagram illustrating a basic configuration of a wireless power transmission device according to a fourth embodiment of the present invention. In FIG. 10, the same components as those in FIG. This embodiment is different from the third embodiment (FIG. 7) in that a primary side capacitor 14 for phase adjustment is added to the primary side (power transmission side) circuit. The capacitance of the primary side capacitor 14 is a _{C 1.} In this case, the parallel resonance angular frequency ω ₂ and the series resonance angular frequency ω ₀ are as follows.

Therefore, it can be seen that two parallel resonance points and two series resonance points occur on the low frequency side and the high frequency side. 11 and 12 are diagrams showing the frequency change of each part current in the circuit of FIG. FIG. 11 (a), the 12 (a) is current amplitude of each current, FIG. 11 (b), the FIG. 12 (b) represents the phase with respect to the power supply voltage _{v 0} of each current. In FIG. 11, the circuit constants are L ₁ = 100 mH, L ₂ = 100 mH, L ₃ = 100 mH, k ₁₂ = 0.8, k ₂₃ = 0.8, k ₃₁ = 0.8, L ₄ = 500 mH, The calculation is performed with C ₁ = 200 pF, C ₂ = 2.53 nF, R _L = 50Ω, and r ₀ = r ₂ = 1Ω. In FIG. 12, the circuit constants are L ₁ = 46 μH, L ₂ = 50 μH, L ₃ = 5 μH, k ₁₂ = 0.8, k ₂₃ = 0.8, k ₃₁ = 0.8, L ₄ = 460 μH, The calculation is made assuming that C ₁ = 144 nF, C ₂ = 66 nF, R _L = 50Ω, and r ₀ = r ₂ = 1Ω. 11 and 12, f ₂ ⁽ⁱ¹⁾ and f ₂ ⁽ⁱ⁴⁾ are parallel resonance frequencies, f _0L is a series resonance frequency on the low frequency side, and f _0H is a series resonance frequency on the high frequency side. From FIG. 11, since the amplitude value of (α ₄₁ i ₄ −i ₁ ) / (β ₂ i ₂ + β ₃ i ₃ ) is constant, the difference current (α ₄₁ i ₄ −i ₁ ) is the secondary current i. _Since the phase of (α ₄₁ i ₄ −i ₁ ) / (β ₂ i ₂ + β ₃ i ₃ ) is always 0 degree, it can be seen that the phases of both are also in agreement. Also, the series resonance point _{f 0L,} since the current _i 1, _{i 2} is caused a phase transition of 180 ° at the vicinity of _{f 0H,} this phase is in the center of the transition series resonance point _{f 0L,} is In _{f 0H} The phases φ ₁ and φ ₂ of the currents i ₁ and i ₂ with respect to the phase φ ₀ of the power supply voltage v ₀ are congruent modulo 180 degrees, that is, the phases φ ₀ , φ ₁ , and φ ₂ are in phase. On the other hand, if you add a primary capacitor C ₁ is near from also primary side capacitor C ₁ is involved in this series resonance, the phase also dummy coil current i _4, the series resonance point f _0L, f _0H In this case, the currents i ₁ and i ₂ cause a phase transition of 180 degrees. Therefore, at the series resonance points f _0L and f _0H at the center of the phase transition, the phase φ ₄ of the current i ₄ with respect to the phase φ ₀ of the power supply voltage v ₀ is congruent modulo 180 degrees, that is, the phase φ ₀ and φ ₄ are in phase.

Therefore, in the present embodiment, the phase difference detection means 8 has a difference between the phase φ _{2 of the} difference current (α ₄₁ i ₄ −i ₁ ) detected by the difference waveform generation means 7 and the phase φ ₀ of the power supply voltage v ₀ . The frequency control unit 9 is configured to detect the phase difference, and the frequency control unit 9 generates an angle of the AC waveform generated by the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection unit 8 corresponds to 0 degree. The frequency ω is feedback controlled. Alternatively, the phase difference detection means 8 has a phase φ _{2 of the} difference current (α ₄₁ i ₄ −i ₁ ) detected by the difference waveform generation means 7 and a phase shifted by 90 degrees from the phase φ ₀ of the power supply voltage v ₀ . The phase difference is detected, and the frequency control unit 9 feeds back the angular frequency ω of the AC waveform generated by the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection unit 8 corresponds to 90 degrees. Control. As a result, the AC waveform output from the waveform generator 5 is adjusted to the series resonance angular frequency ω ₀ .

In the case of FIG. 11, at the series resonance point f _0L , the phase of the primary-side current i ₁ does not become 0 degrees, but is greatly deviated from 80.6 degrees. _{since 0L} is close to the parallel resonance point f _2, due to shifted primary current i ₁ of the phase is large due to the influence of parallel resonance. When the phase of the primary side current i ₁ and the power supply voltage v ₀ as described in Patent Document 4 is used, the phase of the primary side current i ₁ is greatly affected by the parallel resonance. When shifted, proper frequency adjustment cannot be performed. This is because, near the parallel resonance point, the phase of the primary current i ₁ changes by 180 degrees in the opposite direction to the series resonance point, so that the phase transition cancels out when the series resonance point is close to the parallel resonance point. It is. On the other hand, in the present invention, by extracting the secondary current _{(β 2 i 2 + β 3} i 3) from the difference current _{(α 41 i 4 -i 1)} , the secondary current _{(β 2 i 2 + β 3} i ₃ ) and the phase of the dummy coil current i ₄ (or the power supply voltage v ₀ ) may be compared to adjust the power supply frequency. In the vicinity of the parallel resonance point, the 180 ° phase shift of the secondary current does not occur. Therefore, as shown in FIG. 11, even when the phase of the primary current i ₁ is greatly shifted due to the influence of parallel resonance, the phase of the difference current (α ₄₁ i ₄ −i ₁ ) is in series. It becomes ₀ at the resonance point _f0L , and the power supply frequency can be accurately adjusted to the series resonance point _f0L .

From the above analysis, in general, 1 to if there is no primary-side capacitor 14, in series resonance point omega _0, the phase phi _d of the dummy coil current i _d is secondary current i _{2 (secondary} current is more in some cases a state in which the phase phi ₂ and 90 degrees out of the formula (3a) a weighted sum of the left side of the secondary current. hereinafter the same.) If there is a primary-side capacitor 14, the series resonance point, phase phi _d of the dummy coil current i _d it is seen that the phase phi ₂ and phase secondary current i _2. This is due to the following reason.

When the primary side capacitor 14 is not provided, a portion surrounded by a dotted line in FIG. 13 is an independent series resonance circuit. Therefore, the dummy coil 4 does not participate in series resonance. Therefore, in the vicinity of the series resonance point, the phases of the primary side current i ₁ and the secondary side current i ₂ flowing through the power transmission side coil 3 transition 180 degrees, but the dummy coil 4 is not related to this resonance. , the phase of the dummy coil current i _d flowing through the dummy coil 4 is kept at a constant value without transition. Therefore, at the series resonance point ω ₀ , the phase of the primary side current i ₁ and the secondary side current i ₂ with respect to the power supply voltage v ₀ is just intermediate between the transition from +90 degrees to −90 degrees or from −90 degrees to +90 degrees. since the 0-degree, it becomes phase phi ₀ in phase with the power supply voltage v _0, while the phase phi ₃ with respect to the power supply voltage v ₀ of the dummy coil current i _d is maintained in the -90 degrees state. Therefore, the series resonance point omega _0, the dummy coil current i _d of the phase is in a state of 90 degrees with respect to the secondary current i ₂ phase.

On the other hand, when there is the primary side capacitor 14, the primary side capacitor 14 connected in series with the power transmission side coil 3 is also involved in the series resonance. Therefore, the part surrounded by the dotted line in FIG. 14 is a series resonance circuit. As a result, the dummy coil 4 also becomes a part of the series resonance circuit, and in the vicinity of the series resonance point ω ₀ , the phases φ ₁ , 1 of the primary side current i ₁ and the secondary side current i ₂ flowing through the power transmission side coil 3. φ ₂ changes by 180 degrees, and the phase φ _d of the dummy coil current i _d flowing through the dummy coil 4 also changes by 180 degrees together with the phases φ ₁ and φ _{2 of the} primary current i ₁ and the secondary current i ₂ . Therefore, at the series resonance point ω ₀ , the primary side current i ₁ and the secondary side current i _{2 with} respect to the phase φ ₀ of the power supply voltage v ₀ and the phases φ ₁ , φ ₂ , and φ _d of the dummy coil current i _d are +90 Since the transition from the degree to −90 degrees or from −90 degrees to +90 degrees is just in the middle, it is in phase with the phase φ ₀ of the power supply voltage v ₀ . Therefore, at the series resonance point ω ₀ , the phase of the dummy coil current _{id and} the phase of the secondary current i ₂ are in phase.

Therefore, when the primary side capacitor 14 is not provided, the phase difference detecting means 8 can take one of the following configurations.
(A) The phase difference (φ ₂ −φ _d ) between the phase φ _{2 of the} difference current (α _d1 i _d −i ₁ ) detected by the differential waveform generation means 7 and the phase φ _d of the dummy coil current i _d is detected. To be configured.
(B) the differential waveform phase phi ₂ and the power supply voltage _{v 0} of the phase phi ₀ to 90 degree shifted phase phi _0-position of the ± 90 ° of the differential current detected by the generating means _{7 (α d1 i d -i 1} ) The phase difference (φ ₂ −φ ₀ ± 90 °) is detected.
(C) The phase difference (φ ₂ −φ ₀ ) between the phase φ _{2 of the} difference current (α _d1 i _d −i ₁ ) detected by the difference waveform generation means 7 and the phase φ ₀ of the power supply voltage v ₀ is detected. Configure as follows.

  In the cases (a) and (b), the frequency control unit 9 generates the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection unit 8 corresponds to 90 degrees. The configuration is such that the angular frequency ω of the AC waveform is feedback-controlled. In the case of (c) above, the frequency control means 9 has a value of the phase difference detection signal output by the phase difference detection means 8 corresponding to 0 degrees. In this way, the angular frequency ω of the AC waveform generated by the waveform generator 5 may be feedback controlled.

On the other hand, when the primary side capacitor 14 is present, the phase difference detecting means 8 can adopt any one of the configurations (b) and (c). In this case, in the vicinity of the DC resonance point ω ₀ , the phase φ _d of the dummy coil current i _d causes a phase transition of 180 degrees together with the phase φ ₂ of the secondary current i _2. comparison of the phase phi ₂ and the phase phi ₃ of the dummy coil current _{i d} of the differential current (α _d1 i _d -i ₁₎ such as will not make sense. In the case of (b) above, the frequency control means 9 uses the angular frequency ω of the AC waveform generated by the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection means 8 corresponds to 90 degrees. In the case of (c) above, the frequency control means 9 causes the waveform generator 5 so that the value of the phase difference detection signal output from the phase difference detection means 8 is equivalent to 0 degrees. The angular frequency ω of the AC waveform generated by the above may be feedback-controlled.

FIG. 15 is a diagram illustrating a basic configuration of a wireless power transmission device according to the fifth embodiment of the present invention. The present embodiment is an example in which the circuit configuration of the wireless power transmission apparatus (FIG. 7) according to the third embodiment is more concrete. In FIG. 15, the same reference numerals are given to the components corresponding to the components in FIG. In the present embodiment, the current sensors 7a and 7b and the difference current calculation circuit 7c in FIG. 7 are realized by one current transformer CT1. As shown in FIG. 16, the current transformer CT <b> 1 includes a magnetic core 21 formed of an annular magnetic body, and a first conductor 22 in which a winding portion where the magnetic core 21 is linked one or more times is formed. , Second conductive wire 23 and third conductive wire 24. The terminals n1 to n6 of the current transformer CT1 illustrated in FIG. 16 correspond to the terminals n1 to n6 of the current transformer CT1 of FIG. The first conducting wire 22 is connected on the input / output wiring of the power transmission side coil 3 in series with the power transmission side coil 3 and in parallel with the dummy coil 4. The second conducting wire 23 is connected on the input / output wiring of the dummy coil 4 so as to be in series with the dummy coil 4 and in parallel with the power transmission side coil 3. The third conductor 24 is a wiring for detecting a difference current. In the current transformer CT1 of FIG. 15, the first conducting wire 22, the second conducting wire 23, and the third conducting wire 24 are all wound around the magnetic core 21 in the same direction. the ratio _Nt 2 / Nt ₁ and turns Nt ₂ turns Nt ₁ and second conductor 23 of _is the _{Nt 2 / Nt 1 = α 41} = L 4 / L 1. Here, L ₁ is the self-inductance of the power transmission side coil 3, and L ₄ is the self-inductance of the dummy coil 4. The inductances L ₁ and L ₄ are adjusted so that the ratio α ₄₁ = L ₄ / L ₁ is an integer. The connection direction between the power transmission side coil 3 and the first conductive wire 22 and the connection direction between the dummy coil 4 and the second conductive wire 23 are currents from the dummy coil 4 toward the winding portion of the second conductive wire 23. i ₄ to flow, and when a current flows i ₁ flows from the power transmission coil 3 to the winding portion of the first conductor 22, second current i ₄ flowing through the winding portions are magnetic core conductor 23 The direction of the magnetic field H ₄ created in the magnetic field 21 is connected so that the current i ₁ flowing in the winding portion of the first conductor 22 is opposite to the direction of the magnetic field H ₁ created in the magnetic core 21. Yes. Therefore, since the magnetic field in the magnetic core 21 is H ₄ −H ₁ , when the number of turns of the third conductor 24 is Nt ₅ , the current i ₅ induced in the third conductor 24 is as follows: Become.

Therefore, since the current i ₅ proportional to the difference current (α ₄₁ i ₄ −i ₁ ) is output from the third conductor 24 that is the output line, the current transformer CT 1 is the differential waveform generating means. It can be seen that it functions as 7.

Further, the dummy coil current i4 itself flowing through the dummy coil ₄ is detected by a current transformer CT2 provided separately from the current transformer CT1. The output current _{i 5} of the current transformer CT1 is a Zener diode ZD1, ZD2, is rectangular by squaring circuit composed of a capacitor C8, and a diode D1, D2. The output current _{i 6} of the current transformer CT2 are Zener diode ZD3, ZD4, it is rectangular by squaring circuit composed of the capacitor C9, and diodes D3, D4. The rectangularized output currents i ₅ and i ₆ are input to the EXOR circuit IC1, converted into a rectangular wave signal having a pulse width proportional to the phase difference Δφ ₂₄ = φ ₂ −φ ₄ , and then passed to the low-pass filter 8a. The low-pass filter 8a outputs a voltage signal s (Δφ ₂₄ ) proportional to the phase difference Δφ ₂₄ between the output currents i ₅ and i ₆ . Here, phi ₂ is the secondary current i ₂ phase, phi ₄ is the phase of the dummy coil current i _4. The voltage signal s (Δφ ₂₄ ) is input to the frequency control unit 9 (frequency control circuit), and the frequency control unit 9 sets the voltage signal s (Δφ ₂₄ ) to a value s (90 °) corresponding to 90 degrees. In addition, each output frequency ω of the waveform generator 5 is feedback-controlled.

Thus, the circuit is greatly simplified by using the current transformer CT1 as the differential waveform generating means 7. Further, when detecting the difference current (α ₄₁ i ₄ −i ₁ ) between the dummy coil current i ₄ and the primary side current i ₁ , the dummy coil current i ₄ and the primary side current i ₁ are detected. Since the common magnetic core 21 is used, it is possible to detect the difference current (α ₄₁ i ₄ −i ₁ ) with extremely high accuracy.

FIG. 17 is a diagram illustrating a basic configuration of a wireless power transmission device according to a sixth embodiment of the present invention. The wireless power transmission device 1 of the present embodiment is basically the same as the wireless power transmission device (FIG. 15) according to the fifth embodiment,
(I) The adjustment resistor 15 is added in parallel to the power transmission side coil 3 and the dummy coil 4;
(Ii) A difference is that a fourth winding (fourth conductive wire 25) is newly added to the current transformer CT1. The adjustment resistor 15 is configured by connecting a fixed resistor R10 and a semi-fixed resistor R9 in series. Then, by adjusting the resistance value of the semi-fixed resistor R9, the adjustment resistor 15 is in a state where the power receiving side coil 11 is not coupled to the power transmitting side coil 3 (a state where the power receiving side coil 11 is removed far away). The current output to the third conductor 24 of the current transformer CT1 is adjusted to be zero.

  As shown in FIG. 18, the current transformer CT <b> 1 includes a magnetic core 21 formed of an annular magnetic body, and a first conductor 22 formed with a winding portion in which the magnetic core 21 is linked one or more times. , Second conductive wire 23, third conductive wire 24, and fourth conductive wire 25. The terminals n1 to n8 of the current transformer CT1 illustrated in FIG. 16 correspond to the terminals n1 to n8 of the current transformer CT1 of FIG.

In the current transformer CT1 of FIG. 17, the first conductor 22, the second conductor 23, the third conductor 24, and the fourth conductor 25 are all wound around the magnetic core 21 in the same direction. , the number of turns Nt ₁ of the first conductor 22 the ratio _Nt 2 / Nt ₁ and turns Nt ₂ of the second conductor 23 _is a _{Nt 2 / Nt 1 = α 41} = L 4 / L 1. Here, L ₁ is the self-inductance of the power transmission side coil 3, and L ₄ is the self-inductance of the dummy coil 4. The inductances L ₁ and L ₄ are adjusted so that the ratio α ₄₁ = L ₄ / L ₁ is an integer. The connection of the first conducting wire 22, the second conducting wire 23, and the third conducting wire 24 is the same as in the fifth embodiment.

In addition, the connection direction of the adjustment resistor 15 and the fourth conductor 25 is such that a current i ₄ flows from the dummy coil 4 toward the winding portion of the second conductor 23 and the first conductor 22 is connected from the power transmission side coil 3. When the current i ₁ flows toward the winding portion and the current i _ad flows from the adjustment resistor 15 toward the winding portion of the fourth conducting wire 25, the current flowing through the winding portion of the fourth conducting wire 25 The direction of the magnetic field H _aj that i _{aj creates} in the magnetic core 21 is opposite to the direction of the magnetic field H ₁ that the current i ₁ flowing in the winding portion of the first conductor 22 _{creates in} the magnetic core 21; The current i ₄ flowing in the winding portion of the second conductive wire 23 is connected in the same direction as the direction of the magnetic field H ₄ created in the magnetic core 21. Accordingly, since the magnetic field in the magnetic core 21 is H ₄ + H _aj −H ₁ , assuming that the number of turns of the third conductor 24 is Nt ₅ and the number of turns of the fourth conductor 24 is Nt _aj , the third conductor The current i ₅ induced in 24 is as follows:

Therefore, since the current i ₅ proportional to the difference current (α ₄₁ i ₄ + i _aj ′ −i ₁ ) is output from the third conductor 24 that is the output line, the current transformer CT1 is the only difference. It can be seen that it functions as the waveform generation means 7. If Nt _aj = Nt ₁ , then i _aj ′ = i _aj . The value of i _aj is adjusted by the number of turns Nt _aj and the resistance value of the adjusting resistor 15.

In this way, the circuit is greatly simplified by using the current transformer CT1 as the differential waveform generating means. The dummy coil current _{i d} and adjustment current _{i aj} even when detecting the differential current _{(α 41 i 4 + i aj} '-i 1) of the sum and the primary current _{i 1,} the detection of the dummy coil current _{i d} Since the common magnetic core is used for the detection of the adjustment current i _{aj and} the detection of the primary current i ₁ , the difference current (α ₄₁ i ₄ + i _aj ′ −i ₁ ) with extremely high accuracy is used. Can be detected.

Further, when the internal resistance of the dummy coil cannot be ignored, the secondary current i due to the internal resistance of the dummy coil is detected by detecting the difference current (α ₄₁ i ₄ + i _aj −i ₁ ) based on the adjustment current i _{aj. Since} the detection error of ₂ can be canceled, the frequency of the AC waveform output from the waveform generator 5 can be adjusted to the series resonance frequency ω ₀ with extremely high accuracy.

In this embodiment, first, the resonance of the wireless power transmission system and the voltage / current on the power transmission side (primary side) and the power reception side (secondary side) are reconsidered. FIG. 19A is a diagram illustrating a basic configuration of a generalized wireless power transmission system. In FIG. 19 (a), normally, the internal resistance R ₁ of the primary circuit, the power transmission is designed small in order to suppress the loss can be ignored in use frequency domain. In the secondary side circuit, N (N is an integer of 1 or more) power receiving side coils L _2,1 ,..., L _{2, N} are electromagnetically coupled to the power transmitting side coil L ₁ of the primary side circuit. . Transmitting coil _{L 1} and the power receiving side coil _{L 2, k (k = 1} , ..., N) the mutual inductance _{M 12, k,} each receiver coil _{L 2, k with, L 2, l (k,} l = 1, ..., N; a k ≠ l) the mutual inductance between the _{M 2, kl = M 2,} lk. Each of the power receiving coils L _{2, k} (k = 1,..., N ⁾ is connected to a load impedance Z _{2, k} ^(Load) , and a current i _{2, k} flows therethrough. At least one of the load impedances Z _{2, k} ^(Load) (k = 1,..., N) includes a resonance capacitance. The circuit equation of the circuit of FIG. 19A is as follows.

Here, v ₁ is a voltage (primary coil voltage) applied to the power transmission side coil L ₁ , i ₁ is a current (primary coil current) flowing through the power transmission side coil L ₁ , and i _{2, k} (k = 1, ..., N) is the current flowing through each receiver coil _{L 2, k} (2 coil current), a _{Z 1} ≒ _{jωL 1.} Therefore, the primary coil current i ₁ is expressed as follows.

Here, i ₁ ⁽⁰⁾ is the primary coil current in the non-coupled state (M ₁₂ = 0), i ₂ is the load average of each secondary coil current i _{2, k} (k = 1,..., N) (Referred to as “average secondary coil current”), β _{12, k} (k = 1,..., N) is a weighting factor for each of the power receiving coils L _{2, k} , M12 is a mutual inductance M _{12, k} (k = 1). ,..., N). From equation (15a), the circuit of FIG. 19A can be represented by the equivalent circuit of FIG. Here, Z ₂ ^(Load) represents the total load impedance of the secondary circuit.

In the series resonance state (magnetic field phase coupling state), the secondary coil currents i _{2 and k} of any of the secondary side circuits k become dominant in the N secondary side circuits and become dominant. At this time, since the phase of the average secondary coil current i ₂ substantially matches the phase of the secondary coil currents i _{2 and k} , the phase of the average secondary coil current i ₂ is in phase with the phase of the primary coil voltage v _1. It becomes. Accordingly, in the primary side circuit, a similar waveform of the average secondary coil current i ₂ is extracted, and the phase of the extracted average waveform of the average secondary coil current i ₂ is in phase with the phase of the waveform of the primary coil voltage v _1. By adjusting the frequency of the primary coil voltage v _{1 in} such a manner, the frequency of the primary coil voltage v ₁ can be matched with the series resonance frequency.

In FIG. 19, the transmission current detection circuit is a circuit that converts the waveform of the primary coil current i _{1 into} the waveform of the voltage (detection voltage) v ₃ applied to the resistor R ₃ of the transmission current detection circuit using the current transformer CT. Yes, the circuit constant is set so that R ₃ << ωL ₃ in the operating frequency region. Accordingly, the voltage (detected voltage) v ₃ applied to the current (detected current) i ₃ of the transmission current detection circuit and the resistor R ₃ of the transmission current detection circuit is as follows.

From the above equation (16), the detected current i ₃ and the detected voltage v ₃ are a weighted sum of the primary coil current i ₁ ⁽⁰⁾ in the non-coupled state and the average secondary coil current i ₂ . From this, in principle, by subtracting L ₁ / M ₁₂ times the primary coil current i ₁ ⁽⁰⁾ in the non-coupled state from the detected voltage v ₃ , the waveform similarity of the average secondary coil current i ₂ is similar. It can be seen that the waveform can be extracted. The primary coil current i ₁ ⁽⁰⁾ in the uncoupled state is referred to as “reference current”.

The dummy coil system shown in the above Examples 1-6, a similar waveform of the reference current i _{1 ^(0),} using a dummy coil L _d, the current flowing through the dummy coil L _d (dummy coil current) i _d waveform By subtracting a constant α times the dummy coil current i _d from the detection current i _3, a similar waveform of the average secondary coil current i ₂ is extracted as the waveform of the difference current (α i _d −i ₃ ). is doing. At this time, the constant α is such that the difference current (α i _d −i ₃ ) is 0 in a state where M ₁₂ = 0 (i ₂ = 0) by separating the secondary side circuit from the primary side circuit sufficiently far away. It is set to be.

On the other hand, the reference current i ₁ ⁽⁰⁾ can also be expressed as follows from the first expression of Expression (15b).

From Equation (17a), an integrated voltage waveform v _{1, int} obtained by integrating the primary coil voltage v ₁ is generated by an integrating circuit to generate a similar waveform of i ₁ ⁽⁰⁾ , and the integrated voltage waveform v _{1, int} It can be seen that by subtracting the constant γ times from the detection voltage v _3, it is possible to extract a similar waveform of the waveform of the average secondary coil current i _{2 as} the waveform of the weighted difference voltage (γv _d −v ₃ ). At this time, the constant γ is a weighted difference current (γv _{1, int} −v ₃ ) in a state in which the secondary side circuit is sufficiently far away from the primary side circuit and M ₁₂ = 0 (i ₂ = 0 state). Should be set to 0. Incidentally, the equation (17a), the integrated voltage _{v 1, int} is a constant multiple of the reference current _i ^{1 (0),} since the waveform is to be similar to the waveform of the reference current _i ^{1 (0),} the integrated voltage _{v 1 , Int} are referred to as “similar voltage to the reference current”, and the waveform w _ref is referred to as “waveform similar to the reference current waveform”. Thus, a method of extracting a similar waveform of the average secondary coil current i _{2 as} the waveform of the weighted differential voltage (γv _{1, int} −v ₃ ) is referred to as a “transmission voltage integration method”. Hereinafter, in a seventh embodiment, an embodiment of a wireless power transmission device to which the transmission voltage integration method is applied will be described.

FIG. 20 is a diagram illustrating a basic configuration of a wireless power transmission device according to a seventh embodiment of the present invention. In the seventh embodiment, the NS magnetic field resonance method (see FIG. 25A) will be described as an example, but the same applies to other magnetic field resonance methods (see FIGS. 25B to 25F). Applicable in configuration. A wireless power transmission device 1 (hereinafter, “power transmission device 1”) according to the seventh embodiment includes a power transmission side coil 3, a waveform generator 5, a power feeding circuit 6, a power transmission current waveform detection unit 6a, a difference waveform generation unit 7, a phase difference. A detection unit 8, a frequency control unit 9, and a transmission voltage integration unit 10 are provided. The power receiving device 2 includes a power receiving side coil 11, a resonance capacitor 12, and a load resistor 13. In FIG. 20, the same components as those in FIG. 4 of the first embodiment are denoted by the same reference numerals. The transmission current waveform detection means 6a in the present embodiment corresponds to the current sensor 7a in FIG. 4 of the first embodiment. Further, it is assumed that the power transmission side resistance R ₁ is small enough to be ignored with respect to the inductance ωL ₁ of the power transmission side coil 3 in the use frequency region (frequency region around the series resonance angular frequency ω ₀ ).

Transmitting current waveform detecting means 6a is provided with a current transformer CT and the detection resistor R _3. In the current transformer CT, a detection coil (inductance L ₃ ) is wound around an annular magnetic core and a conducting wire through which the primary current i ₁ flows is passed through the ring of the magnetic core. Let L _{4 be} the inductance of the conductive wire passing through the magnetic core, and M _{34 be} the mutual inductance between the inductances L ₃ and L ₄ . The circuit constant is set so that R ₃ << ωL ₃ in the operating frequency range. Further, L ₄ << L ₁ is satisfied. Hereinafter, the voltage v ₃ generated between the terminals of the detection resistor R ₃ is referred to as “transmission current detection voltage”.

The transmission voltage integration means 10 has a waveform of a voltage v ₆ that is similar to the reference voltage of the integrated voltage waveform of the waveform of the transmission voltage v ₁ that is a voltage applied to the transmission coil 3 (corresponding to the reference current similarity waveform w _ref ). Is output as a circuit. In Figure 20, the transmission voltage integration means 10, the resistor _R 5, the transmission voltage detecting circuit 10a composed of _{R 6,} operational amplifier OP _7, the capacitance _{C 7,} and a resistor _R 7, the integrating circuit composed of _{R 8} 10b. Transmission voltage detecting circuit 10a, the dividing circuit for generating a transmission voltage _{v 1} obtained by dividing the divided voltage (hereinafter referred to as "transmission detection _{voltage".) V 8 = v 1 R} 6 / (R 5 + R 6) It is configured. Resistance R ₆ for adjustment is semi-fixed resistor, it is possible to amplitude adjustment of the output voltage v _8. Integrating circuit 10b integrates the waveform of the transmission detected voltage _{v 8,} and outputs a voltage _{v 6.} Here, the resistor R ₆ is a resistor for removing a DC component, and is set to satisfy ωC ₇ R ₈ >> 1 in the use frequency region. The integrated voltage v ₆ is a reference current similar voltage.

The differential waveform generating means 7 is constituted by a subtracting circuit including an operational amplifier OP ₁₁ and resistors R ₉ , R ₁₀ , R ₁₁ , R ₁₂ . The difference waveform generation means 7 is a weighted difference (γ _a v ₆ −γ _b) between the reference current similarity voltage v ₆ output from the transmission voltage integration means 10 and the transmission current detection voltage v ₃ output from the transmission current waveform detection means 6 _a. The voltage v ₃ ) is output as the power receiving side current detection voltage v _out .

The phase difference detection means 8 is the phase of the power reception side current detection voltage v _out = (γ _a v ₆ −γ _b v ₃ ) output from the differential waveform generation means 7 (that is, the phase of the secondary side current i ₂ ) φ _2. And a difference Δφ = (φ ₂ −φ ₆ ) between the reference current similarity voltage v ₆ output from the transmission voltage integrating means 10 and the phase φ _6, and a phase difference detection signal s (proportional to the phase difference Δφ) Δφ) is output. The frequency control unit 9 feedback-controls the angular frequency ω of the AC waveform generated by the waveform generator 5 so that the value of the phase difference detection signal s (Δφ) output from the phase difference detection unit 8 corresponds to 90 degrees. . As a result, the AC waveform output from the waveform generator 5 is adjusted to the series resonance angular frequency ω ₀ .

In the example of FIG. 20, the phase difference detection unit 8 includes the phase φ ₂ of the phase of the difference voltage (γ _a v ₆ −γ _b v ₃ ) detected by the difference waveform generation unit 7 and the reference current similarity voltage v _6. The phase difference Δφ = (φ ₂ −φ ₆ ) of the phase φ ₆ is detected, but the phase difference detection unit 8 detects the difference voltage (γ _a v ₆ −γ _b detected by the difference waveform generation unit 7. The phase difference Δφ = (φ ₂ −φ ₁ ) between the phase φ ₂ of the phase v ₃ ) and the phase φ _{1 of the} AC waveform generated by the waveform generator 5 can also be detected. In this case, the frequency control unit 9 may control the frequency of the AC waveform generated by the waveform generator 5 so that the phase difference detected by the phase difference detection unit 8 becomes 0 degrees.

  Details of the operation of the power transmission device 1 according to the seventh embodiment configured as described above will be described below.

(1) Power transmission / reception circuit The circuit equation of the power transmission / reception circuit configured by the power supply circuit 6, the power transmission side coil 3, the power reception side coil 11, the resonance capacitor 12, and the load resistor 13 is as follows. The internal resistance R ₁ of the primary circuit, the power transmission is designed small in order to suppress the loss can be ignored in use frequency domain.

Where v ₁ is the primary coil voltage, i ₁ is the primary current, i ₂ is the secondary current, Z ₁ is the primary impedance, Z ₂ is the secondary impedance, and X ₂ is the secondary reactance. , M ₁₂ = k ₁₂ √ (L ₁ L ₂ ) is a mutual inductance between the power transmission side coil 3 and the power reception side coil 11, and k ₁₂ is a coupling coefficient. Using Cramer's theorem, the primary current i ₁ and the secondary current i ₂ are expressed as follows.

In the series resonance state, the primary-side current i ₁ is maximized, and therefore the series resonance frequency ω _s0 of the power transmission / reception circuit is expressed as follows under the condition of Δ | _{R2 = 0} = 0.

Here, a non-bonded state (state where M ₁₂ = 0) is considered. At this time, since i ₂ = 0 and Δ = Z ₁ Z ₂ , the primary-side current i ₁ ^{(0) in the} uncoupled state is expressed as follows from the equation (19a).

(2) Transmission current waveform detection means 6a
In the circuit of the inductance L ₃ and the detection resistor R ₃ that constitute the power transmission current waveform detecting means 6a, Applying the laws of Kirchhoff, transmission current detection voltage v ₃ is expressed as follows.

(3) Transmission voltage integrating means 10
The gain A _7v of the integrating circuit of the transmission voltage integrating means 10 is expressed as follows.

Thus, the reference current similar voltage v ₆ is expressed as follows.

Here, the voltage (v ₁ / jω) is an integrated voltage of the primary coil voltage v ₁ .

(4) Difference waveform generation means 7
When R ₉ = R ₁₀ and R ₁₁ = R ₁₂ in the subtraction circuit of the differential waveform generation means 7, the power receiving side current detection voltage becomes v _out = (R ₁₁ / R ₉ ) (v ₆ −v ₃ ), and the differential voltage It is a constant multiple of (v ₆ −v ₃ ). Here, the differential voltage (v ₆ −v ₃ ) is as follows.

Here, the constant A ₇ is a circuit constant that does not depend on the angular frequency ω from the equation (24b), and can be freely adjusted by each circuit constant of the transmission voltage integrating means 10. Therefore, the difference voltage (v ₆ −v ₃ ) when the constant A ₇ is separated from the primary side circuit sufficiently far away from the primary side circuit to be in a non-coupled state (M ₁₂ = 0 state) becomes zero. To be determined. Under this condition, the constant A ₇ is determined as follows.

If Expression (26) is substituted into Expression (25), it is understood that the power receiving side current detection voltage v _out is expressed as follows.

That is, it can be seen that the waveform of the power-receiving-side current detection voltage v _out is a similar waveform that is opposite in phase to the waveform of the secondary-side current i ₂ .

(5) Frequency Characteristics FIG. 21 shows the amplitude and phase of the primary side current i ₁ , secondary side current i ₂ , and receiving side current detection voltage v _out in the wireless power transmission device of FIG. 20 (power supply voltage (primary coil voltage). FIG. 4 is a diagram illustrating a frequency change of a phase with respect to v ₁ . FIG. 22 is a diagram illustrating the frequency change of the amplitude and phase of the ratio v _out / i ₂ of the power receiving side current detection voltage v _out and the secondary side current i ₂ . 21 and 22, f _0p represents a parallel resonance frequency, and f _0s represents a series resonance frequency. In the wireless power transmission device of FIG. 20, the frequency of the primary coil voltage v ₁ output from the power feeding circuit 6 is controlled to be the series resonance frequency f _0s . In the periphery of the series resonance frequency _{f 0 s,} the amplitude of the ratio _v out / _{i 2} is a substantially constant value, the ratio _v out / _{i 2} phase is kept constant at approximately 180 degrees (reverse phase) I understand that. Therefore, it can be seen that the waveform of the secondary current i ₂ is obtained as the power receiving side current detection voltage v _out . Further, in the serial resonance frequency _{f 0 s,} the secondary current _{i 2} of the phase opposite phase to the primary coil voltage _{v 1} of the phase, the power receiving side current detection voltage _{v out} of phase primary coil voltage _{v 1} of the phase It turns out that it is in phase with respect to. From the equation (24a), the phase φ ₆ of the reference current similarity voltage v ₆ is shifted by 90 degrees with respect to the phase of the primary coil voltage v ₁ , so the frequency control means 9 is detected by the phase difference detection means 8. If the frequency of the AC waveform generated by the waveform generator 5 is controlled so that the phase difference Δφ = (φ ₂ −φ ₆ ) is 90 degrees, the frequency of the primary coil voltage v ₁ becomes the series resonance frequency f _0s . It can be seen that they can be controlled to match.

In FIG. 20, a subtraction circuit is used for the difference waveform generation means 7. However, if the connection direction of the current transformer CT is reversed, the difference waveform generation means 7 can be configured using a simpler addition circuit. It is. FIG. 23 shows the basics of the wireless power transmission apparatus according to the seventh embodiment, in which the difference waveform generating means 7 is configured by using an adder circuit composed of an operational amplifier OP ₁₁ and resistors R ₉ , R ₉ ′, R ₁₀ , R _11. The configuration is shown. By the connection direction of the current transformer CT and opposite, mutual inductance _{M 34} between the inductance _L 3, _{L 4} is changed to _{-M 34.} Accordingly, since the issue with the v ₃ in the formula (22) is reversed, by calculating the sum voltage of the voltage v ₃ and the voltage v ₆ by the adder, and the current waveform of the primary coil current i _1, uncoupled A difference waveform from the current waveform of the primary-side current i ₁ ⁽⁰⁾ in the state (M ₁₂ = 0) can be obtained. In the addition circuit of FIG. 23, a weighting factor (weight difference (γ _a w _ref −γ in the sixth configuration of the power receiving side current detection circuit of the present invention) when calculating the weighted sum of the voltage v ₃ and the voltage v ₆ is used. _{b w} difference weights of _{i1) (γ a,} to enable adjusting the equivalent) to gamma _b), using the semi-fixed resistor to the resistor R _{9 'for} adjustment connected in series to the resistor R _9.

Furthermore, in the examples of FIGS. 20 and 23, an example in which a voltage dividing circuit is used as the transmission voltage detection circuit 10a is shown. However, as the transmission voltage detection circuit 10a, besides this, a voltage detection transformer VT or an isolation amplifier is used. A circuit for isolating and detecting the transmission voltage v ₁ with an (insulation amplifier) or the like can also be used. FIG. 24 shows some examples of the transmission voltage detection circuit 10a. FIG. 24 (a) shows a portion around the transmission voltage detection circuit 10a shown in FIGS. FIGS. 24B and 24C are diagrams showing another example in which the part of FIG. 24A is replaced. FIG. 24B is an example in which a voltage detection transformer and a voltage dividing circuit are used in the transmission voltage detection circuit 10a, and FIG. 24C is an example in which a voltage dividing circuit and an isolation amplifier are used in the transmission voltage detection circuit 10a. is there. When the transmission voltage v ₁ is a high voltage, it is necessary to insulate and detect the potential between the transmission voltage v ₁ and the input voltage of the integration circuit 10b. It is better to adopt an isolation configuration as in (1).

In the present embodiment, the transmission voltage integration means 10 and the difference waveform generation means 7 are configured by analog circuits. However, the transmission detection voltage v ₈ output from the transmission voltage detection circuit 10a and the transmission current waveform detection means are shown. 6a is a transmission current detection voltage v ₃ to output AD-converted, it is also possible to configure the integration circuit 10b and the differential waveform generating means 7 by a digital circuit. It is also possible to configure using a reconfigurable logic circuit such as a microcomputer or FPGA (Field-Programmable Gate Array).

1 Wireless power transmission equipment (power transmission equipment)
2 Power receiving device 3 Power transmission side coil 4 Dummy coil 5 Waveform generator 6 Power feeding circuit 6a Power transmission current waveform detection means 7 Difference waveform generation means 7a, 7b Current sensor 7c Difference current calculation circuit 8 Phase difference detection means 9 Frequency control means 10 Transmission voltage Integration means 10a Transmission voltage detection circuit 10b Integration circuit 11 Power receiving side coil 11a Receiving side coil 11b Receiving side load coil 12 Resonant capacitor 13 Load resistor 14 Primary side capacitor 15 Adjustment resistor CT1 Current transformer 21 Magnetic core 22 First Conductor 23 Second conductor 24 Third conductor 25 Fourth conductor

Claims (9)

A power transmission side coil, a waveform generator that generates an AC waveform to be passed through the power transmission side coil, and a power supply circuit that supplies AC power to the power transmission side coil according to the AC waveform generated by the waveform generator, A power receiving side current detection circuit used in a wireless power transmission device that wirelessly transmits power to a power receiving side coil provided in an external power receiving device from a power transmitting side coil,
A power transmission current waveform detecting means for generating a power transmission current detection waveform w _i1 having a waveform similar to the waveform of the power transmission current i ₁ energized from the power feeding circuit to the power transmission side coil;
A reference current having a waveform similar to a current waveform when the power transmission side coil is energized from the power feeding circuit in a state where the power transmission side coil is not magnetically coupled to the power reception side coil (hereinafter referred to as “non-coupled state”). A reference current similar waveform generating means for generating a similar waveform w _ref ;
Differential waveform generation means for generating and outputting a power receiving side current detection waveform w _out that is a waveform of a weighted difference between the transmission current detection waveform w _i1 and the reference current similarity waveform w _ref ,
In a non-coupled state, the differential weight generation value in the differential waveform generation means or the output amplitude of the reference current similarity waveform in the reference current similarity waveform generation means so that the amplitude of the power reception side current detection waveform w _out becomes substantially zero A power receiving side current detection circuit, characterized in that a value is adjusted or adjustable. The reference current waveform similar generating means, the power transmission coil and is connected in parallel, the comprises a dummy coil not magnetically coupled to the power receiving coil, said as the waveform of the dummy coil current i _d is a current flowing through the dummy coil A reference current similarity waveform w _ref is generated,
Said differential waveform generating means outputs a dummy coil current i _d flowing through the dummy coil, the waveform of the weighted difference current between the power current i ₁ flowing through the power transmission coil as the power-receiving-side current detection waveform w _out Is,
When the ratio L _{d /} L ₁ for the inductance L ₁ of the power transmission coil of the inductance L _d of the dummy coil and alpha _d1, the differential waveform generating means as the waveform of the weighted difference current, the dummy coil current i _{2. The} power receiving side current detection circuit according to claim 1, which outputs a waveform of a weighted difference current (α _d1 i _d −i ₁ ) between α _d1 times _{d and} the primary side current i _1. . An adjustment resistor connected in parallel to the power transmission side coil and the dummy coil;
Said differential waveform generating means, the sum of said dummy coil current is a current flowing through the dummy coil i _d and adjustment current i _aj is a current flowing through the adjustment resistor, wherein a current flowing through the power transmission coil the primary A weighted differential current (α _d1 i _d + i _aj −i ₁ ) with respect to the side current i ₁ is detected,
The adjusting resistor is adjusted so that the weighted difference current (α _d1 i _d + i _aj −i ₁ ) becomes zero in a state where the power receiving coil is not coupled to the power transmitting coil. The power receiving side current detection circuit according to claim 2, wherein: A magnetic core, and first, second, and third conductors formed with winding portions where the magnetic core is linked one or more times, and
The first conductive wire is connected in series with the power transmission side coil and in parallel with the dummy coil on the input / output wiring of the power transmission side coil,
The second conductive wire includes a current transformer connected in series with the dummy coil and in parallel with the power transmission side coil on the input / output wiring of the dummy coil,
The power transmission current waveform detection means and the difference waveform generation means are configured integrally with the current transformer,
When a current flows from the dummy coil toward the winding portion of the second conductor, and a current flows from the power transmission side coil toward the winding portion of the first conductor,
The direction of the magnetic field created in the magnetic core by the current flowing in the winding part of the second conducting wire is the direction of the magnetic field created in the magnetic core by the current flowing in the winding part of the first conducting wire. So that the opposite direction
The power transmission coil is connected to the first conductor, the dummy coil is connected to the second conductor,
3. The power receiving side current detection circuit according to claim 2, wherein the number of turns of the winding portion of the second conducting wire is α _d1 times the number of turns of the winding portion of the first conducting wire. A magnetic core, and first, second, third, and fourth conductors formed with winding portions where the magnetic core is linked one or more times, and
The first conductive wire is connected in series with the power transmission side coil and in parallel with the dummy coil and the adjustment resistor on the input / output wiring of the power transmission side coil,
The second conductive wire is connected in series with the dummy coil and in parallel with the power transmission side coil and the adjustment resistor on the input / output wiring of the dummy coil,
The fourth conductor includes a current transformer connected to the adjustment resistor in series with the adjustment resistor and in parallel with the power transmission side coil and the dummy coil on the input / output wiring of the adjustment resistor,
The power transmission current waveform detection means and the difference waveform generation means are configured integrally with the current transformer,
Current flows from the dummy coil toward the winding portion of the second conductor, and current flows from the power transmission side coil toward the winding portion of the first conductor, and from the adjustment resistor When a current is passed toward the winding part of the fourth conductor,
The direction of the magnetic field created in the magnetic core by the current flowing in the winding part of the second conducting wire is the direction of the magnetic field created in the magnetic core by the current flowing in the winding part of the first conducting wire. The direction of the magnetic field created in the magnetic core by the current flowing in the winding portion of the fourth conductor is opposite to that of the fourth conductor, and the current flowing in the winding portion of the first conductor is the magnetic body. In the opposite direction to the direction of the magnetic field created in the core,
The power transmission side coil is connected to the first conductor, the dummy coil is connected to the second conductor, and the adjustment resistor is connected to the fourth conductor;
4. The power receiving side current detection circuit according to claim 3, wherein the number of turns of the winding portion of the second conducting wire is α _d1 times the number of turns of the winding portion of the first conducting wire. The reference current similar waveform generating means comprises a transmission voltage integration means for outputting the integrated waveform of the power transmission side is a voltage applied to the coil transmission voltage v ₁ of the waveform as an operation reference current waveform similar w _ref,
The difference waveform generation means is a weighted difference (γ _a w _ref −) between the reference current similarity waveform w _ref output from the transmission voltage integration means and the transmission current detection waveform w _i1 output from the transmission current waveform detection means. a waveform of γ _b w _i1 ) is output as the power receiving side current detection waveform w _out ,
In a state where the power transmission side coil is not magnetically coupled to the power reception side coil, the differential weight generation value (γ _a , γ in the differential waveform generation means is set so that the amplitude of the power reception side current detection waveform w _out becomes substantially zero. _{2. The} power receiving side current detection circuit according to claim 1, wherein an amplitude of the reference current similarity waveform w _ref in the power transmission voltage integrating means is adjusted or adjustable. A power transmission side coil, a waveform generator that generates an AC waveform to be passed through the power transmission side coil, and a power supply circuit that supplies AC power to the power transmission side coil according to the AC waveform generated by the waveform generator, A wireless power transmission device that wirelessly transmits power from a power transmission side coil to a power reception side coil provided in an external power reception device,
A power receiving side current detection circuit according to any one of claims 1 to 6,
A power transmission frequency adjusting means for adjusting a frequency of the power supply voltage so that a phase of the power reception side current detection waveform and a phase of the power supply voltage supplied to the power transmission side coil are in phase with each other. apparatus. The power transmission frequency adjusting means is
The weight difference current (α _{d1 i} d _{-i 1)} and the phase difference [Delta] [phi _2d between the dummy coil current i _d, or the weighted difference current output by said differential waveform generating means for outputting said difference waveform generating means ( phase difference detecting means for detecting a phase difference Δφ ₂₀ between the waveform of α _d1 i _d −i ₁ ) and the AC waveform generated by the waveform generator;
Based on the phase difference detected by the phase difference detection means,
The phase difference Δφ _2d between the weighted difference current and the dummy coil current is 90 degrees.
Alternatively, the phase difference Δφ ₂₀ between the weighted difference current and the AC waveform generated by the waveform generator is 0 degree.
The wireless power transmission device according to any one of claims 2 to 5, further comprising frequency control means for controlling a frequency of an AC waveform generated by the waveform generator. A primary side capacitor connected in series with the power transmission side coil and the dummy coil between the power transmission side coil and the dummy coil connected in parallel to the power supply circuit, and the power supply circuit. When,
Phase difference detection means for detecting a phase difference Δφ ₂₀ between the weighted difference current detected by the difference waveform generation means and the AC waveform generated by the waveform generator;
Based on the phase difference detected by the phase difference detection means,
The phase difference Δφ ₂₀ between the weighted difference current and the AC waveform generated by the waveform generator is 0 degree.
The wireless power transmission device according to any one of claims 2 to 5, further comprising frequency control means for controlling a frequency of an AC waveform generated by the waveform generator.