JP2019129681A - Dc-dc converter and power supply device - Google Patents

Abstract

To generate an operation voltage of a control circuit from a low input voltage.SOLUTION: A DC-DC converter 12 comprises an oscillator, a first step-up circuit, a second step-up circuit, and a control circuit 14. The oscillator includes a transformer 17 that includes a primary side coil 16A supplied with a first input voltage VIN1 and a secondary side coil 16B supplied with a third input voltage VIN3. The first step-up circuit includes: a first rectification element D1 supplied with a second input voltage VIN2; a second rectification element D2 connected in series to the first rectification element; and a capacitance element C2 whose one end is connected with one end at an output side of the secondary side coil and whose other end is connected with a terminal between the first rectification element and the second rectification element. The second step-up circuit includes: the secondary side coil of the transformer; a first switch SW that connects the one end at the output side of the secondary side coil to a reference potential; and a third rectification element D3 connected with the one end at the output side of the secondary side coil. The control circuit operates by using an output voltage of the first step-up circuit, and controls the first switch of the second step-up circuit.SELECTED DRAWING: Figure 1

Description

  Embodiments described herein relate generally to a DC-DC converter and a power supply device.

  A DC-DC converter operates as a blocking oscillator using a transformer including a primary winding and a secondary winding, and operates as a switching boost converter using the inductance of the secondary winding. There is a DC-DC converter with a mode to perform.

  In this DC-DC converter, an AC voltage is generated by oscillating a blocking oscillator based on an input voltage, and an operating voltage of the control circuit is generated by rectifying the AC voltage with a diode. Then, the control circuit is activated by this operating voltage, and the switching boost converter is activated by the activated control circuit. However, in this DC-DC converter, a high input voltage is required to generate the operating voltage of the control circuit.

A 40-mV Transformer-Reuse Self-Startup Boost Converter With MPPT Control for Thermoelectric Energy Harvesting. Solid-State Circuits, IEEE Journal of, Vol. .47, No. 12, pp. 3055-3067, 2012.

  Embodiments of the present invention provide a DC-DC converter and a power supply device that can generate an operating voltage of a control circuit from a low input voltage.

  The DC-DC converter as an embodiment of the present invention includes an oscillator, a first booster circuit, a second booster circuit, and a control circuit. The oscillator includes a transformer including a primary winding supplied with a first input voltage and a secondary winding supplied with a second input voltage. The first booster circuit includes a first rectifying element to which a third input voltage is supplied, a second rectifying element connected in series to the first rectifying element, and one end of the output side of the secondary winding. And a capacitance element having one end connected and the other end connected to a terminal between the first rectifying element and the second rectifying element. The second booster circuit includes a first switch connecting the secondary winding of the transformer and one end of the output side of the secondary winding to a reference potential, and an output side of the secondary winding. And a third rectifying element connected to the one end. The control circuit operates using the output voltage of the first booster circuit to control the first switch of the second booster circuit.

The circuit diagram of the power supply device provided with the DC-DC converter which concerns on 1st Embodiment. The time waveform of the various voltages and electric current which concern on 1st Embodiment. The circuit diagram of the power supply device provided with the DC-DC converter which concerns on 2nd Embodiment. 7 shows time waveforms of various voltages and currents according to the second embodiment. The circuit diagram of the power supply device provided with the DC-DC converter which concerns on 3rd Embodiment. The time waveform of the various voltages and electric currents which concern on 3rd Embodiment. The circuit diagram of the power supply device provided with the DC-DC converter which concerns on 4th Embodiment. The time waveform of the various voltages and electric current which concern on 4th Embodiment. The circuit diagram of the power supply device provided with the DC-DC converter which concerns on 5th Embodiment. The circuit diagram of the power supply device provided with the DC-DC converter which concerns on 6th Embodiment. The time waveform of the various voltages and electric currents which concern on 6th Embodiment.

Hereinafter, embodiments of the present invention will be described with reference to the drawings.
First Embodiment
FIG. 1 is a circuit diagram of a power supply device according to the first embodiment. This power supply device is an environmental power generation device including a solar battery PV, thermoelectric power generation elements TEG1 and TEG2, and a DC-DC converter 12.

  The thermoelectric generation element TEG1 is modeled by an electromotive force VTEG1 and an output resistance RTEG1. The thermoelectric generation element TEG2 is modeled by an electromotive force VTEG2 and an output resistance RTEG2. The solar cell PV is modeled by an electromotive force VPV and an output resistance RPV.

  The electromotive force of the thermoelectric generator is proportional to the temperature difference between the high temperature side member and the low temperature side member, and is several tens of mV to several 100 mV for a temperature difference of about several degrees Celsius. The output resistance of the thermoelectric generation element is a substantially constant value regardless of the temperature difference, and is a value of several ohms to several hundred ohms.

  The output voltage of the thermoelectric generator TEG1 is input to the input terminal 1 of the DC-DC converter 12 as the input voltage VIN1. The capacitor CIN1 smoothes the input voltage VIN1 input to the input terminal 1. Similarly, the output voltage of the thermoelectric generation element TEG2 is input to the input terminal 3 as the input voltage VIN3. The capacitor CIN3 smoothes the input voltage VIN3 input to the input terminal 3. The output voltage of the solar cell PV is input to the input terminal 2 of the DC-DC converter 12 as an input voltage VIN2. The capacitor CIN2 smoothes the input voltage VIN2 input to the input terminal 2. Here, the thermoelectric generation element TEG1 corresponds to a first power source, the thermoelectric generation element TEG2 corresponds to a second power source, and the solar cell PV corresponds to a third power source. The types of first to third power supplies are not limited to these, and may be arbitrary power supplies.

  The load devices Load1 and Load2 are circuits that operate based on the voltage supplied from the DC-DC converter 12. For example, a circuit that consumes power, such as an LED, a digital circuit, or an analog circuit, a device that stores power such as a storage battery, or a sensor may be used. For example, the load device operates with a power supply voltage of 1V or more. The DC-DC converter 12 boosts the input voltage in order to drive the load device with an appropriate voltage even when the input voltage is small. For example, the input voltage VIN3 is boosted, and the boosted output voltage is supplied to the load device 2.

  The DC-DC converter 12 has a transformer 17. The transformer 17 includes a primary side winding 16A and a secondary side winding 16B. The DC-DC converter 12 also has two operation modes. The first one is the start-up mode, in which the DC-DC converter 12 operates as a blocking oscillator using the transformer 17. The second is a boost mode. In this mode, the secondary winding 16B of the transformer 17 is used as an inductance element to operate as a switching boost converter. After the output voltage VOUT1 of a predetermined value or more is obtained in the start-up mode, the control circuit 14 is started using the output voltage VOUT1 as an operating voltage. Under the control of the control circuit 14 that has been started up, transition to the boost mode is made.

  The upper terminal (terminal on the input side) of the primary winding 16A of the transformer 17 is connected to the input terminal 1, and the lower terminal (terminal on the output side) is the drain of the native nMOS switch (native nMOS transistor) 15. It is connected to the terminal. The native nMOS switch 15 is an nMOS transistor having a threshold at or near 0 [V]. Instead of the native nMOS switch, a depletion type transistor may be used. Native nMOS switches and depletion type transistors are used as, for example, normally on transistors.

  The upper side terminal (terminal on the input side) of the secondary side winding 16B of the transformer 17 is connected to the capacitor C1, the capacitor C2, the switch SW, and the diode D3. The voltage at the upper terminal is represented as VLX. The capacitor is an example of a capacitance element, and any configuration other than a capacitor may be used as long as it is a capacitance element. For example, a series connection of a plurality of capacitors, a parallel connection of a plurality of capacitors, or a series-parallel connection of a plurality of capacitors may be used. A resistor or other element may be connected to the capacitor in series, parallel, or series-parallel. The same applies to capacitors other than the capacitors C1 and C2 used in the present embodiment and other embodiments.

  One end of the switch SW is connected to the upper terminal of the secondary winding 16B of the transformer 17, and the other end is connected to the ground. That is, the switch SW connects the upper terminal of the secondary winding 16B of the transformer 17 to the ground. The control terminal of the switch SW is connected to the control circuit 14. The switch SW is, for example, an nMOS switch (nMOS transistor).

  One end (anode) of the diode D3 is connected to the upper terminal of the secondary winding 16B of the transformer 17, and the other end (cathode) is connected to the output terminal 13B. The diode D3 is an example of a rectifying element, and another type of rectifying element such as a MOS transistor may be used.

  The capacitor COUT2 is a smoothing capacitance element. One end of the capacitor COUT2 is connected to the other end of the diode D3 and the output terminal 13B, and the other end is connected to the ground.

  A load device Load2 is connected to the output terminal 13B. The voltage at the output terminal 13B is represented as VOUT2.

  The lower terminal of the secondary winding 16 B of the transformer 17 is connected to the input terminal 3. A terminal 19 of the capacitor C1 opposite to the transformer 17 is connected to the ground via a resistor R1. Also, the terminal 19 is connected to the control terminal of the native nMOS switch. The terminal 19 is connected to the control circuit 14 by wiring. The voltage of the terminal 19 is denoted as V1.

  If there is no output of the control signal from the control circuit 14 to the terminal 19 (if the output to the terminal 19 of the control circuit 14 is open), the native nMOS switch 15 is on. At this time, a loop composed of the transformer 17, the capacitor C1, and the native nMOS switch 15 operates as a blocking oscillator. An alternating voltage is generated at the upper terminal of the secondary winding 16B of the transformer 17 by the oscillation operation of the blocking oscillator. The voltage at the upper terminal of the secondary side winding 16B of the transformer 17 is denoted by VLX.

  The input terminal 2 is connected to the output terminal 13A via a diode D1 and a diode D2. The diode D1 and the diode D2 are connected in series. One end (anode) of the diode D1 is connected to the input terminal 2, and the other end (cathode) is connected to one end (anode) of the diode D2. The cathode of the diode D2 is connected to the output terminal 13A. One end of a capacitor C2 is connected to a terminal 18 between the diode D1 and the diode D2. The other end of the capacitor C2 is connected to the upper terminal of the secondary winding 16B of the transformer 17, the terminal of the capacitor C1 on the transformer 17, the terminal of the switch SW opposite to the ground, and one end (anode) of the diode D3. Yes.

  The diode D1, the diode D2, and the capacitor C2 constitute a Dixon booster circuit (first booster circuit). The output side terminal (the other end of the diode D2) 7 of the Dickson booster circuit is connected to the power supply terminal of the control circuit 14. The control circuit 14 starts up when the voltage at the terminal 7 becomes equal to or higher than the threshold. The control circuit 14 uses the voltage at the terminal 7 as the operating voltage.

  Each terminal in the present embodiment may be a terminal for connection or may represent an arbitrary point (node) on the wiring.

  The control circuit 14 controls the switch SW and the native nMOS switch 15. In the start-up mode, the control circuit 14 sets the control signal VSW of the switch SW to a low level (L level) voltage, and the switch SW is turned off. When the input voltage VIN2 is boosted in the startup mode and the control circuit 14 is activated by the boosted voltage at the terminal 7, the control circuit 14 enters the boost mode. The control circuit 14 turns off the native nMOS switch 15 to stop the oscillator, and controls the switch SW to function as a boost up converter. That is, the control circuit 14 boosts the input voltage VIN3 and supplies it to the load device Load2. Further, the input voltage VIN2 is boosted and supplied to the load circuit Load1.

  FIG. 2 shows time waveforms of various voltages and currents in the DC-DC converter 12 according to the first embodiment.

  Input power is supplied to the DC-DC converter 12 from the thermoelectric generation elements TEG1 and TEG2 and the solar cell PV. The voltage VLX of the upper terminal of the secondary side winding 16B of the transformer 17 rises. This voltage VLX is an alternating waveform that oscillates around the input voltage VIN3 of the input terminal 3.

  The voltage VLX is rectified by the diode D3, and a current flows on the output terminal 13B side. As a result, the output voltage VOUT2 rises. At the same time, due to the action of the voltage VLX, an alternating current flows into the terminal 18 between the diode D1 and the diode D2 via the capacitor C2 by the Dickson booster circuit constituted by the capacitor C2, the diode D1 and the diode D2. Assuming that the voltage at the terminal 18 is V2, since the voltage VLX is an AC voltage, the voltage V2 via the capacitor C2 is also an AC voltage. On the other hand, a direct current from the input terminal 2 passes through the diode D1. The combined current of the direct current and the alternating current is rectified by the diode D2, flows to the output terminal 13A, and the output voltage VOUT1 also rises.

  While raising the output voltages VOUT2 and VOUT1, the oscillation amplitude of the voltage VLX increases. The increase of the amplitude of the voltage VLX stops at a certain amplitude and becomes maximum. The maximum oscillation voltage amplitude of this voltage VLX is AVLXMAX. In the range of normal use, the maximum oscillation voltage amplitude AVLXMAX becomes larger as the electromotive force VTEG1 is larger (ie, as the input voltage VIN1 is larger).

  The drop voltage of the forward voltage drop of the diode D1, the diode D2, and the diode D3 is VF. The maximum value of voltage VLX is input voltage VIN3 + maximum oscillation voltage amplitude AVLXMAX. The maximum value that the output voltage VOUT2 can take is VIN3 + AVLXMAX−VF, which is lowered by VF below the voltage drop of the diode D3. If the electromotive force VTEG2 is large, the input voltage VIN3 is also large, and the maximum voltage of the voltage VLX is also large. Further, when the electromotive force VTEG1 and the input VIN1 of the thermoelectric generator are large, the maximum voltage of the voltage VLX is large.

  The maximum amplitude of the voltage V2 at the terminal 18 is denoted as AV2MAX. At this time, the maximum value at which the output voltage VOUT1 can be obtained is 2 × (AV2MAX−VF) + VIN2. That is, the voltage drop of two diodes D1 and D2 is obtained by multiplying the maximum amplitude AV2MAX by twice the input voltage VIN2 (adding the upper maximum amplitude AV2MAX and the lower maximum amplitude AV2MAX) to two diodes D1 and D2 (2 A value obtained by subtracting (VF) corresponds to this. In the case of the maximum value VIN3 + AVLXMAX-VF at which the output voltage VOUT2 can be obtained, the voltage VLX is added centering on the input voltage VIN3, and therefore, it is not necessary to double AVLXMAX.

  When the output voltage VOUT1 becomes equal to or higher than the threshold value, the control circuit 14 is activated. Here, when the capacitance value of the capacitor COUT1 is large, it takes time for the output voltage VOUT1 to rise. On the other hand, if the potential is the same, a larger amount of energy can be held in the capacitor COUT1, so that the voltage is unlikely to drop even when the control circuit 14 consumes power. The capacitor COUT1 may be set to an appropriate capacitance value according to the required performance or specification.

  When the electromotive force VTEG1 of the thermoelectric generation element is large, that is, when the input VIN1 is large, the maximum amplitude AV2MAX of the voltage V2 of the terminal 18 also becomes large. Therefore, the maximum value (= 2 × (AV2MAX−VF) + VIN2) at which the output voltage VOUT1 can be obtained is increased. Further, also when the electromotive force VPV of the solar cell and the input voltage VIN2 from the solar cell are large, the maximum value (= 2 × (AV2MAX−VF) + VIN2) at which the output voltage VOUT1 can be obtained becomes large.

  The maximum value (= 2 × (AV2MAX−VF) + VIN2) that the output voltage VOUT1 can take can easily be larger than the maximum value (= VIN3 + AVLXMAX−VF) that the output voltage VOUT2 can take. Therefore, the Dickson booster circuit can provide a large output voltage VOUT1 even if the input voltage (VIN1 and / or VIN3) is small. Therefore, the electromotive force VPV of the solar cell PV may be larger than the electromotive forces VTEG1 and VTEG2. In the boost mode, by using the inductance of the secondary winding 16B of the transformer 17, boosting is performed more efficiently than in the start-up mode, so a large output voltage VOUT2 can be obtained even from small electromotive forces VTEG1 and VTEG2. .

  The control circuit 14 activated when the output voltage VOUT1 becomes equal to or higher than the threshold value sets the voltage V1 at the terminal 19 to be less than the threshold value (here, 0 [V]). The native nMOS switch is turned off. Thereby, the oscillation of the blocking oscillator is stopped. This ends the start-up mode and enters the boost mode.

  The control circuit 14 outputs periodic signals of H level and L level as a control signal (opening / closing signal) VSW of the switch SW. The ratio of periodic signals at H level and L level is determined according to the desired boost ratio. When the control signal VSW is at the H level, a voltage corresponding to the input VIN3 is applied to the secondary winding 16B of the transformer 17, so that the current (current IL) of the secondary winding 16B increases.

  When the control signal VSW becomes L level, the inductance of the secondary side winding 16B of the transformer 17 acts to keep the current flowing, so that the current flows to the output terminal 13B side through the diode D3. At this time, the voltage VLX of the upper terminal of the secondary winding 16B rises to the output voltage VOUT2 + VF, and the voltage of VIN3- (VOUT2 + VF) is applied to the secondary winding 16B of the transformer 17, so the current IL decreases. To go. When the current IL decreases to 0, backflow is prevented by the rectifying action of the diode D3.

  As the electric power that can be supplied from the thermoelectric power generation element TEG2 is larger, the current of the secondary winding 16B of the transformer 17 can be increased, so that larger electric power can be output to the output terminal 13B. By using the inductance of the secondary winding 16B of the transformer 17, it is possible to realize an operation in a boost mode that is more efficient than a startup mode that operates as a blocking oscillator.

  According to this embodiment, since the high output voltage VOUT1 can be obtained through the Dickson booster, the control circuit 14 can be started even when a low input voltage (VIN1, VIN3, or both of them) is used. From the input voltage, the inductance of the secondary winding 16B of the transformer 17 can be boosted with high utilization efficiency.

Second Embodiment
FIG. 3 is a circuit diagram of a power supply device according to the second embodiment. The present embodiment is characterized in that the solar cell PV and the thermoelectric power generation element TEG1 in FIG. 1 in the first embodiment are made common as one power supply device (here, a solar cell PV). Hereinafter, the difference from the first embodiment will be mainly described. The same elements as those in FIG.

  This power supply device is an environmental power generation device including a solar battery PV, a thermoelectric power generation element TEG, and a DC-DC converter 22. The thermoelectric generation element TEG is modeled by an electromotive force VTEG and an output resistance RTEG. The output voltage of the thermoelectric generator TEG is input to the terminal 3 as the input voltage VIN3. Here, the input voltage VIN1 corresponds to the third input voltage and the first input voltage, and the input voltage VIN3 corresponds to the second input voltage.

  An nMOS switch (nMOS transistor) 27 is provided instead of the switch SW in FIG. 1, and a pMOS switch (pMOS transistor) 28 is provided instead of the diode D3. While two load devices are present in FIG. 1, one load device Load is provided in FIG. The load device Load is connected to the output terminal 13.

  The control circuit 24 is connected to the upper terminal of the secondary winding 16B of the transformer 17, and in the start-up mode before start-up, a negative voltage gradually decreasing to the nMOS switch 27 using the voltage of the terminal, pMOS The switch 28 operates to provide a gradually rising positive voltage. That is, in the start-up mode, the control signal VN of the nMOS switch 27 gradually becomes a low value (negative voltage) by the control circuit 24, and the nMOS switch 27 is turned off. Further, the control signal VP of the pMOS switch 28 gradually increases, and the pMOS switch 28 is turned off. In these operations, the nMOS switch 27 and the pMOS switch 28 are turned off more strongly (by applying a voltage that is sufficiently low or large with respect to the threshold value), and leakage current from the upper terminal of the secondary winding 16B of the transformer 17 is reduced. This is done to reduce the voltage swing of the upper terminal. After startup, the control circuit 24 operates using the output voltage of the Dixon booster circuit (the voltage at the terminal 7) as the operating voltage.

  The input terminal 1 is connected to the output terminal 13 via the diode D1 and the diode D2. The terminal 7 on the output side of the Dickson booster circuit is connected to the control circuit 24 and the output terminal 13. The voltage of the output terminal 13 is increased by the voltage output from the Dickson booster circuit.

  FIG. 4 shows time waveforms of various voltages and currents in the DC-DC converter 22 according to the second embodiment.

  From the start of the start-up mode, the control signal VN of the nMOS switch 27 gradually becomes a low value (negative voltage) by the control circuit 24, and the nMOS switch 27 is turned off. The control signal VP of the pMOS switch 28 gradually increases, and the pMOS switch 28 is turned off. The voltage VLX of the upper terminal of the secondary side winding 16B of the transformer 17 is rectified by the body diode of the pMOS switch 28, a current flows to the output terminal 13 side, and the output voltage VOUT rises. At the same time, due to the action of the voltage VLX, the current flows from the input terminal 1 through the diode D1 and the diode D2 to the output terminal 13 side by the Dixon booster circuit constituted by the capacitor C2, the diode D1, and the diode D2. Also, the output voltage VOUT rises.

  The oscillation amplitude of the voltage VLX is increased while the output voltage VOUT is increased. The increase of the amplitude of the voltage VLX stops at a certain oscillation amplitude and becomes maximum. The maximum oscillation voltage amplitude of this VLX is AVLXMAX. In the range of normal use, AVLXMAX increases as the power VPV of the solar cell PV increases, that is, as the input voltage VIN1 increases. Let VF be the forward voltage drop of the body diode of the diode D1, the diode D2, and the pMOS. As in the first embodiment, the maximum value of the voltage VLX is VIN3 + AVLXMAX, and the output voltage VOUT considering only the path via the pMOS body diode is VIN3 + AVLXMAX−VF, as in the first embodiment. . Assuming that the maximum amplitude of the voltage V2 at the terminal 18 is AV2MAX, the maximum value of the output voltage (that is, the voltage that can be taken by the terminal 7) considering only the path through the diode D1 and the diode D2 is the same as the first embodiment. 2 × (AV2MAX−VF) + VIN1. When the voltage at the terminal 7 (equal to VOUT) becomes equal to or greater than the threshold, the control circuit 24 is activated.

  If the capacitance of the capacitor COUT is large, it takes time for the output voltage VOUT to rise, but if the potential is the same, larger energy can be held, so that the voltage is unlikely to drop even if the control circuit 24 consumes power. The capacitor COUT may be set to an appropriate capacitance value according to the required performance or specification.

  When the voltage at the terminal 7 becomes equal to or higher than the threshold value, the control circuit 24 is activated to set the voltage V1 at the terminal 19 to a certain negative voltage. By using a negative voltage, leakage current can be further suppressed even in a native nMOS switch having a threshold value of 0 [V]. This stops the oscillation of the blocking oscillator.

  The control circuit 24 outputs a high level (H level) and low level (L level) periodic signal as the control signal VN of the nMOS switch 27. When the control signal VN is at the H level, the nMOS switch 27 is turned on, and the input VIN1 is applied to the secondary winding 16B of the transformer 17, so that the current IL increases. When the control signal VN becomes L level, the control signal VP of the pMOS switch 28 is set to L level, and the pMOS switch 28 is turned on.

  Since the inductance of the secondary winding 16B of the transformer 17 acts to keep the current flowing, the current flows to the output terminal 13 side through the pMOS switch 28. At this time, the voltage VLX rises to the voltage VOUT, and the voltage of VIN1-VOUT is applied to the secondary winding 16B of the transformer 17, so the current decreases. When the current decreases to 0, the control signal VP is set to H level and the pMOS is turned off. This prevents backflow. As the electric power that can be supplied from the thermoelectric power generation element TEG is larger, the current of the secondary winding 16B of the transformer 17 can be increased. By using the inductance of the secondary winding 16B of the transformer 17, it is possible to operate as a boost mode that is more efficient than the startup mode that operates as a blocking oscillator.

  According to the present embodiment, since the Dickson booster circuit is connected to the input terminal 1, there is an effect that the voltage of the output terminal 13 can be increased by the input voltage VIN1 of the input terminal 1. Further, by making the output of the Dixon booster circuit the same as the output of the switching boost converter using the transformer secondary winding, it is possible to continue supplying power to the control circuit 24 even in the boost mode.

Third Embodiment
FIG. 5 is a circuit diagram of a power supply device according to the third embodiment. In the present embodiment, the thermoelectric power generation elements TEG1 and TEG2 of FIG. 1 in the first embodiment are shared as one power supply device (here, the thermoelectric power generation element TEG1), and the solar cell PV is replaced with the thermoelectric power generation element TEG2. It features. Hereinafter, the difference from the first and second embodiments will be mainly described. The same elements as those in FIG.

  This power supply device is an environmental power generation device provided with the thermoelectric power generation element TEG 1 and the DC-DC converter 32. The thermoelectric generators TEG1 and TEG2 are modeled by electromotive forces TEG1 and VTEG2 and output resistors RTEG1 and RTEG2, respectively.

  Output voltages of the thermoelectric generators TEG1 and TEG2 are input to the input terminals 1 and 2 as inputs VIN1 and VIN2, respectively. The input voltage VIN1 corresponds to the first input voltage and the second input voltage, and the input voltage VIN2 corresponds to the third input voltage.

  An nMOS switch (nMOS transistor) 27 is provided instead of the switch SW of FIG.

  The lower terminal of the secondary winding 16 B of the transformer 17 is connected to the input terminal 1. The upper terminal of the secondary winding 16B of the transformer 17 is connected to the capacitor C1, the capacitor C2, the nMOS switch 27, and the diode D3.

  The control circuit 64 is connected to the upper terminal of the secondary winding 16B of the transformer 17. In the start-up mode before starting, the control circuit 64 applies a gradually decreasing negative voltage to the nMOS switch 27 using the voltage of the terminal. To work. That is, in the start-up mode, the control signal VN of the nMOS switch 27 is gradually decreased (negative voltage), whereby the nMOS switch 27 is turned off. After start-up, it operates with the output voltage (voltage of the terminal 7) of the Dickson booster circuit as the operating voltage.

  FIG. 6 shows time waveforms of various voltages and currents in the DC-DC converter 32 according to the third embodiment.

  From the start of the start-up mode, the control signal VN of the nMOS switch 27 gradually becomes a small value (negative voltage) by the control circuit 34, and the nMOS switch 27 is turned off. As in the first embodiment, the oscillation amplitude of the voltage VLX is increased while the output voltages VOUT2 and VOUT1 are increased. The increase of the amplitude of the voltage VLX stops at a certain oscillation amplitude and becomes maximum. When the output voltage VOUT1 becomes equal to or higher than the threshold value, the control circuit 34 is activated.

  The activated control circuit 34 sets the voltage V1 of the terminal 19 to a certain negative voltage. By setting the voltage to a negative voltage, it is possible to further suppress the leak current even in the native nMOS switch whose threshold voltage is 0 [V]. This stops the oscillation of the blocking oscillator.

  The control circuit 34 outputs periodic signals of H level and L as the control signal VN of the nMOS switch 27. Since the input voltage VIN1 is applied to the secondary winding 16B of the transformer 17 when the control signal VN is at the H level, the current IL increases. When the control signal VN becomes L level, the inductance of the secondary side winding 16B of the transformer 17 acts to keep the current flowing, so that the current flows to the output terminal 13B side through the diode D3. At this time, the voltage VLX rises to VOUT2 + VF, and a voltage of VIN1-VOUT2-VF is applied to the secondary winding 16B of the transformer 17. Therefore, the current decreases. When the current decreases to 0, backflow is prevented by the rectification action of the diode D3. As the electric power that can be supplied from the thermoelectric power generation element TEG1 is larger, the current of the secondary winding 16B of the transformer 17 can be increased, so that larger electric power can be output to the output terminal 13B. By using the inductance of the secondary winding 16B of the transformer 17, it is possible to operate as a boost mode that is more efficient than the startup mode that operates as a blocking oscillator.

  According to this embodiment, even with a low input voltage, a high output voltage can be obtained through the Dickson booster, so that the control circuit 34 can be started with a low input voltage. Further, in the boost mode, high efficiency voltage boosting is possible using the inductance of the secondary side winding 16B of the transformer 17, so that a high output voltage can be obtained from a low input voltage. By sharing the same power supply TEG1 as the power supply of the blocking oscillator and the power supply of the switching boost converter, the area and the volume can be reduced.

Fourth Embodiment
FIG. 7 is a circuit diagram of a power supply device according to the fourth embodiment. This embodiment is characterized in that the thermoelectric generator and the capacitor CIN2 of FIG. 5 in the third embodiment are removed, and one end (anode) of the diode D1 is connected to the ground. That is, a ground voltage is applied to one end of the diode D1. Hereinafter, the difference from the third embodiment will be mainly described. The same elements as in FIG.

  This power supply device includes a thermoelectric generation element TEG and a DC-DC converter 42. The thermoelectric generation element TEG is modeled by an electromotive force VTEG and an output resistance RTEG. The output voltage of the thermoelectric generation element TEG is input to the input terminal 1 as an input voltage VIN1.

  The diode D3 in FIG. 5 replaces the pMOS switch 28. The pMOS switch 28 operates in the same manner as the pMPOS switch 28 in the second embodiment.

  The upper terminal of the secondary winding 16B of the transformer 17 is connected to the capacitor C1, the capacitor C2, the nMOS switch 27, and the pMOS switch 28. The load device Load is connected to the output terminal 13.

  The control circuit 44 is connected to the upper terminal of the secondary winding 16B of the transformer 17. In the start-up mode before start-up, the voltage of the terminal is used to apply a gradually decreasing negative voltage to the nMOS switch 27. , PMOS switch 28 is operated to provide a gradually increasing positive voltage. That is, in the start-up mode, the control signal VN of the nMOS switch 27 is gradually reduced (negative voltage) by the control circuit 44, and the nMOS switch 27 is turned off. The control signal VP of the pMOS switch 28 is also gradually increased by the control circuit 44, and the pMOS switch 28 is turned off. After startup, the control circuit 44 operates using the output voltage of the Dixon booster circuit (the voltage at the terminal 7) as the operating voltage.

  FIG. 8 shows time waveforms of various voltages and currents in the DC-DC converter 42 according to the fourth embodiment.

  At the start of the start-up mode, the control signal VN of the nMOS switch 27 is gradually reduced (negative voltage) by the control circuit 44, and the nMOS switch 27 is turned off. The control signal VP of the pMOS switch 28 is also gradually increased by the control circuit 44, and the pMOS switch 28 is turned off. The voltage VLX of the upper terminal of the secondary winding 16B of the transformer 17 is rectified by the body diode of the pMOS switch 28, a current flows to the output terminal 13B side, and the output voltage VOUT2 rises. At the same time, the voltage VLX causes the current to flow from the input terminal 1 through the diode D1 and the diode D2 to the output terminal 13A side by the Dixon booster circuit constituted by the capacitor C2, the diode D1, and the diode D2. The voltage VOUT1 also rises. The oscillation amplitude of the voltage VLX is increased while the output voltages VOUT2 and VOUT1 are increased. The increase of the amplitude of the voltage VLX stops at a certain oscillation amplitude and becomes maximum.

  When the output voltage VOUT1 becomes equal to or higher than the threshold value, the control circuit 44 sets the voltage V1 at the terminal 19 to a certain negative voltage. By using a negative voltage, the leak current can be further suppressed in the native nMOS switch 15 whose threshold is 0 [V]. This stops the oscillation of the blocking oscillator.

  The control circuit 44 outputs periodic signals of H level and L level as the control signal VN of the nMOS switch 27. When the control signal VN is at the H level, the input voltage VIN1 is applied to the secondary winding 16B of the transformer 17, so the current IL increases. When the control signal VN becomes L level, the inductance of the secondary winding 16B of the transformer 17 acts to keep the current flowing, so the current flows to the output terminal 13B side through the body diode of the pMOS switch 28. . At this time, the voltage VLX rises to the output voltage VOUT2, and since the voltage of VIN1-VOUT2 is applied to the secondary winding 16B of the transformer 17, the current decreases. When the current decreases to 0, backflow is prevented by the rectifying action of the body diode of the pMOS switch 28. The greater the electric power that can be supplied by the thermoelectric power generation element TEG, the larger the current of the secondary winding 16B of the transformer 17 can be, so that larger electric power can be output to the output terminal 13B. By using the inductance of the secondary winding 16B of the transformer 17, it is possible to operate as a boost mode that is more efficient than the startup mode that operates as a blocking oscillator.

  According to this embodiment, even with a low input voltage, a high output voltage can be obtained through the Dickson booster, so that the control circuit 44 can be started with a low input voltage. Further, in the boost mode, high efficiency voltage boosting is possible using the inductance of the secondary side winding 16B of the transformer 17, so that a high output voltage can be obtained from a low input voltage. By sharing the same power supply TEG as the power supply for the blocking oscillator and the power supply for the switching boost converter, the area and volume can be reduced. By connecting the Dickson booster to ground, the fluctuation of the Dickson booster input voltage is reduced.

Fifth Embodiment
FIG. 9 is a circuit diagram of a power supply device according to the fifth embodiment. This embodiment is characterized in that the connection destination of one end (anode) of the diode D1 of FIG. 7 in the fourth embodiment is changed from the ground to the input terminal 1. Hereinafter, the difference from the fourth embodiment will be mainly described. The same elements as those in FIG.

  In the DC-DC converter 52, the input terminal 1 is connected to the output terminal 13A via a diode D1 and a diode D2. An input voltage VIN1 is input to the input terminal 1 from the thermoelectric generation element TEG. The input voltage VIN1 corresponds to the first input voltage, the second input voltage, and the third input voltage. Other configurations are the same as those in FIG. The operation of the power supply device of the present embodiment is the same as that of the fourth embodiment, and thus the description thereof is omitted.

  According to this embodiment, even with a low input voltage, a high output voltage can be obtained through the Dickson booster, so that the control circuit 54 can be started with a low input voltage. Further, in the boost mode, high efficiency voltage boosting is possible using the inductance of the secondary side winding 16B of the transformer 17, so that a high output voltage can be obtained from a low input voltage. By sharing the same power supply TEG as the power supply for the blocking oscillator and the power supply for the switching boost converter, the area and volume can be reduced.

(Sixth embodiment)
FIG. 10 is a circuit diagram of a power supply device according to the sixth embodiment. This power supply device is an environmental power generation device including a thermoelectric power generation element TEG and a DC-DC converter 62. This embodiment corresponds to an extension of the combination of the second and fifth embodiments. Hereinafter, the difference from the second and fifth embodiments will be mainly described. The same elements as those in FIG. 3 and FIG.

  The input terminal 1 is connected to the output terminal 13A via the diode D1 and the diode D2.

  The upper terminal of the secondary winding 16B of the transformer 17 is connected to the capacitor C1, the capacitor C2, the nMOS switch 27, the pMOS switch 28A, and the pMOS switch 28B. The input-side terminals of the pMOS switch 28A and the pMOS switch 28B are connected in parallel to the upper terminal of the secondary winding 16B. Control terminals of the pMOS switch 28A and the pMOS switch 28B are connected to the control circuit 64. Terminals on the output side of the pMOS switch 28A and the pMOS switch 28B are connected to the capacitors COUT1 and COUT2 and the output terminals 13A and 13B. Load devices Load1 and Load2 are connected to the output terminals 13A and 13B.

  Terminals on the output side of the pMOS switch 28A and the pMOS switch 28B are connected to the input terminal of the high voltage selector 61 (High Voltage Selector). The output terminal of the high voltage selection circuit 61 is connected to the respective bulk terminals of the pMOS switch 28A and the pMOS switch 28B via the capacitor 65. One end of the capacitor 65 is connected to the output terminal of the high voltage selection circuit 61, and the other end is connected to the ground. The high voltage selection circuit 61 smoothes the higher one of the voltage VOUT1 and the output voltage VOUT2 with the capacitance of the capacitor 65, and applies it to the bulk terminals of the pMOS switch 28A and the pMOS switch 28B.

  The control circuit 64 is connected to the upper terminal of the secondary winding 16B of the transformer 17, and in the start-up mode before start-up, a negative voltage gradually decreasing to the nMOS switch 27 using the voltage of the terminal, pMOS The switch 28A and the pMOS switch 28B operate to provide a gradually rising positive voltage. That is, in the start-up mode, the control signal VN of the nMOS switch 27 is gradually decreased (negative voltage), and thereby the nMOS switch 27 is turned off. Further, the control signals VP1 and VP2 of the pMOS switch 28A and the pMOS switch 28B are gradually increased, whereby the pMOS switch 28A and the pMOS switch 28B are turned off. These operations are performed by turning off the nMOS switch 27, the pMOS switch 28A, and the pMOS switch 28B more strongly (by applying a voltage that is sufficiently low or large with respect to the threshold value) and from the upper terminal of the secondary winding 16B of the transformer 17. To reduce the leakage current and to prevent the reduction of the voltage amplitude of the upper terminal. After startup, the control circuit 64 operates using the output voltage of the Dixon booster circuit (the voltage at the terminal 7) as the operating voltage.

  FIG. 11 shows waveforms of various voltages and currents in the DC-DC converter 62 according to the sixth embodiment.

  From the start of the start-up mode, the control signal VN of the nMOS switch 27 gradually decreases (negative voltage), whereby the nMOS switch 27 is turned off. Further, the control signals VP1 and VP2 of the pMOS switch 28A and the pMOS switch 28B are gradually increased, whereby the pMOS switch 28A and the pMOS switch 28B are turned off. The voltage at the upper terminal of the secondary winding 16B of the transformer 17 is rectified by the body diodes of the pMOS switch 28A and the pMOS switch 28B, current flows to the output terminal 13A and the output terminal 13B side, and the output voltage VOUT1 rises. At the same time, a current flows from the input terminal 1 through the diode D1 and the diode D2 to the output terminal 13A side by the action of the voltage VLX by the Dixon booster circuit constituted by the capacitor C2, the diode D1 and the diode D2. As a result, the output voltage VOUT1 also rises. The oscillation amplitude of the voltage VLX is increased while the output voltage VOUT1 is increased. The increase of the amplitude of VLX stops at a certain oscillation amplitude and becomes maximum.

  When the output voltage VOUT1 becomes equal to or higher than the threshold value, the control circuit 64 is activated. The activated control circuit 64 sets the control voltage V1 of the nMOS switch 27 to a certain negative voltage. By setting the voltage to a negative voltage, the native nMOS switch 15 whose threshold voltage is 0 [V] is turned off, and the leak current of the native nMOS switch 15 can be further suppressed. This stops the oscillation of the blocking oscillator.

  The control circuit 64 outputs a periodic signal of H level and L level as the control signal VN. Since the input voltage VIN1 is applied to the secondary winding 16B of the transformer 17 when the control signal VN is at the H level, the current IL increases. When the control signal VN becomes L level, either the control signal VP1 of the pMOS switch 28A or the control signal VP2 of the pMOS switch 28B is set to L level. Correspondingly, one of the pMOS switch 28A or the pMOS switch 28B is turned on.

  Since the inductance of the secondary winding 16B of the transformer 17 acts to keep current flowing, the current flows to either the output terminal 13A or the output terminal 13B through one of the pMOS switch 28A and the pMOS switch 28B which is turned on. Flowing. When the pMOS switch 28A is on, the voltage VLX rises to the output voltage VOUT1, and the voltage of VIN1-VOUT1 is applied to the secondary winding 16B of the transformer 17, so the current decreases. When the current drops to zero, the control circuit 64 detects this and the control signal VP1 is set to the H level, whereby the pMOS switch 28A is turned off. This prevents backflow.

  When the pMOS switch 28B is on, the voltage VLX rises to the output voltage VOUT2, and the voltage of VIN1-VOUT2 is applied to the secondary winding 16B of the transformer 17, so the current decreases. When the current drops to zero, the control circuit 64 detects this and the control signal VP2 is set to H level, and the pMOS switch 28B is turned off. This prevents backflow.

  The control circuit 64 determines which of the output terminal 13A and the output terminal 13B to supply the current to control the output voltages VOUT1 and VOUT2 to a certain voltage, and the control signal VN of the nMOS switch 27 becomes L level. At this time, either the control signal VP1 or the control signal VP2 is set to the L level.

  As the electric power that can be supplied from the thermoelectric power generation element TEG is larger, the current of the secondary winding 16B of the transformer 17 can be increased, so that larger electric power can be output to the output terminal 13A or the output terminal 13B. By using the inductance of the secondary winding 16B of the transformer 17, it is possible to operate as a boost mode that is more efficient than the startup mode that operates as a blocking oscillator.

  According to this embodiment, even with a low input voltage, a high output voltage can be obtained through the Dickson booster, so that the control circuit 54 can be started with a low input voltage. Further, in the boost mode, high efficiency voltage boosting is possible using the inductance of the secondary side winding 16B of the transformer 17, so that a high output voltage can be obtained from a low input voltage. By sharing the same power supply TEG as the power supply for the blocking oscillator and the power supply for the switching boost converter, the area and volume can be reduced. In addition, the high voltage selection circuit 61 applies the higher one of the voltage VOUT1 and the output voltage VOUT2 to the bulk terminals of the pMOS switch 28A and the pMOS switch 28B, thereby the pMOS switch 28A and the pMOS switch 28B. The body diode (a body diode (not shown) connected in series in the opposite direction to the body diode shown in FIG. 10) can be turned on and current can be prevented from flowing backward.

  Note that the present invention is not limited to the above-described embodiment as it is, and can be embodied by modifying the constituent elements without departing from the scope of the invention in the implementation stage. In addition, various inventions can be formed by appropriately combining a plurality of components disclosed in the embodiment. For example, some components may be deleted from all the components shown in the embodiment. Furthermore, components in different embodiments may be combined as appropriate.

1, 2, 3: Input terminals 12, 22, 32, 42, 52, 62: DC-DC converters 14, 24, 34, 44, 64: Control circuits 13, 13A, 13B: Output terminals 15 ,: Native nMOS switches 16A: Primary winding 16B: Secondary winding 17: Transformers 28, 28A, 28B: pMOS switch 27: nMOS switch 61: high voltage selection circuits TEG1, TEG2, TEG: thermoelectric generator PV: solar battery SW: switch Load, Load1, Load2: Load devices D, D2, D3: Diode L: Inductors CIN1, CIN2, CIN3, C1, C2, COUT, COUT1, COUT2: Capacitor R1: Resistance

Claims (11)

An oscillator including a transformer including a primary winding supplied with a first input voltage and a secondary winding supplied with a second input voltage;
One end is connected to one end of the first rectifying element to which the third input voltage is supplied, the second rectifying element connected in series to the first rectifying element, and one end of the output side of the secondary winding, and the other end A first booster circuit including a capacitance element connected to a terminal between the first rectifier element and the second rectifier element;
A first switch connecting the secondary winding of the transformer and one end of the output side of the secondary winding to a reference potential, and a first switch connected to the one end of the output side of the secondary winding A second booster circuit including three rectifying elements;
A control circuit that operates using the output voltage of the first booster circuit and controls the first switch of the second booster circuit;
DC-DC converter provided with. The DC-DC converter according to claim 1, wherein the control circuit is activated by an output voltage of the first booster circuit and stops the oscillator after activation. The oscillator is
A second capacitance element having one end connected to a terminal on the output side of the secondary winding;
A second switch having a control terminal connected to the other end of the second capacitance element and one end connected to a terminal on the output side of the primary winding;
The DC-DC converter according to claim 2, wherein the control circuit controls the second switch. The DC-DC converter according to claim 3, wherein the second switch is a native or depletion type transistor. The DC-DC converter according to any one of claims 1 to 4, wherein the first input voltage and the third input voltage are the same voltage. The DC-DC converter according to any one of claims 1 to 4, wherein the third input voltage is a ground voltage. The DC-DC converter according to any one of claims 1 to 4, wherein the first input voltage, the second input voltage, and the third input voltage are the same voltage. The DC-DC converter according to any one of claims 1 to 7, wherein an output terminal of the first booster circuit is connected to an output-side terminal of the third rectifying element. The third rectifying element is a first MOS transistor;
The second booster circuit includes a second MOS transistor connected in parallel with the first MOS transistor at one end on the output side of the secondary winding.
The DC-DC converter according to any one of claims 1 to 8, wherein the control circuit controls the first MOS transistor and the second MOS transistor. The DC-DC converter according to any one of claims 1 to 4, wherein the third input voltage is higher than the first input voltage and the second input voltage. At least one power source for generating the first input voltage, the second input voltage, and the third input voltage;
DC-DC converter according to any one of claims 1 to 10,
Power supply equipment.