JP2019103325A - Motor controller - Google Patents

Abstract

To provide a motor controller capable of reducing resonance level by reducing ripples on a motor drive current before electrical resonance occurs due to the power supply for outputting DC power.SOLUTION: The motor controller has a control unit that includes a carrier frequency control unit. The carrier frequency control unit is configured so as to, when the drive rotation speed of a motor 50 is included in a predetermined rotation speed range Rr including a resonant rotational speed as the number of revolutions of the motor 50, and when the resonance frequency of the power supply matches the frequency of the drive current for the motor 50 at which the resonance occurs electrically, set the carrier frequency Fc of pulse width modulation control to be higher than the case where the drive rotation speed is not included in the rotation speed range Rr.SELECTED DRAWING: Figure 7

Description

  The present invention relates to a motor control device.

  The electric compressor described in Patent Document 1 changes the carrier frequency (carrier frequency) in accordance with the driving rotational speed of the compression mechanism. Specifically, the motor-driven compressor described in Patent Document 1 relates the drive rotational speed or drive rotational speed of the compression mechanism so that the carrier frequency (carrier frequency) decreases as the drive rotational speed of the compression mechanism increases. The carrier frequency (carrier frequency) is changed according to the physical quantity. Thus, the electric compressor described in Patent Document 1 attempts to reduce the switching loss of the inverter.

  The control device for an inverter described in Patent Document 2 includes a motor rotation speed arrival determination unit and a carrier frequency change unit. The motor rotational speed attainment judging means changes the motor rotational speed to a resonance motor rotational speed which is the motor rotational speed when the sideband wave component of the first carrier frequency in the inverter output matches the structural resonant frequency of the motor. On the other hand, it is determined whether or not it has approached a predetermined range. The carrier frequency changing means changes the carrier frequency of the inverter and determines that the first carrier frequency is different from the first carrier frequency, when it is determined by the motor rotational speed attainment determining means that the actual motor rotational speed has approached within the predetermined range. Set to the carrier frequency of 2. Thus, the controller of the inverter described in Patent Document 2 changes the carrier frequency before the motor rotational speed matches the resonant motor rotational speed, and the carrier side band wave component is the structural resonant frequency of the motor (the resonant frequency of the stator Trying to avoid meeting the).

JP, 2014-020266, A JP, 2009-284719, A

  However, in any of the motor-driven compressor described in Patent Document 1 and the control device for an inverter described in Patent Document 2, electrical resonance caused by a power supply outputting DC power is not considered. In particular, when the carrier frequency is set low as in the electric compressor described in Patent Document 1, ripples of drive current of the motor become large. As the ripple of the drive current of the motor is larger when the electrical resonance occurs, the power source or the like is more likely to generate heat. In addition, as the ripple of the drive current of the motor is larger when the electrical resonance occurs, the power supply and the like are more likely to be adversely affected. Furthermore, when electrical resonance occurs, the greater the ripple of the drive current of the motor, the more likely it is that the motor noise (abnormal noise) will occur.

  The present invention has been made in view of such circumstances, and reduces the ripple of the drive current of the motor to reduce the resonance level before the occurrence of electrical resonance caused by the power supply outputting DC power. An object of the present invention is to provide a motor control device that can be reduced.

  A motor control device according to the present invention includes: a power supply that outputs DC power; and a power converter including a plurality of switching elements that convert the DC power output from the power supply into AC power and outputs the AC power to the motor. And a controller configured to control opening and closing of the plurality of switching elements of the converter by pulse width modulation control. The control device is configured to perform predetermined rotation including a resonant rotational speed which is a rotational speed of the motor when an electrical resonance frequency of the power source matches a frequency of a drive current of the motor and an electrical resonance occurs. A carrier frequency control unit which sets the carrier frequency of the pulse width modulation control higher than when the drive rotational speed is not included in the rotational speed range when the drive rotational speed of the motor is included in several ranges. Prepare.

  According to the above-described motor control device, the control device includes the carrier frequency control unit. Therefore, the above-mentioned motor control device can reduce the ripple of the drive current of a motor, and can reduce a resonance level, before the electric resonance resulting from the power supply which outputs direct-current power occurs.

It is a block diagram which shows an example of a motor control apparatus. It is a block diagram which shows an example of a rotor. It is a block diagram which shows an example of a control apparatus. It is a block diagram which shows an example of the control block of a control apparatus. It is a figure which shows an example of the relationship between a torque and armature current. It is a figure which shows an example of the measurement result which measured the impedance (internal resistance) of a power supply. It is a flowchart which shows an example of the control procedure by a carrier frequency control part. It is a figure which shows an example of the amplitude of the fundamental wave component contained in the drive current of a motor, and a harmonic component. It is a flowchart which shows an example of the procedure at the time of calculating the resonant rotation speed (2nd resonance rotation speed) of an electric motor. It is a figure which shows an example of the relationship between the resonant frequency of a power supply, and the rotation speed of a motor. It is a figure which concerns on a reference form and shows an example of the relationship between the carrier frequency before changing a carrier frequency, and the rotation speed of an electric motor. It is a figure which concerns on a reference form and shows an example of the relationship between the carrier frequency after changing a carrier frequency, and the rotation speed of an electric motor.

  Embodiments are described herein based on the drawings. In the drawings, common parts are denoted by common reference numerals, and redundant description is omitted in the present specification. The drawings are conceptual diagrams and do not specify the dimensions of the detailed structure.

<Schematic Configuration of Motor Control Device 10>
As shown in FIG. 1, the motor control device 10 of the present embodiment includes a power supply 20, a smoothing capacitor 30, a power converter 40, and a control device 60. Further, the electric motor 50 is electrically connected to the power converter 40.

  The power supply 20 outputs DC power. The power supply 20 only needs to output DC power, and the type is not limited. For example, a lead storage battery (battery), a lithium ion battery, a power generation device (for example, a fuel cell) or the like can be used as the power supply 20. Further, the power supply 20 can also boost low-voltage DC power using, for example, a known boost converter or the like. In this case, the power supply 20 can include, for example, a known boost chopper converter or the like.

  The smoothing capacitor 30 smoothes the DC power output from the power supply 20. The positive electrode side 20 p of the power supply 20 is connected to the positive electrode side 30 p of the smoothing capacitor 30. The negative electrode side 20n of the power supply 20 is connected to the negative electrode side 30n of the smoothing capacitor 30, and is connected to the power ground (the reference potential of the circuit on the high voltage side including the power supply 20). For example, an electrolytic capacitor can be used as the smoothing capacitor 30. The direct current power supplied from the power supply 20 is smoothed by the smoothing capacitor 30 to reduce ripples.

  The power converter 40 converts the DC power (in the present embodiment, DC power smoothed by the smoothing capacitor 30) output from the power source 20 into AC power, and outputs the converted AC power to the motor 50. As shown in FIG. 1, the power converter 40 includes a plurality of switching elements (three pairs of switching elements 41 in the present embodiment), which are full bridge connected. In each of the three pairs of switching elements 41, positive electrode side switching element 4xp connected to positive electrode side 20p of power source 20 and negative electrode side switching element 4xn connected to negative electrode side 20n of power source 20 are connected in series There is. The power converter 40 of the present embodiment is a three-phase power converter, and x is any one of u, v, and w. For example, the positive electrode side switching element 4up indicates a U phase positive electrode side switching element, and the negative electrode side switching element 4un indicates a U phase negative electrode side switching element.

  A well-known power switching element can be used for the positive electrode side switching element 4xp and the negative electrode side switching element 4xn. For example, a well-known insulated gate bipolar transistor (IGBT: Insulated Gate Bipolar Transistor), a field effect transistor (FET: Field Effect Transistor), etc. can be used for the positive electrode side switching element 4xp and the negative electrode side switching element 4xn.

  As shown in FIG. 1, each of the plurality of (three) positive electrode side switching elements 4xp includes a control terminal 4g, an input terminal 4c, an output terminal 4e, and a free wheeling diode 4d. For example, in an insulated gate bipolar transistor (IGBT), the control terminal 4g corresponds to a gate terminal, the input terminal 4c corresponds to a collector terminal, and the output terminal 4e corresponds to an emitter terminal. In the field effect transistor (FET), the control terminal 4g corresponds to a gate terminal, the input terminal 4c corresponds to a drain terminal, and the output terminal 4e corresponds to a source terminal. The control terminal 4g is connected to the control device 60 via the drive circuit 61b. Each of the plurality of (three) positive electrode side switching elements 4 xp is controlled to open and close based on the drive signal output from the control device 60.

  A voltage between the control terminal 4g and the output terminal 4e is a control voltage Vge. For example, when the control voltage Vge is at a low level (state equal to or lower than a predetermined voltage value), the input terminal 4c and the output terminal 4e are controlled to be in an open state in which the connection is electrically disconnected. On the other hand, when the control voltage Vge is at the high level (the state where the predetermined voltage value is exceeded), the input terminal 4c and the output terminal 4e are controlled to be in a closed state in which electrical conduction is established.

  For example, a body diode (parasitic diode) of a switching element can be used as the free wheeling diode 4d. Also, instead of the body diode, a free wheeling diode may be separately provided and connected in parallel between the input terminal 4c and the output terminal 4e. The free wheeling diode 4d forms a current path from the output terminal 4e side to the input terminal 4c side when the switching element is in the open state. Thereby, the switching element can be protected from the reverse current generated along with the opening and closing of the switching element.

  The above description of the plurality (three) of the positive electrode side switching elements 4xp can be similarly applied to the plurality of (three) negative electrode side switching elements 4xn. The control device 60 controls the power converter 40 by opening and closing a plurality of switching elements of the power converter 40 (the positive side switching element 4xp and the negative side switching element 4xn constituting the three pairs of switching elements 41). Do.

  For example, based on a command from the control device 60, the power converter 40 selects one positive electrode side switching element 4xp of the plurality (three) positive electrode side switching elements 4xp and a plurality (three) negative electrode side switching elements One negative electrode side switching element 4xn of 4xn is closed, and the other switching element is open. The phases (U phase, V phase, W phase) of the positive electrode side switching element 4xp and the negative electrode side switching element 4xn to be closed are different. The power converter 40 converts the DC power (DC power smoothed by the smoothing capacitor 30) output from the power supply 20 into AC power by sequentially changing the combination of switching elements that the control device 60 turns to the closed state. be able to.

  As shown in FIG. 1, an output terminal 42x is provided between the positive electrode side switching element 4xp and the negative electrode side switching element 4xn. The output terminal 42x and the phase terminal 43x of the motor 50 are electrically connected by a power cable 44x. The power cable 44 x supplies the AC power converted by the power converter 40 to the motor 50. Here, x is any one of u, v and w.

  The motor 50 includes a stator 51 and a rotor 52. The electric motor 50 of the present embodiment is a radial air gap type cylindrical electric motor in which the stator 51 and the rotor 52 are disposed coaxially. The motor 50 may be an axial gap type cylindrical motor. Alternatively, the motor 50 may be an inner type cylindrical motor in which the rotor 52 is provided on the inner side (the axial center side of the motor 50) of the stator 51, and the rotor 52 is provided outside the stator 51. It may be an outer type cylindrical motor provided.

  The stator 51 includes a stator core (not shown) in which a plurality of slots are formed, and an armature winding (U-phase coil 51 u, V-phase coil 51 v, W-phase coil 51 w). The stator core is formed by laminating a plurality of thin electromagnetic steel plates (for example, silicon steel plates) in the axial direction. An armature winding is wound around the plurality of slots. The armature winding is formed by winding a conductor (coil) such as copper, and the conductor surface is covered with an insulating layer such as enamel. The cross-sectional shape of the armature winding can be any cross-sectional shape. For example, as the armature winding, conductors (coils) having various cross-sectional shapes such as round wires having a circular cross-sectional shape and square wires having a cross-sectional polygonal shape can be used. Moreover, the armature winding can also use the parallel thin wire which combined the several thinner coil strand. When parallel thin wires are used, the eddy current loss generated in the armature winding is reduced as compared with the case of a single wire, and the efficiency of the motor 50 is improved. In addition, since the force required to form the coil can be reduced, the formability of the coil is improved and the coil can be easily manufactured.

  The armature winding can be wound by known methods such as distributed winding (eg, concentric winding, wave winding, lap winding, etc.) or concentrated winding. Further, as shown in FIG. 1, the armature winding (U-phase coil 51 u, V-phase coil 51 v, W-phase coil 51 w) can be connected by Y connection. In the figure, the neutral point is indicated by a neutral point 51n. The armature winding (U-phase coil 51 u, V-phase coil 51 v, W-phase coil 51 w) can also be connected by Δ connection.

  As shown in FIG. 2, the rotor 52 is provided with a rotor core 52a, permanent magnets 52b for a predetermined number of magnetic pole pairs (four in the present embodiment, and eight magnets), and a shaft 52c. . This figure is a schematic view of the rotor 52 seen in the axial direction (the direction perpendicular to the sheet of the figure), and their arrangement is schematically shown. The rotor core 52a is formed in a cylindrical shape by laminating a plurality of thin electromagnetic steel plates (for example, silicon steel plates) in the axial direction (the direction perpendicular to the sheet of the drawing). The rotor core 52a is provided with a shaft 52c, and the shaft 52c passes through the axial center of the rotor core 52a along the axial direction. Both axial ends of the shaft 52c are rotatably supported by a bearing member (not shown) such as a bearing.

  A plurality (eight) of permanent magnets 52b are embedded in the rotor core 52a. Specifically, the rotor core 52a is provided with a plurality of magnet housings (not shown) at equal intervals in the circumferential direction. In the plurality of magnet housing portions, permanent magnets 52b of a predetermined number of magnetic pole pairs (eight in the present embodiment, which are four magnetic pole pairs) are embedded. For example, a known ferrite magnet or rare earth magnet can be used as the permanent magnet 52b. The manufacturing method of the permanent magnet 52b is not limited. For example, a resin-bonded magnet or a sintered magnet can be used as the permanent magnet 52b. The resin bonded magnet can be formed, for example, by mixing ferrite raw material magnet powder and a resin, and casting and forming the same on the rotor core 52a by injection molding or the like. The sintered magnet can be formed, for example, by press-forming a raw material magnet powder of a rare earth material in a magnetic field and then baking it at a high temperature. The number of slots of the stator 51 and the number of magnetic poles of the rotor 52 are not limited.

  Controller 60 controls a power conversion system including power converter 40. Further, the control device 60 controls the switching of the plurality of switching elements (the positive side switching element 4xp and the negative side switching element 4xn) of the power converter 40 by pulse width modulation (PWM) control. As shown in FIG. 3, the control device 60 includes a known central processing unit 60a, a storage device 60b and an input / output interface 60c, which are connected via a bus 60d.

  The central processing unit 60a is a CPU: Central Processing Unit, and can perform various arithmetic processing. The storage device 60b includes a first storage device 60b1 and a second storage device 60b2. The first storage device 60b1 is a volatile storage device (RAM: Random Access Memory), and the second storage device 60b2 is a non-volatile storage device (ROM: Read Only Memory).

  Further, as shown in FIG. 1, the control device 60 includes a DC voltage detector 61a, a drive circuit 61b, a current detector 61c, and a position detector 61d. The DC voltage detector 61 a detects the DC voltage of the DC power smoothed by the smoothing capacitor 30. Specifically, the DC voltage detector 61 a divides the DC voltage by, for example, a plurality of resistors whose resistance values are known, and outputs the divided DC voltage to the control device 60. Control device 60 obtains a DC voltage value divided by a known A / D converter (not shown) or the like, and DC voltage of DC power smoothed by smoothing capacitor 30 (input to power converter 40 DC voltage) can be obtained.

  The drive circuit 61 b is a drive circuit that amplifies a drive signal output from the control device 60, and can use, for example, a known driver circuit. In FIG. 1, the connection between the control terminal 4g of each switching element of the power converter 40 and the drive circuit 61b is omitted.

  The current detector 61 c detects an output current output from the power converter 40. In the present embodiment, the current detector 61c is provided to the power cable 44u and the power cable 44v, and detects the U-phase current Iu and the V-phase current Iv. In this specification, the U-phase current Iu detected by the current detector 61c is referred to as a U-phase current detection value Iu_fb, and the V-phase current Iv detected by the current detector 61c is referred to as a V-phase current detection value Iv_fb. The W-phase current Iw can be calculated by subtracting the U-phase current detection value Iu_fb and the V-phase current detection value Iv_fb from zero. As the current detector 61c, a known current detector (for example, a current detector using a current transformer, a current detector using a shunt resistor, etc.) can be used.

  The position detector 61 d detects the position of the rotor 52 with respect to the stator 51. The position detector 61 d can use a known position detector (for example, a resolver, an encoder, a Hall sensor, etc.). In addition, the control apparatus 60 can provide various detectors other than the detector mentioned above.

  The central processing unit 60a shown in FIG. 3 reads the control program of the power converter 40 stored in the second storage device 60b2 to the first storage device 60b1, and executes the control program. In addition, the above-described detection value and the like are input to the control device 60 via the insulating unit (not shown) and the input / output interface 60c. The central processing unit 60a outputs an open / close signal to each switching element of the power converter 40 via the input / output interface 60c, the insulator (not shown) and the drive circuit 61b shown in FIG. Control opening and closing. The insulating unit electrically isolates the low voltage side circuit including the control device 60 from the high voltage side circuit including the power supply 20. For example, a known photocoupler or the like can be used as the insulating portion.

<Control by controller 60>
As shown in FIG. 4, when considered as a control block, the control device 60 of the present embodiment is a three-phase / two-phase conversion unit 71, a rotation number calculation unit 72, a current command value setting unit 73, and a current control unit 74, a two-phase / three-phase converter 75, a carrier frequency controller 76, and a pulse width modulation signal generator 77.

  As shown in FIG. 2, the main magnetic flux direction of the permanent magnet 52b is taken as a d-axis direction, and the direction electrically orthogonal to the d-axis direction is taken as a q-axis direction. The voltage equation in the dq coordinate system of the motor 50 can be expressed by the following equation 1.

  However, the voltage in the d-axis direction is represented by the d-axis voltage Vd, and the voltage in the q-axis direction is represented by the q-axis voltage Vq. Further, each winding resistance of the armature winding (U-phase coil 51 u, V-phase coil 51 v, W-phase coil 51 w) is represented by a winding resistance R. Further, the d-axis inductance which is the inductance in the d-axis direction is represented by the d-axis inductance Ld, and the q-axis inductance which is the inductance in the q-axis direction is represented by the q-axis inductance Lq. Further, the current in the d-axis direction is represented by the d-axis current Id, and the current in the q-axis direction is represented by the q-axis current Iq. Furthermore, the angular velocity of the rotor 52 is represented by the angular velocity ω, and the induced voltage constant is represented by the induced voltage constant Φ. Also, the differential operator is represented by the differential operator p (= d / dt).

  The controller 60 controls the power converter 40 by setting the d-axis current command value Id_ref and the q-axis current command value Iq_ref. Specifically, control device 60 controls power converter 40 by setting d-axis voltage command value Vd_ref and q-axis voltage command value Vq_ref based on d-axis current command value Id_ref and q-axis current command value Iq_ref. Do. However, the d-axis current command value Id_ref is a current command value in the d-axis direction, and the q-axis current command value Iq_ref is a current command value in the q-axis direction. The d-axis voltage command value Vd_ref refers to a voltage command value in the d-axis direction, and the q-axis voltage command value Vq_ref refers to a voltage command value in the q-axis direction.

(Three-phase / two-phase converter 71)
The three-phase / two-phase conversion unit 71 calculates the d-axis current calculation value Id_fb and the q-axis current calculation value Iq_fb. The d-axis current calculation value Id_fb is a current calculation value in the d-axis direction, and the q-axis current calculation value Iq_fb is a current calculation value in the q-axis direction. As shown in FIG. 4, the U-phase current detection value Iu_fb and the V-phase current detection value Iv_fb detected by the current detector 61 c are input to the three-phase / two-phase conversion unit 71. As described above, the W-phase current Iw can be calculated by subtracting the U-phase current detection value Iu_fb and the V-phase current detection value Iv_fb from zero, respectively. The detected value (calculated value) of the W-phase current Iw is taken as the W-phase current detected value Iw_fb. Further, the rotation angle (rotational position θ) of the rotor 52 detected by the position detector 61 d is input to the three-phase / two-phase conversion unit 71.

  The three-phase / two-phase converter 71 calculates the d-axis current based on the following equation 2 using the U-phase current detection value Iu_fb, the V-phase current detection value Iv_fb and the W-phase current detection value Iw_fb, and the rotational position θ. The value Id_fb and the q-axis current calculation value Iq_fb are calculated. The rotational position θ can also be estimated from, for example, temporal changes in the U-phase current detection value Iu_fb, the V-phase current detection value Iv_fb, and the W-phase current detection value Iw_fb. The rotational position θ can also be estimated, for example, from the change with time of the induced voltage of each phase.

(Rotation speed calculation unit 72)
The rotation speed calculation unit 72 calculates the rotation speed of the rotor 52 (the driving rotation speed of the electric motor 50). As shown in FIG. 4, the rotation angle calculation unit 72 receives the rotation angle (rotation position θ) of the rotor 52 detected by the position detector 61 d. The rotation speed calculation unit 72 can calculate, for example, the rotation speed of the rotor 52 by time-differentiating the rotation position θ. The control device 60 can also measure the rotational speed of the rotor 52 using a known rotational speed detector. The rotational speed of the rotor 52 calculated by the rotational speed calculation unit 72 is represented by a rotational speed calculation value Nm_fb. The rotation speed calculation value Nm_fb calculated by the rotation speed calculation unit 72 is output to a carrier frequency control unit 76 described later.

(Current command value setting unit 73)
Current command value setting unit 73 calculates d-axis current command value Id_ref and q-axis current command value Iq_ref. The method of calculating the d-axis current command value Id_ref and the q-axis current command value Iq_ref is not limited. The current command value setting unit 73 preferably sets the d-axis current command value Id_ref and the q-axis current command value Iq_ref so that the armature current is minimized with respect to the required torque.

  FIG. 5 shows an example of the relationship between torque and armature current. A curve L11 shows an example of the relationship between the d-axis current command value Id_ref and the q-axis current command value Iq_ref (the relationship between the current vectors of the armature current) when the required torque (torque command value Trq_ref) is constant. The required torque is obtained by setting the current vector of the armature current on the curve L11. An arrow L12 indicates a current vector at which the armature current is minimized when obtaining the required torque indicated by the curve L11. A straight line L13 indicates a tangent at the extreme value of the curve L11. That is, when the current vector of the armature current is orthogonal to the tangent line indicated by the straight line L13, the armature current at the time of obtaining the required torque is minimized.

  When the required torque increases, the torque command value Trq_ref is increased, and the curve L11 moves to the curve L11a. Arrow L12a indicates a current vector at which the armature current is minimized at this time. On the other hand, when the required torque decreases, the torque command value Trq_ref is reduced, and the curve L11 moves to the curve L11 b. Arrow L12 b indicates a current vector at which the armature current is minimized at this time. The horizontal axis in the figure indicates the d axis, and the vertical axis indicates the q axis.

  Control device 60 sets torque command value Trq_ref according to the required torque. As shown in FIG. 4, torque command value Trq_ref is input to current command value setting unit 73. Current command value setting unit 73 sets d-axis current command value Id_ref and q-axis current command value Iq_ref such that the armature current is minimized with respect to torque command value Trq_ref. The d-axis current command value Id_ref and the q-axis current command value Iq_ref corresponding to the required torque (torque command value Trq_ref) are calculated in advance and converted into, for example, a map, a table, a relational expression (polynomial), etc. It is stored in the second storage device 60b2 shown. The current command value setting unit 73 reads a map and the like from the second storage device 60b2 to the first storage device 60b1 at startup, together with the control program of the power converter 40. Thus, current command value setting unit 73 can easily set d-axis current command value Id_ref and q-axis current command value Iq_ref corresponding to the required torque (torque command value Trq_ref).

(Current control unit 74)
The current control unit 74 calculates the d-axis voltage command value Vd_ref based on the d-axis current command value Id_ref. Further, the current control unit 74 calculates a q-axis voltage command value Vq_ref based on the q-axis current command value Iq_ref. Specifically, the d-axis voltage command value Vd_ref is calculated such that the d-axis current calculation value Id_fb matches the d-axis current command value Id_ref. Further, the q-axis voltage command value Vq_ref is calculated such that the q-axis current calculation value Iq_fb matches the q-axis current command value Iq_ref. The current control unit 74 only needs to calculate the d-axis voltage command value Vd_ref and the q-axis voltage command value Vq_ref, and the configuration thereof is not limited.

  As shown in FIG. 4, in the present embodiment, the current control unit 74 includes a subtractor 74a, a PI control unit 74b, a subtractor 74c, and a PI control unit 74d. The d-axis current command value Id_ref and the d-axis current calculation value Id_fb are input to the subtractor 74a. The subtractor 74 a subtracts the d-axis current calculation value Id_fb from the d-axis current command value Id_ref to calculate a deviation ΔId. The deviation ΔId calculated by the subtractor 74a is output to the PI control unit 74b. In the present embodiment, the PI control unit 74b performs proportional control and integral control such that the d-axis current calculation value Id_fb matches the d-axis current command value Id_ref.

  The PI control unit 74b includes a proportional operator 74b1, an integral operator 74b2, and an adder 74b3. The proportional calculator 74b1 outputs an operation result obtained by multiplying the deviation ΔId by the proportional gain Kpd. The integral calculator 74b2 outputs an operation result obtained by multiplying the integral gain obtained by integrating the deviation ΔId by the integral gain Kid. The adder 74 b 3 adds the calculation result of the proportional calculator 74 b 1 and the calculation result of the integral calculator 74 b 2. Then, the PI control unit 74b outputs the calculation result of the adder 74b3 as the d-axis voltage command value Vd_ref. The PI control unit 74b can also include a differential calculator that outputs a calculation result obtained by multiplying the differential gain obtained by differentiating the deviation ΔId by the differential gain. That is, the PI control unit 74b can be a PID control unit that performs proportional control, integral control, and differential control. In this case, the adder 74b3 adds the calculation result of the proportional calculator 74b1, the calculation result of the integral calculator 74b2, and the calculation result of the differential calculator.

  On the other hand, the q-axis current command value Iq_ref and the q-axis current calculation value Iq_fb are input to the subtractor 74c. The subtractor 74c subtracts the q-axis current calculation value Iq_fb from the q-axis current command value Iq_ref to calculate a deviation ΔIq. The deviation ΔIq calculated by the subtractor 74c is output to the PI control unit 74d. In the present embodiment, the PI control unit 74d performs proportional control and integral control such that the q-axis current calculation value Iq_fb matches the q-axis current command value Iq_ref.

  The PI control unit 74d includes a proportional calculator 74d1, an integral calculator 74d2, and an adder 74d3. The proportional calculator 74d1 outputs an operation result obtained by multiplying the deviation ΔIq by the proportional gain Kpq. The integral calculator 74d2 outputs an operation result obtained by multiplying the integral gain obtained by integrating the deviation ΔIq by the integral gain Kiq. The adder 74d3 adds the calculation result of the proportional calculator 74d1 and the calculation result of the integral calculator 74d2. Then, the PI control unit 74d outputs the calculation result of the adder 74d3 as the q-axis voltage command value Vq_ref. Similar to the PI control unit 74b, the PI control unit 74d can also include a differential operator that outputs an operation result obtained by multiplying a differential value obtained by differentiating the deviation ΔIq by a differential gain.

  Thus, for example, the current control unit 74 performs at least proportional control (P control) and integral control (I control) of proportional control (P control), integral control (I control) and differential control (D control). The d-axis voltage command value Vd_ref and the q-axis voltage command value Vq_ref can be calculated. The proportional gain Kpd and the integral gain Kid, and the proportional gain Kpq and the integral gain Kiq are stored in the second storage device 60b2 shown in FIG. These control gains are read out from the second storage device 60b2 to the first storage device 60b1 at startup together with the control program of the power converter 40.

  As the proportional gain Kpd is increased, the deviation ΔId can be reduced in a short time. Further, as the integral gain Kid is increased, the offset (steady state deviation) due to the deviation ΔId can be eliminated in a short time. Furthermore, the vibration of the deviation ΔId can be converged in a short time as the differential gain is increased. These control gains may be obtained in advance, for example, by simulation, verification by a real machine, or the like. The same applies to the control gains (proportional gain Kpq, integral gain Kiq and differential gain) for calculating the q-axis voltage command value Vq_ref.

  The d-axis voltage command value Vd_ref can be expressed by the following equation 3 using the proportional gain Kpd and the integral gain Kid. Further, the q-axis voltage command value Vq_ref can be expressed by the following equation 4 using the proportional gain Kpq and the integral gain Kiq. Each represents the Laplace operator as the Laplace operator s.

(Two-phase / three-phase converter 75)
The two-phase / three-phase conversion unit 75 generates three-phase voltage command values (U-phase voltage command value Vu_ref, V-phase voltage command) from two-phase voltage command values (d-axis voltage command value Vd_ref and q-axis voltage command value Vq_ref). A value Vv_ref and a W-phase voltage command value Vw_ref) are calculated. As shown in FIG. 4, the d-axis voltage command value Vd_ref and the q-axis voltage command value Vq_ref calculated by the current control unit 74 are input to the two-phase / three-phase conversion unit 75. Further, the rotational angle (rotational position θ) of the rotor 52 detected by the position detector 61 d is input to the two-phase / three-phase conversion unit 75. The two-phase / three-phase conversion unit 75 performs reverse conversion of the dq coordinate axis using the d-axis voltage command value Vd_ref, the q-axis voltage command value Vq_ref, and the rotational position θ to generate a U-phase voltage command value Vu_ref, a V-phase voltage command A value Vv_ref and a W-phase voltage command value Vw_ref are calculated.

(Carrier frequency control unit 76)
FIG. 6 shows an example of the measurement result of measuring the impedance (internal resistance) of the power supply 20. The figure measures the impedance (internal resistance) of the lithium ion battery which is one form of the power supply 20 by the electrochemical impedance method, and illustrates the measurement result on the complex plane. The electrochemical impedance method applies an alternating voltage with a constant amplitude to the power supply 20 while changing the frequency, and measures impedance (internal resistance) from the response current. The horizontal axis of the figure shows the real axis Re, and the vertical axis shows the imaginary axis Im. Further, a curve L21 shows an example of the measurement result. The method of measuring the impedance (internal resistance) of the power supply 20 is not limited to the electrochemical impedance method, and the impedance (internal resistance) of the power supply 20 can be obtained by various known methods. Further, the impedance (internal resistance) of the power supply 20 can also be estimated from the output voltage of the power supply 20, the output current, and the like.

  The equivalent circuit of a lithium ion battery can be represented by a series-parallel circuit of a resistor and a capacitor. Therefore, when the frequency of the applied alternating voltage matches the electrical resonance frequency Fb0 determined by the equivalent circuit, electrical resonance may occur. The resonance point P10 shown in FIG. 6 indicates the impedance (internal resistance) at this time, and corresponds to a point at which the curve L21 is maximized. In addition, what was mentioned above is not limited to a lithium ion battery. What has been described above is the same as in the case where the power supply 20 is, for example, a lead storage battery (battery), a power generation device (for example, a fuel cell), etc., except that the characteristics and equivalent circuit are different.

  The control device 60 of the present embodiment controls the opening and closing of the plurality of switching elements (positive electrode side switching element 4xp and negative electrode side switching element 4xn) of the power converter 40 to thereby drive the motor current of the motor 50 (armature winding described above). A wire (U-phase coil 51u, V-phase coil 51v, W-phase coil 51w) is controlled in a sine wave manner. Therefore, similar to the above description, when the frequency Fm0 of the drive current (armature current) of the motor 50 matches the electrical resonance frequency Fb0 determined by the equivalent circuit of the power supply 20, the possibility of the occurrence of electrical resonance There is.

  As the ripples of the drive current (armature current) of the motor 50 increase when electrical resonance occurs, the power supply 20 is more likely to generate heat. In addition, the larger the ripple of the drive current (armature current) of the motor 50 when electrical resonance occurs, the more likely the power supply 20 is adversely affected. Furthermore, when the electrical resonance occurs, the larger the ripple of the drive current (armature current) of the motor 50, the more likely the noise (noise) of the motor 50 is generated.

  On the other hand, the ripple of the drive current (armature current) of the motor 50 can be reduced as the carrier frequency Fc of pulse width modulation (PWM) control is set higher. Therefore, the resonance level is reduced by setting the carrier frequency Fc to a high level before the occurrence of the electrical resonance as compared with the case where the electrical resonance does not occur. As a result, even if electrical resonance caused by the power supply 20 occurs, heat generation in the power supply 20 is suppressed. Further, even if electrical resonance caused by the power supply 20 occurs, the adverse effect of the power supply 20 is suppressed. Furthermore, even if electrical resonance caused by the power supply 20 occurs, the noise (noise) of the motor 50 is suppressed. What has been described above for the power supply 20 applies to the smoothing capacitor 30 as well.

  The control device 60 of the present embodiment includes a carrier frequency control unit 76. The carrier frequency control unit 76 controls the pulse width modulation (PWM) when the drive rotational speed of the electric motor 50 is included in the predetermined rotational speed range Rr compared to when the drive rotational speed is not included in the rotational speed range Rr. The carrier frequency Fc of is set high. The rotational speed range Rr is a resonance rotational speed Rs which is the rotational speed of the motor 50 when the electrical resonance frequency Fb0 of the power supply 20 and the frequency Fm0 of the drive current of the motor 50 coincide and electrical resonance occurs. Including.

  Specifically, the carrier frequency control unit 76 acquires the driving rotational speed of the motor 50 (step S11 shown in FIG. 7). As described above, in the control device 60 of the present embodiment, the driving rotational speed of the electric motor 50 (the rotational speed of the rotor 52) is calculated by the rotational speed calculating unit 72 shown in FIG. Therefore, the carrier frequency control unit 76 acquires the drive rotation speed (rotation speed calculation value Nm_fb) of the motor 50 from the rotation speed calculation unit 72.

  Next, carrier frequency control unit 76 determines whether or not the driving rotational speed (rotational speed calculation value Nm_fb) of electric motor 50 is included in rotational speed range Rr (step S12 shown in FIG. 7). When the drive rotational speed (rotational speed calculation value Nm_fb) of electric motor 50 is included in rotational speed range Rr (in the case of “Yes” in step S12), carrier frequency control unit 76 sets carrier frequency Fc to second carrier frequency Fc2. It sets (step S13). Then, control ends once. On the other hand, when the drive rotational speed (rotational speed calculation value Nm_fb) of motor 50 is not included in rotational speed range Rr (in the case of “No” in step S12), carrier frequency control unit 76 sets carrier frequency Fc to the first carrier. The frequency Fc1 is set (step S14). Then, control ends once. The control shown in FIG. 7 is repeatedly executed each time a predetermined time passes.

  The first carrier frequency Fc1 is the carrier frequency Fc when the drive rotational speed of the motor 50 is not included in the rotational speed range Rr, and corresponds to the carrier frequency Fc when the electrical resonance does not occur. The second carrier frequency Fc2 is higher than the first carrier frequency Fc1. The second carrier frequency Fc2 can be set, for example, to several times (eg, twice) the first carrier frequency Fc1. For example, when the first carrier frequency Fc1 is 5 kHz, the second carrier frequency Fc2 can be set to 10 kHz.

  Further, as the carrier frequency Fc is set higher, the ripple of the drive current (armature current) of the motor 50 is reduced. Therefore, the second carrier frequency Fc2 may be set as high as possible from the viewpoint of reducing the ripple of the drive current (armature current). On the other hand, as the carrier frequency Fc is set higher, the switching loss in the plurality of switching elements (positive electrode side switching element 4xp and negative electrode side switching element 4xn) of the power converter 40 increases. Therefore, it is preferable to determine the second carrier frequency Fc2 in advance by grasping the relationship between the reduction degree of the ripple of the drive current (armature current) and the increase amount of the switching loss by simulation, verification by an actual device or the like.

  Specifically, when the drive rotational speed of motor 50 is included in rotational speed range Rr, carrier frequency control unit 76 determines the required reduction degree of the ripple of the drive current (armature current) and the plurality of switching elements (positive electrode It is preferable to set the carrier frequency Fc so as to satisfy both of the allowable switching loss of the side switching element 4xp and the negative side switching element 4xn). Thereby, the motor control device 10 can reduce the resonance level while suppressing an increase in switching loss in the plurality of switching elements (the positive electrode side switching element 4xp and the negative electrode side switching element 4xn).

  The resonant rotational speed Rs of the motor 50 preferably includes a first resonant rotational speed Rs1 calculated by dividing the resonant frequency Fb0 of the power source 20 by the number of pole pairs Pp of the motor 50. In this case, the first resonance rotational speed Rs1 can be expressed by the following equation 5. Thereby, the resonance level of the resonance by the fundamental wave component (when the electrical angle order n described later is 1) included in the driving current of the motor 50 is reduced.

  As described above, the resonance frequency Fb0 of the power supply 20 can be derived from the measurement result by measuring the impedance (internal resistance) of the power supply 20 in advance. Specifically, a point corresponding to the resonance point P10 shown in FIG. 6 is determined from the measurement result. The frequency of the AC voltage applied at this time corresponds to the resonance frequency Fb 0 of the power supply 20. The resonant frequency Fb0 of the power source 20 is an electrical angle, and the first resonant rotational speed Rs1 is calculated by dividing the resonant frequency Fb0 of the power source 20 by the pole pair number Pp of the motor 50. The number of pole pairs Pp is determined by the specification of the motor 50. For example, the motor 50 of the present embodiment is an eight-pole motor 50 in which the number of magnetic poles of the rotor 52 is eight, and the number of pole pairs Pp of this embodiment is four.

  In the present specification, the resonance frequency Fb0 is described using the impedance (internal resistance) of the power supply 20 alone as an example for simplification of the description, but the present invention is not limited to this. For example, a parallel circuit in which the power supply 20 and the smoothing capacitor 30 are connected in parallel is regarded as the power supply 20, and impedance (internal resistance) is measured as in the case of the power supply 20 alone, and the resonance frequency Fb0 is derived from the measurement result. You can also When the power supply 20 includes a boost converter, the impedance (internal resistance) of the power supply 20 including the boost converter can be measured, and the resonance frequency Fb0 can be derived from the measurement result.

  FIG. 8 shows an example of the amplitudes of the fundamental wave component and the harmonic component included in the drive current of the motor 50. In the figure, the drive current of the motor 50 is subjected to Fast Fourier Transform (FFT), and respective amplitudes of a fundamental wave component and a harmonic component included in the drive current are illustrated by a bar graph. The fast Fourier transform (FFT) is an algorithm for computing discrete Fourier transform at high speed on a computer, and can speed up computation processing for extracting a fundamental wave component and a harmonic component from a drive current. The horizontal axis of the figure indicates the order (electrical angle order n), and the vertical axis indicates the amplitude. The fundamental wave component of the drive current has an electrical angle order n of 1, and the harmonic component of the drive current has an electrical angle order n of 2 or more.

  Among the electrical angle orders n of the harmonic components included in the drive current of the motor 50, the electrical angle order serving as a factor of electrical resonance is taken as a target electrical angle order Tgt_n. For example, from the specification of the motor 50, the amplitude of the harmonic component (fifth harmonic component) of the order (electrical angle order n) 5 shown in FIG. 8 is larger than that of the motor having a different specification for comparison. Assume that 20 is prone to fever. In this case, the target electrical angle order Tgt_n can be set to five.

  The target electrical angle order Tgt_n can also be selected in consideration of an adverse effect (for example, deterioration or the like) in the power supply 20. Also, the target electrical angle order Tgt_n can be selected in consideration of the noise (noise) of the motor 50. That is, the target electrical angle order Tgt_n can be selected in consideration of at least one of heat generation and deterioration of the power supply 20 due to resonance, and noise (noise) of the motor 50 due to resonance. In addition, a smoothing capacitor 30 can be included in selecting the target electrical angle order Tgt_n. Furthermore, as shown in FIG. 8, the harmonic components included in the drive current of the motor 50 have remarkable amplitude magnitudes when the electrical angle order n is odd as compared to when the electrical angle order n is even. Become. Therefore, among the odd electrical angle orders n, the electrical angle order n in which the magnitude of the amplitude is prominent can be set as the target electrical angle order Tgt_n.

  When the target electrical angle order Tgt_n is present, the resonance rotational speed Rs of the motor 50 is calculated by dividing the resonance frequency Fb0 of the power supply 20 by both the target electrical angle order Tgt_n and the pole number Pp of the motor 50. It is preferable to include the resonant rotational speed Rs2. In this case, the second resonance rotational speed Rs2 can be expressed by the following equation 6. Thereby, the resonance level of the resonance by the harmonic component (when the electrical angle order n is 2 or more) included in the drive current of the motor 50 is reduced. Note that the number of second resonance rotational speeds Rs2 corresponds to the number of target electrical angle orders Tgt_n, and may be one or more.

  The resonant frequency Fb0 of the power supply 20 and the pole pair number Pp of the motor 50 are as described above. In the example described above, the second resonant rotational speed Rs2 is obtained by dividing the resonant frequency Fb0 of the power supply 20 by both the target electrical angle order Tgt_n 5 and the pole logarithm Pp of the motor 50 4 (that is, the resonant frequency Calculate Fb0 divided by 20). Thereby, the resonance level of the resonance due to the harmonic component (fifth harmonic component) having the electrical angle order n of 5 is reduced.

  The rotation speed range Rr includes the resonance rotation speed Rs of the motor 50. The rotation speed range Rr preferably includes the first resonance rotation speed Rs1. In this case, it is preferable to set the rotation speed range Rr such that the intermediate rotation speed between the upper limit rotation speed and the lower limit rotation speed of the rotation speed range Rr becomes the first resonance rotation speed Rs1. Furthermore, when the target electrical angle order Tgt_n is present, it is preferable that the rotation speed range Rr includes both the first resonance rotation speed Rs1 and the second resonance rotation speed Rs2. Further, it is preferable that the rotation speed range Rr is set to be capable of suppressing at least one of heat generation and deterioration of the power supply 20 due to resonance, and noise (noise) of the motor 50 due to resonance. A smoothing capacitor 30 can be included in setting the rotation speed range Rr.

  The rotational speed range Rr can be set in advance by simulation, verification by a real machine, or the like. Further, as the rotation speed range Rr becomes wider, the drive range in which the carrier frequency Fc is set to the second carrier frequency Fc2 becomes wider, and the switching loss increases. Therefore, it is preferable to determine in advance the rotational speed range Rr by grasping the minimum necessary rotational speed range by simulation, verification by a real machine or the like.

  Note that regardless of the presence or absence of the target electrical angle order Tgt_n, the resonance rotational speed Rs of the motor 50 can be calculated in advance before the motor 50 is driven. In this case, the rotation speed range Rr can be set in advance based on the resonance rotation speed Rs of the motor 50. At least one of the resonant rotational speed Rs and the rotational speed range Rr of the motor 50 may be stored in the second storage device 60b2 shown in FIG.

  In addition, the resonance rotational speed Rs of the motor 50 can also be calculated sequentially while the motor 50 is driven. For example, as shown in FIG. 9, the carrier frequency control unit 76 acquires the resonance frequency Fb0 of the power supply 20 (step S21). Also, the carrier frequency control unit 76 acquires the target electrical angle order Tgt_n (step S22). Furthermore, the carrier frequency control unit 76 acquires the pole pair number Pp of the motor 50 (step S23). Then, the carrier frequency control unit 76 divides the resonance frequency Fb0 of the power supply 20 by both the target electrical angle order Tgt_n and the pole pair number Pp of the motor 50 to calculate the second resonance rotation number Rs2 of the motor 50 (step S24).

  The order of steps S21 to S23 is not limited. Moreover, about the parameter which is not changed among step S21-step S23, it can also be abbreviate | omitted. Furthermore, the control shown in FIG. 9 can be repeatedly executed each time a predetermined time (which may be the same as or different from the repetition time of the control shown in FIG. 8) elapses. The carrier frequency control unit 76 can also calculate the first resonance rotational speed Rs1 of the motor 50 in the same manner.

  FIG. 10 shows an example of the relationship between the resonance frequency Fb0 of the power supply 20 and the rotational speed of the motor 50. The horizontal axis indicates the frequency, and the vertical axis indicates the number of rotations (the number of rotations of the motor 50). A curved line L31 represents an example of the structural resonant frequency of the motor 50, and a straight line L32 represents an example of the resonant frequency Fb0 of the power supply 20. The control device 60 of the present embodiment includes the carrier frequency control unit 76, so that the resonance level can be reduced. Therefore, as shown to the same figure, even if the curve L31 and the straight line L32 overlap and resonance generate | occur | produces, the noise (noise) of the electric motor 50 is suppressed.

  FIG. 11A relates to the reference embodiment, and shows an example of the relationship between the carrier frequency before changing the carrier frequency and the rotational speed of the motor 50. FIG. 11B relates to the reference embodiment, and shows an example of the relationship between the carrier frequency after changing the carrier frequency and the rotational speed of the motor 50. The horizontal axis of FIG. 11A and 11B shows a frequency, and the vertical axis | shaft has shown rotation speed (rotation speed of the electric motor 50). A straight line L41 and a straight line L44 show an example of the carrier frequency, and a curve L42 and a curve L45 show an example of a sideband wave (harmonic component) of the carrier frequency. The curve L43 and the curve L46 are identical curves and show an example of the structural resonance frequency of the motor 50.

  11A and 11B are diagrams according to a reference embodiment for describing the invention described in Patent Document 2. FIG. As shown in FIG. 11A, the curve L42 and the curve L43 overlap (the frequency of the sideband wave (harmonic component) matches the structural resonance frequency of the motor 50) resonates and the noise of the motor 50 (noise) Can occur. Therefore, as shown in FIG. 11B, in the reference embodiment, when the driving rotational speed of the motor 50 approaches the frequency of the sideband wave (harmonic component), the carrier frequency is changed, and the curve L45 and the curve L46 do not overlap ( The frequency of the sideband wave (harmonic component) does not match the structural resonance frequency of the motor 50). Thus, the present embodiment and the reference form have entirely different technical ideas.

(Pulse width modulation signal generation unit 77)
The pulse width modulation signal generation unit 77 uses the carrier frequency Fc output from the carrier frequency control unit 76 to open and close signals of a plurality of switching elements (positive electrode side switching element 4 xp and negative electrode side switching element 4 xn) of the power converter 40. Generate The carrier frequency Fc (in this embodiment, the first carrier frequency Fc1 or the second carrier frequency Fc2) output from the carrier frequency control unit 76 is set as an output carrier frequency Fc_out. As shown in FIG. 4, in the pulse width modulation signal generation unit 77, three-phase voltage command values (U-phase voltage command value Vu_ref, V-phase voltage command value Vv_ref, and the like) calculated by the two-phase / three-phase conversion unit 75. W-phase voltage command value Vw_ref), DC voltage detection value Vdc_fb detected by DC voltage detector 61a, and carrier frequency Fc (output carrier frequency Fc_out) output from carrier frequency control unit 76 are input.

  The pulse width modulation signal generation unit 77 uses the output carrier frequency Fc_out as the frequency of the carrier wave (triangular wave) in the pulse width modulation (PWM) control. Further, the pulse width modulation signal generation unit 77 divides the three-phase voltage command value by the DC voltage detection value Vdc_fb to calculate the modulation factor. The pulse width modulation signal generation unit 77 generates a pulse signal (switching signal) by pulse width modulation (PWM) control based on the calculated modulation factor and the carrier wave (triangular wave) of the output carrier frequency Fc_out. Specifically, when the voltage command value is larger than the carrier wave, a signal (for example, a high level signal (a state exceeding a predetermined voltage value)) in which the switching element is set to the closed state is generated. In addition, when the voltage command value is smaller than the carrier wave, a signal (for example, a low level (state less than or equal to a predetermined voltage value)) in which the switching element is set to an open state is generated. The generated pulse signal (opening / closing signal) is applied to the control terminal 4g of each switching element of the power converter 40 via the drive circuit 61b shown in FIG.

<One example of effect>
According to the motor control device 10 according to the aspect 1, the control device 60 includes the carrier frequency control unit 76. Carrier frequency control unit 76 has a resonant rotational speed Rs which is the rotational speed of motor 50 when the electrical resonance frequency Fb0 of power supply 20 and the frequency Fm0 of the drive current of motor 50 coincide and electrical resonance occurs. When the drive rotational speed of the electric motor 50 is included in a predetermined rotational speed range Rr including the above, the carrier frequency Fc of pulse width modulation (PWM) control is compared to when the drive rotational speed is not included in the rotational speed range Rr. Set high.
Thereby, the motor control device 10 according to the aspect 1 reduces the ripple of the drive current of the motor 50 to reduce the resonance level before the occurrence of the electrical resonance caused by the power supply 20 that outputs the DC power. Can.

According to the motor control device 10 according to aspect 2, in the motor control device 10 according to aspect 1, the carrier frequency control unit 76 causes the ripple of the drive current when the drive rotational speed of the motor 50 is included in the rotational speed range Rr. The carrier frequency Fc is set so as to satisfy both of the necessary degree of reduction of and the allowable switching loss of the plurality of switching elements (positive side switching element 4xp and negative side switching element 4xn).
Thereby, the motor control device 10 according to the aspect 2 can reduce the resonance level while suppressing an increase in switching loss in the plurality of switching elements (the positive electrode side switching element 4xp and the negative electrode side switching element 4xn).

According to motor control device 10 according to mode 3, in motor control device 10 according to mode 1 or mode 2, the resonance frequency Rs of motor 50 is obtained by dividing resonance frequency Fb0 of power supply 20 by pole number Pp of motor 50. And the first resonance rotational speed Rs1 calculated.
Thereby, the motor control device 10 according to the aspect 3 can reduce the resonance level of the resonance by the fundamental wave component included in the drive current of the motor 50.

According to the motor control device 10 according to aspect 4, in the motor control device 10 according to aspect 3, the resonance rotational speed Rs of the motor 50 corresponds to the resonance frequency Fb0 of the power source 20, the target electrical angle order Tgt_n, and the pole logarithm of the motor 50. The second resonance rotational speed Rs2 calculated by dividing by both Pp is included. The target electrical angle order Tgt_n refers to the electrical angle order that is a factor of resonance among the electrical angle orders n of the harmonic components included in the drive current of the motor 50.
Thereby, the motor control device 10 according to the aspect 4 can reduce the resonance level of the resonance due to the harmonic component included in the drive current of the motor 50.

<Others>
The embodiments are not limited to the embodiments described above and shown in the drawings, and can be appropriately modified without departing from the scope of the invention. For example, control of the motor 50 is not limited to vector control. The control device 60 can perform known drive control (for example, rectangular wave drive and the like). In addition, the motor control device 10 is preferably used in a power conversion system including a driving motor of a vehicle such as, for example, a hybrid car and an electric car.

<Additional clause>
The present invention can also be grasped as a control method of the motor control device 10. The carrier frequency control step corresponds to the control performed by the carrier frequency control unit 76 described above. Also in the control method of the motor control device 10, it is possible to obtain the same effects as the effects described above for the motor control device 10.

(Appendix 1)
A power supply that outputs DC power,
A power converter including a plurality of switching elements that convert the DC power output from the power source into AC power and output the AC power to the motor;
A control device which performs switching control of the plurality of switching elements of the power converter by pulse width modulation control;
A control method of a motor control device comprising the
The motor has a predetermined rotational speed range including the resonant rotational speed which is the rotational speed of the motor when the electrical resonance frequency of the power source matches the frequency of the drive current of the motor and the electrical resonance occurs. An electric motor control device comprising a carrier frequency control step of setting the carrier frequency of the pulse width modulation control higher when the driving rotational speed is included, compared to when the driving rotational speed is not included in the rotational speed range. Control method.

(Appendix 2)
In the carrier frequency control step, when the drive rotational speed of the motor is included in the rotational speed range, both of the required reduction of the ripple of the drive current and the allowable switching loss of the plurality of switching elements The control method of the motor control device according to item 1, wherein the carrier frequency is set so as to satisfy the condition.

10: Motor controller,
20: Power supply,
40: Power converter,
4 xp: positive electrode side switching element, 4 x n: negative electrode side switching element,
50: Electric motor,
60: controller, 76: carrier frequency controller,
Fb0: resonant frequency, Fm0: frequency of driving current, Fc: carrier frequency,
Pp: pole-pair number, Tgt_n: target electrical angle order,
Rs: resonant rotational speed, Rs1: first resonant rotational speed, Rs2: second resonant rotational speed.
However, x is any one of u, v and w.

Claims (4)

A power supply that outputs DC power,
A power converter including a plurality of switching elements that convert the DC power output from the power source into AC power and output the AC power to the motor;
A control device which performs switching control of the plurality of switching elements of the power converter by pulse width modulation control;
Equipped with
The control device is configured to perform predetermined rotation including a resonant rotational speed which is a rotational speed of the motor when an electrical resonance frequency of the power source matches a frequency of a drive current of the motor and an electrical resonance occurs. A carrier frequency control unit which sets the carrier frequency of the pulse width modulation control higher than when the drive rotational speed is not included in the rotational speed range when the drive rotational speed of the motor is included in several ranges. The motor control apparatus provided.   The carrier frequency control unit determines both the required reduction degree of the ripple of the drive current and the allowable switching loss of the plurality of switching elements when the drive rotational speed of the motor is included in the rotational speed range. The motor control device according to claim 1, wherein the carrier frequency is set so as to satisfy the condition.   The motor control device according to claim 1 or 2, wherein the resonant rotational speed of the motor includes a first resonant rotational speed which is calculated by dividing the resonant frequency of the power supply by the number of pole pairs of the motor. Among the electrical angle orders of the harmonic components included in the drive current of the motor, when the electrical angle order causing the resonance is the target electrical angle order,
The said resonant rotation speed of the said motor contains the 2nd resonant rotation speed calculated by dividing the said resonant frequency of the said power supply by both the said object electrical angle order and the pole pair number of the said motor. Motor controller.

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