JP2018201319A - Drive system of double three-phase wiring permanent magnet type synchronous motor - Google Patents

Abstract

To provide a drive system of double three-phase wiring permanent magnet type synchronous motor including a rotor with a permanent magnet and a stator with two three-phase wiring, the drive system being capable of achieving appropriate control of a current of each of the two wiring even strong mutual induction exists between the two wiring.SOLUTION: A current controller 3 constituting a motor-driven system along with a motor 1 and a power converter 2 is constituted using at least either one of a high-speed mode current control unit 33 and a low-speed mode current canceler 34.SELECTED DRAWING: Figure 6

Description

The present invention includes a rotor having a permanent magnet and two three-phase windings (a first three-phase winding and a second three-phase winding, or a self three-phase winding and another three-phase winding). A double three-phase winding permanent magnet synchronous motor comprising at least a double three-phase winding permanent magnet synchronous motor and a power converter capable of simultaneously supplying a three-phase current to two three-phase windings The drive system. In the present invention, the two three-phase windings may be referred to as the self three-phase winding and the other three-phase winding, or the distinction between the self and others is removed, and the first three-way winding and the second Sometimes called three-phase winding. When the first three-phase winding is a self three-phase winding, the second three-phase winding is another three-phase winding. Similarly, when the second three-phase winding is a self three-phase winding, the first three-phase winding is another three-phase winding. In the following description, unless otherwise specified, the invention will be described with the first three-phase winding as its own three-phase winding and the second three-phase winding as the other three-phase winding unless otherwise specified. . Thereby, the generality of the invention is not lost.

In the following description, for the sake of simplicity, “winding” is used synonymously with “three-phase winding”. The above double three-winding permanent magnet synchronous motor is abbreviated as a double synchronous motor for simplicity. Furthermore, for the same reason, the drive system is abbreviated as a double synchronous motor drive system. Applications of the dual synchronous motor drive system of the invention include main drive of battery electric vehicle, fuel cell electric vehicle, hybrid electric vehicle, etc., use of home appliances that require efficient drive over a wide range, or fault tolerance, functional safety Is a required application.

In the present invention, a portion provided with a three-phase winding in a double synchronous motor is referred to as a “stator”. The “stator” in the present invention is synonymous with “armature”. There are Y-type and Δ-type in the three-phase winding applied to the stator. As is well known to those skilled in the art, when evaluated from a three-phase terminal, the characteristics of the Y-shaped winding and the characteristics of the Δ-shaped winding are equivalently converted to each other. In order to ensure the simplicity of the explanation, the technical explanation in this specification is made assuming a Y-shaped connection. This does not lose the generality of the present invention, as is evident from the existence of equivalent transformations.

In the present invention, a two-dimensional plane is taken as a polar coordinate, and three terms of angle, spatial position, and spatial phase are used synonymously. These units are “radians” or “degrees”. In the present invention, the positive direction of the angle, the spatial position, and the spatial phase may be defined as left-handed (counterclockwise) or right-handed (clockwise). However, in this specification, in order to maintain the simplicity of the description, the positive direction of the angle, the spatial position, and the spatial phase is defined as counterclockwise (counterclockwise), and the present invention will be described. Thus, the generality of the present invention is not lost.

In the present invention, a device that supplies AC power to the double synchronous motor is referred to as a power conversion device. Inverters, matrix converters, and the like have been put into practical use as power converters that are main equipment of power converters. Single / six phase, two / three phase, six / single phase power converters and the like may constitute the power converter of the present invention.

In the present invention, in principle, parameters, physical quantities and the like related to the first winding and the second winding are indicated by the reference numerals 1 and 2 respectively. In principle, in a dq synchronous coordinate system (detailed description will be given later with reference to FIG. 5) consisting of two orthogonal axes of d axis and q axis, the symbols d and q are used for parameters, physical quantities and the like associated with each axis. To clearly indicate the relationship with each axis. Further, as a general rule, a signal corresponding to the command value of the response signal is indicated by adding the initial symbol * to the same symbol as that of the response signal to clearly indicate the command value of the corresponding signal.

When the self-inductance of the self-winding is L1, the self-inductance of the other winding is L2, and the mutual inductance between the two windings is M, the leakage coefficient σ is defined according to the following equation (1):
The leakage inductance is defined according to the following equation (2).
The “equivalent value of leakage inductance” in the present invention means the approximate values of true leakage inductances σL1 and σL2. The approximate value in this case also includes a value that is several times the true value (see equation (22) below).

The “equivalent value of mutual inductance” in the present invention means a general term for a wide range of approximate values of the true value of the mutual inductance M. This approximate value includes a value expressed by the following formula (see formula (25) below).
The middle side of the equation (3) indicates the difference between the self inductance and the leakage inductance. Further, the approximation of the right side of the equation is established without any problem when the leakage coefficient σ is small. Naturally, the mutual inductance equivalent value is non-zero.

In the present invention, “current equivalent value” is used as a general term for the true value, command value, estimated value, approximate value, etc. of the stator current.

Similarly, in the present invention, the “speed equivalent value” of the rotor is a true value of the rotor speed, a command value, an estimated value, an approximate value, and a two-axis orthogonal rotating coordinate system that rotates at the same average speed as the rotor. It is used as a general term for the true value and approximate value of the speed. As is well known to those skilled in the art, there is an electrical speed and a mechanical speed in the rotor speed, but there is a one-to-one strict relationship between the two speeds, electric speed to mechanical speed, and mechanical speed to electric speed. Unique conversion to is possible. In the present invention, in consideration of the well-known among those skilled in the art, unless the clarity of explanation is lost, the rotor speed is used as meaning electric speed. In the present invention, “speed multiplication equivalent processing” is used as a generic term for processing involving multiplication of speed equivalent values.

In the present invention, “differential equivalent processing” is used as a general term for processing centered on differential processing, such as processing in which non-linear processing such as limiter processing is added to pure differential processing, approximate differential processing, and linear differential processing.

For example, Patent Documents 1 and 2 and Non-Patent Documents 1 to 5 are prior art related to a double synchronous motor that is driven by the double synchronous motor drive system of the present invention. From the viewpoint of the arrangement of the double three-phase windings of the stator, the already reported double synchronous motor is a three-phase simple synchronous motor (Non-Patent Document 1), a six-phase synchronous motor (Patent Document 2, Non-Patent Documents 2-3). ), Roughly divided into three types of three-phase reverse synchronous motors (Non-Patent Documents 4 to 5).

Referring to Non-Patent Document 1, an outline of a conventional double synchronous motor (three-phase simple synchronous motor) is shown in FIG. 1 by taking as an example a case where the number N of pole pairs is Np = 1. 1 is a double synchronous motor (including a rotor and a stator), 11 is a rotor of the double synchronous motor, 121 is a first winding of the stator of the double synchronous motor, and 122 is a double synchronous motor. Each of the second windings of the stator is shown. In the figure, in order to clarify the distinction between the first winding and the second winding of the stator, the first winding is indicated by a solid line and the second winding is indicated by a broken line. In addition, since the second winding overlaps with the first winding in terms of the winding arrangement, the second winding is intentionally shifted to the right to avoid drawing overlap.

The double synchronous motor (three-phase simple synchronous motor) according to this arrangement has the following characteristics. (A) For both the first winding and the second winding, the u-phase winding, the v-phase winding, and the w-phase winding are sequentially 2π / 3 [rad] spatially in a space based on the number of pole pairs. Arranged at the position of phase advance. (B) In principle, the first winding and the second winding are arranged without a phase difference in space. (C) In principle, the number of pole pairs can be any integer. That is, an odd or even number of pole pairs can be employed. (D) Whether the first winding and the second winding are energized simultaneously, or only one of the windings is energized, the number of phases remains unchanged. (E) In principle, when the first winding and the second winding are energized simultaneously, the currents of the first winding and the second winding need to be synchronized without phase difference.

As a second example of a stator winding arrangement of a double synchronous motor (an example of a six-phase synchronous motor), a case where the number of pole pairs Np is set to Np = 1 with reference to Patent Document 2 and Non-Patent Documents 2 to 3 is an example. FIG. 2 schematically shows the rotor together (drawing of winding resistance is omitted). The meanings of the drawn line numbers 1, 11, 121, and 122 are the same as those in FIG. However, it differs from the example of FIG. 1 in that the arrangement of the second winding is spatially shifted by θ12 = π / 6 [rad] with respect to the first winding in a space based on the number of pole pairs. ing.

The double synchronous motor (six-phase synchronous motor) according to this arrangement has the following characteristics compared to the three-phase simple synchronous motor (FIG. 1). (A) Same as item (a) in Example 1. (B) In principle, the first winding and the second winding are arranged so as to have a spatial phase difference of π / 6 [rad] in a space based on the number of pole pairs. (C) Same as item (c) in Example 1. (D) When energizing the first winding and the second winding simultaneously, it operates as a six-phase motor, and when energizing only one of the windings, it operates as a three-phase motor. (E) In principle, when the first winding and the second winding are energized simultaneously, the currents in the first winding and the second winding have a phase difference corresponding to the spatial phase difference. Synchronization is required.

As a third example of a stator winding arrangement of a double synchronous motor (an example of a three-phase reverse synchronous motor), an example in which the number of pole pairs Np is Np = 2 with reference to Non-Patent Documents 4 to 5 is shown in FIG. Fig. 6 schematically shows together with the rotor (drawing of winding resistance is omitted). The meanings of the drawn line numbers 1, 11, 121, and 122 are the same as those in FIG.

Compared with Example 1 and Example 2, the double synchronous motor (three-phase reverse synchronous motor) by this winding arrangement has the following characteristics. (A) For both the first winding and the second winding, the u-phase winding, the v-phase winding, and the w-phase winding are sequentially 2π / 3 [rad] spatially in a space based on the number of two pole pairs. Arranged at the position of phase lag. (B) In principle, the first winding and the second winding are arranged so as to have a phase difference of ± π [rad] in a space based on the number of two pole pairs. (C) Only an even number of pole pairs can be taken. That is, an odd number of pole pairs cannot be used. (D) Same as item (d) in Example 1. (E) Same as (e) in Example 1.

In the double synchronous motor illustrated in FIGS. 1 to 3, the first winding and the second winding need not necessarily be configured to have the same characteristics. Both windings are configured to have the same characteristics as shown in Patent Documents 1 and 2 and Non-Patent Documents 1 to 4, and are different from each other as shown in Non-Patent Document 5. It can also be configured to have characteristics.

In the double synchronous motor illustrated in FIGS. 1 to 3, the neutral point of the first winding and the neutral point of the second winding are not connected. In the double synchronous motor targeted by the present invention, generally, the neutral point of the first winding and the neutral point of the second winding can be either non-connected or connected.

Subsequently, conventional technologies related to a double synchronous motor drive system, that is, a drive system for a double synchronous motor will be introduced. The present invention relates to a current control device which is one of main components of a double synchronous motor drive system. Based on this point, we introduce the conventional technology related to the current control device for the double synchronous motor drive system. FIG. 9 is a citation of a current control device using an interwinding non-interference device for a double synchronous motor drive system proposed in Non-Patent Document 1 (Patent Document 1 also includes a substantial effect by the same inventor). The same interwinding decoupler is shown). Note that Non-Patent Document 1 targets the three-phase simple synchronous motor of FIG. 1 as a double synchronous motor, and the double synchronous motor at this time is a nonsalient pole.

In FIG. 9, the current control device includes first and second “current control systems” disposed at the left end and a “non-interacting unit” disposed at the center. “Current control system” in FIG. 9 means “feedback current controller” in the correct scientific term (technical term). The configuration of the non-interference part in FIG. 9 is described by the following mathematical formula.
In the equation (4), the output signal of the feedback current controller for the first winding (described as “voltage command-1” in FIG. 9) is expressed by v11 *, and the feedback current controller for the second winding The output signal (described as “voltage command-2” in FIG. 9) is expressed by v22 *. These two output signals (two voltage command values) v11 * and v22 * from the feedback current controller are input signals of the non-interacting unit expressed by the equation (4). The coefficient (M / L) in the equation (4) is a constant determined from the inductance of the non-saliency double synchronous motor. The output signals v12 * and v21 * of the non-interacting unit are generated according to the equation (4).

The output signal v12 * is a non-interfering signal to the first three-phase winding side, and the output signal v21 * is a non-interfering signal to the second three-phase winding side, and a feedback current controller for each winding. Are respectively added to the output signals (voltage command values) to generate final voltage command values v1 * and v2 *. The generation of the final voltage command value is described by the following equation.
Here, v1 * and v2 * mean the final voltage command value for the first winding and the final voltage command value for the second winding, respectively. In FIG. 9, which is a simplified diagram, a power conversion device such as an inverter is omitted and not shown.

As is clear from the equations (4) and (5), the former interwinding non-interferor weights the output signal (voltage command value) of the feedback current controller for each winding with a certain linear relationship. Thus, the final voltage command value for each winding is synthesized. That is, the input signal of the conventional interwinding non-interfering device is a voltage command value, and the conventional interwinding non-interfering device should also be called a “voltage type”. In addition, the current control device including the voltage-type interwinding non-interfering device is constructed for a nonsalient double synchronous motor without performing a strict mathematical analysis. When the mutual induction between the first winding and the second winding (in the present invention, mutual induction and magnetic coupling are used synonymously) is strong, in the case of the current control device as in Patent Document 1, current control is performed. A system destabilization phenomenon or a vibration phenomenon according to this phenomenon easily occurs (for example, see Non-Patent Document 2), and the desired current control performance cannot be exhibited at all.

Akira Satake and Shigeki Mizuno: “Control Device for Multi-winding Motor”, Japanese Patent Laid-Open No. 2001-341135 (2001-11-6) Keiichiro Ban, Kazuyoshi Obayashi: “Electric drive for automobiles”, Japanese Patent Laid-Open No. 2000-41392 (1998-7-23)

Akira Satake, Satoshi Kato, Akira Imanaka: "Modeling and non-interference control method of multi-winding permanent magnet motor", Proceedings of the Institute of Electrical Engineers of Japan, I, pp. 199-202 (2005) Takafumi Imai, Fumiaki Osawa, Satoshi Yamada, Yukio Inaguma: “Possibility of standardization of EV / HEV electric drive system (current ripple suppression of multiphase motor)”, Proceedings of the IEEJ National Convention, 4, pp. 361-362 (2016) Junya Mori and Jun Furukawa: “Pulse pattern to reduce torque ripple in a double inverter with double three-phase PMSM drive 1 shunt current detection”, Proceedings of the Institute of Electrical Engineers of Japan, III, pp. 159-164 (2016) Shinnaka Shinji: “Three-phase permanent magnet synchronous motor (double winding arrangement, dynamic mathematical model, vector simulator) with inverted double three-phase winding with 180 degree spatial phase difference”, 2016 IEEJ Industry Application Division Conference Proceedings, III, pp. 285-290 (2016) Shinji Shinnaka: “Three-phase permanent magnet synchronous motor (double winding arrangement, dynamic mathematical model, vector simulator) with reverse double three-phase winding with 180 degree spatial phase difference”, IEEJ Transactions D, Vol. 137, no. 2, pp. 75-86 (2017)

The present invention has been made under the above background. It is also applicable when the double synchronous motor has a strong mutual induction between the first winding and the second winding, and hence high quality with high stability of the first winding and the second winding. It is another object of the present invention to provide a new current control device for a double synchronous motor drive system that enables current control.

In order to achieve the above object, a first aspect of the present invention is a permanent magnet comprising a rotor having a permanent magnet and a stator having two three-phase windings (a self-three-phase winding and another three-phase winding). A magnet synchronous motor, a power converter that can supply current to two three-phase windings simultaneously, and a current control device that controls the current flowing through the two three-phase windings via the power converter A permanent magnet synchronous motor drive system comprising a controller coefficient determined by using at least an equivalent value of leakage inductance caused by mutual induction of two three-phase windings. A high-speed mode current controller for each three-phase winding that generates a voltage command value for controlling the high-speed mode current generated by the mutual induction, and the mutual inductance due to the mutual induction of the two three-phase windings Value and current phase equivalence of the three-phase winding at least Two different from the low-speed mode current canceller for each three-phase winding that generates a voltage command value for its own three-phase winding to cancel out the low-speed mode current generated by mutual induction of the two three-phase windings The current control device is configured using at least one of the above devices (high-speed mode current controller and low-speed mode current canceller).

The invention according to claim 2 is the permanent magnet synchronous motor drive system according to claim 1, wherein the phase of the N pole of the rotor permanent magnet is the phase of the d axis, and π / 2 [rad with respect to the d axis. ], When a two-axis orthogonal coordinate system having a q axis in phase advance is a dq-synchronous coordinate system and a coordinate system conforming to the dq-synchronous coordinate system is a γδ quasi-synchronous coordinate system, The current deviation of each three-phase winding, which is the difference between the current command value of each three-phase winding on the system and the current response value of each three-phase winding, is processed in a feedback control manner on the dq synchronous coordinate system or γδ The high-speed mode current controller for each three-phase winding is configured to generate a voltage command value for each three-phase winding on the quasi-synchronous coordinate system.

A third aspect of the present invention is the permanent magnet synchronous motor drive system according to the first aspect, wherein the phase of the N pole of the rotor permanent magnet is the d-axis phase, and π / 2 [rad with respect to the d-axis. ], When a two-axis orthogonal coordinate system having a q axis in the phase advance is a dq-synchronous coordinate system and a coordinate system according to the dq-synchronous coordinate system is a γδ quasi-synchronous coordinate system, the current equivalent value of the free-three-phase windings on synchronous coordinate system i1 and ^~, the current equivalent value of the other three-phase windings i2 and ^~, the mutual induction of the three-phase windings of the two (the own and other) the equivalent value of the mutual inductance due to ^{M1d ~, M1q ~, M2d ~} , when the M2Q ^~, to produce an intermediate signal Fai1M ^~ for self three-phase winding according to the following equation,
The intermediate signal for the generated three-phase winding is subjected to at least one of a differential equivalent process and a speed multiplication equivalent process to generate a voltage command value for the own three-phase winding. A low-speed mode current canceller for three-phase winding is constructed.

The effect of the present invention will be described. The present invention is based on the fact that “the currents in the high speed mode and the low speed mode appear due to mutual induction of the self-winding (first winding) and the other winding (second winding)”. Is controlled by the high-speed mode current controller, and the low-speed mode current current is canceled and suppressed by the low-speed mode current canceller, and flows through the two windings of the own winding and the other winding. It is intended to control the current stably. The fact that the high-speed mode and the low-speed mode are generated by mutual induction, which is one of the keys for implementing the present invention, will be clarified.

Consider FIG. 4 (a). This figure schematically shows a state in which two single-phase LR circuits are magnetically coupled (mutual induction). At this time, the primary side and secondary side inductances and resistances are L ₁ , L ₂ , R ₁ , and R ₂ , respectively, and the mutual inductance is M. A characteristic equation governing the characteristics of the circuit is given by the following equation.
The definition of the leakage coefficient σ in the above equation is as in equation (1).

In the case of σ = 1 where there is no mutual induction between the primary side and the secondary side, the characteristic roots s1 and s2 that determine the circuit mode are the following two real roots.
On the other hand, when there is a strong mutual induction between the primary side and the secondary side and the following expression (9a) holds, the two characteristic roots s1 and s2 are approximately 2 in the expression (9b). Become a real root.
When R1 / L1> = R2 / L2, the equations (8) and (9) indicate that the two characteristic roots correspond to the high-speed mode (hereinafter, high-speed root) according to the mutual induction improvement σ → 0. This means changing to s1 and a root corresponding to the low speed mode (hereinafter referred to as a low speed root) s2. That is,
Although the change of the slow root s2 corresponding to the improvement of mutual induction σ → 0 is slow and minute, the change of the fast root s1 shows a divergent increase of the inverse ratio 1 / σ.

The fast root s1 and the slow root s2 at the right end of the equation (10) can be rewritten as the following equation (11).
The middle side of the expression (11) is an expression viewed from the primary side terminal, and the right side is an expression viewed from the secondary side terminal. (ΣL1) and (σL2) in the denominator of the first equation of the equation (11) are developed as the equation (2), which means the leakage inductance viewed from the primary side terminal and the secondary side terminal, respectively. Yes.

In the configuration of the feedback current control system that controls a circuit having strong mutual induction, only the two low-speed roots of the equation (8) are assumed, and the existence of the high-speed root s1 of the equations (9b) and 1 is ignored. In this case, the current control system becomes unstable (generation of high-frequency current ripple, current divergence, etc.).

In order to simplify the explanation of the effects of the present invention, the coordinate system will be explained before the full explanation. Consider FIG. FIG. 5 shows an αβ fixed coordinate system, a dq synchronous coordinate system, and a γδ general coordinate system. The αβ fixed coordinate system is a coordinate system corresponding to the stator, and in general, the α axis is taken at the center of the u-phase winding of the stator first winding (the u-phase winding of the stator second winding). There is no essential difference for the center). The dq synchronous coordinate system is one of the rotary coordinate systems, and in particular, the coordinate system in which the d axis is synchronized with the N pole of the rotor. That is, in the dq synchronous coordinate system, the phase of the d axis is the same as the phase of the rotor magnetic flux. The speed of the dq synchronous coordinate system is the same as the rotor speed ωn instantaneously. The γδ general coordinate system is a general coordinate system having an arbitrary coordinate system speed ωγ. As a special case, the γδ general coordinate system includes an αβ fixed coordinate system and a dq synchronous coordinate system. Further, a γδ quasi-synchronous coordinate system that can have a small phase difference with respect to the dq-synchronous coordinate system is also included as a special case. Regarding the phase of the coordinate system, the phase of the d-axis viewed from the α-axis is θα, the phase of the γ-axis viewed from the α-axis is θαγ, and the phase of the d-axis viewed from the γ-axis is θγ.

The three types of double synchronous motors shown in FIGS. 1 to 3 clearly have different winding arrangements. On the dq synchronous coordinate system, these mathematical models (circuit equations) are commonly expressed by the following equations: (See Non-Patent Document 5).

The meanings of the foot marks 1, 2 and d, q in the mathematical model are as described above. The symbol s means the differential operator d / dt. The physical quantity used in the mathematical model will be described using the first winding as an example. The 2 × 1 vectors v1 and i1 defined on the dq synchronous coordinate system mean the voltage and current of the stator, respectively. I is a 2 × 2 unit matrix. R1 is the resistance of the stator winding, and Φ1 is the rotor magnetic flux intensity (electromotive force constant) viewed from the stator winding. L1d and L1q are a d-axis (self) inductance and a q-axis (self) inductance. Md and Mq are a d-axis mutual inductance and a q-axis mutual inductance, respectively.

In the present invention, the d-axis (self) inductance L1d and the q-axis (self) inductance L1q are used to define the in-phase (self) inductance L1i and the mirror phase (self) inductance L1m as shown in the following equation (13). And use it.
As the equation (13) clearly shows, the in-phase inductance L1i means the average value of the corresponding d-axis and q-axis inductances.

In the double synchronous motor targeted by the present invention, the motor parameters (winding resistance R1, inductance L1d, L1q, etc.) resulting from the two three-phase windings may be the same or different. . The present invention does not depend on differences in motor parameters due to windings.

The relationship of equation (12) is drawn as shown in FIG. 4B as a virtual vector circuit on the dq synchronous coordinate system rotating at ωn. All physical quantities such as voltage and current in the figure are 2 × 1 vectors defined on the dq synchronous coordinate system. In addition, the 2 × 2 inductance matrix in the figure is defined as follows.
In particular, at zero speed ωn = 0, the inter-axis interference between the d-axis and the q-axis disappears, and the virtual vector circuit uses the “single-phase LR circuit (mutual induction circuit) in FIG. To a two-phase circuit arranged independently and in parallel for each axis. As can be understood from the identity at zero speed, the double synchronous motor has a high-speed mode that is approximately proportional to the reciprocal of the d-axis and q-axis leakage coefficients σd and σq. In addition, it has a low speed mode. The leakage coefficients σd and σq of each axis are defined according to the following equation similar to the equation (1).

As understood from the zero speed equivalence of the single-phase LR circuit (mutual induction circuit) and the virtual vector circuit of the double synchronous motor in FIG. 4, the first winding and the second winding have strong mutual induction. In the configuration of the feedback current control system for the double synchronous motor, when assuming only the low-speed mode current due to the low-speed root, the feedback current control system is easily destabilized (high-frequency current ripple generation, current divergence). Etc.).

According to the first aspect of the present invention, there is provided a current (high speed mode current, low speed) in which the double synchronous motor having a strong mutual induction between the first winding and the second winding operates in two different modes, a high speed mode and a low speed mode. It is intended to control them separately with deep recognition of having a mode current. In particular, according to the first aspect of the present invention, the controller coefficient determined by using at least the equivalent value of the leakage inductance caused by the mutual induction of the two three-phase windings is provided, and the two three-phase windings The high-speed mode current is controlled by using a high-speed mode current controller for each three-phase winding that generates a voltage command value for controlling the high-speed mode current generated by the mutual induction. As already revealed analytically, the high-speed mode current is dominated by the leakage inductance. Therefore, according to the first aspect of the invention, the effect that the high-speed mode current can be effectively controlled can be obtained. As a result, instability caused by the high-speed mode current can be prevented. In other words, an effect that the stability of the feedback current control system can be secured can be obtained.

Successful control of the fast mode current does not mean successful control of the slow mode current. Even when the high-speed mode current is appropriately controlled, the low-speed mode current causes a low-frequency pulsation in the winding current. As can be understood from the virtual vector circuit of FIG. 4B, the low-speed mode current is obtained when the self-three-phase winding is the first winding and the mutual inductance matrix M (see equation (14)) and the self-three-phase. It can be seen that the winding current (i.e. the first winding current) i1 is contained in the form of these products. Similarly, when the own three-phase winding is the second winding, the low-speed mode current is the mutual inductance matrix M (see the equation (14)) and the current of the own three-phase winding (that is, the second winding current). It can be seen that i2 is contained in the form of these products. According to the first aspect of the present invention, two three-phase windings are obtained by using at least the equivalent value of the mutual inductance caused by the mutual induction of the two three-phase windings and the current phase equivalent value of the own three-phase winding. Thus, a low-speed mode current canceller for each of the three-phase windings that generates a voltage command value for canceling out the low-speed mode current generated by the mutual induction is constructed. That is, according to the invention of claim 1, the low-speed mode current can be canceled through the voltage command value generated by the low-speed mode current canceller. As a result, the effect that the low frequency pulsation of the winding current caused by the low speed mode current can be suppressed is obtained. In particular, when used in combination with a high-speed mode current controller that contributes to ensuring stability, an effect of providing a high-quality current control effect with low-frequency pulsation suppressed can be obtained. The invention of claim 1 does not provide any restrictions on saliency with respect to the double synchronous motor, and as a matter of course, can be applied to a double synchronous motor having strong saliency.

Note that the effects of the high-speed mode current controller and the low-speed mode current canceller described above focus on the leakage inductance directly related to the high-speed mode current and the mutual inductance directly related to the low-speed mode current. This can also be confirmed by redeploying the circuit equation as follows.
For example, regarding the voltage v1 of the first winding indicated by the equation (16), the left side of the equation (16a) indicates the voltage v1 itself, and the right side of the equation indicates the voltage corresponding to the mode. In particular, v1 ^~ the high-speed mode current
The details of are shown in equations (16b) and (16c), respectively. As reconfirmed from the equations (16b) and (16c), the high-speed mode current is d-axis and q-axis leakage inductances (σd · L1d) and (σq · L1q), and the low-speed mode current is d-axis, q It is governed by the mutual inductance equivalent value ((1-σd) L1d, Md) and ((1-σq) L1q, Mq) of the shaft. The same applies to the voltage v2 of the second winding indicated by the equation (17).

Next, the effect of the invention of claim 2 will be described. The high-speed mode current controller according to the first aspect of the present invention can be constructed on any coordinate system of the dq synchronous coordinate system, the γδ quasi-synchronous coordinate system, and the αβ fixed coordinate system. However, the difficulty of construction is not the same. Construction on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system is the simplest and is expected to be effective. According to the invention of claim 2, each three-phase winding is a difference between the current command value of each three-phase winding on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system and the current response value of each three-phase winding. High-speed mode current control for each three-phase winding so as to generate a voltage command value for each three-phase winding on the dq synchronous coordinate system or γδ quasi-synchronous coordinate system Can be configured. As a result, according to the invention of claim 2, an effect is obtained that a high-speed mode current controller can be easily constructed. As a result, according to the invention of claim 2, the effect that the effect of the invention of claim 1 can be enhanced is also obtained.

Next, the effect of the invention of claim 3 will be described. When the first winding is the self-winding and the second winding is the other winding, one of the ideal configurations of the low-speed mode current canceller for the self-winding is particularly on the dq synchronous coordinate system. One of the configurations is described by the following equation from equation (16c).
The meaning of the prefix * in the equation (18) is as described above. According to the low-speed mode current canceller configured in accordance with the invention of claim 3, when the first winding is the self-winding and the second winding is the other winding, the equation (18b) is included ( 6) generating an intermediate signal Fai1M ^~ for self three-phase winding in accordance with formula for the generation of intermediate signals for its own three-phase windings, differential equivalent process, less speed multiplication equivalent process
The invention includes the configuration of the formula (18). According to the invention of claim 3, the low-speed mode current canceller can adopt one of ideal structures. As a result, the effect that the low-speed mode current canceller can exhibit desirable performance can be obtained.

When the second winding is the self-winding and the first winding is the other winding, one of the ideal configurations of the low-speed mode current canceller for the self-winding is particularly the dq synchronous coordinate system. The above configuration is described by the following equation from equation (17c).

  "Figure showing winding arrangement example (three-phase simple synchronous motor) of a double three-phase permanent magnet synchronous motor"   "Figure showing an example of winding arrangement (six-phase synchronous motor) of a double three-phase permanent magnet synchronous motor"   "Figure showing winding arrangement example (three-phase reverse synchronous motor) of double three-phase winding permanent magnet synchronous motor"   "Figure showing one example each of single-phase mutual induction circuit and double three-phase winding permanent magnet synchronous motor virtual vector circuit"   “Figure showing the relationship between three types of biaxial Cartesian coordinate systems”   “A diagram showing an example of the configuration of a double synchronous motor drive system using a high speed mode current controller and a low speed mode current canceller on a dq synchronous coordinate system according to the present invention”   “A diagram showing an example of the configuration of a low-speed mode current canceller on a dq synchronous coordinate system according to the present invention”.   “A diagram showing an example of the configuration of a double synchronous motor drive system using a low-speed mode current canceller on an αβ fixed coordinate system according to the present invention”   "Figure showing the configuration of a current controller using a conventional voltage-type interwinding decoupler"

Hereinafter, preferred embodiments of the present invention will be described in detail with reference to the drawings.

An embodiment of a double synchronous motor drive system using all the inventions of claims 1 to 3 with respect to the double synchronous motor is shown in FIG. The drive system is mainly composed of a double synchronous motor (including a rotor and a stator) 1, a power converter 2 (dashed line block display), and a current control device 3 (dashed line block display). The internal configuration of the power converter and the internal configuration of the current control device are basically the same for the first winding and the second winding. In consideration of this point, these are basically described mainly for the first winding, and will be described individually only when there is a difference between the first winding and the second winding.

The power conversion device includes a power converter 21 and a current detector 22 for first and second windings. The current control device 3 mainly includes a signal conversion unit 32 (dashed line block display) for converting the stator currents and stator voltage command values of the first and second windings, and a high-speed mode for controlling the high-speed mode current. It comprises a current controller 33 (shown by a broken line block) and a low-speed mode current canceller 34 that is responsible for canceling / suppressing the low-speed mode current. Supplementally, a phase detector 311 for detecting the rotor phase used in the signal conversion unit 32 and a speed detector 312 for detecting the rotor speed used in the low-speed mode current canceller are included. The phase detector 311 and the speed detector 312 are shared by current control of both windings. In the signal converter 32, a three-phase two-phase converter 321a, a two-phase three-phase converter 321b, a vector rotator are provided for each winding so that the current control of the first and second windings can be performed independently. 322a and 322b are configured.

Regarding the three-phase two-phase converter for the second winding (2 × 3 matrix) and the two-phase three-phase converter (3 × 2 matrix), depending on the difference in the winding arrangement of the double synchronous motor, Some changes are required. That is, regarding SR (•) in FIG. 6, in the case of a three-phase simple synchronous motor (see FIG. 1) and a three-phase reverse synchronous motor (see FIG. 3), the following equation (20a) is used, and a six-phase synchronous motor is used. In the case of (see FIG. 2), the following equation (20b) is used.

In the figure, for the sake of simplicity, a plurality of scalar signals are regarded as one vector signal, and the plurality of scalar signal lines are represented by one thick signal line. Note that the symbols r, s, and t of the voltage and current vector signals are a signal on the dq synchronous coordinate system, a signal on the αβ fixed coordinate system, and a signal (three-phase signal) on the uvw coordinate system, respectively. Is shown.

The signal conversion unit 32 constituting the current control device 3 is basically the same as the conventional one. Since this device is well known to those skilled in the art, further explanation is omitted. The core of the present invention is the high-speed mode current controller 33 and the low-speed mode current canceller 34 that constitute the current control device 3. Hereinafter, the high-speed mode current controller 33 and the low-speed mode current canceller 34 shown in FIG. 6 will be described.

The high-speed mode current controller 33 includes a first winding high-speed mode current controller 331 and a second winding high-speed mode current controller 332, and the high-speed mode current control voltage command values v1 ^to *, v2 ^to * are generated. The high-speed mode current controller 33 in FIG. 6 is configured according to the first and second aspects of the invention. This configuration is based on the equations (16b) and (17b) presented in the present invention. This equation shows the dynamic characteristics of the high-speed mode current dominated by the stator resistance and leakage inductance. The high-speed mode current controllers 331 and 332 of the windings can basically adopt any structure. When a simple PI structure is employed, the high-speed mode current controllers 331 and 332 of the windings are described as the following equations (21a) and (21b), respectively.

In order to achieve the intended purpose “control of high-speed mode current”, the controller coefficient (PI gain) in equation (21) is determined according to the invention of claim 1 as the leakage inductance due to the mutual induction of the three-phase windings. It will be determined using the equivalent value of. An example of a controller coefficient design method using an equivalent value of leakage inductance is shown below. The controller coefficients (PI gains) d1d1 and d1d0 of the first winding d-axis in the equation (21a) are designed as follows when the expected bandwidth of the current control system is ωic and the design parameter is w1, for example. Is done.
Defining a controller coefficient specifically (22a), the (22b) type, the "equivalent value of the leakage inductance", in that it uses the ^{(σd ~ · L1d), (} σd ~ · L1i), be noted I want. The selection range of (σd ^~ ) with respect to the d-axis is specified by the expression (22c). Reference value (22c) equation d axis (.sigma.d ^~) is a leakage coefficient (.sigma.d) itself in the d-axis. In addition, the upper limit of the right side of the equation is closely related to the controller design method adopted and is a rough guide. (Σd ^˜ ) should also be referred to as “leakage coefficient equivalent value”. As apparent from the above description, (22) the leakage inductance equivalent value used for setting of the controller coefficients ^{(σd ~ · L1d), (} σd ~ · L1i) is described with reference to (2) The definition of “equivalent value of leakage inductance” is also followed. In addition, the definition of the common-mode inductance Li1 of the first winding is as shown in Equation (13). The in-phase inductance Li2 of the second winding is defined similarly.

The example of the controller coefficient design method according to the equation (22) is for the first equation of the equation (21a), that is, the controller coefficient of the high-speed mode current controller for the first winding d-axis. The design method of the controller coefficient of the high-speed mode current controller for the first winding q-axis can be obtained only by changing the axis corresponding leg in the equation (22) to “d → q”. Similarly, the design method of the controller coefficient of the high-speed mode current controller for the second winding d-axis can be obtained only by changing the winding corresponding leg in the equation (22) to “1 → 2”. The design method of the controller coefficient of the high-speed mode current controller for the second winding q-axis is as follows. In Equation (22), the winding corresponding leg is “1 → 2” and the axis corresponding leg is “d → q”. You can only change it.

It should be pointed out that a structure other than the PI structure may be adopted as the structure of the high-speed mode current controllers 331 and 332 of the windings. In addition, it should be pointed out that there are various controller coefficient design methods using “equivalent value of leakage inductance” in addition to the equation (22). In the embodiment shown in FIG. 6, the high-speed mode current controller 33 is configured on the dq synchronous coordinate system in accordance with the invention of claim 2. Instead, it should be pointed out that the high-speed mode current controller can be configured on the γδ quasi-synchronous coordinate system or the αβ fixed coordinate system. The present invention does not exclude these configurations.

It is comprised according to invention of Claim 3. This configuration is based on the equations (16c) and (17c) presented in the present invention.
did. In practice, it is desirable to change this generation as follows using the mutual inductance equivalent value and the winding current equivalent value.

A detailed configuration of the low-speed mode current canceller 34 in FIG. 6 is shown in FIG. The first winding low-speed mode current canceller 341 in FIG. 7 is configured based on equation (23), and the second winding low-speed mode current canceller 342 is configured based on equation (24). However, (23), (24) a current equivalent value ⁱ¹ ~ used for expression, i2 as the ^~, has become those utilizing current true value. Current equivalent value ⁱ¹ ~, i2 definition of ^~ are as previously described.

Mutual inductance equivalent value ^M1d ～ utilized for the low-speed mode current canceller 341, 342 of each ^winding, M1q ^~, M2d ^~, M2q range ^between are generally as follows.
In equation (25), the upper limit of the mutual inductance equivalent value is the corresponding self-inductance. This is based on the fact that when the mutual induction (magnetic coupling) is strong, the leakage coefficient is small, the leakage inductance is small, and the mutual inductance is approximately equal to the self-inductance. In the low-speed mode current canceller 341 for the first winding of the equation (23) and the low-speed mode current canceller 342 for the second winding of the equation (24), the same mutual inductance equivalent values M1d ^to M1q ^to M2d ^to it is not necessary to use M2Q ^~, you may utilize a different value for each low-speed mode current canceller for each winding. Taken M1d ^～ as an example, more specifically, in a first winding for low-speed mode current canceller 341, (23) and the low-speed mode current canceller 341, for the second winding (24) relates to M1d ^～ Different values may be used. FIG. 7 shows a configuration in which different values are used for each winding low-speed mode current canceller as the mutual inductance equivalent values to be used in each winding low-speed mode current canceller 341 and 342. That is, in FIG. 7, two intermediate signals Fai1M ^~ and Fai2M ^~ has a assumes that not necessarily equal. Two if the intermediate signal ^φ1M ~ and ^φ2M ~ and is selected mutual inductance equivalent value to be equal, the low-speed mode current voltage for offsetting the winding
it can.

Fad (s) in the equations (23) and (24) is an approximate differentiator introduced to eliminate pure differentiation, and can be simply as follows.
The design parameter ωad, which is also the bandwidth of the approximate differentiator, is selected to be equal to or greater than the bandwidth ωic of the current control system configured by the high-speed mode current controller. As a matter of course, another approximate differentiation process other than the expression (26) may be adopted.

In the embodiment using FIG. 7, the same true value (current detection value) is used as the equivalent value of the current of the first winding and the second winding, and the same true value (speed detection value) is used as the speed equivalent value. It is an example. It should be pointed out again that both signals may be replaced by corresponding values. The definitions of the current equivalent value and the speed equivalent value are as given above.

In the embodiment shown in FIG. 7, two processes of a differential equivalent process and a speed multiplication equivalent process are performed together. It should be pointed out that speed multiplication equivalent processing becomes more important during high-speed rotation, so that differentiation equivalent processing can be omitted. On the contrary, it is pointed out that the importance of the speed multiplication equivalent process is relatively reduced during low-speed rotation, and that the speed multiplication equivalent process can be omitted.

In the embodiment shown in FIG. 7, the low-speed mode current canceller 34 is configured on the dq synchronous coordinate system. It should be pointed out that substantially the same low-speed mode current canceller as in FIG. 7 can be constructed on the γδ quasi-synchronous coordinate system.

It should also be pointed out that the low-speed mode current canceller 34 can be configured on an αβ fixed coordinate system. FIG. 8 shows this state. When configuring a low-speed mode current canceller on an αβ fixed coordinate system, it is generally necessary to slightly change the configuration of FIG. However, when the d-axis and q-axis mutual inductance equivalent values are selected to be equal, no additional change is required, and the one obtained by simply removing the speed multiplication equivalent process from that shown in FIG. 7 can be used.

The present invention relates to a double synchronous motor in applications requiring efficient driving over a wide range represented by a main drive motor of a battery electric vehicle, a fuel cell electric vehicle, a hybrid electric vehicle, a high-speed electric motor for home appliances, It is suitable for a drive system of a double synchronous motor in an application that requires functional safety.

1 Double Synchronous Motor 11 Double Synchronous Motor Rotor 121 Double Synchronous Motor Stator First Winding 122 Double Synchronous Motor Stator Second Winding 2 Power Converter 21 Power Converter 22 Current Detection 3 Current controller 311 Phase detector 312 Speed detector 32 Signal converter 321a Three-phase two-phase converter 321b Two-phase three-phase converter 322a Vector rotator 322b Vector rotator 33 High-speed mode current controller 331 First winding High-speed mode current controller 332 High-speed mode current controller for second winding 34 Low-speed mode current canceller 341 Low-speed mode current canceller for first winding 342 Low-speed mode current canceller for second winding

Claims (3)

A permanent magnet synchronous motor comprising a rotor having a permanent magnet and a stator having two three-phase windings (a self-three-phase winding and another three-phase winding);
A power converter that can simultaneously supply current to two three-phase windings;
A permanent magnet synchronous motor drive system comprising a current control device for controlling a current flowing in two three-phase windings via a power converter;
A controller coefficient determined by utilizing at least the equivalent value of the leakage inductance caused by the mutual induction of the two three-phase windings, and for controlling the high-speed mode current generated by the mutual induction of the two three-phase windings. A high-speed mode current controller for each three-phase winding that generates a voltage command value for
Low-speed mode current generated by mutual induction of two three-phase windings using at least the equivalent value of mutual inductance caused by the mutual induction of two three-phase windings and the current phase equivalent value of the own three-phase winding With a low-speed mode current canceller for each three-phase winding that generates a voltage command value for its own three-phase winding for offsetting
A permanent magnet synchronous motor drive system, wherein the current control device is configured by using at least one of two different devices (a high-speed mode current controller and a low-speed mode current canceller). The N-pole phase of the rotor permanent magnet is the d-axis phase, and a 2-axis orthogonal coordinate system having a q-axis phase advance of π / 2 [rad] with respect to the d-axis is a dq-synchronized coordinate system, and dq-synchronized. When a coordinate system according to the coordinate system is a γδ quasi-synchronous coordinate system,
Feedback control of the current deviation of each three-phase winding, which is the difference between the current command value of each three-phase winding and the current response value of each three-phase winding on the dq synchronous coordinate system or γδ quasi-synchronous coordinate system The high-speed mode current controller for each three-phase winding is configured to generate a voltage command value for each three-phase winding on the dq synchronous coordinate system or the γδ quasi-synchronous coordinate system. The permanent magnet synchronous motor drive system according to claim 1. The N-pole phase of the rotor permanent magnet is the d-axis phase, and a 2-axis orthogonal coordinate system having a q-axis phase advance of π / 2 [rad] with respect to the d-axis is a dq-synchronized coordinate system, and dq-synchronized. When a coordinate system according to the coordinate system is a γδ quasi-synchronous coordinate system,
Furthermore, the current equivalent value of the three-phase winding the free-on the dq synchronization coordinate system or γδ quasi-synchronous coordinate system and i1 ^~, the current equivalent value of the other three-phase windings i2 and ^~ 2 (self DOO ^M1d ~ considerable value of the mutual inductance due to mutual induction of the three-phase windings of the ^other), M1q ^~, M2d ^~, when the M2Q ^~,
According to the following equation to generate an intermediate signal Fai1M ^~ for self three phase windings,
The intermediate signal for the generated three-phase winding is subjected to at least one of differentiation equivalent processing and speed multiplication equivalent processing to generate a voltage command value for the own three-phase winding.
2. The permanent magnet synchronous motor drive system according to claim 1, wherein the low-speed mode current canceller for each three-phase winding is configured.