JP2018201118A - Optical transmission/reception system - Google Patents

Abstract

To accurately calculate an optimum value.SOLUTION: A frequency modulation light signal is received so that electric signals of an in-phase component and a quadrature component are outputted by subjecting the frequency modulation light signal to homodyne detection based on local oscillation light. The electric signals of the in-phase component and the quadrature component are each sampled so that they are converted into digital sampling signals of the in-phase component and the quadrature component. The sampling signals of the in-phase component and the quadrature component are subjected to differential detection. Then, a phase average value is calculated on the basis of a fixed pattern having a uniform or substantially uniform occurrence frequency of signal points included in the signals obtained by the differential detection. Based on the phase average value, the phases of the sampling signals of the in-phase component and the quadrature component are reversed.SELECTED DRAWING: Figure 4

Description

  The present invention relates to an optical transmission / reception system.

  In the optical subscriber system, the PON (Passive Optical Network) system standardized by IEEE (The Institute of Electrical and Electronics Engineers, Inc.) and ITU-T (International Telecommunication Union Telecommunication Standardization Sector) is widely used. Yes. In the PON system, each of a plurality of subscriber line terminators (hereinafter referred to as “ONU” (Optical Network Unit)) is coupled to a single optical fiber via an optical splitter installed outside the accommodating station. It has a network configuration connected to a subscriber line terminal device (hereinafter referred to as “OLT” (Optical Line Terminal)) of the accommodation station.

  In the PON system, upstream signals and downstream signals are transmitted bidirectionally on the same optical fiber by different wavelengths. The downlink signal is a continuous signal in which signals addressed to ONUs installed in each of a plurality of subscriber houses are multiplexed using time division multiplexing (hereinafter referred to as “TDM” (Time Division Multiplexing)) technology. The ONU extracts and receives the signal of the time slot assigned to itself from the continuous signal branched in the optical splitter. The upstream signal is a burst signal that is transmitted intermittently by the ONU, and is combined by an optical splitter to become a TDM signal that is received by the OLT. In this way, in the PON system, since the optical fiber from the OLT of the accommodating station to the optical splitter and the OLT arranged in the accommodating station can be shared by a plurality of subscribers, high-speed optical access service exceeding giga is economical. Can be provided.

  In a PON system that is already used as a commercial system, an increase in allowable transmission line loss is considered as one of the major issues. If the permissible transmission line loss can be expanded, the number of accommodated subscribers can be increased by using an optical splitter with a large number of branches, and the accommodation area can be expanded by extending the transmission distance. Is possible.

  To solve this problem, an optical amplifier is placed in the OLT and this optical amplifier is used as a postamplifier to increase the output intensity of the transmitter, while this optical amplifier is used as a preamplifier to improve the sensitivity of the receiver. Therefore, methods for expanding the allowable transmission line loss have been widely studied. However, the effect of improving the reception sensitivity as a preamplifier is smaller than the increase in output intensity as a postamplifier of an optical amplifier. Therefore, in order to obtain an allowable transmission line loss equivalent to that of the downlink signal, it is necessary to further improve the reception sensitivity of the uplink signal.

  In order to improve the reception sensitivity of uplink signals, it has been studied to apply a digital coherent transmission technique to a PON system (for example, see Non-Patent Document 1). By applying the reception technique in the digital coherent transmission technique, it is possible to significantly improve the reception sensitivity as compared with the intensity modulation direct detection method that has been used in the PON system so far.

Further, it is possible to further improve the reception sensitivity by 3 dB by using the phase modulation signal instead of the intensity modulation signal. In recent years, quaternary phase modulation signals, that is, QPSK (Quadrature Phase Shift Keying) signals have been used in large-capacity relay transmission systems. The signal generation of QPSK is performed using a lithium niobate (LiNbO ₃ ) Mach-Zehnder modulator (hereinafter referred to as “LN modulator”). The LN modulator is a device suitable for a long-distance transmission system because it has excellent characteristics in terms of modulation band and coupling with an optical fiber. However, since this device has a length on the order of several centimeters, it is not practical to apply to a subscriber network, particularly to an ONU.

As a small-sized modulator replacing the LN modulator, an indium phosphide (InP) semiconductor Mach-Zehnder modulator (hereinafter referred to as “InP modulator”) has been developed and is beginning to be commercialized. In the InP modulator, since the refractive index change of the material that causes phase modulation is larger than that of LiNbO ₃ , the device can be manufactured with a length on the order of millimeters. Further, since the InP modulator is a semiconductor device, it can be monolithically integrated with an optical component such as a semiconductor laser or a semiconductor optical amplifier (SOA (Semiconductor Optical Amplifier)) to further reduce the size of the transceiver. However, the InP modulator currently has problems such as yield, and it is necessary to develop a further cost reduction technique for application to an access system.

  As a technique for generating a phase modulation signal without using these expensive Mach-Zehnder modulators, there is a direct modulation system in which a semiconductor laser drive current is driven by a transmission signal. However, what can be directly modulated by the transmission signal is not the phase of the light output from the semiconductor laser but the frequency. This frequency modulation is realized by driving the semiconductor laser in a region where the adiabatic chirp is dominant by setting the bias current of the semiconductor laser high.

As an example, FIG. 14A shows an optical spectrum when a semiconductor laser is directly modulated by a binary digital baseband signal. As shown in FIG. 14A, a “1” signal having a large drive current (hereinafter referred to as “mark”) is shifted to the high frequency side by Δf with respect to the center frequency (f ₀ ), and “ The “0” signal (hereinafter referred to as “space”) shifts to the low frequency side by Δf. Actually, since the output intensity of the semiconductor laser changes at the same time by changing the drive current of the semiconductor laser, the light intensity of the optical spectrum is slightly lower than the mark side as shown in FIG. 14B. To do.

  Data is transmitted by adjusting the amount of frequency change so that the phase change after a lapse of a certain time caused by the frequency change becomes π or zero so that the signal can be received by differential detection described later. At this time, when the phase difference from the previous bit is π or zero, it is necessary to differentially encode the transmission signal in advance so as to receive a mark or a space, respectively. As described above, the signal generated by the direct modulation method in which the driving current of the semiconductor laser is driven by the transmission signal is a signal that can be used as a phase modulation signal although it is a frequency modulation signal on the surface.

  As a method of generating a phase modulation signal using this frequency modulation, a method called continuous phase frequency shift keying (hereinafter referred to as “CPFSK”) and a conventional phase modulation, that is, PSK (Phase Shift Keying). There is a technique called). In order to avoid confusion with a general phase modulation method in which the latter PSK directly generates a phase modulation signal corresponding to a transmission signal using the Mach-Zehnder modulator described above, in the following, phase modulation by direct modulation, That is, it is called DM-PSK (Direct Modulation-PSK).

  The relationship between DM-PSK and CPFSK is shown in FIG. In FIG. 15, the horizontal axis represents a frequency modulation modulation index (referred to as FM modulation index in FIG. 15), and the vertical axis represents a duty ratio. Hereinafter, in order to show the difference between DM-PSK and CPFSK, a case where binary digital data is transmitted will be described as an example. When the bit rate of the signal is B, the modulation index m is defined by the following equation (1).

  16 and 17 show the operation principle of DM-PSK. As shown in FIG. 16, an optical pulse according to the duty ratio (D) is output in the mark section, and at the end of the optical pulse, the modulation index of frequency modulation so that the optical phase changes by π with respect to the previous bit, Or the intensity | strength of an optical pulse is adjusted (for example, refer nonpatent literature 2). Since the optical phase does not change after the end of the optical pulse, the optical phase change when 1 bit time elapses remains π. At this time, there is a relationship of the following equation (2) between the modulation index (m) and the duty ratio (D).

  In the method shown in FIG. 16, the optical phase continues to increase with time. On the other hand, as shown in FIG. 17, the intensity of the light pulse in the mark section may be alternately output as the bit that becomes larger than the intensity of the space section (for example, non-patent document). 3). In the method shown in FIG. 17, the optical phase does not increase with the passage of time, and becomes either zero or π. Here, assuming that the bit rate is B, signal reception can be performed by differential detection using a signal sequence delayed by 1 bit time, that is, 1 / B, in any of the systems shown in FIGS. it can.

  On the other hand, as shown in FIG. 15, in CPFSK, the duty ratio (D) is always 1 regardless of the modulation index (m). That is, in CPFSK, the optical phase change at the elapse of 1 bit time exceeds π when the modulation index (m) exceeds 0.5, and the delay time (T) required for differential detection is 1 bit time. That is, instead of 1 / B, it is represented by the following equation (3) (for example, see Non-Patent Document 4).

  When the modulation index (m) is 0.5, a phase modulation signal that can be demodulated with a minimum frequency deviation, that is, B / 4, is generated, and 1 bit time, that is, 1 / B differential detection. Signal reception is performed. This case is particularly referred to as minimum shift keying (hereinafter referred to as “MSK” (Minimum Shift Keying)), and can also be referred to as DM-PSK having a duty ratio of 1, as shown in FIG.

  As described above, a phase modulation signal generated by direct modulation such as CPFSK or DM-PSK can be received by differential detection. However, when coherent reception is performed by homodyne detection, a frequency difference between the signal light and the local oscillation light (hereinafter referred to as “local light emission”) becomes a problem. Hereinafter, the case of CPFSK will be described in detail with reference to mathematical expressions. Note that the processing shown by the following formulas is performed by converting each of the I (In-phase) component and Q (Quadrature) component signals generated by homodyne detection using a 90-degree optical hybrid detector to AD (Analog to Digital) is a process performed on a signal obtained by sampling and digitizing by a converter.

The sampling frequency is set to a value more than twice the signal bit rate B based on Shannon's sampling theorem. In the case of a multilevel signal, B is a symbol rate. In general, when local light having a center frequency of f ₀ and signal light of f ₁ are incident on a 90-degree optical hybrid detector and homodyne detection is performed, the I component and Q of the signal current output from the 90-degree optical hybrid detector The components are expressed by the following formulas (4) and (5), respectively.

Equation (4), in (5), _{f m} is the frequency difference between the local light and the signal light _{(f m = f 0 -f 1} ), φ is the phase difference. T is time, and A is a coefficient for associating the cosine function (cos) and the sine function (sin) with the photoelectric flow rate. When Expression (4) and Expression (5) are expressed in Euler by complex numbers, the following Expression (6) is obtained.

  In CPFSK, a frequency shift of Δf and −Δf is given to each of the mark and the space. Therefore, the received signal current of the mark is expressed by Equation (7), and the received signal current of the space is expressed by Equation (8). Indicated.

  When Expression (7) and Expression (8) are used, when transitioning from mark to mark, the output after differential detection is expressed by the following Expression (9).

  Similarly, the output after differential detection in the case of transition from the mark to the space is represented by Expression (10), and the output after differential detection in the case of transition from the space to the space is represented by Expression (11). The output after differential detection when transitioning from a space to a mark is expressed by equation (12).

  Equations (7) and (8) coincide with each other when the phase is φ at t = 0. Therefore, Equation (10) and Equation (12) show a case where the code transitions while maintaining phase continuity at t = 0, and signal identification is performed at t = T. Substituting t = T into Equation (10) and Equation (12) yields Equation (13) and Equation (14). As Δf, a value obtained by modifying Equation (1) was applied, and as T, the value of Equation (3) was applied.

First, consider the case of f _m = 0. Equations (9) and (14) are for the case where a mark is received, and the received signal converges at the position (0, A ² ) on the IQ plane regardless of the sign of the previous bit. Equations (11) and (13) are cases where a space is received, and the received signal converges at the position (0, -A ² ) on the IQ plane regardless of the sign of the previous bit. .

  However, when the sign is inverted with respect to the previous bit, as is clear from Equation (10) and Equation (12), the signal moves on the circumference of the received signal constellation at angular velocities of 4πΔf and -4πΔf, respectively. To do. FIG. 18A is a diagram showing this state.

Next, when f _m ≠ 0, as shown in FIG. 18B, the received signal constellation rotates by 2πf _m t on the IQ plane as compared with the case of FIG. 18A. .

Actually, the received signal constellation after differential detection has deteriorated due to the lack of bandwidth of the semiconductor laser to be used or the 90-degree optical hybrid detector, so that the predetermined signal point shown in FIG. The received signal does not converge at the position of (0, A ² ), (0, −A ² ). In order to compensate for the degradation of the received signal constellation, digital signal processing technology is used in digital coherent transmission.

A commonly used signal processing algorithm is CMA (Constant Modulus Algorithm). If CMA is used, the amplitude of the received signal constellation is FIR (Finite Impulse Response) so as to converge to a predetermined value, for example, (0, A ² ), (0, −A ² ) in FIG. The filter tap coefficient is adaptively feedback controlled. By filtering the received signal with the FIR filter into which the optimum value of the tap coefficient obtained by the feedback control is substituted, it is possible to compensate for insufficient bandwidth of the used device, that is, adaptive equalization.

  However, the CMA is effective when the circumferential radius of the received signal constellation is reduced due to the influence of insufficient bandwidth of the received signal. For example, this is the case when QPSK generated using a Mach-Zehnder modulator is used as a transmission signal.

  On the other hand, in CPFSK and DM-PSK, even if the influence of the insufficient bandwidth of the device used occurs, the circumferential radius of the received signal constellation is not reduced. Even when the different code generating the highest frequency component appears alternately, the received signal does not reach the predetermined signal point shown in FIG. 18A, but stays on the same circumference. In this case, even if CMA is used, it is not possible to compensate for the lack of bandwidth of the device used.

  Therefore, in the case of CPFSK or DM-PSK, it is necessary to apply an LMS (Least Mean Square) algorithm. When the LMS algorithm is used, the optimum value of the tap coefficient of the FIR filter is obtained by the least square method with a predetermined signal point on the IQ plane as a target value. As a target value, a training signal that is a fixed pattern signal periodically embedded in a transmission signal sequence is generally used.

Dayou Qian, Eduardo Mateo, Ming-Fang Huang, “A 105km Reach Fully Passive 10G-PON using a Novel Digital OLT”, Tu.1.B.2, ECOC (European Conference on Optical Communication) Technical Digest, 2012, OSA. “4 Gbitls, 233-km OPTICAL FIBER TRANSMISSION EXPERIMENT USING NEWLY PROPOSED DIRECT-MODULATION PSK” R.S.VODHANEL, “5 Gbit / s Direct Optical DPSK Modulation of a 1530-nm DFB Laser”, IEEE PHOTONICS TECHNOLOGY LETTERS, VOL. 1, No. 8, AUGUST 1989, pp.218-220. Katsushi Iwashita, Takao Matsumoto, “Modulation and Detection Characteristics of Optical Continuous Phase FSK Transmission System”, JOURNAL OF LIGHTWAVE TECHNOLOGY. VOL. LT-5, NO. 4, APRIL 1987, pp. 452-460.

However, when the LMS algorithm is used for CPFSK and DM-PSK, the phase rotation of the received signal constellation caused by the frequency difference between the signal light and the local light becomes a problem. Without this phase rotation, in the example of FIG. 18A, the LMS algorithm may be operated with the signal point targeting (0, A ² ) or (0, −A ² ). On the other hand, when the received signal constellation is rotated in phase as shown in FIG. 18B, the distance from the target signal point changes according to the amount of phase rotation, so the FIR using the LMS algorithm. There is a problem that the calculation of the optimum value of the tap coefficient of the filter becomes unstable.

  In view of the above circumstances, an object of the present invention is to provide a technique capable of accurately calculating an optimum value.

  One embodiment of the present invention is an optical transmission / reception system including an optical transmitter that transmits an optical signal and an optical receiver that receives the optical signal, the optical transmitter including a semiconductor laser, and transmission data. On the other hand, the semiconductor laser using, as a drive signal, a transmission signal generated by adding a fixed pattern of a plurality of predetermined signal points and the frequency of appearance of each signal point being uniform or substantially uniform A frequency modulation unit that generates and transmits a frequency-modulated optical signal, and the optical receiver receives a local-oscillation light source that outputs local oscillation light, and receives the frequency-modulated optical signal, and the local oscillation Based on the light, homodyne detection of the frequency-modulated optical signal to output an in-phase component and a quadrature component electrical signal, and a 90-degree optical hybrid detector for each of the in-phase component and the quadrature component electrical signal. An analog-to-digital converter that converts the digital sampling signal of the in-phase component and the sampling signal of the quadrature component, a differential detection unit that differentially detects the sampling signal of the in-phase component and the quadrature component, and differential detection A phase average calculation unit that calculates a phase average value based on a fixed pattern of the signal points included in the signal obtained by the step of calculating the phase of the sampling signal of the in-phase component and the quadrature component based on the phase average value. An optical transmission / reception system comprising: a phase rotation unit that rotates reversely.

  One aspect of the present invention is the above-described optical transmission / reception system, wherein the frequency modulation unit changes a transmission intensity in a reverse direction with respect to a change characteristic of an intensity of an optical signal output from the semiconductor laser with respect to the drive signal. An electroabsorption optical modulator having characteristics is provided, and the electroabsorption optical modulator performs intensity modulation on the frequency modulation optical signal based on the transmission signal.

  One aspect of the present invention is the above optical transmission / reception system, wherein the frequency modulation unit has a transmission intensity characteristic in the same direction with respect to a change characteristic of an intensity of an optical signal output from the semiconductor laser with respect to the drive signal. The electroabsorption optical modulator performs intensity modulation on the frequency-modulated optical signal based on a signal obtained by inverting the polarity of the transmission signal.

  One aspect of the present invention is the optical transmission / reception system described above, wherein the optical receiver includes an FIR filter unit that performs filtering based on a tap coefficient, the in-phase component and the quadrature component obtained by rotating the phase rotation unit in a reverse direction. To the FIR filter unit to obtain a demodulated signal, and a position on the IQ plane of the sampling signal included in the demodulated signal and a plurality of predetermined signals that become predetermined convergence target values A distance is calculated based on the position of the point on the IQ plane, and the tap of the FIR filter unit is configured so that the sampling signal converges to the nearest predetermined signal point based on the calculated distance. A tap coefficient calculation unit that calculates an optimum value of the coefficient.

  According to the present invention, it is possible to calculate the optimum value with high accuracy.

It is a figure which shows the structure of the optical transmission / reception system 100 in the 1st Embodiment of this invention. It is a block diagram which shows the structure of the optical receiver in 1st Embodiment. It is a figure which shows the frame structure of the transmission signal of the optical receiver in 1st Embodiment. It is a block diagram which shows the structure of the optical receiver in 1st Embodiment. It is a block diagram which shows the other structural example of the optical receiver in 1st Embodiment. It is a block diagram which shows the other structural example of the optical receiver in 1st Embodiment. It is FIG. (1) which shows the experimental result in 1st Embodiment. It is FIG. (2) which shows the experimental result in 1st Embodiment. It is FIG. (3) which shows the experimental result in 1st Embodiment. It is a block diagram which shows the structure of the optical receiver in the 2nd Embodiment of this invention. It is a graph which shows the characteristic of the semiconductor laser of the optical receiver of 2nd Embodiment. It is a graph which shows the characteristic of the EA modulator of the optical receiver of 2nd Embodiment. It is a block diagram which shows the other structural example of the optical receiver of 2nd Embodiment. It is a figure which shows the optical spectrum at the time of directly modulating using a semiconductor laser. It is a figure which shows the relationship between CPFSK and DM-PSK. It is FIG. (1) for demonstrating the operation principle of DM-PSK. It is FIG. (2) for demonstrating the operation principle of DM-PSK. It is a figure which shows a received signal constellation.

Hereinafter, an embodiment of the present invention will be described with reference to the drawings.
(First embodiment)
FIG. 1 is a diagram showing a configuration of an optical transmission / reception system 100 according to the first embodiment of the present invention. In the following description, a case where the optical transmission / reception system 100 is applied to a PON system will be described as an example. The PON system is an aspect of an optical transmission system. The optical transmission / reception system 100 includes an ONU 1 and an OLT 2. 1 shows a configuration in which the optical transmission / reception system 100 includes one ONU 1, the optical transmission / reception system 100 may include a plurality of ONUs 1.

The ONU 1 is installed, for example, in the home of a subscriber who receives communication service. The ONU 1 includes an optical transmitter 10. The optical transmitter 10 transmits an optical signal.
The OLT 2 is installed in a containment station, for example. The OLT 2 includes an optical receiver 20. The optical receiver 20 receives the optical signal transmitted from the optical transmitter 10.

FIG. 2 is a block diagram illustrating a configuration of the optical transmitter 10 according to the first embodiment. The optical transmitter 10 includes a frequency modulation unit 102.
The frequency modulation unit 102 includes a gate switch 1021 and a semiconductor laser 1022. Based on the transmission signal and the gate signal, the gate switch 1021 outputs a transmission signal in a section corresponding to a time section in which the gate signal is supplied to the semiconductor laser 1022 as a burst signal.

  Here, the transmission signal is a signal having the frame configuration shown in FIG. 3 and is input from the outside. The transmission signal is a continuous signal, and includes a burst signal portion to be transmitted and a fixed idle pattern portion that is superimposed between the burst signals and is not transmitted. The burst signal portion has a configuration in which a preamble (hereinafter referred to as “PA” (Preamble)) portion is added to a payload portion including transmission data.

  The PA portion is a pattern in which a predetermined fixed pattern of L signal points used for calculating the amount of phase rotation is repeated M times (hereinafter, this pattern is also referred to as an L × M symbol fixed pattern). Is included. Further, the portion other than the fixed pattern of the L × M symbol of PA includes, for example, a fixed pattern such as a training signal described later. The portion of the fixed idle pattern is entirely filled with a predetermined pattern that does not include information.

As a fixed pattern of symbols of L signal points, for example, a pseudo-random code sequence (PRBS (Pseudo Random Bit Sequence)) having a code length of 2 ⁿ −1 (n is a natural number) is applied. M is the number of repetitions determined as appropriate so that the fixed pattern has a sufficient length. If the fixed pattern of symbols of L signal points has a sufficient length, M = 1, that is, there may be no repetition.
The gate signal is input from the outside. The time interval in which the gate signal is supplied corresponds to the interval in which the burst signal exists in FIG. The gate signal is intermittently supplied a plurality of times. Therefore, the gate switch 1021 intermittently outputs a plurality of burst signals to the semiconductor laser 1022.

  The semiconductor laser 1022 uses the burst signal output from the gate switch 1021 as a drive signal, drives in a region where adiabatic chirp is dominant, performs direct modulation, and generates and outputs a frequency-modulated optical signal obtained by frequency-modulating the burst signal. . Here, the frequency-modulated optical signal is an optical signal modulated by the above-described CPFSK or DM-PSK method, for example.

  FIG. 4 is a block diagram showing the configuration of the optical receiver 20 in the first embodiment. The optical receiver 20 includes a local oscillation light source 21, a 90-degree optical hybrid detector 22, AD converters 23-1 to 23-2, and a digital signal processing unit 24. The local oscillation light source 21 outputs local light that causes interference with the received signal light.

  The 90-degree optical hybrid detector 22 includes splitters 221-1 to 221-2, a π / 2 delay unit 222, couplers 223-1 to 223-2, and balanced receivers 224-1 to 224-2. The splitter 221-1 receives the frequency-modulated optical signal of the burst signal transmitted by the optical transmitter 10 as received signal light, branches the received received signal light, and outputs it to the couplers 223-1 and 223-2. The splitter 221-2 receives the local light output from the local oscillation light source 21, branches the received local light, and outputs the branched light to the coupler 223-1 and the π / 2 delay device 222. The π / 2 delay unit 222 delays the local light output from the splitter 221-2 by π / 2 and outputs the delayed light to the coupler 223-2.

  The coupler 223-1 generates interference light by combining the reception signal light output from the splitter 221-1 and the local light output from the splitter 221-2 to interfere with each other. Branches to interference light and outputs. The coupler 223-2 generates interference light by combining and interfering the reception signal light output from the splitter 221-1 and the local light output delayed by π / 2 output from the π / 2 delay unit 222. The generated interference light is branched into two interference lights and output.

  The balanced receiver 224-1 converts the two interference lights output from the coupler 223-1 into an electrical signal, and detects and outputs the difference between the converted electrical signals as an in-phase component, that is, an I component. The balanced receiver 224-2 converts the two interference lights output from the coupler 223-2 into an electric signal, and detects and outputs the difference between the converted electric signals as an orthogonal component, that is, a Q component. The AD converter 23-1 samples the analog electrical signal of I component and outputs it to the digital signal processing unit 24 as a digital sampling signal. The AD converter 23-2 samples the analog electrical signal of the Q component and outputs it to the digital signal processing unit 24 as a digital sampling signal.

The digital signal processing unit 24 receives the digital sampling signals output from the AD converter 23-1 and the AD converter 23-2 and demodulates the received signal using a numerical calculation algorithm.
The processing performed by the optical receiver 20 is the same as that of a general homodyne receiver used in digital coherent transmission except for the processing performed by the digital signal processing unit 24. The digital signal processing unit 24 includes a differential detection unit 241, a phase average calculation unit 242, a phase rotation unit 243, a training signal detection unit 244, a tap coefficient calculation unit 245, and an FIR filter unit 246.

  The differential detection unit 241 performs differential detection based on the digital sampling signal of the I component output from the AD converter 23-1 and the digital sampling signal of the Q component output from the AD converter 23-2. Then, the burst signal is demodulated, and the demodulated burst signal PA is output to the phase average calculator 242. The phase average calculating unit 242 performs sampling of the fixed pattern of L signal point symbols for R times (where R ≦ M) among the L × M symbol fixed patterns included in the PA. To calculate the phase average value.

  The phase rotation unit 243 is configured so that the digital sampling signal of the I component output from the AD converter 23-1 and the AD converter 23-2 are output by the phase rotation amount corresponding to the phase average value calculated by the phase average calculation unit 242. The output Q component digital sampling signal is reversely rotated on the IQ plane.

  The training signal detection unit 244 detects the training signal included in the PA of the sampling signal of the I component and the Q component that are reversely rotated by the phase rotation unit 243. The tap coefficient calculation unit 245 obtains a demodulated signal of the training signal by providing the training signal detected by the training signal detection unit 244 to the FIR filter unit 246. Further, the tap coefficient calculation unit 245 causes the position of the training signal included in the obtained demodulated signal to converge to the position of a predetermined target value on the IQ plane that is predetermined for the training signal. Then, the optimum tap coefficient of the FIR filter unit 246 is calculated by adaptive feedback control. Further, the tap coefficient calculation unit 245 uses an LMS algorithm, that is, a least square method in the process of obtaining the optimum value of the tap coefficient by converging the training signal to the position of the predetermined target value on the IQ plane. .

  The FIR filter unit 246 filters and demodulates the I component and Q component sampling signals based on the optimum tap coefficient calculated by the tap coefficient calculation unit 245, and outputs the demodulated I component signal and Q component signal. .

  Here, the calculation method of the phase average value by the phase average calculation unit 242 will be described. As described above, when the CPFSK or DM-PSK signal is differentially detected, the received signal constellation is rotated in phase by the frequency difference between the local light and the received signal light. Due to this phase rotation, calculation of the optimum value of the tap coefficient of the FIR filter using the LMS algorithm becomes unstable.

  On the other hand, the phase average calculation unit 242 applies the following expression (for the L × R symbol sample points among the L × M symbol fixed patterns included in the PA output from the differential detection unit 241) 15) Or, the following equation (16) is calculated. Note that which of Equation (15) and Equation (16) is applied is predetermined for each optical receiver 20.

In Expressions (15) and (16), i indicates the i-th sample point in the L × R symbol, and Σ _i indicates that the sum of the total M sample points to be calculated is calculated. 2πf _m T indicates the amount of phase rotation of the received signal constellation on the IQ plane caused by the frequency difference between the local light and the received signal light. Here, as described above, f _m is represented by f _m = f ₀ −f ₁ when the center frequency of local light is f ₀ and the center frequency of received signal light is f ₁ . T is a delay time required for differential detection represented by Expression (3). φ _i indicates the phase on the IQ plane of the i-th sample point when there is no phase rotation.

  That is, Equation (15) is an equation for calculating the average value of the angular portion of the Euler-labeled sample point, and Equation (16) is the declination of the sum of the exponential portion of the Euler-labeled sample point, That is, it can be said that the angle portion is calculated.

When Expressions (15) and (16) are compared with Expressions (9) to (12), φ _i corresponds to π / 2, π / 2 + 4πΔft, −π / 2, and −π / 2-4πΔft. I understand that The received signal constellation at these sample points is as shown in FIG. FIG. 18A is symmetric with respect to the I axis, and the phase on the IQ plane takes positive and negative values with respect to the I axis. ) Of the portion related to φ _i is zero. As a result, the values of Equation (15) and Equation (16) are approximated to 2πf _m T, and the phase rotation amount 2πf _m T is obtained by calculating φ _ave in Equation (15) and Equation (16). it can.

The phase rotation unit 243 reversely rotates the reception signal constellation of FIG. 18B based on the phase rotation amount 2πf _m T calculated by the phase average calculation unit 242, so that the reception signal constellation of FIG. Can be obtained. Thereby, it is possible to solve the problem that the optimum value calculation of the tap coefficient of the FIR filter unit 246 using the LMS algorithm by the tap coefficient calculation unit 245 becomes unstable.

Note that if the variation in the frequency of appearance of signal points included in the fixed pattern of L × M symbols increases, the received signal constellation in FIG. 18A is no longer the I-axis target, so the average value of the portion related to φ _i Does not become zero, and an error occurs in the calculation of the phase rotation amount. Therefore, the appearance frequency of signal points included in the fixed pattern needs to be uniform or almost uniform. In the case of almost uniformity, the average value of the portion related to φ _i does not become zero, but it is sufficient that the error uniformity is negligible compared to the phase average value 2πf _m T. Further, when the transmission signal is a binary baseband signal, the mark ratio may be a pattern of signal points that is ½ or close to ½. Further, the phase average calculation unit 242 calculates the phase average value by using R times out of the fixed pattern of L × M symbols, but the value of R is the appearance frequency of signal points included in the R times. Is determined to be uniform or substantially uniform.

In addition, for example, when the above-described pseudo-random code sequence having a code length of 2 ⁿ −1 is applied as a fixed pattern of L signal point symbols, the mark rate is halved if the natural number n is small. There are times when it is not close. In this case, a method of adding one zero and setting the code length to ²ⁿ may be applied. Further, a pattern in which 1 and 0 are repeated may be applied as a fixed pattern of symbols of L signal points.

  The training signal detected from the PA by the training signal detection unit 244 may apply a fixed pattern of L × M symbols as a training signal, or apply a fixed pattern other than the fixed pattern of L × M symbols as a training signal. May be. When a fixed pattern of L × M symbols is used as a training signal, all PA portions may be fixed patterns of L × M symbols. Further, when a fixed pattern other than the L × M symbol fixed pattern is applied as a training signal, the training signal is included in the remaining section of the PA as described above.

(Experimental result)
7 and 8 show the results of signal processing of digital data offline. For this digital data, a binary CPFSK signal having a signal bit rate of 10.0 Gbit / s and a modulation index of 0.6 is experimentally generated. This is digital data obtained by sampling at / s.

  FIGS. 7A and 7B show the case where the frequencies of the local light and the signal light are substantially the same, and are diagrams showing the output obtained by differential detection and the output to which the LMS algorithm is applied, respectively. As shown in FIG. 7B, the LMS algorithm operates stably, and binary signal points are clearly separated and demodulation is performed normally.

  On the other hand, FIGS. 8A and 8B are diagrams showing the differentially detected output and the output to which the LMS algorithm is applied when the frequencies of the local light and the signal light are shifted by about 4 GHz. As shown in FIG. 8B, the operation of the LMS algorithm becomes unstable, and binary signal points cannot be separated, and demodulation has failed.

  On the other hand, FIG. 9 shows a result when the technique of the optical transmission / reception system 100 according to the present embodiment is applied to the same digital data as FIG. As shown in FIG. 9A, the phase rotation of the received constellation after differential detection is appropriately compensated. Further, as shown in FIG. 9B, even in an output to which the LMS algorithm is applied, binary signal points are clearly separated and demodulation is performed normally.

  According to the optical transmission / reception system 100 according to the first embodiment configured as described above, in the optical transmitter 10, the frequency modulation unit 102 includes the semiconductor laser 1022, and a plurality of predetermined numbers are determined for transmission data. The frequency modulation light is generated by driving the semiconductor laser 1022 using a transmission signal generated by adding a fixed pattern of signal points of which the frequency of appearance of each signal point is uniform or substantially uniform as a drive signal. A signal is generated, and a frequency-modulated optical signal corresponding to the time interval indicated by the gate signal is transmitted. In the optical receiver 20, the local oscillation light source 21 outputs local light. The 90-degree optical hybrid detector 22 receives the frequency-modulated optical signal, performs homodyne detection on the frequency-modulated optical signal based on the local light, and outputs an in-phase component and a quadrature component electrical signal. Each of the AD converters 23-1 and 23-2 samples each of the in-phase component and quadrature component electrical signals and converts them into digital in-phase component sampling signals and quadrature component sampling signals. In the digital signal processing unit 24, the differential detection unit 241 differentially detects the sampling signal of the in-phase component and the quadrature component. The phase average calculation unit 242 calculates a phase average value based on a fixed pattern of signal points included in a signal obtained by differential detection. The phase rotation unit 243 reversely rotates the phase of the sampling signal of the in-phase component and the quadrature component based on the phase average value.

  With the above configuration, even if there is a frequency difference between the local light and the frequency-modulated optical signal and phase rotation occurs, the appearance frequency of each signal point is uniform or almost uniform. By calculating the phase average value based on this, the phase component of each signal point on the IQ plane becomes zero, and the phase average value represents the amount of phase rotation. Therefore, by reversely rotating the phase based on the calculated phase rotation amount, the position of the received signal constellation can be returned to the position where there is no difference between the frequency of the received signal light and the local light. Thereby, an appropriate optimum value of the tap coefficient of the FIR filter unit 246 using the LMS algorithm by the tap coefficient calculation unit 245 can be calculated. Therefore, when the frequency modulated optical signal generated by directly modulating the semiconductor laser 1022 is subjected to homodyne detection using a 90-degree optical hybrid detector, the received signal constellation at the time of differential detection is the signal light and the local light. It is possible to receive a signal even when the phase is rotated on the IQ plane due to the frequency difference.

<Modification>
The optical transmitter 10 in the first embodiment may be configured as shown in FIG. FIG. 5 is a block diagram showing a configuration of an optical transmitter 10a applied in place of the optical transmitter 10 in the first embodiment. In the optical transmitter 10a, the same components as those of the optical transmitter 10 are denoted by the same reference numerals, and different configurations will be described below.

  The optical transmitter 10a includes a frequency modulation unit 102a. The frequency modulation unit 102 a includes a semiconductor laser 1022 and a semiconductor optical amplifier 1023. The semiconductor optical amplifier 1023 receives the gate signal, changes the drive current according to the gate signal to increase or decrease the gain, and corresponds to the time interval indicated by the gate signal among the continuous frequency modulated optical signals output from the semiconductor laser 1022. Outputs a frequency-modulated optical signal. The time interval in which the gate signal is supplied corresponds to the interval in which the burst signal exists in FIG. Since the gate signal is intermittently supplied a plurality of times, the semiconductor optical amplifier intermittently outputs a plurality of burst signals.

  In the optical transmitter 10 shown in FIG. 2, since the semiconductor laser 1022 is driven and directly modulated using the burst signal as a drive signal, the output wavelength of the semiconductor laser 1022 may drift greatly at the rising edge of the burst signal. On the other hand, in the optical transmitter 10a shown in FIG. 5, since the semiconductor laser 1022 is not driven by the burst signal but is driven by the continuous transmission signal, the output wavelength of the semiconductor laser 1022 is stabilized. Therefore, the optical transmitter 10a is suitable for multiplexing optical signals at high-density wavelength intervals.

Further, the optical receiver 20 in the first embodiment may be configured as shown in FIG. FIG. 6 is a block diagram showing a configuration of an optical receiver 20a applied in place of the optical receiver 20 in the first embodiment. In the optical receiver 20a, the same components as those of the optical receiver 20 are denoted by the same reference numerals, and different configurations will be described below. The optical receiver 20a includes a local oscillation light source 21, a 90-degree optical hybrid detector 22, AD converters 23-1 to 23-2, and a digital signal processing unit 24a. The digital signal processing unit 24a includes a differential detection unit 241, a phase average calculation unit 242, a phase rotation unit 243, a tap coefficient calculation unit 245a, and an FIR filter unit 246.
The tap coefficient calculating unit 245a calculates an optimum value of the tap coefficient by causing a predetermined signal point on the IQ plane to converge as a target value.

  The tap coefficient calculation unit 245a obtains a demodulated signal of the sampling signal by providing the FIR filter unit 246 with the sampling signals of the I component and the Q component that are reversely rotated by the phase rotation unit 243. The tap coefficient calculation unit 245a determines the position on the IQ plane of the sampling signal included in the demodulated signal, that is, the sample point on the IQ plane in order to identify which signal point is the target. A distance between the position of the point and the positions of a plurality of predetermined signal points determined in advance is calculated.

  Based on the calculated distance, the tap coefficient calculation unit 245a selects a predetermined signal point closest to the sample point position as the target value position of the sample point. The tap coefficient calculation unit 245a adaptively performs feedback control by applying an LMS algorithm, that is, a least square method so that the sample points converge at the position of the selected target value, and the tap coefficient of the FIR filter unit 246 Calculate the optimum value.

  In the optical receiver 20 illustrated in FIG. 4, the tap coefficient calculation unit 245 calculates the optimum value of the tap coefficient of the FIR filter unit 246 based on the training signal detected by the training signal detection unit 244. Since it is known at which position on the IQ plane the training signal converges, the optimum value can be calculated in a relatively short time. On the other hand, since the time position accuracy required for training detection by the training signal detection unit 244 is very high, a large amount of resources are required for the signal processing circuit in the digital signal processing unit 24 for realizing this accuracy. In contrast, the tap coefficient calculation unit 245a of the optical receiver 20a calculates the optimum value of the tap coefficient without using the training signal. In this method, the time required for calculating the optimum value increases by the distance calculation, but since no training signal is used, the scale of the signal processing circuit in the digital signal processing unit 24a is set to the digital signal processing unit of the optical receiver 20. It can be made smaller than 24.

  The combinations of the optical transmitters 10 and 10a included in the ONU 1 and the optical receivers 20 and 20a included in the OLT 2 are, for example, the optical transmitter 10 and the optical receiver 20, the optical transmitter 10 and the optical receiver 20a, and the optical transmitter 10a. The optical transmission / reception system 100 may be configured by any combination of the optical receiver 20 and the optical receiver 20 and the optical transmitter 10a and the optical receiver 20a.

(Second Embodiment)
In the second embodiment, the configuration of the optical transmitter in the configuration of the optical transmission / reception system is different from that of the optical transmission / reception system 100 in the first embodiment.
FIG. 10 is a block diagram showing a configuration of the optical transmitter 10b according to the second embodiment of the present invention. The optical transmitter 10b includes a frequency modulation unit 102b. The frequency modulation unit 102b includes a semiconductor laser 1022, a semiconductor optical amplifier 1023, and an electroabsorption modulator 1024. The optical transmitter 10b has a configuration in which an electroabsorption modulator 1024 is newly added to the configuration of the optical transmitter 10a of the first embodiment illustrated in FIG. The same configurations as those of the optical transmitter 10a are denoted by the same reference numerals, and different configurations will be described below. In the following description, the electroabsorption modulator 1024 is referred to as an EA (Electro Absorption) modulator 1024.

  In the optical transmitter 10 and the optical transmitter 10a, when the semiconductor laser 1022 directly modulates according to the drive current, not only the output optical carrier frequency but also the intensity is simultaneously modulated. For this reason, ideal frequency modulation as shown in FIG. 14A is not performed, and there is an intensity difference between the mark shifted to the high frequency side as shown in FIG. 14B and the space shifted to the low frequency side. Will occur.

  In order to accurately obtain the phase rotation amount of the received constellation from the above equations (15) and (16), the sample points need to exist on the circumference of the same radius on the IQ plane. is there. Therefore, if the intensity difference between the mark and the space shown in FIG. 14B is large, an error occurs in the calculation of the phase rotation amount.

  In the optical transmitter 10b shown in FIG. 10, the EA modulator 1024 is used to make the light intensity difference uniform. In the optical transmitter 10b, the same transmission signal, that is, a binary digital baseband signal is input to the semiconductor laser 1022 and the EA modulator 1024 from the outside. The EA modulator 1024 performs intensity modulation on the frequency modulated optical signal output from the semiconductor laser 1022 using the input transmission signal as a control signal.

  FIG. 11 is a graph showing the relationship between the current value of the drive signal given to the semiconductor laser 1022 (hereinafter referred to as “drive current”) and the output light intensity. If an electric resistance is provided in the drive current application section, it is possible to drive with a voltage instead of a current, and therefore the case where the horizontal axis is the drive voltage is also shown.

  As shown in FIG. 11A, when the drive current increases in the positive direction, the output light intensity of the semiconductor laser 1022 increases. Further, as shown in FIG. 11B, when the drive current increases in the negative direction, the output light intensity of the semiconductor laser 1022 increases. Therefore, in any case, it is understood that if intensity modulation is directly performed with a binary digital baseband signal, an unnecessary intensity modulation component is generated together with frequency modulation. In other words, when the drive current shown in FIG. 11A increases in the positive direction, an intensity modulation component having characteristics in the same direction with respect to the characteristics of the transmission signal is generated. When the drive current shown in FIG. 11B increases in the negative direction, the characteristics are inverted, and an intensity modulation component having characteristics in the opposite direction to the characteristics of the transmission signal is generated.

  FIG. 12 is a graph showing the relationship between the control signal voltage and the transmission intensity characteristic in the EA modulator 1024. As shown in FIG. 12A, when the voltage of the control signal increases in the positive direction, the transmission intensity characteristic of the EA modulator 1024 decreases. On the other hand, as shown in FIG. 12B, when the voltage of the control signal increases in the negative direction, the transmission intensity characteristic of the EA modulator 1024 decreases. In other words, when the voltage of the control signal shown in FIG. 12A increases in the positive direction, intensity modulation in which the sign of the applied signal voltage is inverted is performed. When the voltage of the control signal shown in FIG. 12B increases in the negative direction, intensity modulation with the same sign as the applied signal voltage is performed.

  Therefore, by combining the semiconductor laser 1022 having the characteristics shown in FIG. 11A and the EA modulator 1024 having the characteristics shown in FIG. 12A, unnecessary intensity modulation components generated in the semiconductor laser 1022 are generated in the EA modulator 1024. Can be removed. The same effect can be obtained by combining the semiconductor laser 1022 having the characteristics shown in FIG. 11B and the EA modulator 1024 having the characteristics shown in FIG.

  According to the optical transceiver in the second embodiment configured as described above, the optical transmitter 10b transmits in the reverse direction with respect to the change characteristic of the intensity of the optical signal output from the semiconductor laser 1022 with respect to the drive signal. An EA modulator 1024 having intensity change characteristics is provided, and the EA modulator 1024 performs intensity modulation on the frequency-modulated optical signal based on the input transmission signal.

  Although there is a technique to make the strength of the mark and space uniform using an optical filter such as a dielectric multilayer film, etc., since a member different from a semiconductor is used, this technique reduces the size of the optical component by monolithic integration. I can't. On the other hand, in the optical transmitters 10b and 10c according to the second embodiment, the light intensity of the mark and the space is made uniform by using an optical component called the EA modulator 1024 that can be monolithically integrated. The reception sensitivity of the modulation signal can be improved by the configuration. The configuration of the optical transmitters 10b and 10c of the second embodiment is the same as the number of required optical devices as compared to the case where the InP modulator is used, but the EA modulator 1024 is a general-purpose device. It is an optical device and is currently much cheaper than an InP modulator.

<Modification>
The optical transmitter 10b in the second embodiment may be configured as shown in FIG. FIG. 13 is a block diagram showing a configuration of an optical transmitter 10c applied in place of the optical transmitter 10b in the second embodiment. The optical transmitter 10c includes a frequency modulation unit 102c. The frequency modulation unit 102c includes a semiconductor laser 1022, a semiconductor optical amplifier 1023, and an EA modulator 1024 (an electroabsorption modulator 1024 in FIG. 13). The optical transmitter 10c is different from the configuration of the optical transmitter 10b shown in FIG. 10 in that the control signal input to the EA modulator 1024 is a control signal in which the polarity of the transmission signal is reversed. The same configurations as those of the optical transmitter 10b are denoted by the same reference numerals, and different configurations will be described below. A transmission signal, that is, a binary digital baseband signal is input to the semiconductor laser 1022 from the outside, and a control signal with the polarity of the transmission signal reversed is input to the EA modulator 1024 from the outside.

  The optical transmitter 10c includes an EA modulator 1024 having a transmission intensity change characteristic in the same direction with respect to an intensity change characteristic of the optical signal output from the semiconductor laser 1022 with respect to the drive signal. The EA modulator 1024 includes: Based on the signal obtained by inverting the polarity of the input transmission signal, intensity modulation is performed on the frequency modulated optical signal. Thereby, the EA modulator 1024 can remove the intensity modulation component generated when the semiconductor laser 1022 is directly modulated.

  In the optical transmitter 10c, the combination of the characteristics of the semiconductor laser 1022 and the EA modulator 1024 to be applied is the semiconductor laser 1022 having the characteristics shown in FIG. 11A and the EA modulator 1024 having the characteristics shown in FIG. Will be combined. As another combination, the same effect can be obtained by combining the semiconductor laser 1022 having the characteristics shown in FIG. 11B and the EA modulator 1024 having the characteristics shown in FIG.

  In general, systems that handle high-speed signals often use digital baseband signals having different polarities, that is, differential signals. In such a case, it is preferable to apply the optical transmitter 10c.

  Further, in the optical transmitters 10b and 10c, the semiconductor optical amplifier 1023 is connected after the EA modulator 1024, but may be connected between the semiconductor laser 1022 and the EA modulator 1024.

<Modification common to the first embodiment and the second embodiment>
In the first and second embodiments described above, an example in which a binary digital baseband signal is mainly applied as a transmission signal has been described. However, the configuration of the present invention may be limited to the embodiment. Absent. An N-value CPFSK signal or DM-PSK signal may be generated using an N-value (N is an integer of 3 or more) digital baseband signal.
In the first and second embodiments, the configuration in which the optical transmission / reception system 100 is applied to the PON system has been described. However, the optical transmission / reception system 100 may be applied to other optical transmission systems.

DESCRIPTION OF SYMBOLS 10, 10a, 10b, 10c ... Optical transmitter, 102, 102a, 102b, 102c ... Frequency modulation part, 1021 ... Gate switch, 1022 ... Semiconductor laser, 1023 ... Semiconductor optical amplifier, 1024 ... Electroabsorption modulator, 20, 20a ... optical receiver, 21 ... local oscillation light source, 22 ... 90-degree optical hybrid detector, 221-1, 221-2 ... splitter, 222 ... π / 2 delay unit, 223-1, 223-2 ... coupler, 224 -1, 244-2 ... balanced receivers, 23-1, 23-2 ... AD converters, 24 ... digital signal processing unit, 241 ... differential detection unit, 242 ... phase average calculation unit, 243 ... phase rotation unit 244: Training signal detection unit 245 245a Tap coefficient calculation unit 246 FIR filter unit

Claims (4)

An optical transmission / reception system comprising an optical transmitter for transmitting an optical signal and an optical receiver for receiving the optical signal,
The optical transmitter is
A transmission that has a semiconductor laser and is generated by adding a fixed pattern of a plurality of predetermined signal points to the transmission data, each of which has a uniform or almost uniform frequency of appearance. A frequency modulation section for driving the semiconductor laser as a drive signal to generate and transmit a frequency modulated optical signal;
With
The optical receiver is:
A local oscillation light source that outputs local oscillation light;
A 90-degree optical hybrid detector that receives the frequency-modulated optical signal, performs homodyne detection on the frequency-modulated optical signal based on the local oscillation light, and outputs an in-phase component and a quadrature component electrical signal;
An AD converter that samples each of the in-phase component and the quadrature component electrical signal and converts the digital signal into the in-phase component sampling signal and the quadrature component sampling signal;
A differential detector for differentially detecting the sampling signal of the in-phase component and the quadrature component;
A phase average calculating unit that calculates a phase average value based on a fixed pattern of the signal points included in a signal obtained by differential detection;
A phase rotation unit that reversely rotates the phase of the sampling signal of the in-phase component and the quadrature component based on the phase average value;
An optical transmission / reception system comprising: The frequency modulation unit is
An electroabsorption optical modulator having a transmission characteristic change characteristic in the reverse direction with respect to a change characteristic of an optical signal intensity output from the semiconductor laser with respect to the drive signal,
The optical transmission / reception system according to claim 1, wherein the electroabsorption optical modulator performs intensity modulation on the frequency-modulated optical signal based on the transmission signal. The frequency modulation unit includes an electroabsorption optical modulator having transmission intensity characteristics in the same direction with respect to a change characteristic of an intensity of an optical signal output from the semiconductor laser with respect to the drive signal,
The optical transmission / reception system according to claim 1, wherein the electroabsorption optical modulator performs intensity modulation on the frequency-modulated optical signal based on a signal obtained by inverting the polarity of the transmission signal. The optical receiver is:
An FIR filter that performs filtering based on the tap coefficients;
A sampling signal of the in-phase component and the quadrature component rotated reversely by the phase rotation unit is supplied to the FIR filter unit to obtain a demodulated signal, and the position of the sampling signal included in the demodulated signal on the IQ plane The distance is calculated based on the positions on the IQ plane of a plurality of predetermined signal points serving as predetermined convergence target values, and based on the calculated distance, the sampling signal is closest to the sampling signal. A tap coefficient calculation unit that calculates an optimum value of the tap coefficient of the FIR filter unit so as to converge to a predetermined signal point;
The optical transmission / reception system according to any one of claims 1 to 3, further comprising:

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