JP2018094456A - Vibrator drive device and vibration generation device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a vibrator drive device capable of efficiently suppressing with a simple structure, a change of amplitude of vibration following to change of a mechanical load of a vibrator.SOLUTION: A vibrator drive device 2 comprises: a drive voltage generation part 10 for generating an AC drive voltage Vo1 supplied to a vibrator VT; a detection part 20 for detecting a current Io flowing to the vibrator VT; and a control part 30 for controlling the drive voltage generation part 10 so that a frequency of the drive voltage Vo1 is varied. The control part 30 acquires a detection result of the detection part 20 while varying a frequency of the drive voltage Vo1, then, based on the acquired detection result, searches an antiresonant frequency fa in which a current Io flowing to the vibrator VT becomes a minimum value, in a first operation mode, and holds the frequency of the drive voltage Vo1 to the antiresonant frequency fa searched in the first operation mode in a second operation mode.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a vibrator driving device that generates vibration by supplying electric energy to a vibrator and a vibration generator using the vibrator driving device.

  A resonator that converts electric energy into mechanical vibration by the piezoelectric effect generally has a resonance frequency and an anti-resonance frequency. The impedance of the vibrator has a minimum value at the resonance frequency and a maximum value at the anti-resonance frequency.

  A crystal vibrator or a piezoelectric vibrator used for an oscillator such as a clock is usually driven at a resonance frequency. Also, in an apparatus that handles relatively large energy, such as an ultrasonic machine, a bonding machine, or a handpiece, the vibrator is often driven at a resonance frequency.

  In the case of a vibrator used for an oscillator, the mechanical load is substantially constant, but in the case of a vibrator that vibrates an object in an ultrasonic machine or the like, the mechanical load varies greatly depending on the use state. . Since the resonance frequency of the vibrator changes according to the mechanical load, if the drive frequency of the vibrator is constant, the drive frequency deviates from the resonance frequency and the impedance of the vibrator increases. While the impedance of the vibrator increases, if the voltage supplied to the vibrator is constant, the power supplied to the vibrator decreases and the amplitude of vibration decreases.

  In order to avoid such a problem, in the apparatus described in Patent Document 1 below, control is performed so that the drive frequency of the vibrator follows the resonance frequency and the drive current of the vibrator is kept constant.

JP 2003-70798 A

  However, in order to perform control to keep the drive current of the vibrator constant, a circuit that detects the vibrator current or a circuit that varies the drive current of the vibrator according to the difference between the current detection result and the target value A protection circuit against overvoltage during abnormal operation is required. In order to keep the power supplied to the vibrator constant, a more complicated circuit is required.

  Further, in the apparatus described in Patent Document 1, a PLL that controls the drive frequency so that the phase of the voltage and current of the vibrator coincides with each other in order to make the drive frequency of the vibrator follow the resonance frequency. However, since the phase change decreases as the mechanical load of the vibrator increases, the phase is easily unlocked in the PLL control. Therefore, when the load increases, the PLL control that follows the resonance point may not work.

  The present invention has been made in view of such circumstances, and an object of the present invention is to provide a vibrator driving device capable of effectively suppressing a change in amplitude of vibration accompanying a change in mechanical load of the vibrator with a simple configuration. An object of the present invention is to provide a vibration generator using the

  A vibrator driving device according to a first aspect of the present invention is a vibrator driving device that drives a vibrator that converts electrical energy into mechanical vibration energy, and is an alternating drive voltage supplied to the vibrator. A drive voltage generator that generates a current, a detector that detects a current flowing through the vibrator or power supplied to the vibrator, and a control that controls the drive voltage generator so that the frequency of the drive voltage changes Part. In the first operation mode, the control unit acquires the detection result of the detection unit while changing the frequency of the drive voltage, and based on the acquired detection result, the current flowing through the transducer or the transducer The anti-resonance frequency at which the supplied power is a minimum value is searched, and the frequency of the drive voltage is held at the anti-resonance frequency searched in the first operation mode in the second operation mode.

  A vibration generating apparatus according to a second aspect of the present invention includes a vibrator that converts electrical energy into mechanical vibration energy, and a vibrator driving apparatus according to the first aspect that drives the vibrator.

  According to the present invention, it is possible to suppress fluctuations in the amplitude of the vibrator due to fluctuations in the mechanical load of the vibrator, while having a simple configuration without using feedback control.

It is a figure which shows an example of a structure of the vibration generator which concerns on embodiment of this invention. It is a figure which shows an example of a structure of a rectangular wave generation circuit and a waveform conversion circuit. It is a figure which shows an example of a structure of a switching amplifier and a filter. It is a figure for demonstrating the influence of the inductor provided in series with the vibrator | oscillator, and shows the simulation result of the frequency characteristic of a circuit. It is a figure for demonstrating the influence of the inductor provided in series with the vibrator | oscillator, and shows the simulation result of the several frequency characteristic obtained by different inductance value. It is a figure for demonstrating the influence of the mechanical load of a vibrator | oscillator, and shows the simulation result of the frequency characteristic at the time of no load and a load. It is a figure for demonstrating the influence of the mechanical load of a vibrator | oscillator, and shows the simulation result of the other frequency characteristic at the time of no load and with load. It is a figure for demonstrating the influence of the capacitor of a filter, and shows the simulation result of the frequency characteristic of a circuit. It is a figure which shows the signal waveform of each part of the circuit at the time of operation | movement, and shows the simulation result of the transient analysis at the time of no load and with load. It is a flowchart which shows an example of the search process of an antiresonance frequency. It is a figure which shows the signal waveform of each part of a circuit at the time of performing the search process of an antiresonance frequency.

  Hereinafter, a vibration generator according to an embodiment of the present invention will be described with reference to the drawings. FIG. 1 is a diagram illustrating an example of a vibration generator 1 according to the present embodiment. A vibration generator 1 shown in FIG. 1 includes a vibrator VT that converts electrical energy into mechanical vibration energy, and a vibrator driver 2 that supplies driving power to the vibrator VT. The vibration generator 1 is a device that transmits the vibration of the vibrator VT to an object, such as a dental hand tool, an electric knife, or an electric toothbrush. Therefore, the mechanical load of the vibrator VT changes according to the usage state (the presence or absence of the object, the material of the object, how to transmit vibration, etc.). The vibrator VT is, for example, a Langevin vibrator, and has a resonance frequency and an anti-resonance frequency as described later.

  In the example of FIG. 1, the vibrator driving device 2 includes a driving voltage generation unit 10, a detection unit 20, a control unit 30, and a transformer TR.

[Drive voltage generator 10]
The drive voltage generator 10 generates an alternating drive voltage Vo1 supplied to the vibrator VT via the transformer TR. The drive voltage generator 10 adjusts the frequency and amplitude of the drive voltage Vo <b> 1 according to the control of the controller 30.

  For example, as shown in FIG. 1, the drive voltage generation unit 10 includes a rectangular wave generation circuit 11, a waveform conversion circuit 12, a band pass filter 13, a range switching circuit 14, a switching amplifier 15, and a filter 16.

  The rectangular wave generation circuit 11 generates a rectangular wave S11 having an amplitude and a frequency set under the control of the control unit 30. The waveform conversion circuit 12 converts the rectangular wave S11 output from the rectangular wave generation circuit 11 into a triangular wave S12.

FIG. 2 is a diagram illustrating an example of the configuration of the rectangular wave generation circuit 11 and the waveform conversion circuit 12.
In the example of FIG. 2, the rectangular wave generation circuit 11 switches the DC voltage generated by the low-pass filter 111 and the low-pass filter 111 that generates a DC voltage by averaging the PWM signal S30A output from the control unit 30. A switch circuit 112 that generates a rectangular wave S11 is included.

  Low pass filter 111 includes a resistor R1 and a capacitor C connected in series. The PWM signal S30A is input to one end of the resistor R1, the other end of the resistor R1 is connected to one end of the capacitor C1, and the other end of the capacitor C1 is connected to the ground. As the ratio (duty ratio) of the high level period in one cycle of the PWM signal S30A increases, the level of the DC voltage generated in the capacitor C1 increases.

  The switch circuit 112 includes two switches SW1 and SW2 that are complementarily turned on and off according to the control signal S30B output from the control unit 30. The switches SW1 and SW2 are connected in series between one end and the other end of the capacitor C1, and output a rectangular wave S11 from the connection middle point. When the switch SW1 is turned on and the switch SW2 is turned off, the rectangular wave S11 is at a high level, and when the switch SW1 is turned off and the switch SW2 is turned on, the rectangular wave S11 is at a low level.

  In the example of FIG. 2, the waveform conversion circuit 12 includes capacitors C2 and C3, resistors R2 to R6, and an operational amplifier OP1. The reference voltage Vr is input to the inverting input terminal of the operational amplifier OP1 through the resistors R2 and R3 connected in series. The reference voltage Vr is input to the non-inverting input terminal of the operational amplifier OP1 via the resistors R4 and R5 connected in series. A rectangular wave S11 is input to one end of the capacitor C2, and the other end of the capacitor C2 is connected to a connection midpoint N1 of the resistors R2 and R3. The resistor R6 and the capacitor C3 are connected in parallel between the inverting input terminal and the output terminal of the operational amplifier OP1.

  Since the reference voltage Vr is input to the non-inverting input terminal of the operational amplifier OP1, the voltage at the inverting input terminal of the operational amplifier OP1 is approximately equal to the reference voltage Vr. Assuming that the capacitor C2 has a sufficiently large capacitance compared to the capacitor C3, the voltage change of the capacitor C2 accompanying the movement of charges from the capacitor C2 to the capacitor C3 is minute. In this case, the voltage at the connection midpoint N1 during the period when the rectangular wave S11 is at the high level is substantially constant, and the voltage at the connection midpoint N1 during the period when the rectangular wave S11 is at the low level is also approximately constant. That is, the voltage at the connection midpoint N1 is a rectangular wave, and its amplitude is approximately equal to the amplitude of the rectangular wave S11. As a result, a substantially constant current flows through the capacitor R3 through the resistor R3 in each of the period in which the rectangular wave S11 is at the high level and the period in the low level. Further, the direction of the current of the capacitor C3 is reversed according to the change in the level of the rectangular wave S11. Therefore, the output voltage of the operational amplifier OP1 becomes a triangular wave S12 that repeatedly rises and falls with a constant slope as time passes.

Returning to FIG.
The bandpass filter 13 outputs a sine wave S13 obtained by removing harmonic components from the triangular wave S12 generated in the waveform conversion circuit 12. If the frequency change range of the rectangular wave S11 (triangular wave S12) is sufficiently smaller than the harmonics, the passband of the bandpass filter 13 may be constant. The bandpass filter 13 is configured by, for example, a plurality of stages of bandpass filters connected in cascade.

  The range switching circuit 14 switches the amplitude range of the sine wave S14 output to the switching amplifier 15 according to the control signal of the control unit 30. For example, the range switching circuit 14 includes an attenuation circuit that switches an attenuation ratio in accordance with a control signal from the control unit 30, and outputs a sine wave S14 in which the amplitude of the sine wave S13 is attenuated.

  The switching amplifier 15 outputs a drive voltage VD having a pulse waveform in which the pulse width, the pulse density, and the like are modulated according to the sine wave S14. The switching amplifier 15 generates a drive voltage VD having a pulse waveform by switching a DC voltage Vps supplied from a DC power supply, for example, with a switch element (FET or the like). The filter 16 removes the switching component included in the drive voltage VD output from the switching amplifier 15, and outputs a sine wave drive voltage Vo1 corresponding to the sine wave S14.

FIG. 3 is a diagram illustrating an example of the configuration of the switching amplifier 15 and the filter 16.
In the example of FIG. 3, the switching amplifier 15 includes a first amplifier unit A <b> 1 and a second amplifier unit A <b> 2 that switch and output a high level voltage and a low level voltage, respectively, and a triangular wave generator 153.

  The first amplifier unit A1 outputs a first drive voltage VD1 that is pulse-width modulated in accordance with the sine wave S14. The second amplifier unit A2 outputs the second drive voltage VD2 that is pulse-width modulated in accordance with the inverted sine wave S14 obtained by inverting the phase of the sine wave S14. The difference between the first drive voltage VD1 and the second drive voltage VD2 corresponds to the drive voltage VD.

  In the example of FIG. 3, the first amplifier unit A1 includes switch elements M11 and M12, a switch drive circuit 151, and a comparator CP1. The comparator CP1 compares the triangular wave Str having a constant frequency generated by the triangular wave generator 153 with the sine wave S14, and outputs a pulse width modulation signal as a comparison result. The switch drive circuit 151 complementarily drives the switch elements M11 and M12 according to the pulse width modulation signal output from the comparator CP1. The switch elements M11 and M12 are transistors such as FETs, for example, and are connected in series between a power supply line to which the DC voltage Vps is supplied and the ground. When the switch elements M11 and M12 are turned on and off in a complementary manner, the drive voltage VD1 becomes a pulse wave that switches alternately between the DC voltage Vps and the zero voltage (ground level).

  In the example of FIG. 3, the second amplifier unit A2 includes switch elements M21 and M22, a switch drive circuit 152, a comparator CP2, and an inverting amplifier 154. The inverting amplifier 154 outputs an inverted sine wave S14A obtained by inverting the phase of the sine wave S14. The comparator CP2 compares the triangular wave Str and the inverted sine wave S14A, and outputs a pulse width modulation signal as the comparison result. The switch drive circuit 152 drives the switch elements M21 and M22 in a complementary manner according to the pulse width modulation signal output from the comparator CP2. The switch elements M21 and M22 are transistors such as FETs, for example, and are connected in series between a power supply line to which the DC voltage Vps is supplied and the ground. When the switch elements M21 and M22 are turned on and off in a complementary manner, the drive voltage VD2 becomes a pulse wave that is alternately switched between the DC voltage Vps and the zero voltage (ground level).

  In the example of FIG. 3, the filter 16 includes a first choke Lc1 and a second choke Lc2, a first capacitor Cf11, and a second capacitor Cf21. The first choke Lc1 and the second choke Lc2 are provided in a path through which a current flows from the switching amplifier 15 to the primary coil L1 of the transformer TR. That is, the first choke Lc1 is provided in a path through which a current flows from the output of the first amplifier unit A1 to one terminal of the primary coil L1, and the second choke Lc2 is connected to the primary coil L1 from the output of the second amplifier unit A2. It is provided in a path through which current flows to the other terminal.

  The first capacitor Cf11 is connected between a path through which a current flows from the first choke Lc1 to the primary coil L1 and the grant. The second capacitor Cf21 is connected between a path through which a current flows from the second choke Lc2 to the primary coil L1 and the ground.

  In the example of FIG. 3, the filter 16 includes a series circuit of a resistor Rf11 and a capacitor Cf12 connected in parallel to the first capacitor Cf11, and a series circuit of a resistor Rf21 and a capacitor Cf22 connected in parallel to the second capacitor Cf21. .

[Trans TR]
The primary coil L1 of the transformer TR receives the sinusoidal drive voltage Vo1 output from the filter 16. The secondary coil L2 of the transformer TR generates a drive voltage Vo2 that is boosted with respect to the drive voltage Vo1. This drive voltage Vo2 is supplied to the vibrator VT.

[Detection unit 20]
The detection unit 20 detects a current flowing through the vibrator VT. In the example of FIG. 1, the detection unit 20 includes a current transformer CT, a shunt resistor Rs, a current detection amplifier 21, a full-wave rectifier circuit 22, and a low-pass filter 23.

  The primary coil Lt1 of the current transformer CT is provided in the path of the current Io flowing through the vibrator VT. A current Is proportional to the current Io flows through the secondary coil Lt2 of the current transformer CT. The shunt resistor Rs generates a voltage corresponding to the current Is flowing through the secondary coil Lc2.

  The current detection amplifier 21 amplifies the voltage generated in the shunt resistor Rs with a gain corresponding to the control signal of the control unit 30. The full wave rectification circuit 22 performs full wave rectification on the sine wave output signal S 21 of the current detection amplifier 21. The low-pass filter 23 smoothes the full-wave rectified signal S22 output from the full-wave rectifier circuit 22, and outputs a signal S23 that is approximately proportional to the amplitude of the current Io.

[Control unit 30]
The control unit 30 is a circuit that controls the overall operation of the vibration generator 1 and includes, for example, a computer. The control unit 30 controls the drive voltage generation unit 10 so that the frequency and amplitude of the drive voltage Vo1 change. Specifically, the control unit 30 controls the frequency (period) of the rectangular wave S11 by the control signal S30A input to the switch circuit 112 of the rectangular wave generation circuit 11, and inputs the control signal to the low-pass filter 111 of the rectangular wave generation circuit 11. The amplitude of the rectangular wave S11 is controlled by the PWM signal S30A. In addition, the control unit 30 controls the amplitude range of the sine wave S <b> 14 input to the switching amplifier 15 by switching the attenuation ratio of the range switching circuit 14.

  Further, the control unit 30 controls the amplitude of the current Io detected by the detection unit 20 by controlling the gain of the current detection amplifier 21.

  The vibration generator 1 of the present embodiment includes an operation stop mode for stopping the generation of the drive voltage Vo1 in the drive voltage generator 10 and a first operation mode for searching for an anti-resonance frequency of the vibrator VT as modes related to the operation state. And a second operation mode for driving the vibrator VT. When the control signal Sc instructing to generate the drive voltage Vo1 is input from an external control device or the like in the operation stop mode, the control unit 30 shifts from the operation stop mode to the first operation mode.

  In the first operation mode, the control unit 30 acquires the detection result (the output signal S23 of the low-pass filter 23) of the detection unit 20 while changing the frequency of the drive voltage Vo1, and based on the acquired detection result, the vibrator The anti-resonance frequency at which the current Io flowing through the VT becomes a minimum value is searched. For example, the control unit 30 increases the current Io detected by the detection unit 20 from a decrease while changing the frequency of the drive voltage Vo1 from the high frequency side to the low frequency side of a predetermined frequency range in which the anti-resonance frequency can be included. Search for a frequency that turns to. After searching for the antiresonance frequency in the first operation mode, the control unit 30 transitions to the second operation mode.

  In the second operation mode, the control unit 30 generates the drive voltage Vo1 by the drive voltage generation unit 10 and holds the frequency of the drive voltage Vo1 at the anti-resonance frequency of the search result of the first operation mode.

  Further, in the second operation mode, the control unit 30 maintains the set value of the amplitude of the drive voltage Vo1 for the drive voltage generation unit 10 at a constant value according to the control signal Sc. Specifically, the control unit 30 maintains the duty ratio of the PWM signal S30A input to the low-pass filter 111 of the rectangular wave generation circuit 11 and the attenuation ratio in the range switching circuit 14 at constant values according to the control signal Sc. . That is, the control unit 30 does not perform feedback control that keeps the current Io flowing through the vibrator VT constant.

  Here, the function of the inductors (first choke Lc1, second choke Lc2) in the vibration generator 1 having the above-described configuration will be described with reference to FIGS.

  FIG. 4 is a diagram for explaining the influence of the inductor provided in series with the vibrator VT, and shows the simulation result of the frequency characteristics of the circuit. FIG. 4A shows a circuit on which simulation was performed. As shown in FIG. 4A, the electrical equivalent circuit of the vibrator VT includes a series circuit of an inductor Lt, a capacitor Ct1, and a resistor Rt, and a capacitor Ct2 connected in parallel to the series circuit. In the circuit shown in FIG. 4A, a vibrator VT, an inductor Lm, and a resistor Rs are connected in series, and a voltage Vi is supplied to the series circuit from a signal source.

  4B to 4D show simulation results when the value of the inductor Lm is zero, and FIGS. 4E to 4G show simulation results when the value of the inductor Lm is 20 [mH]. 4B and 4E show the frequency characteristics of the impedance Z as seen from the signal source of the voltage Vi, FIGS. 4C and 4F show the frequency characteristics of the current I supplied from the signal source, and FIGS. 4G indicates the frequency characteristic of the power W output from the signal source. In this simulation, the value of the inductor Lt is 0.92 [H], the value of the capacitor Ct1 is 24.3 [pF], the value of the resistor Rt is 332 [Ω], and the value of the capacitor Ct2 is 1.02 [nF. ], The value of the resistor Rs is 50 [Ω].

  When the value of the inductance Lm is zero (FIGS. 4B to 4D), the impedance Z becomes a minimum value (the current I and the power W become a maximum value) and the impedance Z becomes a maximum value (the current I , The electric power W becomes a minimum value). The resonance frequency fr is mainly caused by series resonance of the inductor Lt and the capacitor Ct1, and the anti-resonance frequency fa is mainly caused by parallel resonance of the inductor Lt and the capacitor Ct2. On the other hand, when the value of the inductor Lm is 20 [mH] (FIGS. 4E to 4G), the resonance frequency fx is generated in addition to the resonance frequency fr and the antiresonance frequency fa. The resonance frequency fx is mainly caused by series resonance of the inductor Lm and the capacitor Ct2.

  As can be seen by comparing FIG. 4C and FIG. 4F, the resonance frequency fx is generated at a frequency higher than the anti-resonance frequency fa, so that changes in the current I and power W near the anti-resonance frequency fa are emphasized. That is, the change in the current I and the power W according to the change in the frequency of the voltage Vi is emphasized. Accordingly, by providing the inductor Lm in series with the vibrator VT, it becomes easy to detect the change in the current I with respect to the change in the frequency. Therefore, the search for the anti-resonance frequency fa using the minimum value of the current I is easy. Become.

  FIG. 5 is a diagram illustrating the difference in frequency characteristics depending on the value of the inductor. FIG. 5A shows a circuit on which simulation is performed, and FIGS. 5B to 5G show simulation results of frequency characteristics. The circuit shown in FIG. 5A differs from the circuit shown in FIG. 4A in that the inductor Lc and the vibrator VT are separated on the primary side and the secondary side of the transformer TR. In the circuit shown in FIG. 5A, the equivalent circuit of the vibrator VT is the same as FIG. 4A.

  The transformer TR is a transformer that boosts the voltage Vo2 on the secondary side with respect to the voltage Vo1 on the primary side, the value of the primary coil L1 is 0.4 [mH], the value of the secondary coil L2 is 250 [mH], The coupling coefficient is 0.9993. The impedance on the primary side of the transformer TR is approximately 625 times (250 mH / 0.4 mH) when viewed from the secondary side of the transformer TR. Therefore, the inductor Lm of 20 [mH] in the circuit shown in FIG. 4A corresponds to the inductor Lc of 32 [μH] in the circuit shown in FIG. 5A. By providing the transformer TR, it becomes easy to drive the vibrator VT having a large impedance at the anti-resonance frequency fa with a high voltage, and it is easy to reduce the value of the inductor Lc for generating the resonance frequency fx.

  5B and 5C show simulation results when the value of the inductor Lc is 10 [μH], and FIGS. 5D and 5E show simulation results when the value of the inductor Lc is 20 [μH]. 5E shows a simulation result when the value of the inductor Lc is 33 [μH]. 5B, 5D, and 5F show the impedance Zo of the vibrator VT, and FIGS. 5C, 5E, and 5G show the current Io that flows through the vibrator VT. As can be seen from the simulation results of FIG. 5, the resonance frequencies fr and fx decrease as the value of the inductor Lc increases, while the anti-resonance frequency fa hardly changes depending on the value of the inductor Lc. Therefore, by appropriately setting the value of the inductor Lc, it is possible to set the resonance frequency fx so as to increase the change in the current Io with respect to the frequency change near the antiresonance frequency fa.

  Next, the influence of the mechanical load of the vibrator VT on the frequency characteristics will be described with reference to FIGS.

  FIG. 6 is a diagram for explaining the influence of the mechanical load of the vibrator VT, and shows the simulation result of the frequency characteristics when there is no load and when there is a load. The circuit on which the simulation of FIG. 6 is performed is the same as FIG. 5A. 6A to 6C show simulation results of the impedance Zo and the current Io when there is no load. 6D to 6F show simulation results of the impedance Zo and the current Io when there is a load. In this simulation, the value of the resistance Rt of the vibrator VT is 332 [Ω] when there is no load, and 6.3 [kΩ] when there is a load.

  As can be seen from the simulation result of FIG. 6, even if the mechanical load of the vibrator VT changes, the anti-resonance frequency fa does not change much. Therefore, by maintaining the driving frequency of the vibrator VT at the anti-resonance frequency fa searched for when there is no load, it is generally anti-resonance when there is a load, without performing control (such as PLL) to follow the anti-resonance frequency fa. It is possible to drive the vibrator VT at a frequency close to the frequency fa.

  6A and 6D, the impedance Zo at the antiresonance frequency fa becomes smaller when there is a load than when there is no load. Therefore, the current Io (6.76 mA: FIG. 6C) at the anti-resonance frequency fa when there is a load is larger than the current Io (0.36 mA: FIG. 6F) at the anti-resonance frequency fa when there is no load. As described above, since the current Io increases as the mechanical load increases, the drive power corresponding to the mechanical load can be obtained without performing feedback control to keep the current and power supplied to the vibrator VT constant. Can be supplied to the vibrator VT.

  FIG. 7 is a diagram showing another simulation result regarding the influence of the mechanical load of the vibrator VT. FIG. 7A shows a circuit on which simulation was performed. 7B and 7C show simulation results when there is no load, and FIGS. 7D and 7E show simulation results when there is a load. The circuit illustrated in FIG. 7A includes a parallel circuit of an inductor Ls, a resistor Rs1, and a capacitor Cs as an equivalent circuit of the inductor Lc, and a resistor Rs2 connected in series with the parallel circuit. The circuit shown in FIG. 7A includes capacitors Cf1 and Cf2 connected between the path of the current flowing from the inductor Lc to the primary coil L1 of the transformer TR and the ground. A resistor Rf1 is connected in series to the capacitor Cf2, and this series circuit and the capacitor Cf1 are connected in parallel. The capacitors Cf1 and Cf2 correspond to the capacitors Cf11, Cf12, Cf21, and Cf22 of the filter 16 in FIG. 3, and constitute a low-pass filter together with the inductor Lc.

  In the simulation of FIG. 7, the value of the inductor Ls is 33 [μH], the value of the resistor Rs1 is 56.3 [kΩ], the value of the capacitor Cs is 10.1 [pF], the value of the resistor Rs2 is 67 [mΩ], The values of the capacitors Cf1 and Cf2 are 0.1 [μF], and the value of the resistor Rf1 is 10 [Ω]. In this simulation, the value of the inductor Lt constituting the vibrator VT is 1.58 [H], the value of the capacitor Ct1 is 15.7 [pF], and the value of the capacitor Ct2 is 823 [pF]. The value of the resistance Rt of the vibrator VT is 1.1 [kΩ] when there is no load, and 6.0 [kΩ] when there is a load. Note that the parameters of the transformer TR are the same as those in FIG. 5A.

  FIG. 7B and FIG. 7D show the frequency characteristic (Vo1 / Vi) of the gain of the voltage Vo1 with respect to the voltage Vi of the signal source and the frequency characteristic (Vo2 / Vi) of the gain of the voltage Vo2 with respect to the voltage Vi of the signal source, respectively. 7C and 7E show frequency characteristics of the current Io flowing through the vibrator VT. As can be seen from these frequency characteristics, the change in the antiresonance frequency fa accompanying the change in the mechanical load of the vibrator VT is very small. Further, as can be seen by comparing FIG. 7C and FIG. 7E, the current Io at the anti-resonance frequency fa is minute when there is no load, and increases as the mechanical load increases.

  Next, the influence of the capacitors (Cf11, Cf12, Cf21, Cf22) of the filter 16 on the frequency characteristics will be described with reference to FIG. FIGS. 8A and 8B show the results of simulation of frequency characteristics in a circuit in which capacitors Cf1 and Cf2 and resistor Rf1 are deleted from the circuit shown in FIG. 7A. FIG. 8C and FIG. 8D show the results of simulation of frequency characteristics with the circuit shown in FIG. 7A.

  8A and 8C show the frequency characteristic (Vo1 / Vi) of the gain of the voltage Vo1 with respect to the voltage Vi of the signal source and the frequency characteristic (Vo2 / Vi) of the gain of the voltage Vo2 with respect to the voltage Vi of the signal source, respectively. 8B and 8D show frequency characteristics of the current Io flowing through the vibrator VT. As can be seen from these simulation results, the capacitors Cf1 and Cf2 that constitute the low-pass filter together with the inductor Lc shift the resonance frequency fx to a lower frequency. This is equivalent to the state in which the capacitance of the capacitor Ct2 is apparently increased by connecting the capacitors Cf1 and Cf2 in parallel with the capacitor Ct2 of the vibrator VT via the transformer TR. The inductor Lc and the capacitor Ct2 This is because the series resonance frequency with the above decreases. The antiresonance frequency fa is substantially constant regardless of the presence or absence of the capacitors Cf1 and Cf2. Therefore, by appropriately setting the values of the capacitors Cf1 and Cf2 together with the value of the inductor Lc, the resonance frequency fx can be set so as to increase the change in the current Io with respect to the frequency change near the antiresonance frequency fa. is there.

  Next, signal waveforms at various parts of the circuit during operation will be described with reference to FIG. FIG. 9 shows the simulation results of the transient analysis when there is no load and when there is a load.

  FIG. 9A shows a circuit in which a simulation of transient analysis is performed, and the same reference numerals in FIG. 9A and FIG. 3 indicate the same components. An equivalent circuit of the first choke Lc1 and the second choke Lc2 is the same as the inductor Lc in FIG. 7A. The values of the capacitors Cf11 and Cf21 are 0.1 [μF], the values of the capacitors Cf12 and Cf22 are 0.01 [μF], and the values of the resistors Rf11 and Rf21 are 10 [Ω]. Other configurations of the circuit illustrated in FIG. 9A are the same as those of the circuit illustrated in FIG. 7A.

  9B to 9G show the waveforms of the respective parts when there is no load (Rt = 1.1 [kΩ]), and FIGS. 9H to 9M show the waveforms of the respective parts when there is a load (Rt = 6.0 [kΩ]). Waveform is shown. As can be seen from these waveforms, the filter 16 removes high-frequency components from the rectangular drive first drive voltage VD1 and second drive voltage VD2 output from the switching amplifier 15. Therefore, a sine wave voltage Vo1 (difference between Vo1 + and Vo1-) is input to the transformer TR.

  Next, the search process for the anti-resonance frequency fa will be described. FIG. 10 is a flowchart illustrating an example of a search process for the anti-resonance frequency fa by the control unit 30.

  When the control unit 30 shifts to the first operation mode in which the search process for the anti-resonance frequency fa is performed, each variable (Sa, Sb, Fg, T, Tx) used for the search process for the anti-resonance frequency fa is initialized (ST100). ). Further, the control unit 30 sets the amplitude of the drive voltage Vo1 set in the drive voltage generation unit 10 to the minimum value (ST105).

  First, the control unit 30 outputs a pulse train having a period T from the rectangular wave generation circuit 11 (ST110). In the initial state, the period T is set to a minimum value. For example, the control unit 30 outputs the rectangular wave S11 having the period T from the rectangular wave generating circuit 11 for 16 periods. As a result, the driving voltage Vo2 having the period T is supplied to the vibrator VT over 16 periods.

  In the second half of the pulse train generated in step ST110, control unit 30 integrates the detected value of current Io of vibrator VT by detecting unit 20, and stores the current integrated value as “Sa” (ST115). For example, the control unit 30 integrates the detected value of the current Io corresponding to the last four periods in the 16 periods.

  Control unit 30 compares current integrated value Sa with threshold value TH (ST120). When the integrated current value Sa is equal to or greater than the threshold value TH, the control unit 30 proceeds to step ST150. In step ST150, the control unit 30 stores the result of adding the increment value ΔT to the period T as a new period T. Thereby, since the period T becomes large, the frequency decreases.

  If the period T has reached the maximum value Tmax, the control unit 30 ends the search process for the antiresonance frequency fa (ST155). When the period T is smaller than the maximum value Tmax, the control unit 30 stores the latest current integrated value Sa in “Sb” (ST160), and returns to step ST110. Control unit 30 repeats the processing after step ST110 described above.

  When determining in step ST120 that the integrated value Sa is smaller than the threshold value, the control unit 30 compares the current integrated value Sa calculated in step ST115 with the previous current integrated value Sb stored in step ST160. (ST125). When current integrated value Sa exceeds current integrated current value Sb, control unit 30 determines whether the value of flag data Fg is zero (ST130). In the initial state, the value of the flag data Fg is set to zero. When the value of the flag data Fg is zero in step ST130, the control unit 30 subtracts a value obtained by subtracting the increment value ΔT from the current period T (that is, the period T used to calculate the previous current integrated value Sb). It is stored in “TX” as a candidate for a definite value (ST135). In this case, the control unit 30 sets the value of the flag data Fg to “1” (ST140). After step ST135 and ST140, control unit 30 proceeds to step ST150 and repeats the same processing as described above.

  When the value of the flag data Fg is “1” in step ST130, the control unit 30 acquires the cycle Tx stored in step ST135 as a confirmed value of the search result (ST170). The control unit 30 holds the period of the rectangular wave S11 output from the rectangular wave generation circuit 11 in the second operation mode at the period Tx acquired as a definite value in step ST170.

  When the current integrated value Sa is equal to or less than the previous current integrated value Sb in step ST125, the control unit 30 resets the value of the flag data Fg to zero (step ST145), and proceeds to step ST150. For this reason, when the current integrated value Sa of this time exceeds the previous current integrated value Sb only once, the cycle Tx stored in step ST135 is discarded and is not acquired as a fixed value.

  FIG. 11 is a diagram showing signal waveforms (S14, S23, S22) of each part of the circuit when the search process of the antiresonance frequency fa is executed, and the drive frequency is scanned from a frequency higher than the antiresonance frequency fa to a lower frequency. An example of a signal waveform in the case of the above is shown.

  In the example of FIG. 11A, scanning of the driving frequency is started at time t1, and the driving frequency is continuously scanned after reaching the anti-resonance frequency fa, and scanning of the driving frequency is performed at time t2 when the second minimum value is exceeded. Is stopped. In this example, there are two frequencies at which the current Io has a minimum value, but the anti-resonance frequency fa to be searched is a frequency at which the minimum value is initially set. Therefore, in the example of FIG. 11B, after the scanning of the driving frequency is started at time t3, the operation of the driving frequency is stopped at time t4 when the antiresonance frequency fa (first minimum value of the current Io) is searched. Thus, by scanning the drive frequency from a frequency higher than the anti-resonance frequency fa to be searched toward a lower frequency, another anti-resonance frequency existing at a lower frequency than the anti-resonance frequency fa is erroneously detected. It will never be explored.

(Summary)
According to the present embodiment, in the first operation mode, based on the detection result of the detection unit 20 acquired while changing the frequency of the drive voltage Vo1, the antiresonance frequency fa at which the current Io flowing through the vibrator VT becomes a minimum value. Is searched. In the second operation mode, the frequency of the drive voltage Vo1 is held at the anti-resonance frequency fa searched for in the first operation mode. When the vibrator VT is driven at the anti-resonance frequency fa, the impedance of the vibrator VT decreases as the mechanical load of the vibrator VT increases. When the impedance of the vibrator VT is reduced, the power supplied to the vibrator VT is increased even if the voltage of the vibrator VT remains constant. That is, even if constant current control or the like is not performed, the power supplied to the vibrator VT increases as the mechanical load of the vibrator VT increases. Accordingly, it is possible to effectively suppress a change in amplitude due to a change in the mechanical load of the vibrator VT while having a simple configuration that does not perform feedback control such as constant current control. Further, at the anti-resonance frequency fa, the impedance of the vibrator VT increases as the mechanical load of the vibrator VT decreases, so that power consumption in a state where the mechanical load is small can be reduced.

  According to the present embodiment, since the drive voltage Vo1 is boosted by the transformer TR and applied to the vibrator VT, sufficient power can be supplied to the vibrator VT that is in a high impedance state at the anti-resonance frequency fa. .

  According to the present embodiment, the output impedance of the drive voltage generator 10 as viewed from the secondary side (transducer VT side) of the transformer TR is larger than that in the case where the transformer TR is not provided. As a result, even if the driving frequency of the vibrator VT deviates from the anti-resonance frequency fa and the impedance of the vibrator VT decreases, the voltage applied to the vibrator VT becomes small, so that an excessive current flows through the vibrator VT. Can be effectively prevented.

  According to the present embodiment, the resonator VT corresponding to a change in the frequency of the drive voltage Vo1 due to resonance between the inductor (first choke Lc1, second choke Lc2) and the electrostatic capacitance component (capacitor Ct2) of the resonator VT. Therefore, the accuracy of searching for the anti-resonance frequency in the first operation mode can be improved.

  According to the present embodiment, the inductors (first choke Lc1 and second choke Lc2) are provided in the path through which current flows from the drive voltage generator 10 to the primary coil L1 of the transformer TR. As a result, the inductance of the inductor (first choke Lc1, second choke Lc2) seen from the secondary side (vibrator VT side) of the transformer TR is apparently increased, and therefore the inductor (first choke Lc1, second choke Lc2). Can be reduced in size.

  According to the present embodiment, the choke (first choke Lc1 and second choke Lc2) included in the filter 16 also serves as an inductor that causes resonance with the electrostatic capacitance component (capacitor Ct2) of the vibrator VT. Therefore, an increase in the number of parts can be suppressed.

  According to the present embodiment, the rectangular wave S11 generated in the rectangular wave generating circuit 11 is converted into the triangular wave S12 in the waveform converting circuit 12, and the sine wave S14 is extracted from the triangular wave S12 by the bandpass filter 13. Therefore, compared with the case where the sine wave S14 is directly extracted from the rectangular wave S11 in the band pass filter 13, the harmonic component contained in the extracted sine wave S14 can be reduced.

  According to this embodiment, when the control signal Sc instructing the generation of the drive voltage Vo1 is input in the operation stop mode, the anti-resonance frequency fa is searched in the first operation mode, and then the drive voltage generator 10 drives the drive voltage Vo1. The voltage Vo1 is generated, and the frequency of the drive voltage Vo1 is held at the antiresonance frequency fa of the search result. That is, every time the output of the drive voltage Vo1 is started in the second operation mode, the search for the anti-resonance frequency fa is performed. Therefore, when the operation stop mode and the second operation mode are frequently repeated (for example, when the drive state and the stop state are repeated in a dental hand tool, an electric knife, an electric toothbrush, etc.), the frequency of the drive voltage Vo1 The error from the anti-resonance frequency fa of the vibrator VT can be reduced.

  According to the present embodiment, in the first operation mode for searching for the antiresonance frequency fa, the antiresonance frequency fa is searched from the high frequency side to the low frequency side of the predetermined frequency range. As a result, the secondary resonance mode anti-resonance frequency existing at a frequency lower than the target anti-resonance frequency fa (for example, the frequency of the second current minimum value in FIG. 11A) is erroneously searched. It can be effectively prevented.

  Although one embodiment of the present invention has been described above, the present invention is not limited to the above-described embodiment, and includes various variations.

  For example, in the embodiment described above, the current Io of the vibrator VT is detected by the detection unit 20, but in the other embodiments of the present invention, the power supplied to the vibrator VT is detected by the detection unit 20. Also good. In this case, the control unit 30 may detect the minimum value of power in the search process for the anti-resonance frequency fa.

  Regarding the technical idea of the present invention grasped based on the above-described embodiment, the following additional notes are disclosed.

[Appendix 1]
A vibrator driving device that drives a vibrator that converts electrical energy into mechanical vibration energy, a driving voltage generator that generates an alternating drive voltage supplied to the vibrator, and a current that flows through the vibrator Or a detection unit that detects power supplied to the vibrator, and a control unit that controls the drive voltage generation unit so that the frequency of the drive voltage changes, and the control unit is configured to operate in a first operation mode. , The detection result of the detection unit is acquired while changing the frequency of the drive voltage, and based on the acquired detection result, the current flowing through the vibrator or the power supplied to the vibrator becomes a minimum value. A vibrator drive device that searches for an anti-resonance frequency and maintains the frequency of the drive voltage at the anti-resonance frequency searched for in the first operation mode in the second operation mode.
[Appendix 2]
The vibrator drive apparatus according to appendix 1, further comprising a transformer that boosts the drive voltage generated by the drive voltage generator and applies the boosted voltage to the vibrator.
[Appendix 3]
Provided in a path through which current flows from the drive voltage generation unit to the primary coil of the transformer, resonance occurs with the capacitive component of the vibrator at a frequency higher than the anti-resonance frequency, and the resonance causes the drive voltage to The vibrator driving apparatus according to appendix 2, further comprising an inductor that emphasizes a change in current of the vibrator according to a change in frequency.
[Appendix 4]
The drive voltage generation unit includes a switching amplifier and a filter that removes a switching component included in the output voltage of the switching amplifier, and the transformer outputs an output voltage of the switching amplifier from which the switching component is removed in the filter The vibrator according to claim 3, wherein the filter includes at least one choke provided in a path through which a current flows from the switching amplifier to the primary coil, and the choke functions as the inductor. Drive device.
[Appendix 5]
The switching amplifier includes a first amplifier unit and a second amplifier unit that switch and output a high level voltage and a low level voltage, respectively, and the filter passes from the output of the first amplifier unit to one terminal of the primary coil. A first choke provided in a path through which a current flows, and a second choke provided in a path through which a current flows from the output of the second amplifier unit to the other terminal of the primary coil, the first choke and the The vibrator driving device according to appendix 4, wherein the second choke functions as the inductor.
[Appendix 6]
The drive voltage generation unit includes a rectangular wave generation circuit that generates a rectangular wave, a waveform conversion circuit that converts the rectangular wave into a triangular wave, a bandpass filter that extracts a sine wave from the triangular wave and inputs the sine wave to the switching amplifier; The vibrator driving apparatus according to appendix 5, wherein the control unit controls a frequency of the rectangular wave generated in the rectangular wave generating circuit.
[Appendix 7]
The control unit shifts to the first operation mode when a control signal instructing to generate the drive voltage is input in an operation stop mode in which generation of the drive voltage by the drive voltage generation unit is stopped, Any one of appendix 1 to appendix 6, wherein after searching for the anti-resonance frequency in the first operation mode, the mode is shifted to the second operation mode, and the drive voltage is generated by the drive voltage generator in the second operation mode. The vibrator drive device according to one.
[Appendix 8]
In the first operation mode, the control unit is detected by the detection unit while changing the frequency of the drive voltage from a high frequency side to a low frequency side in a predetermined frequency range in which the anti-resonance frequency can be included. The vibrator driving device according to any one of appendix 1 to appendix 7, wherein a search is made for a frequency at which the current or electric power changes from a decrease to an increase.
[Appendix 9]
A vibration generating device comprising: a vibrator that converts electrical energy into mechanical vibration energy; and the vibrator driving device according to any one of supplementary notes 1 to 8 that drives the vibrator.

DESCRIPTION OF SYMBOLS 1 ... Vibration generator, 2 ... Vibrator drive device, 10 ... Drive voltage generator, 11 ... Rectangular wave generation circuit, 111 ... Low pass filter, 112 ... Switch circuit, 12 ... Waveform conversion circuit, 13 ... Band pass filter, 14 DESCRIPTION OF SYMBOLS ... Range switching circuit, 15 ... Switching amplifier, A1 ... 1st amplifier part, A2 ... 2nd amplifier part, 151 ... Switch drive circuit, 152 ... Switch drive circuit, 153 ... Triangular wave generator, 154 ... Inverting amplifier, 16 ... Filter , Lc1 ... first choke, Lc2 ... second choke, 20 ... detection unit, CT ... current transformer, 21 ... current detection amplifier, 22 ... full wave rectifier circuit, 23 ... low pass filter, 30 ... control unit, TR ... transformer, L1 ... primary coil, L2 ... secondary coil, VT ... vibrator, CP1, CP2 ... comparator, OP1 ... operational amplifier, fr, fx ... both Frequency, fa ... anti-resonance frequency

Claims (9)

A vibrator driving device that drives a vibrator that converts electrical energy into mechanical vibration energy,
A drive voltage generator for generating an alternating drive voltage supplied to the vibrator;
A detection unit for detecting a current flowing through the vibrator or a power supplied to the vibrator;
A controller that controls the drive voltage generator so that the frequency of the drive voltage changes,
The controller is
In the first operation mode, the detection result of the detection unit is acquired while changing the frequency of the drive voltage, and based on the acquired detection result, the current flowing through the vibrator or the power supplied to the vibrator is Search for the anti-resonance frequency that is the minimum value,
In the second operation mode, the frequency of the drive voltage is held at the anti-resonance frequency searched in the first operation mode.
Vibrator driving device. A transformer that boosts the drive voltage generated by the drive voltage generator and applies the boosted voltage to the vibrator;
The vibrator driving apparatus according to claim 1. Provided in a path through which current flows from the drive voltage generation unit to the primary coil of the transformer, resonance occurs with the capacitive component of the vibrator at a frequency higher than the anti-resonance frequency, and the resonance causes the drive voltage to Having an inductor that emphasizes a change in current of the vibrator in response to a change in frequency;
The vibrator driving apparatus according to claim 2. The drive voltage generator is
A switching amplifier,
A filter for removing a switching component included in the output voltage of the switching amplifier,
The transformer boosts the output voltage of the switching amplifier from which the switching component is removed in the filter as the drive voltage,
The filter includes at least one choke provided in a path through which a current flows from the switching amplifier to the primary coil,
The choke acts as the inductor;
The vibrator driving apparatus according to claim 3. The switching amplifier includes a first amplifier unit and a second amplifier unit that switch and output a high level voltage and a low level voltage, respectively.
The filter is
A first choke provided in a path through which current flows from the output of the first amplifier section to one terminal of the primary coil;
A second choke provided in a path through which a current flows from the output of the second amplifier section to the other terminal of the primary coil;
The first choke and the second choke act as the inductor;
The vibrator driving device according to claim 4. The drive voltage generator is
A rectangular wave generating circuit for generating a rectangular wave;
A waveform conversion circuit for converting the rectangular wave into a triangular wave;
A sine wave is extracted from the triangular wave, and includes a bandpass filter that is input to the switching amplifier,
The control unit controls a frequency of the rectangular wave generated in the rectangular wave generating circuit;
The vibrator driving apparatus according to claim 5. The control unit shifts to the first operation mode when a control signal instructing to generate the drive voltage is input in an operation stop mode in which generation of the drive voltage by the drive voltage generation unit is stopped, After searching for the anti-resonance frequency in the first operation mode, the mode shifts to the second operation mode, and the drive voltage is generated by the drive voltage generation unit in the second operation mode.
The vibrator driving device according to any one of claims 1 to 6. In the first operation mode, the control unit is detected by the detection unit while changing the frequency of the drive voltage from a high frequency side to a low frequency side in a predetermined frequency range in which the anti-resonance frequency can be included. Search for the frequency at which the current or power changes from low to high;
The vibrator driving device according to claim 1. A vibrator that converts electrical energy into mechanical vibration energy;
A vibration generator comprising: the vibrator driving device according to claim 1 that drives the vibrator.

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