JP2018014875A - Motor control device, motor driving device, and motor driving system - Google Patents

Abstract

PROBLEM TO BE SOLVED: To make it possible to switch a method for controlling a stepping motor.SOLUTION: A motor control device according to one embodiment is a motor control device that controls a stepping motor on the basis of a first current command value depending on open loop control and a second current command value depending on closed loop control. The motor control device comprises: a control method selection part that selects the open loop control or the closed loop control on the basis of a speed command value; and a command value output part that, when the open loop control is selected, outputs a first target value as a first current command value and when the closed loop is selected, outputs a second target value as a second current command value.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a motor control device, a motor drive device, and a motor drive system.

  Stepping motors having a two-phase coil and a magnetized rotor are known. In this stepping motor, when a driving current having a phase difference of 90 degrees is supplied to a two-phase coil and the phase of the driving current is advanced, the rotor rotates following the phase of the driving current. For this reason, the stepping motor can accurately control the drive by open loop control that does not use a sensor such as an encoder.

  However, in open-loop control, a constant amplitude drive current is supplied to the coil regardless of the load and drive speed, so the rotor cannot follow the phase of the drive current during high-speed drive or high load, resulting in step-out. There is a fear. There is also a problem that the power efficiency at the time of low speed driving or low load is lowered.

  Closed loop control has been proposed as another control method for the stepping motor. Closed loop control is a control method that estimates the rotor angle based on the induced voltage of the coil and controls the drive current based on the estimated angle. According to this closed loop control, the motor can be controlled with high accuracy and low power consumption during high-speed driving and high load.

  However, when the motor is driven at a low speed, the induced voltage generated is small, and it has been difficult to stably control the motor by closed loop control.

  The present invention has been made in view of the above-described problems, and an object of the present invention is to enable switching of a stepping motor control method.

  A motor control device according to an embodiment is a motor control device that controls a stepping motor based on a first current command value that depends on open-loop control and a second current command value that depends on closed-loop control, When the open loop control or the control method selection unit that selects the open loop control or the closed loop control based on the speed command value is selected, the first target value is output as the first current command value. And a command value output unit that outputs a second target value as the second current command value when the closed-loop control is selected.

  According to each embodiment of the present invention, the stepping motor control method can be switched.

The figure which shows an example of a structure of the motor drive system which concerns on 1st Embodiment. The figure which shows an example of a motor. The figure which shows an example of an angle command value calculation part. The figure which shows an example of an angle control part. The figure which shows an example of a d-axis current control part. The figure which shows an example of a q-axis current control part. The graph explaining operation | movement of a 1st vector rotation part. The graph explaining operation | movement of a 1st vector rotation part. The graph explaining operation | movement of a 2nd vector rotation part. The figure which shows an example of an angle estimation part. The state transition diagram explaining the selection method of the control method in 1st Embodiment. The flowchart which shows an example of operation | movement of the motor drive system in 1st Embodiment. The figure which shows the simulation result of the motor drive system which concerns on 1st Embodiment. The figure which shows the structure of the motor drive system which concerns on 1st Embodiment. The figure which shows an example of the angle control part which concerns on 2nd Embodiment. The figure which shows the structure of the motor drive system which concerns on 3rd Embodiment. The state transition diagram explaining the selection method of the control method in 3rd Embodiment. The flowchart which shows an example of operation | movement of the motor drive system in 3rd Embodiment. The figure which shows an example of a structure of the motor drive system which concerns on 4th Embodiment. The graph which shows the relationship between the voltage command value and rotation speed at the time of closed loop control. The graph which shows the relationship between the voltage command value and rotation speed at the time of closed loop control. The figure which shows an example of the 1st selection part which concerns on 4th Embodiment. The state transition diagram explaining the switching method of the field weakening control in 4th Embodiment. The figure which shows an example of the 1st selection part which concerns on 4th Embodiment.

  Hereinafter, embodiments of the present invention will be described with reference to the accompanying drawings. In addition, regarding the description of the specification and the drawings according to each embodiment, constituent elements having substantially the same functional configuration are denoted by the same reference numerals and overlapping description is omitted.

<First Embodiment>
The motor drive system according to the first embodiment will be described with reference to FIGS.

  FIG. 1 is a diagram illustrating an example of a configuration of a motor drive system according to the present embodiment. The motor drive system in FIG. 1 includes a motor 1, an inverter 2, a current detector 3, and a motor control device 100. The inverter 2, the current detector 3, and the motor control device 100 constitute a motor driving device that drives the motor 1.

  The motor 1 is a motor driven by two-phase alternating currents that are 90 degrees out of phase. Each phase of the motor 1 is referred to as an A phase and a B phase. Below, although the motor 1 shall be a two-phase stepping motor (STM), it is not restricted to this. FIG. 2 is a diagram illustrating an example of the motor 1. The motor 1 in FIG. 2 includes a rotor 11, an A-phase coil 12A, and a B-phase coil 12B.

  The rotor 11 is constituted by a permanent magnet arranged circumferentially or a magnetic body magnetized circumferentially. The rotor 11 has p pole pairs (a pair of S poles and N poles). In the example of FIG. 2, the number of pole pairs p is 1. The actual angle (position) of the rotor 11 is referred to as the actual angle θm.

  Hereinafter, an angle with one rotation of the rotor 11 as one cycle is referred to as a mechanical angle, and an angle with one rotation of the rotor 11 as a p cycle is referred to as an electrical angle. The electrical angle corresponds to p times the mechanical angle. Hereinafter, unless otherwise noted, the angle and speed of the rotor 11 are represented by electrical angles. The actual angle θm is an electrical angle.

  The A-phase coil 12A and the B-phase coil 12B are independent coils that are not connected to each other, and are arranged 90 degrees out of phase. Hereinafter, the voltages at the two terminals of the A-phase coil 12A are referred to as terminal voltages A + and A-, respectively. The voltages at the two terminals of the B-phase coil 12B are referred to as terminal voltages B + and B-, respectively. The rotor 11 rotates by passing alternating currents having a phase difference of 90 degrees through the A-phase coil 12A and the B-phase coil 12B.

  The inverter 2 is a circuit that supplies a drive current I to the motor 1 in accordance with a voltage command value input from the motor control device 100. The inverter 2 includes a PWM (Pulse Width Modulation) circuit (control signal generation unit), a driver circuit, and the like. The PWM circuit may be provided in the motor control device 100.

  The inverter 2 converts the a-axis voltage command value Va input from the motor control device 100 into terminal voltages A + and A− and applies them to the terminals of the A-phase coil 12A. Specifically, the PWM circuit generates a control signal (pulse signal) corresponding to the a-axis voltage command value Va and inputs it to the driver circuit. The driver circuit applies terminal voltages A + and A− according to the control signal to the terminals of the A-phase coil 12A, respectively. The difference between the terminal voltages A + and A− is the inter-terminal voltage of the A-phase coil 12A. By applying this inter-terminal voltage, the A-phase drive current Ia corresponding to the a-axis voltage command value Va flows through the A-phase coil 12A. The a-axis voltage command value Va is an a-axis component of the drive voltage and corresponds to the drive voltage value of the A-phase coil 12A. The inter-terminal voltage that can be applied to the A-phase coil 12A by the inverter 2 is equal to or lower than the power supply voltage Vcc of the inverter 2.

  Further, the inverter 2 converts the b-axis voltage command value Vb input from the motor control device 100 into terminal voltages B + and B−, and applies them to the terminals of the B-phase coil 12B. Specifically, the PWM circuit generates a control signal (pulse signal) corresponding to the b-axis voltage command value Vb and inputs it to the driver circuit. The driver circuit applies terminal voltages B + and B− according to the control signal to the terminals of the B-phase coil 12B. The difference between the terminal voltages B + and B− is the inter-terminal voltage of the B-phase coil 12B. By applying this inter-terminal voltage, a B-phase drive current Ib corresponding to the b-axis voltage command value Vb flows through the B-phase coil 12B. The b-axis voltage command value Vb is a b-axis component of the drive voltage and corresponds to the drive voltage of the B-phase coil 12B. The inter-terminal voltage that can be applied by inverter 2 to phase B coil 12B is equal to or lower than power supply voltage Vcc of inverter 2.

  The current detector 3 is a current sensor that detects a current value and outputs it as a digital value. The current detector 3 includes a shunt resistor, a differential amplifier, an AD (Analog to Digital) converter, and the like.

  The current detector 3 detects and outputs an a-axis current value ia that is a current value of the A-phase drive current Ia. The a-axis current value ia is an a-axis component of the detected value of the drive current I. The current detector 3 detects and outputs a b-axis current value ib that is a current value of the B-phase drive current Ib. The b-axis current value ib is a b-axis component of the detected value of the drive current I. The a and b axes correspond to the A and B phases, respectively. The a and b axes will be described later in detail. The a-axis current value ia and the b-axis current value ib output from the current detector 3 are input to the motor control device 100.

  The motor control device 100 includes a processor and a memory. The processor is, for example, a CPU (Central Processing Unit), an MPU (Micro Processing Unit), a DSP (Digital Signal Processor), or the like. The processor may be an application specific integrated circuit (ASIC), a field programmable gate array (FPGA), a programmable logic device (PLD), or the like. The processor executes a program stored in the memory.

  The memory is DRAM (Dynamic Random Access Memory), SRAM (Static RAM), MRAM (Magnetic RAM), flash memory, or the like. The memory stores a program executed by the processor.

  The motor control device 100 is realized by, for example, an IC chip that includes one or more processors and one or more memories. The IC chip may include at least one of the inverter 2 and the current detector 3.

  The motor control device 100 receives the reference pulse refclk. The reference pulse refclk is a pulse for controlling the motor 1 and is the same as the pulse input to the conventional control device. The motor control device 100 controls the motor 1 based on the reference pulse refclk by two types of control methods, closed loop control and open loop control.

  The closed loop control is a control method for controlling the motor 1 based on a current command value calculated based on a feedback control result (angle and speed of the rotor 11). As in the present embodiment, when the motor 1 is sensorlessly controlled, the angle and speed of the motor 1 cannot be directly acquired. For this reason, in this embodiment, the estimated values of the angle and speed of the motor 1 are fed back. According to the closed loop control, the motor 1 can be controlled with high accuracy.

  The open loop control is a control method for controlling the motor 1 with a preset current command value. In the open loop control, unlike the closed loop control, the angle and speed of the motor 1 as a control result are not fed back. According to the open loop control, the motor 1 can be controlled by a simple method.

  As shown in FIG. 1, the motor control device 100 includes an angle command value calculation unit 101, an angle control unit 102, a first selection unit 103, a second selection unit 104, and a d-axis current control unit as functional configurations. 105 and a q-axis current control unit 106. In addition, the motor control device 100 includes a first vector rotation unit 107, a second vector rotation unit 108, a third selection unit 109, an angle estimation unit 110, and a control method selection unit 111. These functional configurations are realized by a processor executing a program.

  The angle command value calculation unit 101 receives a reference pulse refclk from the outside. The angle command value calculation unit 101 calculates and outputs an angle command value θtgt by integrating the reference pulse refclk. The angle command value θtgt is a target value for the angle of the rotor 11. The angle command value θtgt output from the angle command value calculation unit 101 is input to the angle control unit 102 and the third selection unit 109.

  Further, the angle command value calculation unit 101 calculates and outputs a speed command value ωtgt by measuring the frequency of the reference pulse refclk. The speed command value ωtgt is a target value for the speed (angular speed) of the rotor 11. The speed command value ωtgt output from the angle command value calculation unit 101 is input to the control method selection unit 111.

  FIG. 3 is a diagram illustrating an example of the angle command value calculation unit 101. The angle command value calculation unit 101 in FIG. 3 includes a counter 1011, a period measurement unit 1012, and an inverse number calculation unit 1013.

  The counter 1011 receives the reference pulse refclk. The counter 1011 counts the reference pulse refclk and outputs the count value as an angle command value θtgt. The count of the reference pulse refclk corresponds to the integration of the reference pulse refclk.

  The period measurement unit 1012 receives the reference pulse refclk. The period measurement unit 1012 measures and outputs the period tω of the reference pulse refclk. The period tω can be measured by measuring the edge interval of the reference pulse refclk with a fixed frequency clock sufficiently faster than the reference pulse refclk.

  The reciprocal calculation unit 1013 receives the period tω, calculates the reciprocal of the period tω, and outputs a value obtained by multiplying the obtained value by the coefficient K as a speed command value ωtgt. The coefficient K is a coefficient for converting the reciprocal of the period tω into a value corresponding to the frequency of the reference pulse refclk.

  The angle command value calculation unit 101 may calculate the speed command value ωtgt by differentiating the angle command value θtgt. In this case, the angle command value calculation unit 101 can calculate the speed command value ωtgt by calculating the difference between the angle command values θtgt at regular intervals. The difference in the angle command value θtgt corresponds to the speed command value ωtgt.

  The angle control unit 102 receives the angle command value θtgt, the estimated angle value θest, and the speed command value ωest. The estimated angle value θest is an estimated value of the angle of the rotor 11. The speed command value ωest is an estimated value of the speed (angular speed) of the rotor 11. The angle estimated value θest and the speed command value ωest will be described later in detail.

  The angle control unit 102 calculates and outputs a current command value itgt so that the estimated angle value θest matches the angle command value θtgt. The current command value itgt (second target value) is a target value of the q-axis current command value iqt (second current command value) when the motor 1 is closed-loop controlled. The q-axis current command value iqt is a q-axis component (torque component) of the drive current I.

  FIG. 4 is a diagram illustrating an example of the angle control unit 102. 4 includes a comparison unit 1021, an amplification unit 1022, a comparison unit 1023, amplification units 1024 and 1025, an integration unit 1026, and an addition unit 1027.

  The comparison unit 1021 receives the angle command value θtgt and the estimated angle value θest. The angle command value θtgt corresponds to a target value for angle control. Further, the estimated angle value θest corresponds to a detected angle value of angle control. The comparison unit 1021 calculates and outputs the difference between the angle command value θtgt and the angle command value θest. The output value of the comparison unit 1021 is input to the amplification unit 1022.

  The amplification unit 1022 amplifies the output value of the comparison unit 1021 with a predetermined gain G1 and outputs the amplified value. The output value of the amplifying unit 1022 corresponds to an operation amount for angle control. Hereinafter, the output value of the amplifying unit 1022 is referred to as an operation amount C (= G1 × (θtgt−θest)). The operation amount C is input to the comparison unit 1023.

  The comparison unit 1023 receives the operation amount C and the speed estimated value ωest. The comparison unit 1023 calculates and outputs the difference between the operation amount C and the estimated speed value ωest. The speed estimated value ωest corresponds to a detected angle value of speed control. The operation amount C corresponds to a target value for speed control. The output value of the comparison unit 1023 is input to the amplification unit 1024.

  The amplification unit 1024 amplifies the output value of the comparison unit 1023 with a predetermined gain G2 and outputs the amplified value. The output value of the amplifying unit 1024 is input to the amplifying unit 1025 and the adding unit 1027.

  The amplifying unit 1025 amplifies the output value of the amplifying unit 1024 with a predetermined gain G3 and outputs it. The output value of the amplification unit 1025 is input to the integration unit 1026.

  The integrating unit 1026 integrates and outputs the output value of the amplifying unit 1025. The output value of the integrating unit 1026 is input to the adding unit 1055.

  Adder 1027 adds the output value of amplifier 1024 and the output value of integrator 1026, and outputs the result as current command value itgt. The current command value itgt corresponds to a speed control operation amount included in the angle control. The current command value itgt is expressed by the following equation.

  According to the angle control unit 102 of FIG. 4, the speed control loop (the comparison unit 1023 and the amplification unit 1024) is included inside the angle control loop. Since this speed control loop becomes a phase advance element, the phase margin of the angle control unit 102 can be secured by the configuration of FIG. Further, since the angle control unit 102 in FIG. 4 does not include a differential element, the angle control unit 102 is less susceptible to noise and can stably output the current command value itgt.

  The angle control unit 102 may be configured not to include a speed control loop. In this case, the comparison unit 1023 and the amplification unit 1024 may be removed from the configuration of FIG. 4 and the output value of the amplification unit 1022 may be input to the amplification unit 1025 and the addition unit 1027.

  The first selection unit 103 is an example of a command value output unit. The first selection unit 103 receives the current value 0 and the current value iref. The current value iref (first target value) is a value set in advance as a target value for the d-axis current command value idt (first current command value) in the open loop control. The d-axis current command value idt is a d-axis component (magnetic flux component) of the drive current I.

  The first selection unit 103 receives the selection signal sel. The selection signal sel is a 0 or 1 signal indicating a control method selection result by the control method selection unit 111. Hereinafter, the control method selection unit 111 outputs 0 as the selection signal sel when the open loop control is selected, and outputs 1 as the selection signal sel when the closed loop control is selected. Details of the selection signal sel will be described later.

  The first selection unit 103 selects and outputs the current value 0 or the current value iref as the d-axis current command value idt according to the selection signal sel. The d-axis current command value idt output from the first selection unit 103 is input to the d-axis current control unit 105.

  When the selection signal sel is 0 (when the control method is open loop control), the first selection unit 103 selects the current value iref as the d-axis current command value idt, and when the selection signal sel is 1 (the control method is In the case of closed loop control), the current value 0 is selected as the d-axis current command value idt.

  The second selection unit 104 is an example of a command value output unit. The second selection unit 104 receives the current command value itgt and the current value 0. The second selection unit 104 receives the selection signal sel. The second selection unit 104 selects and outputs a current value corresponding to the selection signal sel as the q-axis current command value iqt. The q-axis current command value iqt output from the second selection unit 104 is input to the q-axis current control unit 106.

  When the selection signal sel is 0 (when the control method is open loop control), the second selection unit 104 selects the current value 0 as the q-axis current command value iqt, and when the selection signal sel is 1 (the control method is In the case of closed loop control), the current command value itgt is selected as the q-axis current command value iqt.

  The d-axis current control unit 105 receives the d-axis current value id and the d-axis current command value idt. The d-axis current value id is a d-axis component of the detected value of the drive current I. The d-axis current control unit 105 uses the PI (Proportional Integral) control or the like to control the d-axis voltage command value Vd so that the d-axis current value id matches the d-axis current command value idt. D-axis voltage command value Vd is output. The d-axis voltage command value Vd is a d-axis component of the voltage command value. The d-axis voltage command value Vd output from the d-axis current control unit 105 is input to the first vector rotation unit 107.

  FIG. 5 is a diagram illustrating an example of the d-axis current control unit 105. The d-axis current control unit 105 of FIG. 5 includes a comparison unit 1051, amplification units 1052 and 1053, an integration unit 1054, and an addition unit 1055.

  The comparison unit 1051 receives the d-axis current command value idt and the d-axis current value id. The comparison unit 1051 calculates and outputs the difference between the d-axis current command value idt and the d-axis current value id. The output value of the comparison unit 1051 is input to the amplification unit 1052.

  The amplification unit 1052 amplifies the output value of the comparison unit 1051 with a predetermined gain G4 and outputs the amplified value. The output value of the amplification unit 1052 is input to the amplification unit 1053 and the addition unit 1055.

  The amplifying unit 1053 amplifies the output value of the amplifying unit 1052 with a predetermined gain G5 and outputs the result. The output value of the amplifying unit 1053 is input to the integrating unit 1054.

  The integrator 1054 integrates and outputs the output value of the amplifier 1053. The output value of the integrating unit 1054 is input to the adding unit 1055.

  Adder 1055 adds the output value of amplifier 1052 and the output value of integrator 1054, and outputs the result as d-axis voltage command value Vd. This d-axis voltage command value Vd is expressed by the following equation.

  The q-axis current control unit 106 receives the q-axis current value iq and the q-axis current command value iqt. The q-axis current value iq is a q-axis component of the detected value of the drive current I. The q-axis current control unit 106 uses the PI control or the like to control the q-axis voltage command value Vq so that the q-axis current value iq matches the q-axis current command value iqt. Command value Vq is output. The q-axis voltage command value Vq is a q-axis component of the voltage command value. The q-axis voltage command value Vq output from the q-axis current control unit 106 is input to the first vector rotation unit 107.

  FIG. 6 is a diagram illustrating an example of the q-axis current control unit 106. The q-axis current control unit 106 of FIG. 6 includes a comparison unit 1061, amplification units 1062 and 1063, an integration unit 1064, and an addition unit 1065. These components are the same as the components of the d-axis current control unit 105 in FIG. If the gains of the amplifying units 1062 and 1063 are G6 and G7, respectively, the q-axis voltage command value Vq is expressed by the following equation.

  The first vector rotation unit 107 receives the d-axis voltage command value Vd, the q-axis voltage command value Vq, and the drive angle drv. The drive angle drv is a phase angle for coordinate conversion from the dq coordinate system to the AB coordinate system. The dq coordinate system is a rotating coordinate system including a d-axis corresponding to the magnetic flux component and a q-axis corresponding to the torque component. The AB coordinate system is a fixed coordinate system including an a axis corresponding to the A phase and a b axis corresponding to the B phase. The angle of the rotor 11 corresponds to the angle of the d axis with respect to the a axis. The drive angle drv is input from the third selection unit 109. The drive angle drv will be described later in detail.

  The first vector rotation unit 107 converts the dq coordinate system into the AB coordinate system, thereby converting the a-axis voltage command value Va and the b-axis voltage command value Vb from the d-axis voltage command value Vd and the q-axis voltage command value Vq. Is calculated and output. The a-axis voltage command value Va and the b-axis voltage command value Vb output from the first vector rotation unit 107 are input to the inverter 2 and the angle estimation unit 110.

  Specifically, the first vector rotation unit 107 rotates the vector voltage composed of the d-axis voltage command value Vd and the q-axis voltage command value Vq by the drive angle drv, so that the a-axis voltage command value Va and the b-axis A voltage command value Vb is calculated. The a-axis voltage command value Va and the b-axis voltage command value Vb are expressed by the following equations.

  Expressions (4) and (5) are expressed by the following determinants.

  7 and 8 are graphs for explaining the operation of the first vector rotation unit 107. FIG. Hereinafter, a case of closed loop control will be described as an example. 7 and 8, the broken line indicates the d-axis voltage command value Vd, the alternate long and short dash line indicates the q-axis voltage command value Vq, the dotted line indicates the a-axis voltage command value Va, and the solid line indicates the b-axis voltage command value Vb. The vertical axis represents the voltage level [a. u. The horizontal axis represents the actual angle θm [degree] of the rotor 11.

  In the example of FIG. 7, Vd = 0 and Vq = 1. At this time, the a-axis voltage command value Va (= −sin (θm)) and the b-axis voltage command value Vb (= cos (θm)) have an amplitude of 1 and the phase matches the actual angle θm. On the other hand, in the example of FIG. 8, Vd = −0.342 and Vq = 0.940. At this time, the a-axis voltage command value Va and the b-axis voltage command value Vb have an amplitude of 1, and the phase advances about 20 degrees with respect to the actual angle θm.

  Thus, the phases of the a-axis voltage command value Va and the b-axis voltage command value Vb change according to the d-axis voltage command value Vd and the q-axis voltage command value Vq. In the present embodiment, when the driving speed of the motor 1 increases and the phase delay between the speed a-axis current value ia and the b-axis current value ib increases, the phase of the a-axis voltage command value Va and the b-axis voltage command value Vb advances. As described above, the d-axis voltage command value Vd and the q-axis voltage command value Vq are controlled. For this reason, a decrease in power efficiency accompanying an increase in drive speed is suppressed.

  The second vector rotation unit 108 receives the a-axis current value ia, the b-axis current value ib, and the drive angle drv. The second vector rotation unit 108 converts the AB coordinate system into the dq coordinate system, thereby calculating the d-axis current value id and the q-axis current value iq from the a-axis current value ia and the b-axis current value ib. Output. The d-axis current value id output from the second vector rotation unit 108 is input to the d-axis current control unit 105. Further, the q-axis current value iq output from the second vector rotation unit 108 is input to the q-axis current control unit 106.

  Specifically, the second vector rotation unit 108 rotates the vector current composed of the a-axis current value ia and the b-axis current value ib by the drive angle drv, so that the d-axis current value id and the q-axis current value iq Is calculated. The d-axis current value id and the q-axis current value iq are expressed by the following equations.

  Expressions (7) and (8) are expressed by the following determinants.

  FIG. 9 is a graph for explaining the operation of the second vector rotation unit 108. Hereinafter, a case of closed loop control will be described as an example. In FIG. 9, the broken line indicates the a-axis current value ia, the alternate long and short dash line indicates the b-axis current value ib, the dotted line indicates the d-axis current value id, and the solid line indicates the q-axis current value iq. The vertical axis represents the current level [a. u. The horizontal axis represents the actual angle θm [degree] of the rotor 11.

  In the example of FIG. 9, the phase of the a-axis current value ia and the b-axis current value ib is delayed by about 30 degrees with respect to the actual angle θm. At this time, id = 0.5 and iq = 0.866. On the other hand, when the phase of the a-axis current value ia and the b-axis current value ib matches the actual angle θm, id = 0 and iq = 1.

  Thus, the d-axis current value id and the q-axis current value iq change according to the phase of the a-axis current value ia and the b-axis current value ib. In other words, by controlling the current so that the d-axis current value id and the q-axis current value iq become predetermined values, the phases of the a-axis current value ia and the b-axis current value ib are controlled to desired phases. be able to.

  For example, by controlling the current so that id = 0, the phases of the a-axis current value ia and the b-axis current value ib can be matched with the actual angle θm. In addition, by controlling the current so that id becomes a predetermined value other than 0, the phase of the a-axis current value ia and the b-axis current value ib can be shifted from the actual angle θm by a predetermined value to improve the reluctance torque. It is.

  The third selection unit 109 receives the angle command value θtgt, the angle estimation value θest, and the selection signal sel. The third selection unit 109 selects and outputs the angle command value θtgt or the estimated angle value θest as the drive angle θdrv according to the selection signal sel. Hereinafter, the third selection unit 109 selects the angle command value θtgt as the drive angle θdrv when the selection signal sel is 0, and selects the estimated angle value θest as the drive angle θdrv when the selection signal sel is 1. And

  The angle estimation unit 110 receives the a-axis voltage command value Va, the b-axis voltage command value Vb, the a-axis current value ia, and the b-axis current value ib. The angle estimation unit 110 estimates the angle and speed of the rotor 11 based on these. That is, the angle estimation unit 110 calculates and outputs the angle estimation value θest and the speed estimation value ωest.

  FIG. 10 is a diagram illustrating an example of the angle estimation unit 110. The angle estimation unit 110 in FIG. 10 estimates the actual angle θm using the induced voltage generated by the rotation of the rotor 11. The angle estimation unit 110 in FIG. 10 includes a third vector rotation unit 1101, a fourth vector rotation unit 1102, an estimation error calculation unit 1103, amplification units 1104 and 1105, an integration unit 1106, an addition unit 1107, an integration unit Unit 1108.

  The third vector rotation unit 1101 receives the a-axis voltage command value Va and the b-axis voltage command value Vb. The third vector rotation unit 1101 rotates the vector voltage composed of the a-axis voltage command value Va and the b-axis voltage command value Vb by the estimated angle value θest, so that the d-axis voltage command value Vde and the q-axis voltage command value Vqe. Is calculated and output. The d-axis voltage command value Vde and the q-axis voltage command value Vqe output from the third vector rotation unit 1101 are input to the estimation error calculation unit 1103. The d-axis voltage command value Vde and the q-axis voltage command value Vqe are expressed by the following equations.

  Expressions (10) and (11) are expressed by the following determinants.

  The fourth vector rotation unit 1102 receives the a-axis current value ia and the b-axis current value ib. The fourth vector rotation unit 1102 calculates the d-axis current value ide and the q-axis current value iqe by rotating the vector current composed of the a-axis current value ia and the b-axis current value ib by the estimated angle value θest, Output. The d-axis current value ide and the q-axis current value iqe output from the fourth vector rotation unit 1102 are input to the estimation error calculation unit 1103. The d-axis current value ide and the q-axis current value iqe are expressed by the following equations.

  Expressions (13) and (14) are expressed by the following determinants.

  The estimation error calculation unit 1103 receives the d-axis voltage command value Vde and the q-axis voltage command value Vqe, and the d-axis current value ide and the q-axis current value iqe. Based on these, the estimation error calculation unit 1103 calculates and outputs an estimation error Xerr. The estimation error Xerr is a parameter proportional to an angle error θerr (= θm−θest) that is an error between the actual angle θm of the rotor 11 and the estimated angle value θest. The estimation error Xerr is calculated by the following equation, for example.

  In Equation (16), Ke is a counter electromotive voltage constant (ratio of counter electromotive voltage generated in the coil in proportion to the speed of the rotor 11), R is coil resistance, and L is coil inductance. The values of Ke, R, and L are determined according to the specifications of the motor 1. Further, the estimated speed value ωest is used as the speed ω of the rotor 11.

  The estimation error Xerr calculated by the equation (16) is a sine wave whose phase is the angle error θerr (Xerr = sin (θerr)). Therefore, when the angle error θerr is sufficiently small, the estimation error Xerr is considered to be proportional to the angle error θerr. The estimation error Xerr output from the estimation error calculation unit 1103 is input to the amplification unit 1104.

  The amplifying unit 1104 amplifies the estimation error Xerr with a predetermined gain G8 and outputs it. The output value of the amplifying unit 1104 is input to the amplifying unit 1105 and the adding unit 1107.

  The amplifying unit 1105 amplifies the output value of the amplifying unit 1104 with a predetermined gain G9 and outputs it. The output value of the amplifying unit 1105 is input to the integrating unit 1106.

  The integrating unit 1106 integrates and outputs the output value of the amplifying unit 1105. The output value of the integrating unit 1106 is input to the adding unit 1107.

  The adding unit 1107 adds the output value of the amplifying unit 1104 and the output value of the integrating unit 1106, and outputs the result as a speed estimated value ωest. The estimated speed value ωest output from the adder 1107 is input to the angle controller 102, the integrator 1108, and the estimated error calculator 1103. The estimation error calculation unit 1103 calculates the estimation error Xerr using the speed estimation value ωest.

  The integration unit 1108 integrates the estimated speed value ωest and outputs it as an estimated angle value θest. The estimated angle value θest output from the integration unit 1108 is input to the angle control unit 102, the third selection unit 109, the third vector rotation unit 1101, and the fourth vector rotation unit 1102. Using the estimated angle value θest, the third vector rotation unit 1101 calculates the d-axis voltage command value Vde and the q-axis voltage command value Vqe, and the fourth vector rotation unit 1102 calculates the d-axis current value ide and the q-axis The current value iq is calculated.

  In this way, the estimated angle value θest is fed back to the third vector rotation unit 1101 and the fourth vector rotation unit 1102. Thus, a phase locked loop (PLL) in which the angle error θerr (estimated error Xerr) follows 0, that is, the estimated angle value θerr follows the actual angle θm is configured. As a result, the estimated angle value θest can be estimated with high accuracy.

  Note that the estimation error calculation unit 1103 may use the speed command value ωtgt as the speed ω of the rotor 11 in order to perform the calculation of the equation (16), or the estimated speed value ωest and the speed command value ωtgt. Both may be used. It is preferable that the estimation error calculation unit 1103 uses the speed command value ωtgt when the motor 1 is driven at a low speed (when starting or stopping), and uses the speed estimation value ωest during other periods. This is because when the motor 1 is driven at a low speed, the induced voltage is small, so that the estimation error Xerr becomes unstable, and as a result, the estimation accuracy of the speed estimation value ωest is lowered. During such low-speed driving, the estimation accuracy of the estimated speed value ωest can be improved by using the speed command value ωtgt.

  In the present embodiment, the angle estimation unit 110 may operate during open loop control. Thereby, the power consumption of the motor control apparatus 100 during the open loop control can be reduced. Further, the angle estimation unit 110 may operate during the open loop control. Thereby, even during open loop control, the estimated speed value ωest of the rotor 11 can be used for detecting an abnormal operation of the motor 1 or the like.

  The control method selection unit 111 receives the speed command value ωtgt. The control method selection unit 111 selects a control method based on the speed command value ωtgt and outputs a selection signal sel indicating the selection result. As described above, the control method selection unit 111 outputs 0 as the selection signal sel when the open loop control is selected, and outputs 1 as the selection signal sel when the closed loop control is selected.

  FIG. 11 is a state transition diagram illustrating a control method selection method according to this embodiment. Each node in FIG. 11 indicates a selection state of the control method by the control method selection unit 111 of the motor 1. Hereinafter, the selected states of the nodes N1, N2, and N3 are referred to as states OP1 (OPEN1), OP2 (OPEN2), and CL (CLOSED), respectively.

  The state OP1 is a state in which the control method selection unit 111 selects open loop control and outputs 0 as the selection signal sel. In the state OP1, when the speed command value ωtgt is equal to or greater than a predetermined speed threshold value ω1, the state OP1 transitions to the state OP2.

  The state OP2 is a state in which the control method selection unit 111 selects open loop control and outputs 0 as the selection signal sel. When the speed command value ωtgt becomes less than the predetermined speed threshold value ω2 in the state OP2, the state OP2 transitions to the state OP1. Further, in the state OP2, when the duration t of the state OP2 is equal to or longer than the predetermined period T1, the state OP2 transitions to the state CL.

  The state CL is a state in which the control method selection unit 111 selects closed loop control and outputs 1 as the selection signal sel. In the state CL, when the speed command value ωtgt becomes less than the predetermined speed threshold value ω2, the state CL transitions to the state OP1.

  Note that the speed threshold values ω1 and ω2 are threshold values of the speed command value ωtgt set in advance. The speed threshold ω <b> 1 is set by experiments, simulations, or the like as a lower limit value of the speed command value ωtgt that can stably perform closed-loop control when the motor 1 is accelerated. The speed threshold ω <b> 2 is set by experiment, simulation, or the like as a lower limit value of the speed command value ωtgt that can stably perform the closed loop control when the motor 1 is decelerated. Stable closed-loop control is closed-loop control in which torque vibration is small and synchronization is established.

  FIG. 12 is a flowchart showing an example of the operation of the motor drive system in the present embodiment. FIG. 12 shows the operation at each control timing.

  When the control timing comes, first, the angle command value calculation unit 101 measures the frequency of the reference pulse refclk, and calculates the speed command value ωtgt. The speed command value ωtgt is input to the control method selection unit 111. Further, the angle command value calculation unit 101 integrates the reference pulses refclk from the previous control timing to the current control timing, and calculates the angle command value θtgt (step S101). Further, the angle command value θtgt is input to the angle control unit 102 and the third selection unit 109.

  Next, the control method selection unit 111 determines whether the speed command value ωtgt is less than the speed threshold value ω2 (step S102). When the speed command value ωtgt is less than the speed threshold value ω2 (YES in step S102), the control method selection unit 111 selects open loop control as the control method and outputs 0 as the selection signal sel (step S103). At this time, the selected state corresponds to the state OP1. The selection signal sel is input to the first selection unit 103, the second selection unit 104, and the third selection unit 109.

  When 0 is input as the selection signal sel, the first selection unit 103 outputs the current value iref as the d-axis current command value idt, and the second selection unit 104 outputs 0 as the q-axis current command value iqt. When 0 is input as the selection signal sel, the third selection unit 109 outputs the angle command value θtgt as the drive angle θdrv (step S104). The d-axis current command value idt is input to the d-axis current control unit 105. The q-axis current command value iqt is input to the q-axis current control unit 106. The drive angle θdrv is input to the first vector rotation unit 107 and the second vector rotation unit 108.

  Note that when the open loop control is selected, the angle control unit 102 may or may not operate. By inputting the selection signal sel to the angle control unit 102, the operation of the angle control unit 102 can be stopped when the open loop control is selected.

  Subsequently, the d-axis current control unit 105 calculates the d-axis voltage command value Vd based on the d-axis current command value idt (= iref) and the d-axis current value id input at the previous control timing. To do. Further, the q-axis current control unit 106 calculates the q-axis voltage command value Vq based on the q-axis current command value iqt (= 0) and the q-axis current value iq input at the previous control timing. (Step S105). The d-axis voltage command value Vd and the q-axis voltage command value Vq are input to the first vector rotation unit 107.

  The first vector rotation unit 107 rotates the vector voltage composed of the d-axis voltage command value Vd and the q-axis voltage command value Vq by the drive angle θdrv (= θtgt), thereby causing the a-axis voltage command value Va and the b-axis voltage to be rotated. Command value Vb is calculated (step S106). The a-axis voltage command value Va and the b-axis voltage command value Vb are input to the angle estimation unit 110 and the inverter 2.

  Thereafter, the inverter 2 supplies the A-phase drive current Ia corresponding to the a-axis voltage command value Va to the A-phase coil 12A, and supplies the B-phase drive current Ib corresponding to the b-axis voltage command value Vb to the B-phase coil 12B. (Step S107). Thereby, the motor 1 is driven.

  When the A-phase drive current Ia and the B-phase drive current Ib are supplied to the motor 1, the current detector 3 detects the a-axis current value ia and the b-axis current value ib, respectively (step S108). The a-axis current value ia and the b-axis current value ib are input to the second vector rotation unit 108 and the angle estimation unit 110.

  The second vector rotation unit 108 rotates the vector current composed of the a-axis current value ia and the b-axis current value ib by the drive angle θdrv (= θtgt), thereby changing the d-axis current value Id and the q-axis current value iq. Calculate (step S109). The d-axis current value Id and the q-axis current value iq are input to the d-axis current control unit 105 and the q-axis current control unit 106, respectively.

  Further, the angle estimation unit 110 calculates the angle estimation value θest and the speed estimation value ωest based on the a-axis voltage command value Va and the b-axis voltage command value Vb, and the a-axis current value ia and the b-axis current value ib. Calculate (step S110).

  When the open loop control is selected, the angle estimation unit 110 may or may not operate. By inputting the selection signal sel to the angle estimation unit 110, the operation of the angle estimation unit 110 can be stopped when the open loop control is selected.

  When the driving of the motor 1 is finished (YES in step S111), the process is finished. On the other hand, when the drive of the motor 1 is continued (NO in step S111), the process returns to step S101.

  On the other hand, when the speed command value ωtgt is equal to or higher than the speed threshold value ω2 (NO in step S102), the control method selection unit 111 determines whether the speed command value ωtgt is equal to or higher than the speed threshold value ω1 (step S112).

  When the speed command value ωtgt is less than the speed threshold value ω1 (NO in step S112), the processes in steps S102 to S111 described above are executed.

  On the other hand, when the speed command value ωtgt is equal to or greater than the speed threshold value ω1 (YES in step S112), the control method selection unit 111 determines whether the duration t where the speed command value ωtgt is equal to or greater than the speed threshold value ω1 is equal to or greater than the predetermined period T1. (Step S113).

  If the duration t is less than the predetermined period T1 (NO in step S113), the control method selection unit 111 selects open loop control as the control method and outputs 0 as the selection signal sel (step S114). The selection signal sel is input to the first selection unit 103, the second selection unit 104, and the third selection unit 109. At this time, the selected state corresponds to the state OP2. Thereafter, the processes in steps S104 to S111 described above are executed.

  On the other hand, when the duration t is equal to or longer than the predetermined period T1 (YES in step S113), the control method selection unit 111 selects closed loop control as the control method and outputs 1 as the selection signal sel (step S115). ). The selection signal sel is input to the first selection unit 103, the second selection unit 104, and the third selection unit 109. At this time, the selected state corresponds to the state CL.

  When the closed loop control is selected, the angle control unit 102 calculates the current command value itgt based on the angle command value θtgt, the estimated angle value θest and the estimated speed value ωest input at the previous control timing. (Step S116). The current command value itgt is input to the second selection unit 104.

  When 1 is input as the selection signal sel, the first selection unit 103 outputs the current value 0 as the d-axis current command value idt, and the second selection unit 104 sets the current command value itgt as the q-axis current command value iqt. Output. When 1 is input as the selection signal sel, the third selection unit 109 outputs the angle command value θest as the drive angle θdrv (step S117). The d-axis current command value idt is input to the d-axis current control unit 105. The q-axis current command value iqt is input to the q-axis current control unit 106. The drive angle θdrv is input to the first vector rotation unit 107 and the second vector rotation unit 108. Thereafter, the processes in steps S105 to S111 described above are executed. In the case of closed loop control, the estimated angle value θest is used as the drive angle θdrv.

  In summary, according to the present embodiment, when sel = 0, idt = iref, iqt = 0, and θdrv = θtgt. Therefore, when sel = 0, the a-axis current value ia and the b-axis current value ib are controlled as follows.

  As can be seen from the equations (17) and (18), the a-axis current value ia and the b-axis current value ib are controlled such that the phase is synchronized with the angle command value θtgt and the amplitude is a constant value iref. That is, open loop control is realized in which the a-axis current value ia and the b-axis current value ib are controlled regardless of the angle (position) of the rotor 11.

  On the other hand, when sel = 1, idt = 0, iqt = itgt, and θdrv = θest. Therefore, when sel = 1, the a-axis current value ia and the b-axis current value ib are controlled as follows.

  As can be seen from the equations (19) and (20), the a-axis current value ia and the b-axis current value ib have a current command value itgt whose phase is synchronized with the estimated angle value θest and whose amplitude corresponds to the estimated speed value ωest. It is controlled to become. That is, closed-loop control is realized in which the a-axis current value ia and the b-axis current value ib are controlled according to the estimated value of the angle (position) of the rotor 11.

  As described above, according to the present embodiment, the motor 1 is controlled while switching between open-loop control and closed-loop control. The motor 1 is controlled by open loop control during low speed driving (in the state OP1, OP2), and controlled by closed loop control during high speed driving (in the state CL).

  By performing open loop control during low speed driving, the driving of the motor 1 can be controlled with high accuracy. That is, the rotor 11 can be rotated following the angle command value θtgt. Further, the angle (position) of the rotor 11 when the motor 1 is stopped can be fixed to a predetermined angle.

  Further, by performing the closed loop control during high speed driving, the driving of the motor 1 can be accurately controlled. That is, the rotor 11 can be rotated following the estimated angle value θest.

  Further, according to the closed loop control, the current command value itgt, which is the amplitude of the a-axis current value ia and the b-axis current value ib, can be set to a value corresponding to the estimated speed value ωest. As a result, the current command value itgt can be controlled to a value commensurate with the load, speed, disturbance, and the like. That is, when the load or the like is small, the current command value itgt can be reduced, and when the load or the like is large, the current command value itgt can be reduced. As a result, the power consumption of the motor 1 can be reduced and the power efficiency can be improved.

  Further, since the current command value itgt can take a negative value, the torque in the direction opposite to the rotation direction of the rotor 11 can be generated in the rotor 11. For this reason, even when it is required to generate torque in the direction opposite to the traveling direction, such as when the electric vehicle equipped with the motor 1 is going down a hill, the desired torque is generated according to this embodiment. Can be made.

  Moreover, since the motor control apparatus 100 according to the present embodiment controls the motor 1 based on the reference pulse refclk, like the conventional stepping motor control apparatus, the motor control apparatus 100 is compatible with the conventional control apparatus. As a result, it is not necessary to change the design of the host system associated with the introduction of the motor control device 100, so that the cost for introduction can be reduced and the delivery time can be shortened.

  Further, according to the present embodiment, the A-phase drive current Ia and the B-phase drive current Ib, which are alternating currents, are a DC (low frequency) d-axis voltage command value Vd and a q-axis voltage command value Vq (d-axis current command). It is controlled by the value idt and the q-axis current command value iqt). Therefore, the control band can be lowered as compared with the case where the a-axis current value ia and the b-axis current value ib are directly controlled. As a result, a motor drive system can be configured by an inexpensive motor control device 100 having a low control band.

  Further, according to the present embodiment, it is possible to execute open-loop control and closed-loop control with the same configuration simply by switching the input of each configuration according to the selection signal sel. Therefore, the configuration of the motor control device 100 can be simplified and the motor control device 100 can be configured at low cost.

  Further, according to the present embodiment, since the phase of the a-axis current value ia and the b-axis current value ib can be controlled by the d-axis voltage command value Vd and the q-axis voltage command value Vq, the A-phase drive current Ia and the B-phase The voltage and current of the drive current Ib can be smoothly connected before and after the control method is switched. Thereby, the step-out of the motor 1 resulting from the switching of the control method can be suppressed.

  FIG. 13 is a diagram illustrating a simulation result of the motor drive system according to the present embodiment.

  The upper part of FIG. 13 shows a graph of the speed command value ωtgt (thick line) and the speed estimated value ωest (thin line). In the example of FIG. 13, the speed command value ωtgt is accelerated at a constant acceleration according to a so-called trapezoidal speed profile, becomes a steady speed, and then decelerates at a constant acceleration.

  The middle part of FIG. 13 shows a control method selection state by the control method selection unit 111. The selected state is the state OP1 after the start of the motor 1 while the speed command value ωtgt is less than the speed threshold value ω1. The selected state transitions to state OP2 when the speed command value ωtgt is equal to or greater than the speed threshold value ω1, and transitions to state CL when a predetermined period T1 has elapsed since the transition to the state OP2, and the speed command value ωtgt becomes the speed threshold value ω2. When it is less than, it is transited to the state OP1.

  The lower part of FIG. 13 shows the value of the selection signal sel. The selection signal sel is 0 until the predetermined period T1 elapses after the speed command value ωtgt becomes equal to or higher than the speed threshold ω1 after the motor 1 is started, and then becomes 1. The selection signal sel becomes 0 when the speed command value ωtgt is less than the speed threshold value ω2.

  That is, after the motor 1 is started, the open loop control is performed until the predetermined period T1 elapses after the speed command value ωtgt becomes equal to or higher than the speed threshold ω1, and then the closed loop control is performed. The motor 1 is controlled by open loop control when the speed command value ωtgt is less than the speed threshold value ω2.

  As can be seen from FIG. 13, the estimated speed value ωest is an unstable value until the phase synchronization of the angle estimation is established after the motor 1 is started, and substantially coincides with the speed command value ωtgt after the establishment of the phase synchronization. Yes. Further, the estimated speed value ωest is unstable again when the speed command value ωtgt becomes smaller during deceleration of the motor 1 because angle estimation becomes difficult.

  In the example of FIG. 13, the speed thresholds ω <b> 1 and ω <b> 2 are set so that the open loop control is selected during a period when the value of the estimated speed value ωest is unstable. By setting the speed threshold values ω1 and ω2 in this way, the motor 1 can be controlled in an open loop during a period in which the estimated speed value ωest is unstable (a period in which the motor 1 cannot be accurately controlled by closed loop control). . Further, the motor 1 can be closed-loop controlled during a period in which the estimated speed value ωest follows the speed command value ωtgt (a period during which the motor 1 can be accurately controlled by the closed-loop control). As a result, the motor 1 can be accurately controlled over the entire period during which the motor 1 is being driven.

  Further, as in the example of FIG. 13, by setting the speed thresholds ω1 and ω2 to be lower than the steady speed, the motor 1 can be closed-loop controlled while the motor 1 is at the steady speed. In general, the period during which the motor 1 is driven at a steady speed is significantly longer than the acceleration period or deceleration period of the motor 1. For this reason, by setting the speed threshold values ω1 and ω2 as described above, the motor 1 can be closed-loop controlled during most of the drive period, and the power consumption of the motor 1 can be greatly reduced. Accordingly, the heat generation of the motor 1 can be suppressed, and the motor 1 can be reduced in size and cost.

  The configuration of the motor drive system according to the present embodiment is not limited to FIG. For example, a configuration in which the a-axis current value ia and the b-axis current value ib are directly controlled without using the dq coordinate system is also possible. In this case, the a-axis current value ia and the b-axis current value ib may be controlled so as to satisfy the expressions (17) to (20).

Second Embodiment
A motor drive system according to the second embodiment will be described with reference to FIGS. 14 and 15. In the present embodiment, another example of the angle control unit 102 will be described.

  FIG. 14 is a diagram illustrating a configuration of a motor drive system according to the present embodiment. As shown in FIG. 14, in this embodiment, the speed command value ωtgt output from the angle command value calculation unit 101 is input to the angle control unit 102 and the control method selection unit 111. Other configurations are the same as those of the first embodiment (FIG. 1).

  FIG. 15 is a diagram illustrating an example of the angle control unit 102 according to the present embodiment. As illustrated in FIG. 15, the speed command value ωtgt is input to the comparison unit 1023. The comparison unit 1023 adds the difference between the speed estimated value ωest and the speed command value ωtgt and the operation amount C and outputs the result. Other configurations are the same as those of the first embodiment (FIG. 4). With this configuration, in the present embodiment, the current command value itgt is expressed by the following equation.

  As can be seen from the equations (1) and (21), the current command value itgt in the present embodiment is obtained by adding the speed command value ωtgt to the first term on the right side of the current command value itgt in the first embodiment. In the present embodiment, the speed command value ωtgt is used as a target value for speed control.

  In the angle control unit 102 of the first embodiment (FIG. 4), the speed control is performed with the operation amount C as a target value. Therefore, the operation amount C is a value proportional to the speed command value ωtgt. That is, the difference between the angle command value θtgt and the estimated angle value θest is a value proportional to the speed command value ωtgt. As a result, the higher the motor 1 is driven, the larger the difference between the angle command value θtgt and the estimated angle value θest.

  On the other hand, in the angle control unit 102 of the present embodiment (FIG. 15), the speed control is performed using the speed command value ωtgt as a target value, so the operation amount C is operated independently of the speed command value ωtgt. The As a result, the operation amount C is not a value proportional to the speed command value ωtgt, but is a value near zero. Therefore, according to the angle control unit 102 of the present embodiment, the followability to the angle command value θtgt can be improved.

  Further, in the angle control unit 102 of the first embodiment, the response to the change in the speed command value ωtgt (speed control response) is performed after the response to the change in the angle command value θtgt (angle control response).

  On the other hand, in the angle control unit 102 of the present embodiment, when the speed command value ωtgt changes, the speed control response is instantly started. Therefore, according to the angle control unit 102 of the present embodiment, the response speed with respect to the change in the speed command value ωtgt can be improved, and the followability to the speed command value ωtgt can be improved.

<Third Embodiment>
A motor drive system according to a third embodiment will be described with reference to FIGS. In the present embodiment, another example of a control method selection method by the control method selection unit 111 will be described.

  FIG. 16 is a diagram illustrating a configuration of a motor drive system according to the present embodiment. As shown in FIG. 16, in this embodiment, the estimated speed value ωest output from the angle estimation unit 110 is input to the angle control unit 102 and the control method selection unit 111. Other configurations are the same as those of the first embodiment (FIG. 1).

  FIG. 17 is a state transition diagram illustrating a control method selection method according to the present embodiment. As shown in FIG. 17, in the present embodiment, the conditions (selection method of the control method) for changing the selection state are as follows.

  In the state OP1, when the speed command value ωtgt is equal to or greater than the predetermined speed threshold value ω1, and the difference between the speed command value ωtgt and the estimated speed value ωest (hereinafter referred to as “speed error ωerr”) is less than the error threshold value e, The state OP1 transitions to the state OP2.

  In the state OP2, when the speed command value ωtgt becomes less than the predetermined speed threshold value ω2 or the speed error ωerr is equal to or greater than the error threshold value e, the state OP2 transitions to the state OP1. Further, in the state OP2, when the duration t of the state OP2 is equal to or longer than the predetermined period T1, the state OP2 transitions to the state CL.

  In the state CL, when the speed command value ωtgt is less than the predetermined speed threshold value ω2 or the speed error ωerr is greater than or equal to the error threshold value e, the state CL transitions to the state OP1.

  The error threshold e is a preset threshold value of the speed error ωerr. The error threshold e is set by experimentation or simulation as an upper limit value of the speed error ωerr that can stably perform the closed loop control.

  FIG. 18 is a flowchart showing an example of the operation of the motor drive system in the present embodiment. The flowchart in FIG. 18 includes step S118. Steps S101 to S117 in FIG. 18 are the same as those in the first embodiment (FIG. 12).

  As shown in FIG. 18, in this embodiment, when the speed command value ωtgt is equal to or greater than the speed threshold ω2 (NO in step S102), the control method selection unit 111 determines whether the speed error ωerr is equal to or greater than the error threshold e. (Step S118).

  When the speed error ωerr is equal to or greater than the error threshold e (YES in step S118), the control method selection unit 111 selects open loop control as the control method and outputs 0 as the selection signal sel (step S103). The subsequent processing is as described above.

  On the other hand, when the speed error ωerr is less than the error threshold value e (NO in step S118), the process proceeds to step S112. The subsequent processing is as described above.

  As described above, according to the present embodiment, the closed loop control is selected when the speed error ωerr is small and stable closed loop control can be performed. Therefore, the stability of the control of the motor 1 by the closed loop control can be further improved.

<Fourth embodiment>
A motor drive system according to a fourth embodiment will be described with reference to FIGS. In the present embodiment, field weakening control will be described.

  FIG. 19 is a diagram illustrating a configuration of a motor drive system according to the present embodiment. The configuration of the motor drive system according to the present embodiment is the same as that of the first embodiment except for the first selection unit 103. Hereinafter, the first selection unit 103 will be described.

  As shown in FIG. 19, the first selection unit 103 receives the d-axis voltage command value Vd, the q-axis voltage command value Vq, and the power supply voltage Vcc instead of the current value 0. In the present embodiment, when the selection signal sel is 0 (when the control method is open loop control), the first selection unit 103 selects the current value iref as the d-axis current command value idt. Therefore, the operation of the motor drive system when the selection signal sel is 0 is the same as in the first embodiment.

  On the other hand, when the selection signal sel is 1 (when the control method is closed loop control), the first selection unit 103 uses the d-axis based on the d-axis voltage command value Vd, the q-axis voltage command value Vq, and the power supply voltage Vcc. A value that is a predetermined value times the q-axis current command value iqt is output as the current command value idt. The d-axis current command value idt in the present embodiment will be described in detail later.

  Here, the operation of the first selection unit 103 during the closed loop control will be described. As described above, the d-axis current value id follows the d-axis current command value idt, and the q-axis current value iq follows the q-axis current command value iqt. Therefore, controlling the d-axis current command value idt and the q-axis current command value iqt corresponds to controlling the d-axis current value id and the q-axis current value iq.

  While the motor 1 is driven, a counter electromotive voltage proportional to the speed of the rotor 11 and the strength of the magnetic flux is generated between the terminals of the A-phase coil 12A and the B-phase coil 12B. Therefore, when the closed loop control is established, the d-axis voltage command value Vd and the q-axis voltage command value Vq are required to satisfy the following expressions, respectively.

  In the equations (22) and (23), R and L are the resistance value (ohm) and inductance (H) of the A-phase coil 12A and the B-phase coil 12B, w is the rotational speed (rad / sec) of the rotor 11, Ke is a counter electromotive voltage constant (V / rad / sec) generated in proportion to the rotational speed w.

  As in the first embodiment, when the d-axis current command value idt (d-axis current value id) is controlled to the current value 0 during closed-loop control, the d-axis voltage command value Vd is a negative value according to equation (22). It turns out that it becomes. Further, it can be seen from the equation (23) that when the rotation speed w is large, the q-axis voltage command value Vq becomes a large value.

FIG. 20 is a graph showing the relationship between the voltage command value and the rotation speed during closed-loop control. In the example of FIG. 20, it is assumed that the d-axis current command value idt is controlled to a current value of 0. In FIG. 20, the resistance value R is 9.1 (ohm), the inductance H is 20 mH, the back electromotive voltage constant Ke is 0.008 (V / rad / sec), the d-axis current value id is 0 (A), and the q-axis The current value iq is 0.3 (A), and the power supply voltage Vcc is 24 (V). The broken line indicates the d-axis voltage command value Vd, the solid line indicates the q-axis voltage command value Vq, and the alternate long and short dash line indicates the phase voltage Vab (= (Vd ² + Vq ² ) ^1/2 ).

  As shown in FIG. 20, the d-axis voltage command value Vd, the q-axis voltage command value Vq, and the phase voltage Vab are saturated when the amplitude becomes 24 V (power supply voltage Vcc). In the example of FIG. 20, the d-axis voltage command value Vd is saturated at about 2000 rpm, and the phase voltage Vab is saturated at about 1700 rpm. In general, since the closed loop control is established until either the d-axis voltage command value Vd or the q-axis voltage command value Vq is saturated, it can be understood that the closed-loop control can be performed up to about 2000 rpm in the example of FIG. In other words, in the example of FIG. 20, the closed loop control can be realized only up to about 2000 rpm.

  FIG. 21 is a graph showing the relationship between the voltage command value and the rotation speed during closed-loop control. In the example of FIG. 21, it is assumed that the d-axis current command value idt is controlled to −0.14 (A). In the example of FIG. 21, since the d-axis voltage command value Vd is not saturated even at 2500 rpm, it can be seen that the closed loop control can be performed up to a rotational speed of 2500 rpm or higher.

  In this way, the control for reducing the voltage command value required to establish the closed loop control by controlling the d-axis current value id to a value other than 0 (A) is called field weakening control. By the field weakening control, it is possible to decrease the voltage command value and increase the upper limit value of the rotational speed at which the closed loop control can be normally realized. The first selection unit 103 in the present embodiment realizes this field weakening control by outputting the ratio A between the d-axis current command value idt and the q-axis current command value iqt as the d-axis current command value idt.

  FIG. 22 is a diagram illustrating an example of the first selection unit 103 according to the present embodiment. 22 includes an adder 1031, a subtractor 1032, an amplifier 1033, an integrator 1034, a multiplier 1035, a selector 1036, and a field weakening determination unit 1037. .

  The adder 1031 receives the d-axis voltage command value Vd and the q-axis voltage command value Vq. The adder 1031 adds the d-axis voltage command value Vd and the q-axis voltage command value Vq and outputs the result. The output value of the addition unit 1031 is input to the subtraction unit 1032.

  The subtraction unit 1032 receives the output value of the addition unit 1031 and the predetermined value V0 (second predetermined value). The predetermined value V0 is a constant value set in advance. The subtracting unit 1032 subtracts the output value of the adding unit 1031 from the predetermined value V0 and outputs the result. The output value of the subtraction unit 1032 is input to the amplification unit 1033.

  The amplification unit 1033 receives the output value of the subtraction unit 1032. The amplifying unit 1033 amplifies the output value of the subtracting unit 1032 with a predetermined gain K and outputs it. The gain K is a constant that determines the convergence speed of integration described later. By appropriately setting the gain K, it is possible to avoid a sudden change in the current command value idt and to suppress fluctuations in torque and rotational speed. The output value of the amplifying unit 1033 is input to the integrating unit 1034.

  The integration unit 1034 receives the output value of the amplification unit 1033, the predetermined value A0 (first predetermined value), and the switching signal load. The predetermined value A0 is a constant value set in advance. The switching signal load is a binary signal that switches between enabling and disabling field weakening control, and is input from the field weakening determination unit 1037. Hereinafter, the field weakening determination unit 1037 outputs 0 as the switching signal load when the field weakening control is validated, and outputs 1 as the switching signal load when the field weakening control is invalidated.

  The integration unit 1034 outputs a predetermined value A0 as the output value A when 1 is input as the switching signal load (field weakening control is invalidated). The predetermined value A0 is 0, for example. In this case, since the current value 0 is output as the d-axis current command value idt, the same control as the closed loop control in the first embodiment is performed.

  On the other hand, when 0 is input as the switching signal load (field weakening control is enabled), the integrating unit 1034 outputs a value obtained by integrating the output value of the amplifying unit 1033 with the predetermined value A0. The output value A of the integration unit 1034 is input to the multiplication unit 1035.

  The multiplier 1035 receives the output value A of the integrator 1034 and the current command value itgt. Multiplier 1035 multiplies output value A by current command value itgt and outputs the result. The output value of the multiplier 1035 is input to the selector 1036.

  The selection unit 1036 receives the output value of the multiplication unit 1035, the current value iref, and the selection signal sel. When 0 is input as the selection signal sel, the selection unit 1036 outputs the current value iref as the d-axis current command value idt. On the other hand, when 1 is input as the selection signal sel, the selection unit 1036 outputs the output value of the multiplication unit 1035 as the d-axis current command value idt.

  The field weakening determination unit 1037 receives the d-axis voltage command value Vd, the q-axis voltage command value Vd, and the power supply voltage value Vcc. The field weakening determination unit 1037 determines whether to enable field weakening control based on the d-axis voltage command value Vd, the q-axis voltage command value Vd, and the power supply voltage value Vcc, and switches according to the determination result. The signal load is output. As described above, the field weakening determination unit 1037 outputs 0 as the switching signal load when the field weakening control is validated, and outputs 1 as the switching signal load when the field weakening control is invalidated.

  FIG. 23 is a state transition diagram illustrating a switching method of field weakening control in the present embodiment. Each node in FIG. 23 indicates a determination state of field weakening control by the field weakening determination unit 1037. Hereinafter, the determination states of the nodes N4 and N5 are referred to as states NORM and EN, respectively.

  The state NORM is a state in which the field weakening determination unit 1037 determines that the field weakening control is invalidated and 1 is output as the switching signal load. In the state NORM, when the magnitude (absolute value) of the d-axis voltage command value Vd is equal to or greater than the first threshold value V1, or when the magnitude (absolute value) of the q-axis voltage command value Vq is equal to or greater than the first threshold value V1 NORM transitions to state EN.

  The state EN is a state in which the field weakening determination unit 1037 determines that the field weakening control is to be enabled and outputs 0 as the switching signal load. In the state EN, when the magnitude (absolute value) of the d-axis voltage command value Vd is less than the second threshold value V2, and the magnitude (absolute value) of the q-axis voltage command value Vq is less than the second threshold value V2, State EN transitions to state NORM.

  The first threshold value V1 is set to a value near the power supply voltage value Vcc. The first threshold value V1 is, for example, 0.95 Vcc, but is not limited to this. By setting the first threshold value V1 in this way, when the d-axis voltage command value Vd approaches the power supply voltage value Vcc, the state NORM transitions to the state EN, and field weakening control is enabled. The saturation of the value Vd can be delayed. Further, when the q-axis voltage command value Vq approaches the power supply voltage value Vcc, the state NORM transitions to the state EN and field weakening control is validated, so that the saturation of the q-axis voltage command value Vq can be delayed.

  The second threshold value V2 is set to a value smaller than the first threshold value V1. The second threshold value V2 is, for example, 0.8 Vcc, but is not limited to this. When the magnitudes of the d-axis voltage command value Vd and the q-axis voltage command value Vq are less than the second threshold value V2 that is smaller than the first threshold value V1, the state EN is transitioned to the state NORM, thereby Can be suppressed. Thereby, the fluctuation | variation of a torque and rotation speed resulting from a state transition can be suppressed.

  With the above configuration, when closed loop control is selected (sel = 1) and field weakening control is enabled (load = 0), the integration unit 1034 has the d-axis voltage command value Vd and the q-axis voltage command value. The output value A is output so that the sum of Vq matches the predetermined value A0 (Vd + Vq = A0). For example, when the predetermined value A0 is 0, the d-axis voltage command value Vd and the q-axis voltage command value Vq are controlled by the output value A of the integration unit 1034 so that Vd + Vq = 0.

  At this time, since the q-axis current command value iqt is the current value itgt (iqt = itgt) and the d-axis current command value idt is A times the current value itgt, the output value A is the q-axis current command value. This corresponds to the ratio between iqt and the d-axis current command value idt (idt = A × iqt).

  Thus, when the field weakening control is effective, the integrating unit 1034 causes the d-axis current command value idt and the q-axis current command to be constant so that the sum of the d-axis voltage command value Vd and the q-axis voltage command value Vq is constant. It functions as an adjustment unit that adjusts the ratio A of the value iqt. For example, in the example of FIG. 21, Vd + Vq = 0 near 2500 rpm. This is a result of the output value A being controlled to about −0.46 by the integrating unit 1034 so that Vd + Vq = 0 (iqt = 0.3 (A), idt = −0.14 (A ) = A × iqt).

  As described above, according to the present embodiment, the ratio between the d-axis current command value idt and the q-axis current command value iqt so that the sum of the d-axis voltage command value Vd and the q-axis voltage command value Vq is constant. (Output value) A can be controlled. Thereby, it is possible to delay the saturation of the d-axis voltage command value Vd and the q-axis voltage command value Vq reaching the power supply voltage value Vcc. In other words, it is possible to widen the range of rotation speed and load that can be normally controlled by closed loop control (increase the upper limit value). Further, since the d-axis current command value idt and the q-axis current command value iqt can be linked, the torque utilization efficiency of the motor 1 can be improved.

  Further, according to the present embodiment, the ratio A can be calculated based on the d-axis voltage command value Vd and the q-axis voltage command value Vq without using specification values such as motor constants. Thereby, the influence of the manufacturing dispersion | variation in the motor 1 can be suppressed and stable field-weakening control can be realized.

  Moreover, since the 1st selection part 103 can be comprised by the combination of basic circuit elements, such as an adder and a subtractor, it can be manufactured cheaply.

  FIG. 24 is a diagram illustrating an example of the first selection unit 103 according to the present embodiment. The first selection unit 103 in FIG. 24 corresponds to a modification of the first selection unit 103 in FIG. 22, does not include the multiplication unit 1035, and the output value A of the integration unit 1034 is directly input to the selection unit 1036. Other configurations in FIG. 24 are the same as those in FIG. The configuration of FIG. 24 is the same as that in which a constant 1 is input instead of the current command value itgt to the integration unit 1035 of FIG.

  When the closed loop control is selected (sel = 1) and the field weakening control is enabled (load = 0), the first selection unit 103 in FIG. 24 outputs the output value A as the d-axis current command value idt. That is, the d-axis current command value idt does not depend on the current command value itgt. Therefore, even when the current command value itgt changes transiently due to a disturbance or the like, the influence of the change on the d-axis current command value idt is suppressed by the configuration shown in FIG. Can be suppressed.

  Note that the field-weakening control in the present embodiment is also applicable to a motor drive system including a position sensor such as an optical encoder or a magnetic encoder.

  It should be noted that the present invention is not limited to the configuration shown here, such as a combination with other elements in the configuration described in the above embodiment. These points can be changed without departing from the spirit of the present invention, and can be appropriately determined according to the application form.

1: Motor 2: Inverter 3: Current detector 100: Motor controller 101: Angle command value calculator 102: Angle controller 103: First selector 104: Second selector 105: d-axis current controller 106: q Axis current control unit 107: first vector rotation unit 108: second vector rotation unit 109: third selection unit 110: angle estimation unit 111: control method selection unit

JP 2011-67063 A JP 2011-139583 A

Claims (14)

A motor control device that controls a stepping motor based on a first current command value that depends on open-loop control and a second current command value that depends on closed-loop control,
A control method selection unit that selects the open loop control or the closed loop control based on a speed command value;
When the open loop control is selected, a first target value is output as the first current command value, and when the closed loop control is selected, a second target value is output as the second current command value. Command value output section to output,
A motor control device comprising: The phase of the drive current supplied to the stepping motor in the open loop control is synchronized with an angle command value, and the phase of the drive current is synchronized with an angle estimation value in the closed loop control. Motor control device. 3. The motor control device according to claim 1, further comprising an angle control unit that calculates the second target value based on an angle command value, an angle estimated value, and a speed estimated value. 3. The angle control unit according to claim 1, further comprising an angle control unit that calculates the second target value based on an angle command value, an angle estimation value, a speed estimation value, and the speed command value. Motor control device. The control method selection unit selects the closed-loop control and selects the closed-loop control when the period in which the speed command value is equal to or higher than the first speed threshold becomes a predetermined period or more during the selection of the open-loop control. The motor control device according to any one of claims 1 to 4, wherein the open loop control is selected when the speed command value becomes less than a second speed threshold value. 6. The motor control device according to claim 1, further comprising an estimation unit that calculates an angle estimation value and a speed estimation value based on a drive current supplied to the stepping motor. The motor control device according to any one of claims 1 to 6, further comprising a command value calculation unit that calculates the speed command value and the angle command value based on a reference pulse. When the closed loop control is selected and at least one of the two voltage command values is equal to or greater than a first threshold, the command value output unit outputs the two voltage command values as the first current command value. The motor control device according to any one of claims 1 to 7, wherein a value at which the sum is constant is output. The motor control device according to claim 8, wherein the first threshold value is a value corresponding to a power supply voltage value. The motor according to claim 8 or 9, wherein the command value output unit outputs a first predetermined value as the first current command value when the two voltage command values are less than a second threshold value. Control device. The motor control device according to claim 10, wherein the second threshold value is smaller than the first threshold value. The command value output unit is configured to calculate a ratio between the first current command value and the second current command value based on an integral value of a difference between a sum of the two voltage command values and a second predetermined value. The motor control device according to claim 8, wherein the motor control device is calculated. An inverter for supplying a drive current to the stepping motor;
A current detector for detecting the drive current;
The motor control device according to any one of claims 1 to 12,
A motor drive device comprising: The stepping motor;
A motor driving device according to claim 13;
A motor drive system comprising:

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