JP2016219976A - Communication method, communication system and communication device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a transmission device in which the phase estimation error does not increase and the bit error rate of the received signal does not deteriorate even in a low S/N environment or even when the number of discretized signal components in the transition region of the sub spectrum is small.SOLUTION: The transmission device 50 performs processing for rotating the phases of N sub-spectra before distributed arrangement or after distributed arrangement for each of a plurality of phase sequences having different finite number of phase combinations, selects and transmits the transmission signal of the phase sequence having the minimum peak-to-average power of the transmission signal for each phase sequence. The receiving apparatus 60 estimates the phase difference between the adjacent sub-spectrums with respect to the sub-spectrum after distributed arrangement extracted from the received signal or the sub-spectrum returned before distributed arrangement, selects the phase closest to the estimated phase difference among the finite number of phases to be shared in advance with the transmission device and corrects it by replacing the phase difference with the estimated phase difference, and demodulates the received signal by correcting the phase of the sub spectrum returned from the reception signal after distributed arrangement or before distributed arrangement.SELECTED DRAWING: Figure 4

Description

  The present invention relates to a communication technique using a band dispersion transmission system.

  In recent years, in order to improve the frequency band use efficiency in communication systems, a single-carrier modulation signal is divided into a plurality of sub-spectrums in the frequency domain on the transmission side, and the band dispersion transmission method is distributed in the unused band of the repeater Is considered. However, since the band dispersion transmission system divides and distributes the spectrum, there is a problem that the peak-to-average power ratio (PAPR) of the transmission signal increases, and a technique for reducing the PAPR is studied. (For example, refer nonpatent literature 1). The following is an example of a technique for reducing PAPR in communication using a band dispersion transmission method.

FIG. 9 shows an example of a transmission apparatus 150 that uses the band dispersion transmission method. 9, the transmitter 150 includes a modulator 101, a waveform shaping filter 102, DFT (Discrete Fourier Transform) 103, the split filter 104-1 to _{104-N D,} phaser 105-2-1~105-C-N _D, frequency shifter _{106-1-1~106-C-N D,} adder 107-1~107-C, IDFT (Inverse DFT ) 108-1~108-C, PAPR calculator 109-1 to 109-C And a minimum PAPR selector 110. Here, N _D and C are positive integers. The transmission device 150 includes a control unit 151 that controls the operation of each unit, and a storage unit 152 that stores a plurality of phase sequences.

  The modulator 101 modulates a data signal to be transmitted by a modulation method such as QPSK (Quadrature Phase Shift Keying).

  The waveform shaping filter 102 is a filter for limiting the band of the modulation signal output from the modulator 101.

  The DFT 103 converts the modulation signal output from the waveform shaping filter 102 into the frequency domain (spectrum).

The division filter 104-k (1 ≦ k ≦ N _D ) is a filter for dividing the band of the modulated signal converted into the frequency domain by the DFT 103 into N _D pieces. Splitting filter 104-1 to _{104-N D,} in each of the divided filter, by multiplying the preset filter coefficients to the modulated signal, to generate a _{N D} sub-modulated signal (sub spectrum). Here, division filter 104-k (k is 1 ≦ k ≦ _N a positive integer satisfying _D), the respective output signal is branched to the number C, C frequency shifter the first branch signal among the pieces 106-1 −k (1 ≦ k ≦ N _D ), the second to C-th branch signals among the C signals are input to the phase shifter 105-q-k (2 ≦ q ≦ C, 1 ≦ k ≦ N _D ). Here, q is a positive integer.

The phase shifter 105-q-k (2 ≦ q ≦ C, 1 ≦ k ≦ N _D ) adds a predetermined phase to a signal obtained by branching each output signal of the division filter 104-k. Then, respective output signals of the phase shifter 105-q-k is input to the frequency shifter 106-q-k (2 ≦ q ≦ C, 1 ≦ k ≦ N D).

Here, the processing of the phase shifter 105-qk will be described. The phase shifter 105-q-k, the phase sequence theta ₂ represented by the formula _{(1), Θ 3, ...} , Θ q, ..., using a theta _C, each sub-spectrum input to the phase shifter 105-q-k To the phase of each phase series. Note that the phase sequence given by Equation (1) is stored in, for example, the storage unit 152 and read out by the control unit 151.

Here, in the formula (1), θ _q1 is 0. Thereby, for example, the receiving apparatus 160 illustrated in FIG. 11 can estimate the phase difference with reference to the sub-spectrum of k = 1. Further, the sub-spectrum input to the phase shifter 105-q-k is _assumed to be SS _qk . The phase shifter 105-qk is added to the _{N D} sub spectrum _{SS qk} of the phase of the phase sequence theta _q represented by the formula (2) from the same q-th k = 1 to _{N D.}

The output signal S ^θ _q of the phase shifter 105-q-k after the phase multiplication is expressed by Equation (3).

The frequency shifter 106-q-k (1 ≦ q ≦ C, 1 ≦ k ≦ N _D ) is the division filter 104-k or the phase shifter 105-q-k (2 ≦ q ≦ C, 1 ≦ k ≦ N _D ). Are shifted to a predetermined desired band on the frequency axis. For example, the frequency shifter 106-q-k shifts the sub-spectrum to the unused band of the satellite repeater and distributes the signal band of the transmission signal. The frequency shifter 106-q-k performs processing for each of 1 to C. For example, when q = 1, each sub-spectrum of the frequency shifter 106-1-k (1 ≦ k ≦ N _D ) is distributed and arranged in a desired band on the frequency axis. Similarly, each sub-spectrum of the frequency shifter 106-2-k, each sub-spectrum of the frequency shifter 106-3-k,..., Each sub-spectrum of the frequency shifter 106-C-k is set to 1 of the frequency shifter 106-1-k. from shifting in the same shift amount and shift amount of each sub-spectrum up to N _D. Note that the shift amount of each sub-spectrum is controlled by the control unit 151, for example.

The adders 107-q (1 ≦ q ≦ C) are output from 1 to N which are output by the frequency shifter 106-q-k (1 ≦ q ≦ C, 1 ≦ k ≦ N _D ) for each group (for each same q). adding the N _D sub-spectrum of up to _D. In figure 9, the adder 107-1 adds the _{N D} sub spectrum frequency shifter _{106-1-N D} is output from the frequency shifter 106-1-1. Similarly, the adder 107-C adds _{N D} sub spectrum frequency shifter _{106-C-N D} is output from the frequency shifter 106-C-1.

  IDFT 108-q (1 ≦ q ≦ C) converts the output signal of adder 107-q from a frequency domain signal to a time domain signal. For example, the IDFT 108-1 converts the output signal of the adder 107-1 from a frequency domain signal to a time domain signal.

  The PAPR calculator 109-q (1 ≦ q ≦ C) calculates a PAPR for the signal converted into a time domain signal by the IDFT 108-q. For example, the PAPR calculator 109-1 calculates the PAPR for the signal converted into the time domain signal by the IDFT 108-1. The PAPR is calculated using a well-known technique. For example, the maximum value and the average value of the squared values of the signals input to the PAPR calculators 109-1 to 109-C are calculated. Then, the PAPR calculator 109-q outputs the maximum value / average value (value obtained by dividing the maximum value by the average value) as PAPR.

  Minimum PAPR selector 110 selects and transmits the output signal of IDFT 108-q of the sequence with the smallest PAPR calculated by PAPR calculator 109-q. For example, when the PAPR calculated by the C = 2 PAPR calculator 109-2 is the smallest, the minimum PAPR selector 110 selects and transmits the output signal of the IDFT 108-2 of the same sequence (C = 2).

  Here, the spectrum of the input / output signal of the division filter 104-k, the spectrum of the output signal of the phase shifter 105-q-k, and the spectrum of the output signal of the adder 107-q will be described.

FIG. 10 shows an example of the frequency waveform of each part of the transmission apparatus 150. FIG. 10 shows a case where N _D = 4. FIG. 10A shows an example of amplitude characteristics and phase characteristics of input signals of the division filters 104-1 to 104-4. In FIG. 10A, the amplitude characteristic of (a-1) indicates the frequency f on the horizontal axis and the amplitude A on the vertical axis, and the sub-spectrums of the divided filters 104-1 to 104-4 and the sub-spectra before being divided. A spectrum 141 of the modulated signal is shown. Here, in FIG. 10A-1, the frequency bands of the adjacent division filter 104-1 and division filter 104-2 are superimposed. Similarly, the respective frequency bands are superimposed on the adjacent division filter 104-2 and division filter 104-3, division filter 104-3 and division filter 104-4. 10A, in the phase characteristic of (a-2), the horizontal axis indicates the frequency f, the vertical axis indicates the phase θ, and the phase characteristic 142 of the modulation signal is proportional to the frequency f.

  FIG. 10B shows an example of the amplitude characteristics and phase characteristics of the output signals of the division filters 104-1 to 104-4. (B-1) in FIG. 10B shows an example of the amplitude characteristic and phase characteristic of the output signal of the division filter 104-1. Similarly, in FIG. 10B, (b-2) is the divided filter 104-2, (b-3) is the divided filter 104-3, and (b-4) is the output signal of the divided filter 104-4. An example of the amplitude characteristic and the phase characteristic is shown. As shown in FIG. 10B, the division filters 104-1 to 104-4 divide the output signal of each division filter so that the amplitude characteristics and phase characteristics overlap between adjacent filters.

FIG. 10C shows an example of the amplitude characteristic and the phase characteristic of the output signal of the phase shifter 105-qk. (C-1) of FIG.10 (c) shows an example of the amplitude characteristic and phase characteristic of the output signal of phase shifter 105-q-1. Since the phase θ ₂₁ to be added by the phase shifter 105-q-1 is 0, the phase characteristics are the same as those of the division filter 104-1 shown in (b-1) of FIG. In FIG. 10C, (c-2) shows an example of the amplitude characteristic and phase characteristic of the output signal of the phase shifter 105-q-2. Since the phase shifter 105-q-2 adds the phase θ ₂₂ to the output signal of the division filter 104-2, the phase characteristics shown in (c-2) of FIG. (b-2) phase-splitting filter 104-2 shown in are shifted theta ₂₂ to the paper upward. Similarly, in FIG.10 (c), (c-3) shows an example of the amplitude characteristic and phase characteristic of the output signal of phase shifter 105-q-3. The phase shifter 105-q-3, since adding the phase theta ₂₃ to the output signal of the dividing filter 104-3, the phase characteristics shown in FIG. 10 (c) (c-3) is shown in FIG. 10 (b) (b-3) a phase of dividing filter 104-3 shown in are shifted theta ₂₃ to the paper upward. In FIG. 10C, (c-4) shows an example of the amplitude characteristic and phase characteristic of the output signal of the phase shifter 105-q-4. Since the phase shifter 105-q-4 adds the phase θ ₂₄ to the output signal of the division filter 104-4, the phase characteristic shown in (c-4) of FIG. The phase of the divided filter 104-4 shown in (b-4) is shifted upward by [theta] ₂₄ in the drawing.

  FIG. 10D shows an example of the amplitude characteristic of the output signal of the adder 107-q. In FIG. 10D, the horizontal axis indicates the frequency f, and the vertical axis indicates the amplitude A. For example, the adder 107-1 shown in FIG. 9 has a frequency shifter 106-1-1, a frequency shifter 106-1-2, a frequency shifter 106-1-3, and a frequency shifter 106-1-4 when q = 4. Are added together. In this case, as shown in FIG. 10D, the spectrum of the output signal of the adder 107-1 is the sub-spectrum SS1 of the output signal of the frequency shifter 106-1-1 and the output signal of the frequency shifter 106-1-2. The sub-spectrum SS2, the sub-spectrum SS3 of the output signal of the frequency shifter 106-1-3, and the sub-spectrum SS4 of the output signal of the frequency shifter 106-1-4 are added.

FIG. 11 shows an example of a receiving apparatus 160 that uses a band dispersion transmission method. 11, the receiving device 160, DFT111, extraction filter 112-1 to _{112-N D,} the frequency shifter 113-1 to _{113-N D,} the phase estimator _{114-2~114-N D,} phaser 115-2 115-N _D , adder 116, IDFT 117, and demodulator 118. The receiving apparatus 160 includes a control unit 161 that controls the operation of each unit.

  The DFT 111 converts the received signal into a frequency domain signal.

Extraction filter 112-k (1 ≦ k ≦ N D) is the output signal of DFT111 which is branched to _{the N D} respectively input, extracts the _{N D} sub spectrum.

Frequency shifter 113-k (1 ≦ k ≦ N D) is the respective sub-spectrum extraction filter 112-1 to _{112-N D} outputs, frequency shift by the frequency shifter 106-1 to _{106-N D} of the transmission device 150 Shift to the previous band.

Phase estimator 114-k (2 ≦ k ≦ N D) estimates the phase difference of the sub-spectrum of a signal that adjacent ones of the frequency shifters 113-1 to _{113-N D.} In FIG. 11, for example, the phase estimator 114-2 receives the output signals of the frequency shifter 113-1 and the frequency shifter 113-2 and adds them in the phase shifter 105-q-2 of the transmission device 150. Estimate the phase difference. The phase estimator 114-k receives signals of sub-spectrums (SS _k−1 and SS _k ) adjacent in the frequency domain and calculates a phase difference R _{k_ab} in the transition region of SS _k−1 and SS _k . When the phase difference R _{k_ab} is converted to radians, the phase difference (^ θ _qk ) = atan (R _{k_ab} ) is obtained. Here, as shown in (a-1) of FIG. 10 (a), in the transmission apparatus 150, since the transition areas of the sub-spectrums overlap each other, the transition areas of the adjacent sub-spectrums have the same signal component. have. Therefore, the receiving apparatus 160 estimates the phase difference R _{k_ab} from the signal component of the overlapping transition region. Here, k denotes the number of sub-spectrum from 1 to N _D. Note that θ _q1 is set to 0 as described in Expression (1). That, SS ₁ is because theta ₂₁ is 0, since the transmitting apparatus 150 phase has not been added, the phase estimator 114-2, SS ₁ and SS input to the frequency shifter 113-1 and 113-2 and _2, it is possible to estimate the phase _{R 2_12,} which is added to the SS _2. Then, the phase estimator 114-2 outputs (^ θ _q2 ) = atan (R 2 — _ab ) to the phase shifter 115-2. Hereinafter, from the phase estimator 114-3 _{114-N D,} similarly to the phase estimator 114-2 estimates the phase difference _{R k_ab.} Then, the phase estimator 114-k _{outputs the} phase difference (^ θ _qk ) = atan (R k — _ab ) to the phase shifter 115-k.

Phaser 115-k (2 ≦ k ≦ N D) , the phase difference to be output from the phase estimator _{114-2~114-N D (^ θ qk} ) is inputted, a frequency shifter 113-1～113- by multiplying the _{exp (-j (^ θ qk)} ) to a signal N _D is outputted, to compensate the phase. For example, the phase shifter 115-2 receives the phase difference (^ θ _q2 ) output from the phase estimator 114-2, and uses exp (−j (^ θ _q2 )) as the signal output from the frequency shifter 113-2. To compensate for the phase. Similarly, the phase difference _{R 3_Ab} from the output signal SS ₃ of the phase compensated signal SS ₂ and the frequency shifter 113-3 in the phase estimator 114-3 is calculated by the phase shifter 115-2, the phase shifter 115-3 (^ θ _q3 ) = atan (R ₃ — _ab ) is output. The phase shifter 115-3 uses the output (^ θ _q3 ) of the phase estimator 114-3 to multiply SS ₃ output from the frequency shifter 113-3 by exp (−j (^ θ _q3 )) to obtain the phase. To compensate.

The adder 116 adds the output signal of the frequency shifter 113-1 and phase shifter 115-2～ phaser _{115-N D,} return to the previous signal waveform into a plurality of sub-spectra by the transmission device 150.

  The IDFT 117 converts the output signal of the adder 116 into a time domain signal.

  The demodulator 118 demodulates the modulation signal output from the IDFT 117. For example, when the modulator 101 of the transmission apparatus 150 modulates with QPSK, the received data is demodulated with the same QPSK.

  Here, the input / output signal spectrum of the phase shifter 115-k and the correlation calculation in the phase estimator 114-k will be described.

  FIG. 12 shows an example of the frequency waveform of each part of the receiving apparatus 160. FIG. 12 shows a case where k = 2 to 5. FIG. 12A shows an example of the amplitude characteristics and phase characteristics of the input signals of the phase shifters 115-2 to 115-5. In FIG. 12A, each of the amplitude characteristics from (a-1) to (a-4) shows the frequency f on the horizontal axis and the amplitude A on the vertical axis, and each of the frequency shifters 113-2 to 113-5. The sub-spectrums SS1 to SS4 are shown. Here, in (a-1) of FIG. 12A, the phase 146-2 of the frequency shifter 113-2 is the same as the phase 145 before the shift because the shift amount is zero. On the other hand, the phase 146-3 of the frequency shifter 113-3 shown in (a-2) of FIG. 12A has a phase difference from the phase 145 before the shift. Similarly, both of the phase 146-4 of the frequency shifter 113-4 shown in (a-3) of FIG. 12A and the phase 146-5 of the frequency shifter 113-5 shown in (a-4), There is a phase difference with respect to the phase 145 before the shift.

FIG. 12B shows an example of the amplitude characteristics and phase characteristics of the output signals of the phase shifters 115-2 to 115-5. In FIG. 12B, the amplitude characteristics from (b-1) to (b-4) are the same as those from (a-1) to (a-4) in FIG. In the example shown in (b-1) of FIG. 12B, the phase shifter 115-2 has a phase difference (^ θ ₂₁ ) between the phase 146-2 of the frequency shifter 113-2 and the phase 145 before the shift is 0. Therefore, no phase shift is performed. On the other hand, in the example shown in (b-2) of FIG. 12B, the phase shifter 115-3 includes the phase difference (^ θ ₂₂ ) between the phase 146-3 of the frequency shifter 113-3 and the phase 145 before the shift. To compensate for equal phase. Similarly, in the example shown in (b-3) of FIG. 12B, the phase shifter 115-4 includes the phase difference (^ θ ₂₃₎ between the phase 146-4 of the frequency shifter 113-4 and the phase 145 before the shift. ) To compensate for equal phase. In the example shown in (b-4) of FIG. 12B, the phase shifter 115-5 includes the phase difference (^ θ ₂₄ ) between the phase 146-5 of the frequency shifter 113-5 and the phase 145 before the shift. To compensate for equal phase.

FIG. 12C shows an example of the correlation calculation in the phase estimator 114-k. Here, the signal component of each sub-spectrum is discretized on the frequency axis. In FIG. 12C, (c-1) shows an adjacent low-frequency side sub-spectrum SS _k-1 , and (c-2) shows an adjacent high-frequency side sub-spectrum SS _k- 1. In the sub-spectrum SS _k-1 shown in (c-1) of FIG. 12C, the signal components discretized in the transition region on the low frequency side are a _{(k-1) 1} , a _{( k-1) 2} ,..., a _{(k-1) p} , signal components discretized in the transition region on the high frequency side from the low frequency side b _{(k-1) 1} , b _{(k-1) 2} , ..., b _{(k-1) p} . Further, in the sub-spectrum SS _k shown in (c-2) of FIG. 12C, the signal components discretized in the transition region on the low frequency side are a _k1 , a _k2 ,..., A _kp from the low frequency side. , B _k1 , b _k2 ,..., B _kp from the low frequency side are signal components discretized in the transition region on the high frequency side. Here, p is the number of signal components included in the transition band among the discrete signal points generated in the frequency domain by DFT. FIG. 12C shows an example where p = 3. For example, when the frequency resolution of the DFT 111 is r and the bandwidth of the transition band is B _t , p = [B _t / r]. However, the symbol [x] is the maximum integer not exceeding x.

As shown in Expression (4), the phase difference R _{k — ab} is obtained by _combining the signal component b _(k−1) on the high frequency side and the signal component a ^* _ki on the low frequency side of the adjacent sub-spectrums SS _k−1 and SS _k . It is obtained by multiplying the complex conjugate and smoothing in the frequency axis direction.

Here, in the example of p = 3 shown in _FIG.12 (c), phase difference _{Rk_ab} is represented by Formula (5).

Here, in order to confirm whether or not the phase is added to each sub-spectrum by the phase shifter 105-q-k of the transmission device 150, the reception device 160 has a value (threshold α) that is mounted in advance in the reception device 160. ). For example, when the phase difference R _{k_ab} is greater than or _equal to the threshold value α, it is considered that the phase is added to the k-th sub-spectrum k in the transmission device, and the reception device 160 outputs the phase difference R _{k_ab} from the phase estimator 114-k. The phase difference is compensated by the phase shifter 115. Here, since b _{(k−1) p} and a _kp are the same signal components before the division, the phase difference (^ θ _qk ) = atan generated between b _{(k−1) p} and a _{kp in} the receiving apparatus. It is _{assumed that} (R _{k — ab} ) is due to the phase added by the transmitter 150, and the phase difference can be compensated by multiplying the sub-spectrum SS _k by exp (−j (^ θ _qk )). When the phase difference R _{k_ab} is less than the threshold value α, it is determined that the transmitter 150 has not added a phase, and the phase estimator 114-k outputs 0.

  With the above configuration, the receiving apparatus 160 can sequentially estimate the phase difference between adjacent sub-spectrums and compensate for the phase added by the transmitting apparatus 150.

Satoshi Miyatake, Junichi Abe, Takatoshi Sugiyama “A Study on PAPR Reduction in Bandwidth Distributed Transmission System”, General Conference, IEICE, March 2015.

In the prior art, the receiving apparatus 160, the bandwidth is estimated phase obtained by adding in phase shifter 105-q-1~105-q- N D of the transmission device 150 from the phase difference signal between adjacent sub-spectrum. However, when the number p of discretized signal components existing in a low S / N environment or in the transition region of the sub-spectrum is small, the number of smoothing in the frequency axis direction becomes insufficient, and the estimation error of the phase difference R _{k_ab} Will increase. As a result, the receiving apparatus 160 has a problem that the bit error rate (BER) characteristics of the received signal are deteriorated.

  The present invention improves the phase estimation accuracy on the receiving side, and deteriorates the BER of a received signal that deteriorates in a low S / N environment or due to a lack of the number p of discrete signal components existing in the transition region of the subspectrum. It is an object of the present invention to provide a communication method, a communication system, and a communication device that can improve characteristics.

  A first invention is a communication method for transmitting N (N is an integer of 2 or more) sub-spectrums obtained by dividing a modulation signal in a frequency domain in a distributed manner on the frequency axis and transmitting the transmission signal from a transmission device to a reception device. The transmitter performs the process of rotating the phase of each of the N sub-spectrums before or after distributed arrangement for each of a plurality of phase sequences having different combinations of a finite number of phases including the case where the phases are not added. A peak-to-average power ratio, which is a ratio between the maximum power and average power of the transmission signal for each sequence, is obtained, and a phase-sequence transmission signal that minimizes the peak-to-average power ratio is selected and transmitted to the receiving device. Estimates the phase difference between sub-spectrums whose frequency bands are adjacent to the sub-spectrum after dispersion arrangement extracted from the received signal or the sub-spectrum returned before dispersion arrangement; Of the finite number of phases shared in advance, the phase closest to the estimated phase difference is selected and corrected by replacing it with the estimated phase difference, and after the distributed arrangement extracted from the received signal by the corrected phase difference The received signal is demodulated by correcting the phase of the sub-spectrum or the sub-spectrum returned before the dispersion arrangement.

  In the second invention, the phase sequence is generated by m phases obtained by equally dividing or unequally dividing a circle into m (m is a positive integer), and the receiving device extracts the variance extracted from the received signal. Estimate the phase difference between adjacent sub-spectrums before placement or after dispersion placement, and calculate the Euclidean distance between the point on the estimated phase difference circle and each point on the circle of m phases, A phase having the minimum Euclidean distance is selected from the m phases and replaced with the estimated phase difference for correction.

  According to a third aspect of the present invention, there is provided a modulation unit that modulates transmission data, a division unit that divides a modulation signal output from the modulation unit into N (N is an integer of 2 or more) sub-spectrums in the frequency domain, A case where the first frequency transition unit that disperses and arranges the spectrum on the frequency axis and the process in which the first frequency transition unit rotates the phase of each of the N sub-spectrums before or after the dispersive arrangement is not phase-added. A first phase unit that is performed for each of a plurality of phase sequences having different combinations of a finite number of phases, and a peak pair that is a ratio of the maximum power and the average power of the transmission signal obtained by adding the sub-spectrum after dispersion arrangement for each phase sequence A transmission device including a selection unit that selects a transmission signal of a phase sequence that minimizes the average power ratio and transmits the transmission signal to the reception device, an extraction unit that extracts sub-spectrums distributed from the reception signal, A second frequency transition unit for returning the arranged sub-spectrum to the frequency band before the distributed arrangement, and a frequency band for the sub-spectrum after the distributed arrangement extracted by the extracting unit or before the distributed arrangement restored by the second frequency transition unit The phase estimation unit that estimates the phase difference between adjacent sub-spectrums and the phase estimated by selecting the phase closest to the phase difference estimated by the phase estimation unit from a finite number of phases shared in advance with the transmission device A phase correction unit that corrects the phase difference by replacing it with a phase difference, and a second phase unit that corrects the phase of the sub-spectrum after dispersion arrangement extracted from the received signal or the sub-spectrum returned before dispersion arrangement, by the phase difference corrected by the phase correction section And a demodulator that demodulates received data by adding the N sub-spectrums corrected by the second phase unit.

  In the fourth aspect of the invention, the phase sequence is generated by m phases obtained by equally dividing or unequally dividing a circle into m (m is a positive integer), and the phase correcting unit estimates the phase correcting unit. The Euclidean distance between the point on the circle with the phase difference and the points on the circle with m phases is calculated, and the phase estimated by selecting the phase with the minimum Euclidean distance among the m phases is selected. It is characterized by correcting by replacing with a phase difference.

  According to a fifth aspect of the invention, a modulation unit that modulates transmission data, a division unit that divides a modulation signal output from the modulation unit into N (N is an integer of 2 or more) sub-spectrums in the frequency domain, and N sub-bands A case where the first frequency transition unit that disperses and arranges the spectrum on the frequency axis and the process in which the first frequency transition unit rotates the phase of each of the N sub-spectrums before or after the dispersive arrangement is not phase-added. A first phase unit that is performed for each of a plurality of phase sequences having different combinations of a finite number of phases, and a peak pair that is a ratio of the maximum power and the average power of the transmission signal obtained by adding the sub-spectrum after dispersion arrangement for each phase sequence And a selection unit that selects a transmission signal of a phase sequence that minimizes the average power ratio and transmits the selection signal to the reception device.

  In the sixth invention, a finite number of phases are generated by m phases obtained by equally dividing or unequally dividing a circle into m (m is a positive integer), and the plurality of phase sequences are finite. The combination of phases is different.

  According to a seventh aspect of the present invention, there is provided an extraction unit that extracts a sub-spectrum in which signals received from a transmission device are distributed, a second frequency transition unit that returns the sub-spectrums that are distributed to the frequency band before the distribution, and an extraction unit A phase estimation unit that estimates a phase difference between sub-spectrums adjacent to each other in a frequency band with respect to a sub-spectrum after the distributed arrangement extracted by or before the distributed arrangement restored by the second frequency transition unit, and is shared in advance with the transmission apparatus Of the finite number of phases obtained from the information, the phase correction unit that selects the phase closest to the phase difference estimated by the phase estimation unit and replaces it with the estimated phase difference, and the phase difference corrected by the phase correction unit The second phase unit for correcting the phase of the sub-spectrum after dispersion arrangement extracted from the received signal or the sub-spectrum returned before the dispersion arrangement, and the second phase part corrected And having a demodulator for demodulating a received data by adding the sub-spectrum.

  In the eighth invention, a finite number of phases are generated by m phases obtained by equally dividing or unequally dividing a circle into m (m is a positive integer), and the phase correcting unit includes a phase estimating unit. Is calculated by calculating the Euclidean distance between each point on the circle of the phase difference estimated by and each of the points on the circle of m phases, and selecting the phase having the smallest Euclidean distance among the m phases. The correction is performed by replacing the phase difference.

  A communication method, a communication system, and a communication apparatus according to the present invention add a different phase for each sub-spectrum obtained by dividing a modulated signal into a plurality of bands on the transmission side and transmit the sub-spectrum. The phase can be compensated by estimating the phase added on the transmission side from the transition region of the spectrum and linearizing the phase. Then, the phase candidate value to be added on the transmission side is shared between the transmitting and receiving apparatuses, and the receiving apparatus selects the candidate value closest to the phase estimation value among the phase candidate values. As a result, the phase estimation accuracy in the receiving apparatus is improved, and the BER characteristic of the received signal deteriorates due to the low S / N environment or the lack of the number p of the discrete signal components present in the transition region of the subspectrum. Can be improved.

It is a figure which shows an example of the transmitter which concerns on this embodiment. It is a figure which shows an example of the phase (equal division) added with a phase shifter. It is a figure which shows an example of the phase (unequal division | segmentation) added with a phase shifter. It is a figure which shows an example of the receiver which concerns on this embodiment. It is a figure which shows an example of the phase corrector at the time of equal division. It is a figure which shows an example of the phase corrector at the time of unequal division. It is a figure which shows an example of the transmission method which concerns on this embodiment. It is a figure which shows an example of the receiving method which concerns on this embodiment. It is a figure which shows an example of the conventional transmitter. It is a figure which shows an example of the spectrum of a transmitter. It is a figure which shows an example of the conventional receiver. It is a figure which shows an example of the spectrum of a receiver.

  Hereinafter, embodiments of a communication method, a communication system, and a communication apparatus according to the present invention will be described with reference to the drawings.

  FIG. 1 shows a configuration example of a transmission apparatus 50 that uses a band dispersion transmission method. In FIG. 1, a communication system 70 includes a transmission device 50 and a reception device 60, and transmits a signal modulated by the transmission device 50 to the reception device 60 by a band dispersion transmission method. Here, the transmission device 50 or the reception device 60 is an example of a communication device. Alternatively, in the case of bidirectional communication, the communication device has both functions of the transmission device 50 and the reception device 60.

The transmission device 50 includes a modulator 1, a waveform shaping filter 2, a DFT 3, a division filter 4-1 to 4-N _D , a phase shifter 5-2-1 to 5-C N _D , and a frequency shifter 6-1-1 to 6-1. comprising _{6-C-N D,} adder 7-1~7-C, IDFT8-1~8-C, the PAPR calculator 9-1 to 9-C and minimum PAPR selector 10. Here, N _D and C are positive integers. The transmission device 50 includes a control unit 51 that controls the operation of each unit, a plurality of phase sequences composed of a predetermined finite number of phases, or parameters and formulas for generating a predetermined finite number of phases. , And the like. Alternatively, parameters and the like may be mounted on each block. Here, information on a finite number of phases to be used among a plurality of phase sequences or information such as parameters and mathematical formulas for generating a finite number of phases is shared between the transmission device 50 and the reception device 60. Yes. As a sharing method, information may be notified from the transmission device 50 to the reception device 60, or information such as parameters and mathematical expressions to be shared is standardized in advance, and then the transmission device 50 and the reception device 60 are shared. It may be implemented in. Thereby, it is not necessary to notify the information from the transmission device 50 to the reception device 60.

  The modulator 1 modulates a data signal to be transmitted by a modulation method such as QPSK (Quadrature Phase Shift Keying).

  The waveform shaping filter 2 is a filter for limiting the band of the modulation signal output from the modulator 1.

  The DFT 3 converts the modulation signal output from the waveform shaping filter 2 into the frequency domain (spectrum).

The division filter 4-k (1 ≦ k ≦ N _D ) is a filter for dividing the modulation signal band converted by the DFT 3 into the frequency domain into N _D pieces. Dividing filter 4-1 to 4-N _D, in each of the divided filter, by multiplying the preset filter coefficients to the modulated signal, to generate a N _D sub-modulated signal (sub spectrum). Here, division filter 4-k (k is 1 ≦ k ≦ N a positive integer satisfying _D) is a signal of each sub-spectrum is branched to C number, frequency first branch signal among the C-number shifter 6 −1−k (1 ≦ k ≦ N _D ), the second to C-th branch signals among the C signals are input to the phase shifter 5-q-k (2 ≦ q ≦ C, 1 ≦ k ≦ N _D ). To do. Here, q is a positive integer.

The phase shifter 5-q-k (2 ≦ q ≦ C, 1 ≦ k ≦ N _D ) adds a predetermined phase to a signal obtained by branching each output signal of the division filter 4-k. Then, respective output signals of the phase shifter 5-q-k is input to the frequency shifter 6-q-k (2 ≦ q ≦ C, 1 ≦ k ≦ N D).

Here, the processing of the phase shifter 5-qk will be described. The phase shifter 5-q-k, the phase sequence theta _D2 shown in equation _{(6), Θ D3, ...} , Θ Dq, ..., using a theta _DC, each sub-spectrum input to the phase shifter 5-q-k To the phase of each phase series. Note that the phase sequence given by Equation (6) may be stored in the storage unit 52, for example, or may be generated based on a predetermined parameter or mathematical expression. In the example of FIG. 1, the control unit 51 gives the phases of a plurality of phase sequences to the phase shifter 5-qk.

Here, in Expression (6), θ _Dq1 is set to 0. Thereby, the receiving apparatus 60 can estimate the phase difference between adjacent sub-spectrums with reference to the sub-spectrum of k = 1. Further, the sub-spectrum input to the phase shifter 5-q-k is SS _qk . The phase shifter 5-qk is added to the _{N D} sub spectrum _{SS qk} of the phase of the phase sequence theta _Dq represented by the formula (7) from the same q-th k = 1 to _{N D.}

[Characteristic part of transmitting apparatus 50 according to the present embodiment]
The difference between the transmitter 50 according to the present embodiment and the transmitter 150 of the prior art is that the phase θ _Dqk added by the phase shifter 5- _qk is a finite number of phase values, and the transmitter 50 and the receiver 60 Is shared as known information between the two. Note that the information shared between the transmission device 50 and the reception device 60 may be a finite number of phase values, or may be mathematical formulas or parameters for calculating a finite number of phase values.

For example, as a method for calculating a finite number of phase values, phase values obtained by equally dividing a circle can be used. In this case, any one of m phase values θ _Dqk calculated by Expression (8) is used in the phase shifter 5-qk.

Here, m indicates how much the circle is divided and is called the number of phase divisions. θ _offset represents the initial phase when the circle is _equally divided. l indicates a phase division position. In this case, the parameters (the value of the phase division number m and the value of the initial phase θ _offset ) and the equation (8) may be shared between the transmission device 50 and the reception device 60.

FIG. 2 shows an example of the phase θ _Dqk generated with the number of phase divisions m = 3 and the initial phase θ _offset = 0. The phase shifter 5-q-k selects any one of the m phases generated by the equation (8) and adds the selected phase to each sub-spectrum signal output from the division filter 4-k. At this time, as the same signal at a later stage of the adder 7-1 to 7-C is not outputted, the combination of phases used in the phase shifter 5-q-. 1 to phaser _{5-q-N D,} the phase shifter 5 -s-. 1 to phaser _{5-s-N D (2} ≦ q ≦ C, 2 ≦ s ≦ C, q ≠ s) and the combination of the phase is set differently in advance to be used in. Here, s is a positive integer.

  In the example of FIG. 2, when k = 8, circles are equally divided at m = 3, so that three generated phases (0 degrees, 120 degrees, and 240 degrees) are allocated to 8 sub-spectrums. Many combinations (corresponding to a plurality of phase sequences) are conceivable for the pattern (phase sequence) assigned to the sub-spectrum. Here, assuming that the eight sub-spectrums are “SS1, SS2,..., SS8” in order from the lowest frequency, a pattern (C-1) is obtained from the following pattern 1 as an example. Pattern 1 to pattern (C-1) are respectively provided from phase shifter 5-2-k to phase shifter 5-C-k shown in FIG.

  For example, in the case of pattern 1 (phase shifter 5-2k), SS1: 120 degrees, SS2: 0 degrees, SS3: 0 degrees, SS4: 0 degrees, SS5: 0 degrees, SS6: 0 degrees, SS7: 0 degrees , SS8: 0 degrees can be combined.

  In the case of pattern 2 (phase shifter 5-3-k), for example, SS1: 0 degrees, SS2: 120 degrees, SS3: 0 degrees, SS4: 0 degrees, SS5: 0 degrees, SS6: 0 degrees, SS7: Combinations such as 0 degrees and SS8: 0 degrees are possible.

  Similarly, in the case of pattern 3 (phase shifter 5-4-k), for example, SS1: 120 degrees, SS2: 120 degrees, SS3: 0 degrees, SS4: 0 degrees, SS5: 0 degrees, SS6: 0 degrees, SS7 : 0 degree, SS8: 0 degree combination.

  Similarly, from pattern 4 to pattern (C-2), three phases (0 degrees, 120 degrees, and 240 degrees) are assigned so as to be different from other patterns.

  In the case of the pattern (C-1) (phase shifter 5-Ck), for example, SS1: 240 degrees, SS2: 240 degrees, SS3: 240 degrees, SS4: 240 degrees, SS5: 240 degrees, SS6: 240 Degree, SS7: 240 degrees, SS8: 240 degrees can be combined.

  As described above, combinations of phases having different patterns from the phase shifter 5-2k to the phase shifter 5-C-k are given, and the minimum PAPR selector 10 selects a transmission signal having a pattern with the minimum PAPR. To the receiving device 60.

  Here, although the above is an example of equal division of a circle, a phase that divides the circle equally may be used.

FIG. 3 shows an example of the phase added by the phase shifter 5-qk of the transmission device 50. FIG. 3 shows an example in which the circle is unequally divided into three (phase division number m = 3). The phases to be added are θ _{Dqk_1} = 2π / 9, θ _{Dqk_2} = 17π / 36, and θ _{Dqk_3} = _10π / 9. Either. In this case, between the transmission device 50 and the reception device 60, three phase values: values of 2π / 9, 17π / 36 and 10π / 9 are shared as one phase sequence to be used.

The output signal S ^θ _q of the phase shifter 5-qk after the phase multiplication is expressed by equation (9) as in the conventional equation (3).

The frequency shifter 6-q-k (1 ≦ q ≦ C, 1 ≦ k ≦ N _D ) is the division filter 4-k or the phase shifter 5-q-k (2 ≦ q ≦ C, 1 ≦ k ≦ N _D ). Are shifted to a predetermined desired band on the frequency axis. For example, the frequency shifter 6-qk shifts the sub-spectrum to the unused band of the satellite repeater and distributes the signal band of the transmission signal. The frequency shifter 6-qk performs processing for each of 1 to C. For example, when q = 1, each sub-spectrum of the frequency shifter 6-1-k (1 ≦ k ≦ N _D ) is distributed and arranged in a desired band on the frequency axis. Similarly, each sub-spectrum of the frequency shifter 6-2-k, each sub-spectrum of the frequency shifter 6-3-k,..., Each sub-spectrum of the frequency shifter 6-C-k is set to 1 of the frequency shifter 6-1-k. from shifting in the same shift amount as the respective shift amounts of up to N _D. Note that the shift amount of each sub-spectrum is controlled by the control unit 51, for example.

The adder 7-q (1 ≦ q ≦ C) outputs 1 to N that the frequency shifter 6-qk (1 ≦ q ≦ C, 1 ≦ k ≦ N _D ) outputs for each group (for each same q). adding the N _D sub-spectrum of up to _D. In figure 1, the adder 7-1 adds the _{N D} sub spectrum frequency shifter _{6-1-N D} is output from the frequency shifter 6-1-1. Similarly, the adder 7-C adds _{N D} sub spectrum frequency shifter _{6-C-N D} is output from the frequency shifter 6-C-1.

  IDFT8-q (1 ≦ q ≦ C) converts the output signal of the adder 7-q from a frequency domain signal to a time domain signal. For example, the IDFT 8-1 converts the output signal of the adder 7-1 from a frequency domain signal to a time domain signal.

  The PAPR calculator 9-q (1 ≦ q ≦ C) calculates a PAPR for the signal converted into a time domain signal by the IDFT8-q. For example, the PAPR calculator 9-1 calculates a PAPR for the signal converted into a time domain signal by the IDFT 8-1. The PAPR is calculated using a well-known technique. For example, the maximum value and the average value of squared values of the signals input to the PAPR calculators 9-1 to 9-C are calculated. Then, the PAPR calculator 9-q outputs the maximum value / average value (value obtained by dividing the maximum value by the average value) as PAPR.

  The minimum PAPR selector 10 selects and transmits the output signal of the IDFT 8-q of the phase sequence having the smallest PAPR calculated by the PAPR calculator 9-q. For example, when the PAPR calculated by the C = 2 PAPR calculator 9-2 is the smallest, the minimum PAPR selector 10 selects and transmits the output signal of the IDFT 8-2 having the same phase sequence (C = 2).

  The spectrum of the input / output signal of the division filter 4-k, the spectrum of the output signal of the phase shifter 5-q-k, and the spectrum of the output signal of the adder 7-q are the same as those described with reference to FIG. It is.

In this way, in the transmission device 50 according to the present embodiment, the phase θ _{Dqk of the} phase sequence added by the phase shifter 5-q-k is a finite number of phase values, and a finite number of phase values or a finite number of phase values. Parameters and mathematical formulas for generating a phase value are shared in advance as known information between the transmission device 50 and the reception device 60. Then, the transmission device 50 transmits, as a transmission signal, the sum of the sub-spectrums using the finite number of phases of the phase sequence having the smallest PAPR among the plurality of phase sequences having the finite number of phase values to the reception device 60. Can do.

FIG. 4 shows a configuration example of the receiving device 60 using the band dispersion transmission method. 4, the receiving apparatus 60, DFT11, extraction filter 12-1 to _{12-N D,} the frequency shifter 13-1 to _{13-N D,} the phase estimator _{14-2~14-N D,} a phase corrector 15 2-15-N _D , phase shifters 16-2 to 16-N _D , adder 17, IDFT 18 and demodulator 19. Note that the receiving device 60 may include a control unit 61 that controls the operation of each unit, and a storage unit 62 that stores parameters and the like.

  The DFT 11 converts the received signal into a frequency domain signal.

Extraction filter 12-k (1 ≦ k ≦ N D) is the output signal of DFT11 which is branched to _{the N D} respectively input, extracts the _{N D} sub spectrum.

Frequency shifter 13-k (1 ≦ k ≦ N D) , the extraction filter 12-1 to _{12-N D} each sub spectrum is outputted, a frequency shifter 6-q-1~6-q- N of the transmission device 50 Shift to the band before frequency shift by _D.

Phase estimator 14-k (2 ≦ k ≦ N D) estimates the phase difference of the sub-spectrum signals frequency shifter adjacent among frequency shifters 13-1 to _{13-N D} outputs. The phase difference estimation method estimates the phase difference by calculating the correlation in the transition region where the adjacent sub-spectrum bands overlap as described with reference to FIG. Here, the estimated phase difference corresponds to the phase difference added by the phase shifter 5-q-2 of the transmission device 50. In FIG. 4, for example, the phase estimator 14-2 receives the output signals of the frequency shifter 13-1 and the frequency shifter 13-2 and adds the phase difference added by the phase shifter 5-q-2 of the transmission device 50. Is estimated. The phase estimator 14-k receives sub-spectrums (SS _k−1 and SS _k ) adjacent in the frequency domain and calculates a phase difference R _{k_ab} in the transition region of SS _k−1 and SS _k . Note that when the phase difference R _{k — ab} is converted into radians, the phase difference (^ θ _Dqk ) = atan (R _{k — ab} ) is obtained. Here, as shown in (a-1) of FIG. 10A of the prior art, in the transmission device 50, the transition areas of the sub-spectrums are overlapped with each other, so the transition areas of the adjacent sub-spectrums are the same. Have signal components. Therefore, the receiving device 60 estimates the phase difference R _{k_ab} from the signal component of the superimposed transition region. Here, k denotes the number of sub-spectrum from 1 to N _D. Note that θ _Dq1 is set to 0 as described in Expression (6). That is, since SS ₁ has 0 _Dq1 of 0, the phase is not added by the transmission device 50, and therefore the phase estimator 14-2 receives the SS ₁ and SS input to the frequency shifters 13-1 and 13-2. _2, it is possible to estimate the phase _{R 2_12,} which is added to the SS _2. Then, the phase estimator 14-2 outputs the phase difference (^ θ _Dq2 ) = atan (R 2 — _ab ) to the phase corrector 15-2. Hereinafter, from the phase estimator 14-3 _{14-N D,} similarly to the phase estimator 14-2 estimates a phase difference _{R k_ab.} Then, the phase estimator 14-k _{outputs the} phase difference (^ θ _Dqk ) = atan (R k — _ab ) to the phase corrector 15-2.

Phase corrector 15-k (2 ≦ k ≦ N D) corrects the estimated phase difference phase estimator _{14-2~14-N D} is outputted (^ θ _Dqk). For example, the phase corrector 15-2 performs rounding of the estimated phase difference using a finite number of phases shared with the transmission device 50. In the transmission device 50, any one of the m phases θ _Dqk shown in Expression (8) is added, and parameters and mathematical formulas for generating the phase θ _Dqk or the phase θ _Dqk are as _follows. In advance. Therefore, the phase corrector 15-q selects the phase closest to the phase difference (^ θ _Dqk ) input to the phase corrector 15-q among the m phases θ _Dqk added by the transmission device 50. Then output.

As a method of selecting a point closest to the phase difference (^ θ _Dqk ), for example, a method of selecting a point having the closest Euclidean distance on the complex plane can be considered. Here, the m Street phase _{θ Dqk (θ Dqk> 0)} , θ Dqk_1 in an ascending order of distance from the phase _{0 (rad), θ Dqk_2,} ..., θ Dqk_p, ..., be a finite number of θ _{Dqk_m.} In this case, as shown in equation (10), select the theta _{Dqk_p} that minimizes the evaluation function _{r p} can be output from the phase corrector 15-q.

For example, in an example of equally dividing a circle, when the number of phase divisions m = 3 and the initial phase θ _offset = 0, the phases added by the phase shifter 5-qk of the transmission device 50 as described in FIG. Is either θ _Dqk = 0, 2π / 3, or 4π / 3. In this case, the receiving device 60 performs processing as follows.

FIG. 5 shows an example of processing of the phase corrector 15-q of the receiving device 60. 5A shows an example of the input of the phase corrector 15-q (the phase difference estimated by the phase estimator 14-q), and FIG. 5B shows the input of the phase corrector 15-q. An example of output is shown. 5A and 5B correspond to the case of FIG. 2, and the receiving apparatus 60 has information that the phase division number m = 3 and the initial phase θ _offset = 0, or _Information that the phase to be added is any one of θ _Dqk = 0, 2π / 3, and 4π / 3 is shared with the transmission device 50 in advance.

In FIG. 5A, when the phase difference (^ θ _Dqk ) = π / 6 estimated by the phase estimator 14-q of the receiving device 60 (black circle in FIG. 5A), the above equation ( 10), the Euclidean distances are r ₁ = 0.51, r ₂ = 1.4, and r ₃ = 1.9. In this case, the phase corrector 15-q determines that _{r 1} is the minimum, by replacing the phase θ _{Dqk_1} = 0 for black circles shown in FIG. 5 (b) a black circle in FIG. 5 (a), the phase correction The device 15-2 outputs 0.

Here, in the example of unequal division of the circle, when the phase division number m = 3, as described in FIG. 3, the phase added by the phase shifter 5-q-k of the transmission device 50 is θ _Dqk — ₁ = _Either 2π / 9, θ _Dqk — ₂ = 17π / 36, or θ _Dqk — ₃ = 10π / 9.

FIG. 6 shows an example of processing of the phase corrector 15-q of the receiving device 60. 6A shows an example of the input of the phase corrector 15-q (phase difference estimated by the phase estimator 14-q), and FIG. 6B shows the input of the phase corrector 15-q. An example of output is shown. _{6A and 6B} correspond to the example of FIG. 3, and the receiving apparatus 60 has a phase division number m = 3 and the phase to be added is θ _Dqk — ₁ = 2π / 9, Information indicating that either θ _Dqk — ₂ = 17π / 36 or θ _Dqk — ₃ = 10π / 9 is shared with the transmission apparatus 50 in advance.

_6A, when the phase difference (^ θ _Dqk — 3) = 17π / 18 estimated by the phase estimator 14-q of the receiving device 60 (black circle in FIG. 6A), the above equation ( 10), the Euclidean distances are r ₁ = 1.8, r ₂ = 1.4, and r ₃ = 0.52. In this case, the phase corrector 15-q judges that the _{r 3} is the smallest, replacing black circle in FIGS. 6 (a) to the phase θ _{Dqk_3} = 10π / 9 of the black circle in FIG. 6 (b), the 10π / 9 is output from the phase corrector 15-2 to the phase shifter 16-2.

Phase shifter 16-k (2 ≦ k ≦ N D) , the phase (theta _DQk) respectively output from the phase corrector _{15-2~15-N D} is input, the frequency shifter 13-1 to _{13-N D} Is multiplied by exp (−j (θ _Dqk )) to compensate the phase. For example, the phase shifter 16-2 uses the θ _Dq3 output from the phase corrector 15-2 to multiply the output signal of the frequency shifter 13-2 by exp (−j (θ _Dq3 )) to compensate the phase. To do. Similarly, the phase difference _{R 3_Ab} from the output signal SS ₃ of the phase compensated signal SS ₂ and the frequency shifter 13-3 in the phase estimator 14-3 is calculated by the phase shifter 16-2, the phase corrector 15-3 The phase θ _{Dq3 subjected} to the phase correction is output to the phase shifter 16-3. Using the phase θ _Dq3 output from the phase corrector 15-3, the phase shifter 16-3 multiplies SS ₃ output from the frequency shifter 15-3 by exp (−jθ _Dq3 ) to compensate the phase. Thereafter, the phase estimator 14-k (4 ≦ k ≦ N D) is the phase difference from the SS _k output from the outputted _{SS k-1} and the frequency shifter 13-k from the phase shifter 16- (k-1) Estimate R _{k — ab} . Then, the phase θ _{Dqk subjected} to the phase correction by the phase corrector 15-k is output to the phase shifter 16-k, and the phase shifter 16-k outputs exp (−jθ) to the SS _k output from the frequency shifter 13-k. _Dqk )) is multiplied to perform phase compensation.

The adder 17, the frequency shifter 13-1 and phase shifter 16-2～ phase shifter _{16-N D} of the output signal were respectively added to restore the previous signal waveform into a plurality of sub-spectra by the transmission device 50.

  The IDFT 18 receives the output signal of the adder 17 and converts it into a time domain signal.

  The demodulator 19 demodulates the modulation signal output from the IDFT 18. For example, when the modulator 1 of the transmission device 50 modulates with QPSK, the received data is demodulated with the same QPSK.

  The spectrum of the input / output signal of the phase shifter 16-k and the correlation calculation by the phase estimator 14-k are the same as those described with reference to FIG.

  As described above, the receiving device 60 according to the present embodiment extracts the sub-spectrum divided at the time of transmission by the transmitting device 50, and the sub-spectrum distributed and arranged by the transmitting device 50 by the frequency shift processing before the distributed arrangement. Return to bandwidth. Then, the receiving device 60 estimates the phase difference in the adjacent transition region with respect to the sub-spectrum returned to the band before the distributed arrangement. At this time, since the receiving device 60 shares in advance with the transmitting device 50 information on m phases added by the transmitting device 50 or information such as parameters and mathematical formulas for generating m phases, there are m ways. The phase value closest to the phase estimation value on the receiving side is selected from among the phases, and correction for replacing the phase estimation value is performed. Then, the receiving device 60 compensates the phase of each sub-spectrum with the corrected phase estimation value, adds all the sub-spectrums, calculates the sum of the sub-spectrums, and performs demodulation processing.

As described above, in the present embodiment, the receiving apparatus 60 shares in advance with the transmitting apparatus 50 the information on the m kinds of phases added by the transmitting apparatus 50 or information about parameters and mathematical formulas for generating the m kinds of phases. Keep it. The receiving device 60 selects a value closest to the estimated phase difference between adjacent subspectrums of the received signal from the finite number of phases and replaces the estimated phase difference. Accuracy can be increased. In particular, when the number p of discretized signal components existing in a low S / N environment or in the transition region of the _subspectrum is not sufficient as the smoothing number in the frequency axis direction, the phase difference R _{k_ab} is expressed by Equation (11). ), The signal component b _{(k−1) i} on the high frequency side of the adjacent sub-spectrums SS _k−1 and SS _k is multiplied by the complex conjugate of the signal component a ^* _ki on the low frequency side, and the frequency It is obtained by smoothing in the axial direction.

Here, FIG. 12C shows an example when p = 3. At this time, the phase difference R _{k — ab} is expressed by Expression (12).

Note that the receiving device 60 uses a value (threshold α) that is mounted in advance in the receiving device 60 in order to confirm whether or not the phase is added to the subspectrum by the phase shifter 5-qk of the transmitting device 50. Use and judge. For example, when the phase difference R _{k_ab} is greater than or _equal to the threshold value α, it is considered that the phase is added to the kth sub-spectrum k in the transmission device 50, and the phase difference (^ θ _Dqk ) = atan (R _{k_ab} ) And the phase shifter 16 compensates for the phase difference of the subspectrum k. Here, since b _{(k−1) p} and a _kp are the same signal component before division, the phase difference (^ θ _Dqk ) = b _{(k−1) p} and a _kp generated in the receiving apparatus 60 is equal to It is _{assumed that} atan (R _{k — ab} ) is due to the phase added by the transmission device 50. Then, the receiving apparatus 60, the phase difference (^ θ _Dqk) using the corrected phase (theta _DQk) and compensates the phase by multiplying exp (-j (θ _Dqk)) in the sub-spectrum SS _k. On the other hand, when the phase difference R _{k_ab} is less than the threshold value α, it is determined that the transmitter 50 has not added the phase, and the phase estimator 14-k outputs phase difference (^ θ _Dqk ) = 0. In this case, the output of the phase corrector 15-k is also zero.

With the above configuration, the reception device 60 can sequentially estimate the phase difference between adjacent subspectrums and compensate for the phase added by the transmission device 50. In particular, in the present embodiment, the receiving device 60 shares in advance with the transmitting device 50 information on m phases added by the transmitting device 50 or information about parameters and mathematical formulas for generating m phases, Since the phase value closest to the phase estimation value on the receiving side is selected from the finite number of phases and the phase estimation value is replaced with the selected phase value, the accuracy of the phase estimation value can be improved as compared with the conventional example.
(Transmission process)
FIG. 7 shows an example of transmission processing in the present embodiment. 7 is executed by the transmission device 50 shown in FIG.

  In step S101, the modulator 1 modulates the data signal to be transmitted by a modulation scheme such as QPSK.

  In step S102, the waveform shaping filter 2 limits the band of the modulation signal by multiplying the modulation signal output from the modulator 1 by a predetermined filter coefficient.

In step S103, the modulation signal waveform shaping filter 2 outputs is converted into the frequency domain by DFT3, splitting filter 4-k divides the spectrum of the modulated signal to the N _D.

In step S104, the division filter 4-k stores the spectrum of the modulation signal divided into _ND pieces in an internal buffer. Incidentally, N _D sub spectrum, the phase of the different phase sequences for each set are added by dividing the C-number of pairs. In FIG. 7, the counter i (i is a positive integer) is used as an order in the loop processing from i = 1 to C. However, the phase addition of C sets is performed in parallel. Also good. Here, the counter is initialized to i = 1. The counter i corresponds to the code q of each block described in FIG. For example, the phase shifter 5-qk is rewritten as the phase shifter 5-ik, the frequency shifter 6-qk is rewritten as the frequency shifter 6-ik, and the adder 7-q is rewritten as the adder 7-i. Can do. Similarly, q corresponds to the counter i for the other blocks described below.

  If the counter i is not 1 in step S105, the process proceeds to step S106. If the counter i is 1, the process proceeds to step S107.

  In step S106, the phase shifter 5-i-k adds a predetermined phase to each output signal of the division filter 4-k.

  In step S107, the frequency shifter 6-ik shifts each sub-spectrum output from the division filter 4-i or the phase shifter 5-ik to a predetermined band on the frequency axis.

In step S108, the adder 7-i adds the _{N D} sub spectrum from 1 to frequency shifter 6-i-k outputs to _{N D.}

  In step S109, the PAPR calculator 9-i calculates the PAPR of the output signal of the adder 7-i converted from the frequency domain signal to the time domain signal by the IDFT 8-i. The calculated PAPR is held until the minimum PAPR selector 10 compares.

  If the counter i is not C in step S110, the process proceeds to step S111. If the counter i is C, the process proceeds to step S112.

  In step S111, the counter i is incremented by 1, and the processing returns to step S105, and the same processing is executed for the next set of phase sequences. In this way, the same processing is performed for the number of phase sequences.

  In step S112, the minimum PAPR selector 10 compares the PAPR calculated for each of the counters i from 1 to C, and transmits a set of signals with the minimum PAPR. For example, when the PAPR with the counter i of 3 is the minimum, the minimum PAPR selector 10 transmits a signal obtained by converting the frequency domain output signal of the adder 7-3 into a time domain signal by the IDFT 8-3.

In this way, the transmission device 50 according to the present embodiment has the phase sequence with the smallest PAPR out of a plurality of different phase sequences combined with a finite number of phases shared with the reception device 60. The signal added to the spectrum is transmitted.
(Reception processing)
FIG. 8 shows an example of reception processing in this embodiment. 8 is executed by the receiving device 60 shown in FIG.

In step S201, the received signal in the time domain is converted into a frequency domain signal by DFT11, signal branched to _{the N D} is input to the _{N D} extraction filter 12-k. Then, the extraction filter 12-k extracts the sub-spectrum distributed and arranged by the transmission device 50 from the output signal of the DFT 11.

In step S202, the frequency shifter 13-k are each sub-spectrum extraction filter 12-1 to _{12-N D} is output, the band prior to frequency shift by the frequency shifter 6-1 to _{6-N D} of the transmission device 50 Shift to.

In step S203, the phase estimator 14-k is the correlation calculated in the transition zone of the sub-spectrum that adjacent ones of the sub-spectrum frequency shifter 13-1 to _{13-N D} outputs, estimates the phase difference.

  In step S204, the phase corrector 15-k selects the phase closest to the estimated phase difference output from the phase estimator 14-k from the m phases added by the transmission device 50, and estimates the estimated phase difference. Correct.

  In step S205, the phase shifter 16-k adds the phase output from the phase corrector 15-k to the sub-spectral signal output from the frequency shifter 13-k to compensate the phase.

In step S206, the adder 17, the modulated signal before adding the respective output signals of the frequency shifter 13-1 and phaser _{16-2~16-N D,} into a plurality of sub-spectra by the transmission device 50 return.

  In step S207, the modulation signal output from the adder 17 converted into a time domain signal by the IDFT 18 is demodulated into received data by the demodulator 19.

In this way, the receiving device 60 according to the present embodiment estimates the phase difference in the transition region of the adjacent sub-spectrum with respect to the sub-spectrum returned to the band before the distributed arrangement, and between the transmitting device 50 and the sub-spectrum. The phase closest to the estimated phase value is selected from m shared phases, and the estimated phase value is corrected. The receiving apparatus 60 compensates the phase of each subspectrum with the corrected estimated phase value, calculates the sum of all subspectrums, and performs a process of demodulating received data.
(Modification)
The configurations of the transmission device 50 and the reception device 60 described in the above embodiment are merely examples, and various modifications can be considered, and similar effects can be obtained in the modifications.

  For example, in the transmission apparatus 50 shown in FIG. 1 of the above embodiment, the frequency shifter 6 is arranged between the phase shifter 5 and the adder 7, but the frequency shifter 6 is arranged between the division filter 4 and the phase shifter 5. You may arrange. In this case, the frequency shifter 6 disperses and arranges the sub-spectrums in the target band, and the phase shifter 5 adds the phase to each sub-spectrum that is dispersedly arranged.

  In the above embodiment, the m phase information added by the phase shifter 5 of the transmission device 50 is shared in advance between the transmission device 50 and the reception device 60. The apparatus 50 may transmit m kinds of phase information and information on parameters and expressions for generating m kinds of phases to the receiving apparatus 60. Alternatively, a predetermined combination of fixed phase values may be standardized and mounted in advance in the transmission device 50 and the reception device 60. Alternatively, the receiving device 60 may estimate m phases from the received signal. For example, the receiving device 60 can estimate m phases from the received signal by a method of statistically processing the phase estimation value. Or you may make it the transmission apparatus 50 transmit a training signal before transmission start.

  In the above-described embodiment, the frequency shifter 13 is disposed between the extraction filter 12 and the phase estimator 14 in the receiving device 60, but may be disposed between the phase shifter 16 and the adder 17. In this case, the receiving device 60 performs frequency shift after compensating for the phase difference added by the transmitting device 50.

  As described above, the communication system, the communication apparatus, and the communication method according to the present invention improve the phase estimation accuracy in the receiving apparatus 60, and are discrete in a low S / N environment or in a sub-spectrum transition region. It is possible to improve the BER characteristic of a received signal that deteriorates due to an insufficient number p of signal components.

  Note that the method described in the above embodiment is applicable if communication is performed via a propagation path having frequency selective phase fluctuation. In other words, a channel estimation unit for estimating frequency-selective phase fluctuations that occur in the propagation path and a channel correction unit for compensating for phase fluctuations are provided in the receiver, and parameters for obtaining the phase or phase of the transmission signal between the transmission side and the reception side By sharing the formulas and equations, it becomes easy to estimate and compensate for phase fluctuations in the propagation path of the received signal.

50, 150 ... transmitting device; 60, 160 ... receiving device; 1, 101 ... modulator; 2, 102 ... waveform shaping filter; 3, 11, 103, 111 ... DFT; 104, 105, 115 ... phase shifter; 6, 13, 106, 113 ... frequency shifter; 7, 17, 107, 116 ... adder; 108, 117 ... IDFT; 9, 109 ... PAPR calculator; 10, 110 ... minimum PAPR selector; 12, 112 ... extraction filter; 14, 114 ... phase estimator; ... Phase corrector; 19, 118 ... Demodulator; 51, 61, 151, 161 ... Control unit; 52, 62, 152 ... Storage unit

Claims (8)

A communication method in which N (N is an integer of 2 or more) sub-spectrums obtained by dividing a modulation signal in a frequency domain are distributed on the frequency axis and transmitted from a transmission device to a reception device,
The transmitter is
The process of rotating the phase of each of the N sub-spectrums before or after distributed arrangement is performed for each of a plurality of phase sequences having different combinations of a finite number of phases including the case where no phase addition is performed,
A peak-to-average power ratio, which is a ratio between the maximum power and average power of the transmission signal for each phase sequence, is obtained, and the transmission signal of the phase sequence that minimizes the peak-to-average power ratio is selected to the receiving device. Send
The receiving device is:
Estimate the phase difference between sub-spectrums whose frequency bands are adjacent to the sub-spectrum after dispersion placement extracted from the received signal or the sub-spectrum returned before dispersion placement,
Of the finite number of phases shared in advance with the transmission device, select the phase closest to the estimated phase difference and replace it with the estimated phase difference for correction,
A communication method, comprising: correcting a phase of a sub-spectrum after dispersion arrangement extracted from a received signal or a sub-spectrum returned before dispersion arrangement based on the corrected phase difference, and demodulating the reception signal. The communication method according to claim 1,
The phase sequence is generated by m phases obtained by equally or unequally dividing a circle into m (m is a positive integer).
The receiving apparatus estimates a phase difference between adjacent sub-spectrums before or after distributed arrangement extracted from a received signal, and points on the estimated phase difference circle and the m phase circles. Calculating a Euclidean distance from each of the above points, selecting the phase having the smallest Euclidean distance from among the m phases, replacing the estimated phase difference, and correcting the selected phase difference. Method. A modulator for modulating transmission data;
A division unit that divides the modulation signal output by the modulation unit into N (N is an integer of 2 or more) sub-spectra in the frequency domain;
A first frequency transition unit for distributing and arranging the N sub-spectrums on the frequency axis;
For each of a plurality of phase sequences having different combinations of a finite number of phases, including the case where the first frequency transition unit rotates the phase of each of the N sub-spectrums before or after distributed arrangement, including no phase addition. A first phase portion to be
The transmission signal of the phase sequence that minimizes the peak-to-average power ratio, which is the ratio of the maximum power and the average power of the transmission signal obtained by adding the sub-spectrum after dispersion arrangement for each phase sequence, is selected by the receiving device. A transmission device comprising: a selection unit for transmission;
An extractor for extracting sub-spectrums distributed from the received signal;
A second frequency transition unit for returning the dispersed sub-spectrum to the frequency band before the dispersed arrangement;
A phase estimation unit that estimates a phase difference between sub-spectrums adjacent to each other in a frequency band with respect to a sub-spectrum after dispersion arrangement extracted by the extraction unit or before dispersion arrangement restored by the second frequency transition unit;
A phase correction unit that selects a phase closest to the phase difference estimated by the phase estimation unit from a finite number of phases shared in advance with the transmission device and corrects the phase difference by the estimated phase difference; and
A second phase unit that corrects the phase of the sub-spectrum after dispersion arrangement extracted from the received signal or the sub-spectrum returned before dispersion arrangement based on the phase difference corrected by the phase correction unit;
And a demodulator that demodulates received data by adding N sub-spectrums corrected by the second phase unit. The communication system according to claim 3,
The phase sequence is generated by m phases obtained by equally or unequally dividing a circle into m (m is a positive integer).
The phase correction unit calculates a Euclidean distance between a point on the circle of the phase difference estimated by the phase estimation unit and each point on the circle of m phases, and among the m phases, the phase correction unit calculates the Euclidean distance. A communication system, wherein the phase that minimizes the Euclidean distance is selected and replaced with the estimated phase difference for correction. A modulator for modulating transmission data;
A division unit that divides the modulation signal output by the modulation unit into N (N is an integer of 2 or more) sub-spectra in the frequency domain;
A first frequency transition unit for distributing and arranging the N sub-spectrums on the frequency axis;
For each of a plurality of phase sequences having different combinations of a finite number of phases, including the case where the first frequency transition unit rotates the phase of each of the N sub-spectrums before or after distributed arrangement, including no phase addition. A first phase portion to be
The transmission signal of the phase sequence that minimizes the peak-to-average power ratio, which is the ratio of the maximum power and the average power of the transmission signal obtained by adding the sub-spectrum after dispersion arrangement for each phase sequence, is selected by the receiving device. And a selection unit for transmission. The communication device according to claim 5, wherein
The finite number of phases are generated by m phases obtained by equally or non-equally dividing a circle into m (m is a positive integer).
The plurality of phase sequences have different combinations of the finite number of phases. An extraction unit that extracts sub-spectrums arranged in a distributed manner in a signal received from the transmission device;
A second frequency transition unit for returning the dispersed sub-spectrum to the frequency band before the dispersed arrangement;
A phase estimation unit that estimates a phase difference between sub-spectrums adjacent to each other in a frequency band with respect to a sub-spectrum after dispersion arrangement extracted by the extraction unit or before dispersion arrangement restored by the second frequency transition unit;
A phase correction unit that selects a phase closest to the phase difference estimated by the phase estimation unit from among a finite number of phases obtained from information shared in advance with the transmission device, and corrects the phase difference by replacing it with the estimated phase difference; ,
A second phase unit that corrects the phase of the sub-spectrum after dispersion arrangement extracted from the received signal or the sub-spectrum returned before dispersion arrangement based on the phase difference corrected by the phase correction unit;
And a demodulator that demodulates received data by adding N sub-spectrums corrected by the second phase unit. The communication device according to claim 7.
The finite number of phases are generated by m phases obtained by equally or non-equally dividing a circle into m (m is a positive integer).
The phase correction unit calculates a Euclidean distance between a point on the circle of the phase difference estimated by the phase estimation unit and each point on the circle of m phases, and among the m phases, the phase correction unit calculates the Euclidean distance. The communication apparatus, wherein the phase that minimizes the Euclidean distance is selected and replaced with the estimated phase difference for correction.

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