JP2016166770A - Electric leak detection device - Google Patents

Abstract

PROBLEM TO BE SOLVED: To perform quick electric leak detection even when a coupling capacitor used for detecting electric leak of a high voltage power source is charged/discharged.SOLUTION: An electric leak detection device comprises: a signal generation unit generating a periodic reference signal; a coupling capacitor connecting the signal generation unit to a high voltage power source; a first resistance element connected between the coupling capacitor and the signal generation unit; and an electric leak resistance detection unit to which a voltage signal generated at a junction between the coupling capacitor and the first resistance element is supplied. The electric leak resistance detection unit includes: a determination circuit outputting a determination signal that validates a part of the voltage signal generated at the junction between the coupling capacitor and the first resistance element, falling within a preset determination threshold range, and invalidates the remaining part; and an electric leak ohmic value output circuit detecting an amplitude of a detection target reference signal in a period which the determination signal determines as valid, and converting the amplitude into an electric leak ohmic value and output it.SELECTED DRAWING: Figure 1

Description

  The present invention relates to a leakage detection device.

  The high voltage power supply is electrically insulated from the ground. If the high-voltage power supply and the ground are electrically connected for some reason, a current flows from the high-voltage power supply to the ground, causing leakage.

  In order to detect this leakage, a voltage of the coupling capacitor is detected by supplying a predetermined input pulse signal to the high voltage side terminal or the low voltage side terminal of the high voltage power supply via the coupling capacitor. If there is no leakage, a response pulse signal having the same voltage amplitude as that of the input pulse signal is detected. If there is leakage, a response pulse having a voltage amplitude smaller than the voltage amplitude of the input pulse signal according to the magnitude of the leakage resistance. A signal is detected.

  In the leakage detection device of Patent Document 1, it is noted that leakage detection is performed by comparing peak values of pulse signals, and it is stated that the leakage signal detection time is shortened by reducing the fall time of the pulse signal.

  Patent Document 2 states that the waveform of the response pulse signal from the coupling capacitor changes according to the magnitude of the leakage resistance, and describes that the leakage resistance is obtained at the transition time between the two detection threshold voltages. Yes. Here, pulse signals for slow charge and rapid discharge are used.

  As a matter related to the present invention, Patent Document 2 is in an appropriate range in which leakage can be detected by charging / discharging the coupling capacitor when the leakage equivalent resistance is short-circuited and the voltage at the leakage resistance measurement point fluctuating greatly. It describes that it is necessary to wait until it is out of the range of (0 to Vcc) and pulled back to the proper range by the bypass diode. Even in this case, Patent Document 2 states that rapid discharge is performed, so that the time for returning to an appropriate range can be shortened.

JP 2012-168072 A JP 2013-195136 A

  It is desirable to detect leakage quickly even when charging / discharging occurs in a coupling capacitor used for detecting leakage in a high-voltage power supply.

  A leakage detecting device according to the present invention is connected between a signal generator that generates a periodic reference signal, a coupling capacitor that connects the signal generator to a high-voltage power source, and the coupling capacitor and the signal generator. And a leakage resistance detection unit to which a voltage signal generated at a connection point between the coupling capacitor and the first resistance element is supplied. The leakage resistance detection unit includes the coupling capacitor and the first resistance element. Of the voltage signal generated at the connection point with the resistance element, a determination circuit that outputs a determination signal that determines that a portion within a predetermined determination threshold range is valid and the others are invalid, and the determination signal is valid A leakage resistance value output circuit that detects the amplitude of the detected reference signal during the determination period, converts the detected reference signal into a leakage resistance value, and outputs the leakage resistance value.

  According to the said structure, even if charging / discharging arises in the coupling capacitor used for the leakage detection of a high voltage power supply, a leakage detection can be performed rapidly.

It is a block diagram which shows the structure of the leak detection apparatus of embodiment which concerns on this invention. It is a figure which shows the flow of a charging current when charge arises in the coupling capacitor in the leak detection apparatus of embodiment which concerns on this invention. It is a figure which shows the flow of the discharge current when discharge arises in the coupling capacitor in the leak detection apparatus of embodiment which concerns on this invention. It is a circuit diagram of the determination circuit in the leakage detection apparatus of the embodiment according to the present invention. It is a figure which shows the time change of the to-be-detected reference signal in P1 of FIG. It is a figure which shows the time change of the to-be-detected reference signal in P2 of FIG. It is a figure which shows the time change of the to-be-detected reference signal in P3 of FIG. It is a figure which shows the time change of the to-be-detected reference signal in P4 of FIG. It is a figure which shows the time change of the determination signal in P5 of FIG. It is a figure which shows the to-be-detected reference signal for detecting an amplitude in the earth-leakage resistance value output circuit of FIG. It is a figure which shows the leak detection apparatus of another structure.

  Embodiments according to the present invention will be described below in detail with reference to the drawings. The voltage value, frequency, resistance value, capacitance value, time, and the like described below are examples for explanation, and can be appropriately changed according to the specification of the leakage detection device. Below, the same code | symbol is attached | subjected to the same element in all the drawings, and the overlapping description is abbreviate | omitted.

FIG. 1 is a block diagram showing the configuration of the leakage detection device 10. In FIG. 1, although not a constituent element of the leakage detection device 10, the high voltage power supply 6 to be detected, the first leakage resistance 8 on the high voltage V _H side of the high voltage power supply 6, and the low voltage of the high voltage power supply 6 are illustrated. The second leakage resistance 9 on the voltage _VL side is illustrated. The leakage detection device 10 is a device that detects and outputs the magnitude of the first leakage resistance 8 or the second leakage resistance 9.

The high-voltage power supply 6 is a high-voltage and large-capacity power storage device that combines a plurality of secondary batteries to obtain a desired high-voltage and high-power. An example of the high voltage is about 300V to about 600V. Hereinafter, the high voltage power supply 6 of 250V will be described. 250 V is a difference voltage between the high voltage V _H and the low voltage V _L of the high voltage power supply 6. In other words, the high voltage V _H is 250 V higher than the low voltage V _L.

The high voltage power supply 6 is insulated from the outside in order to prevent electric shock. The housing of the high voltage power supply 6 is floating from both the high voltage V _H side and the low voltage V _L side of the high voltage power supply 6. The leakage resistance value R _L1 of the first leakage resistance 8 is obtained by converting the leakage current to the leakage resistance value when the high voltage V _H side of the high voltage power supply 6 leaks between the casing for some reason. Similarly, the leakage resistance value R _L2 of the second leakage resistance 9 is obtained by converting the leakage current into a resistance value when the low voltage V _H side of the high voltage power supply 6 leaks between the casing for some reason. .

  The leakage detection device 10 includes, as a basic configuration, a signal generation unit 12 that generates a periodic reference signal, a coupling capacitor 16 that connects the signal generation unit 12 to the high voltage power supply 6, a coupling capacitor 16, and a signal generation unit. The first resistance element 18 connected between the first and second resistors 12 and the leakage resistance for obtaining a leakage resistance value from the amplitude of the voltage signal obtained at a connection point between the coupling capacitor 16 and the first resistance element. A detection unit 14 is included. Further, as a protection against a high-voltage leakage current from the high-voltage power supply 6, a set of bypass diodes 31 and 32 provided on the output side of the signal generator 12 and a set of inputs provided on the input side of the leakage resistance detection unit 14. Bypass diodes 33 and 34 are included.

  In the leakage detection device 10 according to the present invention, as pointed out in Patent Document 2, when the leakage resistance is equivalently short-circuited, the coupling capacitor is charged and discharged, and the leakage resistance measurement point To detect leakage in a short time for the problem that the voltage fluctuates greatly and it is out of the range (0 to Vcc) that is the appropriate range where leakage detection is possible, and it is necessary to wait until it is pulled back to the appropriate range by the bypass diode. Make it possible.

Therefore, charge / discharge occurring in the coupling capacitor 16 will be described before describing details of the configuration of the leakage detection device 10. The charging / discharging current flows through the coupling capacitor 16 when the leakage resistance, which is the insulation resistance of the high voltage power supply 6, changes due to an external factor. The leakage resistance of the high voltage power supply 6, for example, the first leakage resistance 8 This is the case when the leakage resistance value R _{L1 of} the second leakage resistance 9 or the leakage resistance value R _L2 of the second leakage resistance 9 is greatly reduced and the first leakage resistance 8 or the second leakage resistance 9 of the high-voltage power supply 6 is equivalently short-circuited. Below, the case where the 1st earth leakage resistance 8 or the 2nd earth leakage resistance 9 equivalently short-circuits is described.

FIG. 2 is a diagram illustrating a path 80 of a charging current that flows through the coupling capacitor 16 when the first leakage resistor 8 of the high-voltage power supply 6 is equivalently short-circuited. When the first leakage resistance 8 of the high voltage power supply 6 is equivalently short-circuited and the high voltage V _H of the high voltage power supply 6 becomes the ground voltage (GND), the relatively low voltage V _L of the high voltage power supply 6 is V The voltage is 250 V lower than _H (−250 V). The voltage on the high voltage power supply 6 side which is one end of the coupling capacitor 16 is also (−250 V). Since the other end of the coupling capacitor 16 is grounded via the second resistance element 20 and the bypass diode 34, it is on the higher voltage side than (−250V). Accordingly, the ground (GND), the bypass diode 34, the second resistance element 20, the coupling capacitor 16, the low voltage _VL side of the high voltage power supply 6, the high voltage power supply 6 to the high voltage _VH side of the high voltage power supply 6, and the second. A current flows through a path 80 of 1 earth leakage resistance 8 to ground (GND). This current charges the coupling capacitor 16 from the (−250 V) side to the 0 V side.

  This charging is performed under a charging characteristic having a time constant determined by the capacitance value C of the coupling capacitor 16 and the resistance value R2 of the second resistance element 20. When actual circuit constants are applied, it takes about 20 s to about 60 s to complete charging. During the transition period, the voltage value at the input point P2 of the leakage resistance detection unit 14, that is, the voltage value generated at the connection point P1 between the coupling capacitor 16 and the first resistance element 18, is not constant. The detection unit 14 cannot detect the amplitude of the reference signal and cannot perform leakage detection.

FIG. 3 is a diagram showing a path 82 of the charging current flowing through the coupling capacitor 16 when the second leakage resistance 9 of the high voltage power supply 6 is equivalently short-circuited. When the second leakage resistance 9 of the high voltage power supply 6 is equivalently short-circuited and the low voltage _VL of the high voltage power supply 6 becomes the ground voltage (GND), the high voltage power supply 6 side, which is one end of the coupling capacitor 16, is connected. The voltage is 0V. Since the other end of the coupling capacitor 16 is connected to the circuit power supply (Vcc) via the second resistance element 20 and the bypass diode 33, it is on the higher voltage side than 0V. Therefore, the ground (GND), the second earth leakage resistance 9, the coupling capacitor 16, the second resistance element 20, the bypass diode 33, the circuit power supply (Vcc), and the circuit load of the signal generator 12 and the earth leakage resistance detector 14 are obtained. Current flows through a ground (GND) path 82. By this current, the coupling capacitor 16 is discharged from the circuit power supply (Vcc) side to the 0V side.

  Similar to the charging described with reference to FIG. 2, this discharging takes about 20 s to about 60 s to complete the discharging. During the transition period, the voltage value at the input point P2 of the leakage resistance detection unit 14, that is, the voltage value generated at the connection point P1 between the coupling capacitor 16 and the first resistance element 18, is not constant. The detection unit 14 cannot detect the amplitude of the reference signal and cannot perform leakage detection.

  As described above, when the coupling capacitor 16 is charged / discharged, the leakage resistance value cannot be accurately detected during the transition time of about 20 s to about 60 s in the related art.

Returning to FIG. 1 again, the detailed configuration of the leakage detection device 10 of the present invention will be described. The coupling capacitor 16 has one end connected to the low voltage _VL side terminal of the high voltage power supply 6, and the other end P1 is connected to the signal generator 12 via the first resistance element 18, and The leakage resistance detection unit 14 is connected via the two-resistance element 20.

The first resistance element 18 divides the voltage of the reference signal generated from the signal generator 12 by the first leakage resistance 8 or the second leakage resistance 9. The first resistance element 18, together with a set of bypass diodes 31 and 32 provided on the output side of the signal generation unit 12, protects the signal generation unit 12 against high voltage leakage current from the high voltage power supply 6 side. It becomes resistance. By using the first resistance element 18, the voltage value at the connection point P <b> 1 between the coupling capacitor 16 and the first resistance element 18 is the resistance value R <b> 1 of the first resistance element 18 and the leakage resistance value of the first leakage resistance 8. It becomes a value divided by R _L1 or the leakage resistance value R _L2 of the second leakage resistance 9. For example, if the charging current flows to the coupling capacitor 16, the amplitude voltage of the reference signal of the signal generator 12 and V _1, the voltage value of the amplitude voltage (P1 of the detected reference signal detected by the leakage resistance detecting section 14 ) = [{R1 / (R1 + R _L1 )} × V ₁ ]. As described above, the first resistance element 18 functions as a divided resistance in the leakage resistance detection.

  The second resistance element 20, together with a set of bypass diodes 33 and 34 provided on the input side of the leakage resistance detection unit 14, protects the leakage resistance detection unit 14 against high voltage leakage current from the high voltage power supply 6 side. The protective resistance to be performed. As shown in FIGS. 2 and 3, when charging / discharging occurs in the coupling capacitor 16, the resistance value R <b> 2 of the second resistance element 20 together with the capacitance value C of the coupling capacitor 16 is a charging / discharging time constant. It works as a charge / discharge resistance that determines

  The two sets of bypass diodes 31, 32, 33, and 34 are protection diodes that allow a high-voltage leakage current from the high-voltage power supply 6 to flow to the GND or Vcc side with a bypass voltage margin. For example, if Vcc = + 12V, GND = 0V, and bypass voltage = 0.6V, even if charging / discharging occurs in the coupling capacitor 16, the voltage of P2 is limited at the lower limit (-0.6V). The upper limit is limited to (+ 12.6V), and the voltage of (± 250V) does not hang on P2.

  The signal generator 12 includes an oscillator 42 built in the microprocessor 40 and a PWM (Pulse Wide Modulation) circuit 44. The signal generator 12 is further provided outside the microprocessor 40, and is denoted by a low-pass filter 46 denoted by LPF and HPF. A high pass filter 48 and an offset circuit 50 are included.

  The PWM circuit 44 generates a PWM signal having a predetermined period. The PWM signal output from the PWM circuit 44 has a low level (L) of 0V and a high level (H) of + 5V. The low-pass filter 46 converts the PWM signal output from the PWM circuit 44 into a sine wave signal of (+ 2.5V ± 2.5V). For example, the frequency of the sine wave signal is 2.5 Hz. The transition times 20 s to 60 s described in FIGS. 2 and 3 correspond to 50 to 150 cycles of the sine wave signal.

  The high-pass filter 48 removes a DC bias voltage (+2.5 V) from a sine wave signal having a frequency of 2.5 Hz at (+2.5 V ± 2.5 V), and a sine wave having a frequency of 2.5 Hz at (0 V ± 2.5 V). Signal.

  The offset circuit 50 is a sine wave having a voltage amplitude of ± 2.5 V and a frequency of 2.5 Hz with a sine wave signal having a frequency of 2.5 Hz at (0V ± 2.5V) and an arbitrarily set DC offset voltage as a center voltage. Convert to wave signal. The DC offset voltage is determined in consideration of the detection voltage range in the leakage resistance detection unit 14. In the case of Vcc = + 12V and GND = 0V, a sine wave signal that fits in this range with a margin is preferable. As an example, it is + 3.3V. In this case, the waveform is (+ 3.3V ± 2.5V), and the voltage range is (+ 5.8V to + 0.8V). In the following, a sine wave signal of (+ 3.3V ± 2.5V) and a frequency of 2.5 Hz is supplied to the coupling capacitor 16. That is, the signal generator 12 generates a sine wave signal with a constant frequency of 2.5 Hz as a reference signal. This reference signal is supplied to the coupling capacitor 16 via the first resistance element 18.

The voltage amplitude of the reference signal supplied to the coupling capacitor 16 changes according to the magnitude of the first leakage resistance 8 or the second leakage resistance 9. The change in voltage amplitude is due to the divided resistance function of the first resistance element 18. Rewriting (P1 voltage value) = [{R1 / (R1 + R _L1 )} × V ₁ ], which is an expression of the divided resistance function of the first resistance element 18, (voltage amplitude of the detected reference signal) = [{R1 / (R1 + R _L1 )} × (reference signal voltage amplitude)]. (Voltage amplitude of the reference signal) is ± 2.5V. Since the resistance value R1 of the first resistance element 18 is known, the leakage resistance value R _L1 of the first leakage resistance 8 can be obtained by detecting (the voltage amplitude of the detected reference signal).

  Since the leakage detection device 10 is intended for charging / discharging of the coupling capacitor 16, (voltage amplitude of the reference signal) is superimposed on a DC voltage component having charging / discharging characteristics determined by a charging / discharging time constant. It is necessary to perform a process of extracting (voltage amplitude of the reference signal) from the superimposed voltage signal. Furthermore, in the case of charging / discharging, it is limited in the range of (−0.6V to + 12.6V) by the bypass diodes 33 and 34, the upper limit is suppressed to (+ 12.6V), or the lower limit is (−0.6V). An accurate detected reference signal can be acquired at an early stage by performing processing to exclude a signal that is cut and becomes a detected reference signal having an incomplete amplitude.

The leakage resistance detection unit 14 includes a noise filter 60, a high-pass filter 62 indicated by HPF, and an offset circuit 64. In addition to this processing route, the leakage resistance detection unit 14 includes a determination circuit 66 for a detected reference signal that has passed through the noise filter 60. Including. Furthermore, the leakage resistance detection unit 14 acquires the output signal of the offset circuit 64 and the output signal of the determination circuit 66, and internally includes an analog-digital converter 70, a memory 72, a V _PP detection circuit 74, and an R _L conversion unit that are indicated as ADCs. And a microprocessor 40.

  The noise filter 60 is a filter that passes a detected reference signal having the same frequency as the reference signal supplied from the signal generator 12 among the voltage signal of the coupling capacitor 16.

  The high-pass filter 62 is a filter that removes a DC voltage component having charge / discharge characteristics determined by a charge / discharge time constant from a voltage signal output from the noise filter 60 and extracts a detected reference signal of an AC component. For example, assuming that the voltage amplitude reduced by the first leakage resistance 8 or the second leakage resistance 9 is ± 1.1 V, the output of the high-pass filter 62 is (0V ± 1), excluding the influence of the bypass diodes 33 and 34. .1V) and a sine wave detected reference signal having a frequency of 2.5 Hz.

  The offset circuit 64 performs a process of adding a DC offset voltage suitable for the signal processing of the microprocessor 40 to the detected reference signal output from the high-pass filter 62 (0V ± 1.1V) and having a frequency of 2.5 Hz. . The DC offset voltage is determined in consideration of the detection voltage range in the leakage resistance detection unit 14. In the case of Vcc = + 12V and GND = 0V, a sine wave signal that fits in this range with a margin is preferable. As an example, it is + 2.5V. In this case, the waveform is (+ 2.5V ± 1.1V), and the voltage range is (+ 3.6V to + 1.4V).

  The determination circuit 66 determines whether or not the signal waveform of the detected reference signal that is superimposed on the DC voltage component having charge / discharge characteristics through the noise filter 60 is within a predetermined determination threshold range. This is a circuit that outputs a determination signal that validates the portion within the threshold range and invalidates the other portion. For example, a high level (H) indicating that the signal waveform portion is within the determination threshold range is output, and a low level (L) indicating that the other signal waveform portion is invalid is output. .

  The determination circuit 66 selects only a signal waveform that falls within the measurable range of the microprocessor 40 from the detected reference signals that have passed through the noise filter 60 as an effective portion for measurement. The microprocessor 40 is used for leakage detection using only the detected reference signal in which the selected effective portion is continuous for one cycle of the signal waveform.

  The determination threshold range can be arbitrarily set within the upper and lower limits of the operating voltage range of the microprocessor 40 which is the leakage resistance detection unit 14. For example, a voltage range narrowed by a predetermined margin voltage for each of the upper and lower limits of the operating voltage range of the microprocessor 40 can be set as the determination threshold range. Assuming that the operating voltage range of the microprocessor 40 is Vcc = + 12.0V to GND = 0V, a range from (0V + lower limit margin voltage) to (+ 12.0V−upper limit margin voltage) can be set as the determination threshold range. When the lower limit margin voltage = the upper limit margin voltage = 0.2V, the voltage range from (0V + 0.2V) = + 0.2V to (+ 12.0V−0.2V) = + 11.8V is the determination threshold range.

  When the determination threshold range is set, a portion fixed to the bypass voltage by the bypass diodes 33 and 34 is not included. This is because if the portion fixed to the bypass voltage, which is the conduction voltage, is validated and input to the microprocessor 40, it is erroneously determined that the voltage amplitude value of the detected reference signal = 0. In the configuration of FIG. 1, the power supply voltage and GND of the bypass diodes 33 and 34 are the same as the power supply voltage and GND of the microprocessor 40. Therefore, in the case of charge / discharge, the detected reference signal is limited by the bypass diodes 33 and 34 in the range of (−0.6 V to +12.6 V), the upper limit is suppressed to (+12.6 V), or the lower limit is ( -0.6V). Both the upper limit (+12.6 V) and the lower limit (−0.6 V) are outside the operating range of the microprocessor 40. Therefore, in the case of the configuration of FIG. 1, by setting the determination threshold range narrower than the operating voltage range of the microprocessor 40, the portion fixed to the bypass voltage by the bypass diodes 33 and 34 can be excluded.

FIG. 4 is an example of a circuit diagram of the determination circuit 66. In the determination circuit 66, a dividing resistor is provided between Vcc and GND, and the Vcc side determination threshold value _VthH and the GND side determination threshold value _VthL are generated. (Vcc−V _thH ) corresponds to the upper limit margin voltage, and (V _{thL −GND} ) corresponds to the lower limit margin voltage. For example, assuming that Vcc = + 12.0V and GND = 0V, and the upper limit margin voltage and the lower limit margin voltage are 0.2V, V _thH = (Vcc−margin voltage) = + 11.8V, V _thL = + 0 .2V.

The determination circuit 66 has two comparators 84 and 86, and compares the detected reference signal output from the noise filter 60 with V _thH and compares the detected reference signal output from the noise filter 60 with V _thL. The respective outputs are input to the OR circuit 88. The output of the OR circuit 88 is a determination signal. Of the signal waveform of the detected reference signal, a portion where the determination signal is high level (H) is a portion determined to be valid, and a portion where the determination signal is low level (L) is a portion determined to be invalid. If it is determined that the signal waveform is valid over one cycle of the signal waveform, the detected reference signal has no missing signal waveform in the voltage range of (+11.8 V to +0.2 V), and the signal waveform of the one cycle. Thus, the voltage amplitude of the detected reference signal can be measured.

  Even if the detected reference signal determined to be valid for one period is only a signal waveform for one period, the leakage resistance value can be obtained by detecting the voltage amplitude. If there are a plurality of detected reference signals that are valid over one period, the leakage resistance value obtained based on the voltage amplitude is further reliable. Preferably, the leakage resistance value is obtained within 10 cycles of the detected reference signal that is valid over one cycle. The detected reference signal of 10 cycles is 400 ms × 10 = 4,000 ms = 4 s. Compared with the necessity of waiting for about 20 s to 60 s in the prior art, the leakage resistance value can be obtained accurately and rapidly. If possible, it is preferable to obtain the leakage resistance value within 5 of the detected reference signals that are valid over one period. In that case, it takes about 2 seconds to obtain the leakage resistance value.

  In the microprocessor 40, the output of the determination circuit 66 is ANDed with respect to the detected reference signal continuously output from the offset circuit 64. The portion invalidated in the signal waveform of the detected reference signal input from the offset circuit 64 by the AND operation becomes the low level (L), but when the low level (L) comes out, the previous operation is reset. I do. Thereby, the calculation for the subsequent leakage detection value output can be continued only while the valid high level (H) continues. When the valid high level (H) is continuous, the output of the AND operation is a continuous complete signal waveform with no chipping over one period. Using these, the leakage resistance value is obtained.

The analog-to-digital converter 70 converts each analog signal waveform of the detected reference signal obtained as a result of the AND operation into a signal waveform having a discrete digital value by sampling. The memory 72 stores a discrete digital value signal waveform output from the analog-to-digital converter 70. The V _PP detection circuit 74 detects the peak value of each signal waveform stored in the memory 72. The detected peak value is a value corresponding to the voltage amplitude of the detected reference signal.

The R _L conversion unit 76 calculates the leakage resistance of the first leakage resistance 8 based on the relational expression of (voltage amplitude of the detected reference signal) = [{R1 / (R1 + R _L1 )} × (voltage amplitude of the reference signal)]. The value R _L1 or the second leakage resistance value R _L2 is calculated and output. The R _L conversion unit 76 may execute a calculation operation process. The relational expression is previously mapped or made into a lookup table, and the output value of the V _PP detection circuit 74 is input to obtain the corresponding leakage resistance value. It may be output. When using a plurality of V _PP values, the average value is input to the R _L conversion unit 76.

  As described above, according to the leakage detection device 10, since the determination circuit 66 is used, even when the coupling capacitor 16 is charged / discharged, it is possible to output the leakage resistance value more quickly than in the prior art.

  The operation of the leakage detection device 10 having the above configuration will be described in more detail with reference to FIGS. FIGS. 5 to 9 are diagrams showing the results of calculating the time transition of the signal waveforms at P1, P2, P3, P4 and P5 in FIG. 1 by simulation. In each figure, the horizontal axis represents time t, and the vertical axis represents the voltage value V. In these drawings, it is assumed that a charging current flows through the coupling capacitor 16. When the discharge current flows through the coupling capacitor 16, the time transition tendency of the signal waveform is the same, but the voltage value and the like are different, so the difference will be described as appropriate.

FIG. 5 is a diagram showing the time transition of the voltage value of the signal waveform at P1 in FIG. As described in FIG. 2, when the coupling capacitor 16 is charged, the voltage value on the low voltage _VL side of the high voltage power supply 6 is about (−250 V) with respect to GND. On the other hand, the reference signal (3.3V ± 2.5V) is supplied from the signal generator 12 to the coupling capacitor 16.

Therefore, the charging characteristic based on the charging time constant determined by the capacitance value C of the coupling capacitor 16 and the resistance value R2 of the second resistance element 20 is a DC component from (−250V) to (+ 3.3V). Thereto, an AC signal (± 2.5V) from the signal generator 12 is correspondingly only a small amplitude which is determined by a division ratio of the resistance value R ₁ of the leakage resistance R _L1 of the first leak resistance 8 first resistive element 18 The AC signal is superimposed.

  FIG. 5 (a) shows the charging characteristics from one scale on the vertical axis to 50V and from (−250V) to (+ 3.3V). In the example of FIG. 5A, it takes about 30 s until the voltage value is stabilized at P1. In addition, when discharge occurs in the coupling capacitor 16, the discharge characteristic is directed from (+ 250V) to (+ 3.3V).

  FIG.5 (b) is the figure which expanded 1 scale of the vertical axis | shaft to 2V about a part of (a). An AC signal of about (± 1.1V) is superimposed on the charging characteristic from (−250V) to (+ 3.3V). The AC signal of about (± 1.1V) is a detected reference signal in the coupling capacitor 16. The leakage resistance value can be obtained from the voltage amplitude (± 2.5V) of the reference signal supplied from the signal generator 12 and the voltage amplitude (± 1.1V) of the detected reference signal at P1, but for this reason, To wait about 30 s until charging is complete.

  FIG.5 (c) is the figure which expanded the horizontal axis about a part of (b). One scale on the vertical axis is 2 V, the same as (b). Although the scale of the time axis is omitted, the frequency of the detected reference signal is 2.5 Hz, which is the same as the frequency of the reference signal.

  FIG. 6 is a diagram showing the time transition of the voltage value of the signal waveform at P2 in FIG. P 2 is the cathode side of the bypass diode 34. The voltage component of the signal waveform at P1 on the minus side of the bypass voltage of the bypass diode 34 (about −0.6 V) is grounded to GND through the bypass diode 34. Therefore, the voltage value of the signal waveform of P2 is obtained by clamping the voltage value of the signal waveform of P1 at about (−0.6V), and a value on the minus side does not appear from about (−0.6V).

  The vertical and horizontal axes in FIGS. 6A, 6B, and 6C are the same as the vertical and horizontal axes in FIGS. 5A, 5B, and 5C, respectively. When these are compared, the minus side of the signal waveform at P2 is fixed at about (−0.6 V). When a discharge occurs in the coupling capacitor 16, the plus side of the signal waveform at P2 is fixed at about (+ 12.6V).

  FIG. 7 is a diagram showing the time transition of the voltage value of the signal waveform at P3 in FIG. P3 is a position after the signal waveform of P2 has passed through the noise filter 60 and the high-pass filter 62. The noise filter 60 passes only the detected reference signal having the same frequency as the reference signal, but no noise component is added in this simulation. Therefore, the voltage value of the signal waveform of P3 is obtained by cutting the DC voltage component of the charging characteristics from the signal waveform of P2.

  The vertical and horizontal axes in FIGS. 7A, 7B, and 7C are the same as the vertical and horizontal axes in FIGS. 6A, 6B, and 6C, respectively. In FIG. 6, a value obtained by superimposing the detected reference signal on the DC voltage component of the charging characteristic is about (−0.6 V) which is a bypass voltage, and the minus side is cut off. In FIG. 7, the portion cut by the bypass voltage in FIG. 6 is shifted to GND = 0V, and the detected reference signal that is not cut by the bypass voltage is superimposed on GND = 0. Therefore, the first detected reference signal that rises from GND = 0 V is not a complete sine wave in one cycle but a signal waveform in which a part of the minus side is missing. For reference, in FIG. 7C, numbers 1 to 9 are assigned in the order in which the detected reference signals appear. At least up to number 6, a part of the signal waveform is missing. After that, it is not clear only in FIG. 7C whether the signal waveform is missing.

  When a discharge occurs in the coupling capacitor 16, the detected reference signal falling from GND = 0V has a signal waveform in which one cycle is not a complete sine wave and the plus side is partially missing.

  FIG. 8 is a diagram showing the time transition of the voltage value of the signal waveform at P4 in FIG. P4 is a position after the offset voltage is applied to the signal that has passed through the high-pass filter 62 by the offset circuit 64. The offset voltage is + 2.5V.

  The vertical axis and the horizontal axis of FIGS. 8A, 8B, and 8C are the same as the vertical axis and horizontal axis of FIGS. 7A, 7B, and 7C, respectively. Since the voltage value of the signal in each diagram of FIG. 8 is obtained by adding +2.5 V to the signal waveform in each diagram of FIG. 7, further detailed description is omitted.

  FIG. 9 is a diagram showing the time transition of the voltage value of the signal waveform at P5 in FIG. The signal at P5 is a determination signal output from the determination circuit 66 through the determination circuit 66 with respect to the signal waveform that has passed through the noise filter 60. Since no noise signal is added in the simulation, the input signal waveform to the determination circuit 66 is the signal waveform at P2.

FIG. 9A is the same diagram as FIG. 6C, and is a time transition diagram of the signal waveform at P2. As described with reference to FIG. 4, the determination circuit 66 is effective for the portion of V _thL = + 0.2 V or more in the signal waveform of FIG. 9A because V _thL = + 0.2 V is the GND-side determination threshold. It outputs a high level (H) indicating that it is present, and outputs a low level (L) indicating that it is invalid for the other parts.

  FIG. 9B is a diagram illustrating a determination signal. The vertical and horizontal axes in FIG. 9B are the same as the vertical and horizontal axes in FIG. Here, the high level (H) = + 5 V and the low level (L) = 0 V. FIG. 9C is a diagram in which FIG. 9A and FIG. 9B are overlapped. For reference, numbers indicating the order of the detected reference signals in FIGS. 7 and 8 are given. As shown in FIG. 9 (c), the detected reference signals up to number 10 output a low level (L) determination signal indicating invalidity, but the detected reference signals after number 11 , A high-level (H) determination signal indicating that all are continuously valid is output. From this, it can be seen that the detected reference signals of No. 11 and after are signal waveforms that can be signal-processed by the microprocessor 40 and that the signal waveforms are not missing over one period.

When the coupling capacitor 16 is discharged, V _thH = + 11.8V is the Vcc-side determination threshold value, so that the signal waveform obtained by taking the noise filter 60 has a portion below V _thH = + 11.8V. On the other hand, it outputs a high level (H) indicating that it is valid, and outputs a low level (L) indicating that it is invalid for the other parts.

  FIG. 10 is a diagram showing an AND operation process of the signal waveform at P4 and the determination signal waveform at P5 in the microprocessor 40. Here, the time transition of the determination signal is superimposed on the time transition of the signal waveforms of FIGS. 8 (a), (b), and (c). In particular, FIG. 10 (c) is a diagram in which FIG. 9 (b) is superimposed on FIG. 8 (c). For reference, numbers indicating the order of the detected reference signals given in FIGS. 7, 8, and 9 are given.

  In FIG. 9C, since the voltage range of the signal that has passed through the noise filter 60 and the determination signal are not common, even if the AND operation processing is performed on these two signals, the detected reference signals of number 11 and later are displayed. I can't choose. On the other hand, in FIG. 10C, the voltage range of the determination signal and the voltage range of the signal obtained by removing the charging DC component from the signal that has passed through the noise filter 60 and performing the offset process are common. By performing an AND operation process on these two signals, it is possible to select the detected reference signals after the number 11.

In this manner, the microprocessor 40 can acquire a detected reference signal having no signal waveform missing over one period in the operation range. The voltage amplitude of the detected reference signal is compared with the voltage amplitude of the reference signal, and the leakage resistance value R _L1 of the first leakage resistance 8 is detected.

  In the example of FIG. 10C, assuming that the voltage amplitude of the detected reference signal of 10 cycles after number 11 is used for detecting the leakage resistance value, the time for acquiring the detected reference signal of 10 cycles is (0. 4s × 10) = 4s. In the prior art, the leakage resistance value can be detected more quickly than when waiting for about 20 s to 60 s from the start of charging.

  The above voltage values, frequencies, etc. are examples for explanation using simulation, and may be other values.

  In the above description, the AND operation is performed inside the microprocessor 40. However, an AND operation circuit may be provided outside the microprocessor 40. FIG. 11 shows an example in which an AND operation circuit 90 that ANDs the output of the offset circuit 64 and the output of the determination circuit 66 is configured by hardware different from the microprocessor 40. The time transition of the signal waveform at P6 in FIG. 11 corresponds to the content described in FIG. By using the configuration of FIG. 11, the number of input ports of the microprocessor 40 can be reduced as compared with FIG.

6 high-voltage power supply, 8 first leakage resistance, 9 second leakage resistance, 10 leakage detection device, 12 signal generation unit, 14 leakage resistance detection unit, 16 coupling capacitor, 18 first resistance element, 20 second resistance element, 31, 32, 33, 34 Bypass diode, 40 Microprocessor, 42 Oscillator, 44 PWM circuit, 46 Low pass filter (LPF), 48, 62 High pass filter (HPF), 50, 64 Offset circuit, 60 Noise filter, 66 Judgment circuit , 70 Analog-digital converter (ADC), 72 memory, 74 V _PP detection circuit, 76 _RL conversion unit, 80 (charge current) path, 82 (discharge current) path, 84,86 comparator, 88 OR circuit, 90 AND operation circuit.

Claims (7)

A signal generator for generating a periodic reference signal;
A coupling capacitor for connecting the signal generator to a high voltage power supply;
A first resistance element connected between the coupling capacitor and the signal generator;
A leakage resistance detection unit to which a voltage signal generated at a connection point between the coupling capacitor and the first resistance element is supplied;
With
The leakage resistance detection unit is
Determination that outputs a determination signal that determines that a portion within a determination threshold range that is set in advance is valid among voltage signals that are generated at a connection point between the coupling capacitor and the first resistance element, and the others are invalid. Circuit,
A leakage resistance value output circuit for detecting the amplitude of the detected reference signal in a period for determining that the determination signal is valid, and converting the detected reference signal into a leakage resistance value;
Including a leakage detecting device. Provided on the output side of the signal generation unit and the input side of the leakage resistance detection unit, respectively, comprising two sets of bypass diodes comprising a positive protection diode for the circuit power supply voltage and a negative protection diode for the ground voltage;
The leakage detection device according to claim 1, wherein the determination threshold range does not include a conduction voltage of the bypass diode within the range.   3. The leakage detection device according to claim 1, wherein the determination threshold range is a voltage range narrowed by a predetermined margin voltage for each of an upper limit and a lower limit of an operating voltage range of the leakage detection resistance detection unit. A second resistance element connected between a connection point of the coupling capacitor and the first resistance element and the leakage resistance detection unit;
The leakage resistance detection unit is
Depending on the DC voltage component due to the discharge time constant determined by the capacitance value of the coupling capacitor and the resistance value of the first resistance element, or by the charge time constant determined by the capacitance value of the coupling capacitor and the resistance value of the second resistance element Equipped with a high-pass filter that eliminates DC voltage components,
The leakage resistance output circuit is
Using the detected reference signal that acquires the detected reference signal that has passed through the high-pass filter and determines that the determination signal is valid over one cycle of the detected reference signal waveform among the acquired detected reference signals The leakage detection device according to claim 1, wherein the leakage resistance value is output. The leakage resistance detection unit is
Depending on the DC voltage component due to the discharge time constant determined by the capacitance value of the coupling capacitor and the resistance value of the first resistance element, or by the charge time constant determined by the capacitance value of the coupling capacitor and the resistance value of the second resistance element A high-pass filter that excludes DC voltage components;
An AND operation circuit having the detected reference signal passed through the high-pass filter and the determination signal as input signals;
Including
The leakage resistance output circuit uses the detected reference signal that determines that the determination signal is valid over one cycle of the detected reference signal waveform among the detected reference signals output by the AND operation circuit. The leakage detection device according to claim 1, which outputs a leakage resistance value. The leakage resistance output circuit is
The leakage detection device according to claim 2 or 3, wherein the leakage resistance value is output based on a detected reference signal within 10 cycles counted from the beginning of the time series within a range where the determination signal is determined to be valid. The determination circuit includes:
The leakage detecting device according to claim 3, comprising a comparator that compares the detected reference signal with a margin voltage value determined for the upper limit and a margin voltage value determined for the lower limit.

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