JP2016111920A - Permanent magnet type motor, position estimation device, and motor drive controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To expand salient pole properties.SOLUTION: There is provided a permanent magnet type motor which includes: a rotor having a rotor magnet where a plurality of N poles and S poles are alternately magnetized on an outer periphery, and a rotary shaft; and first and second respective ring-shaped pole tooth parts which are arranged oppositely to the rotor magnet and in which a plurality of adjacent claw-poles extending in an axial direction of the rotary shaft at prescribed intervals along a circumferential direction are made close to each other so as to be engaged with each other and base end sides of the respective opposed claw-poles are connected in ring-shapes respectively. On a first joint surface with the second ring-shaped pole tooth part at the first ring-shaped pole tooth part, a first opening is formed, and on a second joint surface with the first ring-shaped pole tooth part at the second ring-shaped pole tooth part, a second opening is formed.SELECTED DRAWING: Figure 5

Description

  The present invention relates to a permanent magnet type motor, a position estimation device, and a motor drive control device.

  Conventionally, permanent magnet motors that do not consume power in the field have been widely used as high-efficiency motors (Patent Document 1). Permanent magnet type motors, in particular, interior permanent magnet (IPM) motors in which permanent magnets are embedded in the rotor have a characteristic called saliency in which the coil inductance varies with the rotor angle. . For this reason, since the IPM motor can use not only the magnet torque due to the magnetic flux of the permanent magnet but also the reluctance torque due to the saliency, and has a high efficiency and a wide use speed range, the use range has been expanded in recent years.

  Furthermore, the IPM motor is also used for sensorless angle estimation that uses the saliency to estimate the rotor angle without using a rotation sensor.

  In sensorless angle estimation using the saliency of a conventional IPM motor, the estimation accuracy may depend on the saliency of the IPM motor.

  The disclosed technology has been made in view of the above circumstances, and aims to increase the saliency.

  In the disclosed technique, a rotor magnet having a plurality of N poles and S poles alternately magnetized on the outer periphery and a rotor having a rotating shaft, and the rotor magnet are disposed opposite to each other, and the rotating shaft is arranged at predetermined intervals along the circumferential direction. A plurality of adjacent pole teeth pieces extending in the axial direction are adjacent to each other so that the base end sides of the opposed pole teeth pieces are connected in a ring shape, respectively. A first opening is formed on a first joint surface of the first ring-shaped pole tooth portion with the second ring-shaped pole tooth portion, and the second ring shape. The permanent magnet motor is characterized in that a second opening is formed on a second joint surface of the pole tooth portion with the first ring-shaped pole tooth portion.

  The saliency can be increased.

It is a figure explaining a motor drive control device. It is a figure which shows an example of the structure of a stepping motor. It is a figure which shows an example of the cross section of a stepping motor. It is a figure which shows an example of the relationship between the coil inductance of a motor, and the phase of a rotor. It is a figure explaining the conductor of the stepping motor of 1st embodiment. It is a figure which shows the self inductance and mutual inductance in 1st embodiment. It is a figure which shows the relative magnetic permeability distribution in the stepping motor of 1st embodiment. It is a figure explaining a position feedback control part. It is a figure explaining a d-axis current control part. It is a figure explaining a q-axis current control part. It is a 1st figure which shows the operation | movement concept of a vector rotation part. It is a 2nd figure which shows the operation | movement concept of a vector rotation part. It is a figure which shows the operation | movement concept of another vector rotation part. It is a figure explaining a position estimation part. It is a figure which shows an example of the signal waveform of q-axis current vector and estimated position error th_err. It is a figure explaining the other example of a position feedback control part. It is a figure explaining the conductor of the stepping motor of 2nd embodiment. It is a figure which shows the self-inductance and mutual inductance in 2nd embodiment. It is a figure explaining the conductor of a stepping motor. It is a figure explaining the conductor of the stepping motor of 3rd embodiment. It is a figure which shows the self-inductance and mutual inductance in 3rd embodiment. It is a figure which shows the relative magnetic permeability distribution in the stepping motor of 3rd embodiment. It is a figure which shows the relative magnetic permeability distribution of the inner yoke of 3rd embodiment. It is a figure explaining the conductor of the stepping motor of 4th embodiment. It is a figure which shows the self-inductance and mutual inductance in 4th embodiment. It is a figure which shows the relative magnetic permeability distribution of the inner yoke of 4th embodiment. It is a figure which shows the 1st modification of 4th embodiment. It is a figure which shows the 2nd modification of 4th embodiment. It is a figure which shows the inner yoke of the stepping motor of 5th embodiment, a self-inductance, and a mutual inductance. It is a figure which shows the inner yoke of the stepping motor of 5th embodiment. It is a figure explaining the conductor of the stepping motor of 6th embodiment. It is a figure which shows the self inductance and mutual inductance in 6th embodiment. It is a figure which shows the relative magnetic permeability distribution of the inner yoke of 6th embodiment. It is a figure explaining the conductor of the stepping motor of 7th embodiment. It is a figure which shows the self inductance and mutual inductance in 7th embodiment. It is a figure which shows the mutual inductance and cogging torque of each embodiment.

(First embodiment)
The first embodiment will be described below with reference to the drawings. FIG. 1 is a diagram illustrating a motor drive control device.

  A motor drive control device 100 shown in FIG. 1 controls the drive of the stepping motor 10. In the stepping motor (STM) 10 of the present embodiment, for example, by passing alternating currents having a phase difference of approximately 90 degrees to the two-phase exciting coils A phase and B phase, a rotor made of a permanent magnet rotates. In addition, the stepping motor 10 of the present embodiment (hereinafter simply referred to as the motor 10) has saliency. The saliency is a characteristic that the inductance of the motor coil changes according to the position of the rotor.

  In the present embodiment, the motor 10 utilizes the characteristic that the self-inductance of the A-phase coil and the B-phase coil changes sinusoidally as a function of the rotor angle.

  Furthermore, the inventor of the present invention estimates the position of the rotor when the saliency of mutual inductance between the A-phase coil and the B-phase coil is increased to the same degree as the self-inductance saliency of the A-phase and B-phase coils. We focused on the fact that the estimated position error is generated stably.

  That is, the motor 10 of the present embodiment is configured such that the saliency of mutual inductance between the A-phase coil and the B-phase coil is increased to the same extent as the saliency of the self-inductance of the A-phase and B-phase coils. Details of the motor 10 of this embodiment will be described later.

  The motor drive control device 100 of this embodiment includes a position feedback control unit 101, a d-axis current control unit 102, a q-axis current control unit 103, a position estimation unit 104, an adder 105, vector rotation units 106 and 107, and a high frequency generation unit. 108, amplification units 109 and 110, and current sensors 111 and 112. The current sensors 111 and 112 may be provided outside the motor drive control device 100.

  The motor drive control device 100 according to the present embodiment supplies the motor 10 with a current obtained by superimposing the high frequency component generated by the high frequency generator 108 on the drive current for driving the motor 10. In the motor drive control device 100, the position estimation unit 104 estimates the position of the rotor of the motor 10 according to the response signal of the high frequency component detected by the current sensors 111 and 112.

  Therefore, in the present embodiment, for example, even when the drive current supplied to the motor 10 is very small, it is possible to detect a high-frequency component response signal and estimate the rotor position based on this response signal. it can.

  For this reason, in the motor drive control device 100 of this embodiment, for example, when the rotation of the motor 10 is stopped or when the rotation speed is low, a sensor for detecting the position of the rotor such as an encoder. The position of the rotor can be estimated by closed loop control without using.

  The position feedback control unit 101 of the present embodiment compares the target position command value th_t with position information th_est indicating the current estimated position of the rotor, and outputs drive target amplitude target values idt and iqt according to the comparison result. To do. In this embodiment, by this control, the amplitude of the drive current is controlled so that the target position command value th_t and the position information th_est match, and the position of the rotor is controlled.

  In the present embodiment, when the target position command value th_t increases or decreases by a certain amount per unit time, the position information th_est is also controlled so as to increase or decrease by a certain amount per unit time. Therefore, the rotor of the motor 10 of this embodiment maintains a constant speed of rotation. In the present embodiment, when the target position command value th_t is stationary at a fixed value, the position information th_est is also controlled to be stationary, that is, to maintain the current position.

  The d-axis current control unit 102 according to the present embodiment outputs the d-axis drive voltage Vd so that the d-axis current vector id detected by the vector rotation unit 107 matches the amplitude target value idt of the d-axis drive current. To do. The q-axis current control unit 103 according to the present embodiment outputs the q-axis drive voltage Vq so that the q-axis current vector iq detected by the vector rotation unit 107 matches the amplitude target value iqt of the q-axis drive current. To do. The d-axis current control unit 102 and the q-axis current control unit 103 of the present embodiment are preferably proportional-integral controllers that perform proportional-integral control, for example.

  The position estimation unit 104 of this embodiment estimates the position (electrical angle) and speed of the rotor of the motor 10 from the high-frequency component superimposed on the q-axis current, and position information (angle) th_est indicating the estimated position of the rotor. And speed information w_est indicating the estimated rotational speed of the rotor is output. Details of the position estimation unit 104 will be described later.

  The adder 105 of this embodiment adds the d-axis drive voltage Vd and the high-frequency signal Vh. The frequency of the high-frequency signal Vh in this embodiment is sufficiently higher than the product of the number of rotations of the rotor and the number of magnetic pole pairs (motor coil drive frequency). Details of the frequency of the high-frequency signal Vh will be described later.

  The vector rotation unit 106 of the present embodiment rotates the d-axis drive voltage Vd and the q-axis drive voltage Vq by position information (angle) th_est, and drives the A-phase drive voltage vector Va and the B-phase drive voltage vector Vb. Is output. The following Expression 1 is an arithmetic expression by the vector rotation unit 106.

The drive voltages Vd and Vq are output drive voltages of the d-axis current control unit 102 and the q-axis current control unit 103 and are signals close to direct current. In this embodiment, since this is rotated by an angle th_est corresponding to the angle of the rotor, the drive voltage vectors Va and Vb become AC signals.

  The vector rotation unit 107 of the present embodiment rotates the detected current vector ia detected in the A phase and the detected current vector ib detected in the B phase by an angle th_est, and d-axis current vector id and q-axis current vector iq is output. The following Expression 2 is an arithmetic expression by the vector rotation unit 107.

In the vector rotation units 106 and 107, the direction of vector rotation is opposite. The detected current vectors ia and ib are AC signals corresponding to the coil current and having a frequency of the number of rotations of the rotor × the number of magnetic pole pairs. In the present embodiment, since the detected current vectors ia and ib are rotated by an angle th_est corresponding to the angle of the rotor, the d-axis current vector id and the q-axis current vector iq are signals close to direct current.

  The high frequency generator 108 of this embodiment generates and outputs a high frequency signal Vh that is superimposed on the drive voltage. The high frequency signal Vh has a fixed frequency and is sufficiently higher than the product of the number of rotations of the rotor and the number of magnetic pole pairs (motor coil drive frequency).

  In this embodiment, by generating the high-frequency signal Vh as described above, it becomes easy to separate the drive signal (drive current) and the response signal of the high-frequency component in the position estimation unit 104, and the estimation accuracy of the rotor position is increased. Can be improved. In the present embodiment, since the mechanical response of the motor 10 can be suppressed by generating the high-frequency signal Vh as described above, the influence on the control of the position and speed of the rotor can be suppressed.

  Furthermore, in this embodiment, if the frequency of the high-frequency signal Vh is higher than the human audible range, unpleasant audible noise can be suppressed. The waveform of the high-frequency signal Vh in the present embodiment may be a sine wave or a rectangular wave, or any other periodic signal.

  The amplifying unit 109 according to the present embodiment converts the A-phase driving voltage vector Va into voltages (voltage A + and voltage A− in FIG. 1) that are actually applied to the coil of the motor 10. The amplifying unit 110 converts the B-phase driving voltage vector Vb into voltages (voltage B + and voltage B− in FIG. 1) that are actually applied to the coil of the motor 10. The voltage A +, the voltage A−, the voltage B +, and the voltage B− are signals having opposite phases. Specifically, the amplification units 109 and 110 of the present embodiment may be realized by, for example, a linear power amplifier, a known PWM (Pulse Width Modulation) inverter circuit, or the like.

  The current sensor 111 of the present embodiment detects the A-phase coil current and outputs a detected current vector ia. Further, the current sensor 112 of the present embodiment detects the B-phase coil current and outputs a detected current vector ib. The current sensors 111 and 112 of the present embodiment include, for example, a configuration in which a low resistance is inserted in series with each of the bus lines of the coil drive lines and the amplification units 109 and 110 to differentially amplify both ends, and magnetic sensors such as Hall elements Etc. may be realized.

  Next, the motor 10 of this embodiment will be described. FIG. 2 is a diagram illustrating an example of the structure of the stepping motor.

  The motor 10 of the present embodiment includes an A-phase coil (armature coil) 11, a B-phase coil (armature coil) 12, and a rotor 20. In the motor 10 of the present embodiment, the A phase coil 11 has an A + terminal and an A− terminal as the A phase coil terminal 13. B phase coil 12 has B + terminal and B− terminal as B phase coil terminal 14. In the present embodiment, the A-phase coil 11 and the B-phase coil 12 are not connected to each other and are provided independently. In the rotor 20 of the present embodiment, a permanent magnet is arranged or magnetized on the circumference.

  In the motor 10, the A-phase coil 11 and the B-phase coil 12 are arranged so as to have a relationship of 90 degrees with respect to the direction of the magnetic flux generated by the permanent magnet of the rotor 20. In the motor 10, the rotor 20 rotates by supplying an alternating current whose phase is shifted by 90 degrees to the A-phase coil 11 and the B-phase coil 12. Moreover, in the motor 10, when the phase of the alternating current supplied to the A-phase coil 11 and the B-phase coil 12 is fixed to a predetermined phase, the rotor 20 is maintained in a magnetically balanced position.

  FIG. 3 is a diagram illustrating an example of a cross section of the stepping motor of the present embodiment. In the example shown in FIG. 3, the rotor 20 is multipolarly magnetized.

  The rotor 20 of the present embodiment has a cylindrical shape, and includes a rotor magnet 20A in which a permanent magnet is periodically magnetized on a cylindrical surface, and a rotating shaft 20B. The A-phase coil 11 is wound in an annular shape outside the circumference of the rotor 20. A phase coil terminal 13 is drawn out.

  The A-phase coil 11 is surrounded by a conductor (ring-shaped pole tooth portion) 21. The conductor 21 is disposed so as to surround the A-phase coil 11, and a conductor member extends in a claw-like shape from one direction (upward in the drawing) to the inner diameter (rotor surface side). ; Pole tooth piece) 23. The pitch of the claw poles 23 is equal to that of the pair of magnetic poles of the rotor, and all the claw poles 23 form an N pole or S pole core depending on the coil current direction.

  A similar claw pole 24 extends from the opposite direction of the A-phase coil 11 (downward in the figure) to form a core having a polarity opposite to that of the upper claw pole 23. In FIG. 3, the claw poles 23 and 24 are illustrated as being formed on the B-phase coil 12 side, but are similarly formed on the A-phase coil 11 side. In the following description, claw poles on the A phase coil 11 side are 23A and 24A, and claw poles on the B phase coil 12 side are 23B and 24B.

  The B phase coil 12 and the B phase coil terminal 14 are the same as the A phase. B-phase coil 12 is surrounded by conductor 22. As with the conductor 21, the conductor 22 is formed with an upper claw pole 23B and a lower claw pole 24B.

  In this embodiment, claw poles 23A and 24A in the A phase and claw poles 23B and 24B in the B phase are shifted by 90 degrees when one period of the magnetic pole pair of the rotor 20 is 360 degrees (so-called electrical angle). Be placed. With this arrangement, the motor 10 shown in FIG. 3 has a structure including a two-phase coil equivalent to the A-phase coil 11 and the B-phase coil 12 shown in FIG.

  FIG. 4 is a diagram illustrating an example of the relationship between the coil inductance of the motor and the phase of the rotor. In FIG. 4, the horizontal axis represents the phase of the rotor 20 and is represented by an electrical angle. Here, the unit is [degree]. The relationship between the electrical angle and the mechanical angle of the rotor 20 is expressed by Equation 3 below.

Electrical angle = rotor mechanical angle x number of poled pairs Equation 3
In FIG. 4, the vertical axis represents the coil inductance, and the unit is [mH (Milli Henry)]. The wavy line La indicates the coil inductance of the A-phase coil 11, and the alternate long and short dash line Lb indicates the coil inductance of the B-phase coil 12.

  The claw pole type PM (Permanent Magnet) stepping motor shown in FIG. 3 has a magnetic characteristic that changes depending on the relationship between the magnetization phase of the rotor and the claw pole phase. It is known that the coil inductance changes periodically. Such a characteristic of the stepping motor is called saliency.

  Here, it is assumed that the coil inductance changes in a sine wave shape of two cycles per electrical angle of 360 degrees (one pitch of the magnetic pole pair of the rotor). It should be noted that the period, amount and shape of change of the coil inductance are not limited to this.

  Further, the motor structure that causes the inductance change shown in FIG. 4 is not limited to the claw pole type. For example, it has been found that an inductance change corresponding to the phase of the rotor is also generated by a structure in which the magnet of the rotor is embedded in the cylindrical conductor instead of the cylindrical surface. The claw pole type PM stepping motor can simplify the coil winding method and other members, and can be produced industrially at low cost.

  Next, the shape of the claw pole of the motor 10 of this embodiment will be described with reference to FIG. FIG. 5 is a diagram illustrating conductors of the stepping motor according to the first embodiment. FIG. 5A shows a cross section of the motor 10, and FIG. 5B shows an enlarged view of a portion surrounded by a dotted line in FIG. 5A. FIG. 5C shows a conductor of a general stepping motor as a comparative example of FIG.

  As shown in FIG. 5B, the motor 10 of this embodiment includes an opening formed on a bonding surface 26A (first opening) bonded to the conductor 22 in the conductor 21 surrounding the A-phase coil 11. 25A. In addition, the motor 10 according to the present embodiment has an opening 25 </ b> B formed in a joint surface 26 </ b> B joined to the conductor 21 in the conductor 22 surrounding the B-phase coil 12.

  In the present embodiment, the opening 25A and the opening 25B are formed in the same shape at the same position on the joint surfaces 26A and 26B of the respective conductors so as to overlap when the conductor 21 and the conductor 22 are joined. . Therefore, in the motor 10 of the present embodiment, when the conductor 21 and the conductor 22 are joined, a through-hole penetrating the conductor 21 and the conductor 22 through the opening 25A and the opening 25B is formed.

  In the present embodiment, openings 25A and 25B having the same number of pitches as the number of magnetic pole pairs of the rotor may be formed in each of the conductor 21 and the conductor 22.

  On the other hand, in the general stepping motor shown in FIG. 5C, each of the conductor 21 'surrounding the A-phase coil and the conductor 22' surrounding the B-phase coil is provided with an opening at the joint surface between them. It is not done. Therefore, when the conductor 21 'and the conductor 22' are joined, a through hole penetrating both is not formed.

  The positions and shapes where the openings 25A and 25B are formed need not be the same positions and shapes as long as the changes in self-inductance and mutual inductance approach a sine wave.

  In the motor 10 of the present embodiment, two (two-phase) stators composed of coils and conductors of each phase are stacked, but the number of stators is not limited to this. In the present embodiment, for example, three (three phases) or more may be stacked and an opening may be provided between the stators.

  Below, with reference to FIG. 6, the self-inductance and mutual inductance of the motor 10 of this embodiment, and the self-inductance and mutual inductance in the motor of a comparative example are demonstrated.

  FIG. 6 is a diagram showing the self-inductance and the mutual inductance in the first embodiment. FIG. 6A is a diagram showing the self-inductance and the mutual inductance in the motor 10 of the present embodiment, and FIG. 6B is a diagram showing the self-inductance and the mutual inductance in the motor of the comparative example.

  It can be seen that the mutual inductance of the motor of the comparative example has a very small change in amplitude with respect to the change in the rotor angle compared to the self-inductance (see FIG. 6B). On the other hand, it can be seen that the mutual inductance of the motor 10 of the present embodiment has a change in amplitude comparable to the self-inductance. Therefore, in the motor 10 of this embodiment, the saliency of the mutual inductance can be increased.

  Hereinafter, the reason why the amplitude of the mutual inductance of the motor 10 of the present embodiment is increased will be described with reference to FIG.

  In this embodiment, the mutual inductance is considered from the relative permeability distribution at the rotor angle T1 where the self inductance and the mutual inductance of the motor 10 are small and the relative permeability distribution at the rotor angle T2 where the self inductance and the mutual inductance are large. .

  FIG. 7 is a diagram illustrating a relative permeability distribution in the stepping motor according to the first embodiment. FIG. 7A is a diagram for explaining the viewing direction of FIGS. 7B and 7C, and FIG. 7B shows the relative permeability distribution at the rotor angle T1, and FIG. C) shows the relative permeability distribution at the rotor angle T2.

  The relative magnetic permeability distribution shown in FIGS. 7B and 7C is obtained when the A phase coil 11 and the conductor 21 and the B phase coil 12 and the conductor 22 are viewed from the direction of the arrow Y shown in FIG. The relative permeability distribution.

  It can be seen that, at the rotor angle T1 shown in FIG. 7B, the relative permeability of the portion K1 surrounded by the left dotted line in the figure is remarkably low. Thereby, the magnetic flux around the A phase (arrow Y1) becomes smaller than the current flowing through the A phase coil 11, and the magnetic flux around the B phase (arrow Y2) becomes larger. From these facts, at the rotor angle T1, the self-inductance of the motor 10 decreases and the mutual inductance increases. The portion K1 is a portion where the conductor 21 and the conductor 22 pass through the openings 25A and 25B.

  It can also be seen that the relative permeability of the portion K1 is high at the rotor angle T2. As a result, the magnetic flux around the A phase (arrow Y1) increases and the magnetic flux around the B phase (arrow Y2) decreases with respect to the current flowing through the A phase coil 11. For these reasons, at the rotor angle T2, the self-inductance increases and the mutual inductance decreases.

  As described above, in this embodiment, by providing the openings 25A and 25B on the joint surfaces 26A and 26B of the A-phase side conductor 21 and the B-phase side conductor 22, the relative magnetic permeability can be increased for each rotor angle. Change. Therefore, according to the present embodiment, the change width of the mutual inductance of the motor 10 can be made substantially the same as the change width of the self-inductance, and the saliency can be expressed.

  Next, details of each part of the motor drive control device 100 of the present embodiment will be described. FIG. 8 is a diagram illustrating the position feedback control unit.

  The position feedback control unit 101 according to the present embodiment includes subtractors 501, 503, gain elements 502, 504, 505, an integrator 506, an adder 507, and a fixed value generation unit 508.

  The subtractor 501 of this embodiment subtracts position information (angle) th_est from the target position command value th_t input to the position feedback control unit 101. That is, the subtractor 501 compares the target position of the rotor 20 with the current estimated position, and calculates a position error.

  The gain element 502 amplifies the output (position error) of the subtractor 501 by a predetermined value G7 times and supplies the amplified value to the subsequent subtracter 503. In this embodiment, the output of the gain element 502 is the target speed of the rotor.

  The subtracter 503 subtracts the speed information w_est from the output of the gain element 502. The speed information w_est is speed information on the rotational speed of the rotor 20. That is, the subtracter 503 compares the target speed of rotation of the rotor with the current speed, and calculates a speed error.

  The gain element 504 amplifies the output (speed error) of the subtracter 503 by a predetermined value G8. The speed error amplified by the gain element 504 is supplied to the gain element 505 and the adder 507.

  The gain element 505 amplifies the output of the gain element 504 by a predetermined value G9 and supplies the amplified value to the integrator 506. The output of the integrator 506 (s is a Laplace operator) is supplied to the adder 507.

  The adder 507 adds the output of the gain element 504 and the output of the integrator 506, performs the following calculation (expressing a transfer function) on the speed error, and performs the amplitude target value iqt of the drive current. Is output.

  Hereinafter, the calculation will be described.

Position error between current estimated position of rotor 20 and target position = th_t−th_est
Current rotation speed and target speed of the rotor 20 = position error × G7
Speed error between target speed of rotor 20 and current speed = target speed-speed information w_est
Drive current amplitude target value iqt = speed error × G8 × (1 + G9 × (1 / s))
In the present embodiment, by configuring as described above, the rotational speed of the rotor 20 can be feedback controlled in the inner loop of the position feedback control unit 101. Therefore, in this embodiment, the control of the position of the rotor 20 can be easily stabilized.

  In addition, since the feedback control of the rotational speed in this embodiment is proportional / integral control, a steady speed error does not occur and precise speed control can be performed. Furthermore, in this embodiment, when the position of the rotor 20 reaches the target position and the motor 10 is stationary, the target speed becomes 0 and no steady speed error occurs, so that a deviation from the target position also occurs. Absent.

  The amplitude target value iqt of the drive current may be calculated using only position error amplification. In this case, the calculation using the speed error is not essential. When the amplitude target value iqt is calculated using only the amplification of the position error, for example, the amplitude target value iqt of the drive current obtained by a known PID (proportional / integral / derivative) operation may be used for the position error.

  In the present embodiment, the target amplitude values idt and iqt correspond to a d-axis drive current and a q-axis drive current in vector control. Since the q-axis drive current indicates torque, a simple control method is known that controls only the q-axis drive current and fixes the d-axis drive current to zero. In the present embodiment, the target amplitude value idt of the d-axis drive current is fixed to 0 by the fixed value generation unit 508 using the above method.

  Next, the d-axis current control unit 102 and the q-axis current control unit 103 will be described with reference to FIGS. 9 and 10. FIG. 9 is a diagram illustrating the d-axis current control unit. FIG. 10 is a diagram illustrating the q-axis current control unit.

  The d-axis current control unit 102 illustrated in FIG. 9 includes a subtracter 201, gain elements 202 and 203, an integrator 204, and an adder 205.

  The q-axis current control unit 103 illustrated in FIG. 10 includes a subtracter 301, gain elements 302 and 303, an integrator 304, and an adder 305.

  The operation of each unit shown in FIGS. 9 and 10 is the same as the operation of each corresponding unit in FIG.

  9 and 10 is performed as follows, and proportional / integral control is realized.

d-axis drive voltage Vd = (target amplitude value idt−d-axis current vector id) × G1
× (1 + G2 × (1 / s))
q-axis drive voltage Vq = (target amplitude value iqt−q-axis current vector iq) × G3
× (1 + G4 × (1 / s))
Next, the vector rotation unit 106 of this embodiment will be described with reference to FIGS. 11 to 13.

  FIG. 11 is a first diagram illustrating an operation concept of the vector rotation unit. FIG. 11 shows an operation concept of the vector rotation unit 106. In FIG. 11, the vertical axis indicates the voltage amplitude, and the horizontal axis indicates the phase (electrical angle) th of the rotor 20. In the present embodiment, the phase actually used is not the phase of the rotor 20 itself but the position information th_est estimated by the position estimation unit 104, but the position estimation unit 104 has position information th_est = th. Therefore, it may be considered that they are substantially the same.

  In FIG. 11, the broken line indicates the d-axis drive voltage Vd. The drive voltage Vd shown in FIG. 11 is a signal to which the high frequency signal Vh is not added.

In FIG. 11, when the driving voltage Vd = 0 and the driving voltage Vq = 1, the direct current is as shown in FIG.
A phase drive voltage vector Va = −sin (th)
B phase drive voltage vector Vb = cos (th)
It becomes.

  This is a phase relationship in which the A phase advances by 90 degrees with respect to the B phase, and 0 degrees corresponds to the reference phase (electrical angle) of the rotor of 0 degrees. When the drive voltage Vd = 0, the amplitudes of the drive voltage vectors Va and Vb are determined by the level of the q-axis drive voltage Vq.

  FIG. 12 is a second diagram illustrating the operation concept of the vector rotation unit. In the example of FIG. 12, the drive voltage Vd = 0.342 and the drive voltage Vq = 0.940. The drive voltage Vd shown in FIG. 12 is a signal to which the high frequency signal Vh is not added.

  In the example of FIG. 12, the amplitudes of the drive voltage vectors Va and Vb remain 1, and it can be seen that the phase A is advanced 20 degrees with respect to the reference phase of the rotor.

  In the present embodiment, the relationship between the drive voltage Vd and the drive voltage Vq is controlled based on the relationship between the d-axis current vector id and the q-axis current vector iq in the d-axis current control unit 102 and the q-axis current control unit 103. Therefore, for example, when the rotational speed of the motor 10 is increased and the phase delay of the detection currents ia and ib is increased, the drive voltages Vd and Vq are set so that the phases of the A-phase drive voltage vector Va and the B-phase drive voltage vector Vb advance. Be controlled. For this reason, in this embodiment, the fall of the efficiency by the rotation speed of the motor 10 can be suppressed. The efficiency of the present embodiment indicates the ratio of the machine output to the input power supplied to the motor 10.

  FIG. 13 is a diagram illustrating an operation concept of another vector rotation unit. FIG. 13 shows an operation concept of the vector rotation unit 107. The conditions are the drive voltage Vd = 0.342 and the drive voltage Vq = 0.940, as in FIG.

  FIG. 13 shows a case where the phases of the A-phase detection current ia and the B-phase detection current ib are delayed by 30 degrees (electrical angle) with respect to the reference phase of the rotor. At this time, the d-axis current vector id and the q-axis current vector iq are direct currents with id = 0.5 and iq = 0.866.

  If the phase detection current ia and the phase B detection current ib are 0 degrees behind the reference phase of the rotor 20, id = 0 and iq = 1.

  That is, in this embodiment, if the current is controlled so that id = 0 (target amplitude value idt = 0 of the drive current), the A-phase detection current ia and the B-phase detection current with respect to the reference phase of the rotor 20 The phase delay of ib can be controlled to 0 degree.

  Furthermore, in the present embodiment, the value of the d-axis current vector id (the value of the target amplitude value idt of the drive current) is set to a value other than 0, so that the phases of the detected currents ia and ib become the reference phase of the rotor 20. Can be shifted. Therefore, in this embodiment, by shifting the phases of the detection currents ia and ib with respect to the reference phase of the rotor 20, the reluctance torque can be used and the power efficiency can be improved. The reluctance torque is the torque when the coil electromagnet attracts the rotor conductor.

  As described above, in the present embodiment, the phase of the detected currents ia and ib is set to the reference phase of the rotor 20 by the d-axis current control unit 102, the q-axis current control unit 103, the vector rotation unit 106, and the vector rotation unit 107. On the other hand, it can be controlled to have a predetermined relationship.

  In the present embodiment, the detection currents ia and ib that are alternating currents are converted into direct current (low frequency) dq-axis currents and controlled, so that the current control band can be kept low. For example, when controlling the detection currents ia and ib, which are alternating currents, to follow the target signal, the currents must be controlled in a band sufficiently higher than the frequency of the alternating currents ia and ib. In this case, the cost becomes high. On the other hand, in this embodiment, as described above, the band for controlling the current can be lowered, and the cost can be reduced.

  Next, the position estimation unit 104 of this embodiment will be described with reference to FIGS. 14 and 15. FIG. 14 is a diagram illustrating the position estimation unit.

  The position estimation unit 104 of this embodiment includes a high-pass filter 400, a multiplier 401, gain elements 402 and 403, integrators 404 and 406, and an adder 405.

  In the present embodiment, the high-frequency signal Vh superimposed on the d-axis drive voltage Vd is expressed by the following Expression 4 when the q-axis current vector iq is observed.

iq = K × Vh × sin (2 × (th−th_est)) + drive signal component Equation 4
Here, K is a constant determined by motor characteristics, circuit constants, etc., Vh is a high-frequency signal superimposed on the drive voltage Vd, th is an electrical angle indicating the current position of the rotor 20, and th_est is the rotor. This is position information (electrical angle) indicating 20 estimated positions.

  In Equation 4, the first term is a component obtained by AM (Amplitude Modulation) modulation of the high frequency component with an estimation error. The estimation error is obtained by subtracting the estimated position of the rotor 20 from the current position of the rotor 20, and is represented by sin (2 × (th−th_est)).

  The second term is a motor drive signal component that controls the drive of the motor 10. Therefore, if the estimation error is extracted (demodulated) from the first term, position information indicating the estimated position of the rotor 20 can be acquired.

  The position estimation unit 104 of the present embodiment allows only the high-frequency component of the q-axis current vector iq supplied from the vector rotation unit 107 to pass through the high-pass filter 400. In the present embodiment, this removes the motor drive signal component in the second term of Equation 4, leaving only the first term.

  The position estimation unit 104 multiplies the high-frequency component of the q-axis drive current vector iq by the high-frequency signal Vh supplied from the high-frequency generation unit 108 by the multiplier 401, and outputs an estimated position error th_err.

  The estimated position error th_err includes a high frequency component, but its low frequency component includes an estimated error sin (2 × (th−th_est)). Therefore, in this embodiment, the position estimation unit 104 may extract a low frequency component of the estimated position error th_err.

  The position estimation unit 104 performs proportional integration control using the gain elements 402 and 403, the integrator 404, and the adder 405. Then, the position estimation unit 104 outputs the output of the adder 405 as the estimated speed w_est.

  The output of the adder 405 is supplied to the integrator 406. The integrator 406 integrates the estimated speed w_est and outputs it as position information th_est. The position information th_est is an electrical angle indicating the estimated position of the current rotor 20.

  The angle calculation described above is expressed as a transfer function as follows.

w_est = th_err × G5 × (1 + G6 × (1 / s)) Equation 4
th_est = w_est × (1 / s) Equation 5
In the position estimation unit 104, th_est is supplied to the vector rotation unit 106, and is fed back to the first term of Expression 4. Therefore, in the position estimation unit 104, the gain element 402 to the integrator 406 serve as a control unit that performs feedback control related to the calculation of position estimation. In the present embodiment, since the control unit itself functions as a low-pass filter, the high frequency component included in the estimated position error th_err is removed.

  FIG. 15 is a diagram illustrating an example of signal waveforms of the q-axis current vector and the estimated position error th_err.

  In FIG. 15, the horizontal axis represents error (th−th_est) (electrical angle). In FIG. 15, it is assumed that the drive signal component of the q-axis current vector iq (the second term in Equation 4) has already been removed.

  In FIG. 15, a broken line indicates a high frequency component (response signal) of the q-axis current vector iq, and the high frequency signal Vh is AM-modulated with an estimation error sin (2 × (th−th_est)).

  In FIG. 15, the dotted line indicates the estimated position error th_err, and is the result of multiplying the high frequency component of the q-axis current vector iq by the high frequency signal Vh.

  From FIG. 15, although the high frequency component remains in the estimated position error th_err, if the error (th−th_est) is a positive value, the estimated position error th_err is also a positive value, and the error (th−th_est) is If the value is negative, the estimated position error th_err is also negative. still. The error (th-th_est) is obtained by subtracting the rotor position estimated by the position estimation unit 104 from the current rotor position, and the estimated position estimated by the position estimation unit 104 and the actual rotor. The error from the position of 20 is shown.

  Accordingly, FIG. 15 shows that the estimated position of the rotor 20 matches the current position of the rotor 20 when the error (th−th_est) matches the estimated position error th_err.

  As described above, in the present embodiment, the estimated position error th_err is used for the subsequent feedback control in the position estimating unit 104, so that the error between the estimated position estimated by the position estimating unit 104 and the actual position of the rotor 20, that is, The estimated position error becomes zero. Therefore, the position estimation unit 104 of the present embodiment can converge the position information th_est of the rotor 20 to a position where both (the estimated position estimated by the position estimation unit 104 and the actual position of the rotor 20) match. .

  The solid line in FIG. 15 shows a case where the estimated position error th_err is applied to the low-pass filter. In the solid line, it can be seen that the high-frequency component remains, but is almost similar to the shape of the estimation error sin (2 × (th−th_est)). In the present embodiment, the feedback control configured by the integrator 406 from the gain element 402 of the position estimation unit 104 performs the function of a low-pass filter, so that a high frequency component is removed as shown by a solid line.

  In addition, although the position estimation part 104 of this embodiment was set as the structure which has the high-pass filter 400, it is not limited to this. The position estimation unit 104 may not have the high-pass filter 400.

  When the position estimation unit 104 does not have the high-pass filter 400, the estimated position error th_err is expressed by the following Expression 5.

Estimated position error th_err = K × Vh ² × sin (2 × (th−th_est))
+ Vh × drive signal component.

  In Equation 5, the first term is the same as when the high-pass filter 400 is provided, and the low-frequency component in the first term includes an estimation error sin (2 × (th−th_est)). Since the second term in Equation 5 is multiplied by the high-frequency signal Vh to become a high-frequency component, it is removed by the function of the low-pass filter of the position estimation unit 104 described above.

  As described above, in the present embodiment, the high-pass filter 400 is not essential, but it is preferable to have the high-pass filter 400 from the following viewpoints.

  In the present embodiment, when the high-pass filter 400 is not provided, it is necessary to control a broadband signal from a high frequency to a low frequency with a component other than the high-pass filter 400 included in the position estimation unit 104. . In this case, for example, in the feedback control in the position estimation unit 104, various restrictions may occur in the design of gain elements and the like.

  On the other hand, if the high-pass filter 400 is provided, the drive signal component can be reduced in advance, so that the degree of freedom in design in feedback control is widened, and the accuracy of estimation of the position of the rotor 20 can be improved as a whole.

  Further, when the high-frequency signal Vh of the present embodiment is a rectangular wave, the q-axis current vector iq can be formed into a rectangular wave shape by sampling the q-axis current vector iq at both edges of the high-frequency signal Vh. Therefore, in this case, an estimation error can be extracted without using a filter. Here, a rectangular wave high-frequency signal Vh having an amplitude of 1 is expressed as follows.

Vh = (− 1) ⁿ
Here, n can be a sample number (0, 1, 2, 3,...). Therefore, the first term of the sampled q-axis current vector iq is expressed as follows.

iq = K × (−1) ⁿ × sin (2 × (th−th_est))
When this is multiplied by a high frequency signal Vh (rectangular wave),
Estimated position error th_err = K × sin (2 × (th−th_est))
Thus, the estimated position error can be extracted without passing through a low-pass filter or the like. In addition, such a rectangular wave can be easily generated, and multiplication can be easily performed only by the code logic, and high-speed processing can be performed at low cost.

  Next, a modified example of the position feedback control unit 101 of the present embodiment will be described with reference to FIG. FIG. 16 is a diagram illustrating another example of the position feedback control unit.

  The position feedback control unit 101 </ b> A includes a q-axis target current calculation unit 509 and a d-axis target current calculation unit 510 in addition to the units other than the fixed value generation unit 508 included in the position feedback control unit 101.

  The q-axis target current calculation unit 509 calculates an amplitude target value iqt of the q-axis drive current from the target current amplitude it that is an output of the adder 507 and the target phase ph. The calculation in the q-axis target current calculation unit 509 is shown as follows.

Amplitude target value iqt = it × tan (ph) / sqrt (1 + tan (ph) ² )
The d-axis target current calculation unit 510 calculates an amplitude target value idt of the d-axis drive current from the target current amplitude it that is the output of the adder 507 and the target phase ph. The calculation in the d-axis target current calculation unit 510 is shown as follows.

Amplitude target value idt = it / sqrt (1 + tan (ph) ² )
Note that the target phase ph in FIG. 16 is a phase difference (advance angle) between the reference phase of the A-phase detection current vector ia and the B-phase detection current vector ib and the reference phase of the rotor 20.

  In the above calculation, the right side from it on the right side can be calculated in advance if the target phase ph is determined. In the present embodiment, as described above, by shifting the target phase ph, a motor or the like that can use reluctance torque can be operated with higher efficiency.

  As described above, in the motor drive control device 100 of the present embodiment, the position estimation unit 104 estimates the position of the rotor of the motor 10 according to the response signal of the high frequency component detected by the current sensors 111 and 112. . This response signal is a response signal using the high-frequency signal Vh superimposed on the drive signal supplied to drive the motor 10 as a carrier.

  For this reason, the motor drive control device 100 of the present embodiment detects a response signal that is a high-frequency component even if it is a weak signal that is difficult to detect by the current sensors 111 and 112 supplied to the motor 10, for example. The rotor 20 can be estimated.

  From the above, according to the present embodiment, for example, when the rotation of the motor 10 is stopped or when the rotation speed is low, the closed loop control can be maintained and consumed by performing the open loop control. Electric power can be reduced.

  In the present embodiment, even if the motor 10 is a stepping motor, the drive current can be controlled to a current corresponding to the load in the entire speed range, so that the occurrence of step-out can be suppressed and the motor can be driven efficiently. Can do.

(Second embodiment)
The second embodiment will be described below with reference to the drawings. In the second embodiment, the position of the opening formed in the conductor surrounding the coil of each phase is made more specific than in the first embodiment. In the following description of the second embodiment, those having the same configuration as in the first embodiment are given the reference numerals used in the description of the first embodiment, and the description thereof is omitted.

  FIG. 17 is a diagram illustrating conductors of the stepping motor according to the second embodiment.

  In the present embodiment, the opening 25A is formed so that the center O of the opening 25A formed in the conductor 21 and the center O of the claw pole 23A are on the same straight line. The center of the opening 25B formed in the conductor 22 may be formed on the same straight line as the center O of the claw pole 23A when the conductor 21 and the conductor 22 are joined.

  FIG. 18 is a diagram showing the self-inductance and the mutual inductance in the second embodiment. FIG. 18A is a diagram showing the self-inductance and the mutual inductance in the motor 10 of the present embodiment, and FIG. 18B is a diagram showing the self-inductance and the mutual inductance in the motor of the comparative example.

  In the present embodiment, it can be seen that the change width of the mutual inductance is larger than that of the motor of the comparative example. Moreover, in this embodiment, it turns out that the waveform of a mutual inductance is approaching the sine wave compared with 1st embodiment shown in FIG.

  In this embodiment, the relative permeability distribution is symmetric with respect to the center of the claw pole 23A by forming the opening 25A so that the center thereof is collinear with the center of the claw pole 23A. This is probably because of this.

(Third embodiment)
The third embodiment will be described below with reference to the drawings. In the third embodiment, the shape of the opening formed in the conductor surrounding the coil of each phase is different from that of the first and second embodiments. In the following description of the third embodiment, those having the same functions as those of the first embodiment are given the reference numerals used in the description of the first embodiment, and the description thereof is omitted.

  FIG. 19 is a diagram illustrating conductors of a stepping motor. In FIG. 19, the conductor of the general stepping motor used as the comparative example with the conductor of the stepping motor of this embodiment is shown.

  In the stepping motor of FIG. 19, a conductor (ring-shaped pole tooth portion) 21 'surrounding the A-phase coil 11 is formed by an outer yoke 21'-1 and an inner yoke 21'-2. Similarly, the conductor 22 'surrounding the B-phase coil 12 is formed of an outer yoke 22'-1 and an inner yoke 22'-2.

  In the example of FIG. 19, it is assumed that the positioning opening 40 is formed in each of the inner yokes 21'-2 and 22'-2. The positioning opening 40 may or may not be provided in each of the inner yokes 21′-2 and 22′-2. Further, the position where the positioning opening 40 is formed is not limited to the position shown in FIG.

  In the present embodiment, there are a plurality of joint surfaces 26A of the conductor 21 'with the conductor 22 in the inner yoke 21'-2 and a joint surface 26B of the conductor 22' with the conductor 21 'in the inner yoke 22'-2. An opening is provided, and a magnetic path constriction is provided between the openings. Below, the conductors 21 and 22 of this embodiment are demonstrated with reference to FIG.

  FIG. 20 is a diagram illustrating conductors of the stepping motor according to the third embodiment. FIG. 20A is a perspective view of the conductor of the present embodiment, and FIG. 20B is a plan view of the inner yoke 21-2.

  In FIG. 20A, the outer yoke 21-1 of the A-phase side conductor 21 is omitted for the description of the opening. FIG. 20B is a plan view of the inner yoke 21-2 of the conductor 21.

  In the present embodiment, a plurality of openings 31 are provided at the outer edge portions of the inner yokes 21-2 and 22-2, and the magnetic path constriction portion 32 is formed between the openings 31. In the following description, the inner yoke 21-2 of the conductor 21 will be described, but a similar opening 31 and magnetic path constriction 32 are also formed in the inner yoke 22-2 of the conductor 22.

  In the present embodiment, when the circumferential width of the inner yoke 21-2 of the opening 31 is W1, and the circumferential width of the inner yoke 21-2 of the magnetic path narrowing portion 32 is W2, W1> W2. Thus, the opening part 31 was formed. The width W1 and the width W2 are the widths in the same circumference in the inner yoke 21-2.

  In the opening 31 formed in the inner yoke 21-1, and the opening 31 formed in the inner yoke 22-2, the conductor 21 and the conductor 22 are overlapped, and the joining surface 26A and the joining surface 26B are joined. Sometimes each is formed to penetrate. Therefore, each of the magnetic path constricted portion 32 formed in the inner yoke 21-1 and the magnetic path constricted portion 32 formed in the inner yoke 22-2 is also when the joining surface 26A and the joining surface 26B are joined. It is formed to overlap.

  In the present embodiment, by making the conductors 21 and 22 into the shape shown in FIG. 20, the width of the change in mutual inductance can be increased as compared with the conventional example.

  In this embodiment, the material of the rotor magnet 20A may be ferrite, and the materials of the conductors 21 and 22 may be SECC steel plates.

  In addition, in FIG. 20, although the opening part 31 was made into the shape which has a corner | angular shape, it is not limited to this. The opening 31 of the present embodiment may be, for example, a semicircular shape formed at the outer edge of the inner yoke 21-2. The opening 31 of the present embodiment is formed at the outer edge, and the relationship between the width W1 and the width W2 of the magnetic path narrowing portion 32 may be W1> W2.

  FIG. 21 is a diagram showing self-inductance and mutual inductance in the third embodiment. FIG. 21A is a diagram showing the self-inductance and the mutual inductance in the motor of the present embodiment, and FIG. 21B is a diagram showing the self-inductance and the mutual inductance in the motor of the comparative example.

  It can be seen that the mutual inductance of the motor of the comparative example has a very small change in amplitude with respect to the change in the rotor angle as compared with the self-inductance (see FIG. 21B).

  On the other hand, it can be seen that the mutual inductance of the motor of the present embodiment has a larger amplitude change than the mutual inductance of the motor of the comparative example. Therefore, in the motor of this embodiment, the saliency of mutual inductance can be increased.

  The reason why the amplitude of the mutual inductance of the motor of the present embodiment is increased will be described below with reference to FIG.

  FIG. 22 is a diagram illustrating a relative permeability distribution in the stepping motor according to the third embodiment. FIG. 22A shows the relative permeability distribution at the rotor angle T1, and FIG. 22B shows the relative permeability distribution at the rotor angle T2. The relative permeability distribution shown in FIGS. 22 (A) and 22 (B) is obtained from the direction of arrow Y shown in FIG. 20 (A) when the A phase coil 11 and the conductor 21 and the B phase coil 12 and the conductor 22 are viewed. The relative permeability distribution in the case of

  It can be seen that, at the rotor angle T1 shown in FIG. 22A, the relative permeability of the portion K11 surrounded by the dotted line in the figure is remarkably low. This is because the magnetic flux concentrates on the magnetic path narrowing portion 32 formed adjacent to the opening 31 by providing the opening 31. In this embodiment, this reduces the magnetic flux around the A phase (arrow Y11) and increases the magnetic flux around the B phase (arrow Y12) with respect to the current flowing through the A phase coil 11. From these facts, at the rotor angle T1, the self-inductance of the motor decreases and the mutual inductance increases. The portion K11 is a portion where the conductor 21 and the conductor 22 pass through the opening 31.

  It can also be seen that the relative permeability of the portion K11 is high at the rotor angle T2. Thereby, the magnetic flux around the A phase (arrow Y11) increases with respect to the current flowing through the A phase coil 11, and the magnetic flux around the B phase (arrow Y12) decreases. For these reasons, at the rotor angle T2, the self-inductance increases and the mutual inductance decreases.

  As described above, in the present embodiment, by providing the openings 31 in the joint surfaces 26A and 26B of the A-phase side conductor 21 and the B-phase side conductor 22, the relative permeability is increased for each rotor angle. Change. Therefore, according to the present embodiment, the change width of the mutual inductance of the motor can be made substantially the same as the change width of the self-inductance, and the saliency can be expressed.

  FIG. 23 is a diagram illustrating a relative permeability distribution of the inner yoke according to the third embodiment. 23A shows the relative permeability distribution of the inner yoke 21-2 at the rotor angle T1, and FIG. 23B shows the relative permeability distribution of the inner yoke 21-2 at the rotor angle T2. Show.

  In the present embodiment, at the rotor angle T1, a decrease in the relative magnetic permeability due to the concentration of magnetic flux is observed in the magnetic path constriction portion 32, and at the rotor angle T2, the concentration of magnetic flux is observed in the inner yoke 21-2. No decrease in the relative permeability is observed.

  In the present embodiment, the saliency of the mutual inductance is improved by the reduction of the relative permeability due to the magnetic path constriction portion 32.

  In the present embodiment, the magnetic path constriction portion 32 is made narrower, that is, the magnetic path is narrowed by narrowing the cross-sectional area of the magnetic path in the circumferential direction, thereby concentrating the magnetic flux more and making the relative permeability. And saliency can be improved.

  In the present embodiment, the openings 31 in the A-phase side inner yoke 21-2 and the B-phase side inner yoke 22-2 are formed in the same shape at the same position. However, the present invention is not limited to this. . The openings 31 in the inner yoke 21-2 on the A phase side and the inner yoke 22-2 on the B phase side do not have to be in the same position and shape as long as the change in mutual inductance approaches a sine wave.

(Fourth embodiment)
The fourth embodiment will be described below with reference to the drawings. In the fourth embodiment, the shape of the opening formed in the conductor surrounding the coil of each phase is made different from that of the third embodiment. In the following description of the fourth embodiment, only differences from the third embodiment will be described, and those having functions similar to those of the third embodiment will be denoted by the reference numerals used in the description of the third embodiment. The description is omitted.

  FIG. 24 is a diagram illustrating conductors of the stepping motor according to the fourth embodiment. FIG. 24A is a perspective view of the conductor of this embodiment, and FIG. 24B is a plan view of the inner yoke.

  In FIG. 24A, the outer yoke 21-1A of the A-phase-side conductor 21 is omitted for the description of the opening. FIG. 24B is a plan view of the inner yoke 21-2A of the conductor 21. FIG.

  In the present embodiment, the magnetic path constricted portion 32A is formed by forming the opening 31A in the joint surface 26A of the inner yoke 21-2A.

  In the present embodiment, at least two or more openings 31A and magnetic path constrictions 32A are formed on the joint surface 26A of the inner yoke 21-2A.

  Further, when the circumferential width of the inner yoke 21-2A of the opening 31A of the present embodiment is W11 and the circumferential width of the inner yoke 21-2A of the magnetic path narrowing portion 32A is W21, W11> W21. Thus, the opening 31A was formed. Note that the width W11 and the width W21 are widths on the same circumference of the inner yoke 21-2A.

  Furthermore, in the present embodiment, it is preferable that the width W21 of any of the plurality of magnetic path constrictions 32A is formed to be equal to or less than the width in the circumferential direction of the inner yoke 21-2A of the positioning opening 40. .

  For example, in the example of FIG. 24, the opening 31A is provided on both sides of the positioning opening 40, and the magnetic path narrowing portion 32A is formed between the opening 40 and each opening 31A.

  The width W21 of the magnetic path narrowing portion 32A is preferably equal to or smaller than the circumferential width of the opening 40.

  24, the inner yoke 21-2A of the conductor 21 has been described, but the same opening 31A and magnetic path constricted portion 32A are also formed in the inner yoke 22-2A of the conductor 22. In the present embodiment, the opening 31A and the magnetic path constriction portion 32A in the inner yoke 21-2A on the A phase side and the inner yoke 22-2A on the B phase side are preferably formed in the same shape at the same position.

  In the present embodiment, the width of the change in mutual inductance can be increased by making the conductors 21 and 22 have the shape shown in FIG.

  In the present embodiment, the material of the rotor magnet 20A may be ferrite, and the materials of the conductors 21 and 22 may be SECC (electrogalvanized) steel plates.

  FIG. 25 is a diagram showing the self-inductance and the mutual inductance in the fourth embodiment.

  According to FIG. 25, it can be seen that the mutual inductance of the motor of the present embodiment has a larger change in amplitude than the mutual inductance of the motor of the comparative example (see FIG. 21B). Therefore, in the motor of this embodiment, the saliency of mutual inductance can be increased.

  FIG. 26 is a diagram showing a relative permeability distribution of the inner yoke according to the fourth embodiment. 26A shows the relative permeability distribution of the inner yoke 21-2A at the rotor angle T1, and FIG. 26B shows the relative permeability distribution of the inner yoke 21-2A at the rotor angle T2. Show.

  In the present embodiment, at the rotor angle T1, a decrease in the relative permeability due to the concentration of magnetic flux is observed in the magnetic path constriction portion 32A, and at the rotor angle T2, the concentration of magnetic flux is observed at the inner yoke 21-2A. No decrease in the relative permeability is observed.

  In the present embodiment, the saliency of the mutual inductance is improved by the reduction of the relative permeability by the magnetic path constriction part 32A. Also in this embodiment, similarly to the third embodiment, by reducing the width W21 of the magnetic path constriction portion 32A, that is, by making the magnetic path narrower, the relative permeability is further reduced, and more The polarity can be improved.

<First modification>
Below, the 1st modification of 4th embodiment is demonstrated with reference to FIG. FIG. 27 is a diagram illustrating a first modification of the fourth embodiment. FIG. 27A is a diagram showing a plan view of the inner yoke 21-2B of the first modification, and FIG. 27B is a diagram showing the self-inductance and the mutual inductance of the first modification. .

  In the first modification, a larger number of openings 31B are formed in the inner yoke 21-2B than the number of openings 31A formed in the inner yoke 21-2A in the second embodiment. Further, in the first modification, the inner yoke 21-2B is formed so that the widths of the plurality of openings 31B are different.

  More specifically, in the first modification, the width W12 is wider than the width of the positioning opening 40, and the width W13 is narrower than the width W12 and the width of the positioning opening 40 is larger. An opening 31B was formed.

  In the first modification, the opening 31B is formed so that the width W22 of the magnetic path constriction 32B is equal to or smaller than the width of the positioning opening 40.

  According to the first modification, as shown in FIG. 27B, the width of the mutual inductance amplitude may be larger than the amplitude of the mutual inductance of the motor of the comparative example (see FIG. 21B). Recognize.

<Second modification>
The second modification of the fourth embodiment will be described below with reference to FIG. FIG. 28 is a diagram illustrating a second modification of the fourth embodiment. FIG. 28A is a diagram showing a plan view of the inner yoke 21-2C of the second modified example, and FIG. 28B is a diagram showing the self-inductance and the mutual inductance of the second modified example. .

  In the second modification, a larger number of openings 31C are formed in the inner yoke 21-2C than the number of openings 31A formed in the inner yoke 21-2A in the second embodiment. In the second modification, the width W14 of the opening 31C formed in the inner yoke 21-2C may be equal to the width W13 of the opening 31B in the first modification, for example. That is, in the second modification, a larger number of openings 31C are formed than the number of openings 31B formed in the inner yoke 21-2B in the first modification.

  In the second modification, the opening 31C is formed so that the width W23 of the magnetic path narrowing portion 32C is equal to or smaller than the width of the positioning opening 40.

  The first and second modifications described above increase the number of magnetic path constrictions 32B and 32C compared to the second embodiment, and increase the mechanical strength of the inner yokes 21-2B and 21-2C. Can be bigger. In order to reduce the relative permeability, it is preferable that the number of magnetic path constrictions in the inner yoke is small. Therefore, in the first and second modifications, the mechanical strength and the amplitude of the mutual inductance change are large. Based on the above, the width of the opening and the number of magnetic path constrictions may be determined.

  In the first and second modifications, the material of the rotor magnet 20A may be a rare earth magnet such as neodymium, and the material of the conductors 21 and 22 may be a silicon steel plate or the like. In the first and second modified examples, by using the above-described materials, the amount of magnetic flux from the rotor magnet 20A toward the conductors 21 and 22 can be increased, and the BH (magnetic hysteresis) characteristics of the conductors 21 and 22 can be improved. Specifically, the change in the amount of magnetic flux with respect to the magnetic field can be made steep.

  Therefore, in the first and second modified examples, the magnetic flux can be concentrated on the magnetic path constricted portion, and the relative permeability can be lowered. That is, by using the above-described material, the saliency can be improved while ensuring the mechanical strength.

(Fifth embodiment)
The fifth embodiment will be described below with reference to the drawings. The fifth embodiment is different from the third embodiment in that an opening is formed at the inner edge of the inner yoke. In the following description of the fifth embodiment, only differences from the third embodiment will be described, and those having functions similar to those of the third embodiment will be denoted by the reference numerals used in the description of the third embodiment. The description is omitted.

  FIG. 29 is a diagram illustrating an inner yoke, a self-inductance, and a mutual inductance of the stepping motor according to the fifth embodiment. FIG. 29A is a diagram showing a plan view of the inner yoke 21-2D of the fifth embodiment, and FIG. 29B is a diagram showing the self-inductance and the mutual inductance of the fifth embodiment. .

  In the present embodiment, a plurality of openings 31D are provided at the inner edge of the inner yoke 21-2D, and the magnetic path constriction 32D is formed between the openings 31D.

  The opening 31D of the present embodiment is formed at the inner edge between the claw poles 24A. In the present embodiment, by forming the opening 31D in this way, the magnetic path constriction portion 32D is formed at a position corresponding to the claw pole 24A.

  Further, the width W15 of the opening 31D of the present embodiment is formed such that the width 24 of the magnetic path constriction portion 32D is equal to or less than the widest width Wc of the claw pole 24A. That is, the width W15 of the opening 31D of the present embodiment is formed to be wider than the widest width Wc in the claw pole 24A.

  More specifically, the width W15 of the opening 31D is formed such that the width W24 of the plurality of magnetic path narrowing portions 32D is 1/3 of the widest width Wc in the claw pole 24A. In the present embodiment, it is only necessary to form at least two magnetic path narrowing portions 32D having a width W24 that is 1/3 of the width Wc in the plurality of magnetic path narrowing portions 32D.

  In the present embodiment, it is preferable that the corresponding magnetic path narrowing portion 32D is formed for all the claw poles 24A of the inner yoke 21-2D.

  According to the present embodiment, as shown in FIG. 29B, it can be seen that the change in the amplitude of the mutual inductance is larger than that in the fourth embodiment (see FIG. 25). Moreover, in this embodiment, since the mutual inductance average value has fallen compared with 4th embodiment, it turns out that a cogging torque can be reduced. The cogging torque is a torque that is generated positively or negatively with the rotation of the rotor even in a non-energized state.

  FIG. 30 is a diagram showing an inner yoke of the stepping motor according to the fifth embodiment. 30A is a diagram showing a plan view of the inner yoke 21-2D on the A phase side, and FIG. 30B is a diagram showing a plan view of the inner yoke 22-2D on the B phase side.

  In this embodiment, as shown in FIGS. 30A and 30B, the position of the claw pole 23B formed on the B-phase inner yoke 22-2D is formed on the A-phase inner yoke 21-2D. The position of the claw pole 24A is shifted (see FIG. 19).

  In the inner yoke 22-2D, the opening 31D is formed at the inner edge between the claw poles 23B, and the magnetic path narrowing portion 32D is formed at a position corresponding to the claw pole 23B of the inner yoke 22-2D.

(Sixth embodiment)
The sixth embodiment will be described below with reference to the drawings. The sixth embodiment differs from the fourth embodiment in that a groove is formed in addition to the opening in the conductor surrounding each phase coil. In the following description of the sixth embodiment, only differences from the fourth embodiment will be described, and those having functions similar to those of the fourth embodiment are denoted by the reference numerals used in the description of the fourth embodiment. The description is omitted.

  FIG. 31 is a diagram illustrating conductors of the stepping motor according to the sixth embodiment. FIG. 31A is a perspective view of the conductor of this embodiment, and FIG. 31B is a plan view of the inner yoke 21-2E of the conductor 21. FIG.

  In the present embodiment, the circular groove 33 is formed in the inner yoke 21-2E along the outer circumference circle where the opening 31A is formed. The width W31 of the groove 33 is narrower than the width W32 in the direction (radial direction) orthogonal to the width W11 of the opening 31A, and the width W32> the width W31. The width W31 of the groove 33 is a width in a direction (radial direction) orthogonal to the circumferential direction.

  In the present embodiment, for example, the width W31 may be 1 [mm] and the width W32 may be 1.5 [mm].

  In the present embodiment, by providing the groove 33 in this way, the thickness of the magnetic path constriction portion 32E formed by the opening 31A and the groove 33 is reduced by the depth of the groove 33. That is, the magnetic path narrowing portion 32E is thinner than the magnetic path narrowing portion 32A of the fourth embodiment.

  FIG. 32 is a diagram showing self-inductance and mutual inductance in the sixth embodiment.

  In FIG. 32, it can be seen that the amplitude changes significantly compared to the mutual inductance of the motor of the comparative example (see FIG. 21B). Therefore, in the motor of this embodiment, the saliency of mutual inductance can be increased.

  FIG. 33 is a diagram showing a relative permeability distribution of the inner yoke according to the sixth embodiment. FIG. 33A shows the relative permeability distribution of the inner yoke 21-2E at the rotor angle T1, and FIG. 33B shows the relative permeability distribution of the inner yoke 21-2E at the rotor angle T2. Show.

  In the present embodiment, at the rotor angle T1, a decrease in the relative magnetic permeability due to the concentration of magnetic flux is observed in the magnetic path constriction portion 32E, and at the rotor angle T2, the concentration of magnetic flux is observed at the inner yoke 21-2E. No decrease in the relative permeability is observed.

  In the present embodiment, the saliency of the mutual inductance is improved by the reduction of the relative permeability due to the magnetic path constriction part 32E.

(Seventh embodiment)
The seventh embodiment will be described below with reference to the drawings. The seventh embodiment differs from the sixth embodiment in that the width of the groove formed in addition to the opening is wider than the radial width of the opening. In the following description of the seventh embodiment, only differences from the sixth embodiment will be described, and those having functions similar to those of the sixth embodiment will be denoted by the reference numerals used in the description of the sixth embodiment. The description is omitted.

  FIG. 34 is a diagram illustrating conductors of the stepping motor according to the seventh embodiment. FIG. 34 is a plan view of the inner yoke 21-2F on the A phase side.

  In the present embodiment, the radial width W33 of the groove 33A is formed to be equal to or larger than the radial width W32 of the opening 31A (width W32 ≦ width W33).

  Accordingly, the magnetic path narrowing portion 32F formed by the opening 31A and the groove 33A is thinner by the depth of the groove 33A in a wider range in the radial direction than the magnetic path narrowing portion 32E of the sixth embodiment.

  FIG. 35 is a diagram showing the self-inductance and the mutual inductance in the seventh embodiment.

  The width of the change in the amplitude of the mutual inductance in this embodiment is almost the same as that in the sixth embodiment, but the average value of the mutual inductance is reduced. Therefore, in this embodiment, it can be seen that the cogging torque can be reduced while maintaining the saliency equivalent to that of the sixth embodiment.

  The relationship between the cogging torque and the average value of the mutual inductances of the stepping motor as a comparative example and the stepping motors of the third, fourth, sixth, and seventh embodiments will be described below with reference to FIG. FIG. 36 is a diagram showing the mutual inductance and cogging torque of each embodiment.

  FIG. 36 shows the relationship between the average value of mutual inductance and the value from the positive peak to the negative peak of cogging torque.

  As can be seen from FIG. 36, in the seventh embodiment, the cogging torque can be reduced while maintaining the saliency substantially equivalent to that of the sixth embodiment.

  The stepping motor used for the simulation for measuring the self-inductance and the mutual inductance in the first and second embodiments and the simulation for measuring the self-inductance and the mutual inductance in the third to seventh embodiments. The size is different from the stepping motor used.

  Specifically, the stepping motors of the third to seventh embodiments have a smaller outer diameter than the stepping motors of the first and second embodiments, the magnetic force of the magnet is weak, and the number of turns of the coil is small. did.

  In the third to seventh embodiments, the conductors 21 and 22 are formed of an inner yoke and an outer yoke, but the present invention is not limited to this. The conductors 21 and 22 may be formed integrally.

  As mentioned above, although this invention has been demonstrated based on each embodiment, this invention is not limited to the requirements shown in the said embodiment. With respect to these points, the gist of the present invention can be changed without departing from the scope of the present invention, and can be appropriately determined according to the application form.

DESCRIPTION OF SYMBOLS 10 Stepping motor 20 Rotor 23A, 23B, 24A, 24B Claw pole 25A, 25B, 31, 31A-31F Opening part 32, 32A-32F Magnetic path constriction part 100 Motor drive control apparatus 101 Position feedback control part 102 d-axis current control Unit 103 q-axis current control unit 104, 113 position estimation unit 105 adder 106, 107 vector rotation unit 108 high frequency generation unit 109, 110 amplification unit 111, 112 current sensor 114 selector 115 counter

JP 2014-99996 A

Claims (10)

A rotor magnet having a plurality of alternating N and S poles on the outer periphery and a rotor having a rotation shaft;
A plurality of adjacent pole teeth pieces that are arranged to face the rotor magnet and extend in the axial direction of the rotating shaft at predetermined intervals along the circumferential direction are close to each other so as to mesh with each other, and the bases of the pole teeth pieces that face each other are opposed to each other. The end side includes first and second ring-shaped pole teeth connected to each other in a ring shape,
A first opening is formed on a first joint surface with the second ring-shaped pole tooth portion in the first ring-shaped pole tooth portion,
A permanent magnet type motor, wherein a second opening is formed in a second joint surface of the second ring-shaped pole tooth portion with the first ring-shaped pole tooth portion. The center of the first opening and the second opening is
Of the pole tooth pieces of the first ring-shaped pole tooth portion, it is formed so as to be on the same straight line as the center of the pole tooth piece extending toward the first joint surface. The permanent magnet type motor according to claim 1. Forming a plurality of the first openings and the second openings;
A plurality of magnetic path constrictions are formed between the openings,
The plurality of first openings and the second openings, and the plurality of magnetic path constrictions,
The circumferential width of the magnetic path narrowing portion is formed so as to be equal to or less than the circumferential width of at least one of the first opening and the second opening. The permanent magnet type motor described. The plurality of first and second openings are
The permanent magnet motor according to claim 3, wherein the permanent magnet motor is formed on an outer edge portion of each of the first and second ring-shaped pole teeth. The plurality of first and second openings are
The permanent magnet motor according to claim 3, wherein the permanent magnet motor is formed at an inner edge of each of the first and second ring-shaped pole teeth. The plurality of first and second openings are formed along the circumferential direction of the first and second ring-shaped pole teeth,
In each of the first and second ring-shaped pole teeth, a circular arc is formed along the circumferential direction in which the first and second openings are formed between the first and second openings. The permanent magnet type motor according to claim 1, wherein the permanent magnet type motor has a groove formed.   The permanent magnet motor according to claim 6, wherein a radial width of each of the first and second ring-shaped pole teeth of the groove is equal to or less than a radial width of the first and second openings. .   The permanent magnet type motor according to claim 6, wherein a radial width of each of the first and second ring-shaped pole teeth of the groove is wider than a radial width of the first and second openings. A permanent magnet type motor according to any one of claims 1 to 8,
A harmonic generation unit that generates a harmonic signal supplied to the armature winding of the permanent magnet type motor;
A current detector that detects a harmonic current component that is a response of the harmonic signal;
A position estimation unit that estimates a position of the rotor of the permanent magnet type motor based on the harmonic signal and the harmonic current component;   A motor drive control device comprising the position estimation device according to claim 9.